U.S. patent application number 14/702609 was filed with the patent office on 2015-08-20 for wideband receiver robust to radio frequency harmonics.
The applicant listed for this patent is Broadcom Corporation. Invention is credited to Hooman Darabi, David MURPHY.
Application Number | 20150236736 14/702609 |
Document ID | / |
Family ID | 50931467 |
Filed Date | 2015-08-20 |
United States Patent
Application |
20150236736 |
Kind Code |
A1 |
MURPHY; David ; et
al. |
August 20, 2015 |
WIDEBAND RECEIVER ROBUST TO RADIO FREQUENCY HARMONICS
Abstract
A radio frequency (RF) noise-cancelling receiver includes first
transconductance cells configured to produce respective weighted
current signals proportional to an input voltage signal. The RF
receiver includes frequency conversion cells coupled to the first
transconductance cells and configured to mix the weighted current
signals with a plurality of non-overlapping local oscillator (LO)
signals to produce downconverted current signals. The RF receiver
includes transimpedance amplifiers coupled to the frequency
conversion cells and configured to produce output voltage signals
proportional to the downconverted current signals. The
transimpedance amplifiers include second transconductance cells.
Each of the first and second transconductance cells has an
effective transconductance of a first magnitude for frequency
components of the input voltage signal arising from a first
harmonic and an effective transconductance of a second magnitude
less than the first magnitude for frequency components of the input
voltage signal arising from harmonics at integer multiples of the
first harmonic.
Inventors: |
MURPHY; David; (Costa Mesa,
CA) ; Darabi; Hooman; (Laguna Niguel, CA) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Broadcom Corporation |
Irvine |
CA |
US |
|
|
Family ID: |
50931467 |
Appl. No.: |
14/702609 |
Filed: |
May 1, 2015 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
13736895 |
Jan 8, 2013 |
9059796 |
|
|
14702609 |
|
|
|
|
61737077 |
Dec 13, 2012 |
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Current U.S.
Class: |
455/305 |
Current CPC
Class: |
H04B 15/06 20130101;
H03D 7/18 20130101; H04B 1/10 20130101; H04B 1/1081 20130101 |
International
Class: |
H04B 1/10 20060101
H04B001/10; H04B 15/06 20060101 H04B015/06 |
Claims
1. A radio frequency (RF) receiver comprising: first
transconductance cells configured to produce weighted current
signals proportional to an input voltage signal along a first
downconversion path; a plurality of frequency conversion cells
configured to produce downconverted current signals based on a
plurality of non-overlapping local oscillator (LO) signals, a
current combining component coupled to first frequency conversion
cells of the plurality of frequency conversion cells, the current
combining component configured to combine each output of the first
frequency conversion cells to a current signal bus; and a plurality
of transimpedance amplifiers configured to produce output voltage
signals proportional to the downconverted current signals, the
plurality of transimpedance amplifiers comprising second
transconductance cells, the first transconductance cells and the
second transconductance cells having an effective transconductance
of a first magnitude for frequency components of the input voltage
signal arising from a first harmonic and an effective
transconductance of a second magnitude for frequency components of
the input voltage signal arising from harmonics at integer
multiples of the first harmonic, the first magnitude being greater
than the second magnitude, the plurality of transimpedance
amplifiers having first transimpedance amplifiers coupled to the
current combining component to receive the current signal bus along
the first downconversion path and second transimpedance amplifiers
coupled to at least one second frequency conversion cell of the
plurality of frequency conversion cells along a second
downconversion path, each of the first transconductance cells
coupled to a frequency conversion cell of the first frequency
conversion cells, the at least one second frequency conversion cell
configured to receive the input voltage signal directly.
2. The RF receiver of claim 1, wherein each of the first
transconductance cells and the second transconductance cells is
configured with a transconductance weighting that causes the
effective transconductance of a transconductance cell to be
non-zero when excited by received signals arising from a harmonic
within a frequency band of interest and zero for received signals
arising from harmonics outside the frequency band of interest.
3. The RF receiver of claim 2, wherein the transconductance
weighting corresponds to a current change through the
transconductance cell for a corresponding wanted harmonic.
4. The RF receiver of claim 1, wherein the first transconductance
cells and the second transconductance cells are configured to
amplify the input voltage signal from any harmonic of a fundamental
frequency.
5. The RF receiver of claim 4, wherein the first transconductance
cells and the second transconductance cells are further configured
to amplify the input voltage signal from more than one harmonic
multiple of the fundamental frequency.
6. The RF receiver of claim 1, wherein the plurality of frequency
conversion cells and the first transconductance cells employ
single-ended inputs.
7. The RF receiver of claim 6, wherein each of the first
transconductance cells and the second transconductance cells
comprises a transconductance weighting equal to gmx = k ( 1 + cos (
2 .pi. X M ) ] , ##EQU00011## where X is an integer that represents
each of the transconductance cells, M is a number of LO phases
employed, and k is an arbitrary constant that determines the
effective transconductance of the transconductance cell.
8. The RF receiver of claim 1, wherein the plurality of frequency
conversion cells and the first transconductance cells employ
differential-ended inputs.
9. The RF receiver of claim 8, wherein each of the first
transconductance cells and the second transconductance cells
comprises a transconductance weighting equal to gmx = k ( cos ( 2
.pi. X M ) ] , ##EQU00012## where X is an integer that represents
each of the transconductance cells, M is a number of LO phases
employed, and k is an arbitrary constant that determines the
effective transconductance of the transconductance cell.
10. The RF receiver of claim 1, wherein the plurality of
non-overlapping LO signals have successive phase shifts
substantially equal to 360/M degrees, where M is a number of LO
phases employed.
11. The RF receiver of claim 10, wherein each of the plurality of
non-overlapping LO signals corresponds to a fundamental
frequency.
12. The RF receiver of claim 11, wherein the number of LO phases
corresponds to a number of harmonics relative to the fundamental
frequency.
13. The RF receiver of claim 12, wherein each of the second
transconductance cells receives M downconversion signals including
an input coupled to ground, where M is the number of LO phases
employed based on the number of harmonics corresponding to the
fundamental frequency with a magnitude greater than a threshold
magnitude.
14. The RF receiver of claim 10, wherein each of the second
transconductance cells receives a voltage input equivalent to V IN
.angle. m * ( M - 1 ) M , ##EQU00013## where m is a LO harmonic
from around which the input voltage signal originated, and M
represents the number of LO phases employed.
15. A radio frequency (RF) receiver comprising: a first set of
transconductance cells configured to produce respective weighted
current signals proportional to an input voltage signal; a first
set of frequency conversion cells configured to mix the weighted
current signals with a plurality of non-overlapping local
oscillator (LO) signals to produce first downconverted current
signals, each frequency conversion cell of the first set of
frequency conversion cells being coupled to one transconductance
cell of the first set of transconductance cells along a first
signal path; a current combining component coupled to each output
of the first set of frequency conversion cells, the current
combining component configured to aggregate the first downconverted
current signals into a current signal bus; a first set of
transimpedance amplifiers configured to produce first output
voltage signals proportional to the current signal bus, each input
to the first set of transimpedance amplifiers being coupled to the
current combining component along the first signal path; a second
set of frequency conversion cells configured to receive the input
voltage signal along a second signal path and mix the input voltage
signal with the plurality of non-overlapping LO signals to produce
second downconverted current signals; and a second set of
transimpedance amplifiers configured to produce second output
voltage signals proportional to the second downconverted current
signals, each input to the second set of transimpedance amplifiers
being coupled to the second set of frequency conversion cells along
the second signal path, the first set of transimpedance amplifiers
and the second set of transimpedance amplifiers comprising a second
set of transconductance cells, each of the first set and second set
of transconductance cells having an effective transconductance of a
first magnitude for frequency components of the input voltage
signal arising from a first harmonic and an effective
transconductance of a second magnitude for frequency components of
the input voltage signal arising from harmonics at integer
multiples of the first harmonic, the first magnitude being greater
than the second magnitude.
16. The receiver of claim 15, wherein each of the first set and
second set of transconductance cells is further configured with a
transconductance gain that shifts the weighted current signal with
a defined phase such that a constructive sum of the weighted
current signals provides a resulting current signal of a selected
phase.
17. The receiver of claim 16, wherein the first set of
transconductance cells and the first set and second set of
frequency conversion cells employ single-ended inputs, and wherein
the transconductance gain is shifted to reject any integer multiple
of a wanted signal corresponding to the first harmonic.
18. The receiver of claim 16, wherein the first set of
transconductance cells and the first set and second set of
frequency conversion cells employ differential-ended inputs, and
wherein the transconductance gain is shifted to reject odd integer
multiples of a wanted signal corresponding to the first
harmonic.
19. The receiver of claim 15, wherein each of the second set of
transconductance cells receives a voltage input equivalent to V IN
.angle. m * ( M - 1 ) M , ##EQU00014## where m is a LO harmonic
from around which the input voltage signal originated, and M
represents a number of LO phases employed.
20. A receiver for wideband applications, comprising: an integrated
circuit pin configured to couple an input radio frequency (RF)
signal to a first signal path and a second signal path; a plurality
of transconductance cells configured to produce weighted current
signals proportional to the input RF signal along the first signal
path, each of the plurality of transconductance cells having an
effective transconductance of a first magnitude for frequency
components of the input RF signal arising from a first harmonic and
an effective transconductance of a second magnitude less than the
first magnitude for frequency components of the input RF signal
arising from harmonics at integer multiples of the first harmonic;
a plurality of mixers configured to mix the weighted current
signals with local oscillator signals of successive phase shifts to
generate a plurality of mixer signals, the plurality of mixers
having first mixers along the first signal path and one or more
second mixers along the second signal path, the one or more second
mixers configured to receive the input RF signal directly; a
plurality of feedback impedance cells configured to convert the
plurality of mixer signals into corresponding voltage signals, the
plurality of feedback impedance cells having first feedback
impedance cells along the first signal path and second feedback
impedance cells along the second signal path, the second feedback
impedance cells coupled to the one or more second mixers; and a
switch network configured to switch between outputs of the first
mixers to each input to the first feedback impedance cells.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of U.S. patent
application Ser. No. 13/736,895, entitled "WIDEBAND RECEIVER ROBUST
TO RADIO FREQUENCY HARMONICS, filed Jan. 8, 2013, which claims the
benefit of U.S. Provisional Patent Application No. 61/737,077,
entitled "WIDEBAND RECEIVER ROBUST TO RADIO FREQUENCY AND BASEBAND
HARMONICS," filed Dec. 13, 2012, all of which are hereby
incorporated by reference in their entirety for all purposes.
BACKGROUND
[0002] Wideband receivers are configured to provide filtering to
most out-of-band blocker signals before any baseband amplification.
Wideband receivers can employ passive mixers that immediately
down-convert an input current to baseband. A transimpedance
amplifier (TIA) then converts any baseband current in the receive
band to voltage.
[0003] A received radio frequency (RF) signal may include an
unwanted blocker signal at frequency f.sub.b, and a wanted signal
at frequency f.sub.w, which may be .DELTA.f.sub.b greater than
f.sub.b, e.g., f.sub.w=f.sub.b+.DELTA.f.sub.b. However, these
unwanted blocker signals experience some amplification around the
RF band. This unnecessary amplification of unwanted blocker signals
has potential to saturate the receiver, thus resulting in
degradation of performance.
[0004] Additionally, the output of the TIA may have an unwanted
signal that originates from harmonics of the wanted signal that
experiences some amplification. Although the bandwidth around which
these harmonics experience amplification can be very small, blocker
signals located at these harmonic frequencies potentially cause the
receiver to saturate, introducing distortion. For example, a
receiver gain of 20.times. could potentially amplify a 1V blocker
signal to 20V. This unnecessary amplification of blocker signals at
the TIA output has potential to also saturate the receiver, thus
resulting in further degradation of performance.
SUMMARY
[0005] A circuit and/or method is provided for a noise-cancelling
receiver with baseband harmonic rejection and RF harmonic
rejection, substantially illustrated by and/or described in
connection with at least one of the figures, as set forth more
completely in the claims.
BRIEF DESCRIPTION OF THE DRAWINGS
[0006] Certain features of the subject disclosure are set forth in
the appended claims. However, for purpose of explanation, several
implementations of the subject disclosure are set forth in the
following figures.
[0007] FIG. 1 illustrates a block diagram of a wireless
communication device in accordance with one or more
implementations.
[0008] FIG. 2 illustrates a schematic diagram of a noise-cancelling
receiver front-end in accordance with one or more
implementations.
[0009] FIG. 3 illustrates a waveform diagram of local oscillator
signals for use by the receiver front-end illustrated in FIG. 2 in
accordance with one or more implementations.
[0010] FIG. 4 illustrates a waveform diagram of an input impedance
seen by the receiver front-end illustrated in FIG. 2 in accordance
with one or more implementations.
[0011] FIG. 5 illustrates a schematic diagram of an oversampling
downconversion passive mixer in accordance with one or more
implementations.
[0012] FIG. 6 illustrates a schematic diagram of a receiver
front-end with baseband harmonic rejection in accordance with one
or more implementations.
[0013] FIG. 7 illustrates a schematic diagram of an oversampling
downconversion passive mixer with baseband harmonic rejection
provided in the receiver front-end illustrated in FIG. 6 in
accordance with one or more implementations.
[0014] FIG. 8 illustrates a schematic diagram of a single
transimpedance amplifier used in the oversampling downconversion
passive mixer illustrated in FIG. 7 in accordance with one or more
implementations.
[0015] FIG. 9 illustrates a conceptual diagram of the oversampling
downconversion passive mixer with baseband harmonic rejection in
accordance with one or more implementations.
[0016] FIG. 10 illustrates a schematic diagram of a current-source
downconversion path provided in the receiver front-end illustrated
in FIG. 2 in accordance with one or more implementations.
[0017] FIG. 11 illustrates a schematic diagram of an oversampling
passive mixer provided in the current-source downconversion path
illustrated in FIG. 10 in accordance with one or more
implementations.
[0018] FIG. 12 illustrates a waveform diagram of a gain realized by
the current-source downconversion path illustrated in FIG. 10 in
accordance with one or more implementations.
[0019] FIG. 13 illustrates a schematic diagram of a single-ended
receiver front-end with enhanced harmonic rejection in accordance
with one or more implementations.
[0020] FIG. 14 illustrates a conceptual diagram of a single-ended
current-source downconversion path provided in the receiver
front-end illustrated in FIG. 13 in accordance with one or more
implementations.
[0021] FIG. 15 illustrates a waveform diagram of a gain realized by
the current-source downconversion path illustrated in FIG. 14 in
accordance with one or more implementations.
[0022] FIG. 16 illustrates a schematic diagram of a differential
receiver front-end with enhanced harmonic rejection in accordance
with one or more implementations.
[0023] FIG. 17 illustrates a schematic diagram of a differential
current-source downconversion path provided in the receiver
front-end illustrated in FIG. 16 in accordance with one or more
implementations.
[0024] FIG. 18 illustrates a schematic diagram of a differential
passive mixer provided in the current-source downconversion path
illustrated in FIG. 17 in accordance with one or more
implementations.
DETAILED DESCRIPTION
[0025] It is understood that other configurations of the subject
disclosure will become readily apparent to those skilled in the art
from the following detailed description, wherein various
configurations of the subject disclosure are shown and described by
way of illustration. As will be realized, the subject disclosure is
capable of other and different configurations and its several
details are capable of modification in various other respects, all
without departing from the scope of the subject disclosure.
Accordingly, the drawings and detailed description are to be
regarded as illustrative in nature and not as restrictive.
[0026] According to some implementations, the subject technology
relates to a radio frequency (RF) receiver that includes a first
set of transconductance cells configured to produce respective
weighted current signals proportional to an input voltage signal.
The RF receiver includes multiple frequency conversion cells
coupled to the first set of transconductance cells and configured
to mix the weighted current signals with a plurality of
non-overlapping local oscillator (LO) signals to produce
downconverted current signals. The RF receiver also includes
multiple transimpedance amplifiers coupled to the plurality of
frequency conversion cells and configured to produce output voltage
signals proportional to the downconverted current signals. Each of
the transimpedance amplifiers includes a second set of
transconductance cells. Each of the first set and second set of
transconductance cells has an effective transconductance of a first
magnitude for frequency components of the input voltage signal
arising from a first harmonic and an effective transconductance of
a second magnitude less than the first magnitude for frequency
components of the input voltage signal arising from harmonics at
integer multiples of the first harmonic.
[0027] FIG. 1 illustrates a block diagram of a wireless
communication device 100 in which the subject technology may be
used in accordance with one or more implementations. Referring to
FIG. 1, the wireless communication device 100 includes an antenna
102, a receiver front-end 104, a baseband processing module 106, a
processor 108, a memory 110, and a local oscillator generation
module (LOGEN) 112. The blocks represented in FIG. 1 may be
integrated on semiconductor substrates. For example, the blocks
104-112 may be realized in a single system-on-chip, or realized in
a multi-chip chipset.
[0028] The antenna 102 is configured to transmit and/or receive
wireless signals over a range of frequencies. Although a single
antenna is illustrated, the subject disclosure is not so limited.
The receiver front-end 104 includes logic, circuitry and/or
interfaces that is operable to receive and process signals from the
antenna 102. The receiver front-end 104, for example, may be
operable to amplify and/or down-covert received wireless signals.
The receiver front-end 104 also may be operable to cancel noise
arising from impedance matching and may be linear over a wide range
of frequencies. In this regard, the receiver front-end 104 may
receive signals in accordance with a variety of wireless standards.
The receiver front-end 104 may be applicable to standards,
including but not limited to, Wi-Fi, WiMAX, Bluetooth, and various
cellular standards.
[0029] The baseband processing module 106 includes logic, circuitry
and/or interfaces that is operable to perform processing of
baseband signals. The baseband processing module 106 may, for
example, analyze received signals and generate control and/or
feedback signals for configuring various components of the wireless
communication device 100 such as the receiver front-end 104. The
baseband processing module 106 is operable to encode, decode,
transcode, modulate, demodulate, encrypt, decrypt, scramble,
descramble, and/or otherwise process data in accordance with
wireless standards.
[0030] The processor 108 includes logic, circuitry and/or
interfaces that is operable to process data and/or control
operations of the wireless communication device 100. In this
regard, the processor 108 is configured to provide control signals
to various other portions of the wireless communication device 100.
The processor 108 may control transfers of data between various
portions of the wireless communication device 100. Additionally,
the processor 108 may provide for implementation of an operating
system or otherwise execute code to manage operations of the
wireless communication device 100.
[0031] The memory 110 includes logic, circuitry and/or interfaces
for storage of various types of information such as received data,
generated data, code, and/or configuration information. The memory
110 may include a non-transitory storage medium, including but not
limited to, RAM, ROM, flash, and/or magnetic storage. According to
some implementations of the subject disclosure, information stored
in the memory 110 is utilized for configuring the receiver
front-end 104 and/or the baseband processing module 106.
[0032] The LOGEN 112 includes logic, circuitry and/or interfaces
that is operable to generate oscillating signals at multiple
frequencies. The LOGEN 112 may be operable to generate digital
and/or analog signals. In this regard, the LOGEN 112 is operable to
generate clock signals and/or sinusoidal signals. Characteristics
of the oscillating signals such as the frequency and duty cycle may
be determined based on control signals from the processor 108
and/or the baseband processing module 106.
[0033] In some implementations, the processor 108 configures the
various components of the wireless communication device 100 based
on a wireless standard according to which it is desired to receive
signals. Wireless signals are received via the antenna 102,
amplified and down-converted by the receiver front-end 104. The
baseband processing module 106 may perform noise estimation and/or
noise cancellation, decoding, and/or demodulation of the baseband
signals. In this regard, information in the received signal is
recovered and utilized appropriately. For example, the information
may be audio and/or video to be presented to a user of the wireless
communication device 100, data to be stored to the memory 110,
and/or information affecting and/or enabling operation of the
wireless communication device 100.
[0034] FIG. 2 illustrates a schematic diagram of a noise-cancelling
receiver front-end 200 in accordance with one or more
implementations. The main path of the receiver front-end 204
represents a mixer-first front-end (i.e. path through which
I.sub.RFIN is amplified). The receiver front-end 200 can be used in
several different devices such as cell phones, wireless modems,
wireless routers and access points to receive wireless RF
signals.
[0035] The receiver front-end 200 connects to an antenna (not
shown), and includes two down-conversion paths (e.g., a main
downconversion path 204 and an auxiliary downconversion path 206),
and a weighting and recombination module 230. The receiver
front-end 200 also includes a transconductance cell 222, passive
mixers 216 and 220, and transimpedance amplifiers 224 and 226.
[0036] In one or more implementations, the transconductance cell
222 is included in the auxiliary down-conversion path 206 to
convert an RF signal input 205 (e.g., V.sub.RFIN) into a
corresponding current signal. The transconductance cell 222 can be
implemented, for example, as an inverter. The transconductance cell
222 includes logic, circuitry and/or interfaces that is operable to
output a current that is proportional to a voltage input to the
transconductance cell 222.
[0037] The passive mixers 216 and 220 each include logic and/or
circuitry that may provide for generation of inter-modulation
products resulting from mixing the RF signal input 205 with LO
signals (e.g., LO.sub.0, LO.sub.1, . . . , LO.sub.M-1) generated by
the LOGEN 112 (FIG. 1). In one or more implementations, the passive
mixers 216 and 220 are metal-oxide semiconductor field effect
transistors (MOSFETs). Specifically, the passive mixers 216 and 220
are n-channel MOSFETs (e.g., NMOS). However, the passive mixers 216
and 220 can be implemented using other switching devices. In one or
more implementations, the passive mixers 216 and 220 are configured
to operate in their linear region when powered on.
[0038] The transimpedance amplifier (TIA) 226 includes logic,
circuitry and/or interfaces that is operable to output a voltage
that is proportional to a current input to the TIA 226. Similarly,
the TIA 224 includes logic, circuitry and/or interfaces that is
configured to output a voltage that is proportional to a current
input to the TIA 224.
[0039] In operation, the two down-conversion paths 204 and 206
down-convert an RF signal received by the antenna (not shown) to
baseband or some inter-frequency (IF) to cancel thermal noise
generated by an input matching resistance, while avoiding voltage
gain of the RF signal. The main down-conversion path 204 includes
the passive mixer 216 to down-convert the current signal
I.sub.RF.sub.IN by an amount equal to the frequency of a local
oscillator signal (e.g., LO.sub.0-LO.sub.M-1). The current signal
I.sub.RF.sub.IN includes both the RF signal and the thermal noise
signal generated by the input matching resistance. The auxiliary
down-conversion path 206 includes the passive mixer 220 to
down-convert the voltage signal V.sub.RF.sub.IN by an amount equal
to the frequency of a local oscillator signal used by the passive
mixer 216. The frequency of the LO signal can be tuned, for
example, over a frequency range based on a position of a desired
channel in the spectrum of the received RF signal. Like the current
signal I.sub.RF.sub.IN the voltage signal V.sub.RF.sub.IN includes
both the RF signal and the thermal noise signal generated by an
input matching resistance.
[0040] Although the thermal noise signal and the RF signal appear
in both the current signal I.sub.RF.sub.IN and the voltage signal
V.sub.RF.sub.IN, it can be shown that the thermal noise signal in
V.sub.RF.sub.IN is 180 degrees out of phase with the thermal noise
signal in I.sub.RF.sub.IN, while the RF signal in V.sub.RF.sub.IN
is in-phase with the RF signal in I.sub.RF.sub.IN. Given this, the
transimpedance amplifiers 224 and 226 can be respectively used to
convert and scale the down-converted current signals at the output
of the passive mixers 216 and 220, into two voltage signals
V.sub.OUT1 and V.sub.OUT2 such that, when V.sub.OUT1 and V.sub.OUT2
are added together by the weighting and recombination module 230
(or potentially sensed differentially depending on the relative
polarities of the gains applied by the transfer functions), the
thermal noise generated by input matching resistance is canceled,
while the RF signal received by the antenna 102 is reinforced. In
one or more implementations, the voltage signals V.sub.OUT1 and
V.sub.OUT2 define the In-phase (I) and Quadrature (Q) components of
the RF signal input 205.
[0041] The transfer functions of the transimpedance amplifiers 224
and 226 are determined based on their feedback networks. According
to some implementations, the transimpedance amplifiers 224 and 226
include a resistive element and a capacitive element in parallel in
their feedback networks. Capacitive elements can be included for
stability purposes including resistive elements to set a
current-to-voltage conversion gain for the transimpedance
amplifiers 224 and 226, respectively. The number of transimpedance
amplifiers included in each of the transimpedance amplifiers 224
and 226 may be at least comparable to the number of LO signals
employed.
[0042] By employing the passive mixers 216 and 220 (which are
bidirectional) and the transimpedance amplifiers 224 and 226, the
virtual ground at the negative summing nodes of the transimpedance
amplifiers 224 and 226 appears ideally at the RF side of the
passive mixers 216 and 220 where the RF signal from the antenna 102
is received, thereby suppressing voltage swing prior to baseband or
intermediate frequency (IF) filtering to remove blockers. Thus,
because the receiver front-end 200 suppresses noise, while
minimizing voltage gain, the receiver front-end 200 can be
considered blocker-tolerant. In addition, because the passive mixer
216 is bidirectional, the noise of the transimpedance amplifier 224
up-converts at the input of the receiver front-end 200 and is
canceled along with the thermal noise generated by the input
matching resistance. The noise of the transimpedance amplifier 226
contributes negligibly when driven by a current source, such as the
transconductance cell 222.
[0043] Following down-conversion of the RF signal received by the
antenna 102 by the down-conversion paths 204 and 206 as described
above, the baseband processing module 106 (FIG. 1) processes
V.sub.OUT1 and V.sub.OUT2 using one or more processors and/or
circuits. For example, baseband processing module 106 can low-pass
filter V.sub.OUT1 and V.sub.OUT2, either separately or after having
been added (or subtracted) together by the weighting and
recombination module 230, to remove blockers and other
interference. In addition, baseband processing module 106 can
further perform digitization of V.sub.OUT1 and V.sub.OUT2 either
separately or after having been added (or subtracted) together,
using one or more analog-to-digital converters (ADCs). The ADCs can
be, for example, delta-sigma ADCs. In addition, baseband processing
module 106 can perform low-pass filtering and digitization of
V.sub.OUT1 and V.sub.OUT2 in any order relative to each other.
Further, baseband processing module 106 can demodulate the
down-converted RF signal contained within V.sub.OUT1 and V.sub.OUT2
to recover information.
[0044] It should be noted that, in some implementations, the
receiver front-end 200 can be further constructed as a fully
differential receiver front-end. In other words, the receiver
front-end 200 can be further constructed to process a differential
RF signal received by the antenna 102 by replacing the
transconductance cell 222 with a differential transconductance cell
and by replacing the passive mixers in each down-conversion path
204 and 206 with differential passive mixers.
[0045] FIG. 3 illustrates a waveform diagram of local oscillator
signals for use by the receiver front-end 200 illustrated in FIG. 2
in accordance with one or more implementations. The LO signals 300
each correspond to a fundamental frequency (e.g., .omega..sub.LO).
Additionally, the LO signals 300 have successive phase shifts
substantially equal to 360.degree./M and have duty cycles
substantially equal to 1/M, where M is a number of LO signals
employed.
[0046] According to some implementations, the LO signals 300
approximate a square waveform. Unlike a single-tone sinusoidal
waveform, the square waveform contains harmonics at specific
multiples of the fundamental frequency of the LO signal.
Consequently, the output signal generated by the passive mixers 216
and 220 (FIG. 2) using the square waveform as a LO signal can
contain harmonics corresponding to the LO frequency.
[0047] For example, when the number of harmonics presented around
the fundamental frequency with a magnitude greater than the
threshold magnitude is equal to eight, the LOGEN 112 (FIG. 1) can
be configured to generate eight different LO signals (e.g.,
LO.sub.0, LO.sub.1, LO.sub.2, up to LO.sub.7) to address the eight
harmonics. These eight LO signals each have successive phase shifts
substantially equal to 360/8 degrees or 45 degrees. In addition,
the duty cycle of each LO signal is substantially equal to 1/8 or
12.5%.
[0048] FIG. 4 illustrates a waveform diagram of an input impedance
seen by the receiver front-end 200 illustrated in FIG. 2 in
accordance with one or more implementations. The waveform diagram
400 depicts the input impedance (or resistance (.OMEGA.)) as a
function of the radio frequency (RF) normalized to local oscillator
(LO) frequency. The input impedance seen from an input to the
passive mixers 216 and 220 (FIG. 2), is equivalent to a switch
resistance (e.g., R.sub.SW) except around harmonics of the LO
frequency where the input impedance is equivalent to:
Z IN .apprxeq. R SW + Z BB M ( 1 ) ##EQU00001##
where M is a positive integer and represents the number of LO
phases employed, R.sub.SW is the switch resistance of the passive
mixers 216 and 220, and Z.sub.BB is the baseband impedance of the
transimpedance amplifier 224 (FIG. 2). The harmonics having an
input impedance greater than the switch resistance provide a
pass-band at integer multiples of a fundamental frequency. As such,
harmonics can be seen from the first frequency multiple (e.g.,
1.omega..sub.LO) up to the seventh frequency multiple (e.g.,
7.omega..sub.LO).
[0049] The switch resistance represents the impedance of switching
devices that may be provided in the passive mixers 216 and 220. At
these frequencies, the input impedance is equal to the baseband
impedance frequency shifted to these frequencies (e.g.,
down-conversion, up-conversion). Thus, the input impedance, if
inserted at a node inside the receiver front-end 200 (FIG. 1), all
incoming frequencies at that node except those residing at the
fundamental frequency (e.g., .omega..sub.LO) and its harmonics are
subject to attenuation. While it may be desirable to have an
amplification system centered only around the LO frequency,
components from around 2.omega..sub.LO up to around
7.omega..sub.LO, for example, will receive some amplification
(e.g., with less gain compared with the desired input components at
.omega..sub.LO), but with minimal folding.
[0050] FIG. 5 illustrates a conceptual diagram of an oversampling
down-conversion path 500 in accordance with one or more
implementations. According to some implementations, the
oversampling down-conversion path 500 is composed of switches
502.sub.1-502.sub.M and transimpedance amplifiers (TIAs)
504.sub.1-504.sub.M, where M is a positive integer and represents
the number of LO phases employed. The switches 502.sub.1-502.sub.M
may collectively form the passive mixer 216 as discussed in FIG. 2.
The TIAs 504.sub.1-504.sub.M may collectively form the TIA 224 as
discussed in FIG. 2. The switches 502.sub.1-502.sub.M receive
non-overlapping LO signals that are shifted by 360/M degrees from
one another as illustrated in FIG. 3.
[0051] In operation, each of the switches 502.sub.1-502.sub.M
down-converts an RF current input (i.sub.RF(t)) using an input
impedance profile 506, which defines impedances at harmonics of the
corresponding RF or LO frequency. In a case of a mixer-first
topology, the RF current input represents the input to the receiver
front-end 104 (FIG. 1). Each of the TIAs 504.sub.1-504.sub.M
receives a baseband current (I.sub.BBM(t)) for conversion into a
corresponding voltage output.
[0052] According to some implementations, the impedance at a first
harmonic may be approximated as:
Z IN .apprxeq. R SW + Z BB M * G M ( 2 ) ##EQU00002##
where M is equal to the number of LO phases employed, R.sub.SW is
the switch resistance of a respective one of the switches
502.sub.1-502.sub.M, and 1/G.sub.M is the input impedance of a
respective one of the TIAs 504.sub.1-504.sub.M. This corresponds to
the series sum of the resistance of a single mixer switch and the
up-converted impedance of the baseband TIAs. This is an idealized
and simplified understanding, whereas an actual circuit
implementation may result in a significantly altered impedance
profile.
[0053] A waveform diagram 508 illustrates the conversion gain
(e.g., y-axis) at each of the TIAs 504.sub.1-504.sub.M as a
function of a harmonic multiple (e.g., x-axis). As illustrated,
down-converted harmonics arising from integer multiples of the LO
frequency (e.g., 2.sup.nd through 7.sup.th harmonics) may be
present at the output of the TIAs. Accordingly, the down-conversion
of unwanted signals produces folding terms that can saturate the
receiver front-end 104 due to the gain compression caused by the
blocker signals experiencing a voltage gain. The TIAs
504.sub.1-504.sub.M are typically configured to provide large
voltage amplification. Therefore large blocker signals can
introduce distortion, which can corrupt a wanted signal 510.
[0054] FIG. 6 illustrates a schematic diagram of a receiver
front-end 600 with baseband harmonic rejection in accordance with
one or more implementations. The subject technology provides for a
baseband technique to prevent amplification of down-converted
harmonic blockers by using multiple phases of the passive mixer
output. In addition, the baseband technique improves the harmonic
rejection properties of mixer-first receiver topologies.
[0055] According to some implementations, the subject technology
relates to an apparatus within the receiver front-end 600 that is
configured for baseband amplification with harmonic rejection,
particularly for large harmonic blockers. The apparatus includes
means for receiving a mixer signal down-converted from a radio
frequency signal. The apparatus also includes means for converting
the mixer signal into a weighted current signal proportional to a
voltage corresponding to the mixer signal based on an effective
transconductance. The apparatus also includes means for outputting
the mixer signal with amplification using a feedback impedance such
that a wanted signal receives the amplification and unwanted
signals are rejected without amplification. The effective
transconductance has a first magnitude for frequency components of
the down-converted signal arising from a first harmonic and the
effective transconductance has a second magnitude less than the
first magnitude for frequency components of the down-converted
signal arising from harmonics at integer multiples of the first
harmonic.
[0056] In one or more implementations, the receiver front-end 600
provides transimpedance amplifiers (TIAs) 602 and 604 to be
configured to receive M downconversion signals from respective ones
of the passive mixers 216 and 220. Each of the TIAs 602 and 604
includes multiple transconductance cells with a particular
transconductance weighting to output a weighted current signal.
Outputs of the TIAs 602 and 604 are coupled to a common feedback
impedance, which is coupled to an input of a respective one of the
TIAs 602 and 604 to convert the weighted current signal into a
voltage output with a voltage gain such that the wanted signal
arising from a first harmonic realizes the voltage gain and
unwanted signals corresponding to harmonics at integer multiples of
the first harmonic are rejected without amplification.
[0057] Referring to FIG. 6, the receiver front-end 600 connects to
an antenna (not shown), and includes two down-conversion paths 204
and 206, and a weighting and recombination module 230. The receiver
front-end 600 also includes a transconductance cell 222, passive
mixers 216 and 220, and TIAs 602 and 604. The passive mixers 216
and 220 each include logic and/or circuitry that may provide for
generation of inter-modulation products resulting from mixing an RF
signal input 205 (e.g., RF.sub.IN) with LO signals generated by the
LOGEN 112 (FIG. 1).
[0058] The receiver front-end 600 has a similar structure as the
receiver front-end 200 illustrated in FIG. 2 and operates in
substantially the same manner. However, the receiver front-end 600
provides the transimpedance amplifiers 602 and 604 having enhanced
harmonic rejection circuitry. The transimpedance amplifiers 602 and
604 are configured to receive multiple mixer signals from the
passive mixers 216 and 220 that were mixed with all of the LO
signals generated by the LOGEN 112. That is, each respective TIA
receives a bus of mixer signals to selectively amplify frequency
components of the RF signal input 205 that correspond to a wanted
harmonic.
[0059] FIG. 7 illustrates a schematic diagram of an oversampling
down-conversion path 700 with baseband harmonic rejection in
accordance with one or more implementations. Referring to FIG. 7,
the oversampling down-conversion path 700 includes switches
702.sub.1-702.sub.M and a bank of transimpedance amplifiers 602.
The bank of transimpedance amplifiers 602 is composed of TIAs
704.sub.1-704.sub.M. Each of the TIAs 704.sub.1-704.sub.M includes
a feedback circuit (e.g., 706.sub.1-706.sub.M). Each of the
feedback circuits 706.sub.1-706.sub.M includes one or more
components (e.g., a capacitor, a resistor) for setting a gain
and/or frequency response of the corresponding one of the TIAs
704.sub.1-704.sub.M. According to some implementations, M is equal
to an even integer, including but not limited to, 8, 16, and
32.
[0060] In one or more implementations, the TIAs 704.sub.1-704.sub.M
are configured to receive M down-conversion signals from respective
ones of the switches 702.sub.1-702.sub.M. Each of the TIAs
704.sub.1-704.sub.M includes multiple transconductance cells with a
particular transconductance weighting to output a weighted current
signal. A feedback impedance (e.g., feedback circuits
706.sub.1-706.sub.M) is coupled between an input of a respective
one of the transconductance cells and outputs of the TIAs
704.sub.1-704.sub.M to convert the weighted current signal into a
voltage output with a voltage gain such that the wanted signal
arising from a first harmonic realizes the voltage gain and
unwanted signals corresponding to harmonics at integer multiples of
the first harmonic are rejected without amplification.
[0061] Each of the TIAs 704.sub.1-704.sub.M is configured to
provide baseband amplification with harmonic rejection. The TIAs
704.sub.1-704.sub.M are operable to convert an input current signal
into an output voltage signal proportional to the input current
signal. For example, each of the TIAs 704.sub.1-704.sub.M amplifies
a down-converted signal from an output of one of the switches
702.sub.1-702.sub.M into a corresponding voltage signal.
[0062] In one or more implementations, the switches
702.sub.1-702.sub.M collectively form the passive mixer 216 (FIG.
6) and each include, for example, a single NMOS transistor.
Multiple LO signals (e.g., LO<M-1:0>), each phase-shifted
with respect to one another, are generated by the LOGEN 112 (FIG.
1). Each of the switches 702.sub.1-702.sub.M receives one of the LO
signals. For example, for M=8, 8 LO signals corresponding to 8
phases are generated and each are configured to have a 12.5%
duty-cycle such that only one of the 8 LO signals is in a
logic-high state at a given time instant. As such, the outputs from
the switches 702.sub.1-702.sub.M are identical in magnitude but
shifted in phase by 360/M degrees.
[0063] Each of the TIAs 704.sub.1-704.sub.M receives a different
ordering of the passive mixer outputs. For example, the first TIA
704.sub.1 receives a signal (e.g., IN<0>) via a first input
port and the remaining signals (e.g., IN<1:M-1>) via
respective input ports. The second TIA 704.sub.2 receives a mixed
signal (e.g., IN<1>) via a first input port and the remaining
mixed signals (e.g., ordered as IN<2:M-1>, IN<0>) via
respective input ports. Additionally, the last TIA 704.sub.M
receives a mixed signal (e.g., IN<M-1>) via a first port and
the remaining mixed signals (e.g., IN<0:M-2>) via respective
ports.
[0064] In operation, an input current signal, I.sub.RF.sub.IN is
received and processed via a signal path including the switches
702.sub.1-702.sub.M and the TIAs 704.sub.1-704.sub.M. Each of the
switches 702.sub.1-702.sub.M is configured to mix the input current
signal with a respective LO signal to downconvert the received
input current signal to baseband (e.g., reduce frequency to a
baseband or intermediate frequency). The baseband output of each of
the switches 702.sub.1-702.sub.M is input to a corresponding one of
the TIAs 704.sub.1-704.sub.M. Each of the TIAs 704.sub.1-704.sub.M
converts the current output of a corresponding one of the switches
502.sub.1-502.sub.M to a corresponding voltage. There is a gain
associated with the current-to-voltage conversion.
[0065] The combined output voltage, V.sub.OUT<M-1:0> may be
preconfigured and/or dynamically configured during operation of the
oversampling down-conversion path 700 to maintain, within a
tolerance, the following relationship:
V.sub.OUT=R.sub.RFI.sub.IN, where m=1+/-kM, otherwise 0. (3)
where R.sub.RF is the resistance of the input RF signal, I.sub.IN
is the current of the input RF signal, k is any integer, and M
represents the number of LO phases employed. That is, the effective
transconductance for harmonics around a wanted signal causes a
current change through the transconductance cell to realize a
voltage output; whereas, the effective transconductance for
harmonics at integer multiples of the wanted signal is equal to
zero.
[0066] The effective input impedance is approximately given by:
Z.sub.IN=1/G.sub.m, where m=1+/-kM, otherwise .infin.. (4)
where m is the LO harmonic from where the signal originated (e.g.,
m=1 for a wanted signal, and m=3 from an unwanted blocker from
around the 3.sup.rd LO harmonic), k is any integer, M represents
the number of LO phases employed.
[0067] FIG. 8 illustrates a schematic diagram of a single
transimpedance amplifier used in the oversampling down-conversion
path 700 illustrated in FIG. 7 in accordance with one or more
implementations. The receiver front-end 200, as discussed with
respect to FIG. 2, is enhanced with baseband harmonic rejection.
For example, the TIAs 704.sub.1-704.sub.M (FIG. 7) can each be
expanded to include multiple transconductance cells
802.sub.1-802.sub.M (e.g., labeled gm.sub.0-gm.sub.M-1) to receive
the down-converted signals from respective ones of the switches
702.sub.1-702.sub.M (FIG. 7), where M is a positive integer and
represents the number of LO phases employed. A feedback impedance,
which is coupled between an input of one of the transconductance
cells 802.sub.1-802.sub.M (e.g., 802.sub.1) and outputs of the
transconductance cells, is provided to the transconductance cell
(e.g., gm.sub.0) to convert a down-converted signal into a voltage
output with a voltage gain.
[0068] Each down-converted signal has a wanted signal arising from
a first harmonic and unwanted signals arising from harmonics of the
wanted signal. In one or more implementations, the input voltage,
V.sub.IN, to each transconductance cell is preconfigured and/or
dynamically configured during operation of the single
transimpedance amplifier 800 to maintain, the following
relationship:
V IN .angle. m * ( M - 1 ) M ( 3 ) ##EQU00003##
where m is the LO harmonic from where the signal originated (e.g.,
m=1 for the wanted signal, and m=3 from an unwanted blocker from
around the 3.sup.rd LO harmonic), and M represents the number of LO
phases employed. Here, the input voltage defines a geometric
relationship between the voltages corresponding to respective
harmonics applied to the transconductance cells
802.sub.1-802.sub.M. As such, the input voltages cause each of the
transconductance cells 802.sub.1-802.sub.M to be weighted
accordingly such that their effective transconductance realizes a
current change through the transconductance cell for outputs
corresponding to a wanted signal.
[0069] By appropriately weighting the M transconductance cells in
each of the TIAs 704.sub.1-704.sub.M, the harmonic rejection
circuit can be configured to amplify the wanted down-converted
signal arising from the first harmonic (e.g., m=1), but reject
without amplification all unwanted signals that arise from
harmonics of the wanted signal (up to the M-1.sup.th harmonic).
This is so because the weighting causes the effective
transconductance of the transconductance cell 802.sub.1 to be large
when excited by signals arising from m=1, but zero otherwise. In
other words, the weighting causes certain current signals to shift
in phase such that signals with opposing phases cancel one another
at the output (e.g., V.sub.OUT), thus resulting in no voltage gain
for the canceled current signals. Given that there is no
amplification of harmonic blockers at outputs of the TIAs
704.sub.1-704.sub.M with the enhanced harmonic rejection circuit,
no harmonic blockers up to the (M-1).sup.th harmonic can saturate
the receiver. According to some implementations, the
transconductance cells 802.sub.1-802.sub.M are configured to only
amplify a signal from a harmonic (not just M=1, e.g., M=3), and
further configured to only amplify signals arising from more than
one harmonic multiple of LO (e.g., signals from M=1 and M=3).
[0070] In one or more implementations, each of the multiple
transconductance cells 802.sub.1-802.sub.M has a transconductance
weighting equivalent to:
gm X = k [ 1 + cos ( 2 .pi. X M ) ] , X = { 0 , 1 , 2 , M - 1 ] ( 6
) ##EQU00004##
or possibly,
gm X = k [ cos ( 2 .pi. X M ) ] , X = { 0 , 1 , 2 , M - 1 ] ( 7 )
##EQU00005##
when a fully differential receiver (that uses fully-differential
passive mixers) is employed, where X is an integer that represents
an individual transconductance cell (e.g., 802.sub.1-802.sub.M), M
is the number of LO phases employed, and k is an arbitrary constant
that determines the effective transconductance of the
transconductance cell. As such, the transconductance gain is
shifted to reject integer multiples of the wanted signal. The
transconductance based on equation (6) can provide more immunity to
blockers arising from even-numbered harmonics than the
transconductance based on equation (7), but may utilize more
current for the same noise performance.
[0071] Alternatively, when differential mixers are employed (in
other words fully differential RF inputs are available), each of
the transconductance cells 802.sub.1-802.sub.M has a
transconductance weighting equivalent to equation (7), as shown
above, where M is a positive integer and represents the number of
LO phases employed. As such, the transconductance gain is shifted
to reject odd integer multiples of the wanted signal only.
[0072] The transconductance cells 802.sub.1-802.sub.M can be
implemented using other transconductance devices. The harmonic
rejection circuit can be implemented in a voltage amplifier,
current amplifier, operational amplifier, signal amplifier or any
variation of transconductance. The harmonic rejection circuit also
can be applied to single-stage amplifiers or amplifiers employing
multiple amplification stages. According to some implementations,
the single transimpedance amplifier 800 and the above-described
elements are varied and are not limited to the functions,
structures, configurations, implementations or examples
provided.
[0073] FIG. 9 illustrates a conceptual diagram 900 of the
oversampling down-conversion path 700 with baseband harmonic
rejection illustrated in FIG. 7 in accordance with one or more
implementations. In operation, each of the switches
702.sub.1-702.sub.M receives an RF current signal (e.g.,
I.sub.RF(t)) using an input impedance profile 906 that defines
impedances at harmonics of the corresponding radio frequency or LO
frequency. For example, the resistance at a first harmonic may be
approximated according to equation (2), shown above, where M
represents the number of LO phases employed, R.sub.SW is the switch
resistance of a respective one of the switches 702.sub.1-702.sub.M,
and 1/G.sub.M is the input impedance of a respective TIA in the
transimpedance amplifier 702. This corresponds to the series sum of
the resistance of a single switch and the up-converted impedance of
the baseband TIAs. This is an idealized and simplified
understanding, whereas an actual circuit implementation may result
in a significantly altered impedance profile. Each of the TIAs
provided in the transimpedance amplifier 702 receives a baseband
current (I.sub.BBM(t)) for conversion into a corresponding voltage
output. Here, FIG. 9 assumes M=8.
[0074] Based on the input impedance profile 906, only the first
harmonic excites a voltage gain since the transconductance
weighting ensures that unwanted harmonics do not excite a current
from the transconductance cell (or equivalently the effective
transconductance of the transconductance cell to drop to zero).
Additionally, the input impedance profile 902 defines impedances
between the 2.sup.nd and 7.sup.th harmonics as substantially large
(e.g., infinity). Because the unwanted signals (or folding terms)
do not realize voltage amplification, the gain compression is
reduced around these LO harmonic frequencies.
[0075] A waveform diagram 908 illustrates the conversion gain at
each of the TIAs provided in the transimpedance amplifier 704 as a
function of a harmonic multiple. As illustrated, a wanted signal
910 and minimal feed-through signals are realized at certain
frequencies in the receive band. Accordingly, the folding terms are
reduced such that only unwanted folding terms around the 7.sup.th
hand 9.sup.th harmonics of the wanted signal 910 are realized,
which are of negligible magnitude. The wanted signal 910 around the
1.sup.st harmonic realizes a voltage gain based on the input
impedance profile 902.
[0076] FIG. 10 illustrates a schematic diagram of a current-source
downconversion path 1000 provided in the receiver front-end 200
illustrated in FIG. 2 in accordance with one or more
implementations. The current-source downconversion path 1000
employs a transconductance cell 1002 coupled to an M-phase
oversampling downconversion path 1004. The M-phase oversampling
downconversion path 1004 includes a passive mixer 1006 and
transimpedance amplifiers (TIAs) 1008.
[0077] In one or more implementations, the M-phase oversampling
downconversion path 1004, illustrated in FIG. 10, provides viable
implementations for use within many RF receiver designs. However,
for wideband RF receivers, the higher-order harmonic effects that
result from the passive mixer 1006 of the M-phase oversampling
downconversion path 1004 can cause significant interference with a
desired portion of the RF signal.
[0078] Referring to FIG. 10, the M-phase oversampling
downconversion path 1004 is single-ended and processes a
single-ended RF signal, via a transconductance cell 1002, received
at a node 1001. The passive mixer 1006 is configured to translate
the RF current to a baseband frequency. As the passive mixer is
bidirectional, the passive mixer 1006 is also translates the input
impedance of the TIAs 1008 to a higher frequency substantially
equal to a fundamental frequency (e.g., .omega..sub.LO) of local
oscillator (LO) signals 1010 received by the passive mixer 1006. In
addition, the transconductance cell 1002 realizes a voltage
gain
( e . g . , V OUT V IN ) ##EQU00006##
between the node 1001 and the node 1003, where Vout is considered
at RF, while Vout is considered at baseband.
[0079] Since the impedance of the TIAs 1008 may vary with the
frequency of the signal applied thereon, it follows that the
frequency conversion of the RF signal by the passive mixer 1006
alters the impedance seen by the RF signal at the node 1001.
Specifically, the input impedance of the TIAs 1008 will each appear
translated by .+-..omega..sub.LO (e.g., the fundamental frequency
of the LO signals 1010) as seen by the RF signal at the node
1001.
[0080] The current-source downconversion path 1000 provides
frequency translation of the RF signal at the node 1001 by the sum
(.omega..sub.RF+.omega..sub.LO) and difference
(.omega..sub.RF-.omega..sub.LO or .omega..sub.LO-.omega..sub.RF) in
frequency between the LO signals 1010 and the input signal at the
node 1001. However, due to upper harmonics of the LO signals 1010,
it can be shown that the current-source downconversion path 1000
further provides frequency conversion of the RF signal by the sum
(.omega..sub.RF+(M-1)*.omega..sub.LO) and difference
(.omega..sub.RF-(M-1)*.omega..sub.LO or
(M-1)*.omega..sub.LO-.omega..sub.RF) in frequency between the
(M-1).sup.th harmonic of the LO signals 1010 and the RF signal,
where M defines the number of LO phases employed. Frequency
conversion of the RF signal by the sum and difference in frequency
between the (M-1).sup.th harmonic of the LO signals 1010 and the RF
signal is undesired and can create adverse effects, especially in
wideband applications.
[0081] The RF signal received at the node 1001 illustrates a range
of frequencies, .omega..sub.A-.omega..sub.B, that contain desired
information. The translated baseband impedance Z.sub.IN is centered
within a certain portion of the frequency band
.omega..sub.A-.omega..sub.B, such that desired information
contained at and around that frequency within the RF signal can be
retrieved. Specifically, the translated baseband impedance is
centered at .omega..sub.LO.
[0082] FIG. 11 illustrates a schematic diagram of an oversampling
downconversion path 1100 provided in the current-source
downconversion path 1000 illustrated in FIG. 10 in accordance with
one or more implementations. As noted above, the oversampling
downconversion path 1100 in its single-ended format, is composed of
the passive mixer 1006 (FIG. 10) and the TIAs 1008 (FIG. 10).
According to some implementations, the passive mixer 1006 is
composed of switching devices 1102.sub.1-1102.sub.M and the TIAs
1008 is composed of TIAs 1106.sub.1-1106.sub.M, where M represents
the number of LO signals employed.
[0083] The passive mixer 1006 uses the local oscillator (LO)
signals 1010 (FIG. 10) to down-convert or up-convert an RF signal
input 1112 (e.g., RF.sub.IN). For example, the passive mixer 1006
is configured to down-convert the RF signal input 1112 using
respective ones of the LO signals 1010. Specifically, during
down-conversion, harmonics of the LO signal cause RF input signals
at multiples of the LO frequency to directly interfere with each
other in a resulting baseband or intermediate frequency (IF)
signal.
[0084] The passive mixer 1006 provides multiple frequency
conversion branches. Each frequency conversion branch includes a
switching device coupled at its gate to a respective one of the LO
signals 1010, at its source to the RF signal input 1112, and at its
drain to a respective one of the TIAs 1008. For example, the first
frequency conversion branch illustrated in FIG. 11 includes a
switching device 1102.sub.1 represented as an n-type metal oxide
semiconductor (NMOS) device M.sub.0, which is coupled at its gate
to a LO signal LO.sub.0, at its source to the RF signal input 1112,
and at its drain to a baseband impedance 1106.sub.1. The switching
devices 1102.sub.1-1102.sub.M are not limited to NMOS devices and
can be implemented using any suitable switching device, including
p-type metal oxide semiconductors (PMOS).
[0085] The switching devices 1102.sub.1-1102.sub.M may be clocked
by the LO signals 1010 represented as M periodic non-overlapping
clocks, LO.sub.0, LO.sub.1, . . . , LO.sub.M-1, with a duty-cycle
of 1/M. Given that M represents the number of LO signals employed,
that number of LO signals may be determined according to a number
of harmonics present around a fundamental frequency (e.g.,
.omega..sub.LO). The switching devices 1102.sub.1-1102.sub.M are
switched ON and OFF at a rate substantially equal to the
fundamental frequency of the LO signals 1010. The toggle rate of
the switching devices 1102.sub.1-1102.sub.M effectively multiplies
the RF signal input 1112, coupled to the sources of the switching
devices 1102.sub.1-1102.sub.M, by respective ones of the LO signals
1010. This effective multiplication results in frequency conversion
of the RF signal input 1112 by the sum
(.omega..sub.RF+.omega..sub.LO) and difference
(.omega..sub.RF-.omega..sub.LO or .omega..sub.LO-.omega..sub.RF) in
frequency between the LO signals 1010 and the RF signal input 1112.
The frequency-converted RF signals are each provided to the
baseband impedances 1106.sub.1-1106.sub.M via the drains of the
switching devices 1102.sub.1-1102.sub.M.
[0086] According to some implementations, each of the TIAs
1106.sub.1-1106.sub.M includes a capacitor, a switched capacitor
filter, a switch capacitor resistance, and/or a complex impedance.
Note that the impedance of each of the TIAs 1106.sub.1-1106.sub.M
may be the same or different. Further note that each of the TIAs
1106.sub.1-1106.sub.M may be adjusted to adjust the properties of
the baseband amplification. While the switching devices
1102.sub.1-1102.sub.M are coupled to the baseband impedances
1106.sub.1-1106.sub.M, the switching devices 1102.sub.1-1102.sub.M
are connected to the RF signal input 1112 with a relatively large
output impedance. In one or more implementations, the output
impedance is 100 ohms (.OMEGA.) or greater.
[0087] The TIAs 1106.sub.1-1106.sub.M can include capacitors and/or
resistors that are respectively coupled between the drains of the
switching devices 1102.sub.1-1102.sub.M and the output of the
respective TIA. The input impedance of the TIAs
1106.sub.1-1106.sub.M are each substantially equivalent and their
impedances around DC are given by (ignoring parasitics):
Z.sub.BB=1/GM (8)
where GM is the transconductance of the amplifier used in each of
the TIAs 1106.sub.1-1106.sub.M. However, as the frequency of the
applied signal moves in either the positive or negative direction,
away from DC (e.g., 0 MHz), the impedance Z.sub.BB decreases. Thus,
the input impedance of the TIAs 1106.sub.1-1106.sub.M is
effectively centered at baseband 1110 (e.g., 0 MHz).
[0088] The .omega..sub.LO can be adjusted to any frequency portion
within a frequency band .omega..sub.A-.omega..sub.B. By adjusting
.omega..sub.LO to have a frequency substantially equal to the
center frequency of the desired channel, the translated baseband
impedance 1108 can effectively provide a filter to attenuate
frequency components of the RF signal outside the desired
channel.
[0089] However, the RF signal input 1112 is frequency converted by
the sum (.omega..sub.RF+(M-1)*.omega..sub.LO) and difference
(.omega..sub.RF-(M-1)*.omega..sub.LO or
(M-1)*.omega..sub.LO-.omega..sub.RF) in frequency between the
(M-1).sup.th harmonic of the LO signals 1010 and the RF signal
input 1112, where M is the number of LO phases employed. Therefore,
the frequency components of the RF signal input 1112 at and around
(M-1)*.omega..sub.LO and (M+1)*.omega..sub.LO will be
frequency-translated to .omega..sub.LO and will fall within a
pass-band of the translated baseband impedance 1108.
[0090] The overlap of these components at .omega..sub.LO can
detrimentally affect the desired information or channel centered at
.omega..sub.LO. That is, the overlapping of these frequency
components at .omega..sub.LO can prevent the desired information or
channel centered at .omega..sub.LO from being recovered. Because
these frequency components corresponding to LO harmonics may be
within the frequency band .omega..sub.A-.omega..sub.B, they ideally
receive little or no attenuation from components of the RF
receiver. In other words, because these frequency components
contain information that may be desired at any given point in time,
these components ideally do not receive any significant attenuation
from the components of the RF receiver front-end 104 (FIG. 1).
Therefore, because of the relative strength of these components,
the overlap of these components with the frequency components of
the desired channel centered at .omega..sub.LO can prevent the
desired information from being recovered.
[0091] FIG. 12 illustrates a waveform diagram 1200 of a gain
realized in the current-source downconversion path 1000 illustrated
in FIG. 10 in accordance with one or more implementations. The gain
(e.g., V.sub.OUT/V.sub.IN) realized at the output of the
transconductance cell 1002 (FIG. 10) provided in the current-source
downconversion path 1000 can be defined as:
.apprxeq. G M ( R SW + Z BB M ) ( 9 ) ##EQU00007##
where G.sub.M is the factor of proportionality between an input and
output voltage, M represents the number of LO phases employed,
R.sub.SW is the switch resistance of switching devices
1102.sub.1-402.sub.M, and Z.sub.BB is the input impedance of the
TIAs 1106.sub.1-406.sub.M.
[0092] Illustrated in FIG. 12, a voltage gain can be realized for
harmonics around the fundamental frequency, including unwanted
signals (or blocker signals) at integer multiples of a wanted
signal (e.g., first harmonic). The gain realized at the harmonics
of unwanted blocker signals decreases as the frequency (e.g., RF
frequency, LO frequency) increases. For example, the gain at the
eighth harmonic is given by G.sub.M (R.sub.SW) since the baseband
impedance for the eighth harmonic is substantially equivalent to
zero.
[0093] FIG. 13 illustrates a schematic diagram of a single-ended
receiver front-end 1300 with enhanced harmonic rejection in
accordance with one or more implementations. The receiver front-end
1300 connects to the antenna 102 (FIG. 1), and includes two
down-conversion paths 204 and 206 (FIG. 2), and a weighting and
recombination module 230 (FIG. 2). The receiver front-end 1300 also
includes transconductance cells 1302.sub.1-1302.sub.M, passive
mixers 216 and 1304.sub.1-1304.sub.M, and transimpedance amplifiers
602 and 604. The passive mixers 216 and 1304.sub.1-1304.sub.M each
include logic and/or circuitry that may provide for generation of
inter-modulation products resulting from mixing an RF signal input
205 (FIG. 2) with LO signals generated by the LOGEN 112 (FIG.
1).
[0094] The receiver front-end 1300 has a similar structure as the
receiver front-end 600 illustrated in FIG. 6 and operates in
substantially the same manner. However, the receiver front-end 1300
provides the transconductance cells 1302.sub.1-1302.sub.M and
passive mixers 1304.sub.1-1304.sub.M that collectively form an
enhanced harmonic rejection circuitry. The transconductance cells
1302.sub.1-1302.sub.M are configured to feed to the passive mixers
1304.sub.1-1304.sub.M. As such, the down-conversion path 206 can be
defined as an oversampling downconversion path composed of multiple
transconductance cells and multiple mixers to suppress replication
of the wanted pass-band at harmonics of the LO frequency.
[0095] The noise figure of the receiver front-end 1300 can be
realized as the following:
F = ( 1 + ( K .gamma. G M R SW ) ) 1 CG max 2 ( 10 )
##EQU00008##
where K is a constant dependent on a given implementation, .gamma.
is a constant related to the noise of a given technology, R.sub.SW
is the switch resistance of the passive mixers 206, and CG.sub.MIX
is the conversion gain of the passive mixers 206 used in the
receiver front-end 1300. This is an idealized and simplified
analysis; an actual circuit implementation may differ
significantly. According to some implementations, the receiver
front-end 1300 and the above-described elements are varied and are
not limited to the functions, structures, configurations,
implementations or examples provided.
[0096] According to some implementations, a receiver for wideband
applications includes an integrated circuit pin configured to
couple a radio frequency (RF) signal to a first signal path and a
second signal path. The receiver includes multiple transconductance
cells configured to produce weighted current signals proportional
to the RF signal along the first signal path and the second signal
path. Each of the transconductance cells are configured to provide
an effective transconductance of a first magnitude for frequency
components of the RF signal arising from a first harmonic and an
effective transconductance of a second magnitude less than the
first magnitude for frequency components of the RF signal arising
from harmonics at integer multiples of the first harmonic. The
receiver also includes multiple mixers configured to mix the
weighted current signals with local oscillator (LO) signals of
successive phase shifts to generate mixer signals. The receiver
also includes a feedback impedance coupled to at least a portion of
the transconductance cells along the first signal path to convert
the mixer signals into corresponding voltage signals.
[0097] In one or more implementations, the receiver has each of the
transconductance cells further configured with a transconductance
gain that shifts the weighted current signal with a defined phase
such that a constructive sum of the weighted current signals
provides a resulting current signal of a selected phase. The
receiver has the mixers and the transconductance cells employ
single-ended inputs. As such, the transconductance gain is shifted
to reject any integer multiple of a wanted signal corresponding to
the first harmonic. The receiver also has the mixers and the
transconductance cells employ differential inputs. As such, the
transconductance gain is shifted to reject odd integer multiples of
a wanted signal corresponding to the first harmonic. The receiver
has the transconductance cells along the second signal path receive
a voltage input equivalent to equation (6), shown above, where m
determines the effective transconductance of the transconductance
cell, and M represents the number of LO phases employed. The
receiver also includes multiple transimpedance amplifiers along the
second signal path, in which each of the transimpedance amplifiers
has at least a portion of the transconductance cells. The receiver
has each of the transconductance cells configured with a
transconductance weighting that causes the effective
transconductance of the transconductance cell to be non-zero when
excited by received signals arising from a harmonic within a
frequency band of interest but zero for received signals arising
from harmonics outside the frequency band of interest.
[0098] FIG. 14 illustrates a schematic diagram of a single-ended
current-source downconversion path 1400 provided in the receiver
front-end 1300 illustrated in FIG. 13 in accordance with one or
more implementations. A conventional passive-mixer-based
downconversion path provides voltage amplification of the wanted
signal around the LO fundamental as well as unwanted signals around
harmonics of the LO frequency. The current-source downconversion
path 1400 overcomes this previous limitation and provides a
practical implementation for eliminating unwanted RF
amplification.
[0099] The current-source downconversion path 1400 performs a
filtering technique that eliminates multiple harmonics at RF by
placing multiple stages of oversampling passive mixers at a node
within the receiver front-end 200 (FIG. 2). The current-source
downconversion path 1400 also employs multiple transconductance
cells (e.g., 1402.sub.1-1402.sub.M) coupled to respective frequency
conversion cells 1404.sub.1-1404.sub.M (or passive mixers) to
directly receive the inbound RF voltage signal (e.g., V.sub.IN) for
conversion to a current signal before mixing (or multiplying) the
current signal with multiple phased LO signals (e.g.,
LO<M-1:0>). In this regard, each of the mixed RF signals is
phase-shifted according to the respective LO signal, which may
include an unwanted blocker signal at frequency f.sub.b, and a
wanted signal at frequency f.sub.w, which may be .DELTA.f.sub.b
lower than f.sub.b, e.g., f.sub.w=f.sub.b-.DELTA.f.sub.b. Although
the bandwidth around these LO harmonics is very small, blocker
signals located at these specific harmonic frequencies potentially
saturate the receiver. Although the transconductance cells are
voltage-to-current converters, the outputs of the transconductance
cells can realize a voltage (e.g., V.sub.OUT) using impedances as
seen at inputs of the frequency conversion mixers. Therefore, the
subject technology proposes weighting each transconductance cell
such that the RF current signal corresponding to LO harmonics is
knocked out and the resulting RF current signal passes to
transimpedance amplifiers (TIAs) 1406.sub.1-1406.sub.M to produce a
voltage gain
( e . g . , V OUT V IN ) ##EQU00009##
around the wanted signal. According to some implementations, the
TIAs 1406.sub.1-1406.sub.M may be configured to reject unwanted
harmonics in accordance with the baseband harmonic rejection
technique discussed with respect to FIG. 7.
[0100] The current-source downconversion path 1400 includes M
transconductance cells 1402.sub.1-1402.sub.M configured to convert
an input RF voltage signal at Vs into multiple RF current signals.
In one or more implementations, each of the transconductance cells
1402.sub.1-1402.sub.M is configured with a transconductance gain
relative to one another. That is, each of the multiple
transconductance cells has a transconductance weighting equivalent
to equation (6), where X is an integer that represents each of the
transconductance cells (e.g., 1402.sub.1-1402.sub.M), M is a number
of LO phases employed, and k is an arbitrary constant that
determines the effective transconductance of the transconductance
cell.
[0101] The current-source downconversion path 1400 also includes
multiple frequency conversion cells 1404.sub.1-1404.sub.M. Each of
the frequency conversion cells 1404.sub.1-1404.sub.M is configured
to receive multiple non-overlapping local oscillator (LO) signals
(e.g., LO<M-1:0>). The multiple non-overlapping LO signals
have successive phase shifts substantially equal to 360/M degrees.
The multiple non-overlapping LO signals each have a fundamental
frequency equivalent to .omega..sub.LO. M may be determined based
on a number of harmonics around the fundamental frequency.
[0102] Each of the frequency conversion cells 1404.sub.1-1404.sub.M
is further configured to mix a respective RF current signal with
all LO signals to produce multiple downconverted current signals.
Given that the transconductance weighting produces weighted current
signals, the downconverted current signals corresponding to
harmonic multiples of the wanted signal (i.e. unwanted harmonic
blockers) are not passed to the TIAs 1406.sub.1-1406.sub.M. This is
because the weighting causes the effective transconductance to be
substantially large for harmonics corresponding to the wanted
signal and zero for harmonics corresponding to the unwanted
signals.
[0103] The TIAs 1406.sub.1-1406.sub.M are coupled to the multiple
frequency conversion cells and ground and are configured to provide
a baseband signal (e.g., in-phase and/or quadrature components of
the input RF voltage signal) when frequency translated by the
frequency conversion cells 1404.sub.1-1404.sub.M. In one or more
implementations, the number of TIAs 1406.sub.1-1406.sub.M is the
same and/or different from the number of transconductance cells
1402.sub.1-1402.sub.M and frequency conversion cells
1404.sub.1-1404.sub.M provided within the current-source
downconversion path 1400.
[0104] A translated baseband impedance at an input to the
transconductance cells 1402.sub.1-1402.sub.M is equivalent to the
input impedance of the TIAs 1404.sub.1-1404.sub.M translated in
frequency by .omega..sub.LO. In addition, the input impedance of
the TIAs 1404.sub.1-1404.sub.M is translated as seen by the input
RF voltage signal at an input to the transconductance cells
1402.sub.1-1402.sub.M.
[0105] In some implementations, a wideband receiver includes an
integrated circuit pin that is configured to couple an RF signal to
an RF signal path. The receiver also may include an oversampling
passive mixer coupled to the RF signal path and configured to
translate a baseband impedance to a higher frequency. The wideband
receiver includes multiple transconductance cells each having an
effective transconductance of a first magnitude for frequency
components of the RF signal within a frequency band of interest and
an effective transconductance of a second magnitude that is lesser
than the first magnitude for frequency components of the RF signal
outside the frequency band of interest.
[0106] The receiver also may include multiple frequency conversion
cells configured to receive a multiple non-overlapping local
oscillator (LO) signals. Each of the multiple frequency conversion
cells may be further configured to mix one of the multiple RF
current signals with the multiple non-overlapping LO signals. The
receiver also may include multiple baseband impedances coupled to
the multiple frequency conversion cells and ground.
[0107] Each of the multiple transconductance cells is configured
with a transconductance weighting that causes the effective
transconductance of the transconductance cell to be large when
excited by RF signals arising from a harmonic within the frequency
band of interest but zero for RF signals arising from harmonics
outside the frequency band of interest. The transconductance
weighting provides that only wanted harmonics excite a current from
the transconductance cell. The transconductance cells are
configured to amplify an RF signal from any harmonic of a
fundamental frequency. The transconductance cells are further
configured to amplify an RF signal from more than one harmonic
multiple of the fundamental frequency.
[0108] Referring to FIG. 14, the frequency conversion cells
1404.sub.1-1404.sub.M include metal-oxide semiconductor field
effect transistors (MOSFETs) (not shown). Specifically, the
frequency conversion cells 1404.sub.1-1404.sub.M are n-channel
MOSFETs (NMOS). However, the frequency conversion cells
1404.sub.1-1404.sub.M can be implemented using any suitable
frequency conversion device, including p-channel MOSFETs (PMOS),
bipolar junction transistors (BJTs) and junction gate field effect
transistors (JFETs). The MOSFETs provided in the frequency
conversion cells 1404.sub.1-1404.sub.M are operated substantially
in their linear mode when powered ON.
[0109] According to some implementations, the current-source
downconversion path 1400 is integrated in CMOS IC technology (or
others, e.g., Bipolar, BiCMOS, and SiGe) and applied in wireless
receiver systems including GSM, WCDMA, Bluetooth, and wireless LANs
(e.g., IEEE 1402.11).
[0110] In one or more implementations, a differential oversampling
passive mixer can be designed for use in wideband RF receivers
(e.g. UWB and TV receivers). The differential oversampling passive
mixer may be configured to receive a differential RF signal input
(e.g., RF.sub.IN+ and RF.sub.IN-) at a differential input pair via
a current source (e.g., a transconductance cell). The differential
oversampling passive mixer may include a passive mixer and a
baseband amplifier. The baseband amplifier may include multiple
transimpedance amplifiers and multiple capacitors. The passive
mixer may be configured to translate the baseband impedance (e.g.,
input impedance to the baseband amplifier) to a higher frequency.
Specifically, the passive mixer is configured to translate the
baseband impedance to a higher frequency substantially equal to the
fundamental frequency of LO signals received by the passive
mixer.
[0111] FIG. 15 illustrates a waveform diagram 1500 of a conversion
gain from the RF input to the output of a single TIA realized in
the current-source downconversion path 1400 illustrated in FIG. 14
in accordance with one or more implementations. The x-axis is RF
frequency normalized to the LO frequency. Note that signals
originating from around LO harmonics up to (M-1)th harmonics are
not amplified and, so, cannot saturate the receiver. By
appropriately weighting the transconductance cells
1402.sub.1-1402.sub.M, the transconductance weighting causes the RF
current signals to shift in phase. Thus, the RF current signals,
before reaching the transimpedance amplifiers
1406.sub.1-1406.sub.M, constructively sum to produce a resulting
current signal that corresponds to downconverted signals from
around the LO frequency (e.g., the wanted signal). In this regard,
the phase-shifted current signals with opposing phases in effect
cancel one another at the baseband impedance input. Therefore, no
voltage gain can be realized for the canceled-out current signals
(or unwanted signals). In some implementations, the input to the
transimpedance amplifiers 1406.sub.1-1406.sub.M defines a pass-band
around the wanted signal and minimal gain around the 7.sup.th and
9.sup.th LO harmonics (assuming M=8). That is, the first
feed-through terms may occur at (M-1)*.omega..sub.LO.
[0112] Given that there is no amplification of harmonic blockers at
the output of the current-source downconversion path 1400 with the
enhanced harmonic rejection circuit, no harmonic blockers up to the
(M-1).sup.th harmonic can saturate the receiver.
[0113] FIG. 16 illustrates a schematic diagram of a differential
receiver front-end 1600 with enhanced harmonic rejection in
accordance with one or more implementations. The receiver front-end
1600 connects to the antenna 102 (FIG. 1), and includes two
down-conversion paths 204 and 206 (FIG. 2), and a weighting and
recombination module 230 (FIG. 2). The receiver front-end 1600 also
includes transconductance cells 1602.sub.1-1602.sub.M, passive
mixers 216 and 1604.sub.1-1604.sub.M, and transimpedance amplifiers
602 and 604. The passive mixers 216 and 1604.sub.1-1604.sub.M each
include logic and/or circuitry that may provide for generation of
inter-modulation products resulting from mixing a differential RF
signal input 205 (FIG. 2) with LO signals generated by the LOGEN
112 (FIG. 1).
[0114] The receiver front-end 1600 has a similar structure as the
receiver front-end 1300 illustrated in FIG. 6 and operates in
substantially the same manner. However, the receiver front-end 1600
provides the transconductance cells 1602.sub.1-1602.sub.M and
passive mixers 1604.sub.1-1604.sub.M having differential inputs.
Similarly to FIG. 13, the transconductance cells
1602.sub.1-1602.sub.M are configured to feed to the passive mixers
1604.sub.1-1604.sub.M. As such, the down-conversion path 206 can be
defined as a differential oversampling passive mixer composed of
differential transconductance cells and differential mixers to
suppress replication of the wanted pass-band at harmonics of the LO
frequency. The weightings of the transconductance cells
1602.sub.1-1602.sub.M in this case are given by (7) as opposed to
(6).
[0115] FIG. 17 illustrates a schematic diagram of a differential
current-source downconversion path 1700 provided in the receiver
front-end 1600 illustrated in FIG. 16 in accordance with one or
more implementations. The differential current-source
downconversion path 1700 processes a differential RF signal (e.g.,
RF.sub.IN+ and RF.sub.IN-) received at a differential input pair
(e.g., node V.sub.IN). The differential current-source
downconversion path 1700 includes an oversampling passive mixer
composed of passive mixers 1704.sub.1-1704.sub.M and transimpedance
amplifiers (TIAs) 1706.sub.1-1706.sub.M.
[0116] The differential current-source downconversion path 1700
also employs multiple transconductance cells (e.g.,
1702.sub.1-1702.sub.M) to directly receive the inbound differential
RF voltage signal (e.g., V.sub.IN) for conversion to a differential
current signal before mixing (or multiplying) the current signal
with multiple phased LO signals (e.g., LO<M-1:0>). The
current-source downconversion path 1700 includes M transconductance
cells 1702.sub.1-1702.sub.M configured to convert the input
differential RF voltage signal at V.sub.IN into multiple RF current
signals.
[0117] In one or more implementations, each of the transconductance
cells 1702.sub.1-1702.sub.M is configured with a transconductance
gain relative to one another. That is, each of the transconductance
cells 1702.sub.1-1702.sub.M produces a transconductance gain. By
appropriately weighting the transconductance cells
1702.sub.1-1702.sub.M, the weighting causes the RF current signals
to shift in phase. Thus, the RF current signals, before reaching
the TIAs 1706.sub.1-1706.sub.M, constructively sum to produce a
resulting current signal that corresponds to downconverted signals
from around the LO frequency (i.e. the wanted signal). In this
regard, the phase-shifted current signals with opposing phases in
effect cancel one another at the baseband impedance input.
[0118] FIG. 18 illustrates a schematic diagram of a differential
oversampling passive mixer 1800 provided in the current-source
downconversion path 1700 illustrated in FIG. 17 in accordance with
one or more implementations. According to some implementations, the
differential oversampling passive mixer 1800 includes switching
devices M1-M8, where M is equal to 8. The variable M can be defined
as any positive integer. The switching devices M1-M8 are
metal-oxide semiconductor field effect transistors (MOSFETs).
Specifically, the switching devices M1-M8 are n-channel MOSFETs
(NMOS). However, as will be appreciated by one of ordinary skill in
the art, the switching devices M1-M8 can be implemented using any
suitable switching device, including p-channel MOSFETs (PMOS),
bipolar junction transistors (BJTs) and junction gate field effect
transistors (JFETs). The switching devices M1-M8 may be operated
substantially in their linear mode when ON.
[0119] In operation, the first switching device, M1, of the passive
mixers 1704.sub.1-1704.sub.M (FIG. 17) receives a differential LO
signal (LO<0>). The LO signal has a frequency of
.omega..sub.LO and a duty-cycle substantially equal to 12.5%. The
gates of switching devices M1, M3 up to M-1 are coupled to the
positive LO signal (LO.sub.+), and the gates of switching devices
M2, M4 up to
( M 2 - 1 ) ##EQU00010##
are coupled to the negative LO signal (LO.sub.-). Because the two
LO signals (LO.sub.+ and LO.sub.-) are substantially 180-degrees
out of phase, the switching device pair M1 and M2 are switched ON
and OFF at non-overlapping intervals at the frequency of the LO
signal (.omega..sub.LO). The non-overlapping switching at a
frequency of .omega..sub.LO effectively multiplies the positive RF
signal (RF.sub.IN+), coupled to the source of switching device M1,
and the negative RF signal (RF.sub.IN-), coupled to the source of
switching device M2, by .+-.1. This effective multiplication
results in frequency conversion of the differential RF signal by
the sum (.omega..sub.RF+.omega..sub.LO) and difference
(.omega..sub.RF-.omega..sub.LO or .omega..sub.LO-.omega..sub.RF) in
frequency between the LO signal (LO<0>) and the differential
RF signal. The frequency-converted component of the RF signal is
provided differentially to the TIAs 1706.sub.1-1706.sub.M (FIG.
17). As noted above, the TIAs 1706.sub.1-1706.sub.M may include
capacitors having impedances that are given by (ignoring
parasitics) equation (9).
[0120] It follows that the frequency conversion of the differential
RF signal by the passive mixers 1704.sub.1-1704.sub.M alters the
baseband impedance seen by the differential RF signal at the
differential input pair. Specifically, the impedance of the
capacitors will each appear translated by .+-..omega..sub.LO as
seen by the differential RF signal at the differential input pair,
thus becoming a current-source downconversion path presented at the
differential input pair.
[0121] One or more implementations are performed by one or more
integrated circuits, such as application specific integrated
circuits (ASICs) or field programmable gate arrays (FPGAs). In one
or more implementations, such integrated circuits execute
instructions that are stored on the circuit itself.
[0122] Those of skill in the art would appreciate that the various
illustrative blocks, elements, components, and methods described
herein may be implemented as electronic hardware. Various
illustrative blocks, elements, components, and methods have been
described above generally in terms of their functionality. Whether
such functionality is implemented as hardware depends upon the
particular application and design constraints imposed on the
overall system. Skilled artisans may implement the described
functionality in varying ways for each particular application.
Various components and blocks may be arranged differently (e.g.,
arranged in a different order, or partitioned in a different way)
all without departing from the scope of the subject technology.
[0123] It is understood that any specific order or hierarchy of
blocks in the processes disclosed is an illustration of example
approaches. Based upon design preferences, it is understood that
the specific order or hierarchy of blocks in the processes may be
rearranged, or that all illustrated blocks be performed. Any of the
blocks may be performed simultaneously. In one or more
implementations, multitasking and parallel processing may be
advantageous. Moreover, the separation of various system components
in the implementations described above should not be understood as
requiring such separation in all implementations, and it should be
understood that the described program components and systems can
generally be integrated together in a single software product or
packaged into multiple software products.
[0124] As used in this specification and any claims of this
application, the terms "receiver", "amplifier", "transconductance
cell," and "mixer" all refer to electronic or other technological
devices. These terms exclude people or groups of people.
[0125] The predicate words "configured to" and "operable to" do not
imply any particular tangible or intangible modification of a
subject, but, rather, are intended to be used interchangeably. In
one or more implementations, a receiver configured to receive and
process an operation or a component may also mean the receiver
being operable to receive and process the operation.
[0126] A phrase such as "an aspect" does not imply that such aspect
is essential to the subject technology or that such aspect applies
to all configurations of the subject technology. A disclosure
relating to an aspect may apply to all configurations, or one or
more configurations. An aspect may provide one or more examples of
the disclosure. A phrase such as an "aspect" may refer to one or
more aspects and vice versa. A phrase such as an "implementation"
does not imply that such implementation is essential to the subject
technology or that such implementation applies to all
configurations of the subject technology. A disclosure relating to
an implementation may apply to all implementations, or one or more
implementations. An implementation may provide one or more examples
of the disclosure. A phrase such an "implementation" may refer to
one or more implementations and vice versa. A phrase such as a
"configuration" does not imply that such configuration is essential
to the subject technology or that such configuration applies to all
configurations of the subject technology. A disclosure relating to
a configuration may apply to all configurations, or one or more
configurations. A configuration may provide one or more examples of
the disclosure. A phrase such as a "configuration" may refer to one
or more configurations and vice versa.
[0127] Any implementation described herein as an "example" is not
necessarily to be construed as preferred or advantageous over other
implementations. Furthermore, to the extent that the term
"include." "have," or the like is used in the description or the
claims, such term is intended to be inclusive in a manner similar
to the term "comprise" as "comprise" is interpreted when employed
as a transitional word in a claim.
[0128] All structural and functional equivalents to the elements of
the various aspects described throughout this disclosure that are
known or later come to be known to those of ordinary skill in the
art are expressly incorporated herein by reference and are intended
to be encompassed by the claims. Moreover, nothing disclosed herein
is intended to be dedicated to the public regardless of whether
such disclosure is explicitly recited in the claims. No claim
element is to be construed under the provisions of 35 U.S.C.
.sctn.112, sixth paragraph, unless the element is expressly recited
using the phrase "means for" or, in the case of a method claim, the
element is recited using the phrase "step for."
[0129] The previous description is provided to enable any person
skilled in the art to practice the various aspects described
herein. Various modifications to these aspects will be readily
apparent to those skilled in the art, and the generic principles
defined herein may be applied to other aspects. Thus, the claims
are not intended to be limited to the aspects shown herein, but are
to be accorded the full scope consistent with the language claims,
wherein reference to an element in the singular is not intended to
mean "one and only one" unless specifically so stated, but rather
"one or more." Unless specifically stated otherwise, the term
"some" refers to one or more. Pronouns in the masculine (e.g., his)
include the feminine and neuter gender (e.g., her and its) and vice
versa. Headings and subheadings, if any, are used for convenience
only and do not limit the subject disclosure.
* * * * *