U.S. patent application number 14/561142 was filed with the patent office on 2015-06-11 for method and apparatus for reducing noise in a coded aperture radar.
The applicant listed for this patent is Jonathan J. LYNCH. Invention is credited to Jonathan J. LYNCH.
Application Number | 20150160331 14/561142 |
Document ID | / |
Family ID | 53270945 |
Filed Date | 2015-06-11 |
United States Patent
Application |
20150160331 |
Kind Code |
A1 |
LYNCH; Jonathan J. |
June 11, 2015 |
Method and Apparatus for Reducing Noise in a Coded Aperture
Radar
Abstract
A method and apparatus for reducing noise in a coded aperture
radar system, the coded aperture radar system transmitting signals
which occur in sweeps, with K sweeps utilized to cover field of
view of the coded aperture radar system and Q frequency shifts or
steps occurring each sweep thereof. An array of N antenna elements
is provided, the array of antenna elements each having an
associated two state modulator coupled therewith. Energy received
at the array is modulated according to a sequence of multibit
codes, the number of codes in the sequence of codes comprising at
least K times Q times N, thereby reducing noise in the coded
aperture radar system compared to a coded aperture radar system
radar system having fewer than K times Q times N codes in its
sequence of multibit codes.
Inventors: |
LYNCH; Jonathan J.; (Oxnard,
CA) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
LYNCH; Jonathan J. |
Oxnard |
CA |
US |
|
|
Family ID: |
53270945 |
Appl. No.: |
14/561142 |
Filed: |
December 4, 2014 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
61912990 |
Dec 6, 2013 |
|
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|
Current U.S.
Class: |
342/105 ;
342/128; 342/200 |
Current CPC
Class: |
G01S 2007/356 20130101;
G01S 13/882 20130101; G01S 13/931 20130101; G01S 13/536 20130101;
G01S 7/02 20130101; G01S 13/935 20200101; G01S 7/2813 20130101;
G01S 13/4418 20130101; G01S 13/343 20130101; G01S 13/4463 20130101;
G01S 13/02 20130101 |
International
Class: |
G01S 7/28 20060101
G01S007/28; G01S 13/34 20060101 G01S013/34; G01S 13/58 20060101
G01S013/58 |
Claims
1. A radar system for estimating range, range rate (velocity) and
bearing angles of one or more targets or objects reflecting at
least one transmitted RF signal, the radar system comprising: a
transmitter for transmitting the at least one transmitted RF
signal, the transmitted signal changing in frequency during each
sweep made by the radar system, the transmitter making K sweeps per
acquisition and each sweep having Q frequency shifts per sweep; an
array of N antenna elements for receiving the at least one RF
signal; an array of single bit modulators, each single bit
modulator in said array of single bit modulators being coupled with
a corresponding antenna element or with a corresponding subgroup of
said antenna elements in said array of antenna elements for
modulating signals from the corresponding antenna elements
according to a multibit code, wherein the number unique codes from
which said multibit code is selected is equal to N, with N unique
codes being repeated QK times during each acquisition; a mixer; a
summation network for applying a summation of signals from the
array of single bit modulators to said mixer, the mixer outputting
in-phase and quadrature output signals; one or more analog to
digital convertors for detecting and converting in-phase and
quadrature output signals from the mixer to corresponding digital
signals; means for applying S signal masks to the corresponding
digital signals to thereby generate S different masked versions of
the corresponding digital signals; means for summing said S
different masked versions of the corresponding digital signals to
produced summed versions thereof; and means for performing a two
dimensional FFT processing of the summed versions of the S
different masked versions of the corresponding digital signals, to
estimate the range, range rate (velocity) and bearing angles of
said one or more targets or objects.
2. The apparatus of claim 1 wherein the array of single bit
modulators comprise an array of two state phase shifters.
3. The apparatus of claim 2 wherein the two state phase shifters
are 0.degree./180.degree. phase shifters.
4. The apparatus of claim 1 wherein the means for applying S signal
masks to the corresponding digital signals to thereby generate S
different masked versions of the corresponding digital signals is a
multiplier.
5. The apparatus of claim 4 wherein the means for analyzing the
corresponding digital signals further includes thresholding means
applied to data generated by the two dimensional FFT processing to
discard data which does not exceed a selected threshold.
6. The apparatus of claim 1 wherein the array of single bit
modulators comprises an array of N single bit modulators and
wherein each antenna element in the array of N antenna elements is
coupled to a separate single bit modulator in the array of N single
bit modulators.
7. The apparatus of claim 1 wherein the transmitter transmits N
codes during each frequency shift of the at least one transmitted
RF signal.
8. A method for determining the range of one or more scattering
objects reflecting radar signals which occur in sweeps, with K
sweeps utilized to cover an acquisition of a field of view and Q
frequency shifts occurring during at least one sweep, the method
comprising: utilizing an array of N antenna elements, the array of
antenna elements each antenna element in said array having an
associated two state modulator; coding energy received at said
array according to a sequence of multibit codes, the number of
unique codes in said sequence of codes is equal to N, with N unique
codes being repeated QK times during each acquisition; thereby
allowing the determination of range through digital computation
after the scattered signals have been received.
9. The method of claim 8 wherein transmitted and/or received energy
is 0/180 degree phase encoded with respect to each element of the
array of antenna elements according to the sequence of said
multibit codes.
10. The method of claim 8 wherein the associated two state
modulators form an array of associated two state modulators of size
N so that each of the antenna elements in the array of N antenna
elements is individually coupled to a single associated two state
modulator in the array N associated two state modulators.
11. The method of claim 8 wherein the digital computation includes
Fast Fourier Transform (FFT) processing of the energy received at
said array according to said sequence of multibit codes.
12. A method for reducing noise in a coded aperture radar system,
the radar system transmitting signals which occur in sweeps, with K
sweeps utilized to cover an acquisition of a field of view and Q
frequency shifts occurring during at least one sweep, the method
comprising: utilizing an array of N antenna elements, the array of
antenna elements each having an associated two state modulator
coupled therewith; coding energy received at said array according
to a sequence of multibit codes, the number of unique codes in said
sequence of codes is equal to N', with N' unique codes being
repeated QK times during each acquisition; thereby reducing noise
in said radar system compared to a coded aperture radar system
radar system having fewer than N' codes in its sequence of multibit
codes.
13. The method of claim 12 wherein transmitted and/or received
energy is 0/180 degree phase encoded with respect to each element
of the array of antenna elements according to the sequence of said
multibit codes.
14. The method of claim 12 wherein the associated two state
modulators form an array of associated two state modulators of size
N so that each of the antenna elements in the array of N antenna
elements is individually coupled to a single associated two state
modulator in the array N associated two state modulators.
15. The method of claim 12 wherein the digital computation includes
Fast Fourier Transform (FFT) processing of the energy received at
said array according to said sequence of multibit codes.
16. A method of improving the sensitivity and dynamic range of a
radar system performing radar sweeps during an acquisition of a
field of view, each sweep having a different transmit frequency
associated therewith, the method comprising using a set of N
aperture codes to control N single bit modulators during each
frequency step of the radar system, the N single bit modulators
each being coupled to a single antenna element of an array of N
antenna elements of the radar system.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application claims priority to and the benefit of U.S.
Provisional Patent Application Ser. No. 61/912,990, filed Dec. 6,
2013 and entitled "A Method and Apparatus for Processing Coded
Aperture Radar Signals" and U.S. patent application Ser. No.
14/561,111 filed on the same date as this application and entitled
"A Method and Apparatus for Processing Coded Aperture Radar
Signals" (Attorney Docket 629073). The disclosure of that U.S.
Provisional Patent Application Ser. No. 61/912,990 and the U.S.
patent application Ser. No. 14/561,111 identified above are hereby
incorporated herein by this reference in their entirety.
[0002] This application is related to U.S. patent application Ser.
No. 13/490,607 filed Jun. 7, 2012 and entitled "Coded Aperture Beam
Analysis Method and Apparatus", the disclosure of which is hereby
incorporated herein by reference.
[0003] This application is also related to U.S. patent application
Ser. No. 13/725,621 filed Dec. 21, 2012 and entitled "Coded
Aperture Beam Analysis Method and Apparatus", the disclosure of
which is hereby incorporated herein by reference.
STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT
[0004] None.
TECHNICAL FIELD
[0005] This invention teaches a method of coding for use with Coded
Aperture Radar (CAR) that reduces ambiguity (sometimes called
"multiplicative noise") that is typically present as a result of
aperture coding. The novelty of this invention is that it minimizes
ambiguity in the radar signals while providing complete information
of objects' range, radial velocity, and angular location within a
prescribed field of view. The invention retains all of the
advantages of CAR while improving the radar performance through
reduction of ambiguity.
BACKGROUND
[0006] CAR is different than a conventional phased array radar.
Sensitivity is often limited in CAR compared to a conventional
phased array radar because energy is typically collected in a
relatively short period of time, consisting of a single radar
range/Doppler acquisition when using CAR. Because of this the total
received energy is lower than, say, a conventional phased array
radar that forms a directive beam but then collects energy over a
longer period of time by sequentially moving the beam to all beam
directions within the field of view. For example, if N beams fill
the field of view, then a conventional phased array radar requires
one acquisition period at each beam position, so the total energy
will be N times greater than for CAR. Two type of CAR (Type I and
Type II) are identified in U.S. Provisional Patent Application Ser.
No. 61/912,990, filed Dec. 6, 2013 and entitled "A Method and
Apparatus for Processing Coded Aperture Radar Signals" and U.S.
patent application Ser. No. 14/561,111 filed on the same date as
this application and entitled "A Method and Apparatus for
Processing Coded Aperture Radar Signals" (Attorney Docket 629073).
This disclosure described in greater detail Type II CAR and in
particular how it reduces ambiguity (from multiplicative noise)
compared to Type I CAR.
[0007] If one extends the Type I CAR acquisition period a factor of
N then the sensitivity of CAR will be the same as for a
conventional phased array (other things being equal). However, in
that case the number of Doppler bins increases a factor of N,
increasing the computational overhead of the Type I CAR system.
[0008] Given the typical lower sensitivity of a Type I CAR coded
radar system, it can effectively be employed where a short range
radar can be used, for example, as an automotive radar, for
rotorcraft landing in degraded visual environments, proximity
sensors, aircraft altimeters (used using landing), for aircraft
maneuvering on a taxiway or on an aircraft carrier, etc. However,
as noted above, the sensitivity of a Type I CAR coded radar system
can be increased if needed.
[0009] Type I CAR provides a method and apparatus for acquiring
information about the 3D location and radial velocity of a
continuum of scatterers within a relatively short acquisition
period compared to conventional radar as mentioned above. The Type
I CAR technique has been previously made the subject of patent
applications (see the US patent applications identified above),
along with suggestions for coding the single bit phase shifters
located at each aperture transmitting and/or receiving element.
However, the previously disclosed techniques produce a uniformly
distributed ambiguity, sometimes called residual ambiguity or
multiplicative noise, which reduces the sensitivity and dynamic
range of the radar.
[0010] Type II CAR described herein addresses this residual
ambiguity or multiplicative noise associated with Type I CAR.
BRIEF DESCRIPTION OF THE INVENTION
[0011] This invention reduces the distributed ambiguity noted above
by including additional measurements over the previous CAR coding
schemes described in the two above-identified US patent
applications. The additional measurements provide a sufficient set
for inverting the aperture code and determining the element
signals. Once the element signals are determined, one may use
linear combinations of the signals to computationally define
effective beams in any desired direction and with specific sidelobe
characteristics. The reduction of ambiguity is important for radar
sensors so that maximum sensitivity and dynamic range may be
achieved.
[0012] In one aspect the present invention provides a radar system
for determining range, range rate (velocity) and bearing angles of
a target reflecting at least one transmitted RF signal, the radar
system comprising: a transmitter for transmitting the at least one
transmitted RF signal, the transmitted signal changing in frequency
during each sweep made by the radar system, the transmitter making
K sweeps and each sweep having Q frequency shifts per sweep; an
array of N antenna elements for receiving the at least one RF
signal; an array of single bit modulators, each single bit
modulator in said array of single bit modulators being coupled with
a corresponding antenna element or with a corresponding subgroup of
said antenna elements in said array of antenna elements for
modulating signals from the corresponding antenna elements
according to a multibit code, wherein the number unique codes from
which said multibit code is selected is equal to at least N; a
mixer; a summation network for applying a summation of signals from
the array of single bit modulators to said mixer, the mixer
converting the summation of signals either to baseband or to
intermediate frequency analog signals; an analog to digital
convertor for detecting and converting the baseband or intermediate
frequency analog signals from the mixer to corresponding digital
signals; and means for analyzing the corresponding digital signals
to determine the direction of arrival of the at least at least one
RF signal from the at least one emitting source of the at least one
RF signal.
[0013] In another aspect the present invention provides a method
for determining the range (and preferably also the range rate and
bearing angles) of one or more scattering objects reflecting radar
signals which occur in sweeps, with K sweeps and Q frequency shifts
occurring during at least one sweep, the method comprising:
utilizing an array of N antenna elements, the array of antenna
elements each antenna element in said array having an associated
two state modulator; coded energy received at said array according
to a sequence of multibit codes, the number of codes in said
sequence of codes comprising at least N per frequency step; to
thereby allow the determination of range through digital
computation after the scattered signals have been received.
[0014] In still yet another aspect the present invention provides a
method for reducing multiplicative noise in a coded aperture radar
system, the radar system transmitting signals which occur in
sweeps, with K sweeps utilized to cover field of view and Q
frequency shifts occurring during at least one sweep, the method
comprising: utilizing an array of N antenna elements, the array of
antenna elements each having an associated two state modulator
coupled therewith; coding energy received at said array according
to a sequence of multibit codes, the number of codes in said
sequence of codes comprising at least N times K times Q; to thereby
reduce multiplicative noise in said radar system compared to a
coded aperture radar system radar system having fewer than at least
N times K times Q codes in its sequence of multibit codes.
[0015] In yet another aspect the present invention provides a
method of improving the sensitivity and dynamic range of a radar
system performing radar sweeps of a field of view, each sweep
having a different transmit frequency associated therewith, the
method comprising using a set of N aperture codes to control N
single bit modulators during each frequency step of the radar
system, the N single bit modulators each being coupled to a single
antenna element of an array of N antenna elements of the radar
system.
BRIEF DESCRIPTION OF THE DRAWINGS
[0016] FIG. 1 is a block diagram of an embodiment of the invention
and schematically depicts a homodyne radar with CAR coding on
receive only.
[0017] FIG. 2 shows that the instantaneous transmitted frequency
preferably consists of a series of equal frequency steps that are
repeated with N codes per step, Q steps per sweep, K sweeps per
acquisition.
[0018] FIG. 3 is a block diagram showing the CAR processing
technique described herein where a one to one relationship between
antenna elements and 1-bit phase shifters is utilized.
[0019] FIG. 3a shows an alternative embodiment of the CAR
processing where multiple antenna elements (in a subarray of
antenna elements) share a single 1-but phase shifter.
[0020] FIG. 3b shows another alternative embodiment of the CAR
processing technique described herein where a one to one
relationship between antenna elements and 1-bit phase shifters is
maintained, but where outputs of the antenna elements are the
summed down to more than one port which ports are digitized
independently by more than one A/D convertor.
[0021] FIGS. 4a-4d relate to a simulation of the CAR radar
disclosed herein with a common set of codes used at each frequency
step, FIG. 4a being a graph of the 2D range/azimuth cut with the
reference velocity equal to 10 m/sec. The graphs of FIGS. 4b-4d are
1D cuts of range, azimuth and velocity, respectively.
[0022] FIGS. 5a-5d relate to a simulation of CAR using a coding
technique with a single different code for each frequency step.
FIG. 5a is a graph of the 2D range/azimuth cut with the reference
velocity equals to 10 m/sec, while the graphs of FIGS. 5b-5d are 1D
cuts of range, azimuth and velocity, respectively. The
multiplicative noise is evident throughout range, velocity, and
bearing angle spaces.
[0023] FIGS. 6a-6d relate to another simulation of CAR using a
coding technique with a different code for each frequency sweep.
FIG. 6a is a graph of the 2D range/azimuth cut with the reference
velocity equals to 10 m/sec, while the graphs of FIGS. 6b-6d are 1D
cuts of range, azimuth and velocity, respectively. The
multiplicative noise is now confined to the velocity and angle
spaces and is absent from the range space.
DETAILED DESCRIPTION
[0024] The technique described herein utilizes additional
measurements made in a time period that is constrained by range and
velocity discretization. As a result this technique requires a
faster analog to digital converter (ADC) than the techniques used
in the US patent applications referenced above. But an advantage
compared to the US patent applications referenced above is that the
residual ambiguity is lower than the previous applications.
[0025] FIG. 1 shows a block diagram of a CAR coded radar, with CAR
coding being employed, for simplicity's sake, only on the receive
portion of the radar. The possibility of using CAR on the
transmitted signal as well is discussed towards the end of this
patent. The radar is much simpler to design (and is much less
complicated to implement, and thus less computationally expensive)
if the CAR coding disclosed herein is employed at the only receiver
side of the radar. As such, the receive only embodiments disclosed
herein are preferred for a low cost, close range radar systems such
as might be used in automobiles, for example, or other such
applications where having a relatively short target acquisition
time period compared to conventional radar can be extremely
important.
[0026] A signal is transmitted over a field of view by a radar
transmitter 11 and the scattered energy 8 (from one or more
targets) is received by an array of receiving elements 10 (the
array 10, in practice, is preferably a two dimensional array, but a
one dimensional array is more convenient for analysis and
simulation and may be used in practice). Each of the received
signals is phase shifted (modulated) by either zero or 180 degrees
by an array of 1-bit phase shifters 12. Preferably, there is a
one-to-one relationship between antenna elements 10 and phase
shifters 12. But it is possible, in some embodiments (see FIG. 3a),
to have several antenna elements 10 grouped together (summed in a
subarray 10.sub.SA) that are then coupled with a single phase
shifter 12. The drawback to this approach is the appearance of
grating lobes in the element patterns because the subarrays would
be spaced greater than .lamda./2 apart from one another (where
.lamda. is the wavelength of the nominal frequency of the radar
system). Grating lobes are undesirable because they reduce gain and
produce spurious radiation in unwanted directions. The advantage is
lower cost. Only a few antenna elements 10 and phase shifters 12
are depicted in FIGS. 3 and 3a for ease of illustrational and
explanation, it being understood that actual embodiments of the
inventions described herein would likely employ many more such
antenna elements 10 and phase shifters 12.
[0027] The received scattered signals are phase shifted (or not)
depending on the state of a control word, a bit of which is applied
to each binary phase shifter 10 (thus controlling whether it
performs a 180 degree phase shift (or not) on the received
scattered signals). The control words are preferably generated
pseudo-randomly. The phase shifted (i.e., aperture coded) signals
output from the 1-bit phase shifters 12 are then summed at a summer
14. The output of the summer 14 is an output port. The signal may
be amplified if needed, by an amplifier 15 and then down-converted
at a mixer 16 and digitized by an A/D converter 18. The embodiments
of FIGS. 3 and 3a have a single output port, but it is also
possible to sum down to more than one port (e.g., 2, 4, etc.) and
digitize each port independently using multiple A/D convertors 18
as depicted by the embodiment of FIG. 3b. The advantage to this is
an increase in the collected energy, improving sensitivity. The
disadvantage is increased cost since the number of ASICs (discussed
below) doubles if the number of ports double, quadruples if the
number of ports quadruple, etc.
[0028] The radar transmitter 11 which is in the embodiments of
FIGS. 3, 3a and 3b (with CAR coding on the receiver side only) may
be a standard FMCW radar architecture 11 that is well known to
those skilled in the art. In automotive or other land vehicle
applications, the radar will likely use a "homodyne" downconversion
(swept LO) to baseband, which is typically from DC to a few MHz. In
block 11, element 11a is a voltage controlled oscillator (VCO) that
outputs, in one particular embodiment, a 76.5 GHz frequency
modulated signal (other frequencies may be used). Element 11b is a
coupler that splits the VCO output signal and sends preferably a
portion to the local oscillator port of the mixer 16. The remaining
portion of the VCO power is preferably sent to a power amplifier
11c, and then, after leaving the transmitter block 11, to a
transmitting antenna 11d.
[0029] There are many possible transmitted radar signals, and one
especially convenient one is a repetitive series of equal frequency
steps, with preferably N codes per frequency step, as shown in FIG.
2. The number of codes per step (N) in the transmitted signal is
preferably the same as the number (N) of antenna elements 10. The N
transmitted codes (See FIG. 2) are preferably selected in the same
manner (for example, pseudo-randomly) as the one-bit receive codes
applied to the binary or one-bit phase shifters 12 but the
resulting two matrices should not be chosen to be the same.
Furthermore, since the transmit-receive code combination is just
the product of the separate transmit and receive codes (it is well
known to those skilled in the art that radar response is
proportional to the product of the transmit modulation and receive
modulation), the overall coding matrix is the element by element
product of the transmit code matrix and the receive code matrix.
One should therefore ensure that the overall coding matrix contains
linearly independent columns.
[0030] In the prior patent applications noted above coding is
implemented using pseudo-random phase shifter states with either
one code (set of states) per sweep or one code per frequency step.
One could alternatively use frequency hopping, sinusoidal
modulation, etc. for the transmitted signals, but the use of linear
FM (FMCW) allows the disclosed embodiments to use Fast Fourier
Transform (FFT) processing in the receiver which is depicted in
greater detail by FIGS. 3 and 3b. FIGS. 3 and 3b repeat much of
what is shown in FIG. 1, but shows the FFT processing (preferably
performed by one or more CPUs or Digital Signal Processors (DSPs))
done by the FFT functions 24.sub.0-24.sub.S-1 depicted for the
embodiments of FIGS. 3 and 3b. The CPU of FIG. 1 could conceivably
do all of the required FFT processing, but allocating the FFT
processing to one or more Application Specific ICs (ASICs), each of
which would function as a FFT CPU or FFT DSP, would computationally
make more sense. These ASICs should preferably also perform the
multiplication needed for the S signal mask functions
20.sub.0-20.sub.S-1 which are described in greater detail below.
While there could be a one-to-one relationship between the number
of ASICs utilized and the number of FFT functions
24.sub.0-24.sub.S-1 and the number of S signal mask functions
20.sub.0-20.sub.S-1 depicted for the embodiments of FIGS. 3 and 3b,
it is preferable to implement all S digital channels in a single
ASIC as that would likely lead to a lower cost of implementation
than would the use of multiple ASICs. So for the embodiment of FIG.
3, the total number of ASICs is preferably equal to just one.
[0031] Referring again to FIG. 3, in the receiver, after the down
converted signal is digitized at the A/D converter 18, the signal
is preferably split into S parallel channels, where S is the number
of beams that will be computationally defined in parallel and also
preferably where S=N is the number of antenna elements. Since this
is a digital beamforming approach, the reference here to "beams" is
not to real physical high-gain RF beams, but rather to "effective"
beams that are formed (or defined) digitally after data collection.
The number of channels is preferably equal to the number of
effective beams that are digitally formed at the same time (in
parallel), by, for example, the CPU of FIG. 1 or more preferably by
the single ASIC mentioned above.
[0032] The value of S (the number of channels) is typically equal
to the number of antenna elements 10 (and the number of phase
shifters 12), but one can use a smaller value of S if latency is
not an issue--in which case the FFT processing of the "effective"
beams may be done sequentially as opposed to all as the same time
as suggested by the embodiments of FIGS. 3 and 3a. So if the number
of channels were S=N/2 (where S is the number of beams that will be
computationally defined in parallel and N is the number of antenna
elements), then the processing would take twice as long to process
all beams. One can also use a smaller value of N if only a portion
of the field of view (FOV) is to be initially examined. In this
latter case the signal mask functions 20.sub.0-20.sub.S-1 would
very likely vary over time to focus attention on the portion of the
FOV of initial interest.
[0033] Nevertheless, while a lower value of S (say N/2) can be
utilized for the number of channels (and the number of ASICs), the
embodiment of FIG. 3 is more robust in that all N beams can be
computationally defined in parallel (i.e., at the same time). And
the embodiment of FIG. 3b is still even more robust, but the number
of channels then increases to say 2N, while still processing N
beams in parallel.
[0034] The S signal mask functions 20.sub.0-20.sub.S-1 are each
simply the multiplication of each data sample from the A/D
converter 18 with a value stored in memory of FIG. 1 for each
effective beam position. The data samples are complex (with real
and imaginary parts) because the mixer has in-phase and quadrature
outputs that are separately digitized by A/D converters. It is
common to depict this process with a single mixer and A/D
converter, as done in the figures here, with the understanding that
the signals are complex. There are a number of ways one can
determine the values stored in memory (which are typically also
complex numbers) of FIG. 1 which are used in the multiplication
performed in the mask functions 20.sub.0-20.sub.S-1 and one
(preferred) method for determining these numbers will be explained
below. These masks determine the angular beam pattern performance
of the array, so the masks influence main beam direction, beam
width, sidelobes, etc. They generally do not vary in time unless
one desires to modify the beam pattern of the array over time. This
is possible (and easily done) since all the beamforming is
preferably done digitally.
[0035] During each acquisition there is a string of stored data
samples whose length N' is preferably equal to the number of
antenna elements N. The N' stored data samples (called a signal
mask) multiply N' successive acquired signal samples and the
results are summed to form one signal mask output value that
corresponds to the N' successive input samples. This process is
repeated for the next group of N' input samples to form the next
signal mask output value. As a result, if there are M total samples
in an acquisition, there will be M/N' output values for each signal
mask. The resulting M/N' "masked" samples are then arranged (at
least conceptually) in a 2D matrix, with each row corresponding to
the samples of a particular sweep and the columns corresponding to
sequential sweeps.
[0036] After multiplication by the signal mask, then groups of N'
successive samples are summed together at each block
22.sub.0-22.sub.S-1. Inside each block 22.sub.0-22.sub.S-1 is
depicted a register or memory capable of storing N' successive
multi-bit pieces of data (each piece of data here being the result
of the multiplication of one signal mask value with one acquired
signal sample as described in the preceding paragraph) and a summer
.SIGMA. where the N' successive multi-bit pieces of data are summed
together to "invert" the N-element code and produce a Q.times.K
matrix for each antenna element 10. The real and imaginary parts
are separately summed at summer .SIGMA. to produce complex results.
So N' successive samples are summed together and then this is
repeated so that another N' successive samples are summed together,
as so on. Following each summation, the resulting complex data
value corresponds to a specific frequency step since the N' input
values to the adder correspond to a single frequency step (see FIG.
2). Once this signal masking and addition is completed for the
entire acquisition, the result will be a set of Q times K data
values, where Q is the number of frequency steps per sweep and K is
the number of sweeps per acquisition. This data may then be
organized (at least conceptually) into a matrix with Q rows and K
columns. The subsequent 2D FFT processing at blocks
24.sub.0-24.sub.S-1 is then performed on this matrix of data, which
processing is represented by the S 2D FFT functions
24.sub.0-24.sub.S-1 depicted for the embodiments of FIGS. 3 and 3b.
These operations are carried out in parallel for all of the S
digital channels.
[0037] The result of the FFT processing is a matrix of data values
for each channel whose amplitudes indicate the scattered energy at
a particular range (row number), particular range rate (column
number), and set of bearing angles (channel). This FFT processing
is possible because the transmitted radar signal consists of a
series of linear FM sweeps as shown by FIG. 2. A distinguishing
feature of the CAR processing of FIGS. 3 and 3b is the
multiplication of the signal by a stored signal mask (at the signal
mask functions 20.sub.0-20.sub.S-1) prior to applying the FFT
processing (at the 2D FFT functions 24.sub.0-24.sub.S-1).
[0038] The processing downstream of the A/D convertor 18 is
described above with terms such a register, summer, linear
combinations, FFT processing and are associated with blocks on a
block diagrams, but is should be understood that it is preferred to
embody all of the data processing downstream of the A/D
convertor(s) 18 in an appropriately programmed digital processor as
opposed to by using discrete digital circuits.
[0039] In contrast to the technique described in the prior patent
applications referenced above, this invention utilizes a number (N)
of codes (preferably equal to the number of antenna elements 12 and
phase shifters 12) at each frequency step and repeats the same
codes (usually in the same order each time, but not necessarily so)
at each frequency step and also from sweep to sweep. The codes are
generally not chosen pseudo-randomly, although they can be without
loss of performance as long as the set of codes produces linearly
independent field patterns. If the complex field pattern produced
by the n.sup.th antenna element is denoted e.sub.n(.OMEGA.), where
.OMEGA. is shorthand notation for the spherical coordinate angles
.theta.,.phi., then the complex field pattern produced by the
receive elements, phase shifters, and summation network may be
written
g m ( .OMEGA. ) = n = 0 N - 1 S m , n e n ( .OMEGA. ) Eqn . ( 1 )
##EQU00001##
where S.sub.m,n is a "coding matrix," defined as the complex
transmission coefficient (i.e., S21) from the n.sup.th element to
the summation network output for the m.sup.th code.
[0040] The number N' of codes is preferably equal to the number of
phase shifters 12 and antenna elements 10 as is discussed above
(but not necessarily so, as is also discussed above). Increasing N'
causes the multiplicative noise to drop and therefor one might well
ask oneself if it might be a good idea to further increase N' so
that it is greater than the number of phase shifters? When N' is
equal to the number of phase shifters, one can invert the code and
determine the signals at each element. This allows one to digitally
form a set of beams by forming linear combinations of the phase
shifted antenna element signals in an array, and the resulting
range/Doppler/angle estimates are free from the type of ambiguity
(which may be referred to as "residual ambiguity") that is due to
N' being less than the number of elements (or phase shifters).
Increasing the number N' to a number greater than the number of
phase shifters is certainly possible, but such an embodiment is not
believed to provide any additional performance benefits.
[0041] For a single ideal scatterer at range r, radial velocity v,
and angular position .OMEGA., the mixer output voltage has the
form
v n ' , q , k = - j 2 .omega. q c ( r + v ( n ' + qN ' + kQN ' )
.DELTA. t ) g n ' ( .OMEGA. ) = - j 2 .omega. q c ( r + v ( n ' +
qN ' + kQN ' ) .DELTA. t ) n = 0 N - 1 S n ' , n e n ( .OMEGA. )
Eqn . ( 2 ) ##EQU00002##
where .omega..sub.q are the radian frequency steps and .DELTA.t is
the duration of each code (so that N'.DELTA.t is the duration of
each step, see FIG. 2). We will assume that the number of frequency
steps is Q and the number of sweeps is K. These parameters are
chosen to provide the desired range and velocity resolution. The
range and velocity resolutions are given by
.DELTA. r = c 2 .DELTA. f , .DELTA. v = c 2 f o N ' QK .DELTA. t
Eqn . ( 3 ) ##EQU00003##
[0042] Mathematically, the range and velocity variables run over
positive and negative values, even though the negative range
variables are meaningless in practice, so the maximum range and
velocities (determined by the Nyquist criterion) are given by
r max = 1 2 Q .DELTA. r , v max = 1 2 K .DELTA. v . Eqn . ( 4 )
##EQU00004##
[0043] To determine the elements signals we must invert the
aperture code, and this may be achieved in the following manner. We
multiply the mixer output voltage in Eqn. (2) by the conjugate of
the field pattern produced by the inverse or a pseudo-inverse (a
pseudo-inverse is used when the S matrix is not square; if the
number of unique codes is not equal to the number of elements, then
a pseudo inverse must be used) of the code matrix:
v ~ p , q , k = n ' = 0 N ' - 1 ( S - 1 ) p , n ' v n ' , q , k and
thus Eqn ( 5 a ) = n ' = 0 N ' - 1 ( S - 1 ) p , n ' - j 2 .omega.
q c ( r + v ( n ' + qN ' + kQN ' ) .DELTA. t ) n = 0 N - 1 S n ' ,
n e n ( .OMEGA. ) and thus Eqn ( 5 b ) = - j 2 .omega. q c ( r + v
( qN ' + kQN ' ) .DELTA. t ) n = 0 N - 1 e n ( .OMEGA. ) n ' = 0 N
' - 1 ( S - 1 ) p , n ' S n ' , n - j 2 .omega. q c vn ' .DELTA. t
. Eqn . ( 5 c ) ##EQU00005##
[0044] It may be noted that p appears here for the first time in
the equations above. We will see below (Eqn. 9) that p is the index
for the p.sup.th antenna array element. The matrix S.sup.-1 is just
the inverse of S. One may choose an orthonormal code matrix so that
S.sup.HS is proportional to the identity matrix but this is not
necessary. However, an orthonormal code matrix has the advantage of
being optimally conditioned, so it is more tolerant to numerical
errors.
[0045] For practical radars in accordance with this invention, the
total acquisition time (or acquisition period) N'QK.DELTA.t is
designed to be short enough so that the fastest expected targets
will not move through more than one range bin during the
acquisition period since such movements blurs the radar response.
So it is preferable if the target stays within one range bin during
an acquisition, and thus preferably moves no more than half a range
bin per acquisition period. Thus a range bin is defined as equal to
c/(2*.DELTA.f), where c is the speed of light and .DELTA.f is the
RF bandwidth of the sweep. Given this definition, the movement
during a single code duration .DELTA.t is often negligibly small.
From Eqn. (5), if we assume
2 .omega. o c v max N ' .DELTA. t << 1 Eqn . ( 6 )
##EQU00006##
then the last exponential factor in Eqn. (5c) that depends on n'
may be neglected with little error. Using Eqn. (3) and Eqn. (4),
the condition above in Eqn. (6) may be expressed as
Q>>.pi., Eqn. (7)
a condition that is often satisfied in practice because Q is the
number of range bins and this is typically much larger than 3.
Assuming that Eqn. (7) applies, Eqn. (5c) may be simplified to
v ~ p , q , k = - j 2 .omega. q c ( r + v ( q + kQ ) N ' .DELTA. t
) n = 0 N - 1 e n ( .OMEGA. ) n ' = 0 N ' - 1 ( S - 1 ) p , n ' S n
' , n . Eqn . ( 8 ) ##EQU00007##
[0046] Since the last summation is equal to the identity matrix
(i.e.
n = 0 N - 1 ( S - 1 ) p , n S n , m = .delta. p , m ##EQU00008##
where ##EQU00008.2## .delta. p , m ##EQU00008.3##
is the Kronecker delta function), we have
v ~ p , q , k = - j 2 .omega. q c ( r + v ( q + kQ ) N ' .DELTA. t
) e p ( .OMEGA. ) . Eqn . ( 9 ) ##EQU00009##
[0047] This result clearly shows that the variable p indexes the
pth antenna array element.
[0048] Linear combinations of these functions over the index p may
be taken to form receive beam patterns with the desired
characteristics (pointing direction, sidelobes, etc.). For example,
we may form a beam in direction .OMEGA.' by choosing the elements
weights to be the conjugate of the signals due to a target in the
direction .OMEGA.', with the elements possibly multiplied by an
aperture taper w.sub.p for sidelobe control:
v ~ ~ q , k = - j 2 .omega. q c ( r + v ( q + kQ ) N ' .DELTA. t )
p = 0 N - 1 .alpha. p ( .OMEGA. ' ) e p ( .OMEGA. ) . Eqn . ( 10 )
##EQU00010##
where .alpha..sub.p(.OMEGA.')=w.sub.pe.sub.p*(.OMEGA.') are element
weights.
[0049] Applying the mathematical operations described above, but
retaining the original mixer signal expression v.sub.n',q,k allows
us to identify the signal mask values explicitly:
v ~ ~ q , k = p = 0 N - 1 n ' = 0 N ' - 1 .alpha. p ( .OMEGA. ' ) (
S - 1 ) p , n ' v n ' , q , k = n ' = 0 N ' - 1 p = 0 N - 1 [
.alpha. p ( .OMEGA. ' ) ( S - 1 ) p , n ' ] v n ' , q , k = n ' = 0
N ' - 1 s n ' ( .OMEGA. ' ) v n ' , q , k Eqns . ( 11 )
##EQU00011##
[0050] The numbers s.sub.p, (.OMEGA.') are the N' complex numbers
forming the signal mask, one set for each desired beam direction
.OMEGA.'. From the last equation in Eqns. (11) above, one
multiplies N' successive signal samples v.sub.n',q,k by the mask
values s.sub.n'(.OMEGA.') and adds them up. The choice of the
element weights .alpha..sub.p(.OMEGA.') in Eqns. (11) determine the
main beam width and sidelobe performance, as is well known to those
skilled in the art. The reader should take note that
s.sub.n'(.OMEGA.') (with a lower case s) are the N' complex numbers
forming the signal mask for each direction .OMEGA.' whereas
S.sub.n',n (with an upper case S) is the coding matrix. The
relationship between these two is shown in Eqns. (11).
[0051] The range and velocity estimates are then made in the usual
manner, such as taking a 2D discrete FFT of Eqn. (11) over the
indices q and k. The result will be an ambiguity function that
provides estimates of the range, velocity, and bearing angles for
scatterers (objects) 8 located with the field of view of the
radar.
[0052] FIGS. 4a-4d shows the result of a computer simulation of CAR
using the processing technique described herein. A set of sixteen
orthogonal codes was used on receive only. The codes were inverted
using the techniques described with reference to Eqns, 2 through
10, above. For the simulation we assumed a linear receiving antenna
array of sixteen z-directed dipoles and considered beams only in
the x-y plane so that we needed only consider the polar variable
.phi.. For a single point target one can see that the location of
the target in the range, velocity, and phi (.phi.) spaces is well
defined. The table below shows some of the parameters of the
simulation, including N (codes per frequency step), Q (frequency
steps per sweep), and K (number of sweeps):
TABLE-US-00001 Parameter Value N 16 Q 64 K 64 Range 13 m Vel 10 m/s
Az 0 deg
[0053] For comparison, FIGS. 5a-5d show the simulation results
using a coding scheme with a different code for every frequency
step (and different from sweep to sweep). The parameters set forth
in the preceding table where also used in this simulation. As one
can clearly see the residual ambiguity (multiplicative noise) that
results from this prior coding scheme. This "noise" is due to an
insufficient number of measurements as compared to the number of
unknowns.
[0054] For a further comparison, FIGS. 6a-6d show another
simulation result using a coding scheme, with a different code for
each sweep. The parameters set forth in the preceding table where
also used in this simulation. Now the multiplicative noise is
absent from the range space, but present (and stronger) in the
velocity and azimuth spaces.
[0055] In the prior U.S. patent application Ser. No. 13/490,607
referenced above, at paragraph 0047 thereof, we describe how the
phase shifter states change relative to the frequency sweep. In
that application we described using a fixed set of phase shifter
states per frequency sweep, and changing the code from sweep to
sweep. For that application there are K codes, where K is equal to
the number of sweeps over the field of view (FOV). Increasing the
number of codes to Q times K, where Q equals the number of
transmitted frequency shifts per sweep, by changing the phase
shifter states at each frequency step, reduces the residual
ambiguity further, but does not eliminate it. But including N codes
per frequency step for a total of N' times K times Q codes
eliminates the residual ambiguity that we want to overcome. As N'
increases the residual ambiguity drops. However, one need to not
increase N' to infinity. When N' is equal to the number of phase
shifters 12 (the number of which shifters is equal to N), one can
invert the code and determine the signals at each element. This
allows one to create a set of beams for the array, and the
resulting range/Doppler/angle estimates are free from the type of
ambiguity which occurs when N' is less than the number of receiving
elements (or phase shifters). Thus, preferably N' is equal to N,
that is, N' is preferably equal to the total number of phase
shifters 12, and this is true regardless of whether the receiving
antenna elements 10 are arranged in a 1D or a 2D array.
[0056] Following the 2D FFT processing, the significant scatterers
are typically identified by applying "thresholding" to the data
outputted from the 2D FFT processing where any samples crossing a
chosen threshold are retained and samples falling below that
threshold are omitted. Additional processing may be applied to
group significant samples together in order to identify single,
large objects that may produce many different, but related,
scattering events. Using such processing techniques, the radar
sensor can provide functions such as, for example, identification
of objects on a collision path with the host vehicle.
CAR Coding on Receive Only Vs CAR Coding on Both Receive and
Transmit
[0057] In theory this invention can be also used on the transmit
side of the radar. But in practice there are difficulties in doing
this. To keep the processing simple and fast (with low latency),
FFT processing of the range/Doppler signals is preferred. But to
use FFT processing one must make sure the frequency sweep period is
much longer (>10 times) than the time delay to the furthest
target and back. And one also has to make sure that the sweep
period is short enough to sample the Doppler signal without
aliasing. The result is that one is not free to choose the sweep
period arbitrarily, so one cannot simply increase the sweep period
when one increases the number of codes per frequency step. If many
different transmitted codes are reflected back by targets at
different ranges and are received in a single sweep period, one
must use correlation processing to determine when each code has
arrived. This is much more complicated and computationally
expensive than the receive-side FFT processing techniques described
above, so we prefer to use CAR processing only on the received
radar signals for low cost applications such as using CAR in an
automotive radar application. But in military applications where
cost is of less concern, it is certainly possibly use CAR on
transmit or on both transmit and receive. In fact, one could then
make the coding intervals (on transmit) very short and treating the
radar as a phase coded radar to determine range. When viewed this
way, employing CAR on transmit is essentially a phase coded radar
that transmits a different phase code in every direction (e.g.,
over index p).
[0058] CAR coding on receive does not suffer the timing issues
discussed in the preceding paragraph because the coding is all done
at the same time so there need be no time delays with respect to
the coding (when using CAR on in the radar receiver).
[0059] From these two comparisons one can see the advantages of the
present invention in reducing multiplicative noise and thereby
increasing the sensitivity and dynamic range of radar using
CAR.
[0060] The accompanying Appendix A is an article which will be
published after the filing date of this patent application.
Appendix A, which is incorporated herein by reference, provides
additional background information and addition technical
information regarding the advantages and drawbacks of this
invention compared with other radar schemes.
[0061] This concludes the description of embodiments of the present
invention. The foregoing description of these embodiments has been
presented for the purposes of illustration and description. It is
not intended to be exhaustive or to limit the invention to the
precise form or methods disclosed. Many modifications and
variations are possible in light of the above teachings. It is
intended that the scope of the invention be limited not by this
detailed description, but rather by the claims appended hereto.
* * * * *