U.S. patent application number 14/003536 was filed with the patent office on 2014-07-24 for assisting converter.
The applicant listed for this patent is Victor Marten, Ioannis Milios, Aleksandar Prodic, Mahmoud Shousha. Invention is credited to Victor Marten, Ioannis Milios, Aleksandar Prodic, Mahmoud Shousha.
Application Number | 20140203737 14/003536 |
Document ID | / |
Family ID | 50184263 |
Filed Date | 2014-07-24 |
United States Patent
Application |
20140203737 |
Kind Code |
A1 |
Prodic; Aleksandar ; et
al. |
July 24, 2014 |
Assisting Converter
Abstract
What is described is a battery management architecture that
eliminates previously described problems of the previous solutions
and compensates for the extra cost of a cell-balancing circuit.
These advantages are achieved by integrating the voltage step-up
and balancing functions as well as charging functions inside a
single converter topology. Instead of providing the entire output
voltage and power, the converter in this configuration is merely
assisting the battery by providing a portion of the power delivered
to the load, rather than the entirety of the power delivered to the
load. This portion of power is proportional to the difference
between the output and the battery pack voltages.
Inventors: |
Prodic; Aleksandar;
(Toronto, CA) ; Shousha; Mahmoud; (Toronto,
CA) ; Marten; Victor; (Flushing, NY) ; Milios;
Ioannis; (New York, NY) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Prodic; Aleksandar
Shousha; Mahmoud
Marten; Victor
Milios; Ioannis |
Toronto
Toronto
Flushing
New York |
NY
NY |
CA
CA
US
US |
|
|
Family ID: |
50184263 |
Appl. No.: |
14/003536 |
Filed: |
August 27, 2013 |
PCT Filed: |
August 27, 2013 |
PCT NO: |
PCT/US2013/056917 |
371 Date: |
March 5, 2014 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
61693644 |
Aug 27, 2012 |
|
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|
61867956 |
Aug 20, 2013 |
|
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Current U.S.
Class: |
318/139 ;
307/77 |
Current CPC
Class: |
H02J 7/0014 20130101;
Y02T 10/92 20130101; H02M 3/33584 20130101; Y02T 10/70 20130101;
H02J 7/345 20130101; Y02B 40/00 20130101; H02J 7/0019 20130101;
B60L 58/22 20190201 |
Class at
Publication: |
318/139 ;
307/77 |
International
Class: |
H02J 7/00 20060101
H02J007/00; B60L 11/18 20060101 B60L011/18 |
Claims
1. A system for use with a voltage source and a load and a
capacitor, the voltage source defining first and second nodes, the
capacitor connected to the second node and thereby defining a third
node, the load connected with the first node and the third node,
the voltage source comprising a plurality of series-connected
energy sources; a module connected with the energy sources of the
voltage source, the module connected with the capacitor at the
second node and the third node; the module comprising an inductor;
the module further comprising a plurality of first bidirectional
circuits, each first bidirectional circuit connected with a
respective energy source of the voltage source and connected with
the inductor; the module further comprising a second bidirectional
circuit, the second bidirectional circuit connected with the
capacitor and connected with the inductor; the module further
comprising a controller connected with the first bidirectional
circuits and with the second bidirectional circuit and disposed to
control the bidirectional circuits to pass energy therebetween.
2. The system of claim 1 wherein each bidirectional circuit is an
active-bridge forward converter comprising four semiconductor
switches in a bridge.
3. The system of claim 2 wherein the controller is responsive to
measured information about the energy sources of the voltage
source, for generating control signals to the semiconductor
switches of the active-bridge forward converters of the plurality
of first bidirectional circuits and for generating control signals
to the semiconductor switches of the active-bridge forward
converter of the second bidirectional circuit.
4. The system of claim 1 wherein each of the energy sources
comprises an electrochemical cell.
5. The system of claim 1 wherein each of the energy sources
comprises a photovoltaic module.
6. The system of claim 1 further comprising a respective capacitor
in parallel with each of the energy sources.
7. The system of claim 4 wherein the load comprises a motor of a
vehicle.
8. The system of claim 1 wherein the plurality of energy sources
have different energy content, and wherein the controller draws
current from a first of said plurality of energy sources and passes
the current to a second of said plurality of energy sources.
9. The system of claim 8 wherein the drawing of current from the
first of said plurality of energy sources and the passing of
current to the second of said plurality of energy sources tends to
balance the energy content thereof.
10. The system of claim 1 wherein the controller draws current from
a first of said plurality of energy sources and passes current to
the capacitor.
11. The system of claim 10 wherein the drawing of current from the
first of said plurality of energy sources and the passing of
current to the capacitor tends to keep the voltage between the
first and third nodes nearly constant.
12. A method for use with a voltage source and a load and a
capacitor, the voltage source defining first and second nodes, the
capacitor connected to the second node and thereby defining a third
node, the load connected with the first node and the third node,
the voltage source comprising a plurality of series-connected
energy sources, the method comprising: drawing a first level of
current from at least a first one of the plurality of
series-connected energy sources; drawing a second level of current
from at least a second one of the plurality of series-connected
energy sources, the first level of current being non-identical to
the second level of current; passing the current from the at least
a first one of the plurality of series-connected energy sources
through a respective bidirectional circuit, the respective
bidirectional circuit having an output; passing the current from
the at least a second one of the plurality of series-connected
energy sources through a respective bidirectional circuit, the
respective bidirectional circuit having an output; passing the
outputs of the bidirectional circuit respective to the at least a
first one of the plurality of series-connected energy sources and
of the bidirectional circuit respective to the at least a second
one of the plurality of series-connected energy sources through a
second bidirectional circuit to the capacitor.
13. The method of claim 12 wherein the passing of currents tends to
keep the voltage between the first and third nodes nearly
constant.
14. The method of claim 12 further comprising the step, carried out
with respect to a third one of plurality of series-connected energy
sources having a smaller energy content than the energy content of
the first one of the plurality of series-connected energy sources,
of passing the outputs of the respective first bidirectional
circuits of the first and second ones of the plurality of
series-connected energy sources to the third one of the plurality
of series-connected energy sources through a first bidirectional
circuit respective thereto.
15. The method of claim 14 wherein the drawing of current from the
first at least one energy source and the passing of current to the
third at least one energy source tends to balance the energy
content thereof.
16. A system for use with a voltage source and a load and a
capacitor, the voltage source defining first and second nodes, the
capacitor connected to the second node and thereby defining a third
node, the load connected with the first node and the third node; a
module connected with the voltage source, the module connected with
the capacitor at the second node and the third node; the module
comprising an inductor; the module further comprising a first
bidirectional circuit, the first bidirectional circuit connected
with the voltage source and connected with the inductor; the module
further comprising a second bidirectional circuit, the second
bidirectional circuit connected with the capacitor and connected
with the inductor; the module further comprising a controller
connected with the first bidirectional circuit and with the second
bidirectional circuit and disposed to control the bidirectional
circuits to pass energy therebetween.
17. The system of claim 16 wherein each bidirectional circuit is an
active-bridge forward converter comprising four semiconductor
switches in a bridge.
18. The system of claim 17 wherein the controller is responsive to
measured information about the voltage source, for generating
control signals to the semiconductor switches of the active-bridge
forward converter of the first bidirectional circuit and for
generating control signals to the semiconductor switches of the
active-bridge forward converter of the second bidirectional
circuit.
19. The system of claim 16 wherein the voltage source comprises a
battery comprising electrochemical cells.
20. The system of claim 16 wherein the voltage source comprises a
photovoltaic array.
21. The system of claim 19 wherein the load comprises a motor of a
vehicle.
22. The system of claim 16 wherein the controller draws current
from the voltage source and passes current to the capacitor.
23. The system of claim 22 wherein the drawing of current from the
voltage source and the passing of current to the capacitor tends to
keep the voltage between the first and third nodes nearly
constant.
24. A method for use with a voltage source and a load and a
capacitor, the voltage source defining first and second nodes, the
capacitor connected to the second node and thereby defining a third
node, the load connected with the first node and the third node,
the method comprising: drawing a first level of current from the
voltage source; passing the current from the voltage source through
a first bidirectional circuit, the first bidirectional circuit
having an output; passing the output of the first bidirectional
circuit through a second bidirectional circuit to the capacitor;
wherein the passing of currents tends to keep the voltage between
the first and third nodes nearly constant.
25. A method for use with a voltage source and a current source and
a capacitor, the voltage source defining first and second nodes,
the capacitor connected to the second node and thereby defining a
third node, the load connected with the first node and the third
node, the voltage source comprising a plurality of series-connected
energy sources, each of the series-connected energy sources
associated with a respective first bidirectional circuit, the
method comprising: receiving a charging current from the current
source, thereby tending to charge the capacitor; drawing current
from the capacitor through a second bidirectional circuit to an
inductor; passing a first level of current from the inductor to the
first bidirectional circuit respective to a first one of the
plurality of series-connected energy sources; passing a second
level of current from the inductor to the first bidirectional
circuit respective to a second one of the plurality of
series-connected energy sources, the first level of current being
non-identical to the second level of current.
26. The method of claim 25 wherein the passing of current to the
first at least one of the plurality of series-connected energy
sources and the passing of current to the second at least one of
the plurality of series-connected energy sources tends to balance
the energy content thereof.
27. The system of claim 1 wherein one of several pre-defined
discrete values of relative phase shifts is assigned to each energy
source, depending on its state of charge.
28. The system of claim 16 wherein one of several pre-defined
discrete values of relative phase shifts is assigned to each energy
source, depending on its state of charge.
Description
BACKGROUND
[0001] A typical power management system of an electric or hybrid
vehicle is shown in FIG. 1. It consists of a battery pack 101
(usually comprised of large number of Lithium-Ion or
Lithium-Polymer cells), a step-up stage (usually boost-based
converter) 103, and a motor drive 104 providing power for an
electric motor 102. In many vehicle designs there is regenerative
braking which permits recharging of the battery pack 101, the
details of which are omitted for clarity in FIG. 1.
[0002] In some cases, the power management system also includes a
cell balancing circuit 105, which compensates for different states
of charges (SOC) of individual cells, as shown in FIG. 1. The SOC
variations usually occur due to the aging and variations in the
manufacturing process. Through cell balancing the effective
capacity of the battery pack can be significantly increased.
[0003] The balancing circuits can be divided into two general
categories. The first one is the passive balancing systems, in
which the cells are balanced by dissipating energy from excessively
charged cells, through resistors.
[0004] The second category is the active balancing systems, which
are far more efficient. In these systems, the energy of
over-charged cells is transferred to those with less charge using
dc-dc converters. Even though the benefits of the active cell
balancing are known, their use is relatively sparse, due to the
overly large extra cost and weight the cell balancing circuits add
to the system.
[0005] Balancing systems of possible interest include those
described in the following US patent published applications: [0006]
US 2012-0256593 A1 [0007] US 2012-0249052 A1 and in the following
granted US patents: [0008] U.S. Pat. No. 7,936,150 [0009] U.S. Pat.
No. 8,269,455 and in the following published international patent
application: [0010] WO/2012/172468 all of which are owned by the
same assignee as the assignee of the present patent
application.
[0011] A prior-art balancing circuit 105 might utilize a buck-boost
and a Cuk converter for cell balancing. These topologies can be
implemented with a relatively small number of active components and
regulated with fairly simple controllers. However, these circuits
are fairly large in form factor, which reflects on the overall
physical size of the system. Implementations based on the use of a
bi-directional flyback and a two stage flyback converters have been
proposed. Compared to other solutions, these systems lower
efficiency at high power levels.
[0012] Others have proposed a configurable system for cell
balancing using a large number of switches to transfer the energy
between cells. The main drawback of such a system is that the
balancing becomes too slow for the energy transfer between cells
having similar output voltages.
[0013] Many prior-art approaches direct themselves only to a single
one of the functions suggested by the functional blocks of FIG. 1.
This prompts the alert reader to recall the functional block 103.
The step-up converter 103 of FIG. 1 steps up the voltage from the
battery 101 to the extent that is required to provide a desired DC
voltage to the load 104. It will be appreciated that the step-up
converter 103 as shown in FIG. 1 is required to be able to
accommodate the entirety of the power transferred to the load 104.
This means that the converter 103 must use switches (typically
semiconductor switches) and reactive components (for example
capacitors and inductors) that allow the converter to pass the full
load power, rated for full load currents and rated for the full
reverse voltages that might arise in serving the full load voltage.
In some cases the current and voltage for which the switches must
be rated is larger than the current and voltage at the output, for
example in a flyback circuit. The switches in a boost converter
also conduct larger than the load current.
[0014] It would be desirable if a way could be found to accomplish
the aims of the functional blocks of FIG. 1 in a more integrated
way, using fewer components than in prior-art approaches, and using
components that would not require the full-voltage ratings of the
semiconductor switches in some prior-art approaches. The cost of a
semiconductor switch often increases at least linearly with the
voltage rating of the switch and may increase faster than linearly.
Thus there are rewards for the designer who devises topologies and
approaches that permit use of components with smaller voltage
ratings as compared with those needed in prior-art topologies and
approaches.
SUMMARY OF THE INVENTION
[0015] What is described is a battery management architecture that
eliminates previously described problems of the previous solutions
and compensates for the extra cost of a cell-balancing circuit.
These advantages are achieved by integrating the voltage step-up
and balancing functions as well as charging functions inside a
single converter topology. Instead of providing the entire output
voltage and power, the converter in this configuration is merely
assisting the battery by providing a portion of the power delivered
to the load, rather than the entirety of the power delivered to the
load. This portion of power is proportional to the difference
between the output and the battery pack voltages.
DESCRIPTION OF THE DRAWING
[0016] The invention will be described with respect to a drawing in
several figures.
[0017] FIG. 1 shows a prior-part approach to cell balancing and
providing converted power to a load.
[0018] FIG. 2 shows in functional block diagram form the topology
according to an embodiment of the invention.
[0019] FIG. 3 shows the system of FIG. 2 in greater detail.
[0020] FIG. 4 shows a single cell channel of the system in FIG. 3
in greater detail.
[0021] FIG. 5 shows control waveforms.
[0022] FIG. 6 shows signal levels for various stages during a cycle
of operation in an embodiment of the invention.
[0023] FIG. 7 shows equivalent circuits for a single cell channel,
for various modes of operation.
[0024] FIG. 8 shows the system of FIG. 4 with snubbers.
[0025] FIG. 9 shows the controller 106 in greater detail.
[0026] FIG. 10 shows various inductor topologies that may be
employed.
[0027] Where possible, like reference numerals have been employed
for like elements.
DETAILED DESCRIPTION
[0028] The introduced topology, here referred to as an "assisting
converter" architecture, is shown in FIG. 2. The converter 106
operates at such that the stepped-up output voltage at 111 is
formed as a sum of the battery pack voltage V.sub.batt (across
battery 101) and the output voltage of a bidirectional multi-input
single output converter stage, V.sub.f, (across capacitor C.sub.f)
where the inputs of the converter 106 are connected to the battery
cells of the battery 101.
[0029] As described below, the assisting configuration drastically
reduces power processing requirements needed to achieve required
conversion and, consequently, improves power processing efficiency.
The introduced topology also allows fast cell balancing even when
the cells have similar or equal voltages and allows energy transfer
between any two cells in the battery pack. Furthermore, unlike
other solutions, the topology provides functionality of the system
even when a significant number of cells in the battery pack are out
of function, potentially improving overall system reliability.
[0030] The principle of operation of a general assisting converter
106 shown in FIG. 2 can be described through a comparison with the
conventional system of FIG. 1. For the assisting converter 106 the
power delivered to an output load 104 that takes current I.sub.out
is:
P.sub.out=V.sub.outI.sub.out=(V.sub.batt+V.sub.cf)I.sub.out=P.sub.conver-
ter+P.sub.batt.sub.--.sub.direct,
where P.sub.converter=V.sub.cfI.sub.out is the portion of the
battery power delivered through the converter and
P.sub.batt.sub.--.sub.direct=V.sub.battI.sub.out is the remaining
portion, directly delivered by the battery without any
processing.
[0031] For the conventional converter 103 (FIG. 1),
P.sub.converter=P.sub.out and P.sub.batt.sub.--.sub.direct is 0,
since the step-up converter processes all of the power delivered to
the load. This means that, in comparison with the conventional
system of FIG. 1, the converter 106 in the assisting configuration
of FIG. 2 is required to process V.sub.cf/V.sub.out times lower
power and, thus, can be implemented with smaller components. The
reduction in processing power also makes design of a highly
efficient assisting converter 106 simpler. For the assisting system
of FIG. 2 the overall power processing efficiency can be defined
as:
.eta. = ( P batt _ direct P out 100 % + P conv P out .eta. conv ) =
( V batt V out 100 % + V conv V out .eta. converter ) ,
##EQU00001##
where .eta..sub.converter is the efficiency of the assisting
converter 106. This expression can be explained by looking at the
system of FIG. 2 and noticing that the power processing losses
occur only for the portion of power that is not directly provided
by the battery 101. As described early, this portion of power is
proportional to the ratio of the output and converter-provided
voltages.
[0032] For example, if the assisting converter 106 is providing a
20% of the output voltage and has a very low power processing
efficiency of 50%, a relatively high power processing efficiency of
90% at the system level still can be achieved. The system may be
designed so that the capacitor 107 carries only 20% of the voltage
intended to be delivered to the load, or only 10%, or only 5%.
[0033] This shows that, in order to achieve targeted overall system
efficiency, the assisting converter 106 (FIG. 2) can be designed
with much less stringent power processing requirements than the
conventional converter 103 (FIG. 1), further reducing the overall
system cost and complexity.
[0034] It will be helpful to characterize some of the benefits of
the inventive topology of FIG. 2 by way of a generalization of the
disclosed topology. A series of cells (in battery 101) is seen.
Each cell has a bidirectional converter associated with it (part of
block 106). The bidirectional converter relating to a particular
cell is able to draw from capacitor 107 to charge up the particular
cell. The bidirectional converter relating to a particular cell is
able to draw from the particular cell to charge up capacitor 107.
This provides three distinct functions: [0035] if a charging
current is supplied to the system, the system is able to receive
that current and is able to pass along the charging current to the
various cells each according to its own needs (so that the charging
process leads to a near-balance in the state of charge of the
cells); [0036] when power is being supplied to the load, the system
is able to draw upon the various cells as needed to charge up the
capacitor 107 so as to lead to a nearly constant voltage being
supplied to the load at 111; [0037] in any regime (discharging,
charging, or quiescence) a balancing of cells (in terms of state of
charge) may be accomplished.
[0038] Yet another benefit presents itself, namely that even if a
particular cell fails "open" the system will be able to maintain a
substantial portion of its function despite loss of that cell.
Indeed the system will be able to continue its function even with
loss of two or more cells in an "open" failure mode.
[0039] Finally the alert reader will appreciate that the topology
of FIG. 2 may be generalized to the case of a single cell or a
battery of cells that is employed as a simple two-terminal device.
The cell (or two-terminal battery) may be placed in series with a
capacitor 107 with the assisting converter 106 connected to both.
In such a situation, many of the benefits of the topology are
preserved such as the reduced need for high voltage ratings in the
capacitor and in the semiconductors of the converter, as well as
the need to worry only about the conversion efficiency of the
portion of the power associated with the voltage on the capacitor
107. (More will be said about this below.)
Battery Cell Balancing.
[0040] It will be recalled from FIG. 1 that often a balancing
function (block 105) is provided. In the system of FIG. 2, to
provide battery cell balancing, the input currents of the
bi-directional multi-input converter 106 can be regulated. The
regulation can be performed such that the currents provided by
(drawn from) the individual cells are proportional to their states
of charge. The alert reader will appreciate that it is possible to
achieve an energy transfer from cells with a higher state of charge
to other cells having less charge, through what may be termed an
"indirect" energy transfer. In this process, cells with high SOC
transfer energy to the capacitor, while the cells with low SOC take
energy from it. Both of these cases are demonstrated as
follows.
[0041] An implementation of the assisting converter 106 based on a
multi-phase isolated dual active bridge converter is shown in FIG.
3. In addition to providing galvanic isolation and bi-directional
energy flow, the dual active bridge (DAB) has a number of other
features that make it very attractive for the targeted
applications. Those include high power processing efficiency
(achieved through inherent zero voltage or current switching) and
much smaller inductor volume comparing to the conventional hard
switching and resonant topologies. The small inductance value opens
a possibility for elimination of a discrete inductor through the
utilization of the transformer leakage inductance.
[0042] Another interesting feature of the DAB is that it can
operate with both continuous input and continues output currents,
thus reducing requirements for input and output filters.
[0043] The system of FIG. 3 consists of a number of transformers
(e.g. 123) whose primary windings are connected to the individual
battery cells (e.g. 121) and the secondary windings linked to the
output capacitor 107, through small inductors 127.
[0044] The linkage between the transformer 123 and its respective
cell 121 is by means of an active bridge 122. The active bridge 122
has four semiconductor switches, typically FETs (field-effect
transistors).
[0045] The linkage between the transformer 123 and the capacitor
107 is also by means of an active bridge 124. The active bridge 124
also has four semiconductor switches, typically FETs (field-effect
transistors).
[0046] It is these semiconductor switches that are driven by
control signals 128 from the controller 125. The control signals
have phase relationships which bring about for example a draw of
current from one or another of the cells, or a pumping of current
into one or another of the cells, and which bring about a
charging-up of capacitor 107 or a drawing-down of capacitor
107.
[0047] An implementation based on a multi-winding transformer is
also possible. This is shown in FIG. 10.
[0048] The digital controller 125 implementing phase-shift
modulation regulates the operation of this converter 106. The phase
shift control provides both the output voltage regulation (charging
of capacitor 107 to tend toward a constant voltage available to
load 104) and cell balancing (balancing the energy content of the
various cells) through the regulation of the currents to and from
the individual cells.
[0049] Typical waveforms of a DAB converter connected between two
DC sources are shown in FIG. 5. The power transfer between the
sources (each of the cells and the output port of the converter in
this case) can be derived from the well-known power transfer
formula for sinusoidal systems:
P k = nV A V B sin ( .PHI. k ) .omega. L , ##EQU00002##
where V.sub.A and V.sub.B are the amplitudes of the two sinusoidal
sources, .phi..sub.k is the phase shift (delay) between the
voltages, and .omega.L the impedance value of an inductor placed
between them. In the case of FIG. 5, the waveforms are not
sinusoidal but are square waves, and so the power transfer equation
becomes:
P k = n V cell V cf .PHI. k ( .pi. - .PHI. k ) .pi..omega. L ,
##EQU00003##
where V.sub.cell is the voltage of the battery cell (e.g. 121 in
FIG. 3), n the turns ratio of the transformer, V.sub.cf is the
voltage of the floating capacitor 107 (FIGS. 2, 3, and 4),
.omega.=2.pi.f.sub.sw where f.sub.sw is the switching frequency of
the converter, and .phi..sub.k is the phase shift (delay) between
the voltages v.sub.1(t) and v.sub.2(t), shown in FIGS. 4 and 5.
[0050] FIG. 6 shows realistic key waveforms of the DAB from FIG. 4.
The diagrams show the state of the switching components that
include non-overlapping transistor times t.sub.d, switch nodes
voltages (nv.sub.1(t), i.e. v.sub.1(t) reflected on the secondary
transformer side, and v.sub.2(t)), and the inductor voltage and
current waveforms, v.sub.L(t) and I.sub.L(t), respectively. The
diagrams also show the current supplied by the battery cell
i.sub.E(t). FIGS. 6(a) to (d) shows transistor on-off states. FIG.
6(e) shows switch node voltages. FIG. 6(f) shows voltage across the
inductor. FIG. 6(g) shows inductor current. FIG. 6(h) shows battery
cell current.
[0051] The non-overlapping times prevent simultaneous conduction of
both switches of a single converter branch, that is, they prevent a
short circuit.
[0052] Eight equivalent circuits of FIG. 7, representing different
modes that the left side of the DAB goes through over a switching
period T.sub.s, can be observed. In these equivalent circuits of
FIG. 7, non-conducting MOSFETs are replaced with their parasitic
drain-source capacitances, labeled as C.sub.par. Likewise the
conducting MOSFETs are replaced with short circuits or diodes,
depending on the state of the gate drive signal. The equivalent
circuits also include the leakage inductance of the primary side of
the transformer L.sub.p.
[0053] Mode 1 corresponds to the time interval t.sub.0 to t.sub.1
of FIG. 6(g), during which Q11 and Q14 are turned on while Q12 and
Q13 are switched off, during which and the primary side current
i.sub.PE(t) is negative, that is, is leaving the "dot" on the
inductor.
[0054] Mode 2 occurs during the time interval
t.sub.1<t.ltoreq.t.sub.2. This mode starts when i.sub.PE(t),
that is i.sub.L(t), changes polarity and has the same state of
switches as Mode 1.
[0055] Mode 3 occurs during the transistors' non-overlapping time
(between t.sub.3 and t.sub.4) when all of the switches are turned
off. It can be seen that in this mode a resonant circuit consisting
of the L.sub.p and the capacitive network C.sub.par11 to
C.sub.par14 is formed, meaning that oscillations might occur,
depending on the speed of the body diodes of the MOSFETs. Ideally,
for the case when the antiparallel body diodes are fast, a soft
transition between Mode 3 and Mode 4 occurs. This happens when the
C.sub.par13 is discharged to a value of approximately -V.sub.F and
the charge of C.sub.par11 is approximately equal to
V.sub.cell+V.sub.F, where V.sub.F is the forward voltage drop of
the body diodes, shown in the equivalent circuit of Mode 4.
[0056] In Mode 4, the anti-parallel diodes conduct and the maximum
voltage across the transistors is clamped to a value of
V.sub.cell+2.sub.VF.
[0057] For the case when the body diodes of the MOSFETs are slow,
compared to the period of the resonant circuit oscillations, and,
hence, are not able to react, the circuit does not go through Mode
4. The amplitude of the overshoot is directly proportional to the
energy stored in the leakage inductance at the time instant
t.sub.3, i.e. W.sub.E=1/2i.sub.L(t.sub.3).sup.2L.sub.p, and
inversely proportional to the equivalent of the
C.sub.par11-C.sub.par14 capacitive network.
[0058] Mode 3 (or Mode 4) is followed by Mode 5. Mode 5 starts
immediately after Q12 and Q13 are turned on and occurs during the
time interval t.sub.0<t.ltoreq.t.sub.5. This mode is equivalent
to Mode 1. If prior to this mode the DAB was in Mode 4, both
transistors turn on softly, with zero voltage transition. For a
slow body diode case, i.e. when the previous state is Mode 3, a
soft transition cannot be guaranteed and, consequently, increased
switching losses occur.
[0059] A similar analysis can be carried out for Modes 6 to 8.
[0060] The discussion just given shows that the parasitic drain
source capacitance increases the voltage stress across the switches
and negatively affects the converter efficiency for the case when
the antiparallel diode is slow compared to the frequency of
oscillations. The discussion also indicates that the voltage stress
value and the frequency of oscillations are inversely proportional
to the equivalent capacitance of the C.sub.par11-C.sub.par14
network.
[0061] To minimize this effect a straightforward solution would be
to use faster Schottky diodes connected in parallel with the body
diodes of the transistors. These diodes would provide snubber
action. They would allow the converter 106 to enter Mode 4,
described in the previous discussion, and consequently would
eliminate voltage overshoots while providing zero voltage switching
(ZVS).
[0062] To minimize the cost, in this case, instead of using extra
Schottky diodes, a small ceramic capacitor is placed in parallel
with each of the transistors. These capacitors, labeled as C.sub.s
in FIG. 8, reduce the frequency of the resonant circuit oscillation
during the non-overlapping times. This allows the body diodes to
clamp the voltage.
[0063] The waveform of FIG. 6(g) shows that the current provided by
the battery has large variations. These variations could reduce the
battery life time and, due to a large rms current value, have a
negative effect on the converter efficiency.
[0064] To eliminate this effect a decoupling capacitor C.sub.dec
141 is placed across the primary side bridge, as shown in FIG. 8.
Together with the parasitic inductance of the connecting wires this
capacitor 141 forms a second-order filter which drastically reduces
the input, current ripple.
[0065] In addition to eliminating the large ripple, C.sub.dec 141
can also be potentially used for improving the reliability of the
system in the case of a battery cell failure. In such a situation
the capacitor can act as a replacement for the battery, capable of
maintaining the cell voltage and transferring reactive power.
Controller.
[0066] The main goal of the controller 125 of FIG. 3 is to maintain
the output voltage (at 111) at the desired value while providing
cell balancing. It is shown in more detail in FIG. 9. The control
is performed through phase shift modulation, where the angle on the
secondary side is used for the output voltage and the settings of
the angles on the primary side (relative phase shifts) is used for
cell balancing. In this case, the voltage loop is implemented in a
digital fashion. The voltage at 111 is attenuated (attenuator 151)
yielding an output voltage HV.sub.out(t) which is converted into
its digital equivalent with an analog-to-digital converter (ADC)
152 yielding signal H.sub.vout[n]. This value is than compared (at
comparator 154) to the desired reference V.sub.ref[n] (reference
153) and the resulting error e[n] is passed to a Voltage loop PID
compensator 155. The compensator 155 calculates a value jv[n],
which is the input for the secondary side phase shift modulator 156
(controlling the secondary-side bridge 124). Based on this input
the secondary side phase shift modulator 156 adjusts the phase
shift between the secondary side switches (bridge 124) and one of
the set of primary side switches, i.e. a reference set, such that
the desired output voltage is obtained at 111.
[0067] The relative phase shifts between the DAB switches on the
primary sides (for example bridge 122) are adjusted based on the
cells' state of charge (SOC) (input 126). The calculation of the
relative phase shifts between primary side modules is performed by
the Primary side phase shift calculator (167), which sends four
control signals, jr1[n] to jr4[n], to the Primary side phase shift
modulator 128, for each of the bridges such as 122.
[0068] What was just described is passing the analog voltage to an
ADC and then carrying out a digital difference calculation and
carrying out later steps digitally. The alert reader will
appreciate that there are many ways to provide a controller that
will offer the benefits of the invention. The controller 125 may be
implemented by appropriate firmware in a microcontroller of
suitable bandwidth. Alternatively it may be implemented by an FPGA
with suitable programming. Another approach could be the use of
hardware combining mixed-signal circuits, for example
analog-to-digital converters for measurement, and digital logic for
calculations. An implementation based on the use of application
specific integrated circuits (ASIC) is also possible.
[0069] A simplified balancing method may be employed instead of
using continuously variable phase shifts for the balancing
functions. In this implementation, one of n pre-defined discrete
values of the relative phase shifts is assigned to each cell,
depending on its state of the charge. This way computational
overhead is minimized.
Example
[0070] To verify the previously described concepts, a 4-cell, 200 W
experimental setup was built and tested. At the input, four 6V, 12
AH Lead-Acid cells were used. The DAB stages operated at a
switching frequency of 100 kHz and provided a 42 V regulated
output. The component values for the power stage of FIG. 8 are
shown in this table:
TABLE-US-00001 Component Value Power Transfer Inductance (L.sub.l)
10 .mu.H Primary Leakage Inductance (L.sub.lkp) 0.5 .mu.H (measured
and calculated) Secondary Leakage Inductance 0.5 .mu.H (measured
and calculated) (L.sub.lks) Filter Capacitance (C.sub.dec) 4.4 mF
Transformer Turns Ratio (1:n) 1:4 Input Parasitic Inductance
(L.sub.par) 300 nH (measured) Drain to Source Capacitance 0.75 nF
(estimated) LV side (C.sub.parl1) Drain to Source Capacitance HV
0.45 nF (estimated) side (C.sub.par1) Snubber Capacitance 47 nF
indicates data missing or illegible when filed
[0071] The controller of this setup was implemented with an FPGA
system. For test purposes the system was largely over-designed
allowing an opportunity to verify operation of the assisting
converter at higher power levels. The tests were performed for cell
balancing as well as for developing the voltage assist of the
capacitor 107. Operation of this converter as a battery charger was
also tested. The converter achieved a peak efficiency of 92%.
[0072] The discussion above focuses on a system which is connected
with a battery composed of electrochemical cells. Such a system
might, however, also offer some of its benefits to a solar panel
array of photovoltaic modules.
[0073] The alert reader will likewise appreciate that while the
system is described as connected with a battery of many cells, the
system can likewise offer its benefits in the case of a single
cell. In such a case it provides a more efficient DC-to-DC
converter because less than all of the power is being passed
through the converter.
[0074] It is interesting to consider the ability of this system to
tolerate any of several possible failure modes. As mentioned above,
if a single cell were to fail "open", a suitably sized capacitor
141 may permit continued system function by stepping into the shoes
of the failed cell.
[0075] In many prior-art systems, a capacitor is placed in parallel
with the load, at the output of a step-up (boost) stage (for
example within block 103 in FIG. 1) of a system. In FIG. 1 if a
capacitor is employed in the converter 103 at its output, the
capacitor will need to be rated for the full output voltage. In
contrast, the capacitor 17 need only be rated for the voltages that
it will encounter, which may be only a fraction of the load
voltage. As the price of a capacitor is often proportional to the
square of the voltage rating, the capacitor in block 103 of FIG. 1
will cost much more than the capacitor 107 in FIG. 2. This permits
the disclosed system to be less expensive compared with some
prior-art systems.
[0076] In sum, a new system level architecture for providing both
battery balancing and step-up voltage functions has been described.
The architecture is based on an "assisting converter" concept where
a low-power converter is used merely to provide a voltage that is
added to the battery pack voltage to yield the desired output. The
assisting converter can also provide cell balancing. In comparison
with conventional systems this architecture drastically reduces the
power processing requirements of the step-up power stage and it
relaxes the requirements regarding converter power processing
efficiency. An implementation of this concept based on multi-input
isolated dual active bridge topology (DAB) has been demonstrated.
In comparison with a single step-up stage the multi-input DAB
allows operation at a higher switching frequency, allows
implementation with lower voltage rating low-cost components, and
provides better power processing efficiency.
* * * * *