U.S. patent application number 14/102974 was filed with the patent office on 2014-04-10 for filter, communication module, and communication apparatus.
This patent application is currently assigned to FUJITSU LIMITED. The applicant listed for this patent is Fujitsu Limited. Invention is credited to Xiaoyu Mi, Osamu Toyoda, Satoshi Ueda.
Application Number | 20140097915 14/102974 |
Document ID | / |
Family ID | 41694468 |
Filed Date | 2014-04-10 |
United States Patent
Application |
20140097915 |
Kind Code |
A1 |
Mi; Xiaoyu ; et al. |
April 10, 2014 |
FILTER, COMMUNICATION MODULE, AND COMMUNICATION APPARATUS
Abstract
A filter includes a substrate; a signal line formed on the
substrate and including an input terminal and an output terminal at
either end of the signal line; and a first pair of resonant lines
connected between the signal line and a ground portion, wherein the
first pair of resonant lines are connected to the signal line at
the same point.
Inventors: |
Mi; Xiaoyu; (Kawasaki,
JP) ; Ueda; Satoshi; (Kawasaki, JP) ; Toyoda;
Osamu; (Kawasaki, JP) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Fujitsu Limited |
Kawasaki-shi |
|
JP |
|
|
Assignee: |
FUJITSU LIMITED
Kawasaki-shi
JP
|
Family ID: |
41694468 |
Appl. No.: |
14/102974 |
Filed: |
December 11, 2013 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
12632098 |
Dec 7, 2009 |
|
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14102974 |
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Current U.S.
Class: |
333/205 |
Current CPC
Class: |
H03H 7/0123 20130101;
H03H 7/0153 20130101; H01P 3/082 20130101; H01P 1/2135 20130101;
H01P 1/20381 20130101; H01P 1/2136 20130101; H01P 1/203
20130101 |
Class at
Publication: |
333/205 |
International
Class: |
H01P 1/203 20060101
H01P001/203 |
Foreign Application Data
Date |
Code |
Application Number |
Dec 25, 2008 |
JP |
2008-329360 |
Claims
1. A filter comprising: a substrate; a signal line formed on the
substrate and including an input terminal and an output terminal at
either end of the signal line; a first pair of resonant lines
connected between the signal line and a ground portion, wherein the
first pair of resonant lines are connected to the signal line at a
same point; a second pair of resonant lines connected between the
signal line and the ground portion; a coupling section provided
between the first pair of resonant lines and the second pair of
resonant lines, wherein the coupling section includes: a first
circuit block connected to the signal line and including a first
terminal and a second terminal; a second circuit block connected
between the first terminal of the first circuit block and the
ground portion; and a third circuit block connected between the
second terminal of the first circuit block and the ground
portion.
2. The filter according to claim 1, wherein the coupling section
includes: a first circuit block and a second circuit block
connected in series to the signal line; and a third circuit block
connected between a point between the first circuit block and the
second circuit block, and the ground portion.
3. The filter according to claim 1, wherein the plurality of
resonant lines are connected to the same ground portion.
4. The filter according to claim 1, wherein the resonant lines are
formed into an arc shape.
5. The filter according to claim 1, further comprising: a variable
capacitor including a variable capacitor electrode provided above
the resonant line via an air gap and a drive electrode for changing
distance between the variable capacitor electrode and the resonant
line.
6. The filter according to claim 1, wherein the substrate is a
ceramic substrate including a plurality of laminated internal
wirings.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application is a divisional of U.S. application Ser.
No. 12/632,098 filed on Dec. 7, 2009, which is based upon and
claims the benefit of priority of the prior Japanese Patent
Application No. 2008-329360, filed on Dec. 25, 2008, the entire
contents of which are incorporated herein by reference.
FIELD
[0002] The embodiment discussed herein is related to a filter which
allows a predetermined frequency band signal to pass through.
BACKGROUND
[0003] In recent years, as well as the market of mobile
communication equipment such as a portable telephone growing, the
service has become increasingly sophisticated. Along with this, the
frequency band utilized by the communication network has shifted to
a high frequency band of 1 GHz or higher, and also, there is a
trend toward a multiple number of channels.
[0004] FIG. 1 is a circuit diagram illustrating a configuration of
a related high frequency variable filter. The high frequency
variable filter illustrated in FIG. 1 includes a plurality of
channel filters 101a to 101c, and switches 102a and 102b. The
passbands of the channel filters 101a to 101c differ from one
another. A high frequency signal input from an input terminal 103
is output from an output terminal 104 via one channel filter
selected by the switches 102a and 102b. By switching the switches
102a and 102b, it is possible to change the passband of the high
frequency variable filter.
[0005] For example, Japanese Unexamined Patent Publications JP-A
10-335903 and JP-A 2007-174438 disclose the heretofore described
kind of high frequency variable filter including the plurality of
channel filters and the switches.
[0006] However, the configuration illustrated in FIG. 1 includes a
number of filters equivalent to the number of channels. For this
reason, as well as the size of the high frequency variable filter
increasing, a cost also increases. Also, a loss of signal occurs in
each switch.
[0007] In recent years, attention has been drawn to a small
variable filter using an MEMS (Micro Electro Mechanical Systems)
switch and an variable capacitor. An MEMS device such as an MEMS
switch may be applied to a high frequency band variable filter with
a high Q (quality factor).
[0008] "D. Peroulis et al, "Tunable Lumped Components with
Applications to Reconfigurable MEMS Filters", 2001 IEEE MTT-S
Digest, p341-344", "E. Fourn et al, "MEMS Switchable Interdigital
Coplanar Filter", IEEE Trans. Microwave Theory Tech., vol. 51, NO.
1, p320-324, January 2003", and "A. A. Tamijani et al, "Miniature
and Tunable Filters Using MEMS Capacitors", IEEE Trans. Microwave
Theory Tech., vol. 51, NO. 7, p1878-1885, July 2003" disclose the
heretofore described kind of MEMS device.
[0009] The MEMS device, because of its small size and low loss, is
often used in a CPW distributed constant resonator (CPW: Coplanar
Waveguide).
[0010] "A. A. Tamijani et al, "Miniature and Tunable Filters Using
MEMS Capacitors", IEEE Trans. Microwave Theory Tech., vol. 51, NO.
7, p1878-1885, July 2003" discloses a filter with a structure in
which a plurality of MEMS variable capacitors straddle three
distributed constant lines. In this filter, by the variable
capacitors being displaced to change a gap between the variable
capacitors and distributed constant lines, it is possible to change
the capacitance. By changing the capacitance of the capacitors, it
is possible to change the passband of the filter.
[0011] In "A. A. Tamijani et al, "Miniature and Tunable Filters
Using MEMS Capacitors", IEEE Trans. Microwave Theory Tech., vol.
51, NO. 7, p1878-1885, July 2003", quartz and glass are used as
substrate materials. Also, the drive electrodes of the variable
capacitors are disposed in a gap between a ground line and signal
line formed on a substrate. Also, the length of the lines is
defined by the permittivity of the substrate.
[0012] In the heretofore known distributed constant filter, the
lower the frequency band, the larger the size. For example, the
usable frequency band of principal mobile communication equipment
such as a portable telephone is approximately 800 MHz to 6 GHz.
However, when the frequency band is 800 MHz to 6 GHz, as the
wavelength is long, the size of the distributed constant filter is
too large for practical use. For example, in the event that a
transmission line with an electrical length of .lamda./2 is
fabricated to be a 75.OMEGA. microstrip line working at 800 MHz by
using a ceramic substrate (permittivity .di-elect cons.=9.4), the
physical length being approximately 77 mm, it is difficult to put
the filter into compact handheld wireless communication usage.
[0013] By using a high dielectric substrate, it is possible to
shorten the length of the lines to some extent. However, when the
substrate permittivity becomes higher, it not being possible to
form a distributed constant line with a high characteristic
impedance, there will be no degree of freedom in a filter
configuration. For example, in the event that a microstrip line is
formed using a substrate whose permittivity .di-elect cons. is 80,
even though a distance between the signal line and ground is
increased to 600 .mu.m, a 50.OMEGA. (or other similar resistance)
signal line may only take up a width of 20 .mu.m. For this reason,
a transmission loss increases. Consequently, there is a limit to
reducing the filter size by increasing the substrate
permittivity.
SUMMARY
[0014] A filter includes a substrate, a signal line formed on the
substrate, including an input terminal and an output terminal at
either end of the signal line, and a first pair of resonant lines
connected between the signal line and a ground portion, wherein the
first pair of resonant lines are connected to the signal line at
the same point.
[0015] The object and advantages of the invention will be realized
and attained by means of the elements and combinations particularly
pointed out in the claims.
[0016] It is to be understood that both the foregoing general
description and the following detailed description are exemplary
and explanatory and are not restrictive of the invention, as
claimed.
BRIEF DESCRIPTION OF DRAWINGS
[0017] FIG. 1 is a circuit diagram of a configuration of a related
filter;
[0018] FIG. 2 is a circuit diagram of a filter in the
embodiment;
[0019] FIG. 3 is a circuit diagram of a configuration of a filter
in which a coupling circuit is connected;
[0020] FIGS. 4A to 4N are circuit diagrams of the coupling
circuit;
[0021] FIG. 5 is a circuit diagram illustrating another
configuration of the filter;
[0022] FIG. 6A is a plan view of a configuration wherein resonant
lines are connected to a common ground;
[0023] FIG. 6B is a plan view of a configuration wherein resonant
lines are formed into an arc shape;
[0024] FIG. 7A is a plan view of a resonant line of a variable
filter element taken along the line A-A in FIG. 6B;
[0025] FIG. 7B is a sectional view taken along Z-Z in FIG. 7A;
[0026] FIGS. 8A to 8G are sectional views illustrating a process of
manufacturing the variable filter element;
[0027] FIG. 9 is a block diagram of a configuration of a
communication module;
[0028] FIG. 10 is a block diagram of a configuration of a
communication module including variable filters;
[0029] FIG. 11 is a block diagram of a configuration of a
communication apparatus; and
[0030] FIG. 12 is a block diagram of a configuration of a
communication apparatus including variable filters.
DESCRIPTION OF EMBODIMENTS
1. Configuration of Filter
[0031] 1-1. Filter Including Pair of Resonant Lines
[0032] FIG. 2 is a circuit diagram illustrating a basic
configuration of a bandpass filter which is one example of a filter
of this embodiment. As illustrated in FIG. 2, with the bandpass
filter of the embodiment, an input line 2a is connected to an input
terminal 1. The input line 2a is connected to a contact point 3. A
resonant line 2b and resonant line 2c are connected between the
contact point 3 and ground. That is, the resonant lines 2b and 2c
are connected in parallel, at the same contact point, to a signal
line connecting the input terminal 1 and an output terminal 4. An
output line 2d is connected between the contact point 3 and output
terminal 4. Also, the input line 2a, resonant lines 2b and 2c, and
output line 2d are formed by a distributed constant transmission
line. In the embodiment, two resonant lines connected in parallel
at the same point are defined as a "pair of resonant lines".
[0033] The input-output impedance of the filter illustrated in FIG.
2 is taken to be, for example, 50.OMEGA.. Also, the impedance of
the resonant lines 2b and 2c, being lower than at least the
impedance of the input line 2a and output line 2d, is taken to be,
in the embodiment, 20.OMEGA. as an example.
[0034] The resonant lines 2b and 2c having a length of
(.lamda./8).times.n (n is a positive integer), in the filter
illustrated in FIG. 2, they have a length of .lamda./4 (that is,
n=2). .lamda. is a wavelength (a resonant wavelength), in the
distributed constant transmission line, of the frequency (the
resonant frequency of the resonant lines 2b and 2c) of a signal
extracted in the filter of the embodiment. Also, each of the
resonant lines 2b and 2c means a resonator of which one end is
connected to ground, and which is formed by the distributed
constant transmission line. By connecting one end of each of the
resonant lines 2b and 2c to ground in this way, signals input into
the resonant lines 2b and 2c via the input line 2a are filtered by
being totally reflected from the ground ends of the resonant lines
2b and 2c, enabling a desired frequency signal to be extracted from
the output terminal 4.
[0035] A description has been given of an example in which the
resonant lines 2b and 2c are connected to ground, but it is also
acceptable that they have an open end.
[0036] By connecting the resonant lines in parallel to the signal
line, and connecting the plurality of resonant lines in the same
position in the signal line, as heretofore described, it is
possible to shorten a line length between the input line 2a and
output line 2d in comparison with a filter in which a plurality of
resonators having a line length of .lamda./2 are connected in
series, as in a heretofore known technology, so it is possible to
reduce a size of the filter in a signal line direction.
[0037] Also, with the filter illustrated in FIG. 2, the pair of
resonant lines 2b and 2c are disposed in positions facing each
other across the signal line connecting the input terminal 1 and
output terminal 4. Because of this, it is possible to dispose the
resonant lines in high density, so it is possible to further reduce
the size of the filter in the signal line direction. Even in the
event that the resonant lines are disposed on the same side (one
side) with respect to the signal line, it is possible to reduce the
size of the filter in the signal line direction.
[0038] 1-2. Filter Including Plurality of Pairs of Resonant
Lines
[0039] FIG. 3 is a circuit diagram of a filter including a
plurality of pairs of resonant lines. With the filter illustrated
in FIG. 3, an input line 12a is connected to an input terminal 1.
The input line 12a is connected to a contact point 13a. A resonant
line 12b and resonant line 12c are connected between the connect
point 13a and ground. That is, the resonant lines 12b and 12c are
connected in parallel to the contact point 13a. Also, the resonant
lines 12b and 12c are connected to the same position (the contact
point 13a) in the signal line. Meanwhile, on the output side of the
filter, an output line 12g is connected between a contact point 13b
and output terminal 4. A resonant line 12e and resonant line 12f
are connected between the contact point 13b and ground. That is,
the resonant lines 12e and 12f are connected in parallel to the
contact point 13b. Also, the resonant lines 12e and 12f are
connected to the same position (the contact point 13b) in the
signal line. In the embodiment, a description has been given of an
example in which the resonant lines 12b, 12c, 12e, and 12f are
connected to ground, but it is also acceptable that they have an
open end.
[0040] A coupling circuit 14 is connected between the contact point
13a and contact point 13b. The coupling circuit 14 is a circuit
which couples the contact point 13a and contact point 13b in the
filter. The coupling circuit 14 may be realized by, for example, a
capacitor connected in series between the contact point 13a and
contact point 13b.
[0041] Also, in the filter illustrated in FIG. 3, it is preferable
that the impedance of the resonant lines 12b, 12c, 12e, and 12f is
lower than at least the input-output impedance of the filter. This
is for configuring in such a way that signals (currents) input from
the input terminal 1 flow into the resonant lines 12b, 12c, 12e,
and 12f. In the embodiment, as one example, the input-output
impedance of the filter is taken to be 50.OMEGA., and the impedance
of the resonant lines 12b, 12c, 12e, and 12f to be 20 .OMEGA..
[0042] In the filter illustrated in FIG. 3, the signals input into
the input terminal 1 are input into the resonant lines 12b and 12c
via the input line 12a. As one end of each of the resonant lines
12b and 12c is connected to ground, a signal which meets a resonant
condition of the resonant lines 12b and 12c is totally reflected
from the ground ends of the resonant lines 12b and 12c, but a
signal which does not meet the resonant condition, by being
grounded, or reflected to the input end side, is attenuated (a
filtering). The signal totally reflected from the ground ends of
the resonant lines 12b and 12c is input into the resonant lines 12e
and 12f via the coupling circuit 14. The filtering of the signal is
carried out in the resonant lines 12e and 12f, in the same way as
heretofore described, and the filtered signal is output from the
output terminal 4 via the output line 12g. As only a predetermined
frequency band signal is filtered in each resonant line, it is
possible to output the predetermined frequency band signal from the
output terminal 4.
[0043] Hereafter, a description will be given of a specific
configuration of the coupling circuit 14. The coupling circuit 14
may be realized by only the capacitor, as heretofore described, but
various other forms are also conceivable.
[0044] FIGS. 4A to 4C are diagrams illustrating representative
coupling circuits. The coupling circuit illustrated in FIG. 4A
illustrates a circuit of which the input side and output side are
connected by one circuit block. It is a circuit of which the input
side and output side are connected by, for example, a distributed
constant element 14a as the circuit block. The distributed constant
element 14a has an electrical length of .lamda./4. The impedance of
the distributed constant element 14a is approximate to the
input-output impedance (for example, about 50.OMEGA.) of the
filter, and higher than the impedance (for example, 20.OMEGA.) of
the resonant lines.
[0045] The coupling circuit illustrated in FIG. 4B is an example of
a .pi. type coupling circuit. With the .pi. type coupling circuit,
the input side and output side are connected by a circuit block
141, and both ends of the circuit block 141, and ground, are
connected by circuit blocks 142 and 143.
[0046] The coupling circuit illustrated in FIG. 4C is an example of
a T type coupling circuit. With the T type coupling circuit, the
input side and output side are connected by two circuit blocks 145
and 146, and a point between them, and ground, are connected by a
circuit block 147. These circuit blocks are realized by a
distributed constant element or lumped constant element. The
distributed constant element is, for example, a microstrip line,
and the lumped constant element is an inductor, a capacitor, or the
like. Also the circuit blocks are realized with the individual
element or the combination circuit thereof.
[0047] FIGS. 4D to 4L illustrate examples of the .pi. type coupling
circuit.
[0048] The coupling circuit illustrated in FIG. 4D includes a
capacitor C1 connected to the signal line connecting the input
terminal and output terminal, and two inductors L1 and L2 connected
between the signal line and ground.
[0049] The coupling circuit illustrated in FIG. 4E illustrates a
circuit in which the circuit block 141 includes a plurality of
elements. This coupling circuit includes two distributed constant
elements 14b and 14c, and a capacitor C11, which are connected in
series to the signal line, and two inductors L11 and L12 connected
between the signal line and ground.
[0050] The coupling circuit illustrated in FIG. 4F includes a
distributed constant element 14d connected to the signal line, and
two inductors L21 and L22 connected between the signal line and
ground.
[0051] The coupling circuit illustrated in FIG. 4G illustrates a
circuit in which each of the circuit blocks 141 to 143 includes
combining an inductor and capacitor connected in parallel. This
coupling circuit is a circuit in which a combination of an inductor
L31 and capacitor C31 connected in parallel is connected to the
signal line, and a combination of an inductor L32 and capacitor C32
connected in parallel, and a combination of an inductor L33 and
capacitor C33 connected in parallel, are connected between the
signal line and ground.
[0052] The coupling circuit illustrated in FIG. 4H is a circuit in
which a combination of an inductor L41 and capacitor C41 connected
in parallel is connected to the signal line, and two capacitors C42
and C43 are connected between the signal line and ground.
[0053] The coupling circuit illustrated in FIG. 4I is a circuit in
which a distributed constant element 14e is connected to the signal
line, and a combination of an inductor L51 and capacitor C51
connected in parallel, and a combination of an inductor L52 and
capacitor C52 connected in parallel, are connected between the
signal line and ground.
[0054] The coupling circuit illustrated in FIG. 4J illustrates a
circuit including a circuit block 141 in which an inductor and
capacitor connected in parallel are combined, and furthermore, a
plurality of the combinations are connected. This coupling circuit
is a circuit in which a combination of an inductor L60 and
capacitor C61 connected in parallel, and a combination of an
inductor L61 and capacitor C62 connected in parallel, are connected
in series to the signal line, and a combination of an inductor L62
and capacitor C63 connected in parallel, and a combination of an
inductor L63 and capacitor C64 connected in parallel, are connected
between the signal line and ground.
[0055] The coupling circuit illustrated in FIG. 4K is a circuit in
which the capacitor C61 and the combination of the inductor L61 and
capacitor C62 connected in parallel are connected in series to the
signal line, and the combination of the inductor L62 and capacitor
C63 connected in parallel, and the combination of the inductor L63
and capacitor C64 connected in parallel, are connected between the
signal line and ground.
[0056] The coupling circuit illustrated in FIG. 4L is a circuit in
which an inductor L71 and a combination of an inductor L72 and
capacitor C71 connected in parallel are connected to the signal
line, and a combination of an inductor L73 and capacitor C72
connected in parallel, and a combination of an inductor L74 and
capacitor C73 connected in parallel, are connected between the
signal line and ground.
[0057] FIGS. 4M and 4N are examples of the T type coupling circuit.
FIG. 4M illustrates a circuit in which a combination of a capacitor
C91 and inductor L91 connected in parallel, and a combination of a
capacitor C92 and inductor L92 connected in parallel, are connected
in series to the signal line connecting the input terminal and
output terminal, and a combination of a capacitor C93 and inductor
L93 connected in parallel is connected between the signal line and
ground. The coupling circuit illustrated in FIG. 4N is a circuit
which includes distributed constant elements 14f and 14g connected
in series to the signal line, and a distributed constant element
14h connected between the signal line and ground.
[0058] With a coupling circuit which, being of the .pi. type or T
type, includes lumped elements, as illustrated in FIGS. 4E, 4F, and
4I, it is possible to connect a distributed element to one
portion.
[0059] Also, the .pi. type coupling circuit having two contact
points on the signal line connected between the input terminal and
output terminal, a lumped constant element or distributed constant
element is connected between each contact point and the ground. It
is preferable that the elements connected between the contact
points and ground have a symmetry. For example, it is preferable
that the same element is installed in an a1 section and a2 section
in FIG. 4D. In the event of a configuration wherein the same
element is installed in the a1 section and a2 section, it is
acceptable to install inductors, as illustrated in FIG. 4D or the
like, it is acceptable to install capacitors, as illustrated in
FIG. 4H, and it is acceptable to install a plurality of lumped
constant elements, as illustrated in FIG. 4G or the like.
[0060] As illustrated in FIG. 3, by connecting the resonant lines
in parallel to the signal line, and connecting the plurality of
resonant lines in the same position in the signal line, it is
possible to shorten a line length between the input line 12a and
output line 12d, in comparison with a filter in which a plurality
of resonators having a line length of .lamda./2 are connected in
series, as in a heretofore known technology, so it is possible to
reduce the size of the filter in the signal line direction.
[0061] Also, with the filter illustrated in FIG. 3, the paired
resonant lines 12b and 12c, and the paired resonant lines 12e and
12f, are each disposed in positions facing each other across the
signal line connecting the input terminal 1 and output terminal 4.
By means of this kind of configuration, it is possible to dispose
the lines in high density, so it is possible to further reduce the
size of the filter in the signal line direction.
[0062] With the filter illustrated in FIG. 3, the electrical length
of the resonant lines 12b, 12c, 12e, and 12f is .lamda./4, but it
is possible to make it (.lamda./8).times.n (.lamda. is a resonant
wavelength in the resonator, and n is a positive integer). Even in
the event of using resonant lines whose electrical length is, for
example, .lamda./8 (that is, n=1), it is possible to obtain the
same advantages. In the event of connecting the resonant lines
whose line length is .lamda./8, one end of each resonant line 12b,
12c, 12e, and 12f is connected to ground, as illustrated in FIG. 3.
By connecting the resonant lines 12b, 12c, 12e, and 12f whose line
length is .lamda./8 in this way, signals input from the input
terminal 1 are input into the resonant line 12b via the contact
point 13a, and only a signal which meets the resonant condition of
the resonant line 12b is totally reflected from the ground end,
while a signal which does not meet the resonant condition, by being
grounded, or reflected to the input side, is attenuated. The signal
totally reflected from the ground end of the resonant line 12b has
a phase difference of .lamda./2 from the signals input into the
contact point 13a from the input terminal 1, and interferes
therewith. Because of this, the resonant lines 12b and 12c function
in combination as one resonator. The resonated signal, after being
resonated again in the same way as heretofore described in the
resonant lines 12e and 12f, via the coupling circuit 14, is output
from the output terminal 4 via the output line 12g. By this means,
it is possible to output a desired frequency band signal from the
output terminal 4.
[0063] A signal, among the signals input into the input terminal 1,
which has a wavelength which does not meet the resonant condition
of a resonant line, is attenuated by being grounded, or reflected
to the input end side, and is prevented from being output from the
output terminal 4. By so doing, the filter performs its
function.
[0064] With the filter illustrated in FIG. 3, the pair of resonant
lines on the input side and the pair of resonant lines on the
output side are connected by the one coupling circuit 14. The
number of pairs of resonant lines included in the filter not being
limited to two, it is possible to connect three or a larger number.
A connecting of a much larger number of paired resonant lines may
be realized by connecting them by means of a coupling circuit which
couples a signal between one pair of resonant lines and another
pair of resonant lines, and repeating this kind of connection
structure. By this means, the number of resonators included in the
whole of the filter increasing, it is possible to realize a filter
with a good steepness.
[0065] FIG. 5 is a circuit diagram illustrating another
configuration example of a filter. With the filter illustrated in
FIG. 5, an input line 22a is connected between an input terminal 1
and a contact point 23a. A resonant line 22b is connected between
the contact point 23a and ground. A first coupling circuit 24 is
connected between the contact point 23a and a contact point 23b. A
resonant line 22c and resonant line 22d are connected between the
contact point 23b and ground. A second coupling circuit 25 is
connected between the contact point 23b and a contact point 23c. A
resonant line 22e is connected between the contact point 23c and
ground. An output line 22f is connected between the contact point
23c and an output terminal 4.
[0066] As the first coupling circuit 24 and second coupling circuit
25, it is possible to use the .pi. type coupling circuit or T type
coupling circuit. Also, as the .pi. type coupling circuit or T type
coupling circuit, it is possible to employ one of the coupling
circuits illustrated in FIGS. 4A to 4N. Also, with a coupling
circuit which is connected in the .pi. type or T type, includes a
lumped constant element, as illustrated in FIGS. 4E, 4F, and 4I, it
is possible to connect a distributed constant element to one
portion.
[0067] In this way, by connecting the resonant lines 22c and 22d in
parallel in the same portion in the signal line, it being possible
to shorten the line length of the signal line connecting the input
terminal 1 and output terminal 4, it is possible to reduce the size
of the filter in the signal line direction.
[0068] Also, with the filter illustrated in FIG. 5, a description
has been given of an example in which the resonant lines 22c and
22e are connected to ground, but it is possible to make the
terminals thereof connected to ground an open end. In this case, as
it is possible to attenuate a filter passband low frequency side
signal level in the resonant line 22b, and attenuate a filter
passband high frequency side signal level in the resonant line 22e,
it is possible to improve the steepness of the passband
characteristics of the filter.
[0069] With the filter illustrated in FIG. 5, the input side line
and output side line are connected by the first coupling circuit 24
and second coupling circuit 25 but, by increasing the number of
coupling circuits, it is possible to connect still more paired
resonant lines connected between the signal line and ground.
[0070] 1-3. Ground Sharing of Resonant Lines
[0071] It is acceptable that the ground ends of the resonant lines
12b, 12c, 12e, and 12f illustrated in FIGS. 3 and 5 are connected
one to each independent ground, and it is also acceptable that they
are connected to an identical ground.
[0072] FIG. 6A is a plan view illustrating a specific configuration
of a filter. In the filter illustrated in FIG. 6A, the circuit
illustrated in FIG. 4F is employed as the coupling circuit 14. In
the components illustrated in FIG. 6A, components identical to the
components illustrated in FIGS. 3 and 4F are given identical
reference numerals and characters. Also, in FIG. 6A, resonant lines
12b', 12c', 12e', and 12f' connected one to each independent ground
are depicted by the broken lines, while resonant lines 12b, 12c,
12e, and 12f connected to an identical ground are depicted by the
solid lines. The line lengths of the resonant lines 12b, 12c, 12e,
12f, 12b', 12c', 12e', and 12f' are all taken to be W1. Also, the
resonant lines 12b, 12c, 12e, and 12f, and a distributed constant
element 14d, are each constructed in such a way that a plurality of
capacitor electrodes straddle a signal line (a description of the
capacitor electrodes will be given hereafter). Also, the resonant
lines 12b and 12e are connected to an identical ground G1. Also,
the resonant lines 12c and 12f are connected to an identical ground
G2. Also, the resonant line 12b' is connected to a ground G1'.
Also, the resonant line 12c' is connected to a ground G2'. Also,
the resonant line 12e' is connected to a ground G3'. Also, the
resonant line 12f' is connected to a ground G4'. The grounds G1' to
G4' are physically independent of one another.
[0073] As illustrated by the broken lines in FIG. 6A, in the event
that the resonant lines 12b' and 12e' are disposed so as to be
perpendicular to the signal line of which both ends are connected
to input and output terminals, and connected to the mutually
independent grounds G1' and G3', a disposition space of a dimension
W2 (W2=W1) is needed in a direction perpendicular to the signal
line. In the same way, in the event that the resonant lines 12c'
and 12f' are disposed so as to be perpendicular to the signal line
of which both ends are connected to the input and output terminals,
and connected to the mutually independent grounds G2' and G4', a
disposition space of a dimension the same as the dimension W2 is
needed in the direction perpendicular to the signal line.
Consequently, the size of the filter in the vertical direction (the
direction perpendicular to the signal line) is W4 which is
approximately twice W2.
[0074] As opposed to this, as illustrated by the solid lines in
FIG. 6A, by connecting the resonant line 12b and resonant line 12e
to the ground G1, and disposing them at an angle with respect to
the signal line, it is sufficient that a space for disposing the
resonant lines 12b and 12e is of a dimension W3 (W3<W2) in the
direction perpendicular to the signal line. In the same way, by
connecting the resonant line 12c and 12f to the ground G2, and
disposing them at an angle with respect to the signal line, it is
sufficient that a space for disposing the resonant lines 12c and
12f is of a dimension the same as the dimension W3 in the direction
perpendicular to the signal line. Consequently, it is possible to
make a dimension W5 (W5=W3.times.2) of the filter in the vertical
direction smaller than a dimension W4. In this way, by connecting a
plurality of resonant lines to ground at the same contact point, it
is possible to reduce a space in which the resonant lines are
disposed.
[0075] FIG. 6B illustrates another example of a filter in which
resonant lines are connected to an identical ground. The filter
illustrated in FIG. 6B differs from the filter illustrated in FIG.
6A in that the resonant lines 12b, 12c, 12e, and 12f are formed
into an arc shape. A line length W11 of each resonant line 12b,
12c, 12e, and 12f is the same as the line length W1 of the resonant
lines illustrated in FIG. 6A.
[0076] In this way, by forming the resonant lines 12b, 12c, 12e,
and 12f into the arc shape, it is possible to make a dimension W12
of a resonant line disposition space in the direction perpendicular
to the signal line smaller than the dimension W2 illustrated in
FIG. 6A. Consequently, it is possible to make a dimension W13 of
the filter in the vertical direction (the direction perpendicular
to the signal line) smaller than the dimension W4 illustrated in
FIG. 6A.
[0077] Also, it is possible to make the dimension W12 of the
resonant line disposition space in the direction perpendicular to
the signal line much smaller than the dimension W3 illustrated in
FIG. 6A. Consequently, it is possible to make the dimension W13 of
the filter in the vertical direction (the direction perpendicular
to the signal line) much smaller than the dimension W5 illustrated
in FIG. 6A.
[0078] The heretofore described filter configuration which enables
miniaturization is also advantageous for a loss reduction. A loss
of the filter basically depends on a line conductor loss. By
miniaturizing the filter, it being possible to shorten the line
length of the filter, it is possible to reduce a signal passing
loss.
[0079] Also, by miniaturizing the filter, it being possible to
increase the number (an available number) of filters which may be
fabricated from one wafer at a time of filter manufacture, it is
possible to reduce a cost per element.
[0080] The filter according to the embodiment may be used as, for
example, a small GHz band frequency variable filter using an MEMS
variable capacitor.
2. Configuration of Variable Filter
[0081] With the capacitance of the resonant lines 12b, 12c, 12e,
and 12f illustrated in FIG. 3, as a capacitance between them and
grounds (to be described hereafter) disposed in a substrate is of a
fixed value, the passband of the filter illustrated in FIG. 3 is
fixed. As opposed to this, by mounting a movable capacitor
electrode (to be described hereafter) on the resonant lines 12b,
12c, 12e, and 12f, and coupling circuit 14, illustrated in FIG. 3,
it is possible to realize a variable filter which may vary the
passband. Also, by mounting a movable capacitor electrode on the
resonant lines 22b, 22c, 22d, and 22e, and coupling circuits 24 and
25 in FIG. 5, it is possible to realize a variable filter which may
vary the passband. By mounting the movable capacitor electrode on
the resonant lines, it is possible to shorten the line length and,
as well as it being possible to further miniaturize the filter, it
is possible to vary the passband. Also, by mounting the movable
capacitor electrode on the coupling circuit, it is possible to
equivalently change the electrical length of the resonator in such
a way as to provide a coupling circuit in accordance with the
passband varied in the resonant lines. A description has been given
of an example in which these variable filters have the variable
capacitors, but it is also acceptable to realize them with variable
inductors. Furthermore, it is acceptable to realize the variable
filters by appropriately combining the variable capacitors and
variable inductors.
[0082] In the event that a coupling circuit including only lumped
constant elements is installed in the variable filter, as
illustrated in FIGS. 4D, 4G, 4H, 4J, 4K, 4L, and 4M, it is
sufficient to change at least one lumped constant element, among
the lumped constant elements included in the coupling circuit, to a
variable element. For example, in the event that the coupling
circuit illustrated in FIG. 4D is installed, by changing the
capacitor C1 to a variable capacitor, it is possible to realize the
coupling circuit in accordance with the passband.
[0083] Also, in the event that a coupling circuit including a
lumped constant element and distributed constant element is
installed in the variable filter, as illustrated in FIGS. 4E, 4F,
and 4I, by installing a movable capacitor electrode as the lumped
constant element included in the coupling circuit, it is possible
to realize the coupling circuit in accordance with the
passband.
[0084] Also, in the event that a coupling circuit including a
plurality of distributed constant elements is installed in the
filter, as illustrated in FIG. 4N, by installing a movable
capacitor electrode as at least one distributed constant element,
among the distributed constant elements 14f, 14g, and 14h included
in the coupling circuit, it is possible to realize the variable
filter.
[0085] By installing the variable capacitor electrode in the
resonant lines, as in the embodiment, it being possible to change
the capacitance in the resonant lines, it is possible to change a
signal passband in the resonant lines. By installing the resonant
lines, in which the passband is variable, in the filter in this
way, it is possible to realize the variable filter.
[0086] Hereafter, a description will be given of a specific
configuration of the resonant lines including the movable capacitor
electrode (hereafter referred to as the variable capacitor
element).
[0087] 2-1. Configuration of Variable Capacitor Element
[0088] FIG. 7A is a plan view of the variable capacitor element.
FIG. 7B is a sectional view of a Z-Z section in FIG. 7A.
[0089] The variable capacitor element illustrated in FIGS. 7A and
7B, including a substrate 31, a signal line 32, movable capacitor
electrodes 33, drive electrodes 35a and 35b, a dielectric dot 36,
anchor sections 37a and 37b, electrode pads 38, and a packaging
member 39, is configured as one portion of a filter which allows a
passage of an electromagnetic wave or electrical signal in a
specified high frequency band. The packaging member 39 seals not
only a variable capacitor element section, but the whole of the
filter.
[0090] The substrate 31 is an LTCC wafer (LTCC: Low Temperature
Co-fired Ceramics) including multilayer internal wirings (wiring
patterns 31c). The substrate 31 is formed by mutually bonding a
plurality (five in the substrate illustrated in FIG. 7B) of
insulating layers 31a. A via 31b which includes a conductive
portion in a through hole formed from one principal surface to the
other principal surface is formed in each insulating layer 31a.
Also, each wiring pattern 31c is sandwiched between at least one
pair of adjacent insulating layers 31a. Also, one portion of the
wiring pattern 31c positioned on a side of the substrate 31 closest
to a first surface 31e is a ground line 31d connected to ground.
The ground line 31d faces the signal line 32 across the insulating
layer 31a, and the ground line 31d and signal line 32 have a gap
CG2 between them. Regarding the ground line 31d illustrated in FIG.
7B, an example has been described in which it is disposed in a
position close to the first surface but, it not being limited to
this, it is also acceptable to dispose it on another layer. In this
case, the ground line 31d faces the signal line 32 across a
plurality of the insulating layers 31a. For this reason, the gap
CG2 between the ground line 31d and signal line 32 is equivalent to
a thickness to which the plurality of insulating layers 31a are
stacked. Also, the wiring patterns 31c are connected, and the
wiring patterns 31c and electrode pads 38 are connected, by the
vias 31b. In some cases, it is acceptable that the wiring patterns
31c and signal line 32 are connected by the vias 31b. Also, the
insulating layers 31a are realized by an LTCC. However, their not
being limited to the LTCC, it is acceptable to form them from
another dielectric body.
[0091] The signal line 32, as illustrated in FIG. 7A, including a
terminal 32a and terminal 32b at both ends in its longitudinal
direction, is a conductor pattern in which an electrical signal
passes between the terminals 32a and 32b. The terminals 32a and
32b, by being connected to other elements on the wiring substrate,
or made an open end, are electrically connected to predetermined
electrode pads 38 via predetermined vias 31b and wiring patterns
31c in the wiring substrate 31 (not illustrated). Also, the signal
line 32, being a distributed constant transmission line of which
the impedance is, for example, 20.OMEGA., is formed from a low
resistance metal material such as, for example, Cu, Ag, Au, Al, W,
or Mo. Also, the thickness of the signal line 32 is, for example,
0.5 to 20 .mu.m.
[0092] Both ends of each movable capacitor electrode 33 are fixed
to the anchor sections 37a and 37b formed on the first surface 31e
of the substrate 31, and a main portion thereof excluding both ends
faces the signal line 32 and drive electrodes 35a and 35b across an
air gap. A thick section 33a is formed in a portion of each movable
capacitor electrode 33 facing the signal line 32. The thick
sections 33a and signal line 32 face each other across a gap CG1.
The movable capacitor electrodes 33 are connected to ground via the
anchor sections 37a and 37b, vias 31b, and wiring patterns 31c. The
movable capacitor electrodes 33, being formed from an elastically
deformable material, may be formed from, for example, a low
resistance metal such as, for example, Au, Cu, or Al. The variable
capacitor element whose capacitance changes is realized by the
movable capacitor electrodes 33 being moved to change a distance
between the movable capacitor electrodes 33 and signal line 32.
Also, the gap CG1 between the movable capacitor electrodes 33 and
signal line 32 may be made, for example, 0.1 to 10 .mu.m. Also, the
movable capacitor electrodes 33 and ground line 31d are one example
of a ground wiring section in the embodiment.
[0093] The drive electrodes 35a and 35b, being disposed adjacent to
the signal line 32, face one portion of each movable capacitor
electrode 33. The drive electrodes 35a and 35b generate an
electrostatic attractive force between themselves and the movable
capacitor electrodes 33, enabling the movable capacitor electrodes
33 to be displaced in a direction indicated by an arrow A. By the
movable capacitor electrodes 33 being displaced by the action of
the drive electrodes 35a and 35b, a capacitance between the signal
line 32 and movable capacitor electrodes 33 changes. The drive
electrodes 35a and 35b are formed from a high resistance metal thin
film such as, for example, a SiCr thin film. Also, in order to
suppress an occurrence of a pull-in phenomenon, it is preferable
that a gap between the drive electrodes 35a and 35b and movable
capacitor electrodes 33 is made equal to or more than three times
the gap CG1 between the movable capacitor electrodes 33 and signal
line 32.
[0094] The dielectric dot 36, being provided on the signal line 32,
is formed from a dielectric material such as, for example, Al2O3,
SiO2, SixNy, or SiOC. The dielectric dot 36, as well as being able
to prevent the signal line 32 and movable capacitor electrodes 33
from short circuiting, may increase a capacitance occurring in the
gap CG1 between the signal line 32 and movable capacitor electrodes
33. It is preferable to increase the capacitance because it is
thereby possible to ensure a wide filter frequency variable
range.
[0095] The packaging member 39 seals structures of the filter
which, being bonded to the first surface 31e of the substrate 31,
are formed on the first surface 31e of the substrate 31.
[0096] In the variable capacitor element illustrated in FIGS. 7A
and 7B, a first capacitor is formed by the gap CG2 being formed
between the signal line 32 and the ground line 31d disposed in the
substrate 31. Also, a second capacitor is formed by the gap CG1
being formed between the signal line 32 and movable capacitor
electrodes 33. By forming two capacitors in this way, it is
possible to increase the capacitance. Consequently, with these
capacitors, it is possible to increase the capacitance in
comparison with a microstrip line or distributed constant element
which includes only the first capacitor, as heretofore known. That
is, with these capacitors, it is possible to increase the
capacitance in comparison with a microstrip line or distributed
constant element which includes no movable capacitor electrode. By
increasing the capacitance, it is possible to shorten the physical
signal line length of a resonant line including a variable
distributed constant element. Therefore, by installing this kind of
variable capacitor element in the filter, it being possible to
shorten the line length of the resonant line, it is possible to
miniaturize the filter.
[0097] Also, by applying a voltage to the drive electrodes 35a and
35b via the electrode pads 38, vias 31b and wiring patterns 31c, it
is possible to generate an electrostatic attractive force between
the drive electrodes 35a and 35b and movable capacitor electrodes
33, and elastically displace the movable capacitor electrodes 33 in
the direction indicated by the arrow A. By displacing the movable
capacitor electrodes 33, it is possible to reduce the gap CG1
between the signal line 32 and movable capacitor electrodes 33. By
reducing the gap CG1, it is possible to increase the capacitance in
the second capacitor. By increasing the capacitance, the line
length of the distributed constant element increases equivalently
or essentially, and a resonated frequency band is shifted to a low
frequency side.
[0098] Also, the drive electrodes 35a and 35b, as well as being
divided for each movable capacitor electrode 33, are configured so
that a voltage may be applied to each individual one. Then, by
selectively applying a voltage to the divided drive electrodes 35a
and 35b, the plurality of movable capacitor electrodes 33 are
selectively displaced. The movable capacitor electrodes 33 are
selectively displaced, thereby enabling changes in capacitance to
differ in magnitude.
[0099] Also, as the electrostatic attractive force occurring
between the drive electrodes 35a and 35b and movable capacitor
electrodes 33 is diminished by decreasing the voltage applied to
the drive electrodes 35a and 35b, a displacement amount of the
movable capacitor electrodes 33 decreases, enabling the movable
capacitor electrodes 33 to return in a direction indicated by an
arrow B. By returning the movable capacitor electrodes 33 in the
direction indicated by the arrow B, the gap CG1 between the signal
line 32 and movable capacitor electrodes 33 increases, and the
capacitance in the second capacitor decreases. By the capacitance
decreasing, the electrical length of the distributed constant
element decreases equivalently or essentially.
[0100] In this way, by adjusting the voltage applied to the drive
electrodes 35a and 35b, and displacing the movable capacitor
electrodes 33 in a direction approaching the signal line 32, it is
possible to make the second capacitor a variable capacitor, and it
is possible to change a signal passing frequency band in the
variable filter element. It is possible to realize the variable
filter by installing this kind of variable capacitor element in,
for example, the resonant lines 12b, 12c, 12e, and 12f, and
coupling circuit 14 illustrated in FIG. 3, or the resonant lines
22b, 22c, 22d, and 22e, and coupling circuit 24 illustrated in FIG.
5.
[0101] Also, with a commonly known CPW signal line, a signal line
(one being an example) and ground lines (for example, two) being
provided on the same surface of a substrate, as a drive electrode
for driving a movable capacitor electrode is disposed between the
signal line and the ground lines, there is a limitation on a drive
electrode disposition space, and there is a limit to increasing the
area of the drive electrode. As opposed to this, as the variable
filter element using the microstrip line, illustrated in FIGS. 7A
and 7B has no ground line provided on a surface of the substrate
the same as the surface on which the signal line is formed, it is
possible to secure a large area of the drive electrodes 35a and 35b
on the substrate 31. By securing the large area of the drive
electrodes 35a and 35b, it being possible to reduce the voltage
applied to the drive electrodes 35a and 35b when displacing the
movable capacitor electrode 33, it is possible to secure a wide
movable range of the movable capacitor electrodes 33. Also, by
reducing the drive voltage, it is possible to reduce a power
consumption.
[0102] Also, it is conceivable that, by increasing the area of the
drive electrodes 35a and 35b, it is possible to suppress a
self-actuation phenomenon due to a high frequency signal. That is,
as it is possible, by increasing the area of the drive electrodes
35a and 35b, to increase the electrostatic attractive force
occurring between the drive electrodes 35a and 35b and movable
capacitor electrodes 33, it is possible to form the movable
capacitor electrodes 33 from an elastic body with a high rigidity.
Furthermore, the higher the area ratio of the drive electrodes 35a
and 35b and a capacitor section CAP, a coulomb force occurring
between the signal line 32 and movable capacitor electrodes 33 due
to a high frequency signal passing through the capacitor section
CAP becomes negligible compared with a coulomb force occurring
between the drive electrodes 35a and 35b and movable capacitor
electrodes 33 due to the drive voltage. Consequently, in the
embodiment, it is conceivable that the increase in the area of the
drive electrodes 35a and 35b is advantageous for a suppression of
the self-actuation phenomenon of a parallel plate type variable
capacitor.
[0103] 2-2. Method of Manufacturing Variable Filter Element
[0104] FIGS. 8A to 8G are sectional views illustrating a process of
manufacturing the variable filter element.
[0105] Firstly, as illustrated in FIG. 8A, the electrode pads 38
are formed on a second surface 31f of the substrate 31 including
the multilayer internal wirings. The electrode pads 38 may be
formed by, for example, after forming a predetermined metal
material as a film on the second surface 31f of the substrate 31 by
means of a sputtering method, patterning the metal film by means of
a predetermined wet etching or dry etching. Alternatively, in the
formation of the electrode pads 38, it is possible to employ a
nonelectrolytic plating method or electroplating method. Next, the
drive electrodes 35a and 35b are formed on the first surface 31e of
the substrate 31. The drive electrodes 35a and 35b may be formed
by, for example, after forming a predetermined metal material as a
film on the substrate 31 by means of a sputtering method,
patterning the metal film by means of a predetermined wet etching
or dry etching. It is also acceptable that, after the process of
forming the drive electrodes 35a and 35b, a process of forming an
insulating film is implemented in such a way as to cover the drive
electrodes 35a and 35b. Next, the signal line 32 and anchor
sections 37a and 37b are formed on the first surface 31e of the
substrate 31. The signal line 32 may be formed by, for example,
after forming a resist pattern, which has openings corresponding to
the signal line 32 and anchor sections 37a and 37b, on the
substrate 31 by means of a patterning, depositing a predetermined
metal material (for example, Au), and causing it to grow, in the
openings by means of a plating method (a nonelectrolytic plating or
electroplating).
[0106] Next, as illustrated in FIG. 8B, the dielectric dot 36 is
formed on the signal line 32. The dielectric dot 36 may be formed
by, for example, after forming a predetermined dielectric film on
the first surface 31e side of the substrate 31, patterning the
dielectric film.
[0107] Next, as illustrated in FIG. 8C, a sacrifice layer 40 is
formed. The sacrifice layer 40 is formed from a material which,
being easy to remove, may be selectively etched.
[0108] Next, as illustrated in FIG. 8D, the movable capacitor
electrode 33 is formed on the sacrifice layer 40. The movable
capacitor electrode 33 may be formed by, for example, after forming
a predetermined metal material as a film on the sacrifice layer 40
by means of a sputtering method, patterning the metal film by means
of a predetermined wet etching or dry etching. Alternatively, the
movable capacitor electrode 33 may be formed by employing a
nonelectrolytic plating method or electroplating method
[0109] Next, as illustrated in FIG. 8E, the thick section 33a
forming one portion of the movable capacitor electrode 33 is
formed. The thick section 33a may be formed by, for example, after
forming a resist pattern, which has an opening corresponding to the
thick section 33a, over the movable capacitor electrode 33 and
sacrifice layer 40 by means of a patterning, depositing a
predetermined metal material (for example, Au), and causing it to
grow, in the opening by means of a plating method (a
nonelectrolytic plating or electroplating).
[0110] Next, as illustrated in FIG. 8F, the sacrifice layer 40 is
removed. By this means, it is possible to form an air gap between
the movable capacitor electrode 33 and the signal line 32, drive
electrodes 35a and 35b, and dielectric dot 36.
[0111] Next, as illustrated in FIG. 8G, the packaging member 39 is
bonded to the first surface 31e side of the substrate 31. As a
method of bonding the packing member 39 to the substrate 31, it is
possible to propose, for example, an anodic bonding method, a
direct bonding method, a room temperature bonding method, and a
eutectic bonding method. The packaging member 39 being one
fabricated by processing an LTCC, a concavity 39a is provided in
advance in a portion corresponding to each variable filter
formation section of the substrate 31.
[0112] Next, the substrate 31 and packaging member 39 are cut into
individual variable filters.
[0113] By the above means, the variable filter is completed.
[0114] The LTCC has been used for the packaging member, but it is
also possible to use a dielectric body, such as a resin or ceramic,
or a high resistance silicon.
3. Configuration of Communication Module
[0115] FIG. 9 illustrates an example of a communication module
including the bandpass filter of the embodiment. As illustrated in
FIG. 9, a duplexer 62 includes a reception filter 62a and
transmission filter 62b. Also, for example, receiving terminals 63a
and 63b corresponding to balance outputs are connected to the
reception filter 62a. Also, the transmission filter 62b is
connected to a transmitting terminal 65 via a power amplifier 64.
Herein, the bandpass filter of the embodiment is included in the
reception filter 62a and transmission filter 62b.
[0116] When carrying out a receiving operation, the reception
filter 62a allows only a predetermined frequency band signal, among
received signals input via an antenna terminal 61, to pass through,
and outputs it from the receiving terminals 63a and 63b to the
exterior. Also, when carrying out a transmitting operation, the
transmission filter 62b allows only a predetermined frequency band
signal, among transmission signals input from the transmitting
terminal 65 and amplified by the power amplifier 64, to pass
through, and outputs it from the antenna terminal 61 to the
exterior.
[0117] Also, FIG. 10 illustrates a communication module which
includes a variable reception filter 66a in place of the reception
filter 62a in the communication module illustrated in FIG. 9, and a
variable transmission filter 66b in place of the transmission
filter 62b. The variable reception filter 66a and variable
transmission filter 66b include the variable filter described in
the section "2. Configuration of Variable Filter" in the present
specification. With a reception filter and transmission filter
which cannot vary the passband, in the event of attempting to
realize a multiband compatible communication module which may
transmit and receive a plurality of high frequency signals in
differing frequency bands, the communication module includes
reception filters and transmission filters corresponding to the
frequency bands, and a switch circuit which switches between the
filters for each of the frequency bands in which to transmit and
receive the signals, meaning that the communication module is
larger in size. As opposed to this, according to the communication
module illustrated in FIG. 10, by its including one variable
reception filter 66a and one variable transmission filter 66b, it
is possible to reduce the number of filters, and it is possible to
downsize the multiband compatible communication module.
[0118] By including the passband filter of the embodiment in the
reception filter 62a and transmission filter 62b of the
communication module, as heretofore described, it is possible to
downsize the communication module. That is, with a heretofore known
filter, as a configuration has been employed wherein a plurality of
resonant lines are connected in series, the size in the signal line
direction has been increased but, in the embodiment, as a
configuration is employed wherein the plurality of resonant lines
are connected in parallel in the same position, it is possible to
reduce the size of the filter in the signal line direction.
Consequently, by mounting the miniaturized filter, it is possible
to downsize the communication module. In particular, with a
communication module which carries out a high frequency band
communication, the number of filters becomes larger, by mounting a
filter of which the size in the signal line direction is small, as
in the embodiment, it is possible to downsize the communication
module. In particular, as the number of filters becomes larger with
the communication module which carries out the high frequency band
communication, by mounting a filter of which the size in the signal
line direction is small, as in the embodiment, it is possible to
downsize a communication module compatible with the high frequency
band communication.
[0119] Also, as it is possible to reduce the passing loss by
miniaturizing the filter, it is possible to realize a communication
module with superior communication characteristics.
[0120] The configurations of the communication modules illustrated
in FIGS. 9 and 10 being one example, it is also possible to obtain
the same advantages when mounting the passband filter of the
embodiment in a communication module of another form.
4. Configuration of Communication Apparatus
[0121] FIG. 11 illustrates an RF block of a portable telephone
terminal as one example of a communication apparatus including the
passband filter or communication module of the embodiment. Also,
the communication apparatus illustrated in FIG. 11 is illustrated
as one example of a portable telephone terminal compatible with a
GSM (Global System for Mobile Communications) communication system
and W-CDMA (Wideband Code Division Multiple Access) communication
system. Also, the GSM communication system in the embodiment is
compatible with a 850 MHz band, 950 MHz band, 1.8 GHz band, and 1.9
GHz band. Also, although the portable telephone terminal includes a
microphone, a speaker, a liquid crystal display, and the like,
apart from the configuration illustrated in FIG. 11, as they are
unnecessary in a description in the embodiment, an illustration
thereof is omitted. Herein, the bandpass filter in the embodiment
is included in reception filters 73a, 77, 78, 79, and 80, and a
transmission filter 73b.
[0122] Firstly, an LSI to be operated is selected by an antenna
switch circuit 72 depending on whether a communication system
compatible with a received signal input via an antenna 71 is of the
W-CDMA or GSM. In the event that the received signal input is
compatible with the W-CDMA communication system, the received
signal is switched in such a way as to be output to a duplexer 73.
The received signal input into the duplexer 73 is limited to a
predetermined frequency band by the reception filter 73a, and a
balanced type received signal is output to an LNA 74. The LNA 74
amplifies the input received signal, and outputs it to an LSI 76.
The LSI 76, based on the input received signal, carries out a
process of demodulation into a sound signal, and controls the drive
of each section in the portable telephone terminal.
[0123] Meanwhile, when transmitting signals, the LSI 76 generates
transmission signals. The generated transmission signals are
amplified by a power amplifier 75, and input into the transmission
filter 73b. The transmission filter 73b causes only a predetermined
frequency band signal, among the input transmission signals, to
pass through. The transmission signal output from the transmission
filter 73b is output from the antenna 71 to the exterior, via the
antenna switch circuit 72.
[0124] Also, in the event that the input received signal is a
signal compatible with the GSM communication system, the antenna
switch circuit 72 selects one of the reception filters 77 to 80 in
accordance with the frequency band, and outputs the received
signal. The received signal subjected to a band limitation by the
selected one of the reception filters 77 to 80 is input into an LSI
83. The LSI 83, based on the input received signal, carries out a
process of demodulation into a sound signal, and controls the drive
of each section in the portable telephone terminal. Meanwhile, when
transmitting signals, the LSI 83 generates transmission signals.
The generated transmission signals are amplified by a power
amplifier 81 or 82, and output from the antenna 71 to the exterior,
via the antenna switch circuit 72.
[0125] Also, FIG. 12 illustrates a communication apparatus which
includes a variable reception filter 84 in place of the reception
filter 73a in the communication apparatus illustrated in FIG. 11,
and includes a variable transmission filter 85 in place of the
transmission filter 73b. Also, the communication apparatus includes
a variable reception filter 86 in place of the reception filters
77, 78, 79, and 80. The variable reception filters 84 and 86, and
variable transmission filter 85, include the variable filter
described in the section "2. Configuration of Variable Filter" in
the embodiment. Although not illustrated, the passbands in the
variable reception filters 84 and 86, and variable transmission
filter 85, are adjusted by a separately provided control
circuit.
[0126] As heretofore described, by installing in the communication
apparatus the filter of which the size in the signal line direction
is reduced, it is possible to downsize the communication apparatus.
That is, with the heretofore known filter, as a configuration has
been employed wherein the plurality of resonant lines are connected
in series, the size in the signal line direction has been increased
but, in the embodiment, as a configuration is employed wherein the
plurality of resonant lines are connected in parallel in the same
position, it is possible to reduce the size of the filter in the
signal line direction. Consequently, by mounting the miniaturized
filter, it is possible to downsize the communication apparatus. In
particular, as the number of filters becomes larger with the
communication module which carries out a high frequency band
communication, by mounting the filter of which the size in the
signal line direction is small, as in the embodiment, it is
possible to downsize the communication apparatus compatible with
the high frequency band communication.
[0127] Also, by installing the variable filter, it is possible to
selectively transmit and receive a plurality of frequency band
signals using one filter, meaning that, it being possible to reduce
the number of filters, it is possible to downsize the communication
apparatus. Also, as it is possible to reduce the passing loss by
miniaturizing the filter, it is possible to realize the
communication apparatus with superior communication
characteristics.
[0128] The communication apparatus of the embodiment is useful for
a mobile communication apparatus of which the usable frequency band
is approximately 800 MHz to 6 GHz, in particular, a mobile
communication apparatus which carries out communication using a
frequency band of 2 GHz or higher.
5. Advantages of Embodiment, and Other
[0129] According to the embodiment, by connecting a resonant line
functioning as a resonator between a signal line and ground, and
connecting a plurality of the resonant lines in the same position
in the signal line, it is possible to shorten the line length in
the signal line direction in comparison with a configuration
wherein a plurality of resonant lines are connected in series in
the signal line direction, as in the heretofore known technology,
so it is possible to reduce the size of the filter in the signal
line direction.
[0130] Also, by connecting one end of each of the resonant lines to
ground, it is possible to shorten the line length of the resonant
lines. In the event that the resonant lines are connected in series
to the signal line, as in the heretofore known technology, the line
length of the resonant lines has been .lamda./2 but, by connecting
the resonant lines in parallel to the signal line, and connecting
one end of each of the resonant lines to ground, as in the
embodiment, it is possible to totally reflect a signal which meets
the resonant condition, so it is possible to make the length of the
resonant lines .lamda./8.times.n (n is a positive integer).
[0131] Also, by adopting a configuration wherein the resonant lines
are connected in parallel to the signal line, it is possible to
mount them in high density on the substrate, so it is possible to
miniaturize the filter.
[0132] Also, by adopting a configuration wherein a plurality of the
resonant lines are connected to a common ground, it is possible to
dispose the resonant lines at an angle with respect to the signal
line, so it is possible to reduce the size of the filter in the
vertical direction (the direction perpendicular to the signal
line).
[0133] Also, as well as connecting a plurality of the resonant
lines to a common ground, by forming the resonant lines, of each of
which one end is connected to the signal line, and the other end is
connected to the ground, into an approximate arc shape, it is
possible to reduce the size of the filter in the vertical direction
(the direction perpendicular to the signal line).
[0134] Also, by suspending the capacitor electrodes on the resonant
lines, it is possible to increase the capacitance in the resonant
lines, meaning that, it being possible to shorten the physical line
length of the resonant lines, it is possible to reduce the size of
the filter in the vertical direction (the direction perpendicular
to the signal line).
[0135] Also, by suspending the variable capacitor electrodes on the
resonant lines, and changing the capacitance in the resonant lines
by displacing the variable capacitor electrodes, it is possible to
equivalently change the electrical length, so it is possible to
realize the variable filter. By employing the variable filter as
the reception filter and the transmission filter in the
communication module and communication apparatus, there is no need
to install the transmission filter and reception filter for each
passband in the multiband compatible communication module and
communication apparatus, so it is possible to downsize the
communication module and communication apparatus.
[0136] Also, by miniaturizing the filter, it being possible to
increase the number of filter modules which may be obtained from
one wafer at the time of filter manufacture, it is possible to
reduce a manufacturing cost.
[0137] All examples and conditional language recited herein are
intended for pedagogical purposes to aid the reader in
understanding the principles of the invention and the concepts
contributed by the inventor to furthering the art, and are to be
construed as being without limitation to such specifically recited
examples and conditions, nor does the organization of such examples
in the specification relate to a showing of the superiority and
inferiority of the invention. Although the embodiments of the
present inventions have been described in detail, it should be
understood that the various changes, substitutions, and alterations
could be made hereto without departing from the spirit and scope of
the invention.
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