U.S. patent application number 14/083202 was filed with the patent office on 2014-03-06 for apparatus and method for sensing of isolated output.
This patent application is currently assigned to Power Integrations, Inc.. The applicant listed for this patent is Power Integrations, Inc.. Invention is credited to Alex B. Djenguerian, Arthur B. Odell, Henson Wu.
Application Number | 20140063867 14/083202 |
Document ID | / |
Family ID | 44276331 |
Filed Date | 2014-03-06 |
United States Patent
Application |
20140063867 |
Kind Code |
A1 |
Djenguerian; Alex B. ; et
al. |
March 6, 2014 |
APPARATUS AND METHOD FOR SENSING OF ISOLATED OUTPUT
Abstract
A switched-mode power supply includes an energy transfer element
coupled between a primary side and a secondary side. A first main
terminal of a switch is coupled to the energy transfer element and
a second main terminal of the switch is coupled to an input of the
primary side. A driver circuit is coupled to drive the switch to be
open at a first one of a plurality of levels and closed at a second
one of the plurality of levels. The driver circuit is coupled to
drive the switch to be substantially independent of a voltage
between the first and second main terminals at a third one of the
plurality of levels. A current conducted between the first and
second main terminals at the third one of the plurality of levels
is sufficient to only partially discharge a capacitance that is
coupled to the first main terminal.
Inventors: |
Djenguerian; Alex B.;
(Saratoga, CA) ; Odell; Arthur B.; (Morgan Hill,
CA) ; Wu; Henson; (Saratoga, CA) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Power Integrations, Inc. |
San Jose |
CA |
US |
|
|
Assignee: |
Power Integrations, Inc.
San Jose
CA
|
Family ID: |
44276331 |
Appl. No.: |
14/083202 |
Filed: |
November 18, 2013 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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13721906 |
Dec 20, 2012 |
8593832 |
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14083202 |
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13398000 |
Feb 16, 2012 |
8355266 |
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13721906 |
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12770478 |
Apr 29, 2010 |
8144487 |
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13398000 |
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Current U.S.
Class: |
363/21.17 |
Current CPC
Class: |
H02M 3/33523 20130101;
H02M 3/156 20130101; H02M 3/33507 20130101; H02M 2001/0045
20130101 |
Class at
Publication: |
363/21.17 |
International
Class: |
H02M 3/335 20060101
H02M003/335 |
Claims
1. A switched-mode power supply comprising: an energy transfer
element coupled to transfer energy between a primary side of the
switched-mode power supply and a secondary side of the
switched-mode power supply; a switch having a first main terminal,
a second main terminal, and a control terminal, wherein the first
main terminal is coupled to the energy transfer element and the
second main terminal coupled to an input of the primary side of the
switched-mode power supply; a driver circuit coupled to drive the
switch at one of a plurality of levels, wherein the driver circuit
is coupled to drive the switch to be open between the first and
second main terminals at a first one of the plurality of levels,
wherein the driver circuit is coupled to drive the switch to be
closed between the first and second main terminals at a second one
of the plurality of levels, wherein the driver circuit is coupled
to drive the switch to be substantially independent of a voltage
between the first and second main terminals at a third one of the
plurality of levels, and wherein a current conducted between the
first and second main terminals at the third one of the plurality
of levels is sufficient to only partially discharge a capacitance
that is coupled to the first main terminal; and a controller
coupled to receive a current sense signal representative of the
current conducted between the first and second main terminals, the
controller further coupled to receive a feedback signal
representative of a regulated output of the switched-mode power
supply, wherein the controller is coupled to output a selection
signal to the driver circuit to drive the switch at one of the
plurality of levels in response to the current sense signal and the
feedback signal.
2. The switched-mode power supply of claim 1 wherein the driver
circuit is coupled to drive the switch at one of the plurality of
levels to control a transfer of energy from the primary side of the
switched-mode power supply to the secondary side of the
switched-mode power supply.
3. The switched-mode power supply of claim 1 wherein the switch
comprises a transistor coupled to the energy transfer element and
the input of the primary side of the switched-mode power
supply.
4. The switched-mode power supply of claim 3 wherein the transistor
is coupled to be OFF in response to the driver circuit driving the
switch at the first one of the plurality of levels.
5. The switched-mode power supply of claim 3 wherein the transistor
is coupled to be ON in response to the driver circuit driving the
switch at the second one of the plurality of levels.
6. The switched-mode power supply of claim 1 wherein the transistor
comprises a MOSFET coupled to operate in a saturation region of the
MOSFET in response to the driver circuit driving the switch at the
third one of the plurality of levels.
7. The switched-mode power supply of claim 1 wherein the feedback
signal is coupled to be generated in response to a reflected signal
representative of the regulated output of the switched-mode power
supply.
8. A switched-mode power supply comprising: an energy transfer
element coupled to transfer energy between a primary side of the
switched-mode power supply and a secondary side of the
switched-mode power supply, the energy transfer element having a
bias winding coupled to produce a bias winding voltage responsive
to an output voltage of the switched-mode power supply when a
rectifier coupled to the secondary side conducts a secondary
current; one or more switches coupled between the energy transfer
element and an input of the primary side of the switched-mode power
supply; switch driver circuitry coupled to drive the one or more
switches in a plurality of modes, wherein in a first one of the
plurality of modes, current conduction across the one or more
switches is substantially zero, wherein in a second one of the
plurality of modes, current conduction across the one or more
switches is sufficient to drive the bias winding voltage to a
magnitude that is representative of the output voltage of the
switched-mode power supply, and wherein in a third one of the
plurality of modes, current conduction across the one or more
switches has a sufficiently low magnitude and duration that the
rectifier coupled to the secondary side conducts the secondary
current without the bias winding voltage being driven to a
magnitude that is representative of an input voltage of the
switched-mode power supply; and feedback circuitry coupled to
receive a feedback signal representative of the bias winding
voltage and a current sense signal representative of the current
conduction across the one or more switches, wherein the feedback
circuitry is coupled to generate a mode select signal to which the
switch driver circuitry is responsive to drive the one or more
switches in the plurality of modes to control the transfer of
energy between the primary side of the switched-mode power supply
and the secondary side of the switched-mode power supply.
9. The switched-mode power supply of claim 8 wherein the feedback
circuitry includes a first feedback circuit coupled to the energy
transfer element to generate a first signal in response to a
reflected signal representative of the output voltage of the
switched-mode power supply.
10. The power converter of claim 8 wherein the feedback circuitry
includes a second feedback circuit coupled to the energy transfer
element to generate a second signal in response to a portion of a
decaying oscillation in a reflected signal representative of the
output voltage of the switched-mode power supply.
11. The switched-mode power supply of claim 8 wherein in the third
one of the plurality of modes, the current conduction across the
one or more switches is sufficient to only partially discharge a
capacitance that is coupled to the one or more switches.
12. The switched-mode power supply of claim 8 wherein switch driver
circuitry is coupled to drive the one or more switches in the
plurality of modes to control the transfer of energy between the
primary side of the switched-mode power supply to the secondary
side of the switched-mode power supply.
13. An integrated circuit for use in a switched-mode power
converter, the integrated circuit comprising: a first terminal to
be coupled to an energy transfer element that is coupled between a
primary side of the power converter and a secondary side of the
switched-mode power converter; a second terminal to be coupled to
an input of the switched-mode power converter; a third terminal to
be coupled to receive a current sense signal representative of a
current conduction between the first terminal and the second
terminal; a fourth terminal to be coupled to receive a feedback
signal representative of a regulated output of the switched-mode
power converter; a current control circuit coupled to the first
terminal, the second terminal, the third terminal, and the fourth
terminal, the current control circuit coupled to operate in one of
a first mode, a second mode, and a third mode in response to the
current sense signal and the feedback signal, wherein in the first
mode the current control circuit is coupled to conduct
substantially no current between the first terminal and the second
terminal, wherein in the second mode the current control circuit is
coupled to conduct current between the first terminal and the
second terminal that is sufficient to fully discharge a capacitance
coupled to the first terminal, and wherein in the third mode the
current control circuit is coupled to conduct current between the
first terminal and the second terminal that is sufficient to only
partially discharge the capacitance coupled to the first terminal.
Description
REFERENCE TO PRIOR APPLICATION(S)
[0001] This is a continuation of U.S. application Ser. No.
13/721,906, filed Dec. 20, 2012, now pending, which is a
continuation of U.S. application Ser. No. 13/398,000, filed Feb.
16, 2012, now U.S. Pat. No. 8,355, 266, which is a continuation of
U.S. application Ser. No. 12/770,478, filed Apr. 29, 2010, now U.S.
Pat. No. 8,144,487. U.S. application Ser. Nos. 13/721,906,
13/398,000 and 12/770,478 are hereby incorporated by reference.
BACKGROUND INFORMATION
[0002] 1. Field of the Disclosure
[0003] This invention is related to the control of switched-mode
power supplies. Specifically, it is related to low-cost power
supplies with regulated isolated outputs that must meet standards
for maximum power consumption when the output has no load, and yet
must keep the output within specified limits when a load is
suddenly applied.
[0004] 2. Background
[0005] Low-cost solutions to regulate an isolated output voltage of
a switching power supply typically rely on the magnetic coupling
between isolated windings of an energy transfer element to provide
information about the output to a control circuit. The control
circuit typically receives a signal representative of the output
voltage immediately after a switching event that delivers energy to
the output. The signal is typically received from a
primary-referenced winding of an energy transfer element instead of
from an optocoupler. This type of control is often referred to as
"primary-side control" or control using primary-side feedback.
[0006] Although these solutions eliminate the cost and the power
consumed by an optocoupler, they cannot sense the output voltage in
the absence of switching. A problem arises when the load on the
output of the power supply approaches zero. The power supply must
provide the specified regulated output voltage but almost no power.
Under such conditions, the power lost in the operation of the power
supply itself is a significant part of the total power consumed.
Requirements to limit the consumption of power by the power supply
under conditions of near zero external loading discourage the use
of a dummy internal load in a power supply. A dummy internal load,
sometimes called a pre-load, can be useful in a power supply to
help provide overvoltage protection, improve regulation between
multiple outputs, and prevent the switching frequency from going
below a minimum value. In particular, a dummy internal load is a
small permanent minimum load inside a power supply. However, a
penalty for using a dummy internal load is that the power supply
becomes less efficient because the dummy load dissipates power that
is not measured as output power. Also, the controller has to switch
more often to power the additional internal load, which results in
the power supply consuming additional power even though there is no
load connected to the power supply's output. To avoid these
drawbacks, it is useful to increase the time between switching
events under no-load conditions to reduce the losses inherent in
switching. However, under such conditions, the controller is unable
to sense the output voltage during the relatively long intervals
between switching events.
[0007] When a substantial load is suddenly applied to the output
during one of the relatively long intervals between switching
events, the output voltage can easily fall outside the specified
limits of regulation before the controller is able to respond to
the condition. A typical remedy for such a condition is the
addition of costly bulk capacitance to the output to provide the
energy required by a load that could be applied during the time
when the controller cannot sense the output.
BRIEF DESCRIPTION OF THE DRAWINGS
[0008] Non-limiting and non-exhaustive embodiments of the present
invention are described with reference to the following figures,
wherein like reference numerals refer to like parts throughout the
various views unless otherwise specified.
[0009] FIG. 1 shows an example power converter including a
controller in accordance with the teachings of the present
invention that provides sensing of an isolated output.
[0010] FIG. 2 is an example of a power converter including a
controller that uses a winding on a coupled inductor to sense
output voltage in accordance with the teachings of the present
invention and that provides sensing of an isolated output.
[0011] FIG. 3 shows voltage and current waveforms from an example
power converter that illustrates the operation of a controller in
accordance with the teachings of the present invention that
provides for sensing of an isolated output voltage.
[0012] FIG. 4 shows one example of a power converter that
illustrates one example of a current controller in accordance with
the teachings of the present invention.
[0013] FIG. 5 shows one example of a power converter that
illustrates another example of a current controller, which uses a
transistor for sensing of an isolated output voltage in accordance
with the teachings of the present invention.
[0014] FIG. 6 is flow diagram that shows one example method to
control a power converter in accordance with the teachings of the
present invention that provides for sensing of an isolated output
voltage.
DETAILED DESCRIPTION
[0015] Methods and apparatuses for implementing a power supply
controller that provide relatively low cost solutions that
accomplish sensing of an isolated output of a power converter are
disclosed. In the following description, numerous specific details
are set forth in order to provide a thorough understanding of the
present invention. It will be apparent, however, to one having
ordinary skill in the art that the specific detail need not be
employed to practice the present invention. In other instances,
well-known materials or methods have not been described in detail
in order to avoid obscuring the present invention.
[0016] Reference throughout this specification to "one embodiment",
"an embodiment", "one example" or "an example" means that a
particular feature, structure or characteristic described in
connection with the embodiment or example is included in at least
one embodiment of the present invention. Thus, appearances of the
phrases "in one embodiment", "in an embodiment", "one example" or
"an example" in various places throughout this specification are
not necessarily all referring to the same embodiment or example.
Furthermore, the particular features, structures or characteristics
may be combined in any suitable combinations and/or subcombinations
in one or more embodiments or examples. Particular features,
structures or characteristics may be included in an integrated
circuit, an electronic circuit, a combinational logic circuit, or
other suitable components that provide the described functionality.
In addition, it is appreciated that the figures provided herewith
are for explanation purposes to persons ordinarily skilled in the
art and that the drawings are not necessarily drawn to scale.
[0017] FIG. 1 is a schematic diagram that shows generally one
example of a switching power converter 100 that uses a flyback
topology in accordance with the teachings of the present invention.
In the illustrated example, power converter 100 is shown as a power
supply having flyback topology for explanation purposes. It is
noted, however, that there are many other known topologies and
configurations for switching power supplies. It is appreciated that
the example flyback topology illustrated in FIG. 1 is adequate for
explaining principles in accordance with the teachings of the
present invention and that the principles may apply also to other
types of switching regulators in accordance with the teachings of
the present invention. Details that will be discussed in greater
detail below are omitted from FIG. 1 to avoid obscuring teachings
in accordance with the present invention.
[0018] The example power converter in FIG. 1 controls the transfer
of energy from an unregulated input voltage V.sub.IN 102 at the
input of the power converter 100 to a load 122 at the output of the
power converter 100. The input voltage V.sub.IN 102 is coupled to
an energy transfer element T1 105 and a current controller 148. In
the example of FIG. 1, the energy transfer element T1 105 is a
coupled inductor, sometimes referred to as a transformer, with a
primary winding 108 and a secondary winding 112. In the example of
FIG. 1, primary winding 108 may be considered an input winding, and
secondary winding 112 may be considered an output winding. A clamp
circuit 104 is coupled to the primary winding 108 of the energy
transfer element T1 105 to control the maximum voltage on the
current controller 148.
[0019] In the example of FIG. 1, input voltage V.sub.IN 102 is
positive with respect to an input return 110, and output voltage
V.sub.O 120 is positive with respect to an output return 124. The
example of FIG. 1 shows galvanic isolation between the input return
110 and the output return 124 because input return 110 and output
return 124 are designated by different symbols. In other words, a
dc voltage applied between input return 110 and output return 124
will produce substantially zero current. Therefore, circuits
electrically coupled to the primary winding 108 are galvanically
isolated from circuits electrically coupled to the secondary
winding 112.
[0020] In the illustrated example, current controller 148 either
conducts current or does not conduct current in response to a
control circuit 144 that is included in a controller 142. Current
controller 148 and controller 142 may include integrated circuits
and discrete electrical components. In some examples, current
controller 148 and controller 142 may be integrated together in a
single monolithic integrated circuit.
[0021] In the example of FIG. 1, current controller 148 controls a
current Ip 126 in response to controller 142 to meet a specified
performance of the power converter 100. In operation, current
controller 148 produces pulsating current in primary winding 108
and in secondary winding 112. Current in secondary winding 112 is
rectified by rectifier D1 114 and then filtered by capacitor C1 116
to produce a substantially constant output voltage V.sub.O 120 or
output current I.sub.O 118 at the load 122. The operation of
current controller 148 also produces a time varying voltage V.sub.P
106 between the ends of primary winding 108. By transformer action,
a scaled replica of the voltage V.sub.P is produced between the
ends of secondary winding 112, the scale factor being the ratio
that is the number of turns of secondary winding 112 divided by the
number of turns of primary winding 108.
[0022] The example illustrated in FIG. 1 shows a capacitor C.sub.P
150 in broken lines at the node between one end of primary winding
108 and current controller 148. Capacitor C.sub.P 150 in the
example of FIG. 1 represents all the capacitance that couples to
current controller 148. Capacitor C.sub.P 150, which could be
referred to as the primary switching node capacitance, may include
natural capacitance internal to energy transfer element T1 105 as
well as the natural internal capacitance of current controller 148.
Capacitor C.sub.P 150 may also include discrete capacitors placed
intentionally in various parts of the circuit to filter noise and
to slow transitions of switching voltages. Capacitor C.sub.P 150
has a voltage V.sub.CP 128 that is the voltage at one end of
primary winding 108 with respect to the input return 110. The
importance of capacitor C.sub.P 150 will become apparent later in
this disclosure.
[0023] In the example of FIG. 1, a sensor 134 receives a sense
signal 132 that is representative of the output quantity to be
regulated at the output of power converter 100. The output quantity
to be regulated by controller 142 is typically the output voltage
V.sub.O 120, but in some examples is the output current I.sub.O
118, and in other examples may be a combination of output voltage
V.sub.O 120 and output current I.sub.O 118. Controller 142 receives
a feedback signal U.sub.FB 136 from sensor 134. Feedback signal
U.sub.FB 136 may be either a voltage or a current.
[0024] Since circuits electrically coupled to the secondary winding
112 are galvanically isolated from the circuits electrically
coupled to the primary winding 108, either the sense signal 132 is
galvanically isolated from the load 122, or sensor 134 provides
galvanic isolation between sense signal 132 and controller 142. In
other words, galvanic isolation may reside in either the sensor 134
or in another part of the path of the sense signal 132 not shown in
FIG. 1.
[0025] In the example of FIG. 1, controller 142 receives a current
sense signal 130 that is representative of the current I.sub.P 126.
Current sense signal 130 may be either a voltage or a current and
may be obtained using known methods. For example, current sense
signal 130 may be the output of a current transformer, the voltage
across a current sense resistor, or the voltage across the
on-resistance of a metal oxide field-effect transistor MOSFET that
conducts either the entire current I.sub.P 126 or a portion of the
current I.sub.P 126.
[0026] In the example of FIG. 1, controller 142 receives feedback
signal U.sub.FB 136 and current sense signal 130 to produce a mode
select signal 146 that is received by current controller 148. In
one example, current controller 148 may have three modes of
operation. A first mode may be one that does not conduct current,
such that current I.sub.P 126 is substantially zero when current
controller 148 is the first mode. A second mode may be one that
conducts as much current as external circuitry allows, such as for
example the condition where the current in the primary winding 108
of energy transfer element T1 105 is determined by the input
voltage V.sub.IN, the inductance of primary winding 108, and the
time that current controller 148 remains in the second mode. A
third mode may be one that restricts conduction of current to a
relatively small value during the time the current controller 148
remains in the third mode in accordance with the teachings of the
present invention. In one example, the relatively small value for
the current is a constant current value that is substantially less
than the current value during the second mode. In one example, the
relatively small constant current value of the third mode is 5
percent of the maximum current conducted in the second mode.
[0027] In the example of FIG. 1, feedback signal U.sub.FB 136 has
substantially different characteristics that depend on the changes
in modes of current controller 148 in accordance with the teachings
of the present invention. For example, when current controller 148
changes between the second mode and the first mode, the feedback
signal U.sub.FB 136 contains features that are not present when
current controller 148 changes between the third mode and the first
mode. Therefore, controller 142 includes a first feedback circuit
138 and a second feedback circuit 140 to interpret the feedback
signal U.sub.FB 136 appropriately for the different modes of
current controller 148 in accordance with the teachings of the
present invention. Examples of other controllers may include more
than two feedback circuits as required to interpret feedback
signals that arise from different modes of operation.
[0028] In the example of FIG. 1, control circuit 144 included in
controller 142 receives signals from first feedback circuit 138 and
second feedback circuit 140 to control the output of the power
converter as desired. Feedback circuits included in controller 142
may use any analog and digital circuits such as filter circuits,
sample and hold circuits, and comparators, to extract necessary
information from feedback signal U.sub.FB 136. Control circuit 144
included in controller 142 may use any analog and digital circuits,
such as oscillators, comparators, digital logic, and state
machines, or the like, to interpret and respond as required to
information received from the feedback circuits.
[0029] FIG. 1 shows mode select signal 146 as a single line that in
another example may represent several individual analog or digital
signals. For example, two binary digital signals lines for control
signal 146 may select as many as four distinct modes of current
controller 148 in accordance with the teachings of the present
invention.
[0030] FIG. 2 is a schematic diagram that shows another example of
a switching power converter 200 that uses a flyback topology in
accordance with teachings of the present invention. The example of
FIG. 2 includes a coupled inductor 205 that has a primary winding
108, a secondary winding 112, and a bias winding 210. Bias winding
210 may also be referred to as an auxiliary winding. In one
example, bias winding 210 in FIG. 2 is the sensor 134 introduced in
FIG. 1 that produces feedback signal U.sub.FB 136. Bias winding 210
produces a voltage V.sub.B 215 that is responsive to the output
voltage V.sub.O 120 when rectifier D1 114 on secondary winding 112
conducts. Sense signal 132, shown in FIG. 1 but not visible in FIG.
2, is the magnetic flux that couples bias winding 210 to secondary
winding 112 of the coupled inductor 205. In another example, bias
winding 210 may also provide a source of power to the circuits
within controller 142.
[0031] It is appreciated that many variations are possible in the
use of a bias winding to sense an output voltage and for providing
sensing while also providing power to a controller with galvanic
isolation. For example, a bias winding may apply a rectifier and a
capacitor similar to rectifier D1 114 and capacitor C1 116,
respectively, to produce a dc bias voltage while providing an ac
feedback signal from the anode of the rectifier. As such,
additional passive components such as resistors may be used on the
bias winding to scale the voltage from the winding to a value that
is more suitable to be received by controller 142.
[0032] Use of bias winding 210 to sense output voltage V.sub.O 120
has the advantages of providing galvanic isolation between output
voltage V.sub.O 120 and controller 142 without the expense of an
optocoupler. A disadvantage of using a winding on coupled inductor
205 to sense output voltage V.sub.O 120 is that the voltage V.sub.B
215 at bias winding 210 is representative of output V.sub.O 120
only when output rectifier D1 114 is conducting, whereas an
optocoupler can provide continuous sensing of output voltage
V.sub.O 120. Output rectifier D1 114 is conducting only while there
is a pulse of current in secondary winding 112. Therefore, the time
between pulses of current in secondary winding 112 is the time when
controller 142 cannot sense output voltage V.sub.O 120. In other
words, in contrast to sensing output voltage V.sub.O 120
continuously with an optocoupler, sensing output voltage V.sub.O
120 with a winding on coupled inductor 205 is limited to pulses
that may not occur often enough to provide the necessary
information for the desired control of output voltage V.sub.O 120.
Since secondary winding 112 has a pulse of current only after
primary winding 108 has a pulse of current, it is desirable to
decrease the time between pulses of current in primary winding 108
so that controller 142 can sense output voltage V.sub.O 120 more
often.
[0033] The rate and magnitude of pulsating current in primary
winding 108 is controlled by controller 142 to provide the power
required to maintain the desired output voltage V.sub.O 120 over a
range of values of load 122. As the load approaches zero, less
current in primary winding 108 is needed to maintain the desired
output voltage V.sub.O 120. As such, controllers may reduce the
magnitude of the current in primary winding 108 as well as increase
the time between pulses of current.
[0034] Controller 142 may produce pulses of current in primary
winding 108 by providing current controller 148 with a mode select
signal 146 that changes current controller 148 from the first mode
to the second mode, allowing current I.sub.P 126 to increase with a
linear slope to a desired maximum before returning to the first
mode. Operation of current controller 148 in the second mode fully
discharges capacitor C.sub.P 150 so that voltage V.sub.P 106 on
primary winding 108 is equal to input voltage V.sub.IN 102.
[0035] All the energy stored on capacitor C.sub.P 150 is lost when
current controller 148 operates in the second mode, even if the
maximum current I.sub.P 126 is allowed to increase to the lowest
practical value before returning to the first mode. The only way to
reduce the power dissipated from the full discharge of capacitor
C.sub.P 150 is to increase the time between discharges. In other
words, increasing the time between pulses of primary current will
reduce the power lost in the power converter as the load approaches
zero at the expense of increasing the time where the controller 142
cannot sense the output voltage V.sub.O 120. As a consequence, a
sudden increase in the load 122 may reduce the output voltage
V.sub.O 120 to an unacceptable value before controller 142 can
sense the voltage and respond to it.
[0036] A solution is discussed below that allows the controller 142
to sense the output voltage V.sub.O 120 frequently enough to
respond adequately to a sudden increase in the load 122 while also
reducing power dissipation at near zero load. This solution
produces pulses of current in primary winding 108 without fully
discharging capacitor C.sub.P 150. The solution is realized by the
introduction of a third mode of operation for current controller
148 in accordance with the teachings of the present invention. In
one example, the third mode of current controller 148 operates to
put only enough current into primary winding 108 so that output
rectifier D1 114 will conduct after current controller 148 returns
to the first mode. The third mode of current controller 148
conducts current with a sufficiently low magnitude and duration to
put the desired current into the primary winding while only
partially discharging capacitor C.sub.P 150 in accordance with the
teachings of the present invention. The determination of the proper
value of current for a given application is discussed in detail
later in this disclosure.
[0037] FIG. 3 shows voltage and current waveforms from the example
power converter of FIG. 2 that illustrates the operation of an
example controller in accordance with the teachings of the present
invention that provides sensing of an isolated output voltage. As
shown in the depicted example, pulses of current I.sub.P 126 that
fully discharge capacitor C.sub.P 150 begin at times t.sub.0 310,
t.sub.(N+1) 350, and t.sub.(N+2) 360. Pulses of current I.sub.P 126
that partially discharge capacitor C.sub.P 150 begin at times
t.sub.1 320, t.sub.2 330, and t.sub.N 340 in accordance with the
teachings of the present invention.
[0038] The distinction between pulses of current I.sub.P 126 that
fully discharge capacitor C.sub.P 150 and pulses of current I.sub.P
126 that partially discharge capacitor C.sub.P 150 is clear in the
waveform of V.sub.CP 128 that is the voltage on capacitor C.sub.P
150. Capacitor C.sub.P 150 is fully discharged when the voltage
V.sub.CP 128 is substantially zero. Capacitor C.sub.P 150 is only
partially discharged when the voltage V.sub.CP 128 remains
substantially greater than zero when current I.sub.P 126 is greater
than zero.
[0039] As shown, at the end of each full-discharge pulse and each
partial-discharge pulse of current I.sub.P 126, voltage V.sub.CP
128 rises above the input voltage V.sub.IN 102 while energy from
the energy transfer element (e.g., energy transfer element T1 105
in FIG. 1 and coupled inductor 205 in FIG. 2) charges capacitor
C.sub.P 150. Voltage V.sub.CP 128 rises until the output rectifier
D1 114 conducts, clamping the voltage V.sub.CP to the input voltage
V.sub.IN plus the reflected output voltage V.sub.OR, where the
reflected output voltage V.sub.OR is the voltage on secondary
winding 112 multiplied by the number of turns on primary winding
108 and divided by the number of turns on the secondary winding
112.
[0040] As shown, voltage V.sub.CP 128 remains clamped at the value
of V.sub.IN plus V.sub.OR until output rectifier D1 114 stops
conducting, which happens when the current from secondary winding
112 falls to zero. The energy stored in capacitor C.sub.P 150 that
raised its voltage above V.sub.IN 102 then dissipates in a decaying
oscillation with the self-inductance of primary winding 108 and the
effective parasitic resistance of the circuit.
[0041] FIG. 3 also shows the voltage V.sub.B 215 in FIG. 2 that
provides feedback signal U.sub.FB 136 to controller 142. Controller
142 may sense input voltage V.sub.IN 102 as well as output voltage
V.sub.O 120 from bias voltage V.sub.B 215. During full-discharge
pulses of current I.sub.P 126, bias voltage V.sub.B 215 goes
negative to a magnitude V.sub.INS that is representative of input
voltage V.sub.IN 102. After a full-discharge pulse of current
I.sub.P 126, output rectifier D1 114 conducts to allow sensing of
output voltage V.sub.O 120 as a positive voltage V.sub.OS on bias
winding 210 that is representative of output voltage V.sub.O 120.
After a pulse of current I.sub.P 126 that only partially discharges
capacitor C.sub.P 150, output rectifier D1 114 conducts just enough
to allow sensing of output voltage V.sub.O 120 with a decaying
oscillation of bias voltage V.sub.B 215 as shown in FIG. 3.
[0042] In one example, when the load 122 is large enough to require
full-discharge pulses of current I.sub.P 126 to maintain output
voltage V.sub.O 120 at a desired value, the full-discharge pulses
may occur as often as every switching period T.sub.S. An example
switching period T.sub.S is the time between t.sub.(N+1) 350 and
t.sub.(N+2) 360 in FIG. 3. Typically, light to moderate loads may
require patterns of full-discharge pulses separated by several
switching periods of no current pulses.
[0043] In the example, when the load 122 is near zero, only
partial-discharge pulses are used to sense the output voltage
V.sub.O 120 at intervals much shorter than the period between
full-discharge switching pulses in accordance with the teachings of
the present invention. It is not necessary to use partial-discharge
pulses to sense the output voltage V.sub.O 120 when the load 122 is
sufficiently greater than zero because full-discharge pulses occur
often enough at loads sufficiently greater than zero to provide
adequate sensing of the output voltage. The partial-discharge
pulses may be considered wake-up pulses that determine whether or
not a full-discharge pulse is required. The times between
partial-discharge pulses may be considered wake-up periods.
[0044] In the illustrated example, an example wake-up period
T.sub.W1 is the time between t.sub.1 320 and t.sub.2 330 in FIG. 3.
In one example, wake-up period T.sub.W1 is 16 switching periods
T.sub.S. In another example, wake-up pulses may be separated by
wake-up periods of different durations. The first partial-discharge
pulse in a train of partial-discharge pulses may follow a
full-discharge pulse by a period that is different from any wake-up
period within a train of wake-up pulses. FIG. 3 shows a period
T.sub.1 that is the time between t.sub.0 310 at the start of a
full-discharge pulse and time t.sub.1 320 that is the start of the
first partial-discharge pulse in a train of partial-discharge
pulses. In one example, the period T.sub.1 is 9 switching periods
whereas T.sub.W1 is 16 switching periods.
[0045] In one example, control circuit 144 included in controller
142 may determine the need for full-discharge pulses, partial
discharge pulses, or no pulse within a switching period T.sub.S
according to the value of feedback signal U.sub.FB 136 immediately
after each pulse of current I.sub.P 126 in accordance with the
teachings of the present invention. For example, if a sequence of
full-discharge pulses causes the sensed output voltage V.sub.OS to
rise beyond a first threshold value, control circuit 144 may set
mode select signal 146 such that current controller 148 conducts no
current for several subsequent switching periods T.sub.S. If sensed
output voltage V.sub.OS remains above the first threshold value
after the next full-discharge pulse, controller 142 may conclude
that the load is near zero and begin using partial discharge pulses
to sense the output voltage V.sub.O 120. The example of FIG. 3
illustrates an example in which control circuit 144 determined that
sensed output voltage V.sub.OS after the partial-discharge pulse at
time T.sub.N 340 was too low, requiring consecutive full-discharge
pulses at times t.sub.(N+1) 350 and t.sub.(N+2) 360.
[0046] In the example of FIG. 2, control circuit 144 interprets a
signal received from first feedback circuit 138 after a
full-discharge pulse, and control circuit 144 interprets a signal
received from second feedback circuit 140 after a partial-discharge
pulse in accordance with the teachings of the present invention. In
the example of FIG. 2, first feedback circuit 138 samples feedback
signal U.sub.FB 136 during the time when output rectifier D1 114 is
conducting. In the example of FIG. 2, second feedback circuit 138
samples feedback signal U.sub.FB 136 during a decaying oscillation
of the bias voltage V.sub.B 215 after output rectifier D1 114 stops
conducting.
[0047] In the example, the peaks of the decaying oscillation in the
bias voltage V.sub.B 215 are representative of output voltage
V.sub.O 120 after output rectifier D1 114 stops conducting because
capacitor C.sub.P 150 charges to a value representative of output
voltage V.sub.O 120 when output rectifier D1 114 conducts after a
partial-discharge pulse. The maximum voltage on capacitor C.sub.P
150 sets the initial condition for the decaying oscillation after
output rectifier D1 114 stops conducting. Therefore, each peak in
the decaying oscillation of bias voltage V.sub.B 215 is determined
by the maximum voltage on capacitor C.sub.P 150 after a
partial-discharge pulse.
[0048] In the example of FIG. 2, first feedback circuit 138 samples
feedback signal U.sub.FB 136 to regulate output voltage V.sub.O 120
over a wide range of loads. In contrast to first feedback circuit
138, second feedback circuit 140 in one example does not sample
feedback signal U.sub.FB 136 to regulate output voltage V.sub.O 120
over a wide range of load. Instead, second feedback circuit 140 in
the example is used only to determine whether or not there has been
a sufficient change in the output voltage V.sub.O 120 during a
train of partial-discharge pulses to require a change in operating
mode in accordance with the teachings of the present invention.
[0049] Specifically, in one example, second feedback circuit 140
holds the value of the second peak in the decaying oscillation of
feedback signal U.sub.FB 136, as illustrated for example in FIG. 3
with the decaying oscillations in V.sub.B 215 after the first
partial-discharge pulse in a train of consecutive partial-discharge
pulses, and compares it to samples of the second peak in the
decaying oscillation of feedback signal U.sub.FB 136 after each
subsequent partial-discharge pulse in the train of consecutive
partial-discharge pulses. When the value of a subsequent sample is
less than the value of the first sample by a threshold value,
control circuit 144 determines that output voltage V.sub.O 120 is
too low, and sets mode select signal 146 to start a sequence of
full-discharge pulses. It is appreciated that any peak value in the
decaying oscillation may be sampled for use in the comparison. In
one example, the second peak value may be a preferred peak because
it has the highest magnitude and is relatively free from noise and
distortion that may be present on the first peak while the output
rectifier D1 114 is conducting. In one example, the threshold value
is 30 millivolts.
[0050] In one example, the magnitude and duration of a
partial-discharge pulse are just sufficient to allow output
rectifier D1 114 to conduct at the end of the partial-discharge
pulse. In another example, the magnitude and duration of a
partial-discharge pulse are more than sufficient to allow output
rectifier D1 144 to conduct at the end of the partial-discharge
pulse. The output voltage V.sub.O 120 may be sensed with greater
accuracy when output rectifier D1 144 is allowed to conduct until a
transient voltage associated with non-ideal coupling of the
windings of coupled inductor 205 reduces to a negligible value. The
non-ideal coupling, sometimes quantified as a leakage inductance,
may produce a voltage between output rectifier D1 144 and secondary
winding 112 when diode D1 144 begins to conduct. The transient
voltage owing to leakage inductance may also distort the first peak
of the decaying oscillation. Therefore, it is desirable to allow
the voltage from the leakage inductance to reduce to a negligible
value so that capacitor C.sub.P 150 charges to a voltage that more
accurately represents output voltage V.sub.O 120 before output
rectifier D1 144 stops conducting. It is also desirable not to
sample the first peak of the decaying oscillation to avoid
distortion from the effects of leakage inductance.
[0051] In one example, the magnitude of the partial-discharge pulse
is 16 milliamperes whereas the peak current of a full-discharge
pulse is 250 milliamperes. As such, the energy transferred to the
output by the partial-discharge pulse may be considered
insignificant in comparison to the energy transferred to the output
by the full-discharge pulse because the energy transferred is
proportional to the square of the peak current in primary winding
108. It will be appreciated that since the partial-discharge pulse
may transfer finite energy to the output, controllers that have a
minimum switching frequency, however small, may require the power
supply to have a dummy internal load to keep the output voltage
V.sub.O 120 from going higher than desired as the output current
I.sub.O 118 approaches zero.
[0052] It may be determined either analytically or experimentally
that there is a magnitude and a duration for a partial-discharge
pulse that gives a minimum power loss in the power converter for a
particular set of circumstances. The duration of the
partial-discharge pulse is typically less than half of one period
of the decaying oscillation of feedback signal U.sub.FB 136 as
illustrated in FIG. 3. In one example, the duration of the
partial-discharge pulse is approximately one quarter of one period
of the decaying oscillation of feedback signal U.sub.FB 136. In one
example where the inductance of the primary winding 108 of a
coupled inductor is 2.2 millihenries, capacitor C.sub.P 150 is
approximately 70 picofarads, the partial discharge pulse is 16
milliamperes for a duration of approximately 600 nanoseconds. It
will be appreciated that in one example control circuit 144 may
adjust the magnitude and duration of partial-discharge pulses to
achieve minimum power loss in the power converter and to guarantee
that output rectifier D1 114 conducts in accordance with the
teachings of the present invention. This adjustment may be done,
for example, in response to an external signal received by
controller 144. The adjustment may also be done by choosing values
of discrete components within control circuit 144. In examples
where control circuit 144 is included in an integrated circuit, the
adjustment may be done by trimming the internal parameters of the
integrated circuit.
[0053] FIG. 4 is a schematic diagram of a power converter 400 that
illustrates one example of current controller 148 in greater
detail. In the example of FIG. 4, current controller 148 includes a
mode selector 410 that receives mode select signal 146 from
controller 142. In the example of FIG. 4, mode selector 410 closes
either switch S1 420, or switch S2 440, or neither switch in
response to mode select signal 146.
[0054] In the example of FIG. 4, the first mode opens switch S1 420
and switch S2 440 such that current I.sub.P 126 is substantially
zero when current controller 148 is in the first mode. In the
example of FIG. 4, the second mode closes switch S1 420 and opens
switch S2 440 to fully discharge capacitor C.sub.P 150. In the
example of FIG. 4, the third mode opens switch S1 420 and closes
switch S2 440 such that current I.sub.P 126 is the value of current
source 430 to partially discharge capacitor C.sub.P 150 in
accordance with the teachings of the present invention. It is
appreciated that in other examples current source 430 could be a
variable current source that varies in accordance with the degree
of partial discharge of capacitor C.sub.P 150 desired.
[0055] FIG. 5 is a schematic diagram of a power converter 500 that
shows another example of current controller 148 that includes a
metal oxide semiconductor field-effect transistor (MOSFET) 520 and
a tri-level driver 510 to produce full-discharge and
partial-discharge pulses of current I.sub.P 126 in response to the
mode select signal 146. In the example of FIG. 5, tri-level driver
510 responds to signals from mode selector 410 to produce at least
three distinct values of a voltage between the gate terminal and
the source terminal of n-channel MOSFET 520 in response to mode
select signal 146. In one example, the gate terminal of MOSFET 520
may be considered as a control terminal of MOSFET 520.
[0056] In the example of FIG. 5, the first mode of current
controller 148 applies a gate-to-source voltage substantially less
than the threshold voltage of n-channel MOSFET 520. As a result,
MOSFET 520 is substantially switched OFF in the first mode of
operation of current controller 148. In the example of FIG. 5, the
second mode of current controller 148 applies a gate-to-source
voltage substantially greater than the threshold voltage of
n-channel MOSFET 520. As a result, MOSFET 520 is substantially
switched ON in the second mode of operation of current controller
148. In the example of FIG. 5, the third mode of current controller
148 applies a gate-to-source-voltage slightly higher than the
threshold voltage of n-channel MOSFET 520. As a result, the
gate-to-source voltage at the control terminal of MOSFET 520 in the
third mode corresponds to MOSFET 520 providing a partial-discharge
current pulse for the magnitude of current I.sub.P 126. In other
words, when in the third mode of operation, MOSFET 520 operates not
as a switch that may be either open or closed, but in its
saturation region, sometimes called the active region, where the
drain current is controlled primarily by the gate-to-source voltage
and is substantially independent of the drain-to-source voltage. In
examples where MOSFET 520 and tri-level driver 510 are included in
an integrated circuit, tri-level driver 510 may be designed such
that the voltage applied to the gate during the third mode of
current controller 148 tracks the threshold voltage of MOSFET 520,
thereby reducing the change in partial-discharge current over a
range of temperature and process variations. It is appreciated that
in other examples, tri-level driver 510 could have four or more
drive levels to select different MOSFET 520 saturation
characteristics according to the degree of partial discharge of
capacitor C.sub.P 150 desired.
[0057] FIG. 6 is a flow diagram 600 that shows one example method
to control a power converter in accordance with the teachings of
the present invention providing for sensing of an isolated output.
After starting in block 605, a current controller is operated in
full capacitance discharge mode in block 610 to produce a current
pulse that fully discharges capacitance on a node of the current
controller.
[0058] Next, a first feedback circuit senses the isolated output
voltage in block 615. In block 620, information from feedback
circuits is processed to estimate the condition of the load. Then,
in decision block 625, the flow continues to block 630 if the load
is near zero, or branches back to block 610 if the load is not near
zero. In one example, the load is considered near zero when the
full-discharge pulses occur at intervals greater than the wake-up
period T.sub.W1 of FIG. 3.
[0059] In block 630, the current controller is operated in partial
capacitance discharge mode to produce a current pulse that only
partially discharges a capacitance on a node of the current
controller. Then a second feedback circuit senses the isolated
output voltage in block 635 before returning to block 620 where the
information from feedback circuits is processed.
[0060] The above description of illustrated examples of the present
invention, including what is described in the Abstract, are not
intended to be exhaustive or to be limitation to the precise forms
disclosed. While specific embodiments of, and examples for, the
invention are described herein for illustrative purposes, various
equivalent modifications are possible without departing from the
broader spirit and scope of the present invention. Indeed, it is
appreciated that the specific voltages, currents, frequencies,
power range values, times, etc., are provided for explanation
purposes and that other values may also be employed in other
embodiments and examples in accordance with the teachings of the
present invention.
[0061] These modifications can be made to examples of the invention
in light of the above detailed description. The terms used in the
following claims should not be construed to limit the invention to
the specific embodiments disclosed in the specification and the
claims. Rather, the scope is to be determined entirely by the
following claims, which are to be construed in accordance with
established doctrines of claim interpretation. The present
specification and figures are accordingly to be regarded as
illustrative rather than restrictive.
* * * * *