U.S. patent application number 13/470709 was filed with the patent office on 2013-11-14 for method and apparatus for generating dedicated data channels in backscatter rfid systems.
The applicant listed for this patent is Tajinder MANKU. Invention is credited to Tajinder MANKU.
Application Number | 20130299579 13/470709 |
Document ID | / |
Family ID | 49547883 |
Filed Date | 2013-11-14 |
United States Patent
Application |
20130299579 |
Kind Code |
A1 |
MANKU; Tajinder |
November 14, 2013 |
METHOD AND APPARATUS FOR GENERATING DEDICATED DATA CHANNELS IN
BACKSCATTER RFID SYSTEMS
Abstract
An antenna apparatus for backscattering an incoming radio
frequency (RF) signal includes an antenna for backscattering the
incoming RF signal in accordance with a reflection coefficient
characteristic of the antenna. A variable impedance circuit
includes an output electrically connected to the antenna. A low
pass delta sigma modulator is coupled to the variable impedance
circuit and digitally controls the output of the variable impedance
circuit, such that the reflection coefficient of the antenna is
adjusted based on the output of the variable impedance circuit.
Inventors: |
MANKU; Tajinder; (Waterloo,
CA) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
MANKU; Tajinder |
Waterloo |
|
CA |
|
|
Family ID: |
49547883 |
Appl. No.: |
13/470709 |
Filed: |
May 14, 2012 |
Current U.S.
Class: |
235/439 ;
343/745 |
Current CPC
Class: |
G06K 19/0723 20130101;
H01Q 1/2225 20130101; G06K 19/0701 20130101 |
Class at
Publication: |
235/439 ;
343/745 |
International
Class: |
H01Q 1/50 20060101
H01Q001/50; G06K 7/00 20060101 G06K007/00 |
Claims
1. An antenna apparatus for backscattering an incoming radio
frequency (RF) signal comprising: an antenna for backscattering the
incoming RF signal in accordance with a reflection coefficient
characteristic of the antenna; a variable impedance circuit having
an output electrically connected to the antenna; and at least one
low pass delta sigma (.DELTA..SIGMA.) modulator coupled to the
variable impedance circuit and digitally controlling the output of
the variable impedance circuit; wherein the reflection coefficient
(.GAMMA.) of the antenna is adjusted based on the output of the
variable impedance circuit.
2. The antenna apparatus of claim 1 wherein an output of the at
least one low pass delta sigma modulator switches the output of the
variable impedance circuit between two states to adjust the
reflection coefficient.
3. The antenna apparatus of claim 1 wherein an input signal applied
to the low pass delta sigma modulator consists of one of a complex
modulation signal offset from the incoming radio frequency signal
by +/-.omega..sub.0.
4. The antenna apparatus of claim 3 wherein the complex modulation
signal consists of one of a GMSK, QPSK, nPSK, nQAM, and an OFDM
signal.
5. The antenna apparatus of claim 1 wherein an output of the low
pass delta sigma modulator is one of a return to zero (RTZ) and a
non-return to zero (NRZ) type signal.
6. The antenna apparatus of claim 1 further comprising at least a
second low pass delta sigma modulator coupled to the variable
impedance circuit, wherein the output of the variable impedance
circuit is further digitally controlled by the at least a second
low pass delta sigma modulator.
7. The antenna apparatus of claim 6 wherein input signals to the
first and second low pass delta sigma modulators comprise in-phase
(I) and quadrature (Q) signals respectively.
8. The antenna apparatus of claim 6 wherein a combined output of
the first and second said low pass delta sigma modulators switches
the output of the variable impedance circuit between four states to
adjust the reflection coefficient of the antenna.
9. The antenna apparatus of claim 8 wherein the reflection
coefficient comprises four states which are relative from each
other by .GAMMA..sub.oexp(j0.degree.)
.GAMMA..sub.oexp(j180.degree.), .GAMMA..sub.oexp(j90.degree.), and
.GAMMA..sub.oexp(j270.degree.).
10. The antenna apparatus of claim 9 wherein the first said low
pass delta sigma modulator ((.DELTA..SIGMA.).sub.I) switches the
states between 0 degrees and 180 degrees.
11. The antenna apparatus of claim 10 wherein the second low pass
delta sigma modulator ((.DELTA..SIGMA.).sub.Q) switches the states
between 90 degrees and 270 degrees.
12. The antenna apparatus of claim 11 wherein outputs of the first
and second low pass delta sigma modulators alternately switch
between each other, wherein if (.DELTA..SIGMA.).sub.I generates 0,
180, 180, 0, 0, 180 . . . and (.DELTA..SIGMA.).sub.Q generates 90,
90, 270, 270, . . . , .GAMMA. is controlled to adjust by 0, 90,
180, 90, 180, 270, 0, 270, . . . .
13. The antenna apparatus of claim 6 wherein the input signals
applied to the low pass delta sigma modulators comprise sine and
cosine wave forms offset from a frequency of the incoming RF signal
by .omega..sub.1, where .omega..sub.1 can be either positive or
negative.
14. The antenna apparatus of claim 6 wherein the input signals
applied to the low pass delta sigma modulators consists of a
complex modulation signal offset from a frequency of the incoming
RF signal by one of +.omega..sub.o, -.omega..sub.o and zero.
15. The antenna apparatus of claim 14 wherein the complex
modulation signal consists of one of a GMSK, nPSK, QPSK, nQAM, and
OFDM signal.
16. The antenna apparatus of claim 6 wherein outputs of the low
pass delta sigma modulators consist of one of a return to zero
(RTZ) and a non-return to zero (NRZ) type signal.
17. The antenna apparatus of claim 7 wherein the I and Q signals
are adjusted to compensate for errors that may arise in generating
.GAMMA..sub.oexp(j0.degree.), .GAMMA..sub.oexp(j180.degree.),
.GAMMA..sub.oexp(j90.degree.), and
.GAMMA..sub.oexp(j270.degree.).
18. The antenna apparatus of claim 17 wherein the errors are
compensated for in a radio frequency identification (RFID) reader
device electromagnetically coupled to the antenna.
19. The antenna apparatus of claim 1 further comprising at least
one filter device at the variable impedance circuit to filter out
of band noise output from the at least one low pass delta sigma
modulator.
20. The antenna apparatus of claim 1 wherein the antenna apparatus
is included a tag terminal of a radio frequency identification
(RFID) system including a reader device, wherein the antenna
apparatus is activated for backscattering RF signals only when the
tag terminal is within a predetermined critical distance of the
reader device.
21. The antenna apparatus of claim 1 wherein the antenna comprises
part of a tag terminal, the tag terminal electromagnetically
coupled to a reader device within a radio frequency identification
(RFID) system, the RFID system comprising clocking the low pass
delta sigma modulator, generation of the clocking consisting of one
of a clock circuit within the tag reader and a clock circuit
generated based on a frequency of the incoming RF signal.
Description
FIELD OF THE INVENTION
[0001] The present invention relates generally to a method and
apparatus for generating dedicated data transmission channels in
backscatter radio frequency communication networks.
BACKGROUND OF THE INVENTION
[0002] Radio Frequency Identification (RFID) systems are commonly
used to locate and track items in a near-field communication
network including a reader device and at least one wireless
terminal, or tag. Energized time-varying electromagnetic radio
frequency (RF) waves, which comprise the carrier signal, are
transmitted from the reader to the tags in a given RFID network or
system. Tags use backscatter technology to reflect the reader's RF
signal back to the reader, modulating the signal to encode and
transmit data.
[0003] FIG. 1 depicts a prior art RFID system in which data
transmission from tags 101a-c to reader device 103 is performed on
a same frequency channel or spectrum 104. Using the established
backscattering technology, each of the plurality of tags typically
in the RFID system or network sends RF signals on the same
backscattered carrier signal. Hence, the backscattered RF signals
from each tag overlap those of other tags within the same RF
spectrum associated with that given reader device/RFID network.
[0004] As a consequence, tag collision in RFID systems occur when
the multiple tags are energized by the same RFID reader device, and
simultaneously reflect their respective, overlapping signals back
to the reader using the given frequency channel. Thus the tag
collision problem is exacerbated whenever a large number of tags
must be read together in the same RF field. The reader is unable to
differentiate these signals when the simultaneously generated
signals collide. The tag collisions confuse the reader, generate
data transmission errors, and generally reduce data throughput
within the RFID system or network.
[0005] Various systems have been proposed to isolate individual
tags. For example, in one technique aimed at reducing collision
errors, when the reader recognizes that tag collision has taken
place, it sends a special "gap pulse" signal. Upon receiving this
signal, each tag consults a random number counter to determine the
interval to wait before sending its data. Since each tag gets a
unique number interval, the tags send their data at different
times. The adverse impact on overall RFID system performance, in
terms of data throughput rate, however, still exists.
[0006] Modulating the signal received by the tag and re-radiating
the modulated signal backscattered to the reader device is known,
using such signal modulation schemes, such as phase shift keying
(PSK) and amplitude shift keying (ASK), where the tag changes its
reflection coefficient by changing the impedance match between
states. However, the adverse effects of tag collisions resulting
from overlapping backscattered signals on a given frequency channel
still remain.
SUMMARY OF THE INVENTION
[0007] Provided is an antenna apparatus for backscattering an
incoming radio frequency (RF) signal. The antenna apparatus
comprises an antenna for backscattering the incoming RF signal in
accordance with a reflection coefficient characteristic of the
antenna, a variable impedance circuit having an output electrically
connected to the antenna, and at least one low pass delta sigma
(.DELTA..SIGMA.) modulator coupled to the variable impedance
circuit and digitally controlling the output of the variable
impedance circuit, wherein the reflection coefficient (.GAMMA.) of
the antenna is adjusted based on the output of the variable
impedance circuit.
[0008] In one embodiment, an output of the at least one low pass
delta sigma modulator switches the output of the variable impedance
circuit between two states to adjust the reflection
coefficient.
[0009] In another embodiment, an input signal applied to the low
pass delta sigma modulator consists of one of a complex modulation
signal offset from the incoming radio frequency signal by
+/-.omega..sub.o.
[0010] The complex modulation signal may consists of any of a GMSK,
QPSK, nPSK, nQAM, and an OFDM signal.
[0011] In yet another embodiment, the antenna apparatus further
comprises at least a second low pass delta sigma modulator coupled
to the variable impedance circuit, wherein the output of the
variable impedance circuit is further digitally controlled by the
second low pass delta sigma modulator.
[0012] In a further embodiment, input signals to the first and
second low pass delta sigma modulators of the antenna apparatus
comprise in-phase (I) and quadrature (Q) signals respectively.
BRIEF DESCRIPTION OF THE DRAWINGS
[0013] The invention will now be described by way of example only
with reference to the following drawings in which:
[0014] FIG. 1 shows a prior art RFID system in which data
transmission from tag to reader is performed on a same frequency
channel;
[0015] FIG. 2 shows, in one embodiment, an apparatus for generating
the varying impedance for backscattering;
[0016] FIG. 3 shows, in one embodiment, an apparatus for generating
an IQ signal offset by the frequency of a digital signal source
such as a Direct Digital Synthesizer;
[0017] FIG. 4 shows, in one embodiment, an apparatus for generating
an OFDM signal offset by the frequency of a digital signal source
such as a Direct Digital Synthesizer;
[0018] FIG. 5a shows, in one embodiment, a modulator apparatus for
generating interleaved SSB signals based on IQ signal inputs;
[0019] FIG. 5b shows a representative output signal of the SSB
signals generated using the modulator apparatus of FIG. 5a;
[0020] FIG. 6 shows an embodiment of an RFID system in which data
transmission, via backscatter from tag to reader, is performed on
dedicated frequency channels;
[0021] FIG. 7 shows, in one embodiment, an apparatus for generating
a QAM signal;
[0022] FIG. 8a shows, in one embodiment, a modulator apparatus for
generating a GMSK signal;
[0023] FIG. 8b shows a representative output of the GMSK signal
generated using the modulator apparatus of FIG. 5a;
[0024] FIG. 8c shows a representation of quadrature errors which
may be produced in a reflected signal generated using the modulator
apparatus of FIG. 5a;
[0025] FIG. 9a shows a representation of controlling data rate
versus power; and
[0026] FIG. 9b shows a representation of link budget versus
distance between tag and reader.
DETAILED DESCRIPTION
[0027] The term modulation as used herein refers to the process by
which the radio frequency identification (RFID) wireless terminal,
or tag, changes the carrier radio frequency (RF) signal of the
reader antenna to convey information. For instance, in phase
modulation, data being transmitted from the reader device to the
tag is encoded in changes in the phase of the carrier wave sent out
by the RFID reader device.
[0028] FIG. 2 shows, in one embodiment, an antenna apparatus 200 in
a wireless communication system, such as a radio frequency
identification (RFID) communication network, which may be passive
or semi-passive, for generating a varying impedance 205 at antenna
203 for backscattering an incoming radio frequency (RF) signal,
such as from a reader device of the RFID network. Antenna 203,
which may be part a tag terminal of the RFID communication network,
backscatters the incoming RF signal in accordance with its
reflection coefficient (.GAMMA.) characteristic.
[0029] Still with regard to FIG. 2, illustrated is one embodiment
for designing the varying impedance circuit 205 used to generate
the backscattering wave from the antenna 203. Here the impedance
Z.sub.L is switched between two states depending on the control
bit. When the control bit is high Z.sub.L looks like zero impedance
at f.sub.rf, hence the backscattering .GAMMA.(f.sub.rf)=1. Note
that Z.sub.L is designed to have a high impedance other than
f.sub.rf. At 2f.sub.rf the impedance helps to reduce folding of
2f.sub.rf. When the control is low, Z.sub.L>>50 ohms.
Therefore .GAMMA.=0, and no signal is back scattered.
[0030] The varying impedance can also be designed to produce a
phase shift in the backscattered wave. That is,
.GAMMA..sub.i=.alpha.e.sup.j.phi..sub.1
where .phi. has two states, .phi..sub.1 and .phi..sub.2 and .alpha.
is a constant The back scattering impedance is then given by:
Z i = Z s ( 1 + .alpha. j .phi. i ) ( 1 - .alpha. j .phi. i )
##EQU00001##
where Z.sub.1 has two states, Z.sub.1 and Z.sub.2. Here .phi..sub.i
can be designed to have states .phi..sub.1=0.degree. and
.phi..sub.2=180.degree.. Here, Z.sub.s is the impedance of the
antenna. Since the antenna impedance adjusts given its environment,
the effective .GAMMA. is simply rotated and scaled. This can be
illustrated by assuming Zs changes to Z.sub.s.beta.exp(j.phi.)
where .beta. is the scaling factor and .phi. is the rotation.
Therefore, .GAMMA. changes to:
.GAMMA. = Z - Z s .beta. j .PHI. Z + Z s .beta. j .PHI.
##EQU00002## Or , .GAMMA. = Z .beta. - 1 - j .PHI. - Z s Z .beta. -
1 - j .PHI. - Z s ##EQU00002.2##
Given this, a change in Z.sub.s results in scaling and rotating Z
by .beta..sup.-1 and -.phi., respectively. In this complex
modulation scheme, phase changes in r rather than amplitude changes
may be utilized.
[0031] In an embodiment, a one filter or more filters may be in the
variable impedance circuit to filter out of band noise output from
the low pass delta sigma modulator.
[0032] FIG. 3 shows, in one embodiment, apparatus 300 for
generating an In-Phase--Quadrature (IQ) signal (308, 309) offset by
the frequency of a digital signal source which in one embodiment
may be Direct Digital Synthesizer (DDS) 307. A low pass delta sigma
(.DELTA..SIGMA.) modulator 302 may be applied to generate a complex
modulation signal. As referred to herein, the low pass delta sigma
modulator generates an output bit stream that represents the input
data from a DC level to some predetermined design bandwidth, BW.
Beyond the predetermined design bandwidth BW, quantized noise of
the low pass delta sigma increases until at some design cutoff
point, the signal would be deemed to have too much quantization
noise.
[0033] Still with regard to FIG. 3, the signals to the mixers are
generated by DDS 307.
[0034] FIG. 4 shows, in one embodiment, apparatus 400 for
generating an orthogonal frequency division multiplexing (OFDM)
signal offset by the frequency of a digital signal source.
[0035] In the examples of FIGS. 3 and 4, the complex modulation
signals are generated at f.sub.rf+.delta.f and f.sub.rf-.delta.f;
i.e. they are double side banded and have a lower and upper side
band.
[0036] FIG. 5a shows, in one embodiment, modulator apparatus 500
for generating interleaved SSB signals based on IQ signal inputs
508,509.
[0037] Single side band (SSB) signals can also be generated, but
two low pass .DELTA..SIGMA. modulators 502a, 502b are required. The
two .DELTA..SIGMA. modulators 502a, 502b provide signals that
either change .GAMMA. by 0, 90, 180, or 270.degree. (or in general
offset+0, offset+90, offset+180, or offset+270); see FIG. 5a. The
input signals to the first and second low pass delta sigma
modulators 502a, 502b may comprise in-phase (I) and quadrature (Q)
signals 508, 509 respectively. In one embodiment, the input signal
applied to the low pass delta sigma modulators 502a, 402b consists
of a complex modulation signal offset from the incoming radio
frequency signal by +.omega..sub.o or -.omega..sub.o or zero.
[0038] Still in regard to FIG. 5a, the first .DELTA..SIGMA. (i.e.
(.DELTA..SIGMA.).sub.1) has an output that either changes .GAMMA.
by 0 or 180.degree. and the other .DELTA..SIGMA. (i.e.
(.DELTA..SIGMA.).sub.Q) by 90 or 270.degree.. However, the outputs
are interleaved, alternately switching between the first
.DELTA..SIGMA. and the second .DELTA..SIGMA.. So if
(.DELTA..SIGMA.E).sub.I generates 0, 180, 180, 0, 0, 180 . . . and
(.DELTA..SIGMA.).sub.Q generates 90, 90, 270, 270, . . . then
.GAMMA. is controlled to change by 0, 90, 180, 90, 180, 270, 0,
270, . . . . By using this architecture SSB signals may be
generated.
[0039] FIG. 5b shows a representative output signal 510 of the SSB
signals generated using the modulator apparatus 500 of FIG. 5a.
FIG. 5b shows the output of such a structure where the signal
applied to it is sin .omega..sub.bbt and cos .omega..sub.bbt to the
(.DELTA..SIGMA.).sub.I and (.DELTA..SIGMA.).sub.Q modulator,
respectively. Here .omega..sub.bb is being changed to three
different frequencies.
[0040] The impedance corresponding to the phases may be determined
via the equations above. For example if .alpha.=1/sqrt(2), .phi.=0,
90, 180, 270, f.sub.rf=1 GHz, Z.sub.s=50.OMEGA., the impedances
become 50+100j, 10+20j, 10-20j, and 50-100j, respectively.
[0041] If there are any errors in Z, this will result in an
effective IQ offset in the reflected signal. However, this can be
corrected within the reader device using known IQ correction
schemes. If the antenna impedance changes, one can apply
equalization on the RFID reader.
[0042] FIG. 6 shows an embodiment of an RFID communication network
600 in which data transmission, via backscatter from tag to reader,
is performed on dedicated frequency channels using the complex
modulation apparatus and method for low pass delta sigma
modulation, by generating separate channels 605, 606, 607 for each
of the tags 601a-c used in the RFID communication network 600. The
complex modulation method and apparatus for low pass delta sigma
modulation are herein referred to, and denoted, as "the
.GAMMA.-.DELTA..SIGMA. scheme". An antenna 603a-c in respective
ones of tag terminals 601a-c backscatters the incoming RF signal,
such as from reader device 602, in accordance with a reflection
coefficient characteristic of the antenna 603a-c. A variable
impedance circuit (not shown in FIG. 6) has an output electrically
connected to the antenna 603a-c. A low pass delta sigma modulator
is coupled to an input of the variable impedance circuit to
digitally control the output of the variable impedance circuit,
such that reflection coefficient .GAMMA. of antenna 603a-c may be
adjusted by changing the output of the variable impedance
circuit.
[0043] FIG. 7 shows, in one embodiment, modulator apparatus 700 for
generating quadrature amplitude (QAM) signals. Data bits are
applied to LUT (Look Up Table) 701 and then applied to the
.DELTA..SIGMA. modulator 702a, 702b.
[0044] FIG. 8a shows, in one embodiment, modulator apparatus 800
for generating a Gaussian minimum shift keying (GMSK) signal. By
applying the SSB scheme, complex modulation signals like GMSK,
nPSK, quadrature phase shift keying (QPSK), OFDM, nQAM, etc. may be
generated, where n represents an integer.
[0045] In one embodiment, the output of the low pass delta sigma
modulators 802a-b may be a return to zero (RTZ), so if the data is
1101101, the output would be 10100010100010; note there is a zero
between each bit. In an alternate embodiment, the output of low
pass delta sigma modulator 802a-b may be a non-return to zero (NRZ)
type signal; for example, if the data is 1101101, the output is
1101101, and nothing is added to the data stream.
[0046] FIG. 8b shows a representative output of the GMSK signal
generated using modulator apparatus 800 of FIG. 8a. Here, a first
order .DELTA..SIGMA. is used. One can easily improve the spectrum
by applying a higher order .DELTA..SIGMA. modulator. The center
frequency is 2.179 normalized units. The phases of the reflection
coefficient may have errors; i.e. .GAMMA..sub.oexp(j0.degree.),
(.GAMMA..sub.o+.epsilon..sub.1)exp(j(180.degree.+.phi..sub.1),
(.GAMMA..sub.o+.epsilon..sub.2)exp(j(90.degree.+.phi..sub.2), and
(.beta..sub.o+.epsilon..sub.3)exp(j(270.degree.+.phi..sub.3), where
.epsilon..sub.1, .phi..sub.1, .epsilon..sub.2, .phi..sub.2,
.epsilon..sub.3, and .phi..sub.3 represent the errors. These errors
produce a quadrature error in the signal reflected back by the
antenna.
[0047] FIG. 8c is a representation of the reflected signal if a SSB
is generated at an offset of .delta.f. The error tone at -.delta.f
is produced due to this error; ideally the error signal would not
exist. This quadrature error can be corrected by adjustments either
(i) to the I and Q signals applied to the low pass delta sigma
modulators, or (ii) within the reader of the RFID communication
network itself.
[0048] For instance, in the reader what is measured is E(Q 2)-E(I
2) and E(IQ), where E(x) is the average expected value. The term
E(Q 2)-E(I 2) is a measure of the gain mismatch, and E(IQ) is a
measure of the phase mismatch. The gain on the I (or Q) channel may
be corrected until E(Q 2)-E(I 2)=0, and the phase so E(IQ)=0. This
may be done in a closed loop scheme, for example using a Least Mean
Square filter.
The matrix that is used:
Icorrected=I*D
Qcorrected=sin(phase_error)*I+cos(phase_error)*Q
where D is a measure of the gain mismatch between I and Q, and
phase_error is the error in phase between I and Q. Without any
error, D=1 and phase_error=0.sup.o.
[0049] With regard to the clocking function utilized by the
wireless tag terminal, such as for driving the low pass delta sigma
modulator, generation of the clocking function may be provided by a
clock circuit within the tag reader, or via a clock circuit
generation based on the frequency of the incoming RF signal
provided by the reader device of the RFID network.
[0050] For example, in the instance of using the signal from the
read as the clock, if the reader is at frf, the clock used by the
tag will frf, or some frequency, frf/N, where N is some integer
(that is frf is divided by N to generate a clock).
[0051] FIG. 9a shows a representation of controlling data rate
versus power. A power management system may be introduced that
depends on the data rate and the modulation type based on the
distance between the tag and the reader, r. As the reader gets
close enough (i.e. r<r.sub.min) the tag gets enough power to
turn on. From r.sub.min to r.sub.critical the tag transmits using a
slow clock and ASK or PSK. As .sub.r<r.sub.critical the tag can
start transmitting using 64QAM. For RF backscattering technology,
the power received by the tag (i.e. P.sub.RX(r)) is given by:
P RX ( r ) = ( .lamda. 4 .pi. r ) 2 P TX G TX G RX ##EQU00003##
where .lamda. is the wavelength of the carrier signal, r is the
distance between the tag and the reader, P.sub.TX is the power of
the transmitter, G.sub.TX is the antenna gain of the reader, and
G.sub.RX is the antenna gain of the tag. The modulated power from
the tag is then received by the reader.
[0052] FIG. 9b shows a representation of link budget versus
distance between tag and reader. The link budget between the tag
and reader is shown as a function of the tag-reader distance. The
power received by the tag is decreased as the tags moves further
away form the reader. At some such position, the backscatter power
of the tag is attenuated as it travels back to the reader. The
signal to noise ratio (SNR) is given by the power received by the
reader over the phase noise of the oscillator within the
reader.
[0053] For example, since the power coming from the reader falls
off as (1/r 2) the complex modulation technology may be applied
when the reader is closer. As the reader gets further, lower
modulations can be used.
[0054] It is understood that application of the complex modulation
requires a higher SNR and more power.
[0055] Although preferred embodiments of the invention have been
described herein with regard to passive and semi-passive RFID
communication networks, it is contemplated, and indeed it will be
understood by those skilled in the art, that the solutions
presented herein may be applied to other aspects of wireless
communication. Accordingly, a person of ordinary skill in the art
would understand that the specific embodiments described herein,
while illustrative are not necessarily comprehensive. Thus, other
various modifications may be made those skilled in the art without
departing from the scope of the invention as defined by the
claims.
* * * * *