U.S. patent application number 13/674873 was filed with the patent office on 2013-08-22 for optical transmission apparatuses, methods, and systems.
This patent application is currently assigned to Level 3 Communications, LLC. The applicant listed for this patent is Level 3 Communications, LLC. Invention is credited to David J. Copeland, Alistair J. Price, Raymond Zanoni.
Application Number | 20130216232 13/674873 |
Document ID | / |
Family ID | 43016007 |
Filed Date | 2013-08-22 |
United States Patent
Application |
20130216232 |
Kind Code |
A1 |
Zanoni; Raymond ; et
al. |
August 22, 2013 |
OPTICAL TRANSMISSION APPARATUSES, METHODS, AND SYSTEMS
Abstract
Apparatuses, systems, and methods are disclosed that provide for
an agile coherent optical modem that can generate agile RF
waveforms and data rates on a generic opto-electronic hardware
platform. An "agile coherent optical modem" [ACOM] approach to
optical communications by employing a software configurable and
adaptive technologies to the transport system. The ACOM generate
agile RF waveforms and data rates on a generic opto-electronic
hardware platform. By employing advanced communication techniques
to the optical domain such as wavelength agility, waveform agility,
and symbol rate agility, it is possible to enable robust optical
communications. The ACOM allows for the transport capacity of a
communications link to be varied, thereby accommodating variations
in transport conditions, range, opacity, etc.
Inventors: |
Zanoni; Raymond; (Columbia,
MD) ; Copeland; David J.; (Silver Spring, MD)
; Price; Alistair J.; (Ellicott City, MD) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Level 3 Communications, LLC; |
|
|
US |
|
|
Assignee: |
Level 3 Communications, LLC
Broomfield
CO
|
Family ID: |
43016007 |
Appl. No.: |
13/674873 |
Filed: |
November 12, 2012 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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12917395 |
Nov 1, 2010 |
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13674873 |
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11446392 |
Jun 2, 2006 |
7826752 |
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12917395 |
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60711306 |
Aug 25, 2005 |
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60686551 |
Jun 2, 2005 |
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Current U.S.
Class: |
398/79 ;
398/115 |
Current CPC
Class: |
H04B 10/65 20200501;
H04B 10/11 20130101; H04B 10/5561 20130101; H04B 10/614 20130101;
H04B 10/6151 20130101; H04B 10/677 20130101; H04B 10/60 20130101;
H04B 10/5055 20130101; H04B 10/541 20130101 |
Class at
Publication: |
398/79 ;
398/115 |
International
Class: |
H04B 10/11 20060101
H04B010/11 |
Claims
1. An agile coherent optical system comprising: an agile optical
transmitter including: a client interface configured to receive
electrical signals; a signal processing unit configured to process
the electrical signals and produce electrical RF signals as
waveforms having both amplitude and phase characteristics; and an
optical transport unit receiving the electrical RF signals and
including a vector modulator capable of transporting information as
optical signals having waveforms and data rates corresponding to
the electrical RF signals provided by the signal processing unit,
while preserving the amplitude and phase characteristics of the
electrical RF signals; an optical transport unit including a
polarization diversity coherent receiver capable of receiving
information as an optical signal having waveforms and data rates
from the agile optical transmitter and producing electrical RF
signals preserving the amplitude and phase characteristics of the
waveform; a signal processing unit configured to process the
electrical RF signals from the optical transport unit and produce
electrical signals containing the information carried by the
waveforms; and a client interface configured to receive and
transmit the electrical signals from the signal processing
unit.
2. The agile coherent optical system of claim 1, wherein the agile
coherent optical system is at least part of a WDM system.
3. The agile coherent optical system of claim 1, wherein the agile
coherent optical system is at least part of a free-space optical
transport system.
4. The agile coherent optical system of claim 1, wherein the agile
coherent optical system is at least part of an optical fiber
optical transport system.
5. The agile coherent optical system of claim 1, wherein the agile
coherent optical system operates at data rates of up to 160
gigabits per second.
6. The agile coherent optical system of claim 1, wherein the agile
coherent optical system operates at data rates of up to 10 gigabits
per second.
7. The agile coherent optical system of claim 1, wherein the agile
coherent optical system is at least a part of an optical transport
system including at least one of an optical amplifier, an optical
switch, and an optical add-drop multiplexer.
8. The agile coherent optical system of claim 1, wherein the agile
coherent optical system is at least a part of a hybrid
fiber-free-space optical transport system.
9. The agile coherent optical system of claim 8, wherein the
optical transport system includes wavelength translation of optical
signals being transported from the agile optical transmitter to the
agile optical receiver.
10. The agile coherent optical system of claim 1, wherein at least
part of one of the agile optical transmitter and agile optical
receiver is integrated on a common substrate.
11. The agile coherent optical system of claim 1, wherein the agile
optical receiver is integrated as a transceiver.
12. The agile coherent optical system of claim 1, wherein a
plurality of the agile optical transmitters are provided in a
common module.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of and claims benefit of
priority to U.S. Non-Provisional patent application Ser. No.
12/917,395, filed on Nov. 1, 2010, which is incorporated herein by
reference in its entirety for all purposes. Application Ser. No.
12/917,395 is a divisional of and claims benefit of priority to
U.S. Non-Provisional patent application Ser. No. 11/446,392, filed
Jun. 2, 2006, which is incorporated herein by reference in its
entirety for all purposes. Application Ser. No. 11/466,392 claims
benefit of priority to U.S. Provisional Patent Application No.
60/686,551, filed on Jun. 2, 2005, and U.S. Provisional Patent
Application No. 60/711,306, filed on Aug. 25, 2005, both of which
are incorporated herein by reference in their entirety for all
purposes.
STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT
[0002] Not applicable.
BACKGROUND OF THE INVENTION
[0003] The present invention is directed generally to the
transmission of information in communication systems including
fiber and free-space systems. More particularly, the invention
relates to transmitting information via optical signals in optical
transmission systems and transmitters and receivers for use
therein.
[0004] The development of digital technology has provided resources
to store and process vast amounts of information. While this
development has greatly increased information processing
capabilities, it was soon recognized that in order to make
effective use of information resources, it was necessary to
interconnect and allow communication between information resources.
Efficient access to information resources requires the continued
development of information transmission systems to facilitate the
sharing of information between resources.
[0005] The continued advances in information storage and processing
technology has fueled a corresponding advance in information
transmission technology. Information transmission technology is
directed toward providing high speed, high capacity connections
between information resources. One effort to achieve higher
transmission capacities has focused on the development of optical
transmission systems for use in conjunction with high speed
electronic transmission systems. Optical transmission systems
employ fiber networks to provide high capacity, low error rate
transmission of information over long distances at a relatively low
cost.
[0006] Optical transmission of information is performed by
imparting the information in some manner to a lightwave carrier by
varying the characteristics of the lightwave. The lightwave is
transmitted to a receiver at a destination for the information. At
the receiver, a photodetector is used to detect the lightwave
variations and convert the information carried by the variations
into electrical form.
[0007] In most optical transmission systems, the information is
imparted by using the information data stream to either modulate a
lightwave source to produce a modulated lightwave or to modulate
the lightwave after it is emitted from the light source. The former
modulation technique is known as "direct modulation", whereas the
latter is known as "external modulation", i.e., external to the
lightwave source. External modulation is more often used for higher
speed transmission systems, because the high speed direct
modulation of a source often causes undesirable variations in the
wavelength of the source. The wavelength variations, known as
chirp, can result in transmission and detection errors in an
optical system.
[0008] Data streams can be modulated onto the lightwave using a
number of different schemes. The two most common schemes are return
to zero [RZ] and non-return to zero [NRZ]. In RZ modulation, the
modulation of each bit of information begins and ends at the same
modulation level. In NRZ schemes, the modulation level is not
returned to a base modulation level, i.e., zero, at the end of a
bit, but is directly adjusted to a level necessary to modulate the
next information bit. Other modulation schemes, such as duobinary
and PSK, encode the data in a waveform.
[0009] In many systems, the information data stream is modulated
onto the lightwave at a carrier wavelength, .lamda.c, to produce an
optical signal carrying data at the carrier wavelength. The
modulation of the carrier wavelength also produces symmetric lobes,
or sidebands, that broaden the overall bandwidth of the optical
signal. The bandwidth of an optical signal determines how closely
spaced successive optical signals can be spaced within a range of
wavelengths.
[0010] Alternatively, the information can be modulated onto a
wavelength proximate to the carrier wavelength using subcarrier
modulation [SCM]. SCM techniques, such as those described in U.S.
Pat. Nos. 4,989,200, 5,432,632, and 5,596,436, generally produce a
modulated optical signal in the form of two mirror image sidebands
at wavelengths symmetrically disposed around the carrier
wavelength. Generally, only one of the mirror images is required to
carry the signal and the other image is a source of signal noise
that also consumes wavelength bandwidth that would normally be
available to carry information. Similarly, the carrier wavelength,
which does not carry the information, can be a source of noise that
interferes with the subcarrier signal. Modified SCM techniques have
been developed to eliminate one of the mirror images and the
carrier wavelength, such as described in U.S. Pat. Nos. 5,101,450
and 5,301,058.
[0011] Initially, single wavelength lightwave carriers were
spatially separated by placing each carrier on a different fiber to
provide space division multiplexing [SDM] of the information in
optical systems. As the demand for capacity grew, increasing
numbers of information data streams were spaced in time, or time
division multiplexed [TDM], on the single wavelength carrier in the
SDM system as a means to provide additional capacity. The continued
growth in transmission capacity has spawned the transmission of
multiple wavelength carriers on a single fiber using wavelength
division multiplexing [WDM]. In WDM systems, further increases in
transmission capacity can be achieved not only by increasing the
transmission rate of the information via each wavelength, but also
by increasing the number of wavelengths, or channel count, in the
system.
[0012] There are two general options for increasing the channel
count in WDM fiber transport systems. The first option is to widen
the transmission bandwidth to add more channels at current channel
spacings. The second option is to decrease the spacing between the
channels to provide a greater number of channels within a given
transmission bandwidth. The first option currently provides only
limited benefit, because most optical systems use erbium doped
fiber amplifiers [EDFAs] to amplify the optical signal during
transmission. EDFAs have a limited bandwidth of operation and
suffer from non-linear amplifier characteristics within the
bandwidth. Difficulties with the second option include controlling
optical sources that are closely spaced to prevent interference
from wavelength drift and nonlinear interactions between the
signals.
[0013] A further difficulty in WDM fiber transport systems is that
chromatic dispersion, which results from differences in the speed
at which different wavelengths travel in optical fiber, can also
degrade the optical signal. Chromatic dispersion is generally
controlled in a system using one or more of three techniques. One
technique is to introduce the optical path of the different
wavelengths to offset the dispersion of the different wavelengths
in the transmission fiber using through the use of optical
components such as Bragg gratings or arrayed waveguides that vary
the relative optical paths of the wavelengths. Another technique is
to intersperse different types of fibers that have opposite
dispersion characteristics to that of the transmission fiber. A
third technique is to attempt to offset the dispersion by
prechirping the frequency or modulating the phase of the laser or
lightwave in addition to modulating the data onto the lightwave.
For example, see U.S. Pat. Nos. 5,555,118, 5,778,128, 5,781,673 or
5,787,211. These techniques require that additional components be
added to the system and/or the use of specialty optical fiber that
has to be specifically tailored to each length of transmission
fiber in the system.
[0014] New fiber designs have been developed that substantially
reduce the chromatic dispersion of WDM signals during transmission
in the 1550 nm wavelength range. However, the decreased dispersion
of the optical signal allows for increased nonlinear interaction,
such as four wave mixing, to occur between the wavelengths that
increases signal degradation. The effect of lower dispersion on
nonlinear signal degradation becomes more pronounced at increased
bit transmission rates.
[0015] The many difficulties associated with increasing the number
of wavelength channels in WDM and free-space systems, as well as
increasing the transmission bit rate have slowed the continued
advance in communications transmission capacity. In view of these
difficulties, there is a clear need for transmission techniques and
systems that provide for higher performance optical communication
systems.
BRIEF SUMMARY OF THE INVENTION
[0016] The present invention introduce a paradigm shift from the
conventional approach of hardware defined optical transport to an
"agile coherent optical modem" [ACOM] approach to optical
communications by employing a software configurable and adaptive
technologies to the transport system. The ACOM generates agile RF
waveforms and data rates on a generic opto-electronic hardware
platform. By employing advanced communication techniques to the
optical domain such as wavelength agility, waveform agility, and
symbol rate agility, it is possible to enable robust optical
communications. The ACOM allows for the transport capacity of a
communications link to be varied, thereby accommodating variations
in transport conditions, range, opacity, etc.
[0017] The ACOM includes four key elements: a) an optical vector
modulator in the transmitter, b) a coherent optical receiver, c) a
programmable electronics platform, and d) wavelength translation,
if necessary. The transmitter combines arbitrary waveform
generation with programmable electronics and the capability to map
an RF waveform (i.e. both amplitude and phase) into the optical
domain. A coherent receiver allows the recovery of the amplitude
and phase information, which together with programmable electronics
and adaptive communications between the transmitter and receiver,
enables the ACOM.
BRIEF DESCRIPTION OF THE DRAWINGS
[0018] Embodiments of the present invention will now be described,
by way of example only, with reference to the accompanying
drawings, wherein:
[0019] FIG. 1 shows an agile coherent optical modem [ACOM].
[0020] FIG. 2 shows a Software Defined Optical Transmitter.
[0021] FIG. 3 shows a Software Defined Optical Receiver.
[0022] FIG. 4 shows a Software Defined Optical Transmitter.
[0023] FIG. 5 shows an I-Q phase diversity mode of the agile
coherent optical receiver [ACOR].
[0024] FIG. 6 shows a basic mode operation of the ACOR.
[0025] FIG. 7 shows an ACOM Transmitter and Receiver
embodiment.
[0026] FIG. 8 shows an idealized ACOM.
[0027] FIG. 9 shows a diagram of a communications model for the
ACOM.
[0028] FIG. 10 shows a front end receiver architecture.
[0029] FIG. 11 shows an I/Q intradyne receiver.
[0030] FIG. 12A shows transmitter options for transmitter
architectures.
[0031] FIG. 12B shows transmitter options for insertion losses for
various theoretical modulation formats.
[0032] FIG. 13 shows modulator concepts and theoretical
performance.
[0033] FIG. 14 shows an experimental ACOM platform.
[0034] FIG. 15 shows an ACOT architecture.
[0035] FIG. 16 shows a multi-electrode architecture.
[0036] FIG. 17 shows a schematic or 2-electrode DQPSK
modulator.
[0037] FIG. 18 shows transmit electronics.
[0038] FIG. 19 shows an ACOR architecture.
[0039] FIG. 20 shows receiver electronics.
[0040] FIG. 21 shows a receiver DSP architecture.
[0041] FIG. 22 shows an integrated O/E BGA package.
[0042] FIG. 23 shows an ACOM system approach.
[0043] FIG. 24 shows a receiver front-end.
[0044] FIG. 25 shows a phase recovery subsystem.
[0045] FIG. 26 shows a clock recovery subsystem.
[0046] FIG. 27 shows a frequency error recovery.
[0047] FIG. 28 shows a joint phase and frequency recovery.
[0048] FIG. 29 shows a phase error characteristic for QPSK.
[0049] FIG. 30 shows a signal phase and the local digital
reference.
[0050] FIG. 31 shows a phase error transient.
[0051] FIG. 32 shows a zoom-view of the residual phase error
(degrees) in steady state.
[0052] FIG. 33 shows A/D clock phase as a percentage of the symbol
time T.sub.s in zero noise.
[0053] FIG. 34 shows A/D clock phase in noise: SNR=5 dB.
[0054] FIG. 35 shows QPSK probability of error results.
[0055] FIG. 36 illustrates the residual phase noise limited 64-QAM
constellation.
[0056] FIG. 37 illustrates the phase noise limited 64-QAM
constellation with 6-bit A/D.
DETAILED DESCRIPTION OF THE INVENTION
[0057] FIG. 1 illustrates an ideal ACOM with an "agile optical
transmitter" [AOT] and a "agile coherent optical receiver" [ACOR].
The AOT (FIG. 2) is comprised of a client interface, a
signal-processing unit [SPU], and an optical transport unit
generally including a vector modulator or other device that
converts any RF waveform into an optical signal at any wavelength.
See U.S. Pat. No. 6,118,566, which is incorporated herein by
reference, for an example of optical upconversion of RF waveforms.
The signal-processing unit is a flexible digital signal-processing
[DSP] unit that can generate any RF waveform in any shape and at
any data rate. The signal-processing unit also has the capability
of scrambling data or pre-distorting the signals for security
purposes, forward error correction [FEC] for improved link
communication, adaptive modulation for optimizing the data rate and
modulation format, and polarization control of the signals in an
appropriately designed vector modulator. The client interface can
be any custom or standard protocol that mates the AOT to the system
of interest.
[0058] The ACOR (FIG. 3) consists of a receive optical transport
unit generally including polarization diversity coherent receiver
front end or other device that allows the transmitted optical
signal to be converted into an RF signal whose amplitude, phase and
polarization are recovered. That signal is fed into a
signal-processing unit that demodulates the incoming signal,
descrambles the scrambled signal and/or post compensates for any
pre-distortion for secure transmission, FEC, and link performance
monitoring. The client interface can be any custom or standard
interface to the system.
[0059] The ACOM is predicated on an architecture that employs a
generic electro-optics platform to maximize information
transmission combined with programmable signal-processing units and
distinct client interface assemblies. This architecture allows a
scalable, flexible platform that can accommodate a wide range of
deployment scenarios, as well as continuing advancement of the
technology. The architecture of the ACOM is designed to exploit the
strengths and compensate for the limitations of the platform. For
example, while the vector integrated optical modulator will have a
finite bandwidth, the SPU employs various modulation formats
ranging from NRZ to QAM to provide the required capacity through
the system. As an example, it is conceivable that if a 50 GHz
bandwidth, I-Q optical modulator is driven at 40 Gbaud, then it is
possible to transmit 160 Gb/s at QAM16.
[0060] The AOT embodiment shown in FIG. 4 includes a client
interface, signal processing unit, an optical transport unit
including a vector modulator and a narrow line-width, low noise
tunable laser. The optical vector modulator should be able to
support any of the following modulation formats: BPSK, QPSK, and
QAM-2.sup.N in order to increase the line rate and the spectral
efficiency. There are various implementations of the vector optical
modulators with tradeoffs based on loss, drive voltage and ease of
generating multiple modulation formats. The I-Q modulator is
capable of being driven with an "in-phase" [I] RF signal and a
"quadrature phase" [Q] RF signal to generate any modulation format
analogous to RF communications transmitters. Although the insertion
loss is higher than that of external modulators, the benefit is the
ease with which any optical waveform can be generated makes it
extremely suitable for combination with DSP technology to produce
software definable data rates and modulation formats.
[0061] In one embodiment, the client interface will be a 1.times.10
Gb/s synchronous serial (for BPSK) and 2.times.10 Gb/s synchronous
serial (for QPSK), which can employ high speed Silicon Germanium
CMOS DACs to drive the IQ modulator. Vendors for the high speed
DACs include Pulse Link Inc, whereas high speed FPGAs are available
from Xilinx and others, which when combined with the I-Q modulator
will enable the generation of a wide range of formats, such as:
NRZ, BPSK, QPSK, or QAM-2.sup.N with data rates up to or exceeding
40 Gb/s.
[0062] The ACOR includes a polarization controller, low phase
noise, local oscillator, such as from a Princeton Optronics Inc.,
that can be combined with an input signal using an optical
processors to generate under software control either a heterodyne
or homodyne signal.
[0063] FIGS. 5 and 6 illustrate the operation of the optical
receiver. In FIG. 5, a temperature controlled Mach-Zehnder
interferometer can be tuned to act as a 3 dB coupler with signals
directed to both balanced receivers. The local oscillator is also
split equally to mix with the signals at the balanced receivers. In
this mode, the receiver acts as phase diversity I-Q receiver where
the I and Q signals are separated optically.
[0064] In FIG. 6 the Mach-Zehnder interferometer can be temperature
tuned under software control to pass the signal through to the
upper balanced receiver only. In this mode, the receiver can be
used as a true homodyne receiver for BPSK, or general purpose
heterodyne receiver. Since the lower half of the receiver is
unused, it can receive a second signal applied to the `Aux` input
of the Mach-Zehnder. Note that although FIG. 5 has the advantage of
separating I and Q signals optically, it suffers the disadvantage
of an inherent 3 dB sensitivity penalty for BPSK reception.
Therefore, switching the Mach-Zehnder to pass through mode allows
the receiver to operate at the highest receiver sensitivity for
BPSK.
[0065] Separate polarization controllers can be used to minimize
the complexity of the receiver; however, it is desirable to
integrate the polarization control features into the receiver
signal-processing unit.
[0066] In order to increase further the transmission capacity of
the system, polarization combining the signals from two separately
modulated transmitters as illustrated in FIG. 7. FIG. 7 also
illustrates Phase 2 implementation of a 20 Gbaud, QAM-16
transmitter. The full 40 Gbaud transmitter would consist of two
polarization multiplexed transmitters as is shown.
[0067] A single generic opto-electronic transport platform can
provide the capability to dynamically manage the capacity and
system margin in the link. On the transmit side, a vector modulator
can be used in the optical transport unit to transport both
amplitude and phase information over the data rates of interest
using suitable modulation formats. On the receive side, a coherent
receiver can be used in the receive optical transport unit to
recover the amplitude and phase information. Because these
technologies enable the transmission and reception of both
amplitude and phase information, a single generic transport
platform is possible over the range of interest. The resulting
architecture is illustrated in FIG. 8.
[0068] A single generic transport platform further allows the
signal processing units to be "in-service" definable. The modular
architecture support the dynamic control of the modulation format
with the signal processing module, which is integrated with the
transport platform via a well-defined interface, irrespective of
the modulation format.
[0069] While the description of the present invention will be
generally describing the client interface, signal processing unit,
and optical transport units separately, it will be appreciated that
various levels of integration can be employed for the agile optical
transmitter and agile optical receiver. Multiple functions can be
performed on a common substrate and/or board, or the main functions
can be separated depending upon the design and application
objectives. In addition, the agile optical transmitter and agile
optical receiver can be made as separate module or integrated in a
transceiver with the transmitter and receiver having the same or
differing level of functional integration. Also, one or more
transmitters and receivers can be included in a module, which can
be configured as a single board or as multiple boards, such as
mother and daughter boards, which may or may not be
hot-swappable.
[0070] The ACOM provides the following key innovations: 1) an ACOT
designed to reduce the need for high speed DACs, 2) a novel
coherent receiver front-end that allows for homodyne, heterodyne,
or intradyne coherent detection, and 3) a signal processing
architecture that remediates channel impairments in a free space
coherent optical communications link.
[0071] Free Space Coherent Optical [FSO]Communications Impairments:
The ACOM can incorporate channel impairments, realistic hardware,
and moving platforms (FIG. 10). A description of a model for the
ACOM that includes: multiple modulation formats, variable symbol
rate, coherent receiver, signal processing algorithms for
unraveling the rotating multi-level constellations, realistic ADCs
with quantization error, optical phase lock loops, Doppler shifts
due to moving platforms, and algorithms for handling diverse
polarization in signal processing is provided below.
[0072] For FSO communications, the primary impairments are due to
scattering and atmospheric turbulence. In the case of scattering
there are several mechanisms: molecular scattering, molecular
absorption, and aerosol scattering. Turbulence induced refractive
index fluctuations causes distortion of the optical beam by
distorting the optical phase front that results in random
modulations in the received optical power (intensity
scintillations). Clouds, smoke, or fog can completely obscure a
line of sight communications link.
[0073] There are multiple methods for dealing with dense scattering
(i.e. clouds, smoke, and fog) by either using site diversity or by
using very long wavelengths such as 3.8 .mu.m. Atmospheric
turbulence, which is caused by rapid, spatial and temporal
fluctuations of various scales in clear air, is the most
significant impairment. Atmospheric turbulence causes scintillation
(i.e. fluctuations in the optical power at the receiver), beam
broadening, loss of spatial coherence, and fluctuations in the
angle of arrival. From an optical communications point of view, if
the incident signal is focused into a single mode fiber, which
spatially filters the beam, then the primary optical communications
impairment is fading. Fading is essentially, the loss of signal for
periods of 1-10 msec. Unlike RF communications, FSO has an
advantage in that inter symbol interference (ISI) is not an issue
because the atmosphere is not dispersive, whereas in RF
communications, multipath propagation leads to ISI. It is
anticipated that the capacity/margin tradeoff capability of the
ACOM will mitigate fading by using "burst communication" during
periods of clear FSO communication.
[0074] An atmospheric model that can be used in combination with
the communications model to calculate SNR and BER based on
atmospheric parameters such as the refractive index structure
constant C.sub.n.sup.2, the Greenwood frequency F.sub.g, and the
Fried parameter R.sub.0. (The Fried parameter is a measure of the
characteristic correlation length of a turbulence induce phase
distortions and the Greenwood frequency depends on both the
turbulence strength and cross wind speed and is associated with the
characteristic frequency of turbulence change.) These models will
be used to develop signal processing methods to recover the data
more reliably. The communications model will also include account
for the Doppler shift caused by transmitters and receivers that are
housed in aircraft, missiles, and satellites. Moving platforms are
considered in the DSP modeling discussed herein.
[0075] Receiver Rationale: The key elements for the ACOR are: 1)
coherent receiver, 2) approach the theoretical limit for homodyne
BPSK, and 3) detect and demodulate modulation formats of interest,
with the symbol demodulation and data recovery functions
implemented in re-configurable DSP blocks These requirements impose
the following conditions: 1) software reconfigurable receiver, 2) 9
photons/bit sensitivity for homodyne BPSK, 3) narrow line width
local oscillator, 4) high speed ADCs, 4) fast/highly paralleled
signal processing, and 5) optical phase lock loops to lock the
local oscillator.
[0076] Table 1 tabulates the theoretical receiver sensitivities and
theoretical laser line-widths for various modulation formats. The
first row of the table indicates the theoretical sensitivity that
is achievable with a BPSK, homodyne optical receiver. At 1 GSym/sec
the theoretical sensitivity is as low as -65 dBm, and the
sensitivity increases to -49 dBm at 40 GSym/sec. The first line of
Table 1 assumes that a local oscillator is phase locked to the
incoming signal with an optical phase lock loop that has a
bandwidth of 40 kHz and a local oscillator line width less than 1
kHz. As an example, the ACOM can operate homodyne BPSK at any lower
data rate, e.g. 155 Mb/s, which has a receiver sensitivity of -73
dBm. For a system designed to operate up to 50 GSym/sec, the
required receiver sensitivity can range from -73 dBm to -48 dBm by
merely changing the symbol rate under software control.
[0077] Table 1 also shows the theoretical performance with an
intradyne coherent receiver. See also A. W. Davis, M. J. Pettitt,
J. P. King, and S. Wright, "Phase Diversity Techniques for Coherent
Optical Receivers," Journal of Lightwave Technology, Vol. LT-5, No.
4, 1987, pp. 561-572. For all modulation formats other than BPSK
homodyne, the theoretical receiver sensitivity is the same for both
homodyne and heterodyne coherent detection. Homodyne BPSK provides
the maximum theoretical receiver sensitivity, but there is an added
complexity because the local oscillator must be phase locked to the
received signal. In operation, it may be prudent to operate the
receiver in BPSK homodyne when high receiver sensitivity is
required, but change modulation formats by moving to higher order
multi-level modulation formats when the bandwidth is limited. In
order to switch between different coherent detection modes, a
flexible receiver front-end that is used to support any mode of
coherent detection.
TABLE-US-00001 TABLE 1 Shot Noise Limited Receiver Sensitivities
and Local Oscillator Line-width Requirements ##STR00001##
##STR00002##
[0078] The receiver front-end is illustrated in FIG. 11. The
splitting ratio is governed by the Mach-Zehnder interferometer on
the input to the receiver. This device can be temperature-tuned to
provide a continuously variable coupling ratio. For k=0.5 the
receiver operates as a standard I/Q phase diversity receiver where
both PIN diode pairs are illuminated with the input and local
oscillator [LO] signals. This mode is used for QPSK and
higher-order modulation formats, which require I/Q detection. For
k=0, the receiver front-end acts as a single-phase receiver, only
one PIN diode pair is illuminated. This mode provides the highest
receiver sensitivity for BPSK operation by removing the 3 dB
slitting penalty on the input signal. In this manner, the receiver
front-end topology can be dynamically re-configured to match
modulation format.
[0079] For optimum sensitivity, the LO laser will be phase-locked
when receiving BPSK. The error signal for this loop can be derived
digitally within the Rx DSP. For QPSK and higher-order
constellations, the LO does not need to be phase-locked, and the
receiver operates in an intradyne mode. In this case, carrier
recovery is accomplished using a de-rotation stage in the Rx DSP.
Another advantage of the input MZ structure is that by operating
the device at a small coupling ratio k=.epsilon., it is possible to
implement a decision driven phase locked loop (see L. G. Kazovsky,
"Decision-Driven Phase-Locked Loop for Optical Homodyne Receivers:
Performance Analysis and Laser Linewidth Requirements," Journal of
Lightwave Technology, vol. LT-3, No. 6, December 1985, pp.
1238-1247) or a Costas loop. If the modulation bandwidth of the
lasers is limited, various methods of external frequency
translation can be employed with the intent of ultimately
integrating the device into the receiver front end. The receiver
front-ends can be made using fiber components or integrated into
substrate, such as LiNbO.sub.3.
[0080] The receiver DSP functions are critical for achieving the
agility required of the ACOM. FIG. 12 illustrates an intradyne
receiver with the digital signal processing architecture in place.
The output from the receiver front-end is detected with photodiodes
in a balanced receiver configuration and the signals are amplified
with programmable gain trans impedance amplifiers [TIAs]. The
output of the TIAs are digitized with high-speed ADCs. The digital
signals from the ADCs can be demultiplexed to a data rate that can
be handled by FPGAs. Implemented in FPGAs are digital signal
processing routines that recover the phase, the symbol rate (i.e.
clock) and the difference frequency between the local oscillator
and the received carrier. In the I/Q phase diversity receiver, the
constellation will rotate at the difference frequency between the
local oscillator and the received carrier frequency.
[0081] The parameters I and {circumflex over (Q)} represent the
non-rotating constellation. Both I and {circumflex over (Q)} are
passed into a decision circuit that recovers the transmitted data.
The recovered local oscillator frequency difference will be used to
control the local oscillator frequency. A greater reliance on DSP
for the carrier recovery enables more flexibility in shaping the
recovery loop response, and adapting it to compensate for system
impairments. The clock rate for the A/Ds and subsequent DSP
processing will be locked to the symbol rate. Current technologies
can generally operate with 1 sample per symbol at the highest
symbol rate. At lower symbol rates, this clock rate will be a
multiple of the symbol rate. By maintaining the high clock rate,
multiple samples per symbol can now be used to implement matched
filtering and equalization.
[0082] Materials Rationale: The choice of materials for the
receiver architecture is a strategic choice in terms of integration
of the receiver. It may be convenient to implement the receiver
front-end in lithium niobate, the ADC's in Silicon-Germanium, the
high speed photo detectors in Galium Arsenide, and the high speed
analog switch in Indium phosphide. This approach may not lend
itself to micro-miniaturization, due to the differing capabilities
of the processes and the use of different substrates, although
continued advances in the these technology may obviate this
concern.
[0083] Given the recent advances in InP HBT technology with
f.sub.t>600 GHz (see J. E. Kloeppel, "New material structure
produces world's fastest transistor," News Bureau University of
Illinois at Urbana-Champaign, Apr. 11, 2005) whereas the record for
SiGe was set in 2002 at f.sub.t.about.400 GHz, B.sub.vceo>4 V
compared to 2 V for SiGe, high B.sub.vcbo>8 V compared to 5 V
for SiGe, reduced current density at max F.sub.t, and excellent
linearity, InP HBT technology may make sense for developing mixed
signal circuits such as ADCs. Even though there are devices in SiGe
at 3 Bit ADCs at 40 GSa/sec and there are 20 GSa/sec devices, it
may be desirable to implement a high speed 10 GSa/sec InP ADC with
the intention of integrating the technology together with the high
speed photodetectors and TIA into a single chip. Since it is
possible to achieve f.sub.t.about.600 GHz in InP today, it is
conceivable that this technology can lead to ADCs that operate at
data rates on the order of 60 GSa/sec in the foreseeable
future.
TABLE-US-00002 TABLE 2 HBT Attributes for High Dynamic Range
DACs/ADCs (See C. Zolper, "Challenges and Opportunities for InP HBT
Mixed Signal Circuit Technology," International Conference on
Indium Phosphide and Related Materials 2003.) Attribute SiGe HBT
InP DHBT Breakdown voltage: <2 V, 5 V >4 V, >8 V Bvceo,
BVbco Gain at 20 GHz 22 (0.12 um .times. >20 dB (0.4 um .times.
11 um) 11 um MESA) >35 dB (0.4 um .times. 6 um TS) Substrate
cross talk moderate low Linearity (Third order Good excellent
intercept) 1/f noise corner 400 Hz ~ 1 KHz frequency 20 GHz Noise
Figure 1.4 dB 0.7 dB F.sub.t, f.sub.max (best reported) 350, 170;
270, 260 370, 280 (0.35 um .times. 5 um); (GHz) (0.12 um .times.
2.5 um) 300, 300 (0.4 um .times. 11 um) Current density at 2
MA/cm.sup.2 0.5 MA/cm.sup.2 max F.sub.t V.sub.be matching Excellent
excellent
[0084] Transmitter Rationale: Four options are illustrated in FIG.
12A for potential electro-optic modulators in the ACOT. The first
is a phase modulator, the second is a Mach-Zehnder (MZ), the third
is a combination of a phase and MZ, and the fourth is a dual
parallel MZ (also known as an I/Q optical modulator).
[0085] The Phase modulator can generate any PSK modulation format,
but it cannot generate QAM signals. On the other hand, the MZ
modulators cannot easily generate higher order PSK or QAM signals.
The last two candidates can generate any modulation format, but the
advantage to the I/Q modulator is that it is relatively easy to
generate any modulation format even though the insertion loss of
the modulator is slightly higher than that of the combination
phase/MZ modulators. Furthermore, the I/Q modulator allows for
easier implementation of the waveform generation electronics, in
that a large number of constellations allow I and Q to be generated
independently. Thus the I and Q data streams may be independent,
parallel processes. Both drive signals for the combination phase/MZ
modulator require knowledge of both I and Q simultaneously. (See
"Digital Modulation in Communication Systems--An Introduction,"
Application Note 1298 by Agilent Technologies, 2001.)
[0086] Three embodiments of the I/Q modulator section are
illustrated in FIG. 13: a) a standard I/Q modulator, b) a segmented
electrode version, and c) a parallel architecture. For the standard
I/Q modulator, each modulator arm is driven with a D/A and linear
amplifier. The modulators in b) and c) integrate the basic D/A
function with the optical modulation process. Multi-electrode
modulators have been pursued in the past for different purposes.
For this application, these structures offer several advantages
over more traditional designs:
[0087] The high-speed DAC is eliminated
[0088] Binary drivers can be used--the modulator drivers do not
need to amplify a multi-level electronic signal, thus their
performance within their linear range is less critical
[0089] The weighting of each bit may be individually tuned for a
desired constellation using low-speed A/Ds controlling the driver
output amplitude.
[0090] Simulations were conducted on the bandwidth tolerance and
expected performance of the three scenarios. The first type has a
bandwidth limit of 20 GHz, and it is driven at 40 GSym/sec with a
QPSK modulation format. The second version is a segmented electrode
design where the shortened electrodes have a higher bandwidth (for
this model we estimated a 50% improvement in the bandwidth.) The
third concept is one where a series of four parallel Mach-Zehnder
modulators are connected in series such that the total phase
response is in a ratio of 1, 1/2, 1/4, and 1/8. Each modulator is
driven to V.sub.x.
[0091] In the second column of FIG. 13, the eye diagrams of each of
the modulator concepts are illustrated. The cleanest performance is
with the third parallel structure. The third column of FIG. 13
illustrates the constellation diagrams for each modulator. Again,
the central constellation points for the third modulator are
clearly the best. The reason for the broadening of the central
constellation points in modulator types A and B is that the optical
output level depends on the previous symbol state due to the
limited bandwidth of the modulator. Similarly, the BER curves that
include modeling results for a QPSK homodyne system operating at 40
GSym/sec further indicate the improved performance shown in the eye
diagrams. Clearly, with an appropriately designed modulator with a
bandwidth of 40 GSym/sec, this modeling shows that it may be
possible to realize I/Q modulator concepts and performance that may
approach or exceed 80 GSym/sec.
[0092] Optical sub carriers can be used; however, there are
challenges with subcarrier implementation, such as mixing products
because of the nonlinear response of the electro-optic modulator,
multiple sub-carriers, which require wide band RF amplifiers unless
mixing occurs after RF amplification, and sub band filtering at the
receiver unless images are rejected electronically after detection.
Alternatively, the sidebands can be generated in software.
[0093] One of the challenges in developing optical transmission
architectures beyond 100 GHz is that the PD, TIA and ADC are
limited by state of the art electronic technology. An alternative
approach is the possibility of using optical ADCs where the ADC
function is incorporated into the optical component. The Terahertz
Optical Asymmetric Demultiplexer (TOAD) can provide signal
processing capabilities that far exceed the speeds of electronics
and other all optical switching technologies. The TOAD has been
demonstrated to operate at symbol rates as high as 250 GSym/sec, so
it may provide an option for ACOM technologies beyond the 40
GSym/sec.
[0094] The basic architecture for the ACOM that is a paradigm shift
from existing optical transport technology based on direct
detection. By adding two additional requirements: 1) the ability to
redefine the functionality of the ACOM "in-service", and 2) the
goal of integrating the ACOR we arrive at the guidelines that drive
the innovations. The receiver front-end is software -reconfigurable
to maintain optimum performance for different modulation formats.
The receiver components can be chosen with the intent of
integrating the components. The concept of "parallelism" leads to
novel ODAC designs for the ACOT, and the same concept leads to the
interest in exploring OADCs with the intent of ultimately moving
towards ACOM technology that exceed 100's of GSym/sec.
[0095] System Approach
[0096] The primary goals in this architecture are to provide agile
hardware that is not specific to a given modulation format or data
rate, and to exploit the flexibility inherent in digital signal
processing for optical communications. To this end, the ACOM can
employ thin optical front-ends, which allow the direct writing and
recovery of arbitrary I and Q information on the optical carrier.
Format- and rate-specific tasks such as waveform generation, symbol
recovery, and timing recovery are performed in using high-capacity
electronic processing. This approach is shown in FIG. 14.
[0097] The transmit optical transport unit generally includes a
laser and an electro-optical modulator. The Tx electronics include
the drive components and bias control for the modulator. The
receive optics optical transport unit generally includes a local
oscillator laser, coherent optical front-end, and high speed
detectors. These optics stages can be implemented using fiber
technology as integrated. The Rx Electronics variable gain TIA,
receive ADCs, and data interface from the ADCs to the first stages
of the Digital Signal Processing data-path will be described
further in following sections.
[0098] The present invention allows operation over a large range of
modulation formats and data rates through future software-driven
reconfiguration. It is expected that 20 Gb/s data throughput is
achievable with lower order modulation formats and 160 Gb/s data or
more of throughput is achievable with higher order modulation
formats and more capable electronics.
[0099] Agile Coherent Optical Transmitter (ACOT): In one ACOT
embodiment, the transmitter, shown in FIG. 15, a continuous wave
[cw] signal from the source laser is split into two paths for the
orthogonal polarizations. Each path contains a dual-arm I/Q
modulator which is driven from the transmit electronics block. The
transmit electronics processing block maps the incoming client data
to the desired modulation waveform in the signal processing
unit.
[0100] The embodiment in FIG. 15 uses two DACs and two linear
modulator drivers. This approach offers flexibility in waveform
generation, especially for systems at lower ultimate symbol rates.
At 10 GSym/s, for example, 6 bits or more resolution should be
easily achievable (for example, the PulseLink Inc. 6 bit, 12 GSym/s
PLK12106), and linear driver stages (for example, the Triquint
TGA4819). The scalability of these technologies, however, to 40
GSym/s while maintaining this flexibility may be limited by the
availability of high speed electronics.
[0101] An alternative embodiment, shown in FIG. 16, integrates the
D/A function into the modulator using a multiple-electrode
modulator. In this embodiment, each modulator arm has several
shorter electrodes as opposed to a single electrode. The electrode
lengths are weighted to produce different phase shifts.
[0102] Multi-electrode optical digital to analog converters [ODACs]
can be made from lithium niobate due to its large electro-optic
coefficients, high optical quality, and mature waveguide
technology, although other integrated technologies are possible. As
a starting point and performance baseline for future improved
designs, commercially available DQPSK modulators, such as from
COVEGA, can be used. Advanced modulators that employ cascaded RF
electrodes in the sub-MZIs can be also used in lieu of conventional
design.
[0103] FIG. 17 shows an embodiment of a 2-electrode design. Each
sub-MZI has 2 RF electrodes in series and a separate bias section.
The main MZI has a similar bias section, but is not shown to allow
a clear presentation of the sub-MZI details. Each RF electrode
imparts a predetermined amount of phase modulation and the
contributions from the two electrodes are added coherently. Each
electrode is therefore required to produce less modulation than
what would be necessary of a single electrode. The cascaded
electrode approach can thus allow for higher Vpi's of the cascaded
electrodes and their shorter interaction lengths. The latter result
in electrodes with lower RF loss, and, as such, broader bandwidth
due to a decreased roll-off of the frequency response.
[0104] A two-electrode structure is theoretically capable of
supporting up to 4 points per phase axis. Thus, the modulation
formats shown in Table 3 are possible.
TABLE-US-00003 TABLE 3 Modulation Format Support for 2-Element
Modulator Signal Amplitude Element phases Element Drive Format (I
and Q) (.phi.0, .phi.1) States {(.phi.1, .phi.0)} BPSK (1, -1)
.phi.0 + .phi.1 = .pi./2 {(1, 1), (-1, -1)} QPSK (1, -1) .phi.0 +
.phi.1 = .pi./2 {(1, 1), (-1, -1)} 8-ary PSK (1, 0.414, -0.414, -1)
(1.00, 0.5720) rad {(1, 1), (1, -1), (-1, 1), (-1, -1)} 16-QAM (1,
0.333, -0.333, -1) (0.955, 0.616) rad {(1, 1), (1, -1), (-1, 1),
(-1, -1)}
[0105] For BPSK and QPSK, the two electrodes are driven in phase,
and act as a single electrode. For 8-ary PSK and 16-QAM, two
additional drive states are included, which correspond to
.phi.1-.phi.0 and .phi.0-.phi.1. Higher constellations will require
more electrodes. An alternative hybrid approach, where a lower
resolution D/A drives each electrode can also be considered. Note
that a 2-bit D/A would not be able to support both 8-ary PSK and
16-QAM due to the different required bit weights. The ability to
individually adjust the phase value of each electrode via the drive
level is a primary advantage of this architecture.
[0106] Transmitter Output Power: The optical power budget for the
transmitter appears in Table 4. The output power of the transmitter
is limited by the maximum input power to the modulator, the
modulator loss, and the loss intrinsic to the modulation format.
Higher powers will require the inclusion of an amplifier following
the modulator. For high-order constellations, it is apparent that
amplification will most likely be necessary even if the I/Q
modulator itself were lossless, until higher levels of performance
and integration can be achieved.
TABLE-US-00004 TABLE 4 Transmitter Output Power Calculations Item
Loss/Power Source Laser 23 PM splitter -3.75 Modulator loss -5.5
PBC -0.5 CW power available (dBm) 13.25 Modulation Intrinsic Output
Format Format Loss (dB)* Power (dBm) BPSK 0 13.25 QPSK -3 10.25
8-PSK -5.33 7.92 16-QAM -5.563 7.687 32-QAM -7 6.25 64-QAM -6.69
6.56 NRZ -3 10.25 *Loss when implemented through I and Q amplitude
control. It is recognized that strictly phase modulation would have
0 dB intrinsic format loss for N-ary PSK
[0107] This power calculation illustrates the basic tradeoff
between agility and output power. A design based on a phase
modulator alone could make an output power goal of +20 dBm.
However, this modulator choice would limit the available modulation
formats to phase-only formats.
[0108] Transmitter Electronics: The basic transmit DSP
functionality is not especially complex. The primary function is to
convert the client data to the required line code and create the I
and Q data for the modulators. For lower data rates and more bits
per symbol it could also be desirable shape the date using a raised
cosine filter to improve transmission and ease demodulation. A
possible implementation for Phase 1 is shown in FIG. 18. In
addition to the DSP processor, a multiplexer is used to adapt the
625 MHz 96 bit wide data bus to a pair of 10 GHz streams to suit
the D/A converter timing requirements. Other critical functions are
the high frequency, low jitter clock generation. The D/A converters
have been described elsewhere in this document.
[0109] Agile Coherent Optical Receiver (ACOR): It is desirable that
the ACOR be a compact coherent receiver capable of near theoretical
sensitivity, while maintaining the flexibility to handle different
data rates and modulation formats and a receiver architecture which
includes the ability to reconfigure the optical front-end for
single-phase or I/Q operation. This architecture is shown in FIG.
19.
[0110] The primary goal with this architecture is to enable generic
I/Q demodulation while not sacrificing sensitivity for BPSK
homodyne operation. The ACOR can implement the receiver front-end
using fiber or substrate based components. Commercially deployable
fiber-based Mach-Zehnder filters with free-spectral ranges of 8 GHz
are available or the structure can be integrated into a packaged
lithium niobate device.
[0111] In the receive optical transport unit, the LO laser source
must be phase-locked to the incoming signal for optimum sensitivity
for BPSK reception. The linewidth of the laser sources used in the
proposed architecture is expected to be 1 kHz to 100 Hz range.
Given these linewidths, the OPLL must have a bandwidth on the order
of 10 kHz to maintain acceptable phase error. The phase error
signal can be generated as part of the Rx digital signal
processing: this loop bandwidth is well within the capabilities of
the digital electronics. As mentioned previously, the proposed
architecture can support different loop topologies such as the
decision-directed loop by adjusting the MZ to couple a small amount
of the input signal to the quadrature PIN pair. The OPLL bandwidth
for higher-order constellations is proportionately wider, with no
gain in sensitivity. Therefore, the LO laser does not need to be
phase-locked for higher-order constellations. For these
constellations the receiver will operate in an intradyne mode. True
heterodyne operation is also possible, but may be limited to lower
bit rates by the bandwidth of the electronics.
[0112] In FIG. 19, a single polarization channel is shown for
clarity: polarization diversity may be achieved by adding a
polarization-beam splitter after the polarization controller and
using the splitter to drive a second receiver. A polarization
controller can be used to minimize the complexity of the receiver.
Ultimately, the polarization control features can be integrated
into the receiver signal-processing unit.
[0113] Receiver Electronics: An exemplary embodiment of the
receiver electronics is shown in FIG. 20. The optical signals are
received by the differential photo-diode detectors in the optical
transport unit. The I and Q electrical signals are retrieved and
then amplified to a level sufficient to give a full-scale signal,
but no more, at the A/D converter. The A/D converter output rate
may be too high to feed directly to the DSP processor so it must be
de-multiplexed down to a rate that the logic inputs can handle. For
a synchronous bus, the rate is generally limited to below 1 GHz
with a rate of 625 MHz being more conservative. This requires a
16:1 demultiplexer. It is expected that electronics performance
will continue to improve. As such, the actual embodiment of the
function, such as the need to demultiplex may be diminished or
eliminated over time.
[0114] For higher data-rates, FPGAs may not provide an optimal
solution for the transmitter. Again, as with the transmitter it may
be best to define a new device using deep sub-micron CMOS, which is
optimized for this type of application. This could be full custom,
ASIC or a custom FPGA depending on how much investment and/or
return on investment is warranted. As with other functions, the
function can be enable separately or integrated with other
functions.
[0115] Additional electronics is generally required for clock
generation, gain control etc, as well as a general purpose or DSP
microprocessor for set-up, control and management of the receiver
subsystem. In combined transceiver, the same microprocessor could
control the transmitter as well as the receiver.
[0116] Receiver DSP: An overview of the receiver DSP processing is
shown in FIG. 21. The most critical functions are the control of
system gain, followed by the tracking of the modulated carrier
phase and frequency. The A/D sampling point is then also optimised
to center it with the data eye. The recovered clock and I/Q data
are then processed by the data-slicer/decision block to recover the
original data. As stated earlier it is expected that multiple high
performance FPGAs will be required. A more precise estimate will be
made early in the development program based on simulations and
trial mapping of this functionality required to available FPGA
devices.
[0117] ADC Development: The proposed system will include a 10
GSa/s, 6-bit ADC and a 6:96 DMUX. As an option, the companion 10
GSa/s, 6-bit DAC and 96:6 MUX will also be developed. The A/D and
optional D/A converters can use InP HBT process technology, which
for lower speeds would be a very low risk. For the DMUX and MUX, it
may be desirable to use a deep-submicron CMOS process, such as 0.13
um, although other process options are possible. 6 Gsa/s 6-bit DAC
are already available (Part No. RAD006 from Rockwell Scientific
Corp.), which can be used for lower data rates.
[0118] A 40 GSa/s, 4-bit ADC can be developed (see FIG. 22) for
higher data rates using InP HBT process technology but may be
somewhat more risky with current implementations. It requires the
use of several innovative circuit techniques, which have been
reported though not applied to this problem. Packaging of the
device would also be critical and custom multi-chip module would be
developed to incorporate the differential photo-diodes, TIA and
data-demux.
[0119] Laser Technology: The laser is a critical component for both
the receiver and transmitter. To facilitate a coherent
communications link, the laser used for the Tx signal source and
the Rx local oscillator should have a narrow linewidth, low noise,
high wavelength stability, and high power at high efficiency (see
Table 5). The laser design can employ an external cavity approach
pumping of Yb:Er doped phosphate glass with 980 nm beam. The output
wavelength can be tuned to any wavelength between 1528 to 1565 nm.
The lasers can be incorporated in package suitable for small form
factor transmitters.
TABLE-US-00005 TABLE 5 ACOM Laser Specifications Specification
Lower Speeds Higher Speeds Output Power 200 mW >500 mW
Efficiency 10% 20% Linewidth 1 kHz 100 Hz Stability 5 kHz 1 kHz
Noise (1 MHz to 40 GHz) -170 dB/Hz Size 1.5'' .times. 1.5'' .times.
0.5''
[0120] Solid state lasers have very narrow line widths compared to
other lasers like semiconductor lasers. Since the line width of the
laser is very low, the measured line width will depend on the
stability of the laser/locker over 1 second which is typically
required for such systems. The short term stability is limited by
the noise of the structure mainly induced by vibration.
[0121] The important criteria for high performance of the locker is
temperature uniformity, stress minimization and noise immunity. The
TEC controller is able to control the temperature of the locker
assembly to about 0.007 deg C. over >6 hours and this enables in
very high stability of the locker. The locker is undergoing further
improvement which involves increasing the accuracy and finesse of
the locker etalon as well as improvements to temperature control
and vibration isolation as well as putting the locker under
vacuum.
[0122] The primary capabilities for a high-capacity
software-defined optical communications system are:
[0123] Coherent optical processing exploiting both amplitude and
phase information
[0124] High capacity, e.g., 10-40 GSym/sec
[0125] Data rate and modulation constellation agility: support
software-controlled data rate and modulation format reconfiguration
on a common hardware platform
[0126] Near-theoretical receiver sensitivity
[0127] The present invention exploits the basic system approach
taken in advanced RF communications systems and provides novel
implementations that enable robust optical communications. This
approach is shown in FIG. 23. The architecture provides an agile
hardware that is not specific to a given modulation format or data
rate, and to exploit the flexibility inherent in digital signal
processing for optical communications. The ACOM will employ thin
optical front-ends, i.e., optical transport units, which allow the
direct writing and recovery of arbitrary I and Q information on the
optical carrier, as well as generic client interfaces. Format- and
rate-specific tasks such as waveform generation, symbol recovery,
and timing recovery are performed in using high-capacity electronic
processing in software definable signal processing units.
[0128] The ACOM is architected to work in conjunction with optical
wavelength translation stages to form a complete optical
communications system. These translation stages are shown in gray
in FIG. 23.
[0129] There currently exists a mismatch between the bandwidth
capability of optoelectronic components and electronic processing
components, which may be overcome as technology advances.
Modulators and detectors are commercially available with bandwidths
exceeding 40 GHz. However, advanced digital signal processing
components cannot currently achieve such rates. A/D and DACs are
now becoming available up to 20 Gbps. DSP operations must currently
be performed at sub-GHz rates. For this reason, the electronics
used at the physical layer in optical systems remains comparatively
simple: one prime example is in data recovery, where the vast
majority of deployed systems employ simple binary hard decision.
The proposed architecture deals with this mismatch in several
ways:
[0130] Processing algorithms for the highest symbol rates will use
a minimum number of samples per symbol. It is expected that only
one sample per symbol will be available.
[0131] Sampling rates are held constant as the symbol rate changes,
allowing lower symbol rates to exploit greater processing
capability.
[0132] Data is broken into parallel data paths as soon as possible.
Both the optoelectronics and signal processing blocks are optimized
to maintain independence between data paths.
[0133] Advanced optical structures which integrate the A/D and D/A
processes will be explored.
[0134] The approach taken in the present invention allows operation
over a large range of modulation formats and data rates through
future software-driven reconfiguration. The exemplary modulation
formats and data rates are shown in FIG. 24.
[0135] Description of the Tracking Loops
[0136] The tracking loops for time, frequency and phase shown in
FIGS. 25 through 28 are described. It is to be noted that these
designs have been used as baseline approaches for the purpose of
estimating and bounding technological limits on performance (such
as samples/symbol and bits/sample limits arising from the high
clock rates). These functions can be further analyzed and optimized
as technology advances. The present discussion will cover an
illustrative design and where appropriate indicate some areas of
further improvement.
[0137] All of the estimation algorithms/architectures share a
common canonical form. This form is sometimes referred to as the
extended Kalman filter and is well known. The principle of
operation is to approximate a recursive maximum likelihood
estimation procedure. Maximum likelihood [ML] methods are known to
be efficient in the sense of minimizing parameter variance (jitter
power). In addition, the ML method provides insight into the
appropriate error signal computation when estimation is performed
in the presence of random data. This is the manner in which the
function in FIG. 29 was determined.
[0138] The procedure is to attempt to adjust the parameter under
discussion (for example phase) in such a way that at each update
the parameter estimate moves in a direction of increasing
likelihood of the observations. The likelihood is the probability
of observing the received signal given a particular parameter
value. Thus, the error signal is taken to be the derivative with
respect to the parameter. This error signal is then input to a
filter structure that is designed to match the dynamics expected to
be affecting the parameter. In the case of phase estimation, the
dynamics include a step function (fixed phase shift between signal
and reference oscillator) and a ramp (a frequency offset between
signal and reference).
[0139] Thus the canonical algorithm decomposes into 3 parts: 1) A
gradient computation for determining the direction of increasing
likelihood, 2) A dynamical system suitable to the dynamics expected
and 3) a feedback connection that realizes the recursion. Of
course, it is the feedback that makes the loop track a parameter.
The closed loop is also capable of noise reduction because the
closed loop bandwidth is smaller than the open loop bandwidth. On
the other hand feedback introduces issues of stability and
transport delay arising from computation. Techniques for organizing
the loop in a manner that circumvents transport delay have been
developed and are applicable to the present problem.
[0140] Phase and Frequency Tracking Subsystem--In order to support
phase modulation, high order QAM, or higher data rates, it is
necessary to provide signal phase recovery with minimal error. This
must be accomplished in the presence of frequency offsets on the
order of 1 GHz. The latter requirement implies the use of frequency
error feedback in the phase locked loop because such large
frequency offsets are beyond the loop pull-in capability. Finally,
the phase acquisition and tracking must be possible in the presence
of data modulation. For PSK operation, in particular BPSK and QPSK,
a phase detector that is transparent to the data is available (FIG.
29). This detector is nonlinear but can be easily implemented in a
look-up table [LUT].
[0141] Higher order system design consideration--QAM--When
considering modulations such as QAM, whose signal constellations
are not periodic in angle, the phase detection process exhibits
"self-noise" created by the data randomness. To optimize
performance in this more ambitious circumstance two statistical
approaches are available: 1) the use of generalized phase detector
based on the maximum average likelihood principal (ALP) and 2) a
generalized likelihood processor (GLP) approach in tandem with the
first approach. The GLP is a decision directed process and it
performs adequately if the occurrence of decision errors is small
and the lock condition is maintained. Thus, the GLP cannot be used
by itself without a means of bootstrapping. The combination of an
ALP and GLP provides the benefit of GLP (which exhibits lower
residual phase noise) while insuring the ability to lock and relock
automatically. In the dual processing approach the loop
periodically enters the ALP mode. As an illustration of a typical
system design at a 10 GHz symbol (and sampling) rate, one might
provide a 1.5 microsecond period in which 100 nS is dedicated to
ALP while the remaining 1.4 .mu.S are operated in GLP mode. During
the ALP period the demodulated symbols will exhibit a somewhat
greater jitter. This can be mitigated by the use of interleaving.
Continuing with the example of a 1.4 .mu.S period, a square 10K by
10K (100 megasymbols) interleaver arrangement would distribute 1
degraded symbol per 15-symbol string. A simple (15,11) Hamming code
could then be used to correct the occasional error. This is merely
an illustration of a system approach and further optimization of
the design is warranted.
[0142] Returning to the basic processing it is shown in FIGS. 30-32
that phase tracking is feasible under stringent technological
constraints and maximum signal degradation. In particular, the
system must operate with limited A/D resolution, extreme frequency
offset and low SNR. To illustrate the robustness of the digital
phase recovery subsystem a bit-exact simulation has been developed.
The simulation incorporates a variable A/D that can be programmed
to any desired number of bits. The simulation conditions
represented in the figures are: SNR=5 dB, Frequency Offset=1 GHz
and A/D bit-width of 3 bits. The SNR is equal to the Eb/No ratio of
an equivalent RF system. One sample per data symbol is also
assumed. This represents a lower limit of available technology.
[0143] In FIGS. 30 and 31 it may be seen that the system
acquisition time is dominated by the frequency acquisition time
(the AFC action). This process has a time constant .tau. of
approximately 3 microseconds (100,000 samples at 10 GHz corresponds
to 3.tau.). This time can be reduced by amplification of the
frequency error at the expense of increased residual phase noise.
The bandwidth of the phase tracking process is wider than the
frequency acquisition bandwidth (thus, on the order of MHz). This
is consistent with the need to "track-out" local laser phase
noise.
[0144] OPTIMIZATION--In 1955 Jaffe and Rechtin (Jaffe, R., and E.
Rechtin, "Design and Performance of Phase-Lock Circuits Capable of
Near-Optimum Performance over a Wide Range of Input Signal and
Noise Level," IRE Trans. Information Theory, Vol. IT-1, pp. 66-76,
March 1955) published a technique (essentially a Wiener technique)
for optimizing a control loop operating with conflicting criteria:
minimal dynamic error and minimal fluctuation due to noise. The
former is reduced by increasing loop bandwidth while reducing it
minimizes the latter. The technique was presented in the s-plane
(continuous system) but it has also been adapted successfully to
the z-plane (discrete time). An interesting application to the
present problem entails the solution of the optimization problem
including the phase noise process of the light sources, a component
of the noise not present in the classical problem. Here it is known
that it possible to remove a portion of the line-width dependent
phase noise by increasing PLL bandwidth. Because residual phase
noise will be seen to be a limiting factor on attainable high-order
constellations, this is a central theoretical consideration, which
can also be addressed by bounding performance.
[0145] Another aspect of the problem is nonlinear A/D conversion.
Not only is the conversion likely to be coarse (perhaps as little
as 3 or 4 bits), but also the A/D is now part of the "channel".
This means that the phase measurement is distorted and the error
characteristic shown in FIG. 29 is appropriate for perfect
digitization--it is only approximate when the nonlinear distortion
mentioned is present. It is possible to define an error function
specific to the A/D output. This function would then provide a
signal that serves to estimate and correct the phase at the output
of the A/D rather than at the input. The function can be
implemented in a look-up table (LUT), which provides a
straightforward approach.
[0146] CLOCK RECOVERY--The second critical parameter that must be
recovered is sampling time phase. Robust algorithms capable of
digital clock tracking utilizing low resolution A/D (3 bits as a
bounding case) and assuming only 1 sample per symbol are desired.
With the limitation of 1 sample it becomes necessary to employ
dither techniques rather than relative amplitude (early-late)
comparator techniques available when multiple staggered samples are
available. Further, it is assumed for the sake of establishing
basic feasibility that the high-speed A/D clock phase cannot be
divided into controlled sub-phases digitally. This implies that the
clock must be modulated to create a sampling phase sensitive error
signal. Specifically, the clock can be frequency modulated by a
local square wave.
[0147] As indicated in FIG. 33, the amplitudes of samples taken
during the low frequency shift intervals are multiplied by -1 while
during the high frequency shift they are multiplied by 1. The
alternation between low and high shifts is at a rate much higher
than the tracking loop bandwidth so that the effective (mean) value
of these polarity modulated samples is a measure of the imbalance
between amplitudes taken "early" and those taken "late". In other
words, a phase error signal is created.
[0148] The clock recovery process has been tested via bit exact
simulation of the algorithm operating on randomly modulated QPSK
signals. (The driving binary data is random.) Once again the
question of mathematical optimization of the loop parameters
presents itself. In the interest of time and space this analysis is
deferred and we present here only a demonstration of the
performance possible based on modest empirical optimization.
[0149] FIGS. 33 and 34 show the limiting cases of low SNR (5 dB)
and noise free operation. The conditions are given by a clock
offset of 20% of a symbol period. The plots show the phase
adjustment to the local clock in response to the algorithm. The
residual jitter is seen to be approximately 5%. Note that in the
noise free case the coarse quantization (3 bit A/D) limits the
performance. In fact the behavior of the algorithm is sensitive to
the location of A/D quantization levels (i.e., the scaling of the
input vs. the A/D reference voltage). It is desirable to have an
A/D code threshold near the peak of the signal. The system will
employ a digital AGC loop that includes the ADCs within the loop so
that the optimal scaling can be maintained. When the noise is
significant this sensitivity is lessened considerably because of
the dithering effect. Moreover, the loop response time is actually
shorter in the high noise case.
[0150] Timing and Phase Jitter Losses
[0151] It has been shown that phase and frequency can be recovered
digitally even under stringent implementation limits. We now
characterize predicted system behavior including final (uncoded)
data decisions. To this end, simulation of the demodulation process
in conjunction with time synchronization and phase recovery has
been performed. The results are summarized in the FIGS. 35-37.
[0152] The conditions of the simulation are: variable SNR, 3-bit
A/D or 6-bit A/D, frequency offset 1 GHz. It may be seen that a
significant portion of the departure from ideal performance is
attributable to coarse A/D conversion. At 3 bits the system loss is
2.3 dB while at 6 bits the loss drops to 0.68 dB.
[0153] Illustrating the Potential for 64 QAM
[0154] FIGS. 36-37 show the feasibility of higher order
constellations such as 64 QAM. The figures are not typical of
system behavior under extreme conditions as in the previous
figures, but rather serve to indicate that, at least, given the
availability of 6-bit devices, high order constellations may be
achievable. The results in FIGS. 36 and 37 show basic limits of A/D
and data-induced phase noise that would be faced when employing
M'ary QAM.
[0155] Polarization Tracking
[0156] Polarization tracking may be effected using the exact same
canonical structure as described above. In this case the parameter
space is multidimensional and comprises the polarization parameters
and the signal phase parameters that are coupled with the
polarization parameters. It is entirely feasible to construct
digital phase and polarization estimators that operate on diversity
branches in both the polarization domain (polarization beam
splitting and orthogonalization) and the temporal domain (I and Q).
The use digital processing permits the exploitation of diversity
processing and removes nonlinear and difficult to calibrate optical
components from the design replacing them with stable computational
equivalents.
[0157] It will be appreciated that the present invention provides
for improved transmission systems with increased reliability and
performance. Those of ordinary skill in the art will further
appreciate that numerous modifications and variations that can be
made to specific aspects of the present invention without departing
from the scope of the present invention. It is intended that the
foregoing specification and the following claims cover such
modifications and variations.
* * * * *