U.S. patent application number 13/720507 was filed with the patent office on 2013-05-23 for method and apparatus for performing optical imaging using frequency-domain interferometry.
This patent application is currently assigned to THE GENERAL HOSPITAL CORPORATION. The applicant listed for this patent is THE GENERAL HOSPITAL CORPORATION. Invention is credited to Brett Eugene Bouma, Johannes Fitzgerald De Boer, Guillermo J. Tearney, Seok-Hyun Yun.
Application Number | 20130128274 13/720507 |
Document ID | / |
Family ID | 34590115 |
Filed Date | 2013-05-23 |
United States Patent
Application |
20130128274 |
Kind Code |
A1 |
Yun; Seok-Hyun ; et
al. |
May 23, 2013 |
METHOD AND APPARATUS FOR PERFORMING OPTICAL IMAGING USING
FREQUENCY-DOMAIN INTERFEROMETRY
Abstract
An apparatus and method are provided. In particular, at least
one first electro-magnetic radiation may be provided to a sample
and at least one second electro-magnetic radiation can be provided
to a non-reflective reference. A frequency of the first and/or
second radiations varies over time. An interference is detected
between at least one third radiation associated with the first
radiation and at least one fourth radiation associated with the
second radiation. Alternatively, the first electro-magnetic
radiation and/or second electro-magnetic radiation have a spectrum
which changes over time. The spectrum may contain multiple
frequencies at a particular time. In addition, it is possible to
detect the interference signal between the third radiation and the
fourth radiation in a first polarization state. Further, it may be
preferable to detect a further interference signal between the
third and fourth radiations in a second polarization state which is
different from the first polarization state. The first and/or
second electro-magnetic radiations may have a spectrum whose mean
frequency changes substantially continuously over time at a tuning
speed that is greater than 100 Tera Hertz per millisecond.
Inventors: |
Yun; Seok-Hyun; (Cambridge,
MA) ; Bouma; Brett Eugene; (Quincy, MA) ;
Tearney; Guillermo J.; (Cambridge, MA) ; De Boer;
Johannes Fitzgerald; (Amstelveen, NL) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
THE GENERAL HOSPITAL CORPORATION; |
Boston |
MA |
US |
|
|
Assignee: |
THE GENERAL HOSPITAL
CORPORATION
Boston
MA
|
Family ID: |
34590115 |
Appl. No.: |
13/720507 |
Filed: |
December 19, 2012 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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13191885 |
Jul 27, 2011 |
8355138 |
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13720507 |
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12795529 |
Jun 7, 2010 |
8384909 |
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13191885 |
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10577562 |
Apr 27, 2006 |
7733497 |
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PCT/US2004/029148 |
Sep 8, 2004 |
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12795529 |
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60514769 |
Oct 27, 2003 |
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Current U.S.
Class: |
356/450 |
Current CPC
Class: |
G02F 1/093 20130101;
H01S 3/08009 20130101; A61B 5/0066 20130101; G01J 3/021 20130101;
G01B 2290/45 20130101; G02F 1/11 20130101; G01J 3/0264 20130101;
H01S 3/0071 20130101; G01B 2290/70 20130101; A61B 5/0059 20130101;
G01B 9/0209 20130101; G01B 9/02043 20130101; A61B 5/7257 20130101;
G01B 9/02075 20130101; H01S 5/1025 20130101; H01S 5/146 20130101;
H01S 5/5045 20130101; H01S 5/14 20130101; G01B 9/02002 20130101;
G01J 3/02 20130101; G01B 9/02083 20130101; G01B 9/02081 20130101;
H01S 3/0078 20130101; H01S 5/0078 20130101; G01J 3/0218 20130101;
G01N 21/4795 20130101; G01J 3/453 20130101; H01S 5/141 20130101;
G01J 9/0215 20130101; G01B 9/02004 20130101; G02B 6/3604 20130101;
H01S 3/08063 20130101; G01B 9/02091 20130101; H01S 3/105 20130101;
G02B 27/48 20130101; G01J 3/0208 20130101; G01B 9/02084 20130101;
H01S 5/0071 20130101; G02B 26/12 20130101; H01S 3/1068
20130101 |
Class at
Publication: |
356/450 |
International
Class: |
G01B 9/02 20060101
G01B009/02 |
Claims
1-93. (canceled)
94. An apparatus, comprising: at least one first arrangement
configured to provide a radiation which includes at least one first
electro-magnetic radiation directed to a sample and at least one
second electro-magnetic radiation directed to a reference, wherein
a frequency of the radiation provided by the at least one first
arrangement varies over time; and at least one second arrangement
configured to detect an interference between at least one third
radiation associated with the at least one first radiation and at
least one fourth radiation associated with the at least one second
radiation to generate an electrical first signal, wherein the at
least one second arrangement is configured to obtain a second
signal associated with at least one phase of at least one frequency
component of the electrical signal, and wherein the at least one
second arrangement includes a computing arrangement which compares
the second signal to at least one particular information.
95. The apparatus according to claim 1, wherein the at least one
second arrangement is configured to determine a third signal
associated with at least one further phase of at least one further
frequency component of the electrical signal, and wherein the at
least one particular information is the third signal.
96. The apparatus according to claim 1, wherein the at least one
second arrangement is configured to determine a third signal
associated with at least one further phase of at least one further
frequency component of a further electrical signal, the further
electrical signal being different from the electrical signal, and
wherein the at least one particular information is the third
signal.
97. The apparatus according to claim 96, wherein the electrical
signal and the further electrical signal are obtained at different
times.
98. The apparatus according to claim 96, wherein the electrical
signal and the further electrical signal are obtained at different
locations of the sample.
99. The apparatus according to claim 94, further comprising at
least one third arrangement is configured to generate a third
signal associated with the radiation, wherein the at least one
second arrangement is configured to provide the particular
information based on the third signal.
100. The apparatus according to claim 99, wherein at least one
third arrangement includes an interferometric arrangement.
101. The apparatus according to claim 94, further comprising a
third arrangement configured to receive at least one portion of the
radiation and provide a further radiation so as to generate a third
signal, wherein at least one of the second signal or the particular
information is associated with the further radiation.
102. The apparatus according to claim 94, wherein the at least one
second arrangement configured to detect a third signal
corresponding to the electrical signal between the at least one
second and third radiations, and wherein the second signal is
associated with at least one phase of at least one frequency
component of an additional electrical signal associated with the
third signal.
103. A method comprising: providing a radiation which includes at
least one first electro-magnetic radiation directed to a sample and
at least one second electro-magnetic radiation directed to a
reference, wherein a frequency of the radiation varies over time;
detecting an interference between at least one third radiation
associated with the at least one first radiation and at least one
fourth radiation associated with the at least one second radiation
to generate an electrical first signal; obtaining a second signal
associated with at least one phase of at least one frequency
component of the electrical signal; and comparing the second signal
to at least one particular information.
104. The method according to claim 103, further comprising
determining a third signal associated with at least one further
phase of at least one further frequency component of the electrical
signal, and wherein the at least one particular information is the
third signal.
105. The method according to claim 104, further comprising:
determining a third signal associated with at least one further
phase of at least one further frequency component of a further
electrical signal, the further electrical signal being different
from the electrical signal, wherein the at least one particular
information is the third signal.
106. The method according to claim 105, wherein the electrical
signal and the further electrical signal are obtained at different
times.
107. The method according to claim 105, wherein the electrical
signal and the further electrical signal are obtained at different
locations of the sample.
108. The method according to claim 104, further comprising
generating a third signal associated with the radiation; and
providing the particular information based on the third signal.
109. The method according to claim 103, wherein the second signal
is an interferometric signal.
110. The method according to claim 103, further comprising:
receiving at least one portion of the radiation and provide a
further radiation, wherein at least one of the second signal or the
particular information is associated with the further
radiation.
111. The method according to claim 103, further comprising
detecting a third signal corresponding to the electrical signal
between the at least one second and third radiations, wherein the
third signal is associated with at least one phase of at least one
frequency component of an additional electrical signal associated
with the third signal.
112. A system comprising: at least one arrangement configured to
(i) detect an interference between at least one first radiation
associated with a first split portion of a radiation received from
a sample and a second split portion of the radiation received from
a reference and (ii) to generate an electrical first signal
associated with the interference, wherein a frequency of the
radiation varies over time, and wherein the at least one
arrangement is configured to obtain a second signal associated with
at least one phase of at least one frequency component of the
electrical signal, and wherein the at least one arrangement
includes a computing arrangement which compares the second signal
to at least one particular information.
113. The apparatus according to claim 112, wherein the at least one
first arrangement is configured to detect a third signal
corresponding to the electrical signal between the first and second
split portions of the radiation, and wherein the second signal is
associated with at least one phase of at least one frequency
component of an additional electrical signal associated with the
third signal.
Description
CROSS REFERENCE TO RELATED APPLICATION(S)
[0001] The present application is a continuation of U.S. patent
application Ser. No. 13/191,885 filed on Jul. 27, 2011, which is a
continuation of U.S. patent application Ser. No. 12/795,529 filed
on Jun. 7, 2010, which is a continuation of U.S. patent application
Ser. No. 10/577,562 filed Apr. 27, 2006, which issued as U.S. Pat.
No. 7,733,497 issued on Jun. 8, 2010, which is a national phase of
PCT/US2004/029148 filed on Sep. 8, 2004, the entire disclosures of
which are incorporated herein by reference. This application also
claims priority from U.S. Provisional Patent Application Ser. No.
60/514,769 filed on Oct. 27, 2003, the entire disclosure of which
is incorporated herein by reference.
FIELD OF THE INVENTION
[0002] The present invention relates generally optical imaging, and
more particularly to method and apparatus for performing optical
imaging using frequency-domain interferometry.
BACKGROUND OF THE INVENTION
[0003] As is known in the art, optical interferometric
reflectometry is a powerful tool for performing non-invasive,
high-resolution (.about.10 .mu.m), cross-sectional imaging of a
biological or other sample, to visualize micro-structural optical
properties such as reflection, absorption, scattering, attenuation,
birefringence, and spectroscopic analysis. There are a number of
interferometric imaging techniques that are known in the art. These
techniques in general can be divided into two major categories: (i)
time-domain technique, and (ii) frequency-domain technique.
[0004] Low coherence interferometry ("LCI") is one of the
time-domain techniques. This technique uses a scanning system to
vary the reference arm length and acquire the interference signal
at a detector. Then, the fringe pattern is demodulated to obtain
the coherence envelope of the source cross correlation function.
Optical coherence tomography ("OCT") is a technique for obtaining
two- or three-dimensional images using LCI. OCT is described in
U.S. Pat. No. 5,321,501 issued to Swanson et al. Multiple variants
of the OCT techniques have been described, but many suffer from
less than optimal signal to noise ratio ("SNR"), resulting in
non-optimal resolution, low imaging frame rates, and poor depth of
penetration. Power usage is a factor in such imaging techniques.
For example in ophthalmic uses, only a certain number of milliwatts
of power are tolerable before thermal damage can occur. Thus,
boosting power is not feasible to increase SNR in such
environments. Nevertheless, it would be desirable to have an
imaging method with superior SNR without appreciably increasing
power requirements.
[0005] Insufficient SNR can also prevent the OCT technique from
being used at a high frame rate which is important to avoid motion
artifacts and overcome the short measurement time window available,
for example, for in-vivo vascular imaging. Therefore, a way to
improve SNR and imaging speed (e.g., the frame rate) is desired.
Spectral interferometry, or spectral radar, is one of the
frequency-domain imaging techniques. In spectral radar, the real
part of the cross spectral density of sample and reference arm
light is measured with a spectrometer. Depth profile information
can be encoded on the cross-spectral density modulation.
[0006] The use of spectral radar concepts to increase SNR of LCI
and OCT has been described previously. This technique uses a charge
coupled device ("CCD") with a large number of pixels (an order of
1,000) to reach scan ranges on the order of a millimeter. The fast
readout of the CCD device makes high-speed imaging possible.
[0007] There are, however, a number of disadvantages associated
with using a CCD device. First, CCD devices are relatively
expensive compared to a single-element photo-receiver. Secondly,
the previously described method uses a single CCD to acquire the
data. Since the charge storage capacity is limited, it requires a
reduction of the reference arm power to approximately the same
level as the sample arm power, giving rise to auto correlation
noise on the sample arm light. In addition, since no carrier is
generated, the 1/f noise will dominate the noise in this system.
Thirdly, even with the short integration times of state of the art
CCD technology, phase instabilities in the interferometer reduce
fringe visibility of the cross spectral density modulation. This
shortcoming makes the technique vulnerable to motion artifacts.
[0008] Coherent frequency-modulated continuous-wave reflectometry
(C-FMCW) is another frequency domain technique known in the art.
U.S. Pat. Nos. 5,956,355 and 6,160,826 issued to Swanson et al.
describes an optical imaging method and apparatus using this
technique. The previously described imaging method is based on
using a continuously-tuned single-frequency laser as an optical
source. The tuning wavelength range is required to be several tens
of nanometers to achieve ranging resolution of less than 100
microns. The instantaneous linewidth of the laser must be less than
approximately 0.1 nm to achieve a detection range on the order of
1.0 mm. The tuning rate should be greater than 10 kHz for high
speed (e.g., video-rate) imaging. Although an external-cavity
semiconductor laser can be configured to achieve mode-hop-free
single-frequency tuning over several tens of nanometer, the tuning
rate has been less than 1 Hz due to stringent requirement on
mechanical stability. A way to overcome this speed difficulty is
preferable.
[0009] It would, therefore, be desirable to provide a system and
method to overcome the source availability and scan speed
shortcomings of conventional LCI and OCT.
SUMMARY OF THE INVENTION
[0010] In accordance with exemplary embodiments of the present
invention, an exemplary optical frequency domain imaging ("OFDI")
system can include a multiple-frequency-mode (or multiple
longitudinal or axial-mode) wavelength-swept laser source optically
coupled to an interferometer containing a sample under study. The
system can further include an arrangement which is configured to
produce interferometric signals in quadrature between light
reflected from a sample and a reference light and a detector
disposed to receive the interferometric signals.
[0011] With such exemplary particular arrangement, an OFDI system
which can operate with source powers that are relatively low
compared with source powers of conventional systems and/or which
operate at acquisition rates which are relatively high compared
with acquisition rates of conventional systems may be provided. The
use of a swept source results in an imaging system having reduced
shot noise and other forms of noise which allows for much lower
source powers, or much higher acquisition rates than conventional
systems. This can lead to an increased detection sensitivity which
results in the ability to provide real time imaging. Such imaging
speed can assist practitioners in gastrointestinal, ophthalmic and
arterial imaging fields, where motion artifacts are a continuing
problem. By increasing a frame rate while maintaining or improving
the signal to noise ratio such artifacts can be minimized or in
some cases eliminated. Exemplary embodiments of the present
invention may also enable the screening of large areas of tissues
with OFDI and allows enables the use of clinically viable screening
protocols.
[0012] In one exemplary embodiment of the present invention, the
wavelength-swept laser can be provided that may use an optical
band-pass scanning filter in the laser cavity to produce a
rapidly-swept multiple-frequency-mode output. By using an optical
band-pass scanning filter in the laser cavity, it is not necessary
to tune the laser cavity length to provide synchronous tuning of
the laser spectrum. In other words, it does not require tuning the
longitudinal cavity mode of the laser at the same rate as the
center wavelength of the laser.
[0013] In another exemplary embodiment of the present invention,
the detector can be a dual-balanced receiver disposed to accept
interferometric signals and to suppress the relative intensity
noise in the interferometric signals.
[0014] The gain in signal-to-noise ratio ("SNR") according to an
exemplary embodiment of the present invention is advantageous over
time-domain approaches such as OCT via a performance of the signal
processing in the Fourier-domain. The SNR enhancement is by a
factor of N, the ratio of the depth range to the spatial
resolution. The enhancement factor N can reach a few hundreds to
several thousand. This increase in SNR enables the imaging by a
factor of N times faster, or alternatively allows imaging at the
same speed with a source that has N times lower power. As a result,
the exemplary embodiment of the present invention overcomes two
important shortcomings of conventional LCI and OCT, e.g., source
availability and scan speed. The factor N may reach more than
1,000, and allows construction of OFDI systems that can be more
than three orders of magnitude improved from OCT and LCI technology
currently in practice.
[0015] The gain in SNR is achieved because, e.g., the shot noise
has a white noise spectrum. The signal intensity present at the
detector at frequency .omega. (or wavelength .lamda.) contributes
only to the signal at frequency .omega., but the shot noise is
generated at all frequencies. By narrowing the optical band width
per detector, the shot noise contribution at each frequency can be
reduced, while the signal component remains the same.
[0016] Exemplary embodiments according to the present invention
improves current data acquisition speeds and availability of
sources compared with OCT. Shot noise is due to the statistical
fluctuations of the current that are due to the quantized or
discrete electric charges. The reduction of shot noise allows for
much lower source powers or much higher acquisition rates.
Limitations in current data acquisition rates (.about.4 frames/sec)
are imposed by available source power and availability of fast
mechanisms for scanning delay. An increase in the sensitivity of
the detection by a factor of 8 would allow real time imaging at a
speed of about 30 frames per second. An increase of the sensitivity
by a factor of about 1,000-2,000 allows for the use of sources with
much lower powers and higher spectral bandwidths which are readily
available, cheaper to produce, and can generate higher resolution
OFDI images.
[0017] For ophthalmic applications of OFDI, efficient detection
preferably allows for a significant increase of acquisition speed.
One limitation in ophthalmic applications is the power that is
allowed to enter the eye according to the ANSI standards
(approximately 700 microwatts at 830 nm). Current data acquisition
speed in ophthalmic applications is approximately 100-500 A-lines
per second. The power efficient detection technique of the present
invention would allow for A-line acquisition rates on the order of
about 100,000 A-lines per second, or video rate imaging at about
3,000 A-lines per image.
[0018] To achieve at least some of the goals of the present
invention, an apparatus and method according an exemplary
embodiment of the present invention are provided. In particular, at
least one first electro-magnetic radiation may be provided to a
sample and at least one second electro-magnetic radiation can be
provided to a non-reflective reference. A frequency of the first
and/or second radiations varies over time. An interference is
detected between at least one third radiation associated with the
first radiation and at least one fourth radiation associated with
the second radiation. Alternatively, the first electro-magnetic
radiation and/or second electro-magnetic radiation have a spectrum
which changes over time. The spectrum may contain multiple
frequencies at a particular time. In addition, it is possible to
detect the interference signal between the third radiation and the
fourth radiation in a first polarization state. Further, it may be
preferable to detect a further interference signal between the
third and fourth radiations in a second polarization state which is
different from the first polarization state. The first and/or
second electro-magnetic radiations may have a spectrum whose mean
frequency changes substantially continuously over time at a tuning
speed that is greater than 100 Tera Hertz per millisecond.
[0019] In one exemplary embodiment of the present invention, the
third radiation may be a radiation returned from the sample, and
the at least one fourth radiation is a radiation returned from the
reference. The frequency of the first, second, third and/or fourth
radiation may be shifted. An image can be generated based on the
detected interference. A probe may be used which scans a transverse
location of the sample to generate scanning data, and provides the
scanning data to the third arrangement so as to generate the image.
The scanning data may include the detected interference obtained at
multiple transverse locations on the sample. At least one
photodetector and at least one electrical filter may be used which
follow a photodetector, which is followed by an electrical filter.
The electric filter ma be a bandpass filter having a center
frequency that is approximately the same as a magnitude of the
frequency shift by the frequency shifting arrangement. A
transmission profile of the electrical filter can vary
substantially over its passband. The probe may include a rotary
junction and a fiber-optic catheter. The catheter can be rotated at
a speed higher than 30 revolutions per second. At least one
polarization modulator may be provided.
[0020] At least one polarization diverse receive and/or a
polarization diverse and dual balanced receiver may be used. It is
further possible to track the phase difference between: [0021] the
first electromagnetic radiation and the second electromagnetic
radiation, and/or [0022] the third electromagnetic radiation and
the fourth electromagnetic radiation.
[0023] According to still another exemplary embodiment of the
present invention, the first and second electro-magnetic radiations
can be emitted, at least one of which has a spectrum whose mean
frequency changes substantially continuously over time at a tuning
speed that is greater than 100 Tera Hertz per millisecond.
[0024] According to still a further exemplary embodiment of the
present invention, an apparatus is provided. Such apparatus
includes at least one first arrangement providing at least one
first electro-magnetic radiation to a sample and at least one
second electro-magnetic radiation to a reference. The apparatus
also includes t least one second arrangement adapted for shifting
the frequency of the first electro-magnetic radiation and the
second electromagnetic radiation, and an interferometer interfering
the first and second electro-magnetic radiations to produce an
interference signal. Further, the apparatus includes
[0025] at least one second arrangement detecting the interference
between the first and second electro-magnetic radiations.
[0026] Further, according to another exemplary embodiment of the
present invention, a system, method, software arrangement and
storage medium are provided for determining particular data
associated with at least one of a structure and composition of a
tissue. In particular, information associated with an
interferometric signal is received which is formed from at least
one first electro-magnetic radiation obtained from a sample and at
least one second electro-magnetic radiation obtained from a
reference. The first and/or second electro-magnetic radiations
is/are frequency-shifted. The information is sampled to generate
sampled data in a first format. Further, the sampled data is
transformed into the particular data that is in a second format,
the first and second format being different from one another.
BRIEF DESCRIPTION OF THE DRAWINGS
[0027] For a more complete understanding of the present invention
and its advantages, reference is now made to the following
description, taken in conjunction with the accompanying drawings,
in which:
[0028] FIG. 1 is a block diagram of a time-domain optical coherence
tomography ("OCT") system;
[0029] FIG. 2 is a block diagram of a system which performs
frequency-domain imaging using a spectral radar technique;
[0030] FIG. 3A is a block diagram of a system which performs
frequency-domain imaging using a coherent single-frequency tuning
source according to one exemplary embodiment of the present
invention;
[0031] FIGS. 3B and 3C are graphs of wavelength versus amplitude
which taken together illustrate the occurrence of a frequency shift
generated by the system of FIG. 3A;
[0032] FIG. 3D is a graph of a beat signal generated by the system
of FIG. 3A;
[0033] FIG. 4A is a block diagram of a system which performs
frequency-domain imaging using a multiple-longitudinal-mode
wavelength-swept source according to another exemplary embodiment
of the present invention;
[0034] FIGS. 4B and 4C are graphs of a wavelength spectrum which
taken together illustrate the occurrence of a shift in the spectrum
generated by the system of FIG. 4A;
[0035] FIG. 4D is a graph of a beat signal generated by the system
of FIG. 4A;
[0036] FIG. 5 is a block diagram of a system which performs
frequency-domain imaging using a wavelength-swept source according
to another exemplary embodiment of the present invention;
[0037] FIG. 6 is a block diagram of an optical wavelength tunable
filter arrangement according to an exemplary embodiment of the
present invention;
[0038] FIG. 7 is a block diagram of a wavelength-swept laser
arrangement according to an exemplary embodiment of the present
invention;
[0039] FIG. 8A is an exemplary graph of a laser output spectrum as
measured at an output of the wavelength-swept laser arrangement of
FIG. 7;
[0040] FIG. 8B is an exemplary graph of a laser output as measured
at an output of the wavelength-swept laser of FIG. 7;
[0041] FIG. 9A is a block diagram of a wavelength tunable filter
arrangement bearing a polygonal mirror according to yet another
exemplary embodiment of the present invention;
[0042] FIG. 9B is a block diagram of a wavelength tunable filter
arrangement having reflective disk according to still another
exemplary embodiment of the present invention;
[0043] FIG. 10A is a block diagram of an optical frequency domain
imaging ("OFDI") system which includes a wavelength-swept laser and
a polarization diversity-balanced detection ("PDBD") circuit
according to a further exemplary embodiment of the present
invention;
[0044] FIG. 10B is a block diagram of an exemplary probe
arrangement shown in FIG. 10A;
[0045] FIG. 10C is a plurality of graphs illustrating exemplary
outputs of a carrier-frequency heterodyne detection using the
system of FIG. 10A;
[0046] FIG. 11 is an exemplary in-vivo image of a human finger tip
obtained using exemplary embodiments of the present invention;
[0047] FIG. 12 is a block diagram of a phase tracker arrangement
according to an exemplary embodiment of the present invention;
[0048] FIG. 13 is a block diagram of an exemplary embodiment of the
OFDI system according to the present invention having the phase
tracker;
[0049] FIGS. 14A-14C are flow diagrams which illustrate an
exemplary technique for a phase tracker operation according to the
present invention;
[0050] FIG. 15 is a simplified diagram of the OFDI system according
to another exemplary embodiment of the present invention;
[0051] FIGS. 16(a) and 16(b) are graphs of effects of a frequency
shift according to the present invention, i.e., depth versus signal
frequency;
[0052] FIG. 17 is a block diagram of the OFDI system employing two
acousto-optic frequency shifters according to still another
exemplary embodiment of the present invention;
[0053] FIGS. 18(a) and 18(c) are graphs of point spread functions
measured without a mapping process according to the present
invention;
[0054] FIGS. 18(b) and 18(d) are graphs of point spread functions
measured with the mapping process according to the present
invention; and
[0055] FIGS. 19A and 19B are exemplary illustrations of images
and/or graphs of experimental results obtained using exemplary
embodiments of the present invention.
[0056] Throughout the drawings, the same reference numerals and
characters, unless otherwise stated, are used to denote like
features, elements, components, or portions of the illustrated
embodiments. Moreover, while the present invention will now be
described in detail with reference to the Figures, it is done so in
connection with the illustrative embodiments.
DETAILED DESCRIPTION
[0057] FIG. 1 shows an exemplary prior art time domain optical
coherence tomography ("OCT") system 10 which includes a broadband
source 12 that provides a signal to a first aim 14a of two-by-two
splitter 14. The splitter divides the signal provided thereto at
port 14a, and provides a first portion of the signal at a port 14b
coupled to a reference arm 16. The splitter 14 also provides a
second portion of the signal at a port 14c coupled to a sample arm
18.
[0058] The sample arm 18 terminates at a sample volume 19 and an
arrangement 22 for providing a lateral scan of the sample volume is
disposed in the sample arm 18 prior to the sample volume 19. The
reference arm 16 terminates in an arrangement 20 for providing an
axial scan. The arrangements 20 and 22 operate as is generally
known in the art.
[0059] Signals reflected from the means 20 and sample volume 19
back along the reference and sample arms 16, 18 respectively, are
coupled back into respective ports 14b, 14c of the splitter 14 and
are coupled to a detector 24 which produces axial scan data 26 as
is generally known. U.S. Pat. No. 6,341,036, the entire disclosure
of which is incorporated herein by reference, describes systems
similar to the one described above and shown in FIG. 1.
[0060] In general, on scanning the reference arm path length 16,
interference fringes are formed corresponding to positions that
match the distance to the three structures 19a, 19b, 19c in the
sample volume 19. The single detector 24 is used to detect the
interference fringes. By envelope detection of the fringe patterns,
an image 26 is constructed that maps tissue reflectivity to a given
location.
[0061] As will be apparent from certain exemplary embodiments
described herein below, an exemplary embodiment of the present
invention relates to a system which utilizes a detection principle
based upon Spectral Radar concepts (further referred to as Spectral
Domain OCT) and/or a hybrid method between Spectral Domain and Time
Domain OCT that is preferably more sensitive than current state of
the art Time Domain OCT, allowing a substantial increase in the
acquisition speed to resolution ratio.
[0062] Analysis of the Signal to Noise Ratio ("SNR") in Time Domain
OCT has been previously described in related publications. The
interference fringe peak amplitude in time domain OCT is given
by
I.sub.peak= {square root over (P.sub.refp.sub.sample)}, (1)
with P.sub.ref, P.sub.sample the reference and sample arm power in
Watts, respectively. In terms of electrical power at the detector,
the signal in units [A.sup.2] is defined as
S=.eta..sup.2e.sup.2P.sub.refP.sub.sample/E.sub.v.sup.2, (2)
with .eta. the quantum efficiency, e the charge quantum and
E.sub.v=hc/.lamda. the photon energy. The reference and sample arm
powers are given by the respective reflected spectral
densities,
P.sub.ref,sample=.intg.S.sub.ref,sample(.omega.)d.omega.. (3)
[0063] Assuming that the reference and sample spectral densities
are equal to the source spectral density S(.omega.), where the
sample arm spectral density is attenuated by a large factor, i.e.,
S.sub.ref(.omega.)=S(.omega.),
S.sub.sample(.omega.)=.alpha.S(.omega.) with .alpha.<<1, and
inserting the above expression of reference and sample arm into the
original definition of the signal gives,
S=.eta..sup.2e.sup.2.alpha..left
brkt-bot..intg.S(.omega.)d.omega..right
brkt-bot..sup.2/E.sub.v.sup.2. (4)
[0064] Three contributions to the total noise of OCT signals are:
(i) thermal noise, (ii) shot noise and (iii) relative intensity
noise. Thermal noise is generated by the feedback resistor, shot
noise is related to the finite nature of the charge quantum
resulting in statistical fluctuations on the current, and relative
intensity noise is related to the temporal fluctuations due to
chaotic character of classical light sources. These three
contributions to the noise density in units [A.sup.2/Hz] are given
by,
N noise ( f ) = 4 kT R fb + 2 .eta. e 2 P ref E v + 2 ( .eta. eP
ref E v ) 2 .tau. coh , ( 5 ) ##EQU00001##
k is Boltzmann's constant, T the temperature in Kelvin, R.sub.fb
the value of the feedback resistor, and .tau..sub.coh the coherence
time of the source. Coherence time is related to the full spectral
width at half maximum .DELTA..lamda. of a Gaussian source by the
following relation, .tau..sub.coh= {square root over (2 In
2/.pi.)}.lamda..sub.0.sup.2/(c.DELTA..lamda.). Shot noise limited
detection is achieved when the second term in Eq. (5) dominates the
other noise contributions.
[0065] The signal to noise ratio (SNR) is given by
S N R = S N noise ( f ) BW , ( 6 ) ##EQU00002##
with BW the signal bandwidth, and parameters S and N.sub.noise(f)
as described above.
[0066] Spectral Domain OCT Using a Spectrometer and CCD Array
Detector
[0067] The best signal to noise performance of Time Domain OCT
systems is obtained when the noise is shot noise limited. Shot
noise can be reduced significantly by replacing the single element
detector with a multi-element array detector. When the detection
arm light is spectrally dispersed on the array detector, each
element of the array detects a small wavelength fraction of the
spectral width of the source. The shot noise is preferably reduced
by a factor equal to the number of elements of the array. The
principle of the signal to noise improvement is based on the white
noise characteristic of shot noise and the observation that only
electromagnetic waves of the same wavelength produce interference
fringes.
[0068] The shot noise power density N.sub.noise(f) (in units
[W/Hz], [A.sup.2/Hz] or [V.sup.2/Hz]) is proportional to the
current (or equivalently the optical power times the quantum
efficiency) generated in the detector. For a monochromatic beam of
wavelength .lamda..sub.1 entering the interferometer, the fringe
frequency or carrier f at the detector is determined by the
velocity v of the mirror, f.sub.1=2v/.lamda..sub.1. The shot noise
is proportional to the power (or spectral density S(.omega.)) at
wavelength .lamda..sub.1. A second wavelength .lamda..sub.2 is
preferably coupled into the interferometer. A second fringe
frequency or carrier at frequency f.sub.2=2v/.lamda..sub.2 is
simultaneously present. The shot noise at this second frequency is
preferably the sum of the shot noise generated by the optical power
at wavelength .lamda..sub.1 and .lamda..sub.2. Also, at frequency
f.sub.1 the shot noise is the sum of the shot noise generated by
the optical power at wavelength .lamda..sub.1 and .lamda..sub.2.
Thus, at both frequencies a cross-shot noise term is generated by
the simultaneous presence of both wavelengths at the detector. By
spectrally dispersing each wavelength to a separate detector, the
cross shot noise term can be eliminated. In this way, Spectral
Domain OCT offers a significant improvement of signal to noise
ratio over Time Domain OCT systems.
[0069] The OCT signal is most easily described in the space domain.
For a single object in the sample arm, the interference term of the
OCT signal is proportional to the real part of the Fourier
transform of the source spectrum S(.omega.),
I(.DELTA.z)oc Re.intg.exp(ik.DELTA.z)S(k)dk, (7)
with .DELTA.z the path length difference between sample and
reference arm and k the wave vector. As a function of time, the OCT
signal is given by,
I(t)oc Re.intg.exp(2i.omega.rv/c)S(.omega.)d.omega., (8)
with v the reference arm mirror velocity. The frequency spectrum of
the signal is given by a Fourier transform of the signal in the
time domain, resulting in a complex function. The absolute value of
this function is equal to the spectral density,
|I(f)|=|.intg.I(t)e.sup.2i.pi.ftdt|=S(.pi.fc/v), (9)
which shows that the signal bandwidth is directly proportional to
the source spectral width and scales linearly with the reference
arm mirror velocity, i.e., imaging speed. Eq. (9) also preferably
directly relates the absolute value of the frequency spectrum,
|I(f)| to the signal S (see Eq. (4)). Eq. (9) also demonstrates
that each angular frequency of the light source or equivalently
each wavelength of the source is represented at its own frequency
in the measured interferometric signal. The depth profile
information I(t) can be obtained from the complex cross spectral
density, |I(f)| by a Fourier transform.
[0070] The complex cross spectral density can also be obtained by
splitting the signal I(t) in several spectral bands using a
dispersive or interferometric element. At each detector, only part
of the complex cross spectral density is determined. Combining the
cross spectral densities of each detector, the full spectral
density of the signal are retrieved. Thus, the same information can
be obtained by separating spectral components to individual
detectors. Combining the signal of all detectors in software or
hardware would result in the same signal as obtained with a single
detector.
[0071] In the detection arm, the spectrum can be split into two
equal halves, where two detectors each detect one half of the
spectrum. According to Eq. (9), the frequency spectra at detectors
1 and 2 are given by |I.sub.1(f)|=S(.pi.fc/v) for f<f.sub.0,
I.sub.1(f)=0 for f>f.sub.0 and I.sub.2(f)=0 for f<f.sub.0,
|I.sub.2(f)|=S(.pi.fc/v) for f>f.sub.0, respectively. The
frequency spectrum as would be acquired by a single detector in
time domain OCT is given by the sum of I.sub.1(f) and I.sub.2(f);
I(f)=I.sub.1(f)+I.sub.2(f). Thus, the signal S after combining the
spectra is equal, however I.sub.1(f)=0 for f>f.sub.0 and
I.sub.2(f)=0 for f<f.sub.0, the bandwidth BW per detector can be
reduced by a factor of 2.
[0072] The noise is determined by the sum of the shot noise
contributions at detectors one and two. From Eqs. (5) and (6), the
shot noise per detector is proportional to the reference arm power
at the detector times the bandwidth for the detector. Since the
spectrum was split in equal halves, the reference power at
detectors 1 and 2 is, respectively,
P.sub.ref.sup.1=0.5P.sub.ref, P.sub.ref.sup.2=0.5P.sub.ref.
(10)
[0073] The sum of the shot noise contribution for the two detectors
is,
N.sub.noise.sup.SDoc
P.sub.ref.sup.1.times.0.5BW+P.sub.ref.sup.2.times.0.5BW=0.5P.sub.refBW,
(11)
which may compared with the shot noise of a single detector in time
domain OCT,
N.sub.noise.sup.TDoc P.sub.refBW. (12)
[0074] Thus, by spectrally dispersing the detection and light over
two separate detectors, the signal remains the same, while the
noise is reduced by a factor of 2, resulting in a net SNR gain by a
factor of 2.
[0075] Extending the above analysis, it can be demonstrated that
the shot noise contribution is reduced by a factor equal to the
number of detectors. The sum of shot noises for N detector
elements, where each detector element receives one Nth of the total
reference power, is given
N noise = 2 .eta. e 2 P ref E v BW N . ( 13 ) ##EQU00003##
[0076] The signal is the same as in Time Domain OCT, and the SNR
ratio for Spectral Domain OCT is given by,
S N noise = .eta. P sample N 2 E v BW . ( 14 ) ##EQU00004##
[0077] Thus Spectral Domain OCT enables a SNR improvement over Time
Domain OCT of a hundred to a thousand fold, depending on the number
of detector elements N. Using a charge coupled array or an
integrating device as a detector, such as, but not limited to, a
line scan camera, the ratio N/BCW is replaced by the integration
time .tau..sub.i of the array, which results in,
S N noise = .eta. P sample .tau. i 2 E v . ( 15 ) ##EQU00005##
[0078] FIG. 2 shows an exemplary Spectral Domain OCT system 100
which includes an interferometer 102 with a source arm 104, a
sample arm 106, a reference arm 108, and a detection arm 110 with a
spectral separating unit 112, a detector array 114 comprised of a
plurality of detectors and a like plurality of amplifiers 116. The
amplifiers 116 are coupled through optional analog processing
electronics (not shown, but known to those having ordinary skill in
the art), and A/D converters (not shown, but known to those skilled
in the art) for conversion of signals and through digital band pass
filtering ("BPF") units 122 to a processing and display unit
124.
[0079] The processing and display unit 124 executes data processing
and display techniques, and can optionally include the digital band
pass filtering ("BPF") units 122 as well as Digital Fast Fourier
Transforms ("DFFTs") circuits (not shown), in order to provide
coherent combination of signals and to perform the data processing
and display functions. The detector array 114 may be 1.times.N for
simple intensity ranging and imaging and/or Doppler sensitive
detection, 2.times.N for dual balanced detection, 2.times.N for
simple intensity ranging and/or polarization and/or Doppler
sensitive detection, or 4.times.N for combined dual balanced and
polarization and/or Doppler sensitive detection. Alternatively, an
M.times.N array may be used for arbitrary number "M" of detectors
114 to allow detection of transverse spatial information on the
sample 130.
[0080] Electro-magnetic radiation (e.g., light) is transmitted from
the source along the source arm 104 to the splitting unit via and
is split between the reference arm 108 and the sample arm 106. The
light propagates along the sample arm to the tissue sample 130 and
through the reference arm 108 to a wavelength dependent phase
arrangement. The light is reflected from the sample and the
wavelength dependent phase arrangement back toward the splitting
unit where at least portions of the reflected light are directed
toward the spectral separating unit 112 (which may be provided as a
grating for example). The detection arm light is dispersed by the
spectral separating unit 112 and the spectrum is imaged onto the
detector array 114. By stepping the reference arm 108 length over a
distance .lamda./8, the cross spectral density of reference arm 108
and sample arm 106 light can be determined. The processing and
display unit received the signals fed thereto and performs a
Fourier transform of the cross spectral density to generate depth
profile information.
[0081] FIG. 3A shows a block diagram of an exemplary system
according to the present invention which illustrates basic
principles of a coherent frequency modulated continuous Wave
("C-FMCW") system using a single-frequency tuning source. A
monochromatic laser light 70 operable as a frequency chirped laser
provides a light signal to an input 72a of a coupler 72. The
coupler 72 divides the light signal into a reference arm 80 which
terminates in a reference mirror 82 and a sample arm 84 which
terminates in a sample 86. The light propagates down paths 80, 84
and reflects from the reference mirror 82 and sample mirror 86 to
provide, via coupler 72, interference signals which are detected by
a photo-detector 88.
[0082] As shown in graphs of FIGS. 3B-3D, when there is an optical
delay between two reflected light signals 90 (FIG. 3B) and 92 (FIG.
3C), respectively, a beat signal 94 (see FIG. 3D) having a
frequency f may be detected at the photo detector 88. Where there
are multiple reflection points in the sample along the axis, the
interference consists of beat notes having frequencies which are
proportional to the optical delay difference between the reflection
(scatter) point in the sample and the reference mirror. The power
of each beat frequency component is proportional to the
reflectivity of the scatter. Therefore, the image of the sample can
be constructed by Fourier transform of the interference data.
[0083] Referring now to FIGS. 4A-4D, in which like elements
described above and shown in FIGS. 3A-3D are provided having the
same reference designations, an optical frequency domain imaging
("OFDI") system according to an exemplary embodiment of the present
invention includes a wavelength-swept laser source 95 (also
referred to herein as a frequency swept source 95) which provides a
laser output spectrum comprised of multiple longitudinal modes to
an input of a coupler 72. The coupler 72 divides the signal fed
thereto into the reference arm 80 which terminates in the reference
mirror 82 and the sample arm 84 which terminates in the sample 86.
The optical signals reflect from the reference mirror 82 and the
sample 86 to provide, via coupler 72, a spectrum of signals which
are detected by a photo-detector 88.
[0084] The center (or mean) wavelength of the signal spectrum is
tuned in time by the creation of new longitudinal modes at the
leading side of the spectrum and the annihilation of the modes at
the trailing side of the spectrum.
[0085] The same principles described above with reference to FIGS.
3A-3D also apply to the OFDI technique using a wavelength-swept
laser source 95. Similar to the case of a C-FMCW system (e.g., the
system of FIG. 3A described above), a beat signal 94 can be
produced. In the case of the OFDI system that uses a
wavelength-swept laser source, the beat signal 94 can be generated
having a beat frequency f which corresponds to the difference in
the center frequency of the lights, 96 and 98, from the reference
and sample, respectively.
[0086] The frequency spacing between longitudinal modes should be
substantially larger than the detection bandwidth. The mode beat
frequency (relative intensity noise peaks) can be removed by a
proper electronic filter, such as low pass filter, prior to
digitization. The interference signal 94 contains a frequency
component that is proportional to the optical delay. Furthermore,
the image of the sample can be constructed by Fourier transform of
the digitized interference data.
[0087] In one exemplary embodiment of the present invention, the
wavelength-swept laser 95 can be provided which utilizes an optical
band-pass scanning filter in the laser cavity to produce a
rapidly-swept multiple-frequency-mode output. Exemplary filters
according to the present invention is described below in
conjunction with FIGS. 6 and 9A. By using an optical band-pass
scanning filter in the laser cavity, it is not necessary to tune
the laser cavity length to provide synchronous tuning of the laser
spectrum. Indeed, such arrangement does not require tuning the
longitudinal cavity mode of the laser at the same rate as the
center wavelength of the laser.
[0088] Using the OFDI techniques, a single pixel of the image can
be constructed from the signal that is recorded as a function of
time over the duration of one A-scan through Fourier transform.
This is different from the TD OCT where a single pixel is
constructed from the data measured at a short period time within
one A-scan. The detection bandwidth to acquire the same number of
data within the same A-scan period is approximately the same for
both TD and FD OCT. However, the Fourier transform used for the
OFDI technique effectively improves the signal-to-noise ratio
compared to TD OCT by constructing a single image pixel from many
data points acquired over the whole A-scan period. This effect can
result in an "effective" detection bandwidth that is N-times larger
than the actual detection bandwidth. Therefore, the SNR may be
improved by N times, where N is the number of (digitized) data
points in the Fourier transform. It can be shown that SNR in a shot
noise limited case is given by:
S N noise = .eta. P sample N 2 E v BW ( 16 ) ##EQU00006##
[0089] Due to the narrowband output spectrum of the
wavelength-swept source, however, the relative intensity noise
(RIN) can be significantly higher than that of a CW broadband light
source. For a thermal light, RIN is given by 1/.DELTA.v where
.DELTA.v=c.DELTA..lamda./.lamda..sup.2 is the optical bandwidth of
the (instantaneous) source output. For a laser light, RIN results
from different statistics and therefore has a different value than
the thermal light. For FD-OCT, a wavelength-swept laser with a low
RIN level is preferred. The laser light with multiple longitudinal
modes may have a similar RIN level as the thermal light with the
same linewidth. In this case, a means to suppress the RIN is
critical to have sufficient SNR, such as the dual balanced
detection.
[0090] Use of a swept source results in a system having reduced
shot noise and other forms of noise which allows for much lower
source powers, or much higher acquisition rates than current
systems. The increased detection sensitivity allows for real time
imaging. Such imaging speed can assist with a problem of motion
artifacts, such as in gastrointestinal, ophthalmic and arterial
imaging environments. By increasing the frame rate while
maintaining or improving the signal to noise ratio such artifacts
can be minimized. The present invention also enables one to screen
large areas of tissues with the OFDI technique, and allows clinical
viable screening protocols using this method.
[0091] For ophthalmic applications of OFDI, the efficient detection
preferably allows for a significant increase of acquisition speed.
A possible limitation in ophthalmic applications is the power that
is allowed to enter the eye according to the ANSI standards
(approximately 700 microwatts at 830 nm). Current data acquisition
speed in ophthalmic applications is approximately 100-500 A-lines
per second. The power efficient detection would allow for A-line
acquisition rates on the order of about 100,000 A-lines per second,
or video rate imaging at about 3,000 A-lines per image.
[0092] The gain in SNR is achieved because the shot noise has a
white noise spectrum. The signal intensity present at the detector
at frequency .omega. (or wavelength .lamda.) contributes only to
the signal at frequency .omega., but the shot noise is generated at
all frequencies. By narrowing the optical band width per detector,
the shot noise contribution at each frequency can be reduced, while
the signal component remains the same.
[0093] FIG. 5 shows an exemplary embodiment of a system 99 for
performing optical imaging using frequency-domain interferometry
("OFDI") which includes a frequency swept source 100 that emits a
narrowband spectrum of which the center wavelength is tuned
continuously and repeatedly in time across the bandwidth of the
gain medium in the source. The instantaneous emission spectrum
consists of a plurality of frequency modes of the light source. The
frequency swept source 100, may be provided in a variety of
different ways, some of which are described below. The source 100
may, for example, be provided from various gain media, tunable
wavelength filters, cavity configurations. Devices and methods are
known in the art to provide a rapidly-tuned wavelength-swept laser
source, such as solid-state lasers, active-ion-doped waveguide
lasers, and fiber lasers. A wavelength-swept laser in a mode-locked
regime can also be used with potential advantage of a lower
relative intensity noise (RIN) in a frequency region between
harmonics of longitudinal-mode beat frequencies. An optical
saturable absorber may be incorporated inside a laser cavity or
after the output port of the source to lower RIN level.
[0094] The light provided from swept source 100 is directed toward
a fiber-optic coupler 102 which divides the light fed thereto into
a reference arm 103 and a sample arm 104. In this exemplary
embodiment, the coupler 102 has a 90:10 power splitting ratio with
90% of the power being directed toward the sample arm. Those of
ordinary skill in the art would understand, however, that other
coupling ratios for the coupler 102 may also be used. The
particular coupling ratio to use in any particular application
should be selected such that an amount of power is provided to both
the reference aim and the sample arm to allow for proper operation
of the exemplary system according to the present invention.
[0095] The power provided to the sample arm passes though a
circulator 111, and illuminates a sample 136 to be imaged through a
transverse-scanning imaging probe. The reference arm provides
preferably a fixed optical delay. The lights reflected from a
reference mirror 124 and from within the sample 136 can be directed
through the respective circulators 110, 111 toward a fiber-optic
beam splitter (or fused coupler) 150 and interfere between each
other to produce interference signals.
[0096] It is desirable that the combining coupler 150 have an equal
splitting ratio with minimal polarization dependence and wavelength
dependence over the wavelength tuning range of the source. A
deviation from equal splitting results in reduction of common mode
rejection ratio ("CMRR") of the dual balanced detection. In one
embodiment, the combining coupler 150 is preferably provided as a
bulk broadband beam splitter. Those of ordinary skill in the art
would understand that other types of couplers (including but not
limited to wavelength-flattened fiber fused couplers) may also be
used.
[0097] The interference signals are received by a dual balanced
receiver 151. Output of the receiver 151 is provided to a computing
arrangement (e.g., a data acquisition board and computer 160), such
that the output is digitized and processed by the computer
arrangement to produce an image. The data acquisition, transverse
scanning, and wavelength tuning are synchronously controlled.
[0098] FIG. 6 shows an exemplary light source 100' which may, for
example, be adapted for use as a frequency swept source (such as
frequency swept source 100 described above with reference to FIG.
5) is provided from an optical filter 170, coupled through a lens
172 and a light path 174 to a light source/controller 176
(hereinafter referred to as "light controller 176"). The light
controller 176 may, in turn, be coupled to one or more applications
178. The applications 178 may, for example, correspond to optical
imaging processes and/or optical imaging systems, laser machining
processes and systems, photolithography and photolithographic
systems, laser topography systems, telecommunications processes and
systems. Thus, the exemplary light source 100' provided from the
filter 170 and the light controller 176 may be used in a wide
variety of different applications, certain general examples of
which are described herein.
[0099] As shall be described in further detail below, the filter
170 allows the light source 100' to operate as a frequency swept
source which emits a spectrum of which the center wavelength can be
tuned continuously and repeatedly in time across the bandwidth of
the light controller 176. Thus, light source 100' may have an
instantaneous emission spectrum comprised of a plurality of
frequency modes of the light source/controller 176. In this
exemplary embodiment, the optical wavelength filter 170 is
configured as a reflection-type filter in that the input and output
ports are identical. Thus, light path 174 may be provided, for
example, as an input/output optical fiber and lens 172 may
correspond to a collimating lens. Although the filter 170 in FIG. 6
is shown coupled to one or all of applications 178 through the
light controller 176, it is possible to directly couple the filter
170 to one or more of the applications 178. Alternatively, it is
possible to couple the filter 170 to one or more of the
applications 178 through a device other than a light
controller.
[0100] In the exemplary embodiment according to the present
invention, the light controller 176 can include a number of systems
that are specifically adapted to transmit a beam of light (in one
embodiment, a collimated beam of light) having a broad frequency
(f) spectrum. In particular, the beam of light can include a
plurality of wavelengths, within the visible light spectrum (e.g.,
red, blue, green). The beam of light provided by the light
controller can also include a plurality of wavelengths that are
defined outside of the visible spectrum (e.g., infrared).
[0101] As shall be described in greater detail below with reference
to FIG. 7, in one exemplary embodiment of the present invention,
the light controller 176 can include a unidirectional light
transmission ring. In another exemplary embodiment to be described
in detail in conjunction with FIG. 9 below, the light controller
176 can include a linear resonator system. The filter 170 includes
a wavelength dispersing element 180 adapted to receive the beam of
light from the light controller 176 and to separate the beam of
light into a plurality of different wavelengths of light each
directed along a light path as is known. The wavelength dispersing
element 180 can include one or more elements adapted to receive the
beam of light from the light controller 176, and to separate the
beam of light into a plurality of wavelengths of light each
directed along a light path. The wavelength dispersing element 180
is further operative to direct the plurality of wavelengths of
light in a plurality of angular directions or displacements with
respect to an optical axis 182. In one exemplary embodiment of the
present invention, the wavelength dispersing element 180 can
include a light dispersion element, such as a reflection grating
184. The wavelength dispersing element 180 could alternatively be
provided as a transmission grating (e.g. a transmission type
grating such as Dickson-type holographic grating), a prism, a
diffraction grating, an acousto-optic diffraction cell or
combinations of one or more of these elements.
[0102] The wavelength dispersing element 180 directs light at each
wavelength towards a lens system 186 along paths which are at an
angle with respect to the optical axis 182. Each angle is
determined by the wavelength dispersing element 180. The lens
system 186 can include one or more optical elements adapted to
receive the separated wavelengths of light from the wavelength
dispersing element 180 and to direct or steer and/or focus the
wavelengths of light to a predetermined position located on a beam
deflection device 188. The beam deflection device 188 can be
controlled to receive and selectively redirect one or more discrete
wavelengths of light back along the optical axis 182 through the
lens system 186 to the wavelength dispersing element 180 and back
to the light controller 176. Thereafter, the light controller 176
can selectively direct the received discrete wavelengths of light
to anyone or more of the applications 178. The beam deflecting
device 188 can be formed and/or arranged in a number of ways. For
example, the beam deflecting device 188 can be provided from
elements including, but not limited to, a polygonal mirror, a
planar minor disposed on a rotating shaft, a minor disposed on a
galvanometer, or an acousto-optic modulator.
[0103] In the exemplary embodiment shown in FIG. 6, the dispersing
element 186 includes a diffraction grating 184, a lens system 186
(which has first and second lenses 190, 192 to form a telescope
193), and the beam deflecting device 188 which is shown as a
polygon mirror scanner 194. The telescope 193 is provided from the
first and second lenses 190, 192 with 4-f configuration. The first
and second lenses 190, 192 of the telescope 193 are each
substantially centered on the optical axis 182. The first lens 190
is located at a first distance from the wavelength dispensing
element 180 (e.g., diffraction grating 184), which is approximately
equal to the focal length F1 of the first lens 190. The second lens
192 is located a second distance from the first lens 190, which is
approximately equal to the sum of the focal length F1 of the first
lens 190 and the focal length F2 of the second lens 192. In this
exemplary arrangement, the first lens 190 can receive the
collimated discrete wavelengths of light from the wavelength
dispersing element 180, and may effectively perform a Fourier
Transform on each one of the collimated one or more discrete
wavelengths of light to provide an equal one or more converging
beams projected onto an image plane (see designated IP of FIG. 6).
The image plane IP is located between the first and second lenses
and at a predetermined distance from the first lens, which
predetermined distance is defined by the focal length F1 of the
first lens. After propagating through the image plane IP, the
converging beam(s) form an equal one or more diverging beams that
are received by the second lens. The second lens operates to
receive the one or more diverging beams and to provide an equal
number of collimated beams having predetermined angular
displacements with respect to the optical axis 182 for directing or
steering the collimated beams to predefined portions of the beam
deflection device 188.
[0104] The telescope 193 is configured to provide a number of
features, as described above, and further to convert diverging
angular dispersion from the grating into converging angular
dispersion after the second lens 192, which is desired for proper
operation of the filter 170. In addition, the telescope 193
provides a useful degree of freedom, which controls the tuning
range and reduces the beam size at the polygon mirror 194 to avoid
a beam clipping.
[0105] As is illustrated in FIG. 6, the polygon mirror 194 reflects
back preferably only the spectral component within a narrow
passband as a function of the angle of the front mirror facet of
the polygon with respect to the optic axis. The reflected
narrowband light is diffracted and received by the optical fiber
174.
[0106] The orientation of the incident beam with respect to the
optic axis and the rotation direction 198 of the polygon mirror 194
determine the direction of wavelength tuning: wavelength up
(positive) scan or down (negative) scan. The arrangement in FIG. 6
produces a positive wavelength sweep. It should be understood that
while the mirror 194 is shown in FIG. 6 as having twelve facets,
fewer or more than twelve facets can also be used. The particular
number of facets to use in any application depends upon the desired
scanning rate and scanning range for a particular application.
Furthermore, the size of the mirror is selected in accordance with
the needs of a particular application taking into account factors
including, but not limited to, manufacturability and weight of the
mirror 194. It should also be appreciated that lenses 190, 192 may
be provided having different focal lengths. The lenses 190, 192
should be selected to provide a focal point at approximately the
center point 200 of the mirror 194.
[0107] Consider a Gaussian beam with a broad optical spectrum
incident to the grating from the fiber collimator 172. The
well-known grating equation is expressed as .lamda.=p(sin
.alpha.+sin .beta.) where .lamda. is the optical wavelength, p is
the grating pitch, and .alpha. and .beta. are the incident and
diffracted angles of the beam with respect to a nominal axis 202 of
the grating, respectively. The center wavelength of tuning range of
the filter is given by .lamda..sub.0=p(sin .alpha.+sin
.beta..sub.0) where .lamda..sub.0 is the angle between the optic
axis 38 of the telescope and the grating normal axis. It can be
shown that FWHM bandwidth of the filter is given by
(.delta..lamda.).sub.FWHM/.lamda..sub.0=A(p/m)cos .alpha./W where
A= {square root over (4 In 2/.pi.)} for double pass, m is the
diffraction order, and W is 1/e.sup.2-width of the Gaussian been at
the fiber collimator. When the real part of the complex spectral
density is determined, ranging depth z is defined by
z = .lamda. 0 2 4 ( .delta..lamda. ) FWHM . ##EQU00007##
[0108] Tuning range of the filter is fundamentally limited by the
finite numerical aperture of lens1 20. The acceptance angle of
lens1 without beam clipping is given by .DELTA..beta.=(D.sub.1-W
cos .beta..sub.0/cos .alpha.)/F.sub.1, where D.sub.1 and F.sub.1
are the diameter and focal length of lens1. It relates to the
filter tuning range via .DELTA..lamda.=p cos
.beta..sub.0.DELTA..beta.. An importance design parameter of the
filter, originated from multiple facet nature of the polygon
mirror, is the free spectral range, which is described in the
following. A spectral component after propagating through lens1 20
and lens2 22 will have a beam propagation axis at an angle .beta.'
with respect to the optic axis 38:
.beta.'=-(.beta.-.beta..sub.0)(F.sub.1/F.sub.2) where F.sub.1 and
F.sub.2 are the focal lengths of lens1 and lens2, respectively. The
polygon has a facet-to-facet polar angle given by
.theta.=2.pi./N.apprxeq.L/R where L is the facet width, R is the
radius of the polygon, and N is the number of facets. If the range
of .beta.' of incident spectrum is greater than the facet angle,
i.e. .DELTA..beta.'=.DELTA..beta.(F.sub.1/F.sub.2)>.theta., the
polygon minor could retro-reflect more than one spectral component
at a given time. The spacing of the multiple spectral components
simultaneously reflected, or the free spectral range, can be shown
to be (.DELTA..lamda.).sub.FSR=p cos
.beta..sub.0(F.sub.2/F.sub.1).theta..
[0109] In the application as an intracavity scanning filter, the
tuning range of the laser cannot exceed the free spectral range if
the gain medium has homogenous broadening, since the laser chooses
the wavelength of highest gain. The duty cycle of laser tuning by
the filter can be, in principle, 100% with no excess loss caused by
beam clipping if two necessary conditions are met as follows:
W < cos .alpha. F 1 cos .beta. 0 F 2 L and W < cos .alpha.
cos .beta. 0 ( F 2 - S ) .theta. ( 17 ) ##EQU00008##
[0110] The first equation is derived from the condition that the
beamwidth after lens 192 should be smaller than the facet width.
The second equation is from that the two beams at the lowest and
highest wavelengths 204, 206 respectively of the tuning range
should not overlap each other at the polygon mirror S in Equation
(1) denotes the distance between the lens 192 and the front mirror
of the polygon.
[0111] In one experiment, optical components with the following
parameters were selected: W=1.9 mm, p=1/1200 mm, .alpha.=1.2 rad,
.beta..sub.0=0.71 rad, m=1, D.sub.1=D.sub.2=25 mm, F.sub.1=100 mm,
F.sub.2=45 mm, N=24, R=25 mm, L=6.54, S=5 mm, .theta.=0.26 rad,
.lamda..sub.0=1320 nm. From the parameters, the theoretical FWHM
bandwidth, tuning range and free spectral range of the filter could
be calculated: (.delta..lamda.).sub.FWHM=0.09 nm,
.DELTA..lamda.=126 nm and (.DELTA..lamda.).sub.FSR=74 nm. Both
conditions in (1) are satisfied with margins. The characteristics
of the filter were measured using broadband amplifier spontaneous
emission light from a semiconductor optical amplifier (SOA) and an
optical spectrum analyzer. The optical spectrum analyzer recorded
the normalized throughput (reflected) spectrum in peak-hold mode
while the polygon mirror was spinning at its maximum speed of 15.7
kHz. The measured tuning range was 90 nm which is substantially
smaller than the theoretical value of 126 nm. The discrepancy was
due to the aberration of the telescope, primarily field curvature,
associated with relatively large angular divergence of the beam
from the grating. It is expected that the aberration would be
improved by using optimized lenses. The free spectral range was
73.5 nm in agreement with the theoretical calculation. The FWHM
bandwidth was measured to be 0.12 nm. The discrepancy with
theoretical limit of 0.11 nm may be reasonable considering the
aberration and imperfection of the optical elements.
[0112] FIG. 7 shows an extended-cavity semiconductor laser 208
according to an exemplary embodiment of the present invention which
can include a filter 210 that may, for example, be similar to the
filter 170 described above with reference to FIG. 6. The filter 210
is coupled through a light directing element 212 and a light path
214 to a Faraday circulator 216. In this exemplary embodiment, the
filter 210 includes a grating 232 and a polygonal mirror 236. Thus,
the filter 210 may correspond to a polygon-based filter. A motor
234 drives the mirror.
[0113] The Faraday circulator 216 of this exemplary embodiment is
coupled through polarization controllers 220, 222 to a gain medium
224 which in one exemplary embodiment can be a semiconductor
optical amplifier (e.g., SOA, Philips, CQF 882/e) having coupled
thereto a current source 226 which provides an injection current to
the SOA 224. The intracavity elements may be connected by
single-mode optical fibers, for example. The two polarization
controllers 220, 222 can align the polarization states of the
intracavity light to align to the axes of maximum efficiency of the
grating 232 and of maximum gain of the SOA 224.
[0114] A laser output 228 may be obtained through a 90% port of a
fiber-optic fused coupler 230. To generate a sync signal useful for
potential applications, 5% of the laser output may be coupled
through a variable wavelength filter 237 having a bandwidth of 0.12
nm and is directed toward a photodetector 238. In one exemplary
embodiment, the center wavelength of the filter may be fixed at
1290 nm. The detector signal generates short pulses when the output
wavelength of the laser is swept through the narrowband passband of
the fixed-wavelength filter. The timing of the sync pulse is
controlled by changing the center wavelength of the filter.
[0115] FIG. 8A shows a graph 240 of an output spectrum of a laser
of the type described above with reference to FIG. 7 as measured by
an optical spectrum analyzer in peak-hold mode, when the polygon
mirror (i.e. mirror 236 in FIG. 7) spins at a rate of 15.7 kHz. The
edge-to-edge sweep range may be from 1282 nm to 1355 nm over 73
nm-width equal to the free-spectral range of the filter. The
Gaussian-like profile of the measured spectrum, rather than a
square profile, is likely due to the polarization-dependent cavity
loss caused by polarization sensitivity of the filter and the
birefringence in the cavity. It is preferable to adjust the
polarization controllers to obtain the maximum sweep range and
output power.
[0116] FIG. 8B shows a curve 242 of a laser output in a time
domain. The upper trace 244 corresponds to a sync signal obtained
through the fixed-wavelength filter. The amplitude of power
variation from facet to facet was less than 3.5%. The peak and
average output power was 9 mW and 6 mW, respectively. It should be
mentioned that the y-axis scale of curve 240 had to be calibrated
from the time-domain measurement, because the optical spectrum
analyzer only recorded a time-averaged spectrum due to the laser
tuning speed much faster than the sweep speed of the spectrum
analyzer.
[0117] A frequency downshift in the optical spectrum of the
intracavity laser light may arise as the light passes through an
SOA gain medium (e.g. SOA 224 in FIG. 7), as a result of intraband
four-wave mixing phenomenon. In the presence of the frequency
downshift, the positive wavelength scan can facilitate tuning of
the laser spectrum, and thereby produce higher output powers. The
peak power of the laser output can be measured as a function of the
tuning speed. The negative tuning speed may be obtained by flipping
the position of the collimator and the orientation of the grating
with respect to an optic axis (e.g., axis 182 in FIG. 6). It is
preferable to make the physical parameters of the filter
approximately identical in both tuning directions. Thus, the
combined action of self-frequency shift and positive tuning allows
higher output to be obtained and enables the laser to be operated
at higher tuning speed. Therefore, the positive wavelength scan may
be the preferred operation. The output power may decrease with
increasing tuning speed. Thus, a short cavity length may be desired
to reduce the sensitivity of the output power to the tuning speed.
In this case, a free-space laser cavity is preferred.
[0118] FIG. 9A shows an exemplary embodiment of a free-space
extended-cavity semiconductor tunable laser 250 according to an
exemplary embodiment of the present invention that includes a
semiconductor waveguide 252 fabricated on a substrate chip 254
coupled to a polygon scanning filter 255 through a collimating lens
256. A front facet 258 may be anti-reflection coated, and an output
facet 260 is cleaved or preferably coated with dielectrics to have
an optimal reflectivity. An output 262 of the laser is obtained
through the output coupling lens 264. The collimating lenses 256,
264 are preferably provided as aspheric lenses.
[0119] The filter 255 includes a wavelength dispersing element 180'
adapted to receive the beam directed thereto from the lens 256. The
wavelength dispersing element 180' may be similar to wavelength
dispersing element 180 described above with reference to FIG. 6. A
lens system 186' can be disposed between the wavelength dispersing
element 180' and a beam deflection device 188'. The wavelength
dispersing element 180' and a beam deflection device 188' may be
similar to wavelength dispersing element 180 and a beam deflection
device 188 described above with reference to FIG. 6. The lens
systems 186' includes a pair of lenses 187a, 187b which are
preferably provided as achromats having low aberration particularly
in field curvature and coma.
[0120] A sync output may be obtained by using a lens 266, a pinhole
268, and a photodetector 270 positioned on the 0-th order
diffraction path for the light which is on retro-reflection from a
polygon scanner 272. The detector generates a short pulse when the
focus of the optical beam of a particular wavelength sweeps through
the pinhole 268. Other types of gain medium may include, but are
not limited to, rare-earth-ion doped fiber, Ti:Al.sub.2O.sub.3, and
Cr.sup.4+:forsterite.
[0121] FIG. 9B, shows another exemplary embodiment of a wavelength
tunable filter 280 according to the present invention which may
include an optical fiber 281 coupled to an input collimating lens
282, optically coupled to a diffraction grating 284, a focusing
lens 286, and a spinning disk 288. The diffraction grating 284 may
be replaced by other angular dispersive elements such as a prism.
In one exemplary embodiment, the diffraction grating 284 can have a
concave curvature with a focal length selected such the focusing
lens 286 is not needed.
[0122] Preferably more than one reflector 290 may be deposited on a
surface 288a of the spinning disk 288. Preferably, the reflectors
290 comprise multiple narrow stripes periodically and radially
patterned. The material for the reflectors is preferably gold. The
disk 288 can be composed of a lightweight plastic or silicon
substrate. Instead of the reflectors deposited on the top surface
of the disk, the disk can have a series of through holes followed
by a single reflector attached to the back surface of the disk.
Incident from the optical fiber 281, the optical beams of different
wavelengths may be illuminated on the surface of the disk into a
line after being diffracted by the grating and focused by the lens
286 (in those systems which include lens 286). Preferably, only the
beam that impacts the reflectors of the spinning disk may be
retro-reflected and received by the optical fiber 281. A mirror 292
may be used to facilitate the access of the beam onto the disk.
[0123] The distance from the lens 286 to the reflectors of the disk
288 is equal to the focal length, F, of the lens. It can be shown
from the grating equation that the tuning range of the filter is
given by .DELTA..lamda.=p cos .beta..sub.0(D/F) where D denotes the
distance between the stripes. The width of the strip, w, is
preferably made to be substantially equal to the beam spot size
w.sub.s, at the surface of the disk:
w s = W cos .beta. 0 cos .alpha. F / z 1 + ( f / z ) 2
##EQU00009##
where z=.pi.w.sub.s.sup.2/.lamda.. It leads to a FWHM filter
bandwidth given by
(.delta..lamda.).sub.FWHM/.lamda..sub.0=A(p/m)cos .alpha./W where
A= {square root over (4 In 2)}/.pi.. For the w>w.sub.s, filter
bandwidth becomes greater, and for w<w.sub.s, the efficiency
(reflectivity) of the filter is decreased by beam clipping. The
orientation of an incident beam 294 with respect to the optic axis
of the lens 286 and the spinning direction 288 preferably
determines the sense of wavelength tuning. The positive wavelength
scan is preferable, which is achieved by spinning the disk 288 in a
clockwise direction as shown in FIG. 9B. a. Interferometer
[0124] FIG. 10A shows an exemplary embodiment of an OFDI system 300
according to the present invention for performing optical imaging
using frequency-domain interferometry includes a frequency swept
source 301 which emits a light signal having an instantaneous
emission spectrum comprised of a plurality of frequency modes of
the light source. Source 301 may, for example, be provided as one
of the sources described above with reference to FIGS. 4A, 5, 6, 7,
9 and 9B. The light from source 301 can be directed toward a
fiber-optic coupler 302 which divides the light fed thereto into a
reference arm 303 and a sample arm 304.
[0125] The reference arm 303 preferably includes a polarization
circuit 306 and a circulator 308. Thus, light propagates from
source 301 through the coupler 302, the polarization circuit 306
and the circulator 308 to an optional motion artifact circuit 309.
The optional motion artifact circuit 309 may be provided from a
lens 310 which directs the light toward a frequency shifter 311, a
phase tracker 312 and a dispersion compensator 314. The light
passes through optional circuit 309 and is incident upon a
reference mirror 316. It should be appreciated that circuit 309
functions to remove or reduce motion artifacts. It should also be
appreciated that circuit 309 may include all of the elements
310-314, and/or one or more of the circuit elements 310-314.
[0126] The sample arm 304 may include a circulator 318. Thus, a
light signal transmitted from the source 301 propagates from source
301 through the coupler 302 and the circulator 308 to a lens 320
which directs the light toward a scanning mirror 322. The scanning
mirror 322 may be provided from a wide variety of optical elements
including but not limited to, a galvanometer, a piezoelectric
actuator or another functionally equivalent device. A transverse
scanner 324 is coupled to the scanning mirror 322 and a data
acquisition board and computer 326. The data acquisition board and
computer 326 is also coupled to the frequency swept source 301.
[0127] The OFDI system 300 shown in FIG. 10A can also include a
polarization diversity balanced detection ("PDBD") circuit 334
configured to receive signals from the reference arm 303 and/or the
sample arm 304. In particular, the reference arm 303 is connected
through circulator 308 and polarization control circuit 330 to a
reference port of the PDBD circuit 334. Similarly, sample arm 304
is connected through circulator 318 and polarization control
circuit 332 to a sample port of the PDBD circuit 334.
b. Interferometer
[0128] The sample arm 304 collects light reflected from a tissue
sample 328 and is combined with the light from the reference arm
303 in the polarization diversity balanced detection (PDBD) circuit
334 to form interference fringes.
[0129] For example, the OFDI technique does not require that the
optical path length in the reference arm be scanned in time. Thus,
in certain exemplary embodiments of the present invention, it may
be preferable to provide the reference arm as a fixed delay
reference arm. Such fixed delay reference arms may have various
configurations that are known to those having ordinary skill in the
art.
[0130] The reference arm 303 can be either of reflective and/or
transmission type, and can return light back from the mirror 316.
The returned light is directed toward the polarization control
circuit 330 via the circulator 308. Similarly, the reflected light
from the sample 338 can be directed toward a polarization control
circuit 332 via the circulator 318. The reference arm can also be
transmission with no reflection. The polarization control circuit
330 can be used to match the polarization state of the
reference-arm light to that of the sample-arm. The total
birefringence in the interferometer should be minimized not to
induce wavelength-dependent birefringence. The polarization
controller may include, but is not limited to, a fiber-optic
polarization controller based on bending-induced birefringence or
squeezing.
[0131] Preferably, the chromatic dispersion should be matched
substantially between the reference and sample arm. The result of
strong dispersion mismatch may be a loss in the axial resolution.
Any residual dispersion can likely be compensated by appropriate
signal processing, such as nonlinear mapping based on interpolation
of the detector data before the Fourier transform. This mapping may
also be accomplished, at least in part, by adjusting the optical
layout of the wavelength-swept source. In one example in which the
source 301 includes a polygon scanner and a telescope, the distance
between the polygon scanner and the telescope can be adjusted to
convert wavelength space to wave vector space prior to Fourier
transformation.
c. Sample Arm
[0132] For certain OFDI applications, the sample arm may be
terminated by an optical probe comprising a cleaved (angled, flat,
or polished) optical fiber or free space beam. A lens 336 (such as,
but not limited to, aspherical, gradient index, spherical,
diffractive, ball, drum or the like) may be used to focus the beam
on or within the sample. Beam directing elements (such as, but not
limited to, mirror, prism, diffractive optical element or the like)
may also be contained within the probe to direct the focused beam
to a desired position on the sample. The position of the beam may
be changed on the sample as a function of time, allowing
reconstruction of a two-dimensional image. Altering the position of
the focused beam on the sample may be accomplished by the scanning
mirror 322. The scanning mirror 322 may be provided, for example,
from a number of different devices including, but not limited to, a
galvanometer, piezoelectric actuator, an electro-optic actuator or
the like.
[0133] The sample arm probe may be a fiber optic probe that has an
internally moving element, such that the motion is initiated at a
proximal end of the probe and the motion is conveyed by a motion
transducing arrangement (such as, but not limited to, wire,
guidewire, speedometer cable, spring, optical fiber and the like)
to the distal end. The fiber optic probe may be enclosed in a
stationary sheath which is optically transparent where the light
exits the probe at the distal end. Thus, scanning way also be
accomplished by moving the optical fiber. For example, by rotating
the optical fiber, or linearly translating the optical fiber. FIG.
10B shows an exemplary embodiment of the probe 359 which includes
an inner cable 361 (that may rotate or linearly translate along the
axis of the probe), an outer transparent or semi-transparent sheath
362, distal optics 364, and remitted light 366 (which may be at any
angle with respect to axis of catheter).
d. Detection
[0134] The PDBD circuit 334 may include a plurality of detectors
370 disposed to provide dual balanced detection. Dual balanced
detection may be preferred in certain applications for the
following reasons. First, most light sources generate 1/f noise
(f=frequency) at relatively low frequencies and balanced detection
will eliminate 1/f source noise. Second, an interference term of
the sample arm light with itself (i.e. an auto-correlation term)
can be present on top of the true signal term, which is preferably
the interference between sample and reference arm. Such
auto-correlation term can be eliminated by a differential technique
and balanced detection may eliminate this auto-correlation term
from the measured signal. Third, RIN can be reduced.
[0135] The detectors 370 may preferably include photodiodes (such
as, but not limited to, silicon, InGaAs, extended InGaAs, and the
like). Balanced detection can be implemented by subtracting diode
signals that are exactly out of phase with respect to the maxima
and minima pattern. The difference between two detector signals is
obtained by a differential circuit included in PDBD circuit 334 and
amplified by trans-impedance amplifiers ("TIA") 360. The dual
balanced receiver may be further followed by a low-pass or
band-pass filter to reject noise outside the detection
bandwidth.
[0136] In this exemplary embodiment of the present invention, the
dual balanced detection can be implemented as follows. The
polarization beam splitter 362 receives signals from the reference
and sample arms and provides two output signals. The two output
signals are further split by two non-polarizing beam splitters
364a, 364b, respectively. The outputs from each beam splitter 364a,
364b are detected by a dual balanced receiver provided from the
four detectors 370. Furthermore, the two outputs of the dual
balanced receivers are digitized and processed in a computer
arrangement to obtain a polarization diversity.
[0137] The receiver output is provided to circuit 326 which
acquires and digitizes the signals fed thereto via A/D converters,
and stores the digitized signals in a computer for further
processing. The bandwidth of the TIA is preferably matched to half
the sampling rate. Gain of the TIA is preferably selected such that
the maximum receiver output range is matched to the voltage range
of the A/D converter.
e. Processing
[0138] If more than two detectors are used, the signals can be
selectively subtracted and complex spectral density can be
obtained. Using the Fourier transform, the complex cross spectral
density can be converted to a depth profile in the tissue. Several
methods to process the complex spectral density to obtain depth
profile information are known to those skilled in the art, such as,
but not limited to, by obtaining at least two signals with a Pi/2
phase shift in the reference arm and then reconnecting the complex
spectral density by some linear combination of the two signals, or
by squaring the spectral density.
[0139] Following the detection, analog processing can include a
trans-impedance amplifier, low pass (band pass) filter, and
digitization of the signal. This signal may then be converted to
reflectivity as a function of depth by the Fourier transform
operation. Digital processing includes digitization, digital band
pass filtering in either the frequency domain or time domain (FIR
or IIR filter) and inverse Fourier transformation to recover the
tissue reflectivity as a function of depth.
[0140] Prior to the Fourier transformation, the detected non-linear
wavelength coordinates is preferably converted to regularly sampled
wave-vector space. Typically zero padding the signal, Fourier
transformation, and inverse Fourier transformation with re-sampling
can be utilized for remapping. Other interpolation methods known in
the art, such as linear, bi-linear, and cubic spline interpolation
of the data may also be used to convert wavelength space into
regularly sampled k space. This mapping may also be accomplished in
part by adjusting the optical layout of the wavelength-swept
source. In one example, the distance between the polygon scanner
and the telescope may be adjusted to convert wavelength space to
wavevector space prior to Fourier transformation.
[0141] Another exemplary embodiment of the present invention can
utilize one or more techniques described below to further enhance
the performance and functionality of imaging. These techniques are
not limited to the OFDI techniques that use a
multiple-frequency-mode tuned source, but can be applied in the
OFDI technique using a single-frequency tuned source.
a. Polarization Diversity
[0142] For an application where polarization fading is a problem, a
polarization diversity scheme may be used. Various configurations
for polarization diversity are known in the art.
[0143] In the system shown in FIG. 10A, the polarization diversity
circuit operates as follows. The polarization beam splitter 362
separates the reference-arm and sample-arm light signals depending
upon their polarization states. The polarization controller 330 is
preferably adjusted so that the reference-arm power is split with
an equal magnitude by the polarization controller. The polarization
state of the sample arm power can be assumed to vary randomly due
to the probe or sample motion, therefore the separating ratio of
the sample arm power by the polarization splitter can vary in time.
However, the two output signals at the two output ports of the
polarization beam splitter 362 can be detected by a photo receiver,
e.g., squared and summed. The resulting signal is independent of
the polarization state of the sample arm light.
b. Carrier-Frequency Heterodyne Detection
[0144] The optical frequency shifter 311 may be situated in the
reference arm 303 to shift the optical frequency for
carrier-frequency heterodyne detection. As a result, the signal
frequency band is shifted by the magnitude of the frequency shift.
In this manner, relatively large 1/f noise (f=frequency) and RIN
around DC can be avoided. The frequency shifter can be, but not
limited to, an acousto-optic frequency shifter. In the detection, a
proper electronics should be used to demodulate the carrier
frequency.
[0145] One of the benefits of using the frequency shifter 311 is
that the effective ranging depth can be doubled. This can be
illustrated in the electrical frequency domain, as shown in FIG.
10C in which a graph 380 depicts the fringe visibility curve given
by the instantaneous output spectrum of the source. The visibility
curve has a Gaussian profile if the source's instantaneous spectrum
is Gaussian. A curve 390 depicts the transmission efficiency
profile of an electrical filter, which is optimized for a given
Nyquist frequency defined as the half of the sampling frequency.
Section (a) of FIG. 10C shows a typical case where there is no
frequency shifter in the interferometer and the electrical filter
is a low pass filter. Because the positive and negative frequency
band is not differentiable, the images associated with the positive
and negative frequency band, respectively, are overlapped. Because
of this ambiguity, only half of the frequency range (zero to
f.sub.N) or (zero to -f.sub.N) is usable in this case. However,
using a frequency shifter results in a shift of the visibility
curve by f.sub.FS, as shown in portion (b) of FIG. 10C. With a
bandpass filter (or a low pass filter), both sides of the frequency
band centered at f.sub.FS produce images without ambiguity,
resulting in a twice larger ranging depth compared to section (a)
of FIG. 10C.
[0146] Instead of a square-top bandpass filter, it is possible to
use a slope filter. In an example shown in FIG. 10C section (c),
the transmission efficiency curve of the filter, 390, has an
exponentially-rising (falling) slope in its low frequency band.
This filter may be useful in which attenuation is significant and
the resulting signal strength decays with depth. The slope filter
can improve the dynamic range of the detection by effectively
suppressing the large signal from the surface relative to that at
greater depths.
c. Reference Arm Delay (Phase Tracking and Auto-Ranging)
[0147] As described above, the OFDI technique does not: require the
optical path length in the reference arm to be scanned in time. A
fixed-delay reference arm can be made in various configurations
that are known to those having ordinary skill in the art. The
reference arm can be of either reflective or transmission type.
[0148] In certain applications, the capability of varying the
optical delay in the reference arm may be useful when a larger
ranging depth is desired, without increasing the data acquisition
rate or reducing the instantaneous linewidth of the optical source.
Such ability is useful in a clinical study where the distance from
the imaging lens and the front surface of the sample can varies
significantly. Such variation can result from the motion or from
the uncontrolled position of a probing catheter. For example, a
rotating catheter inside a blood vessel can have distance variation
by a couple of millimeter over a single A-scan.
[0149] A mechanism in the reference arm 303 may allow for scanning
the group delay of the reference arm 303. This group delay can be
produced by any of a number of techniques known to those having
ordinary skill in the art, such as, but not limited to, stretching
an optical fiber, free space translational scanning using a
piezoelectric transducer, or via a grating based pulse shaping
optical delay line. Preferably, the delay can be introduced by a
non-mechanical or motionless arrangement. By the term
"non-mechanical", what is meant is that there are no mechanically
moving parts being utilized. The absence of the mechanically moving
parts is believed to reduce the known deficiencies of using
mechanical devices to introduce delay. In contrast to traditional
LCI or OCT systems, the reference arm 303 according to an exemplary
embodiment of the present invention does not necessarily need to
scan over the full ranging depth in the sample, and can preferably
scan over at least a fraction of the ranging depth equal to one
over the number of detectors (1/N). This scanning feature is
different from the conventional delay scanning schemes used in the
known LCI and OCT systems. The reference arm 303 optionally has a
phase modulator mechanism, such as but not limited to, an
acoustooptic modulator, electro-optic phase modulator or the like,
for generating a carrier frequency.
[0150] Phase tracking is preferable performed to eliminate phase
instabilities in the interferometer. Phase instabilities can cause
individual interferometric fringes to shift in location. If
detection is slow relative to the shifting of the fringes, the
resulting averaging results in chirping of the interference signal.
A-scan rate of 10 to 40 kHz results in an effective integration
time of 100 to 25 .mu.s. Phase instabilities arising on a time
frame shorter than the integration time should be compensated.
Phase locking circuitry is commonly used in electronics, and is
frequently used in radar and ultrasound. Active phase tracking can
be implemented by modulating the interferometer path length
difference at 10 MHz with an electro-optic phase modulator in the
reference arm over a fraction of the wavelength. By demodulating
the intensity measured by one detector at the output of the
interferometer at the frequency of the path length modulation, an
error signal can be generated indicating in which direction the
phase modulator should shift to lock onto a fringe amplitude
maximum. By adding an offset to the phase modulator as determined
by the error signal, the phase tracker actively locks onto a fringe
maximum.
[0151] The phase modulator can modulate the path length difference
over a few wavelengths. The processing unit can determine if the
phase modulator has reached its range limit, and jump by a full
wave in phase to maintain lock on a different fringe maximum. This
approach exploits the fact that phase should be controlled only
modulo 2.pi.. In addition, the processing drives a slower component
(e.g., the Rapid Scanning Optical Delay ("RSOD") line) to extend
the path length range of the phase modulator/RSOD combination over
several millimeters. Phase locking can be performed on a fringe
maximum, minimum, or zero crossing, based on the type of mixing
performed in the demodulation circuit.
[0152] Another exemplary embodiment of the present invention can
also use autoranging techniques and technology, including
processing techniques described in U.S. patent application
publication no. 2002/0198457, the disclosure of which is hereby
incorporated herein by reference in its entirety. The autoranging
mechanism may, in one exemplary embodiment, include a processor
unit for (a) obtaining a first scan line; (b) locating a surface
location "S" of a sample; (c) locating an optimal scan range "R" of
the sample; (d) modifying a reference arm delay waveform to provide
an output; (e) outputting the output to a reference arm; (f)
determining whether the image is complete; and/or (g) moving to the
next scan line if the image is not complete or remapping the image
using the surface S data and the waveform data stored in the memory
storage device if the image is complete.
[0153] If the light signal returned from the sample has a low
amplitude, phase locking may be unstable due to the presence of
noise. In another exemplary embodiment of the present invention, a
separate, preferably monochromatic, light source can be transmitted
into the interferometer. The separate source wavelength may be
within the wavelength tuning range of the OFDI source or may be
centered at a different wavelength than the OFDI source spectrum.
The separate source is preferably of higher power, and may be
combined with the source arm (using wavelength division,
multiplexer, grating, prism, filter or the like) travel to the
reference and sample arms and return back to the beam recombining
element. The returned separate source light can then be separated
from the OFDI light following transmission back through the beam
recombining element (i.e. beam splitter output). A separation
arrangement can perform spectral separation by a dispersing
element, such as a dichroic mirror, filter, grating, prism,
wavelength division multiplexer or the like. The separate source
will be detected separately from the OFDI light using one or more
detectors. The higher power provided by this separate source can
enable detection of a higher amplitude interference pattern, and
provide an improved input to the phase tracker, thus enabling more
stable phase tracking.
[0154] Referring now to FIG. 11, an in vivo image of a subject's
fingertip (300*500 pixels) acquired at an A-line scan rate of 15.7
kHz is shown using the exemplary embodiment of the system and
process according to the present invention. The optical sensitivity
was measured to be about -100 dB. The SNR is superior to an
equivalent TD OCT of the same A-line scan rate. The vertical line
noise arises due to an un-optimized detection when there is a
strong mirror-like reflection from the surface of the tissue, but
should preferably be eliminated substantially by a detection
optimization and/or an appropriate signal processing.
[0155] FIG. 12 shows another exemplary embodiment of a phase
tracker system 600 according to the present invention having an
extended phase lock range is provided. This done by combining a
fast element 602 (which may be provided, for example, as an
electro-optic (EO) phase modulator 602) to modulate the path length
difference over a small range, and a slower element 604 (which may,
for example, be provided as a Rapid Scanning Optical Delay (RSOD)
line 604) to modulate the path length over an extended range. The
detector 606 signal can be mixed with the phase modulator
modulation frequency 608 by a mixer 610 and low pass filtered
(filter not shown) to generate an error signal. The processing unit
612 preferably processes the error signal to generate an offset
voltage, and adds this offset voltage to the modulation signal 608,
so as to generate the output for the phase modulator driver 614. In
addition, the processing unit 612 can generate a signal to the RSOD
604 to provide extended range tracking of the phase over distances
of several millimeters. Light source 616, fiber splitter 618,
sample arm 620 and reference arm 622 are shown, and are described
herein.
[0156] The intensity I(t) at the detector at a given moment within
a single oscillation of the fringe pattern is given by
I(t)=cos [.phi.(t)]
where the phase .phi. gives the position in the fringe. For
.phi.=0, the signal is at a fringe maximum, for .phi.=.pi., the
signal is at a fringe minimum. At an arbitrary moment t, the phase
.phi.(t) is given by,
.phi.(t)=.alpha.+sin(.omega.t)
where .alpha. describes the position within a single oscillation of
the fringe pattern, and .beta.*sin(.omega.t) is the phase
modulation introduced by the phase modulator, with .beta. the
amplitude of the phase modulation, and .omega. the frequency of the
phase modulation signal. The intensity at the photodetector I(t)
can be mixed with a carrier at frequency .omega. and 2.omega.,
resulting in the mixer signal MixerC(t), MixerS(t),
Mixer2.omega.C(t) and Mixer2.omega.S(t),
MixerC(t)=cos(.omega.t)*cos(.alpha.+.beta. sin(.omega.t));
MixerS(t)=sin(.omega.t)*cos(.alpha.+.beta. sin(.omega.t));
Mixer2.omega.C(t)=cos(2.omega.t)*cos(.alpha.+.beta. sin(.omega.t));
Mixer2.omega.S(t)=sin(2.omega.t)*cos(.alpha.+sin(.omega.t))
[0157] The time average over a single oscillation of the carrier
frequency .omega. of MixerC, MixerS, Mixer2.omega.C and
Mixer2.omega.S is given by, MixerC(t)=0;
MixerS(t)=sin(.alpha.)*J.sub.1(.beta.);
Mixer2.omega.C(t)=cos(.alpha.)*J.sub.2(.beta.); Mixer2wS(t)=0,
where J.sub.1(.beta.) and J.sub.2(.beta.) are a Bessel functions of
the first kind; its value depends on .beta., the amplitude of the
phase modulation. Thus, the signal MixerS(t) and Mixer2.omega.C(t)
are proportional to sin(.alpha.) and cos(.alpha.), respectively,
with .alpha. the position within a single oscillation of the fringe
pattern. The mixer outputs MixerS(t) and Mixer2.omega.C(t) are used
as an error signal to generate an offset voltage to steer the phase
modulator to a new center position that minimizes the error signal,
and locks the interferometer output on a fringe maximum or minimum,
or a zero crossing, respectively. The complex spectral density can
now be determined by two consecutive tuning scans, one where the
error signal sin(.alpha.) is minimized, and the next where the
error signal cos(.alpha.) is minimized, resulting in a 90 degrees
phase shift between the two interference patterns. Using this
mixing arrangement, the complex spectral density can be obtained
rapidly and without resorting to an additional mechanical
arrangement for changing the phase of the reference arm light.
[0158] FIG. 13 shows a further exemplary embodiment of an OFDI
system 700 which includes a phase tracker for providing balanced
detection according to the present invention. In this exemplary
embodiment, a source 702 provides an electro-magnetic radiation
(e.g., light) which passes through a splitter 704, that sends part
of the light to a sample probe 706 and the remainder of the light
to a Rapid Scanning Optical Delay ("RSOD") line 708. Light is
passed from the RSOD 708 to the phase modulator PM 710. Light from
the phase modulator PM 710 is transmitted through a splitter 712,
and then through two additional splitters 714 and 716, a portion of
the output of which is sent as balanced detection outputs to
spectral detection units (not shown, but as described elsewhere
herein) and the remainder of the output is sent to the phase
tracker assembly 720. In the phase tracker assembly 720, phase
tracker detectors D.sub.1 and D.sub.2, 722 and 724, receive the
partial output of the pair of splitters 714 and 716, which in turn
send signal to a mixer 726 to generate an error signal. A
processing unit 728 processes the error signal, where the sum
generation of offset voltage and adds this to the modulation signal
730 to generate the output for the phase modulator driver 732.
Modulation signal, shown at box 730, is forwarded to the mixer 726
and the processing unit 726. In addition, the fringe amplitude
could be too small for the phase tracker to lock. Alternatively, a
secondary source with longer coherence length can be coupled to the
system 700, e.g., to provide a larger fringe amplitude to the phase
tracker.
[0159] FIGS. 14A-14C show an exemplary embodiment of a method for
tracking phase in an imaging system begins in processing blocks 750
and 752 according to the present invention by measuring a signal
received from the sample arm and then increasing a phase of the
signal. Processing of this exemplary method then proceeds to block
754, in which a first signal partition of the signal defined as
x.sub.1 is measured at least one peak of the signal. In decision
block 756, a determination is as to whether the signal defined as
x.sub.1 has been measured at least one peak of the signal. If in
decision block 756, it is determined that the signal defined as
x.sub.1 has been measured at at least one peak of the signal, then
processing returns to block 754 and the signal is again
measured.
[0160] On the other hand, if in decision block 756, it is
determined that the signal defined as x.sub.1 has not been measured
at at least one peak of the signal, then processing flows to a
decision block 758, where a determination is made as to whether to
adjust the signal. The adjustment may be, e.g., an increase or a
decrease in the phase of the signal by an incremental amount as
shown in blocks 760 and 762. Regardless of whether an increase or a
decrease in the phase of the signal is made, processing returns to
processing block 754, where a second signal partition of the signal
is measured at its peak. Blocks 756-762 are then repeated for such
measured signal. It should be noted that the functions of blocks
750-762 may be performed in parallel and/or series with other
imaging processes.
[0161] The adjustment of phase ".phi." can be defined as
A(x.sub.2-x.sub.1), where "A" is a constant and that the process of
determining whether to increase or decrease the phase of the signal
by an incremental amount may further comprise the substeps of (1)
determining whether A(x.sub.2-x.sub.1) is within range of the phase
modulator; and (2) changing .phi. by an amount equal to
A(x.sub.2-x.sub.1) if A(x.sub.2-x.sub.1) is within the range or
changing .phi. by an amount equal to A(x.sub.2-x.sub.1)-m2.pi. if
A(x.sub.2-x.sub.1) is outside of the range, where M is an integer
greater than 1. The method may optionally further comprise a
substep (3) re-measuring signal x.sub.1.
d. Data Processing
[0162] In general, the data recorded by the detector in time may
not be sampled as a strictly linear function of the optical
frequency .omega. or wave number k. The Fourier transform, however,
can link z and k space (or t and .omega.). Because of the
non-linear sampling in k, the acquired spectrum is interpolated to
create evenly spaced samples in the k domain. Alternatively, the
tuning slope of the laser could be adjusted in such a way that the
light is samples in equal intervals in k space, such that the
interpolation becomes obsolete. Alternatively, the detection timing
could be designed to sample the light evenly spread in the k
domain, such that the interpolation becomes obsolete. To achieve
the optimal point spread function, dispersion in the sample and
reference arm of the interferometer should preferably be balanced.
Dispersion imbalance can also be corrected by digital processing.
Phase chirping induced by motions can also be corrected by digital
processing. For the motion artifact correction, the axial movement
of the sample is measured, and a proper nonlinear mapping can be
calculated from the velocity of the movement.
[0163] Various interpolation techniques are known to those having
ordinary skill in the art. This includes, but is not limited to,
simple two-point interpolation, FFT zero-padding followed by
two-point interpolation, and rigorous interpolation with the sinc
function dictated by the Nyquist theorem.
[0164] An exemplary embodiment of the present invention may also
provide a probe for locating atherosclerotic plaque in a blood
vessel, comprising: an interferometer; a spectral separating unit
which splits signal received from the interferometer into a
plurality of optical frequencies; and a detector arrangement
capable of detecting at least a portion of the optical frequencies
received from the spectral separating unit.
e. Frequency Shifting Technique
[0165] For high-speed OFDI techniques, the maximum ranging depth
can likely be limited by the finite width of the coherence function
of the laser output because the coherence length is often
compromised to obtain higher tuning speed, higher output power, or
wider tuning range. The finite coherence length may cause the
visibility of the interference fringe to decrease as the path
length difference of the interferometer increases. This result in
the degradation of SNR, and therefore limits the maximum ranging
depth. Furthermore, the inability to distinguish between a positive
and negative electrical frequency in a conventional interferometry
may lead to the ambiguity between positive and negative depths. To
avoid the imaging folding artifact, the reference delay of the
interferometer should be adjusted so that the image presents at
only either positive or negative depth. This further may limit the
ranging depth for a given coherence length of the source.
[0166] To avoid such possible limitation, quadrature interference
signals have been measured based on active or passive phase biasing
using a piezoelectric actuator, birefringence plate or 3.times.3
coupler. These techniques may provide otherwise overlapping images
associated with positive and negative depths, but tended to leave
significant residual artifacts due to the difficulty of producing
stable quadrature signals. In this paper, we propose and
demonstrate a simple technique that effectively eliminates the
ambiguity between positive and negative depths.
[0167] The exemplary technique according to the exemplary
embodiment of the present invention uses an optical frequency
shifter in the interferometer to provide a constant frequency shift
of the detector signal. This allows both sides of the coherence
range to be used without crosstalk, and can double the ranging
depth. The same concept has been described above in the context of
1-dimensional optical frequency domain reflectometry using rotating
birefringence plates at 58 Hz or a recirculating frequency shifting
loop. In this exemplary embodiment, an acousto-optic frequency
shifter is used, and the technique is applied to high-speed OFDI
with several orders of magnitude faster ranging speed. Furthermore,
a signal processing technique according to a further exemplary
embodiment of the present invention is provided to accommodate a
nonlinear tuning slope of the swept source in the frequency
shifting technique.
A. Principle
[0168] Frequency Shift
[0169] FIG. 15 shows a high level diagram of the OFDI system
according to the present invention which includes a
wavelength-swept source 95, singlemode-fiber interferometer
employing an optical frequency shifter 311 in a reference arm 80, a
photodetector 88, and a signal processor 160. With a roundtrip
frequency shift of .DELTA.f in the reference arm, the photocurrent
associated with the interference between the reference and sample
light can be expressed as
t s ( t ) = .eta. P r ( t ) P s ( t ) .intg. R ( z ) G ( z ) cos [
4 .pi. c v ( t ) z + .phi. ( z ) + 2 .pi..DELTA. ft ] z ,
##EQU00010##
where .eta. denotes the quantum efficiency of the detector, Pr(t)
and Ps(t) the optical powers of the reference and sample arm light,
respectively, R(z) the reflectivity profile of the sample, G(|z|)
the coherence function corresponding to the fringe visibility, c
the speed of light, v(t) the optical frequency, and .phi.(z) the
phase of backscattering. In the case of a linear tuning, i.e.
v(t)=0-1t, the frequency of the detector signal is given by
f s = v 1 2 z c - .DELTA. f ##EQU00011##
[0170] The zero signal frequency corresponds to a depth
z=c.DELTA.f/(2v1). Therefore, by choosing the direction of
frequency shifting same as the tuning direction of the swept
source, the zero signal frequency can be made to point to a
negative depth. FIGS. 16(a) and 16(b) illustrate the effect of the
frequency shift. The fringe visibility or the coherence function
has a peak value at the zero depth and decrease as the depth
increases. The coherence length z.sub.c indicates the depth where
the visibility drops to 0.5 and thereby the SNR drops by 6 dB. One
may arguably define the effective ranging depth as the maximum
depth span where the SNR penalty is less than 6 dB. For example, in
FIG. 16(a), a single side of the coherence range can be used due to
the sign ambiguity of the signal frequency (hatched region). In
contrast, as shown in FIG. 16(b), with an appropriate frequency
shift, both sides of the coherence range from -z.sub.c to z.sub.c
can be utilized without any image crosstalk between the negative
and positive depths.
[0171] Nonlinear Tuning
[0172] Nonlinearity in v(t) with respect to time results in
frequency chirping of the signal at a constant depth and causes the
degradation of axial resolution. As a solution to this problem, the
detector signal may be sampled with nonlinear time interval
compensating for the frequency chirping. Alternatively, the
detector signal can be sampled with a constant time interval, and
then the sampled data be mapped to a uniform v-space by
interpolation prior to discrete Fourier transform ("DFT"). Both
methods have been demonstrated to yield a transform-limited axial
resolution. However, these methods are not applicable directly in
the frequency shifting technique. Both the nonlinear sampling and
interpolation method can result in artificial chirping of the
frequency shift, leading to sub optimal axial resolution. Thus, a
modified interpolation method can be used to achieve nearly
transform-limited axial resolution over the entire ranging depth.
The exemplary technique may be as follows: [0173] Step 1. Obtain N
samples of the signal with uniform time interval during each
wavelength sweep of the source. [0174] Step 2. Produce DFT of N
data points in the electrical frequency domain. [0175] Step 3.
Separate two frequency bands below and above .DELTA.f corresponding
to negative and positive depths, respectively. [0176] Step 4. Shift
each frequency band such that the zero depth is aligned to the zero
electrical frequency. [0177] Step 5. Apply zero-padding to each
frequency band and perform inverse DFT resulting in an array of
increased number of samples in the time domain with smaller time
interval for each frequency band. [0178] Step 6. Interpolate each
array in the time domain into a uniform v space using a proper
mapping function given by the nonlinearity of the swept source.
[0179] Step 7. Conduct DFT of each interpolated array. [0180] Step
8. Combine the two arrays (images) by shifting the array index.
[0181] As a result, the zero depth lies at the electrical frequency
of .DELTA.f.
B. Experiment
[0182] OFDI System
[0183] FIG. 17 depicts the experimental setup of an exemplary OFDI
system employing two acousto-optic frequency shifters (FS1 800 and
FS2 802, Brimrose Inc. AMF-25-1.3) according to an exemplary
embodiment of the present invention. The two frequency shifters may
be driven with voltage controlled oscillators to produce a net
shift of .DELTA.f=FS2-FS1. The use of two frequency shifters
balanced the material dispersion of the acousto-optic crystals
automatically. The insertion loss of each device including fiber
coupling may be less than 2.5 dB. The sampling rate of the
digitizer can be 10 MHz. The swept laser 100 may be constructed to
provide a tuning range of 108 nm centered swept from 1271 nm to
1379 nm (.nu.1=135 GHz/.mu.s). Although a repetition rate up to 36
kHz could be achieved, the laser was operated at a reduced rate of
7 kHz and 1300 samples were acquired during a single wavelength
sweep. This resulted in a depth span of 5.8 mm in the image
corresponding to the Nyquist frequency of 5 MHz. The probe 810 may
include a galvanometer mirror and an imaging lens produced a probe
beam with a confocal parameter of 1.1 mm. An optical tap coupler
820 can be used in conjunction with a narrowband filter 830 and a
photodetector 834 to generate a TTL trigger signal in an electrical
circuit 836. The TTL signal may be used as a trigger in analog to
digital conversion.
[0184] The interference signal can be measured using a dual
balanced receiver 151. The detector signal was further processed
prior to digitization using a low pass electrical filter 840. Other
types of electrical filters such as a band pass filter and a slope
filter. The transmission of the slope filter may have an
exponentially-rising (falling) slope in its low frequency band.
This filter may be useful in which attenuation is significant and
the resulting signal strength decays with depth. The slope filter
can improve the dynamic range of the detection by effectively
suppressing the large signal from the surface relative to that at
greater depths.
[0185] To characterize the coherence function of the swept laser
100, the point spread function of the system may be measured at
.DELTA.f=0 (FS1=-25 MHz, FS2=-25 MHz) with a partial reflector at
various locations of the reference mirror. For comparison, the
sampled data acquired at each depth was processed with and without
the mapping process. FIGS. 18(a) and 18 (b) show exemplary results,
where the y-axis represents the square of the DFT amplitudes
normalized to the value at zero frequency, and the bottom and top
x-axes represent the signal frequency and the depth z,
respectively. Without mapping, the point spread function suffers
from significant broadening and large degradation of the peak power
as the depth increases, because of the nonlinearity of our swept
laser [see FIG. 18(a)]. With the mapping process, however, the
spread function exhibits nearly transform-limited axial resolution
as shown in FIG. 18(b). The finite coherence length of the laser
output accounts for the decrease of the signal power depth. Over
the entire depth span of 5.8 mm, the SNR is reduced by more than 11
dB. According to the criterion for the effective ranging depth
introduced earlier, the depth corresponding to the coherence length
may be only 2.9 mm, a half the total in the image. The same
experiment was conducted with a nonzero frequency shift of
.DELTA.f=-2.5 MHz (FS1=-22.5 MHz, FS2=-25 MHz). FIGS. 18(c) and
18(d) show the point spread functions measured with and without the
mapping process, respectively. As shown in these figures, the peak
of the signal power occurring at the zero depth present at a
frequency of 2.5 MHz is at least approximately equal to the net
acousto-optic frequency shift. The nearly transform-limited axial
resolution observed in FIG. 18(d) validates the mapping technique.
The reduction in the signal power is less than 5 dB over the entire
depth span of 5.8 mm, demonstrating the benefit of the frequency
shifting technique in terms of extending the ranging depth.
[0186] Image
[0187] Exemplary imaging of a human lung tissue ex vivo was
conducted with the OFDI system. FIGS. 19A and 19B depict two
images/graphs, obtained under identical experimental conditions
except that .DELTA.f=0 for the image/graph in FIG. 19A and
.DELTA.f=-2.5 MHz for the image/graph in FIG. 19B. Each image/graph
was obtained using the mapping technique described above. The
surface of the tissue was placed with an angle with respect to the
probe beam axis, and the reference mirror was positioned such that
the signal was present at both positive and negative depths in the
image. In FIG. 19A, the tissue image is contained within the
effective ranging depth of 2.8 mm, i.e. the top half of the total
depth span. However, the relatively large variation in the sample
location resulted in the imaging folding artifact. In contrast, in
FIG. 19B the entire positive and negative depths could be displayed
without ambiguity taking advantage of the ranging depth increased
to 5.8 mm by the frequency shifting technique.
[0188] The foregoing merely illustrates the principles of the
invention. Various modifications and alterations to the described
embodiments will be apparent to those skilled in the art in view of
the teachings herein. It will thus be appreciated that those
skilled in the art will be able to devise numerous systems,
arrangements and methods which, although not explicitly shown or
described herein, embody the principles of the invention and are
thus within the spirit and scope of the present invention.
* * * * *