U.S. patent application number 13/158010 was filed with the patent office on 2012-12-13 for touch screen.
This patent application is currently assigned to Texas Instruments Incorporated. Invention is credited to Marco Corsi, Brian P. Ginsburg, Baher S. Haroun, Srinath M. Ramaswamy, Vijay B. Rentala, Eunyoung Seok.
Application Number | 20120313895 13/158010 |
Document ID | / |
Family ID | 47292780 |
Filed Date | 2012-12-13 |
United States Patent
Application |
20120313895 |
Kind Code |
A1 |
Haroun; Baher S. ; et
al. |
December 13, 2012 |
TOUCH SCREEN
Abstract
A method for determining the location of an object on a touch
panel is provided. Initially, a pulse of terahertz radiation is
transmitted through a touch panel, which formed of a dielectric
material such that the pulse generates a evanescent field in a
region adjacent to a touch surface of the touch panel. A reflected
pulse is generated by an object located within the region adjacent
to the touch surface of the touch panel, and a position of the
object on the touch surface of the touch panel is triangulated at
least in part from the reflected pulse.
Inventors: |
Haroun; Baher S.; (Allen,
TX) ; Corsi; Marco; (Allen, TX) ; Ginsburg;
Brian P.; (Allen, TX) ; Rentala; Vijay B.;
(Plano, TX) ; Ramaswamy; Srinath M.; (Murphy,
TX) ; Seok; Eunyoung; (Plano, TX) |
Assignee: |
Texas Instruments
Incorporated
Dallas
TX
|
Family ID: |
47292780 |
Appl. No.: |
13/158010 |
Filed: |
June 10, 2011 |
Current U.S.
Class: |
345/175 |
Current CPC
Class: |
G06F 3/0421 20130101;
G06F 3/046 20130101; G06F 2203/04109 20130101 |
Class at
Publication: |
345/175 |
International
Class: |
G06F 3/042 20060101
G06F003/042 |
Claims
1. A method comprising: transmitting a pulse of terahertz radiation
through a touch panel formed of a dielectric material such that the
pulse generates a evanescent field in a region adjacent to a touch
surface of the touch panel; generating a reflected pulse from an
object located within the region adjacent to the touch surface of
the touch panel; and triangulating a position of the object on the
touch surface of the touch panel at least in part from the
reflected pulse.
2. The method of claim 1, wherein the step of triangulating further
comprises: receiving the reflected pulse by a first receiver and a
second receiver that are separated from one another by a distance;
and calculating the position of the object based at least in part
on a first elapsed time between transmission and reception at the
first receiver, a second elapsed time between transmission and
reception at the second receiver, and the distance.
3. The method of claim 1, wherein the reflected pulse further
comprises a first reflected pulse, and wherein the step of
triangulating further comprises: generating a second reflected
pulse from a reflector included in the touch panel; receiving the
first and second reflected pulses by a receiver; and calculating
the position of the objected from the first and second reflected
pulses.
4. The method of claim 3, wherein the reflector is located along
the periphery of the touch panel.
5. The method of claim 1, wherein the pulse further comprises a
first pulse, and wherein the reflected pulse further comprises a
first reflected pulse, and wherein the first pulse is transmitted
by a first transceiver, and wherein the method further comprises:
transmitting a second pulse of terahertz radiation through a touch
panel by a second transceiver; and generating a second reflected
pulse from an object located within the region adjacent to the
touch surface of the touch panel.
6. The method of claim 5, wherein the step of triangulating further
comprises triangulating the location of the object on the touch
surface of the touch panel from the first and second reflected
pulses.
7. An apparatus comprising: a touch panel formed of a dielectric
material that is configured to carry terahertz radiation and having
a touch surface; and a touch controller that is optically coupled
to touch panel, wherein the touch controller is configured to
transmit a pulse of terahertz radiation through the touch panel so
as to generate a evanescent field in a region adjacent to the touch
surface, and wherein the touch controller is configured to receive
a reflected pulse that is generated by an object located within the
region, and wherein the touch controller is configured to
triangulate a position of the object on the touch surface at least
in part from the reflected pulse.
8. The apparatus of claim 7, wherein the touch controller further
comprises: a signaling circuit that is configured to generate and
receive terahertz radiation; and a control circuit that is coupled
to the signal circuit.
9. The apparatus of claim 8, wherein the signaling circuit further
comprises: a transceiver; a local oscillator that is coupled to the
transceiver; and a receiver circuitry that is coupled to the
transceiver.
10. The apparatus of claim 9, wherein the receiver circuitry
further comprises an analog baseband circuit that averages the
combined signal for a plurality of sampling periods within a
digitization window to generate a plurality of averaged signals and
that converts the plurality of averaged signals to a digital
signal.
11. The apparatus of claim 10, wherein the analog baseband circuit
further comprises: a clock circuit; a low noise amplifier (LNA)
that is coupled to the summing circuit; an averager that is coupled
to the LNA and the clock circuit; an analog-to-digital converter
(ADC) that is coupled to the LNA and the clock circuit; and an
output circuit that is coupled to the ADC.
12. The apparatus of claim 11, wherein the transmitter further
comprises: a transmitter that is coupled to the local oscillator;
and a plurality of receivers that are spaced apart from one another
and that are each coupled to the receiver circuitry.
13. The apparatus of claim 12, wherein the touch panel further
comprises a reflector.
14. A method comprising: transmitting a pulse of terahertz
radiation through a touch panel formed of a dielectric material
such that the pulse generates a evanescent field in a region
adjacent to a touch surface of the touch panel; generating a
plurality of reflected pulses, wherein a first reflected pulse is
generated by a object located within the region adjacent to the
touch surface of the touch panel; and triangulating a position of
the object on the touch surface of the touch panel at least in part
from the plurality of reflected pulses.
15. The method of claim 14, wherein the object further comprises a
first object, and wherein the position further comprises a first
position, and wherein a second reflected pulse of the plurality of
reflected pulses is generated by a object located within the region
adjacent to the touch surface of the touch panel and at a second
location.
16. The method of claim 15, wherein the step of triangulating
further comprises: receiving the plurality of reflected pulses by a
first receiver and a second receiver that are separated from one
another by a distance; and calculating the position of the object
based at least in part on the plurality of reflected pulses.
17. The method of claim 16, wherein a third reflected pulse of the
plurality of reflected pulses is generated by a reflector.
18. The method of claim 17, wherein the reflector is located along
the periphery of the touch panel.
Description
TECHNICAL FIELD
[0001] The invention relates generally to a touch screen and, more
particularly, to a terahertz-enabled touch screen.
BACKGROUND
[0002] Touch screens have become ubiquitous, being included in
mobile devices (i.e., phones) and other devices (i.e., tablet
computers). There is however difficulty in engineering larger
displays (such as for "black boards). Resistive and capacitive
touch panels for large scale applications can be expensive and
"power hungry," while projector based solutions suffer from
occlusion. Thus, there is a need for a touch sensitive system that
is scalable.
[0003] Some examples of conventional circuits and systems are:
Williams, "Filling the THz Gap," doi:10.1088/0034-4885/69/2/R01;
Heydari et al., "Low-Power mm-Wave Components up to 104 GHz in 90
nm CMOS," ISSCC 2007, pp. 200-201, February 2007, San Francisco,
Calif.; LaRocca et al., "Millimeter-Wave CMOS Digital Controlled
Artificial Dielectric Differential Mode Transmission Lines for
Reconfigurable ICs," IEEE MTT-S IMS, 2008; Scheir et al., "A 52 GHz
Phased-Array Receiver Front-End in 90 nm Digital CMOS" JSSC
December 2008, pp. 2651-2659; Straayer et al. "A Multi-Path Gated
Ring Oscillator TDC With First-Order Noise Shaping," IEEE J. of
Solid State Circuits, Vol. 44, No. 4, April 2009, pp. 1089-1098;
Huang, "Injection-Locked Oscillators with High-Order-Division
Operation for Microwave/Millimeter-wave Signal Generation,"
Dissertation, Oct. 9, 2007; Cohen et al., "A bidirectional TX/RX
four element phased-array at 60 HGz with RF-IF conversion block in
90 nm CMOS processes," 2009 IEEE Radio Freq. Integrated Circuits
Symposium, pp. 207-210; Koh et al., "A Millimeter-Wave (40-65 GHz)
16-Element Phased-Array Transmitter in 0.18-.mu.m SiGe BiCMOS
Technology," IEEE J. of Solid State Circuits, Vol. 44, No. 5, May
2009, pp. 1498-1509; York et al., "Injection- and Phase-locking
Techniques for Beam Control," IEEE Transactions on Microwave Theory
and Techniques, Vol. 46, No. 11, November 1998, pp. 1920-1929;
Buckwalter et al., "An Integrated Subharmonic Coupled-Oscillator
Scheme for a 60-GHz Phased Array Transmitter," IEEE Transactions on
Microwave Theory and Techniques, Vol. 54, No. 12, December 2006,
pp. 4271-4280; PCT Publ. No. WO2009028718; and U.S. Pat. No.
3,673,327.
SUMMARY
[0004] An embodiment of the present invention, accordingly,
provides a method. The method comprises transmitting a pulse of
terahertz radiation through a touch panel formed of a dielectric
material such that the pulse generates a evanescent field in a
region adjacent to a touch surface of the touch panel; generating a
reflected pulse from an object located within the region adjacent
to the touch surface of the touch panel; and triangulating a
position of the object on the touch surface of the touch panel at
least in part from the reflected pulse.
[0005] In accordance with an embodiment of the present invention,
the step of triangulating further comprises: receiving the
reflected pulse by a first receiver and a second receiver that are
separated from one another by a distance; and calculating the
position of the object based at least in part on a first elapsed
time between transmission and reception at the first receiver, a
second elapsed time between transmission and reception at the
second receiver, and the distance.
[0006] In accordance with an embodiment of the present invention,
the reflected pulse further comprises a first reflected pulse, and
wherein the step of triangulating further comprises: generating a
second reflected pulse from a reflector included in the touch
panel; receiving the first and second reflected pulses by a
receiver; and calculating the position of the objected from the
first and second reflected pulses.
[0007] In accordance with an embodiment of the present invention,
the reflector is located along the periphery of the touch
panel.
[0008] In accordance with an embodiment of the present invention,
the pulse further comprises a first pulse, and wherein the
reflected pulse further comprises a first reflected pulse, and
wherein the first pulse is transmitted by a first transceiver, and
wherein the method further comprises: transmitting a second pulse
of terahertz radiation through a touch panel by a second
transceiver; and generating a second reflected pulse from an object
located within the region adjacent to the touch surface of the
touch panel.
[0009] In accordance with an embodiment of the present invention,
the step of triangulating further comprises triangulating the
location of the object on the touch surface of the touch panel from
the first and second reflected pulses.
[0010] In accordance with an embodiment of the present invention,
an apparatus is provided. The apparatus comprises a touch panel
formed of a dielectric material that is configured to carry
terahertz radiation and having a touch surface; and a touch
controller that is optically coupled to touch panel, wherein the
touch controller is configured to transmit a pulse of terahertz
radiation through the touch panel so as to generate a evanescent
field in a region adjacent to the touch surface, and wherein the
touch controller is configured to receive a reflected pulse that is
generated by an object located within the region, and wherein the
touch controller is configured to triangulate a position of the
object on the touch surface at least in part from the reflected
pulse.
[0011] In accordance with an embodiment of the present invention,
the touch controller further comprises: a signaling circuit that is
configured to generate and receive terahertz radiation; and a
control circuit that is coupled to the signal circuit.
[0012] In accordance with an embodiment of the present invention,
the signaling circuit further comprises: a transceiver; a local
oscillator that is coupled to the transceiver; and a receiver
circuitry that is coupled to the transceiver.
[0013] In accordance with an embodiment of the present invention,
the receiver circuitry further comprises an analog baseband circuit
that averages the combined signal for a plurality of sampling
periods within a digitization window to generate a plurality of
averaged signals and that converts the plurality of averaged
signals to a digital signal.
[0014] In accordance with an embodiment of the present invention,
the analog baseband circuit further comprises: a clock circuit; a
low noise amplifier (LNA) that is coupled to the summing circuit;
an averager that is coupled to the LNA and the clock circuit; an
analog-to-digital converter (ADC) that is coupled to the LNA and
the clock circuit; and an output circuit that is coupled to the
ADC.
[0015] In accordance with an embodiment of the present invention,
the transmitter further comprises: a transmitter that is coupled to
the local oscillator; and a plurality of receivers that are spaced
apart from one another and that are each coupled to the receiver
circuitry.
[0016] In accordance with an embodiment of the present invention,
the touch panel further comprises a reflector.
[0017] In accordance with an embodiment of the present invention, a
method is provided. The method comprises transmitting a pulse of
terahertz radiation through a touch panel formed of a dielectric
material such that the pulse generates a evanescent field in a
region adjacent to a touch surface of the touch panel; generating a
plurality of reflected pulses, wherein a first reflected pulse is
generated by a object located within the region adjacent to the
touch surface of the touch panel; and triangulating a position of
the object on the touch surface of the touch panel at least in part
from the plurality of reflected pulses.
[0018] In accordance with an embodiment of the present invention,
the object further comprises a first object, and wherein the
position further comprises a first position, and wherein a second
reflected pulse of the plurality of reflected pulses is generated
by a object located within the region adjacent to the touch surface
of the touch panel and at a second location.
[0019] In accordance with an embodiment of the present invention,
the step of triangulating further comprises: receiving the
plurality of reflected pulses by a first receiver and a second
receiver that are separated from one another by a distance; and
calculating the position of the object based at least in part on
the plurality of reflected pulses.
[0020] In accordance with an embodiment of the present invention, a
third reflected pulse of the plurality of reflected pulses is
generated by a reflector.
[0021] The foregoing has outlined rather broadly the features and
technical advantages of the present invention in order that the
detailed description of the invention that follows may be better
understood. Additional features and advantages of the invention
will be described hereinafter which form the subject of the claims
of the invention. It should be appreciated by those skilled in the
art that the conception and the specific embodiment disclosed may
be readily utilized as a basis for modifying or designing other
structures for carrying out the same purposes of the present
invention. It should also be realized by those skilled in the art
that such equivalent constructions do not depart from the spirit
and scope of the invention as set forth in the appended claims.
BRIEF DESCRIPTION OF THE DRAWINGS
[0022] For a more complete understanding of the present invention,
and the advantages thereof, reference is now made to the following
descriptions taken in conjunction with the accompanying drawings,
in which:
[0023] FIG. 1 is an example of touch screen system in accordance
with an embodiment of the present invention;
[0024] FIG. 2 is diagram an example of a portion of the operation
of the system of FIG. 1;
[0025] FIG. 3 is a cross-sectional diagram of FIG. 2 along section
line I-I;
[0026] FIG. 4 is diagram an example of a portion of the operation
of the system of FIG. 1;
[0027] FIG. 5 is a cross-sectional diagram of FIG. 2 along section
line II-II;
[0028] FIG. 6 is a block diagram of an example of a phased array
system that can be employed with the signaling circuit of FIGS.
1-5;
[0029] FIG. 7 is a block diagram of an example of the local
oscillator (LO) of FIG. 6;
[0030] FIG. 8-11 are block diagrams of examples of the transceivers
of FIG. 6;
[0031] FIG. 12 is a circuit diagram of an example of the
multipliers of FIGS. 8-11;
[0032] FIG. 13 is a circuit diagram of an example of the phase
shifters of FIGS. 8-11;
[0033] FIG. 14 is a circuit diagram of an example of the injection
locked voltage controlled oscillator (ILVCO) of FIGS. 8-11;
[0034] FIG. 15 is a circuit diagram of an example of the power
amplifier (PA) and low noise amplifier (LNA) of FIGS. 8-11;
[0035] FIG. 16 is block diagram of an example of the radiator of
FIGS. 8-11;
[0036] FIG. 17 is a block diagram of the phased array system of
FIG. 6;
[0037] FIGS. 18 and 19 are timing diagrams that depict examples of
the operation of the phased array system of FIG. 6;
[0038] FIG. 20 is a circuit diagram of an example of the switches
of FIG. 6;
[0039] FIG. 21 is a circuit diagram of an example of the
analog-to-digital converters (ADCs) of FIG. 6;
[0040] FIG. 22 is a circuit diagram of the low pass/band pass
filter of FIG. 17;
[0041] FIGS. 23 and 24 are circuit diagrams of an example of a time
to digital converter used with the ADCs of FIG. 6;
[0042] FIG. 25 is a timing diagram depicting an example of the
operation of the time to digital converter of FIGS. 23 and 24;
[0043] FIG. 26 is a circuit diagram of an example of the summing
circuit for the receiver circuitry of FIG. 6;
[0044] FIG. 27 is a diagram of an example of the receiver circuitry
of FIG. 6;
[0045] FIG. 28 is a diagram of an example of the analog baseband
circuit of FIG. 27;
[0046] FIG. 29 is a diagram of an example of the averagers of FIG.
28,
[0047] FIG. 30 is a diagram of an example of the switched capacitor
banks of FIG. 29; and
[0048] FIGS. 31 and 32 are diagrams demonstrating the operation of
the system of FIG. 6.
DETAILED DESCRIPTION
[0049] Refer now to the drawings wherein depicted elements are, for
the sake of clarity, not necessarily shown to scale and wherein
like or similar elements are designated by the same reference
numeral through the several views.
[0050] This application incorporated by reference co-pending U.S.
patent application Ser. No. 12/871,626, entitled "DOWNCONVERSION
MIXER," filed on Aug. 30, 2010; co-pending U.S. patent application
Ser. No. 12/878,484, entitled "TERAHERTZ PHASED ARRAY SYSTEM,"
filed on Sep. 9, 2010; and co-pending U.S. patent application Ser.
No. 12/871,626, entitled "ANALOG BASEBAND SYSTEM FOR A TERAHERTZ
PHASED ARRAY SYSTEM," filed on Apr. 12, 2011.
[0051] Turning to FIG. 1, an example of a system 100 in accordance
with an embodiment of the present invention can be seen. As shown,
the system 100 generally comprises a touch panel 102, control
circuit 104, signaling circuit 106, and reflectors 107-1 to 107-4.
Generally, the touch panel 102 is a large panel (i.e., 2 m.times.2
m) formed of a sheet of dielectric material (such as glass or
polyethylene) that is capable of carrying terahertz radiation
(which typically has a frequency range between 0.1 THz and 10 THz)
and functioning as a dielectric waveguide for this terahertz
radiation. Typically, this touch pane 102 can also include a
transparent cladding 112 (as shown in FIG. 3) located between the
touch panel 102 and display (i.e., liquid crystal display or LCD
screen). The signaling circuit generally comprises transmission and
receiver circuit (i.e., such as the phased array system 200
detailed below) that is optically coupled to the touch panel so as
to transmit and receive terahertz radiation through the touch panel
102. There are a number of configurations that the signaling
circuit 106 may have such as a single transceiver (i.e., system
200), multiple transceivers, or a signal transmitter with multiple
receivers. The control circuit 104 may includes logic or a
processor (such as a digital signals processor or DSP) that is
configured to calculate object location on the touch surface of the
touch panel 102 and to control the signaling circuit. Additionally,
reflectors 107-1 to 107-4 may be optionally included to assist in
triangulating an object location and may be included at any
position (including the periphery as shown) on the touch panel 102.
There may be any number of reflectors, but one likely configuration
is to include the reflectors 107-1 to 107-4 at each corner of the
touch panel 102.
[0052] Because terahertz radiation exhibits optical behavior, it
can be transmitted through the touch panel 102 in a similar manner
infrared radiation being transmitted through an optical fiber. As a
result, the terahertz radiation generates an evanescent field in an
evanescent field region 114 that is adjacent to the touch surface
of the touch panel 102. When transmitted through fibers or other
transmission media, infrared, visible spectrum, and ultraviolet
radiation also generate an evanescent field in an evanescent field
region, but this region for infrared, visible spectrum, and
ultraviolet radiation is much smaller than evanescent field region
114 because of the frequencies. Thus, because the evanescent field
region 114 is so much larger, it is much easier for interference
within this region 114 to be detected, which can be seen in FIGS.
2-5.
[0053] In the example shown in FIGS. 2-5, an object an object
(i.e., a finger or pointer) 110 is located with in the evanescent
field region 114 at a location on the touch surface of the touch
panel 102. When the signing circuit 106 generates a pulse 108 of
terahertz radiation, this transmitted pulse 108 propagates through
the touch panel 102 and generates an undistorted evanescent field
116 within region 116. When the undistorted evanescent field 116
encounters the object 110 that is located within region 114,
evanescent field 116 becomes distorted (as shown with the distorted
evanescent field 120 of FIG. 5), and a reflected pulse 118 is
generated. This reflected pulse 118 then propagates back to the
signaling circuit 106. Based at least in part on the reception of
this reflected pulse 118, the control circuit 104 can triangulate
the position of the object on the touch surface of the touch panel
102. Typically, two or more receivers (which are spaced apart from
one another) are employed and the time-of-flight (or time elapsed
from transmission to reception) along with the spacing between
receivers within signaling circuit 106 can be used for
triangulation. Alternatively or additionally, reflectors 107-1 to
107-4 (which have been omitted from FIGS. 2-5) may be employed to
assist in performing triangulation.
[0054] Turning to FIG. 6, a phased array system 200 (which can be
used as part of signaling circuitry 106) can be seen. The phase
array system 200 generally comprises a LO 202, a phased array 224,
a distribution network 226, receiver circuitry 228, and controller
208. The phased array 224 generally comprises several transceivers
204-1 to 204-N arranged in an array. The distribution network 226
generally comprises amplifiers 206 and 208-1 to 208-N. Additionally
the receiver circuitry generally comprises a summing circuit 210, a
mixer 212, amplifier 214, filter 216, switches 218-1 to 218-N,
variable selector 220, and ADCs 222-1 to 222-N. While a phased
array may not be necessary (i.e., a signal transceiver 204 may be
employed), it may be preferable in order to allow for optical
alignment with the touch panel 102. Additionally, detection of a
single object (i.e., object 110) is shown in FIGS. 2-5, but
detection of multiple objects (so-called "multi-touch") can be
performed (which typically uses multiple reflectors to assist in
eliminated background noise and more accurately determining the
positions for the objects).
[0055] In operation, phased array system 200 (which is generally
incorporated into an integrated circuit or IC) can generate a short
range radar system that operates in the terahertz frequency range
(which is generally between 0.1 THz and 10 THz). To accomplish
this, local oscillator 202 generates a high frequency signal FL01
that is on the order of tens to hundreds of gigahertz (i.e., 40
GHz, 50 GHz, 67 GHz, and 100 GHz.) and a pulse signal TPUSLE. The
distribution network 226 then provides signal FL01 to each of the
transceivers 204-1 to 204-N such that the signals received by each
of transceivers 104-1 to 204-N are substantially in-phase. A
controller 208 provides a control signal to array 224, which
phase-adjusts the transceivers 204-1 to 204-N with respect to one
another to direct a beam of terahertz frequency radiation. The
transceivers 204-1 to 204-N can then receive reflected radiation
back from a target, which is provided to summing circuit 210. The
output of summing circuit 210 is the converted to a digital signal
by a mixer 212, amplifier 214, filter 216, switches 218-1 to 218-N,
variable selector 220, and ADCs 222-1 to 222-N. Additionally, mixer
212 can receive a divided signal from LO 202 (i.e., FL01/2 or
another synthesized signal) or can be removed (typically for 40
GHtz or less).
[0056] Generally, this phased array system 200 has several
different types of operational modes: pulsed, continuous, and
stepped frequency. For a pulsed operational mode, a pulse of
terahertz radiation is directed toward a target. The continuous
operational mode uses a continuously generated beam, which is
generally accomplished by effective "shutting off" the pulse signal
TPULSE. Finally, stepped frequency allows to frequency of the
terahertz beam to be changed, which can be accomplished by
employing a bank of local oscillators (i.e., 202). For the pulsed
operational mode, in particular, the range of the system 200 is
governed by the following equation:
R = .sigma. PG 2 .lamda. nE ( n ) ( 4 .pi. ) 3 kTBF ( S N ) 4 , ( 1
) ##EQU00001##
where: [0057] R is distance that can be measured or range; [0058] a
is the radar cross section of the target (usually not equal to the
physical cross section); [0059] S/N is single pulse SNR at the
intermediate frequency IF filter output (envelope detector input);
[0060] kTB is the effective incoming noise power in receiver
bandwidth B (B.apprxeq.1/pulsewidth); [0061] F is noise figure of
the receiver (derived parameter); [0062] P is the peak transmitter
power; [0063] G is the antenna power gain; [0064] .lamda. is
wavelength of the radiation (i.e., for 200 GHz, .apprxeq.1.5 mm);
[0065] n is number of integrations of pulses in the receiver
(multi-pulse averaging); and [0066] E(n) is the efficiency of
integration. For a monolithically integrated, low power IC that
includes system 200, this range is generally less than one
meter.
[0067] Turning to FIG. 7, an example of the LO 202 can be seen in
greater detail. Generally, this LO 202 employs a phase locked loop
(PLL) 326 that generates signals FL01 and FL02 from reference
signal REF and employs counter 322 and pulse generator 324 to
produce the pulse signal TPULSE. PLL 326 is generally comprised of
a phase detector 302, charge pump 304, low pass filter 304,
amplifiers 310 and 312, voltage controlled oscillator (VCO) 308,
and dividers 320, 318, 316, and 314. In operation, the phase
detector 302 receives a feedback signal from divider 314 and the
reference signal REF, and (along with charge pump 304 and low pass
filter 306) generates a tuning voltage for VCO 308. Typically, VCO
308 generates a high frequency signal (i.e., 100 GHz, 67 GHz, 50
GHz, or 40 GHz) which is amplified by amplifiers 310 and 312,
producing signal FL01. Divider 320 (which is generally an
injection-locked, divide-by-2 frequency divider) receives the
output of amplifier to output signal FL02. Signal FL02 is then
provided to divider 318 (which is generally a divide-by-2 current
mode logic divider). The output of divider 318 is provided to
divider 316 (which is generally a divide-by-8 current mode logic
divider), and the output of divider 316 is provided to divider 314
(which is generally a divide-by-M CMOS divider) to generate the
feedback signal. The counter 322 generates a count signal based on
a control signal CNTL and the feedback signal from divider 314, and
the pulse generator 234 produces the pulse signal TPULSE based at
least in part on the count signal from counter 322 and the outputs
of dividers 318 and 320.
[0068] In FIG. 8, an example of one of transceivers 202-1 to 202-N
(referred to as 202-A) can be seen in greater detail. As shown,
transceiver 202-A generally includes a transmit path 402-A and a
receive path 404-A that are each coupled to radiator 426 (i.e.,
antenna). During transmission, phase shifter 404 (which is
generally controlled by controller 230) receives signal FL01 from
LO 202 and phase-shifts signal FL01 accordingly. This phase shifted
signal is amplified by amplifier 408 and multiplied by multiplier
410-A (which is typically a multiply-by-3 multiplier) that allows
the signal FL01 to be increase to the desired frequency range. For
example, if signal FL01 is about 67 GHz, then multiplier 410-A
would output a signal having a frequency of about 201 GHz. This
multiplied signal is provided to ILVCO 412, which is generally used
to compensate for losses from multiplier 410-A. Additionally, ILVCO
412 receives the pulse signal TPULSE. Power amplifier (PA) 414 then
amplifies the output of ILVCO 412 for transmission through radiator
426. Typically, the pulse widths of pulse signal TPULSE are about
30 ps, 60 ps, or 90 ps when the signal FL01 has a frequency of
about 67 GHz. During reception, radiator 426 receives a signal,
which is amplified by amplifier 420. This amplified signal is mixed
with a signal having a frequency that is a multiple of signal FL01.
Typically, multiplier 416 (which is generally a multiply-by-2
multiplier) receives an output from amplifier 408, and the result
is amplified by amplifier 418 and provided to mixer 422. The mixed
output is then amplified by amplifier 424 and provided to summing
circuit 210. Additionally, mixer 422 is described in co-pending of
U.S. patent application Ser. No. 12/871,626 entitled
"DOWNCONVERSION MIXER."
[0069] Looking to FIG. 9, an alternative configuration for one of
transceivers 202-1 to 202-N (referred to as 202-B) can be seen in
greater detail. The transmit path 402-B is similar to transmit path
402-A; however, multiplier 410-B has replaced multiplier 410-B.
Generally, multiplier 410-B has a large range than multiplier 410-B
to accommodate a lower frequency signal FL01. For example, if
signal FL01 has a frequency of 50 GHz, then multiplier 410-B can be
a multiply-by-4 multiplier to generate a signal that is on the
order of 200 GHz. Additionally, to accommodate a lower frequency
signal FL01, receive path 404-B includes a mixer 428 that mixes the
outputs of amplifiers 424 and 408 and an amplifier 430. Also, the
pulse widths of pulse signal TPULSE can be about 40 ps or 80 ps
when the signal FL01 has a frequency of about 50 GHz.
[0070] Turning to FIG. 10, yet another alternative one of
transceivers 202-1 to 202-N (referred to as 202-C) can be seen in
greater detail. Here, D flip-flop 432 has been included in the path
for the pulse signal TPULSE; namely, the input terminal of
flip-flop 432 receives the pulse signal TPULSE, while flip-flop is
clocked by the output of amplifier 408. Additionally, multiplier
416 and amplifier 418 have been replaced by amplifier 434. This
arrangement is generally useful for even lower frequency ranges
(i.e., 40 GHz), which can produce pulse widths for pulse signal
TPULSE are about 50 ps or 100 ps.
[0071] In FIG. 11, another alternative one of transceivers 202-1 to
202-N (referred to as 202-D) can be seen in greater detail. Here,
the transmit path 402-D is similar to path 204-A; however,
multiplier 410-A has been replaced with multiplier 410-D, while
amplifier 408 has been removed. Multiplier 410-D generally has a
lower range to accommodate a signal FL01 with a high frequency. For
example, if signal FL01 has a frequency of about 100 GHz, then
multiplier 410-D can be a multiply-by-2 multiplier. Additionally,
for receive path 404-D, multiplier 416 and amplifier 418 have been
removed so that mixer 422 mixes the output of LNA 420 with the
output of phase shifter 406.
[0072] Turning now to FIG. 12, a circuit diagram of an example of
multipliers 410 and/or 416 can be seen. This type of multiplier 410
and/or 416 is generally employed within transceivers 202-1 to 202-N
to produce very high frequencies (i.e., 200 GHz) because direct
production of these high frequency signals is very difficult.
Generally, multiplier 410 and/or 416 employs a differential choke
802, a rectifying interleaver 804, and a VCO 806. Typically, VCO
806 uses two oscillator tanks to generate two pairs of output
signals from differential in-phase signals VIP and VIM and
differential quadrature signals VQM and VQP. Typically, VCO 806
comprises MOS transistors Q5 through Q12, inductors L3 through L6,
and capacitors C1 and C2. Rectifying interleaver 804 employs two
differential pairs of transistors Q1/Q2 and Q3/Q4 and current
sources 810 and 812 to interleave the outputs from VCO 806 to
generate a single-ended output signal OUT. Additionally, a
termination 808 and inductors L1 and L2 (from differential choke
802) are coupled to the rectifying interleaver 804. Typically,
power output is sufficient to lock ILCVO 412 (i.e., -20 bBm).
[0073] In FIG. 13, an example of phase adjuster 406 can be seen.
Here, a differential input signal IN (which is generally signal
FL01 from LO 202) is provided to differential pairs of MOS
transistors Q13/14, Q15/Q16, Q17/Q18, and Q19/Q20 (which are also
coupled to inductors L7 and L8). Based on control signals VC1
through VC4 received from controller 236, transistors Q21 through
Q24 can activate the differential pairs Q13/14, Q15/Q16, Q17/Q18,
and Q19/Q20 to generate a phase rotation of the differential input
signal IN, having a total phase shift range of less than about
.+-.22.5.degree.. Typically, phase shifting is performed in the
lower frequency domain (i.e., 50 GHz) to generally ease any
bandwidth requirements and efficiently recover power losses.
[0074] Turning to FIG. 14, a circuit diagram of an example of ILVCO
412 can be seen. ILVCO 412 is generally employed because of the
losses from multiplier 410. Theoretically, ILVCO 412 can provide an
infinite gain if the center frequencies match with a finite gain
throughout the locking range. Typically, MOS transistors Q25 and
Q28 are coupled at their respective gates to balun 1002, which
receives an output from multiplier 410 (i.e., 410-A, 410-B, or
410-C). In an alternative configuration, MOS transistor Q28 can
receive receives an output from multiplier 410 (i.e., 410-A, 410-B,
or 410-C) at its gate, while MOS transistor Q28 receives the pulse
signal TPULSE at its gate. These transistors Q25 and Q28 are
generally coupled in parallel to a gain stage (which is generally
comprised of cross-coupled MOS transistors Q26 and Q27) and the
oscillator tank (which is generally comprised of capacitors C3 and
C4 and inductors L9 and L10). Alternatively, the second harmonic of
the output can be used instead of first harmonic to relax any
tuning range requirements, but with reduced output power. As an
illustration, the properties of ILVCO 412 can be seen in Table 1
below using both the first and second harmonics.
TABLE-US-00001 TABLE 1 Targets First Harmonic Second Harmonic Input
Frequency [GHz] 200 100 Output Frequency [GHz] 200 200 Power Output
[dBm] -12 -12 Phase locking @200 GHz @100 GHz
[0075] In FIG. 15, a circuit diagram for an example of PA 414
and/or LNA 420 can be seen. Generally, the PA 414 and/or LNA 420
can provide linear amplification and isolation, and one of the
features of PA 414 and/or LNA 420 is its ability to be power gated
with a fast pulse time (i.e., tens of picoseconds). PA 414 and/or
LNA 420 generally comprise inductors L11 through L15, capacitors C5
through C7, and transistors Q29 and Q30. Here, the capacitors C5
through C7 are resonated by series or shunt inductors L11 through
L15 to provide the amplification with transistors Q29 and Q30.
Additionally, the input and output of PA 414 and/or LNA 420 can be
matched input or output impedances. For example, for PA 414, the
output impedance can be matched to the radiator 426. Moreover, the
circuit shown in FIG. 11 can be cascaded in multiple stages, where
the gain can be between 0 and 2 dB per stage.
[0076] Turning to FIG. 16, an example of a radiation 426 can be
seen. Here, radiator 426 is shown as being a patch antenna formed
over a substrate 210. This patch antenna generally comprises a
patch 1204 having slots 128 that are generally parallel to ground
strips and radiating edges 1202. For a frequency of about 410 GHz
(which has a wavelength of about 0.75 mm in air), the width W and
length L of patch 1204 are each about 200 .mu.m, while the slots
are 2 .mu.m wide. The proportions of the patch antenna can then be
varied so as to accommodate a desired emission frequency (and
wavelength). These radiators 426 (i.e., patch antennas) can then be
formed into an array as shown in FIG. 17. Alternatively, radiator
426 can be a bondwire Yagi-Uda antenna.
[0077] Because the data bandwidth of system 200 is very high (i.e.,
on the order of tens of gigahertz), it is generally impractical to
employ an ADC that digitizes the signals receives through by the
receiver circuitry 228. In FIGS. 18 and 19, timing diagrams can be
seen that generally depict the operation of the receiver circuitry
228, where each uses a trigger signal to reconstruct the received
signal. For FIG. 18, variable selector 220 actuates switches 218-1
to 218-N at various periods (i.e., .DELTA..sub.1 to .DELTA..sub.4)
following the trigger signal to allow each of the ADCs 222-1 to
222-N to resolves a portion of the received signal. FIG. 19, on the
other hand, use an envelop signal following the periods (i.e.,
.DELTA..sub.1 to .DELTA..sub.4) as part of the control mechanism
for switches 218-1 to 218-N.
[0078] To accomplish this, there are several approaches that can be
taken. In FIG. 20, an example for one arrangement can be seen. In
this arrangement, the switches 218-1 to 218-N are comprised of
zener diodes D1 to DN, capacitors CS1 to CSN, and pulse circuits
1602-1 to 1602-N (which are generally controlled by the variable
selector 220). These switches 218-1 to 218-N operate as an input
sampling network where each capacitor CS1 to CSN is coupled to a
"slow" ADC 222-1 to 222-N. Generally, this approach may require
very small apertures and very accurate clock generation.
[0079] Another arrangement can be seen in FIG. 21. For this
arrangement, ADCs 222-1 to 222-N (referred to as 222) are low
pass/band pass sigma-delta converters that can directly digitize
about a 10 GHz bandwidth with a clock of about 100 GHz. ADC 222
generally comprises a filter 1702, a quantizer 1704, a delay 1712,
a digital-to-analog converter (DAC) 1714, and amplifiers 1716 and
1718. The quantizer 1704 generally comprises quantizers 1706-1 and
1706-2, clock divider 1710, and multiplexer 1708. In operation, a
feedback signal (which is amplified by amplifier 1718) is combined
with the input signal and filtered by filter 1702. This filtered
output is combined with the feedback signal (which is amplified by
amplifier 1716). Quantizer 1704 (which is generally an 2-bit, 2-way
interleaved quantizer operating at 1.5 GHz) quantizes the signal
(which is then delayed by delay 1712 and converted to a feedback
signal by DAC 1714).
[0080] The filter 1702 can be seen in greater detail in FIG. 22. In
particular, the filter 1702 operates as amplifier and LC filter. To
accomplish this, filter 1702 generally comprises a trasconductor
cell 1804 (which generally comprises transistors Q31 through Q36,
linearizer 1802 and switches S1 and S2) and a negative
transconductor cell 1806 (which generally comprises transistors Q37
through Q40) that are each coupled to an LC circuit 1808 (which
generally comprises inductors L16 and L17 and capacitor C8).
[0081] Yet another approach can be seen in FIGS. 23, 24, and 25.
Here, a time to digital converter 1902 is coupled to each ADC 222-1
to 222-N; only one ADC, labeled 222, is shown, however. This
converter 1902 has sub-picosecond resolution and, in operation,
enabled when the input signal transitions to logic high or "1."
This activates the gated ring oscillator 1904 so that the counters
1906 can performed counting operations from the taps of the
oscillator 1904. The outputs from the counters 1904 can then be
summed and stored in register 1904.
[0082] Turning to FIG. 26, a circuit diagram of an example of
summing circuit 210 can be seen. Typically, summing circuit 210 is
a summing amplifier that is formed as a summing amplifier tree. As
shown in FIG. 20, each summing circuit or summing amplifier 2002
receives a pair of input signals. At the first stage 2004-1 of the
tree each summing circuit 2002 is coupled to a pair of transceivers
(i.e., 204-1 and 204-1). Then each subsequence stage (i.e., 2004-2)
receives input signals from a pair of summing circuits 2002 from
the previous stage (i.e., 2004-1). As a result the tree has a depth
of log.sub.2 N, where N is the number of transceivers 204-1 to
204-N.
[0083] As stated above, for a monolithically integrated, low power
IC that includes system 200, this range is generally less than one
meter. Thus, it should be apparent that in the terahertz frequency
range, there is a shortage of available power, which results in
decreased sensitivity, and with other frequency range systems being
available that have fewer limitations than terahertz systems,
transmission and reception in the terahertz range usually becomes
attractive when there is a large increase in available bandwidth.
However, transmitting, receiving, and digitizing such large
bandwidths (i.e., >10 GHz) can be problematic due at least in
part on analog-to-digital converter (ADC) performance
requirements.
[0084] These issues, though, are addressed in system 200. In
particular, system 200 generally employs an increased pulse
repetition frequency (PRF) of the terahertz radar so as to reduce
coherency losses due to target motion. By making use of a high PRF,
a small portion (subset) of the total available time for reception
can be digitized, and by scanning this subset rapidly, it is
possible to generate the full reception interval, reducing the
overhead for a very high sampling frequency on the ADC. The high
PRF can also generally ensure that it is possible to digitize the
desired reception interval very quickly. Additionally, because of
the lack of signal power, most signals should include baseband
averaging of pulse reception, in system 200 some averaging is
performed in the analog domain so as to reduce the ADC and
digitization conversion rate to be equal to the PRF, which is an
easily manageable task.
[0085] Turning to FIGS. 27 and 28, alternative receiver circuitry
228 can be seen. As shown, this circuitry 228 includes an analog
baseband circuit 2116, which performs the analog averaging and
digitization for system 200. The analog baseband circuit 2116
generally comprises an in-phase or I channel 3001, a quadrature or
Q channel 3003, a clock circuit 3005, and an output circuit 3014.
Each of these channels 3001 and 3003 generally and respectively
includes a low noise amplifier (LNA) 3002-1 and 3002-2, an averager
3004-1 and 3004-2, an amplifier 3006-1 and 3006-2, and an ADC
3008-1 and 3008-2. The clock circuit 3005 generally comprises a
clock generator 3010 (which can generate an ADC clock signal
ADCCLK[L] and a clear signal CLR[L]) and a DLL 3012 (which can
generate a sample clock signal SAMPLECLK[L]).
[0086] In operation, a digital output signal RXDATA and clock
signal ADCCLKOUT are generated from the baseband input signals BBI
and BBQ and DLL clock signal RXDLL. Typically, BBI and BBQ are
differential signal (as shown), but may also be single-ended. These
I and Q baseband signals BBI and BBQ (which are generally received
from the summing circuitry 210) are respectively amplified by
amplifiers 3002-1 and 3002-2. Because there are difficulties in
digitizing the high bandwidth (as explained above), the performance
requirements for ADCs 3008-1 and 3008-2 can be reduced by averaging
the output of LNAs 3002-1 and 3002-1 with averagers 3004-1 and
3004-2.
[0087] The averagers 3008-1 and 3008-2 (which can be seen in
greater detail in FIGS. 29 and 30) generally comprise switched
capacitor banks 4002-1 to 4002-R with each bank having several
branches 5002-1 to 5002-J; for example and as shown in FIG. 30,
each branch (which is labeled 4002) has J branches. As with the
baseband signals BBQ and BBI, branches 5002-1 to 5002-J are
arranged to receive differential signals, but branches 502-1 to
502-J can be arranged to receive single-ended signals. These
branches 5002-1 to 5002-J generally and respectively comprise
sample switches S1-1 to S1-J and S5-1 to S5-J, capacitors C1-1 to
C1-J and C2-1 to C2-J, clear switches S3-1 to S3-J and S4-1 to
S4-J, and output switches S2-1 to S2-J and S6-1 to S6-J. The sample
switches S1-1 to S1-J and S5-1 to S5-J are each generally coupled
to a tap of the DLL 3014 so as to receive branch sample signals
SAMPLE1 to SAMPLEL, respectively (where sample clock signal
SAMPLECLK[L] is generally comprised of clock signals SAMPLE1 to
SAMPLL). Moreover, the clear signal CLR[L] (which generally
comprises branch clear signals CLR1 to CLRL) can actuate switches
S3-1 to S3-J and S4-1 to S4-J to discharge capacitors C1-1 to C1-J
and C2-1 to C2-J, while the output switches S2-1 to S2-J and S6-1
to S6-J are actuated by the ADC clock signal ADCCLK[L] (which
generally comprises branch readout signals ADCCLK1 to ADCCLKL).
[0088] Turning to FIG. 31, an example of the operation of the
analog baseband circuit 2116 (and system 200) can be seen.
Typically, the controller 230 adjusts the phase shift for each of
the transceivers 206-1 to 206-N (for this example) to direct a beam
of terahertz radiation emitted from the phased array 204 so as to
be aligned and optically coupled with the touch panel 102. This
emitted radiation is in the form of a pulse that can be directed
toward the touch panel 102 so that reflected radiation (i.e., from
the object 110) can be received by the transceivers 206-1 to 206-N.
These transmitted pulses TXPulse can (for example) each a width of
about 100 ps that would correspond to a distance of about 1.5 cm
and can be separated from one another by at least an unambiguous
range or duration 6002 (which allows ample time for reset and
detection) between times TO and TPRI (which is the pulse repetition
interval). This unambiguous range 6002 can, for example, be 9.9 ns
or 1.485 m, which can correspond to a 100 MHz pulsing frequency.
Within this unambiguous range 6002, there is a scan range 6004
between the minimum and maximum target distances and unused ranges
6007 and 6008. The minimum target distance is generally dictated by
far field conditions and may be, for example, about 3 cm, while the
maximum target distance is generally limited by the available power
reflected by the target and sensitivity of the transceivers 204-1
to 204-N (which may be, for example, about 24 cm). The scan range
6004 can be divided into number of range cells (not shown in FIG.
31 for the sake of simplicity) that each have approximately the
same width as the transmitted pulse TXPulse (i.e., 100 ps), and a
set (i.e., 4) of the range cells can be arranged into a
digitization window 6006, having a total width of (for example)
about 400 ps. The digitization window 6006 allows for the reflected
and received radiation to be digitized. Additionally, the setup
period 6010 following the scan range 6004 can be used as setup time
for analog transmission.
[0089] In FIG. 32, the structure and operation of digitization
window 6006 can be seen in greater detail. As described above, the
digitization window 6006 is generally comprised of a set of range
cells; in this example, there are four range cells 7004-1 to 7004-4
in window 6006. Each of the range cells 7004-1 to 7004-4 can then
be subdivided into sampling instants (i.e., 7006). Again, in this
example, there are four sampling instants per range cell 7004-1 to
7004-4 (with a total of 16). Since each sampling instant (i.e.,
7006) is generally associated with a branch (i.e., 5002-1), it can
be assumed for this example that there are four transceivers (i.e.,
206-1 to 206-4), four switched capacitor banks (i.e., 4002-1 to
4002-4) with four branches each (i.e., 5002-1 to 5002-4), sixteen
branch sample signals (i.e., SAMPLE1 to SAMPLE16), and sixteen dump
branch signals (i.e., SAMPLE1 to SAMPLE16). Additionally, the
sampling instants (i.e., 608) can, for example, be separated from
one another by 25 ps.
[0090] During digitization window 6006, averaging of the baseband
signals BBI and BBQ is performed. The branch sample signals SAMPLE1
to SAMPLE16 (for the example of FIG. 32) are asserted on each
successive sampling instant (i.e., 6008) within digitization window
6006 so as to actuate sample switches S1-1 to S1-4 and S5-1 to S5-4
for each branch 4002-1 to 4002-4. These branch sample signals
SAMPLE1 to SAMPLE16, in this example, are asserted for
substantially the same duration as each of sub-range cell or
sampling period (i.e., time between sampling instants which can be
about 25 ps). This process is then repeated over a predetermined
number (i.e., 16) of transmitted pulses TXPulse (generally in
consecutive cycles) such that the each capacitor C1-1 to C1-4 for
each branch 4002-1 to 4002-4 measures the same sub-range cell or
same sampling period during each of the repeated cycles. This
allows the capacitors C1-1 to C1-4 for each branch 4002-1 to 4002-4
to "average" its amplified baseband signal (i.e., BBI or BBQ) for
its sub-range cell or sampling period over the predetermined number
of cycles. Following the completion of the predetermined number of
cycles, the ADC clock signal ADCCLK[L] (which is generally
synchronized with the sample signal SAMPLECLK[L]) can be asserted
so as to actuate output switches S2-1 to S2-4 and S6-1 to S6-4 for
each branch 4002-1 to 4002-4 in order so that ADCs 3008-1 and
3008-2 can readout and digitize the averaged voltages from each of
capacitors C1-1 to C1-4 for each branch 4002-1 to 4002-4. Once the
ADCs 3008-1 and 3008-2 readout the averaged voltages from each of
capacitors C1-1 to C1-4 for each branch 4002-1 to 4002-4, the
branch clear signals CLR1 to CLR16 are asserted so as to actuate
clear switches S3-1 to S3-46 and S4-1 to S4-4 for each branch
4002-1 to 4002-4 to discharge capacitors C1-1 to C1-4 and C2-1 to
C2-4 for each branch 4002-1 to 4002-4.
[0091] Having thus described the present invention by reference to
certain of its preferred embodiments, it is noted that the
embodiments disclosed are illustrative rather than limiting in
nature and that a wide range of variations, modifications, changes,
and substitutions are contemplated in the foregoing disclosure and,
in some instances, some features of the present invention may be
employed without a corresponding use of the other features.
Accordingly, it is appropriate that the appended claims be
construed broadly and in a manner consistent with the scope of the
invention.
* * * * *