U.S. patent application number 12/421623 was filed with the patent office on 2012-09-20 for remote detection of electronic devices.
This patent application is currently assigned to Los Alamos National Security , LLC. Invention is credited to Clifford M. Fortgang, David C. Guenther, Stephen L. Judd.
Application Number | 20120236905 12/421623 |
Document ID | / |
Family ID | 46828423 |
Filed Date | 2012-09-20 |
United States Patent
Application |
20120236905 |
Kind Code |
A1 |
Judd; Stephen L. ; et
al. |
September 20, 2012 |
REMOTE DETECTION OF ELECTRONIC DEVICES
Abstract
An apparatus and method for detecting solid-state electronic
devices are described. Non-linear junction detection techniques are
combined with spread-spectrum encoding and cross correlation to
increase the range and sensitivity of the non-linear junction
detection and to permit the determination of the distances of the
detected electronics. Nonlinear elements are detected by
transmitting a signal at a chosen frequency and detecting higher
harmonic signals that are returned from responding devices.
Inventors: |
Judd; Stephen L.; (Los
Alamos, NM) ; Fortgang; Clifford M.; (Los Alamos,
NM) ; Guenther; David C.; (Los Alamos, NM) |
Assignee: |
Los Alamos National Security ,
LLC
Los Alamos
NM
|
Family ID: |
46828423 |
Appl. No.: |
12/421623 |
Filed: |
April 9, 2009 |
Related U.S. Patent Documents
|
|
|
|
|
|
Application
Number |
Filing Date |
Patent Number |
|
|
61123697 |
Apr 9, 2008 |
|
|
|
Current U.S.
Class: |
375/130 ;
375/E1.001; 455/73 |
Current CPC
Class: |
G01S 13/18 20130101;
G01S 13/887 20130101; G01S 13/284 20130101; H04B 1/707
20130101 |
Class at
Publication: |
375/130 ; 455/73;
375/E01.001 |
International
Class: |
H04B 1/38 20060101
H04B001/38; H04B 1/69 20110101 H04B001/69 |
Goverment Interests
STATEMENT REGARDING FEDERAL RIGHTS
[0002] This invention was made with government support under
Contract No. DE-AC52-06NA25396 awarded by the U.S. Department of
Energy. The government has certain rights in the invention.
Claims
1. An apparatus for remotely locating solid-state electronics,
comprising in combination: an RF generator for generating a chosen
frequency RF signal; a signal splitter for dividing the RF signal
into a transmitted portion and a reference portion; a modulator for
encoding a chosen code onto the transmitted portion of the RF
signal; a controller for generating the chosen code, and for
directing the code to said modulator; a power amplifier for
amplifying the encoded RF signal; an antenna for transmitting the
amplified, encoded RF signal; an antenna for receiving a similarly
encoded second harmonic frequency of the encoded RF signal; a
demodulator for receiving the second harmonic frequency and the
chosen code delayed by a selected time interval from said
controller, and for removing the modulation from the second
harmonic frequency; a frequency doubler for doubling the frequency
of the reference portion of the RF signal; a mixer for comparing
the doubled frequency of the reference portion of the RF signal
with the demodulated second harmonic frequency, and for generating
a DC signal if the doubled frequency of the reference portion of
the RF signal is correlated with the demodulated second harmonic
frequency; and a computer for receiving the DC signal, for
directing said controller and for calculating distance between
located solid-state electronics and said transmitting antenna.
2. The apparatus of claim 1, wherein said modulator and said
demodulator comprise quadrature phase modulators.
3. The apparatus of claim 2, wherein the chosen code is imparted to
the transmitted portion of the RF signal by binary phase shift
keying spread-spectrum modulation.
4. The apparatus of claim 3, wherein the code is chosen from
M-sequences, Gold codes, Kasami sequences, and q-ary codes.
5. The apparatus of claim 3, wherein the binary phase shift keying
spread-spectrum modulation has a modulation frequency of less than
about 100 MHz.
6. The apparatus of claim 1, wherein said chosen code is imparted
to the transmitted portion of the RF signal by frequency shift
keying spread-spectrum modulation.
7. The apparatus of claim 1 wherein said controller comprises a
field programmable gate array.
8. The apparatus of claim 1, wherein said transmitting antenna is a
directional antenna.
9. The apparatus of claim 1, wherein said receiving antenna is a
directional antenna.
10. The apparatus of claim 1, wherein said transmitting antenna and
said receiving antenna are disposed on a movable vehicle.
11. The apparatus of claim 1, wherein said transmitting antenna and
said receiving antenna are disposed on an elevated platform.
12. The apparatus of claim 1, wherein said chosen frequency is
between about 870 MHz and about 920 MHz.
13. The apparatus of claim 1, further comprising at least one RF
filter for reducing interference of harmonics of the chosen
frequency generated in said apparatus.
14. The apparatus of claim 1, wherein the DC signal is averaged
over a selected number of cycles of said apparatus.
15. The apparatus of claim 1, further comprising a low-noise
amplifier for amplifying the encoded second harmonic frequency of
the encoded RF signal received by said receiving antenna.
16. A method for remotely locating solid-state electronics,
comprising the steps of: generating a chosen frequency RF signal;
dividing the RF signal into a transmitted portion and a reference
portion; encoding a chosen code onto the transmitted portion of the
RF signal; amplifying the encoded RF signal; transmitting the
amplified, encoded RF signal; receiving a similarly encoded second
harmonic frequency of the encoded RF signal; removing the encoding
from the second harmonic frequency; doubling the frequency of the
reference portion of the RF signal; comparing the doubled frequency
of the reference portion of the RF signal with the second harmonic
frequency for which the encoding has been removed; generating a DC
signal if the doubled frequency of the reference portion of the RF
signal is correlated with the second harmonic frequency for which
the encoding has been removed; and calculating a distance between
located solid-state electronics and the location of said step of
transmitting the amplified, encoded RF signal.
17. The method of claim 16, wherein the code in said step of
encoding the chosen code is imparted to the transmitted portion of
the RF signal by binary phase shift keying spread-spectrum
modulation.
18. The method of claim 17, wherein the code is chosen from
M-sequences, Gold codes, Kasami sequences, and q-ary codes.
19. The method of claim 17, wherein the binary phase shift keying
spread-spectrum modulation has a modulation frequency of less than
about 100 MHz.
20. The method of claim 16, wherein the code in said step of
encoding the chosen code is imparted to the transmitted portion of
the RF signal by frequency shift keying spread-spectrum
modulation.
21. The method of claim 16, wherein said step of transmitting the
amplified, encoded RF signal is performed using a directional
transmitting antenna.
22. The method of claim 16, wherein said step of receiving an
encoded first harmonic frequency of the encoded RF signal is
performed using a directional receiving antenna.
23. The method of claim 21, further comprising the step of
disposing the directional transmitting antenna on a movable
vehicle.
24. The method of claim 22, further comprising the step of
disposing the directional receiving antenna on a movable
vehicle.
25. The method of claim 21, further comprising the step of
disposing the directional transmitting antenna on an elevated
platform.
26. The method of claim 22, further comprising the step of
disposing the directional receiving antenna on an elevated
platform.
27. The method of claim 16, wherein the chosen frequency is between
about 870 MHz and about 920 MHz.
28. The method of claim 16, further comprising the step of
filtering for reducing interference of generated harmonics of the
chosen frequency.
29. The method of claim 16, further comprising the step of
averaging the DC signal over a selected number of cycles of said
method.
30. The method of claim 16, further comprising the step amplifying
the similarly encoded second harmonic frequency of the encoded RF
signal.
Description
RELATED CASES
[0001] The present application claims the benefit of provisional
patent application Ser. No. 61/123,697 entitled "Device for
Detecting Electronics at Long Ranges" by Stephen L. Judd et al.
filed on Apr. 9, 2008, which provisional application is hereby
incorporated by reference herein for all that it discloses and
teaches.
FIELD OF THE INVENTION
[0003] The present invention relates generally to detecting
electronics utilizing solid-state junctions and, more particularly,
to increasing the range and sensitivity of non-linear junction
detection technology and providing range resolution thereto.
BACKGROUND OF THE INVENTION
[0004] Non-Linear Junction Detection (NLJD) is a well-known
technique for detecting electronics that utilize semiconductor
(solid-state) junctions. The current state of the art for finding
hidden electronics such as electronic eavesdropping devices using
this technology has a maximum range of about 2 m, and more
typically between 6 in. and 12 in. A bare diode may be viewed as a
dipole antenna having a nonlinear junction separating the two
antenna elements. The response of a nonlinear junction to an
applied voltage follows the IV curve for the junction, and may be
described by I=I.sub.0 (e.sup.qV/kT-1) where q is the electronic
charge, k is Boltzman's constant, T is the temperature of the
device in Kelvin, V is the applied voltage, and I is the current
flowing through the junction. The first two terms in the expansion
of this expression are: I=I.sub.0 (qV/kT+1/2(qV/kT).sup.2), the
second term being responsible for generating the second-harmonic
(doubled) frequency which is determinative of the RF radiation from
the sought electronics. It is this frequency which is detected as
an indicator of the presence a nonlinear or semiconductor junction
associated with electronics. In the presence of a RF field, the
voltage, V, is determined by the applied field (the transmitted
power). When it exceeds the bias voltage, current I flows through
the device and may be re-radiated. Because the current is a
nonlinear function of the applied voltage, the re-radiated energy
contains harmonics of the fundamental applied RF frequency. In its
simplest form, then, a nonlinear junction detector irradiates an
area using frequency f, and detects returning electromagnetic
radiation at frequency 2f (and possibly 3f, etc.).
[0005] Electronic devices typically contain multiple nonlinear
junctions linked by wires or traces to other components. Therefore,
energy may couple into and out of the device through multiple
paths; moreover, the path(s) into a device may be different than
the path(s) out of the device. It is to be noted that powering a
device may alter its coupling characteristics (that is, biasing a
diode of interest will place a signal farther up the IV curve).
[0006] A popular commercial device is the ORION (See, e.g.,
http://www.tscm.com/orion.html.). The ORION is effective, but has a
range of only about 12 in. Simply increasing the transmitted power
to several Watts with the hope of increasing the range for locating
targets on the ground at several tens of meters has been found to
be ineffective because of false positives (self-detection), low
sensitivity, and severe attenuation of RF propagation along the
ground.
[0007] Spread spectrum techniques are commonly used in
communications, as they provide high sensitivity for low power
requirements. Examples include cell phones (Code Division Multiple
Access (CDMA)), and GPS (the latter uses a 50 W transmitter 20,000
km away). Pseudo-random encoding at the transmitter and
cross-correlation at the receiver is used to detect and locate
extremely weak signals, even in a noisy RF environment (this is how
dozens of cell phones can work in close proximity without
interfering with one another). Multiple techniques exist, including
phase shift keying (PSK), frequency shift keying (FSK), amplitude
shift keying (ASK), and the like.
SUMMARY OF THE INVENTION
[0008] Accordingly, it is an object of the present invention to
provide an apparatus and method for increasing the sensitivity of
non-linear junction detection measurements.
[0009] Another object of the invention is to provide an apparatus
and method for increasing the range of non-linear junction
detection measurements.
[0010] Still another object of the invention is to provide an
apparatus and method for generating range information for
non-linear junction detection measurements.
[0011] Yet another object of the invention is to provide an
apparatus and method for discriminating among targets for
non-linear junction detection measurements.
[0012] Additional non-limiting objects, advantages and novel
features of the invention will be set forth in part in the
description which follows, and in part will become apparent to
those skilled in the art upon examination of the following or may
be learned by practice of the invention. The objects and advantages
of the invention may be realized and attained by means of the
instrumentalities and combinations particularly pointed out in the
appended claims.
[0013] To achieve the foregoing and other objects, and in
accordance with the purposes of the present invention as embodied
and broadly described herein, the apparatus for remotely locating
solid-state electronics, hereof, includes in combination: an RF
generator for generating a chosen frequency RF signal; a signal
splitter for dividing the RF signal into a transmitted portion and
a reference portion; a modulator for encoding a chosen code onto
the transmitted portion of the RF signal; a controller for
generating the chosen code, and for directing the code to the
modulator; a power amplifier for amplifying the encoded RF signal;
an antenna for transmitting the amplified, encoded RF signal; an
antenna for receiving a similarly encoded second harmonic frequency
of the encoded RF signal; a demodulator for receiving the second
harmonic frequencies and the chosen code delayed by a selected time
interval from the controller, and for removing the modulation from
the second harmonic frequency; a frequency doubler for doubling the
frequency of the reference portion of the RF signal; a mixer for
comparing the doubled frequency of the reference portion of the RF
signal with the demodulated second harmonic frequency, and for
generating a DC signal if the doubled frequency of the reference
portion of the RF signal is correlated with the demodulated second
harmonic frequency; and a computer for receiving the DC signal, for
directing the controller and for calculating distance between
located solid-state electronics and the transmitting antenna.
[0014] In another aspect of the invention and in accordance with
its objects and purposes the method for remotely locating
solid-state electronics, hereof includes the steps of: generating a
chosen frequency RF signal; dividing the RF signal into a
transmitted portion and a reference portion; encoding a chosen code
onto the transmitted portion of the RF signal; amplifying the
encoded RF signal; transmitting the amplified, encoded RF signal;
receiving a similarly encoded second harmonic frequency of the
encoded RF signal; removing the encoding from the second harmonic
frequency; doubling the frequency of the reference portion of the
RF signal; comparing the doubled frequency of the reference portion
of the RF signal with the second harmonic frequency for which the
encoding has been removed; generating a DC signal if the doubled
frequency of the reference portion of the RF signal is correlated
with the second harmonic frequency for which the encoding has been
removed; and calculating a distance between located solid-state
electronics and the location of said step of transmitting the
amplified, encoded RF signal.
[0015] Benefits and advantages of the present invention include,
but are not limited to, sufficient performance enhancement that the
apparatus will perform at distances of 100 m on the ground.
Moreover, the manner in which detection is performed provides range
information, thereby greatly increasing the ability to discriminate
targets, as well as precisely locating targets.
BRIEF DESCRIPTION OF THE DRAWINGS
[0016] The accompanying drawings, which are incorporated in and
form a part of the specification, illustrate the embodiments of the
present invention and, together with the description, serve to
explain the principles of the invention. In the drawings:
[0017] FIG. 1 is a block diagram illustrating the components of an
embodiment of the apparatus of the present invention.
[0018] FIG. 2 is a schematic representation of an example of the
time relationship between chips transmitted in a modulated RF
signal and chips received from an electronics package upon which
the RF signal is incident.
[0019] FIG. 3 illustrates processed signals received using the
embodiment of the present apparatus described in FIG. 1 hereof from
four electronic devices placed at different locations on the
ground.
DETAILED DESCRIPTION OF THE INVENTION
[0020] Briefly, embodiments of the present invention include an
apparatus and method for remotely locating solid-state electronics.
Non-linear junction detection techniques (NLJD) are combined with
spread-spectrum encoding and cross correlation to increase the
range and sensitivity of the NLJD and to permit the determination
of the distances of the detected electronics. Nonlinear elements
are detected by transmitting a signal at a chosen frequency and
detecting higher harmonic signals that are returned from responding
devices. Spread-spectrum techniques including a hardware correlator
increase sensitivity (decrease effective noise bandwidth). As will
be set forth in detail hereinbelow, the present invention provides
high sensitivity, wide dynamic range, and excellent clutter
rejection/discrimination.
[0021] Primary applications of the present invention include, but
are not limited to, detection of eavesdropping devices, and
standoff (remote) detection of unusual electronics. For
eavesdropping devices, the increased sensitivity of embodiments of
the present invention allows for detecting a greater variety of
devices, while the enhanced range allows for much more rapid
sweeping of rooms; and the range resolution assists in the precise
location of such devices. For electronics associated with
suspicious packages, the appreciable standoff (remote) capability
allows detection of such devices from safe distances, as well as
the ability to establish the distance of the electronics.
[0022] As mentioned hereinabove, known difficulties encountered in
nonlinear junction detection include: [0023] a. Weak return
signals: The conversion from the fundamental frequency which is
incident on the target to a second harmonic (doubled) frequency is
an inefficient and weak process and contributes to the short range
of commercial NLJ Detectors. Propagation losses add to this problem
since near the ground, RF propagates effectively as 1/r.sup.4
instead of 1/r.sup.2, giving rise to a substantial loss: for
two-way propagation along the ground this implies that the
difference between 1 m and 100 m is a factor of 10.sup.-16. [0024]
b. Detection in the presence of large signals: The transmitter may
have a second harmonic component, though small, that masks the
second harmonic being sought from a target; the transmitter
fundamental may also generate a second harmonic signal in the NLJD
electronics. Further, the transmitted power may enter the receiver
along with any return signals. A large dynamic range permits the
NLJD to detect both weak signals from distant targets
simultaneously with strong signals from nearby targets. [0025] c.
High false-positive rate (rusted pipes are one commonly cited
example). Unlike noise, which is random, "clutter" refers to
deterministic, unwanted in-band signals, for example, from other
transmitters such as nearby cell phones, or self-generated
harmonics. Clutter leads to false positives in the detector,
masking true positives.
[0026] These problems, individually and in combination increase the
difficulty of obtaining the desired measurements, and nonlinear
junction detection has generally been considered to be a
short-range technique. The present invention combines a transceiver
architecture, transmitting at frequency f and receiving at
frequency 2f, with modulating of the outgoing signal (using a
pseudo-random spread spectrum code), and matching
(cross-correlating) the return modulation at the receiver, as may
be understood by referring to FIG. 1 hereof.
[0027] A spread-spectrum modulation code made up of chips having a
chosen length is used to modulate the transmitted frequency. Each
chip may be a -1 or +1. The code length may be varied. However,
using longer code lengths may reduce, the effective noise bandwidth
of the apparatus to provide increased sensitivity. Therefore, the
code is essentially a series of +1's and -1's which appear to be
random. An example of a 31-bit M-sequence is: -1 -1 -1 -1 1 -1 1 -1
1 1 1 -1 1 1 -1 -1 -1 1 1 1 1 1 -1 -1 1 1 -1 1 -1 -1 1. Note the
sequence is random and the sum of the bits is +1, which is a
property of all M-sequences regardless of their length. The
pseudo-random codes for the present invention were chosen to be
M-sequences which may be generated in hardware using a Galois
Linear Feedback Shift Register which is well known. The
spread-spectrum modulation imparted to the RF signal is known as
Binary Phase Shift Keying (BPSK). A +1 corresponds to a 0.degree.
phase shift and a -1 corresponds to a 90.degree. phase shift. When
the transmitted signal illuminates a semiconductor junction
(electronics) the reflected (or scattered signal) is doubled in
frequency, as discussed hereinabove, and the phase shift is
similarly doubled. Therefore, a +1 still represents 0.degree., but
the -1 now represents 180.degree.. A 180.degree. phase shift means
the signal is inverted which is equivalent to multiplying it by -1.
Thus, each received chip may be represented by either A
sin(2.pi.2f.sub.xmit t) if the chip is a +1 or by -A sin (2.pi.2
f.sub.xmit t) if the chip is a -1, where 2f.sub.xmit is the
frequency of the received signal.
[0028] A time delay between when the coded signal is transmitted,
and the time the return signal is received depends on the distance
the detected electronics reflecting the signal are from the
transmitter. A sample relationship between chips returning to the
receiver and being transmitted is shown schematically in FIG. 2,
hereof, where the RF signal, 32a, having twice the transmitted
frequency of encoded RF signal, 14, is scattered from device, 66a,
and the RF signal, 32b, having twice the transmitted frequency of
encoded RF signal 14 is scattered from device, 66b.
[0029] The received signal is correlated with the transmitted
signal in the following manner. A reference signal having the same
pseudo-random encoding as the modulated transmitted signal is used
to demodulate the received signal. The modulated transmitted signal
is repeatedly generated and transmitted. The modulated signal
consists of a chosen number of chips, and when all the chips are
transmitted the sequence begins again and continues to repeat. The
reference signal (which is the chip sequence) is delayed (shifted)
by a chosen number of chips each time the transmitted signal is
generated. The apparatus multiplies the received signal by the
reference chips and adds the results. If the delay is incorrect
such that there is no correlation among the chips, on average, 50%
of the time a result of +1 is observed (from either 1.times.1 or
-1.times.1), and 50% of the time a result of -1 is observed (from
either 1.times.-1 or -1.times.1). When the observed results are
added, an approximate sum of zero results (actually -1 because of
the odd number of chips). If the reference signal is appropriately
delayed, all the -1's and +1's line up such that every one of the
multiplications gives a result of +1 (either -1.times.-1 or
1.times.1). When the results are added, the final result is
significantly larger than when the time delay is not correct.
[0030] In summary, when the correct delay is applied, the two
signals are correlated and the resulting sum may be large, and a
target is detected. The result of such correlation is equal to the
number of chips, longer codes generating larger sums, whereas the
result for uncorrelated signals is -1, independent of the code
length. The ratio of the correlated result (the code length in
chips) and the uncorrelated result (-1) is commonly called the
processing gain.
[0031] The delay applied to the reference modulated signal and
which gave rise to the correlation may be used to calculate the
distance of the target. Since the reference signal is delayed
one-half chip at a time, the resolution in the distance to the
target is determined by the distance associated with a single,
half-chip. The ability to determine the range reduces the false
positives which are common in conventional NLJDs. The principal
sources of false positives were discussed hereinabove as being the
second harmonic leakage from the transmitting antenna, and the
detection of second harmonics by the apparatus from within the
apparatus. Since these sources of false positive are located at the
apparatus, they generally are detected with a delay of zero or one
chip in the reference signal. If the minimum time delay is set to 2
or more chips, then the false positives in the immediate vicinity
of the apparatus cannot be correlated and therefore cannot be
"detected." Further, the correlation and time delay associated with
the correlation process interrogates only a chosen range of
distances at a time; therefore, large signals from nearby targets
(which also may be false positives), do not mask weak signals from
real, more distant targets. Thus, the ranging capability allows the
present apparatus to separate weak signals from distant targets
from strong signals of nearby targets, and the modulation permits
the discrimination between a signal of interest and clutter
generated elsewhere. With a transmitted power of 20 W, the
embodiment of the present apparatus has consistently measured -155
dBm signals (with some averaging), post-antenna, representing a 198
dB spread. The invention has detected electronics at a distance of
greater than 100 m which is well beyond the range of the existing
technology. An example of an embodiment of the invention
simultaneously detecting and locating four electronic devices
dispersed on the ground is shown in FIG. 3 hereof.
[0032] The present apparatus may be mounted on a vehicle with
directional transmitting and receiving antennas having fields of
view tailored by both the antenna beam width and/or physical
rotation of the antenna(s). The antennas would be directed to sweep
both along and off to the sides of a road in the direction of
forward of motion of the vehicle. Another application of the
invention may be to mount the system on an elevated platform
disposed on a post, at least 10 m high. The post might either be
further mounted on a slowly moving platform or attached to the
ground as a fixed post, and would continuously monitor a specific
area. The apparatus might also be mounted on a moving platform with
an extendable mast.
[0033] Reference will now be made in detail to the present
illustrative embodiments of the invention, examples of which are
illustrated in the accompanying drawings. It will be appreciated
that in the development of actual embodiments, numerous
implementation-specific decisions will perforce be made to achieve
a developer's specific goals that will vary from one implementation
to another. Moreover, it will be appreciated that such a
development effort may be complex, but would nevertheless be a
routine undertaking for those having ordinary skill in the art
having the benefit of the present disclosure. In the FIGURES,
similar structure will be identified using identical reference
characters.
[0034] Returning now to FIG. 1, a block diagram of an embodiment of
the present apparatus is illustrated. Apparatus, 10, includes an RF
signal generator, 12, the output of which is divided into RF
carrier signals, 14, and, 16, by splitter, 18. Chosen pseudo-random
code (for example, a phase-shift-keying code), 20, generated by
Field Programmable Gate Array (FPGA), 22, is impressed on signal 14
by quadrature phase modulator, 24, after which signal 14 is
amplified by power amplifier, 25, and transmitted by transmitting
antenna, 26 (XMT), which may be a directional antenna. Return
signal, 32, is received by antenna, 34 (RCV), which may be a
directional antenna, directed into quadrature phase demodulator,
36, driven by an identical, but time-delayed pseudo-random code,
38, from FPGA 22. Output, 39, of demodulator 36 is directed to
mixer, 40. When the time-delayed code from FPGA 22 matches
(correlates with) the modulation on received signal 32, output 39
of the demodulator is a sine wave having frequency which is doubled
from that of the original carrier signal 14 and is equal to
frequency, 30, exiting frequency doubler, 28. If the time-delayed
code from FPGA 22 does not match the round trip delay of the
received signal, then output 39 of demodulator 36 still has the
modulation on it, indicating that the signals are uncorrelated.
Mixer 40 down converts demodulator output 39. For the correlated
case mixer output, 42, and, 44, is a low-frequency DC pulse, while
for the uncorrelated case mixer output 42 and 44 contains the same
high-frequency modulation that exited modulator 36. For the
correlated case, the low-frequency pulse is passed without
attenuation though low-pass filters, 46, and 48, and on to A-D
converters, 50, and, 52. For the uncorrelated case, the
high-frequency modulation is attenuated by low-pass filters 46 and
48 before being digitized by A-D converters 50 and 52. Some
additional signal processing (integration) is performed by FPGA
22.
[0035] The output from FPGA is sampled at appropriate times by PC,
54, which both controls FPGA 22 and extracts and displays the
target information from return signal 32 through interface, 56,
which permits the cooperation of FPGA 22 therewith. Low-pass
filters, 58a-58d, band-pass filters, 60a-60d, amplifiers, 62a-62f,
and low-noise amplifier, 64, may be added to improve the apparatus
response and reduce the generation of second harmonics.
[0036] As stated hereinabove, the correlation process may be
mathematically expressed as multiplication and integration of the
two signals; it may also be implemented by multiplication and
filtering. The correlation process may be implemented in hardware
(before digitization) or digitally (after digitization at the
fundamental frequency). The latter method is more efficient (higher
throughput), but has lower sensitivity and dynamic range which
limits the maximum chipping rate. The present apparatus utilizes
the digital spread-spectrum technique, but makes an analog
measurement (I/Q phase, amplitude). This provides greatly increased
sensitivity through background rejection, long integration times,
and range resolution. Although phase shift keying (PSK) is
employed, frequency shift keying (FSK) may be used individually or
in combination therewith, and is supported by embodiments of the
invention.
[0037] Codes having perfect or near-perfect correlation properties
that may be used include M-sequences and derivatives such as Gold
codes, Kasami sequences, and q-ary codes. The transmitted code is
chosen such that, when doubled (or tripled, etc.) the returned
signal is an M-sequence, q-ary sequence, etc.
[0038] Again referring to FIG. 1 hereof the method of the present
invention may be understood as follows. A CW sine wave which may
have a frequency between about 870 MHz and about 920 MHz due to
bandwidth limitations is generated using a function generator. The
bandwidth is limited by the bandwidth of the commercially available
antennas and other RF components used in the apparatus. The signal
is then divided, thereby preserving the exact frequency of the
transmitted carrier signal for the receiver (mixer). The sine wave
is amplified and filtered (to eliminate 2.sup.nd harmonics that are
inherent to the amplification process), and directed to the
modulator. The modulator may be a digital (TTL) Quadrature-Phase
(0.degree., 90.degree., 180.degree., and 270.degree.) modulator,
having two digital inputs labeled I and Q (In-phase and Quadrature
phase, respectively). The truth table for the modulator is provided
in the TABLE.
TABLE-US-00001 TABLE I Q Output Phase 0 0 0.degree. 0 1 90.degree.
1 0 180.degree. 1 1 270.degree.
For the modulator employed, only 0.degree. and 90.degree. are used;
thus, the I input is always LOW, the Q input being either HIGH or
LOW. The state of the Q input is the M-SEQUENCE discussed
hereinabove. Therefore, a sine wave exits the modulator having its
phase modulated between 0.degree. and 90.degree. with the
pseudo-random M-sequence. The length of the M-sequence and the
frequency of the modulation or chip rate are chosen inputting these
parameters into the FPGA through the USB port. The length of the
M-sequence is typically 2.sup.10-1 (=1023) but may be chosen to be
between 2.sup.4-1 (=15) to 2.sup.14-1 (=16383). The modulation (or
chip) rate is also variable from as low as desired to about 100
MHz, but is typically 20 MHz. As explained hereinbelow, the error
bar for assigning a range to a detected target is inversely
proportional to the chip rate; that is, the higher the chip rate
the more accurately a target can be located; therefore, very low
chip rates are not useful.
[0039] The modulated signal exits the modulator, is again amplified
and filtered, amplified using a high-power amplifier and again
filtered, and transmitted. The modulated transmitted signal is
repeatedly sent. When the M-sequence is completed (in the example
hereinabove, 1023 chips), the process is repeated. Signals from
responding electronics as a result of the non-linear junction
phenomena are both doubled in frequency and phase compared to the
RF exiting the transmitter. Thus, for example, if the transmitted
signal is centered at 900 MHz with a 0.degree. and 90.degree. phase
shift modulation at 20 MHz, as an example, then the scattered
signal is centered at 1800 MHz with a 20 MHz phase shift modulation
of 0.degree. and 180.degree.. Note that the modulation rate of the
scattered signal does not double to 40 MHz. The received signal
(from a second antenna) is first amplified using a low noise
amplifier, and again with another amplifier before entering the
demodulator which has the same truth table as the modulator except
now the two states used are 0.degree. and 180.degree.. Referring to
the TABLE, the Q input is held LOW and the I input changes between
LOW and HIGH. The manner in which the I input to the demodulator
changes is again the M-sequence. The M-sequence that is directed to
the demodulator (I input) is the same as that which was directed
into the modulator (Q input), except that the M-sequence going to
the demodulator is delayed relative to that which was directed to
the modulator.
[0040] This delay has several consequences. The delay time
increment is equal to 1/2 a chip. For example, if the chip rate is
20 MHz, the delay increment of the M-sequence to the modulator is
25 ns (=0.51/(20 MHz)). An operator using the computer graphical
user interface, or GUI, through the USB interface instructs the
FPGA to set a minimum and maximum delay. The delay can be expressed
in one of three ways: (1) the number of chips; (2) time; or (3)
range to the target. Range is often the most useful to the user.
Range=ct.sub.delay/2, where c is the speed of light. When a target
is detected, the distance to the target is calculated from this
equation where t.sub.delay is the time delay that produced the
correlation. The minimum range is usually set to at least about 5 m
since electronics associated with the apparatus will be detected if
the minimum range is set too low. The maximum range is generally
less than about 100 m because the signals coming from targets
further than 100 m are often too weak to detect. Therefore, typical
minimum and maximum distances are between about 5 and about 100 m,
although these distances may be contracted or expanded depending on
the application intended.
[0041] The apparatus is designed to detect electronics in the
direction the antennas are pointed because high-gain antennas
having a relatively narrow beam are typically employed. The range
of detection is stepped through from the minimum to the maximum in
half-chip increments. A range increment is equal to
cT.sub.half-chip/2, the factor of 1/2 arising because the signal
makes a round trip. T.sub.half-chip=1/2.times. (1/chip rate);
therefore, as an example, if the chip rate is 20 MHz, then
T.sub.half-chip=25 ns and the range increment is 3.75 m. Typical
operating conditions might include: 1000 chips, 20 MHz chip rate,
minimum range=30 m, the maximum range is 100 m, and the range
increment is 3.75 meters. For 1000 chips in a M-sequence, it takes
1000.times.50 ns=50 .mu.s to interrogate a single range bin. There
are (100-30)/3.75=.about.19 bins to be interrogated which takes 50
.mu.s.times.19=950 .mu.s to scan from 30 m to 100 m. Because the
system is fast (typically 1 ms to scan the desired range) a user
may request an Averaging Mode where the entire range is scanned N
times and the average value for each range bin is displayed. This
is especially useful for weak signals, because averaging increases
the Signal-to-Noise ratio. Also, regarding range accuracy, as
explained hereinabove, the accuracy of locating a given target is
determined by the range increment which in turn is determined by
how fine the time delay is. The smaller the time-delay step the
higher the accuracy. A time-delay step of one-half chip at a time
is typical, but the time delay can be smaller if desired so that
the accuracy of locating a detected target is less than 1 m. The
finer the time delay, the higher the accuracy, but the time it
takes to scan a given range is increased because the number of
range bins is increased.
[0042] At the demodulator, the incoming signal is a pseudo-random
bi-phase (0.degree. and 180.degree.) modulated signal. The purpose
of the demodulator is to "strip off" the modulation if the signal
is coming from a target that is located at the range being
interrogated. When the signal is coming from the range being
interrogated, the M-sequence on the received signal will exactly
match the delayed M-sequence coming from the FPGA. For this
situation, a sine wave at exactly twice the frequency produced by
the signal generator exits the modulator. This is what is meant by
the signals being "correlated." If there is no target at the range
being interrogated, then the M-sequence on the received signal does
not match up with the delayed M-sequence from the FPGA, and a
doubled carrier signal that is still phase modulated with a
pseudo-random sequence exits the demodulator. For this situation,
the two signals are uncorrelated.
[0043] The mixer is an analog device (all 4 ports), whereas the
described modulator and demodulator are digital devices, and down
converts the signal from the modulator. Further, I and Q for the
mixer are outputs not inputs as for the modulator and demodulator.
One input is a sine wave at double the frequency which was used for
the transmitted carrier signal, and is generated by the doubler
labeled. If there is correlation from the demodulator, the
demodulator produces a sine wave having twice the transmitted
frequency. For the situation where there is correlation, two sine
waves enter the mixer. These two sine waves are at exactly the same
frequency, but do not necessarily have the same phase. For the
correlated situation a DC level from the I and Q ports exits the
mixer. The exact proportion of I and Q depends on the phase between
the two sine waves. Of importance is that there is a DC level at
either I or Q or both for as long as there is correlation. For the
example parameters provided hereinabove, it takes 50 .mu.s to
interrogate a single range bin, and there will be a step pulse of
constant amplitude for 50 .mu.s duration. The frequency content of
the pulse is .about.1 /code length=1/50 .mu.s=20 kHz. Note the
frequency content of the correlated signal exiting the mixer (20
kHz) is much lower than the modulation frequency (20 MHz). In fact,
it is lower by a factor equal to 1/number of chips (1000 for this
example). If there is no correlation, the signal emerging from the
mixer no longer has the 1800 MHz carrier because of the down
conversion, but still has the pseudo-random modulation on it. It is
a noisy pseudo-random signal having a bandwidth equal to the chip
rate. For the present example, it will appear to be random white
noise out to about 20 MHz.
[0044] The mixer output is filtered by a low pass filter which
passes the low-frequency correlated signal, but not the
high-frequency uncorrelated signal. If there is correlation, a
"pulse" having 50 .mu.s duration exits the low pass filter. If
there is no correlation, then a small quantity of pseudo-random
noise exits the low pass filter. This small amount of noise has an
average value of 0 volts. The I and Q signals out of the mixer are
amplified, digitized and directed into the FPGA. Typically, the
digitizer has a digitizing rate of about 1 MHz, but may be higher.
The FPGA is synchronized to the digitizer in order to identify
which samples are associated with which range bin. At 1 MHz
sampling and M-sequences that are 1000 chips and therefore 50 .mu.s
long (for a 20 MHz chip rate) there are therefore 50 samples/range
bin. If the signal is correlated, then there is a DC pulse exiting
from the mixer, through the low pass filter, and after
digitization, into the FPGA. The FPGA integrates (or adds) the
samples (for both I and Q separately) over the 50 .mu.s which
produces a result that is 50 times the amplitude of the DC pulse.
If the signal is not correlated, then there is random
high-frequency noise exiting the low pass filter that is being
digitized. The noise is sampled 50 times, has + and - values, and
the result after integration is close to zero, or at least much
smaller than the correlated value. This process is performed
separately for both I and Q, and for each range bin there is a SUM
I value and a SUM Q value. For each range bin the single number
that is passed to the user for display at the computer is Range Bin
Magnitude=SQRT [(SUM I).sup.2+(SUM Q).sup.2], and if there is a
detected signal from any particular range bin it appears as a peak
at that range bin. FIG. 3 is an example where 4 targets at 4 having
different range bins were detected.
[0045] The foregoing description of the invention has been
presented for purposes of illustration and description and is not
intended to be exhaustive or to limit the invention to the precise
form disclosed, and obviously many modifications and variations are
possible in light of the above teaching. The embodiments were
chosen and described in order to best explain the principles of the
invention and its practical application to thereby enable others
skilled in the art to best utilize the invention in various
embodiments and with various modifications as are suited to the
particular use contemplated. It is intended that the scope of the
invention be defined by the claims appended hereto.
* * * * *
References