U.S. patent application number 13/209607 was filed with the patent office on 2011-12-29 for quadratic amplitude control circuit for cosite interference cancellation.
This patent application is currently assigned to BAE Systems Information and Electronic Systems Integration Inc.. Invention is credited to Raymond J. Lackey.
Application Number | 20110319046 13/209607 |
Document ID | / |
Family ID | 42223271 |
Filed Date | 2011-12-29 |
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United States Patent
Application |
20110319046 |
Kind Code |
A1 |
Lackey; Raymond J. |
December 29, 2011 |
Quadratic Amplitude Control Circuit For Cosite Interference
Cancellation
Abstract
A quadratic amplitude matching system and associated method with
an associated tuning control system is provided for continuously
and automatically tuning a quadratic amplitude matching filter
(QAMF) to a band center of an interfering signal to provide
improved rejection of an interfering signal coupled from a
transmission antenna into a local receive antenna in the presence
of local multi-path, thereby providing improved interference
cancellation system performance. The matching control system is
provided as an element of an interference cancellation system.
Inventors: |
Lackey; Raymond J.;
(Bohemia, NY) |
Assignee: |
BAE Systems Information and
Electronic Systems Integration Inc.
Nashua
NH
|
Family ID: |
42223271 |
Appl. No.: |
13/209607 |
Filed: |
August 15, 2011 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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12315431 |
Dec 3, 2008 |
8023921 |
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13209607 |
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Current U.S.
Class: |
455/307 |
Current CPC
Class: |
H04B 1/525 20130101 |
Class at
Publication: |
455/307 |
International
Class: |
H04B 1/10 20060101
H04B001/10 |
Claims
1. A method for continuously and automatically tuning a quadratic
amplitude matching filter (QAMF) to an inserted signal to allow
tracking of the inserted signal to match a dynamically changing
quadratic amplitude distortion of the inserted signal for improved
interference cancellation system performance, the method
comprising: a) forming an imaginary broadband RF lobed filter
having a single quiescent null within a frequency band of interest;
b) dynamically adjusting a delay time (T) for tuning the single
quiescent null of the imaginary broadband RF lobed filter to
effectively block the inserted signal to be tracked; c) forming a
first narrowband RF lobed filter with a quiescent lobe peak
centered on a quiescent null of the inserted signal to be tracked,
wherein an output of the first narrowband RF lobed filter output
has a near-linear and flat amplitude shape to match a dynamically
changing quadratic amplitude distortion of the inserted signal to
be tracked; d) forming a second narrowband RF lobed filter with a
quiescent lobe peak centered on the quiescent null of the inserted
signal to be tracked, wherein an output of the second narrowband RF
lobed filter has a downward quadratic amplitude shape to match a
dynamically changing quadratic amplitude distortion of the inserted
signal to be tracked; e) forming an FIR filter with a quiescent
lobe peak centered on the quiescent null of the inserted signal to
be tracked, wherein an output of the FIR filter has an upward
quadratic amplitude shape to match a dynamically changing quadratic
amplitude distortion of the inserted signal to be tracked; f)
adjusting an in-line path delay of the first narrowband RF lobed
filter, the second narrowband RF lobed filter and the FIR filter to
have the same throughput delay; and g) adjusting combining weights
of the respective filter outputs of the first narrowband RF lobed
filter, the second narrowband RF lobed filter and the FIR filter to
implement a corrective quadratic amplitude shaping of the inserted
signal, thereby matching a dynamically changing quadratic amplitude
distortion of the inserted signal to be tracked.
2. The method of claim 1, wherein the respective outputs of the
first and second narrowband RF lobed filters and the FIR filter are
dependent upon the dynamically adjusted delay time (T) for tuning
the single quiescent null of the imaginary broadband RF lobed
filter.
3. The method of claim 1, wherein the first and second narrowband
RF lobed filters and the FIR filter are orthogonal to the imaginary
broadband RF lobed filter in a quiescent state.
4. The method of claim 1, wherein said step (g) of adjusting
combining weights of respective filter outputs of the first
narrowband RF lobed filter, the second narrowband RF lobed filter
and the FIR filter parallel filters is performed under control of
an external amplitude control signal.
5. The method of claim 1, wherein the first and second narrowband
RF lobed filters and the FIR filter are comprised of a plurality of
lobes formed within the frequency span of a single lobe of the
imaginary broadband RF lobed tuning filter.
6. The method of claim 5, wherein one of the plurality of lobes of
the narrowband RF lobed filter is peaked on the inserted signal to
be tracked.
7. The method of claim 6, wherein said one of said plurality of
lobes peaked on the inserted signal to be tracked is a quiescent
lobe of an amplitude sloped matching filter (ASMF) function.
8. The method of claim 6, wherein said one of said plurality of
lobes is peaked on the interfering signal to be tracked and is
centered at a null of the broadband RF lobed filter.
9. The method of claim 5, wherein one of the plurality of lobes of
the imaginary broadband RF lobed tuning filter is centered on the
inserted signal to be tracked has an amplitude shape to correct a
dynamically changing quadratic amplitude distortion of the inserted
signal to be tracked.
10. The method according to claim 1, wherein the broadband RF lobed
filter has a null to null bandwidth substantially twice the desired
tuning bandwidth.
11. The method according to claim 1, wherein said step (b) of
dynamically adjusting a delay time is performed via a control
process.
12. The method according to claim 11, wherein the control process
comprises: a) measuring an RF energy output from the broadband RF
filter; b) applying a delta increment to the delay time (T); c)
re-measuring the RF energy output from the broadband RF filter; d)
comparing the measured and the re-measured values of RF energy; and
e) determining if the applied delta increment to the delay time (T)
improves the tuning as measured by an energy output from the
broadband RF filter; f) if improved, updating the delay time (T) by
adding a positive delta increment as T=T+.DELTA.T; g) if
unimproved, updating the delta delay time (.DELTA.T) by changing
sign of delta delay time increment as .DELTA.T=-.DELTA.T.
13. The method of claim 1, wherein increasing the control voltage
causes the delay T to be increased resulting in a narrowing of the
filter lobes.
14. The method of claim 13, wherein narrowing the filter lobes
results in a downward frequency shift of the individual lobes and
nulls of the filter.
15. The method of claim 1, wherein decreasing the control voltage
causes the delay T to be decreased resulting in a widening of the
filter lobes.
16. The method of claim 15, wherein widening the filter lobes
results in an upward frequency shift of the individual lobes and
nulls of the filter.
17. An interference cancellation system, comprising: (A) an
adaptively tuned quadratic control (ATQC) module (200) for
providing a tuning procedure directed to a quadratic amplitude
matching filter (QAMF) (100) control function to the band of
interference and for externally controlling the QAMF control
function subsequent to said tuning, said adaptively tuned quadratic
control (ATQC) module (200) comprising: a) an inline variable lobe
filter structure (VLFS) (201) for providing controlled variable
time delay for generating a broadband RF tuning filter formed by a
delay T and a first narrowband inline control filter for providing
a near-flat quadratic control filter lobe formed by a first delay
(T) and a second delay (2nT) yielding a total time delay of (2n+1)T
centered in the null of the broadband RF tuning filter and weighted
by an external amplitude control signal; b) a second narrowband
inline control filter as a slaved lobe filter structure-down (104)
for providing a down quadratic amplitude control filter lobe formed
by a delay of (2m+1)T centered in the null of the RF tuning filter
and weighted by an external amplitude control signal; c) a third
narrowband inline control filter as a slaved lobe filter
structure-up (105) for providing a up quadratic amplitude control
filter lobe formed by a simple FIR filter with inter-tap spacing
delay of (2o+1)T centered in the null of the tuning filter and
weighted by an external amplitude control signal; d) an offline
time delay tuning control (TDTC) element (202) for receiving signal
samples output from the inline variable lobe filter structure
(VLFS) 201 to control a first variable time delay element (T) 205
of the inline variable lobe filter structure (VLFS) 201 to provide
said controlled variable time delay to form said broadband RF
tuning filter and a second variable time delay element (2nT) (208)
of the inline variable lobe filter structure (VLFS) 201 to provide
said controlled variable time delay to generate said slope control
filter lobe. (B) an adaptive control loop (6) for adjusting a
complex weighting of the delayed coupled signal (57) to maximally
cancel a propagated reference signal.
18. The interference cancellation system of claim 17, wherein the
inline variable lobe filter structure (VLFS) 201 comprises: a) said
first variable time delay line (T) (205) for providing broadband
tuning of an imaginary tuning filter lobe; and b) a second variable
time delay line (2nT) (208) for providing more narrowband tuning of
the imaginary tuning filter lobe relative to said first delay
element; wherein said first variable time delay element (205) and
second variable time delay element (2nT) (208) yield a total time
delay of (2n+1)T centered in the null of the tuning filter and
skewed by an external slope control signal.
19. The interference cancellation system of claim 17, wherein the
second variable time delay line (2nT) 208 is an integer multiple of
said first time delay line (T) 205.
20. The interference cancellation system of claim 17, wherein the
adaptive control loop (6) comprises: a reference port (9) for
receiving the an antenna signal (30); an auxiliary port (8) for
receiving a delayed and matched coupled signal (57); a complex
correlator (66) for generating error correlation signal (72) an
integrator (67) to smooth transients on the error correlation
signal (72) to form the adaptive weight control signals (73); a
complex phase and amplitude weighting device (68) having a first
input and a second input, said first input receiving said delayed
and matched coupled signal (57), said second input receiving a
complex adaptive weight control signal (73) to weight the delayed
and matched coupled signal (57) to produce a weighted delayed
coupled signal (65); and forming a weighted delayed coupled signal
(65) as a mirror image of a propagated reference signal, contained
in antenna signal 30; a summing junction (70) having a first and
second input, said first input for receiving said weighted delayed
coupled signal (65) output from said complex phase and amplitude
weighting device (68), said second input for receiving the antenna
reference signal (71) to yield a protected output signal (58).
21. The interference cancellation system of claim 17 wherein said
propagated reference signal comprises at least a transmission
signal (40) propagated through an uncontrolled path from a first
antenna (2) and received at a second antenna (4).
22. The interference cancellation system of claim 20, wherein
forming the weighted delayed coupled signal (65) as a mirror image
of the transmitted reference signal indicates that it is equal in
amplitude and 180.degree. out of phase with a portion of the
transmitted signal (40) in the antenna reference signal (71).
23. The interference cancellation system of claim 17, wherein said
antenna signal (30) includes both said propagated reference signal
and at least one other signal.
24. The interference cancellation system of claim 23, wherein the
at least one other signal is a desired signal anticipated by a
protected receiver (25).
25. A method for implementing a first-order quadratic correction to
the amplitude of an input signal across its band by the use of a
quadratic amplitude matching filter (QAMF), the method comprising:
a) dividing an input signal into three parallel signal paths; b)
dynamically adjusting a delay time (T) in a first signal path from
among said three parallel signal paths for tuning a first
narrowband RF lobed filter with one of its quiescent lobes peaked
on an interfering signal to be tracked; c) forming a second more
narrowband RF lobed filter in a second signal path from among said
three signal paths dependent upon the delay time (T) wherein one of
the quiescent lobes of the second narrowband RF filter is peaked on
the interfering signal to be tracked; d) forming a simple FIR
filter in the third branch dependent upon the delay time, (T),
having a filter configuration in the form of an upward quadratic
shape centered on the interfering signal to be tracked, e) matching
each of the first, second and third signal paths to have a uniform
path delay dependent upon the delay time (T); f) weighting each of
the first, second and third signal paths according to an external
control function, and g) combining the first, second and third
signal paths into a single output to allow facilitate the
implementation of a first-order quadratic amplitude distortion of
the input signal via the QAMF to match the delayed coupled signal
to that of the propagation path for improved interference
cancellation of the inserted signal in an interference cancellation
system.
26. The method of claim 25, wherein the first narrowband RF lobed
filter is sufficiently broad enough in bandwidth to implement a
first path characterized by a near-linear and flat amplitude
shape.
27. The method of claim 25, wherein the second more narrowband RF
lobed filter is controlled to be more narrow than the first
narrowband RF lobed filter to implement said second signal path for
downward quadratic amplitude adjustment of the inserted signal.
28. The method of claim 25, wherein the FIR filter upward quadratic
area is centered on the interfering signal to be tracked to
implement said third signal path for upward quadratic amplitude
adjustment of the inserted signal.
Description
FIELD OF THE INVENTION
[0001] The invention relates to the field of radio communication
and, in particular, to the reduction of interference in signals
coupled from a transmission antenna into a local receive antenna in
the presence of a local multipath.
DESCRIPTION OF THE RELATED ART
[0002] Unwanted (i.e., interfering) signals manifest themselves in
several ways. Interference can cause a reduction in the sensitivity
of a receiver (receiver desensitization), masking of a desired
signal, tracking of an undesired interfering signal and loss of the
desired signal, and processing of the unwanted interfering signal
instead of the desired signal. Each of these manifestations of
interference limits the communication capabilities of the radio
system afflicted by this problem. The effects of interference can
be some combination of the absence of usable output from a
receiver, false signals from a receiver, and malfunction of a
device which is operated by the receiver. During emergency
situations, the loss and corruption of the desired signal can be
critical.
[0003] Unwanted signal interference is generally caused by
modulation of signals provided to the receiver by the carrier
waves, or by the wideband noise, generated by collocated
transmitters. Unwanted signal interference also occurs when
frequency-hopping transmitters are transmitting signals at
frequencies that are substantially close to the frequency of the
desired receiver signal (i.e., co-channel operation). Unwanted
signal interference can also be caused by "pseudo white-noise"
generated by transmitters over a wide band of frequencies on either
side of the transmitter's operating frequency. It is often found in
collocated transceiver systems that this "pseudo white-noise"
reaches unacceptable levels within the operating band of adjacent
receivers. Unwanted signal interference is also attributed to
signals (i.e., spurious emissions) generated by transmitters at odd
harmonics of the fundamental frequency of the transmitter output
signal. This is caused by the non-linear transfer characteristics
of amplifiers in the transmitter chain, or by passive
inter-modulation at the transmitter or receiver antenna
connectors.
[0004] In order to substantially reduce and eliminate the undesired
interfering signals while maintaining the spatial benefits afforded
by proximately locating transceivers, especially frequency-hopping
transceivers, several signal processing techniques have been
proposed. These techniques include agile filtering, agile filtering
with multicoupling and interference cancellation.
[0005] When the signal noise and spurious sidebands generated by
the interfering transmitter are strong, broadband, and scenario
dependant, standard interference cancellation is inadequate.
Changes in the scenario surrounding the platform may vary the
coupling between the transmitter and the protected receiver and
thus require adjustment of system parameters in an adaptive
process.
[0006] Interference cancellation involves sampling the transmitter
output signal in order to eliminate from the received signal, any
interfering signal having a frequency proximate to the receiver
carrier frequency. In co-site environments, a collocated source
usually interferes with the receiver due to the finite isolation
between transmit and receive antennas. This interference in a
co-site environment is a combination of several factors,
desensitization caused by one or more nearby high-power transmitter
carriers and wideband moderate to low-power interference components
associated with those carriers. These interference components are
received by the collocated radio and degrade system operation. The
nearby high-power transmitter carrier signals could simply exist as
a part of the platform signal environment. Further, the interfering
signals may be classified as either cosite or remote interferers. A
cosite interferer is physically collocated with the receiver on a
platform permitting a physical circuit connection from the
interference generator to the receiver. A remote interferer is
located far enough from the receiver to preclude a physical circuit
connection.
[0007] A typical Interference cancellation system utilizes a
correlation-based adaptive controller using feedback derived after
the cancellation process. The system takes a sample of an
interference signal and adjusts the magnitude and phase such that
the result is equal in amplitude and 180.degree. out of phase with
the interference signal at the input of the receiver. The vector
sum of the two signals will cancel, leaving only the signal of
interest. In practice, however, the two signals are not identical,
due to unwanted distortion in the reference path, as well as
differences in signal path lengths and non-ideal components in the
Tx/Rx signal paths. Cancellation performance is a function of
amplitude and phase match between the interference signal and the
sampled signal. Transmission path distortions include time delay,
magnitude amplitude and phase distortion, linear amplitude and
phase distortion, and quadratic distortion, correction of each
adding a level of performance enhancement but also adding to system
complexity and difficulty in implementing the corrections.
[0008] To suppress a wideband interference signal, the performance
of a cancellation system is directly proportional to the match
between the sampled transmission cancellation signal and the
receive path interference signal across the signal bandwidth. The
interference cancellation system (ICS) compensates for minor
corrections and component drift by controlling a complex weight
that implements flat phase and amplitude controls in the adaptive
control loop (ACL) to correct the magnitude amplitude and phase
errors between the two. The receive path interference signal
provided to the ICS is disrupted by signal distortions in time of
arrival, linear and non-linear (i.e., quadratic) amplitude, and
linear and non-linear phase. The sampled transmission cancellation
signal must be adjusted to match this distorted receive path signal
as closely as possible to achieve complete nulling of the received
interference signal. The present disclosure addresses these
concerns by focusing on minimizing mismatch errors caused by first
order non-linear amplitude distortions.
[0009] As is well known, cosite interference cancellation requires
amplitude slope matching across the signal bandwidth to achieve a
deep null across the band. U.S. Pat. No. 6,693,971 "Wideband
co-site interference reduction apparatus" (Kowalski) issued on Feb.
17, 2004 and assigned to BAE Systems Information and Electronic
Systems Integration Inc. (Greenlawn, N.Y.), incorporated by
reference herein in its entirety, discloses a method of
implementing a near-linear correction of the amplitude slope using
an amplitude slope-matching filter. However, a drawback of the
system and method of Kowalski is that it also imparts a quadratic
shape to the amplitude across the band.
[0010] Similarly, the propagation path can also impart a quadratic
amplitude modulation across the band that will be time varying with
the changing environment. Together, these two distortions limit the
nulling performance of the cosite interference cancellation
system.
[0011] A need therefore exists for a system and method for
continuously adjusting the quadratic amplitude of a coupled
co-sited transmitter signal before subtracting it from the
propagated and received signal with multipath dispersive
distortions to achieve required nulling. Such a system would also
have to be tuned with the transmitter frequency and adjust to
changes in the propagation path distortion.
SUMMARY OF THE INVENTION
[0012] It is therefore an object of the present disclosure to
provide a method and apparatus for reducing the effects of
interference between collocated transceivers.
[0013] It is an object of the present disclosure to provide a
method and apparatus in which proximately located transceivers can
simultaneously transmit and receive independent signals without
substantially affecting the quality of a desired signal
reception.
[0014] It is another object of the present disclosure to eliminate
the effects of interference between collocated transceivers
utilizing interference cancellation.
[0015] It is a more particular object of the present disclosure to
provide a method and apparatus for providing a quadratic amplitude
matching capability to an interference cancellation system by
implementing a quadratic amplitude matching filter (QAMF).
[0016] It is a more particular object of the present disclosure to
provide a method and apparatus for automatically tuning a bank of
lobed filters of the QAMF such that the signal tracked is near the
center of each lobing structure to generate quadratic shaping
structures in the region of a tracked signal spectrum.
[0017] It is yet another object of the present disclosure to
provide a method and apparatus for tuning this quadratic amplitude
matching filter (QAMF) over as large of a band as possible without
external interface or control.
[0018] The present disclosure provides a quadratic amplitude
matching filter (QAMF) architecture and a tuning control system as
an element of an interference cancellation system and associated
method for continuously and automatically tuning a quadratic
amplitude matching filter (QAMF) to a band center of an inserted
coupled transmitted signal for improved interference cancellation
system performance and adjusting to match propagation path
distortion. More particularly, the QAMF system provides improved
rejection of an interfering signal coupled from a transmission
antenna into a local receive antenna in the presence of local
multipath.
[0019] The tuning control system and associated method of the
present disclosure provide improved signal rejection over other
possible tuning approaches by continuously tuning (adjusting) a
lobed filter of the tuning control system so that the QAMF has a
quiescent flat shape in the region of the tracked signal
spectrum.
[0020] In accordance with one embodiment of the present disclosure
a tuning control system is provided for reducing interference in
signals coupled from a transmission antenna into a local receive
antenna in the presence of a local multi-path. The tuning control
system interfaces with a time-delay based lobed filter architecture
including delay means for forming synchronously locked lobed
filters for both a tuning filter for tracking to a predominant
interfering signal inserted at an input port and a bank of filters
capable of applying a first order quadratic amplitude matching to
effect the amplitude shape desired for distortion matching. The
system further includes control means, associated with the delay
means, for tuning the QAMF to track the inserted signal and center
it at the center of the filter, thereby eliminating the need to
interface the control means with the transmitter.
[0021] In accordance with one embodiment of the present disclosure,
a method is provided for implementing a first-order quadratic
correction to the amplitude of an input signal across its band by
the use of a quadratic amplitude matching filter (QAMF), the method
comprising: dividing an input signal into three parallel branches,
dynamically adjusting a delay time (T) in the first branch for
tuning a first narrowband RF lobed filter with one of its quiescent
lobes peaked on an interfering signal to be tracked, wherein the
first narrowband RF lobed filter is broad enough in bandwidth to
implement a first path with near-linear and flat amplitude shape,
forming a second, more narrowband RF lobed filter in the second
branch dependent upon the delay time (T) wherein one of the
quiescent lobes of the more narrowband RF filter is peaked on the
interfering signal to be tracked but controlled to be more narrow
to implement a second path for downward quadratic amplitude
adjustment of the inserted signal, forming a simple FIR filter in
the third branch dependent upon the delay time, (T), wherein its
central form is shaped to form an upward quadratic shape centered
on the interfering signal to be tracked, and wherein the FIR filter
upward quadratic area is centered on the interfering signal to be
tracked to implement a third path for upward quadratic amplitude
adjustment of the inserted signal, matching each of the first,
second and third branches to have a uniform path delay dependent
upon the delay time (T), weighting each of the first, second and
third branches according to an external control function, and
combining the first, second and third paths into a single output to
allow the QAMF to implement a first-order quadratic amplitude
distortion of the input coupled transmitted signal to match the
delayed coupled signal to that of the propagation path for improved
interference cancellation of the inserted signal in an interference
cancellation system
[0022] Also, in accordance with one embodiment of the present
disclosure, a method is provided for continuously and automatically
tuning a quadratic amplitude matching filter (QAMF) to a band
center for improved interference cancellation system performance,
the method comprising: a) forming a broadband RF lobed filter
having a single quiescent null within a frequency band of interest;
b) dynamically adjusting a delay time (T) for tuning the single
quiescent null of the broadband RF lobed filter to effectively
block the inserted signal to be tracked; c) forming a first
narrowband RF lobed filter with a quiescent lobe peak centered on a
quiescent null of the inserted signal to be tracked, wherein an
output of the first narrowband RF lobed filter output has a
near-linear and flat amplitude shape to match a dynamically
changing quadratic amplitude distortion of the inserted signal to
be tracked; d) forming a second narrowband RF lobed filter with a
quiescent lobe peak centered on the quiescent null of the inserted
signal to be tracked, wherein an output of the second narrowband RF
lobed filter has a downward quadratic amplitude shape to match a
dynamically changing quadratic amplitude distortion of the inserted
signal to be tracked; e) forming an FIR filter with a quiescent
lobe peak centered on the quiescent null of the inserted signal to
be tracked, wherein an output of the FIR filter has an upward
quadratic amplitude shape to match a dynamically changing quadratic
amplitude distortion of the inserted signal to be tracked; f)
adjusting an in-line path delay of the first narrowband RF lobed
filter, the second narrowband RF lobed filter and the FIR filter to
have the same throughput delay; and g) adjusting combining weights
of the respective filter outputs of the first narrowband RF lobed
filter, the second narrowband RF lobed filter and the FIR filter to
implement a corrective quadratic amplitude shaping of the inserted
signal, thereby matching a dynamically changing quadratic amplitude
distortion of the inserted signal to be tracked.
[0023] According to one aspect of the method described above,
dynamic adjustment of the time-delay element considers both
direction and degree in dependence upon the most recent nulling
filter output comparison result.
[0024] In different embodiments, the system may be implemented in
discreet components or alternatively as a MMIC. Time delays can be
implemented as either a switched delay or a continuously variable
delay through an analog control voltage.
BRIEF DESCRIPTION OF THE DRAWINGS
[0025] These and other objects, features and advantages of the
invention will be apparent from a consideration of the following
Detailed Description Of The Invention considered in conjunction
with the drawing Figures, in which:
[0026] FIG. 1 illustrates the general block diagram of an improved
cosite interference cancellation system, according to one
embodiment.
[0027] FIG. 2a is a circuit diagram of the general structure of a
lobed filter for use in an improved cosite interference
cancellation system, according to one embodiment.
[0028] FIG. 2b illustrates the general structure of the lobed
filter, for use in an improved cosite interference cancellation
system, according to one embodiment.
[0029] FIGS. 3a-b are exemplary output waveforms of a lobed filter
for illustrating that subtraction, rather than summation, of the
output of two signal paths of the lobed filter forms an orthogonal
filter of the same repetitive bandwidth as the output of the lobed
filter from a basic delay (T).
[0030] FIG. 4 are exemplary resultant output waveforms of a lobed
filter formed from the summed outputs of a lobed filter having
signal paths characterized by delays which are odd integer
multiples of a basic delay (T), the resultant output waveforms
illustrating that a lobe of the summed outputs is always aligned
with a lobe of a lobed filter formed from the basic delay (T).
[0031] FIG. 5a-c illustrate three different exemplary tuning
scenarios of a generated wideband RF lobed filter that is
orthogonal to the lobe of an imaginary (unformed) wideband RF
tuning filter.
[0032] FIG. 6 is a block diagram of a five-tap finite impulse
response (FIR) filter whose pass-band amplitude repeats in the
frequency domain, according to one embodiment.
[0033] FIG. 7a is a plot comprising four curves, a first curve
representing the output of a lobed broadband tuning filter, the
second curve representing a lobed narrowband tuning filter, a third
curve representing a narrower lobed narrowband filter and a fourth
curve representing an FIR filter.
[0034] FIGS. 7b & 7c are plots representing the four curves of
FIG. 7a in increasingly expanded form.
[0035] FIG. 8 is a circuit diagram of a quadratic amplitude
matching filter (QAMF) structure where the tuning control comes
from an external controller, according to one embodiment.
[0036] FIG. 9 is a circuit diagram of an adaptively tuned quadratic
control (ATQC) structure where the time delay tuning control
circuit has been integrated, according to one embodiment.
[0037] FIG. 10 is a circuit diagram of the adaptively tuned
quadratic control (ATQC) of FIG. 9 incorporated into an
interference cancellation system to improve the cancellation of a
local transmitter signal that is received in the receive antenna
with a time varying modulation due to changes in local
multipath.
[0038] FIG. 11 illustrates one embodiment of an improved cosite
interference cancellation system 20 for elimination of interfering
signals between three or more co-located transceivers
[0039] FIG. 12 illustrates an improved cosite interference
cancellation system for elimination of interfering signals between
a single co-located transceiver and a plurality of receivers to be
protected.
DETAILED DESCRIPTION OF THE INVENTION
[0040] In the following discussion, numerous specific details are
set forth to provide a thorough understanding of the present
invention. However, those skilled in the art will appreciate that
the present invention may be practiced without such specific
details. In other instances, well-known elements have been
illustrated in schematic or block diagram form in order not to
obscure the present invention in unnecessary detail. Additionally,
for the most part, details concerning network communications,
electromagnetic signaling techniques, and the like, have been
omitted inasmuch as such details are not considered necessary to
obtain a complete understanding of the present invention and are
considered to be within the understanding of persons of ordinary
skill in the relevant art.
[0041] The present description illustrates the principles of the
present disclosure. It will thus be appreciated that those skilled
in the art will be able to devise various arrangements that,
although not explicitly described or shown herein, embody the
principles of the disclosure and are included within its spirit and
scope.
[0042] All examples and conditional language recited herein are
intended for pedagogical purposes to aid the reader in
understanding the principles of the disclosure and the concepts
contributed by the inventor to furthering the art, and are to be
construed as being without limitation to such specifically recited
examples and conditions.
[0043] Moreover, all statements herein reciting principles,
aspects, and embodiments of the disclosure, as well as specific
examples thereof, are intended to encompass both structural and
functional equivalents thereof. Additionally, it is intended that
such equivalents include both currently known equivalents as well
as equivalents developed in the future, i.e., any elements
developed that perform the same function, regardless of
structure.
[0044] The functions of the various elements shown in the figures
may be provided through the use of dedicated hardware as well as
hardware capable of executing software in association with
appropriate software. When provided by a processor, the functions
may be provided by a single dedicated processor, by a single shared
processor, or by a plurality of individual processors, some of
which may be shared. Moreover, explicit use of the term "processor"
or "controller" should not be construed to refer exclusively to
hardware capable of executing software, and may implicitly include,
without limitation, digital signal processor ("DSP") hardware, read
only memory ("ROM") for storing software, random access memory
("RAM"), and nonvolatile storage.
[0045] Other hardware, conventional and/or custom, may also be
included. Similarly, any switches shown in the figures are
conceptual only. Their function may be carried out through the
operation of program logic, through dedicated logic, through the
interaction of program control and dedicated logic, or even
manually, the particular technique being selectable by the
implementer as more specifically understood from the context.
Overview
[0046] The present disclosure is directed to a tuning control
system and associated method for continuously and automatically
tuning a quadratic amplitude matching filter (QAMF) to a band
center of a reference input signal for improved interference
cancellation system performance in a cosite interference
cancellation system. In some embodiments, the tuning process may be
performed off-line to preclude the interruption of processing,
during an operation stage, with intermediate or final control
signals being transferred to an inline structure to implement the
same control.
[0047] The present disclosure provides an automated system and
method that performs dynamic adjustment of the delay time, tuning a
quadratic amplitude matching filter (QAMF) that centers the QAMF
filter on the frequency of its reference input signal as a
pre-requisite to adjusting the QAMF filter for quadratic amplitude
control (i.e., amplitude matching). More particularly, the present
disclosure provides a novel quadratic amplitude-matching filter
(QAMF) to implement dynamic, real-time correction to the quadratic
amplitude mismatch. The present disclosure further provides a time
delay tuning control 202, coupled to the quadratic amplitude
matching filter (QAMF) to provide frequency tuning to the quadratic
amplitude matching filter (QAMF) without the need for an external
tuning control signal (e.g., a tuning control signal from the
transmitter as practiced in the prior art). It should be
understood, however, that quadratic amplitude control is required,
as a further processing step beyond performing dynamic, real-time
tuning. As is well known, Quadratic amplitude control is performed
to adjust the quadratic amplitude-matching filter (QAMF) to the
proper weights to match the quadratic amplitude distortion of a
sampled transmission signal to that of the propagation path.
[0048] Referring now to the drawings, FIG. 1 is a circuit diagram
for illustrating an improved cosite cancellation circuit 20 for
eliminating interfering signals between radio transmitter 21, as an
element of transceiver 1, and receiver 25, as an element of
transceiver 5, where system dynamics cause changes in the coupling
between transmit antenna 2 and receive antenna 4, co-located on a
platform, according to one embodiment.
[0049] It should be understood that each of the transceivers 1, 5
function independent of the other such that they alternate in being
viewed as either the interfering transmitter or protected receiver
depending upon the specific needs of the user. However, the system
description will only address a single functional aspect for ease
of explanation. The transceivers 1, 5 can operate at any RF
frequency including, for example, in the high frequency (HF), very
high frequency (VHF) and ultra-high frequency (UHF) spectrums.
[0050] The improved cosite cancellation circuit 20 for the
elimination of interfering signals between radio transceivers 1, 5,
is adapted to be coupled to transceiver 5, in the illustrative
embodiment, or other type of device, known or envisioned, capable
of receiving electronic signals. The transceiver 1 operating in the
transmission mode produces electronic signals for transmission via
antenna 2 of transceiver 1. Substantially contemporaneously to this
signal transmission, other electronic signals are received by
antenna 4 and provided to at least transceiver 5 operating in the
receiving mode. As is known to happen, in addition to the signals
intended to be received by antenna 4, the co-located transmitter 21
also generates noise and distortion signals which interfere with
the electronic signals received by the antenna 4 that are to be
provided to a receiver 5.
[0051] In order to substantially eliminate the effect of the
interfering signals generated from transceiver 1, the novel
cancellation circuit 20 is electrically coupled to transmission
signal 40. In a preferred form of the present invention, a
directional coupler 7 is operatively coupled to the output port of
transmitter 21. The cancellation circuit 20 receives a sample of
the filtered transmission signal corresponding to the transmitter 1
to which it is coupled.
Operation
[0052] In operation, transmitter 21 transmits RF transmission
signal 40 through antenna 2 which couples spatially 3 either
directly or through a multipath environment into a second antenna 4
connected to a receiver 25 on the same platform as interfering
transmitter 21. This coupled energy interferes with the reception
in the receiver 25 of its desired reception of a distant
transmission. The interfering transmitter 21 thus becomes a
collocated source of interference. It is desired to protect the
receiver 25 from the interfering transmitter 21. The addition of a
simple Interference Cancellation System (ICS) consisting of only a
coupled adaptive control loop (ACL) 6 can reduce this interference
to a limited extent by sampling the transmission signal 7 and
feeding it into the auxiliary port 8 of the ACL 6 while antenna
signal 30, including both the interfering propagated reference
signal and the desired signal, is fed into the reference port 9 of
the ACL 6.
[0053] In an environment clear of reflective obstacles (e.g., no
multi-path sources present), the spatially coupled signal 3 from
antenna 2 to antenna 4 would be received unchanged except for the
propagation delay and the quadratic amplitude matching filter
(QAMF) 100 would not be required. However, in a typical multi-path
laden environment, the spatially coupled signal 3 is distorted
across the band in a number of ways, one of them being an
undesirable quadratic amplitude distortion across the band of
interest which is constant in a stable environment but varies with
a changing multipath environment of a platform in motion.
Static v. Dynamic Environments
[0054] In a static environment, the cable delay, T.sub.D 277,
between sample point 7 and point 8, the input to ACL 6, is ideally
adjusted to be the typical coupling delay through space from source
antenna 2 to receive antenna 4. This delay, T.sub.D 277, is
implemented to include the delay of QAMF 100 and any other in-line
delays. The next level of correction is the amplitude slope
matching 278 which is ideally adjusted to match the amplitude slope
distortion through space from source antenna 2 to receive antenna
4. These corrections will change with time in a dynamic environment
but are not the subject of this disclosure. In a dynamic
environment, as environmental conditions change with time in an
unpredictable manner, a variable quadratic amplitude distortion can
be affected upon the propagated signal resulting in an undesirable
mismatch between the coupled transmission (i.e., the signal coupled
via path 7 to 8) and the propagated transmission (i.e., the signal
coupled via path 2-9) limiting the effectiveness of the applied
cancellation.
[0055] To correct a dynamically changing quadratic amplitude
mismatch between the afore-mentioned signals, the present
disclosure provides, in one aspect, a quadratic amplitude matching
filter (QAMF) 100, generally shown in FIG. 1 and illustrated in
more detail in FIG. 7, to implement a dynamic correction to the
amplitude slope mismatch between the delayed coupled signal 57 and
antenna signal 30, including both the interfering propagated
reference signal and the desired signal.
[0056] To successfully track and match the distortion introduced by
the dispersive propagated interfering propagated reference signal,
contained in antenna signal 30, interference cancellation circuit
20 must first continuously and automatically tune a quadratic
amplitude matching filter (QAMF) 100, to the reference input
interfering transmitted signal 40 band center. This continuous and
automatic tuning process comprises a key feature of the invention.
In a preferred embodiment, the tuning process is continuously and
automatically performed by a local tuning control system (i.e.,
time delay tuning control 202), as a quiescent starting point for
performing subsequent operations such as quadratic amplitude
adjustment.
[0057] It should be understood that the present disclosure is
primarily directed to: (1) an architecture that implements a
quadratic amplitude correction, (2) the tuning of the quadratic
amplitude matching filter (QAMF) 100 as a pre-requisite to
performing quadratic amplitude adjustment, and (3) quadratic
amplitude adjustment under control of an adaptive amplitude
controller 225 (see FIG. 1a). It should be understood that Adaptive
amplitude control adjustment 225 uses standard control algorithms
and processes, which are well known in the art, and peripheral to
the teachings of the present disclosure. However, Adaptive
amplitude control adjustment is briefly discussed as follows.
Adaptive Amplitude Adjustment
[0058] As is well known, RF spectral amplitude adjustment may be
implemented by forming a filter of desired shape. Filters of
differing shape can be formed in parallel and a controller can
select the best match for the application but there is often no way
of knowing a priori which filter will best match the application.
Another way of selecting a filter output, or even generating a new
filter from a composite of a number of filters, is to weight and
sum each of the filter outputs in a variable weighting structure. A
controller is provided which has a feedback mechanism such that it
can change the weighting and summation network weight values and
then evaluate the change. Adaptive amplitude control 225 implements
this process by monitoring the protected output 58 of ACL 6 (See
FIG. 1a) while dithering control lines that adjust the weights of
the quadratic amplitude matching filter (QAMF) under a sequence
determined by its algorithm and loop feedback.
[0059] Referring now to FIG. 2a, a circuit structure is shown for
forming a tunable variable lobed filter 250, according to one
embodiment. In this embodiment, the tunable variable lobed filter
250 is implemented using a power divider 252, a variable delay 254
and a summing junction 256. The tunable variable lobed filter 250
is tunable by changing the variable delay value [T] 254. Tuning the
variable delay value [T] 254 causes expansion and contraction of
each lobe from zero and thus a shift of every lobe, beyond the
first, up or down in frequency.
[0060] FIG. 2b is an alternative circuit structure 260 of the
tunable variable lobed filter 250 implemented with a difference
hybrid 258 as a substitute for the summing junction 256. This
creates a functionally similar tunable variable lobed filter 250 as
described above but has orthogonal lobes to the structure of FIG.
2a, providing an important mathematical relationship to be used in
control of the tuning process, as discussed immediately below and
also further below with reference to FIG. 3.
[0061] The inventor has recognized two important mathematical
relationships that together allow tuning over a large bandwidth and
control of a more narrowband filter to provide the desired
amplitude shaping effect. The first important mathematical
relationship relates to the orthogonal nature of the sine and
cosine function of two RF filters simultaneously formed from the
same power divider 252 and time delay structure when combined in
either a sum or difference port of the tunable variable lobed
filter 250, as briefly discussed above. The first recognized
mathematical relationship allows the use of a null at one frequency
in a sine filter to align with the lobe of the cosine filter, or
vice versa, and can be used as a sensitive tuning control, as
illustrated in FIG. 3, and described below.
[0062] The second important mathematical relationship is the
recognition that two RF filters, one tuned with time delay T and
the other tuned with a further time delay (2n+1)T, where n is an
integer, will always have lobes co-aligned at the center of the
wider band lobe. It is noted that the relationship is one of the
further time delay being an odd multiple of a basic delay T. The
implications of such a relationship are described in more detail
further below with respect to FIG. 4.
[0063] Referring now to FIGS. 3a-3b, there is shown an output of a
lobed filter, such as, for example, the tunable variable lobed
filter 250 of FIGS. 2a and 2b. The output is represented as curve
390 in FIG. 3a (and further illustrated in expanded form in FIG.
3b).
[0064] Referring to FIG. 3a, the output 390 of tunable variable
lobed filter 250 is shown as a magnitude (cosine) function of the
delay difference in the two paths, i.e., path A and path B, shown
in FIG. 2a. The lobed filter amplitude of output curve 390 of FIG.
3a repeats at a regular spacing of BW.sub.n equal to (2T).sup.-1.
As stated above, in an alternate embodiment, a difference hybrid
258 (See FIG. 2b) can be substituted for the summing junction 256
(See FIG. 2a) of the tunable variable lobed filter 250 FIG. 2a. In
such an embodiment, the output 390 of the tunable variable lobed
filter 250 follows a magnitude (sine) function, represented as
curve 391 in FIG. 3a. Thus, a time delay can be selected to have
the tunable variable lobed filter 250, 260 extend beyond a band of
interest and a corresponding orthogonal filter will have a null
within the tuning bandwidth. For example, by extending tunable
variable lobed filter 250 of FIG. 2a beyond a band of interest it
will have a null 390 within the tuning band of interest. As a
further example, by extending tunable variable lobed filter 260 of
FIG. 2b beyond a band of interest, it will have a null 391 within
the tuning band of interest.
[0065] It should be appreciated that the null to null bandwidth,
BW.sub.n of the lobed tunable variable lobed filter 250 is
inversely proportional to the time delay, T 254, as shown in FIGS.
2a and 2b. Therefore, an increase in the time delay T 254 reduces
the bandwidth BW.sub.n of the tunable variable lobed filter 250.
Further, by changing the time delay to be an odd multiple of a
basic delay T, for example, by (2n+1)T, where n an integer, the
original tunable variable lobed filter 250 is effectively split
into (2n+1) lobes. As this always results in an odd number of
lobes, one lobe 402, necessarily is always centered with the tuning
lobe 404 of a broadband tuning filter, as shown in FIG. 4. This
single centered lobe 402 becomes useful in the quadratic amplitude
matching filter (QAMF) structure 100 (See FIGS. 7 and 8) to be
weighted by the adaptive amplitude control 225 controller to shape
the coupled signal to match the propagated interfering propagated
reference signal, contained in antenna signal 30 in an interference
cancellation system. The value of n used to effectively split the
output of the filter into 2n+1 lobes can be adjusted to achieve the
desired flatness at quiescent with minimal propagation path
distortion as required for slaved lobe filter structure-flat 103
but can also be adjusted to the value m to provide the required
quadratic down shaping required for slaved lobe filter
structure-down 104. It is also contemplated to use the values of n
and m as variables for finer tuning control in future envisioned
implementations of the improved cosite interference cancellation
system.
[0066] FIGS. 5a-5c illustrate, by way of example, plots of
different exemplary tuning scenarios to further illustrate the
concept of generating a lobed filter orthogonal to the lobe of a
broadband tuning filter. It should be understood that, in
accordance with invention principles, a tuning filter lobe of the
broadband tuning filter is not necessary for actual operation, and
is not necessarily formed in actual operation. It will therefore be
referred to hereafter as a so-called imaginary tuning filter lobe.
It should be understood, however that the quadratic amplitude
matching filter (QAMF) will track the center of the so-called
imaginary tuning filter lobe by use of the orthogonal null formed
off-line in the timing delay tuning control (TDTC) 202, as shown in
FIGS. 1, 9 and 10. Herein, inline refers to an action or process
that generates an immediate change, upon signals passing through,
at the output of the circuit where offline refers to action or
processes that may use samples of signals passing through but do
not impact the signals passing through until a result is reached
and a change is made to the inline processes.
[0067] Each of the plots of FIGS. 5a-5c illustrates a common
insertion signal 511 to be tracked. The insertion signal represents
the sample of transmitted signal 40 to be matched to an undesirable
multipath signal received in antenna signal 30 to be cancelled by
the improved cosite cancellation circuit 20 of FIG. 1.
[0068] Referring first to FIG. 5a, four output filter curves are
shown 511, 512 N.sub.O, 513 N.sub.L, 514 N.sub.H. Output filter
curves 512 N.sub.O, 513 N.sub.L and 514 N.sub.H represent three
different filters tuned with a so-called imaginary tuning filter
lobe but orthogonal to the imaginary tuning filter lobe such that
nulls of the orthogonal filter are aligned with the peak of a lobe
of the original filter formed by the same delay, T. A first filter
output curve 512 N.sub.O represents the null portion of a lobed
filter, N.sub.O, formed by current value of delay T, orthogonal to
the tuning filter tuned on frequency with the imaginary tuning
filter by using the same delay T used to form the tuning filter.
Using the same delay used to form both the first filter output
curve 512 N.sub.O and the imaginary tuning filter, results in a
null of the first filter output curve 512 N.sub.O aligned with the
imaginary tuning lobe of the tuning filter, as shown in FIG. 3.
[0069] A second filter output curve 513 N.sub.L represents the null
portion of a lobed filter, N.sub.L, formed by delay T+.DELTA.T, an
incremental step of delay time T 254 of the circuit of FIG. 2 tuned
low in frequency with a path delay difference of T+.DELTA.T and
results in a null below, or lower than the current center frequency
of the imaginary tuning lobe of the broadband lobed filter.
[0070] A third filter output curve 514 N.sub.H represents the null
of the lobed filter, N.sub.H, is tuned high in frequency with a
path delay difference of T-.DELTA.T and results in a null above, or
higher than, the current center frequency of the imaginary tuning
lobe of the broadband lobed filter.
[0071] With continued reference to FIG. 5a, there is shown the
condition in which the filter, N.sub.O, is centered at a frequency
that is below the frequency of the insertion signal 511. In this
case, the filter N.sub.H allows more of the inserted signal energy
of the inserted signal 511 through, than the filter N.sub.L thus
providing feedback to the interference cancellation system to move
the tuning filter higher in frequency by decreasing the delay,
T.
[0072] FIG. 5b illustrates the case in which the filter, N.sub.O,
is centered at a frequency that is above the frequency of the
insertion signal 511. In this case, the filter N.sub.L allows more
of the inserted signal energy of the inserted signal 511 through,
than the filter N.sub.H thus providing feedback to the interference
cancellation system to move the tuning filter lower in frequency by
increasing the delay, T.
[0073] FIG. 5c illustrates the case where conditions when the
filter, N.sub.O, is centered on the inserted signal. In this case,
the low and high filters, N.sub.H and N.sub.L, will pass equal
amounts of the inserted signal energy, thus providing no feedback
to change frequency by changing the delay, T. This state represents
a point of stability in tuning such that, as the null of the
orthogonal filter is aligned with the inserted signal and thus
aligned with the peak of the center of the lobe of the imaginary
tuning filter, the inserted signal is thus aligned with the peak of
the quadratic amplitude matching filter (QAMF) center.
[0074] It should be understood that the direction of the null
shifts as a function of the time delay introduced by the
interference cancellation system is inherent to lobed filters which
are comprised of a plurality of nulls originating at zero Hz and
repeating at a regular spacing of (2T).sup.-1. Thus an increase in
delay T reduces the effective BW.sub.n, thereby compressing the
lobing and shifts the current null to the left, i.e., lower in
frequency.
[0075] Referring again to FIG. 5a, the center null 512 N.sub.O is
representative of a filter output null which is orthogonal to the
corresponding filter output formed by the summation of the output
of filter signal paths with path delay differences formed by the
inline delay T.
[0076] The left null 513 N.sub.L represents the null of a filter
output having a path delay T+.DELTA.t, the output exhibiting a
slightly more narrow lobed structure than the output of a filter
signal path having a path delay T, and thus the repetitive lobing
shifts to the left, lower in frequency, moving the null below the
nominal location using delay T.
[0077] The right null 514 N.sub.H represents the null of a filter
output having a path delay T-.DELTA.t, the output exhibiting a
slightly wider lobed structure than the output of a filter signal
path having a path delay T, and thus the repetitive lobing shifts
to the right, higher in frequency, moving the null above the
nominal location using delay T.
[0078] It should be appreciated that these two filter output curves
513 N.sub.L, 514 N.sub.H advantageously allow different amounts of
the incident signal energy to pass through them. In this manner,
measurement of the energy from the respective filter outputs
provides information on a corrective direction in frequency of the
tuning lobe orthogonal filter required for proper tuning.
[0079] With continued reference to FIG. 5a, this figure further
illustrates a set of undesirable image nulls 515. It is appreciated
that these undesirable image nulls 515 are a limitation to the
tuning bandwidth of the tuning control system. They arise by using
too large of a value of delay T, resulting in an excess of narrow
lobes for tuning. It therefore follows that it is desirable to have
as large a tuning bandwidth as possible to preclude the creation of
these image nulls. It is preferred that tuning to the low edge of
the frequency tuning band cannot allow image nulls to approach the
high band limit for inserted signal, or vice versa, or the system
may shift lobes of the tuning filter upon a jump in transmitted
signal frequency, and cause significant change in subsequent filter
bandwidths and thus shaping amplitude factors. The narrowband
filter cannot be used for tuning because of this limitation. This
shows the importance of the recognition of the lobe alignment for
filters formed by T and (2n+1)T delays so that the tuning filter
lobe can be very broad for a broad tuning range but still be used
to focus a much more narrow lobe for quadratic amplitude matching
filter (QAMF) function.
[0080] As stated above, a primary objective of the tuning control
system of the present disclosure is to continuously and
automatically tune a quadratic amplitude matching filter (QAMF) to
an interferer band center as a quiescent starting point for
performing quadratic amplitude control adjustment. While it is
understood that amplitude control adjustment is not central to the
teachings of the present disclosure, it is understood that it is
implemented by controlling the weights of the tuned quadratic
amplitude matching filter (QAMF), tuned in accordance with
invention principles.
[0081] The tunable variable lobed filter 250 structure cannot
provide the necessary shaping required for the slaved lobe filter
structure-up 105 of the quadratic amplitude matching filter (QAMF).
A finite impulse response filter with structure shown in FIG. 6 can
provide the shape required while maintaining linear phase.
[0082] Referring now to FIGS. 7a-7c, there is shown, by way of
example, a plot of four curves. For ease of explanation, each
signal has been offset in level for clarity and each successive
plot is an expansion of the center area of the previous plot, as
indicated by the common centerline.
[0083] The first curve 610 is representative of an imaginary
broadband tuning filter formed by a delay interval T, corresponding
to an off-line lobed tuning filter with an orthogonal null to allow
it to track an incoming signal of interest. The second curve 612 is
representative of a lobed filter tracking the tuning filter with a
delay interval (2n+1)T, where n is some integer multiplier of T. In
this case, the lobed filter tracks the off-line broadband tuning
filter null, as is true of the first curve 610 formed with a delay
T, however, in the present case, the filter is more narrow in
bandwidth although still nearly flat in the region of the bandwidth
of signal of interest, as generated by Slaved lobe filter
structure-flat 103 and output at 112 (See FIGS. 1 and 8).
[0084] The third curve 614 is representative of the lobed filter
tracking the tuning filter with a multiplier value of m=28 in the
present example such that it tracks the imaginary broadband tuning
filter lobe center and the slaved lobe filter structure-flat 103
(See FIGS. 1 and 8). This lobe structure is even more narrow in
bandwidth than the lobe structure generated by slaved lobe filter
structure-flat 103 and output at 112 (See FIGS. 1 and 8). The
presently described lobe structure has amplitude shaped as a down
quadratic in the region of the bandwidth of signal of interest,
i.e., the "signal bandwidth" region, as generated by slaved lobe
filter structure-down 104 and output at 118.
[0085] The fourth curve is representative of a more complex FIR
filter (e.g., formed using 5 taps by way of example and not
limitation). In this embodiment, weights having values of 1.0, 1.0,
-1.0, 1.0, 1.0 are used to create an upward quadratic curve in the
region of the signal of interest as generated by slaved lobe filter
structure-up 106 and output at 122. Further, the tuning filter is
tracked using a tap spacing of T with multiplier of o (o=14 in this
example) such that it tracks the tuning filter lobe center, the
slaved lobe filter structure-flat 103 and the slaved lobe filter
structure-down 104.
[0086] Referring now to FIG. 7c, there is shown a region labeled
"Signal Bandwidth" for illustrating the alternate signal path
amplitude filter shapings, before weighting and combining,
implemented upon the coupled transmitted signal 40 (see FIG. 1a) in
the quadratic amplitude matching filter (QAMF) 100. The coupled
transmission signal 40 is desirably shaped by a cosite cancellation
circuit 20 of the protected receiver 25 for the purpose of matching
the distortion introduced in the propagated interfering propagated
reference signal, contained in antenna signal 30 for improved
interference cancellation.
[0087] FIG. 8 is a more detailed circuit diagram of the quadratic
amplitude matching filter (QAMF) 100 of FIG. 1. In the presently
described embodiment, a quadratic amplitude adjustment 100 is
implemented as a block of three parallel filters 103, 104, 105,
each respectively formed in standard finite impulse response
filters having different characteristics of amplitude shapings
across the band of interest and each being formed based upon
different odd integer multiples of a basic delay interval T which
tunes such structures to a central band of interest of an
interfering signal.
[0088] The three parallel filter blocks include the slaved lobe
filter structure-flat block 103, the slaved lobe filter
structure-down block 104, and the slaved lobe filter structure-up
block 105. Each filter block 103, 104, 105 uses a common digital
control signal W.sub.T 129 as a tuning signal to track a center
frequency with different relative bandwidths, to be described as
follows.
[0089] Slaved lobe filter structure-flat 103 includes an equalizing
delay block and a simple filter. The signal enters a time delay
t.sub.a 111 controlled by the W.sub.T 129 but internal circuitry is
designed to adjust the setting of the delay implemented in t.sub.a
111 to cause the total delay through the slaved lobe filter
structure-flat 103 to that matching the simultaneous delays through
slaved lobe filter structure-down 104 and slaved lobe filter
structure-up 105. The delayed signal enters the simple filter at a
power divider 108 forming two paths, one of which feeds directly
into one port of a summing junction 110 while the second path
enters a controlled delay line 109 which is set to a delay (2n+1)T
by the same control W.sub.T 129 before entering a second port of
the summing junction 110. This delay corresponds to the delay that
would tune a similar lobed filter used for amplitude slope control
to the band of interest and bandwidth such that it is nearly flat
in the region of the signal of interest. The signal exits the
summing junction and enters a weighting device
[0090] Slaved lobe filter structure-down block 104 is the same
structure as slaved lobe filter structure-flat block 103 except
that the filter time constant for Slaved lobe filter structure-down
block 104 is increased to narrow the lobing of the filter to
generate a quadratic shape in the area of the signal of
interest.
[0091] Slaved lobe filter structure-down block 104 includes an
equalizing delay block 117 and a simple filter. The signal enters a
time delay t.sub.b 117 controlled by the same W.sub.T 129. Internal
circuitry (not shown) is included in slaved lobe filter
structure-down 104 to adjust the setting of the delay implemented
in t.sub.b 117 to cause the total delay through the slaved lobe
filter structure-down block 104 to match the simultaneous delays
through slaved lobe filter structure-flat block 103 and slaved lobe
filter structure-up block 105. The delayed signal enters the simple
filter at a power divider 114 forming two paths, one of which feeds
directly into one port of a summing junction 116 while the second
path enters a controlled delay line 115 which is set to a delay
(2m+1)T by the control W.sub.T 129 before entering a second port of
the summing junction 116. This delay is based upon the same tuning
interval T but has a multiplier of (2m+1). Multiplier m establishes
the relative bandwidth of the quadratic filter shaping.
[0092] Slaved lobe filter structure-up block 105 includes an
equalizing delay block 121 and a simple FIR filter. The signal
enters a time delay t.sub.c 121 controlled by the same W.sub.T 129
but internal circuitry is designed to adjust the setting of the
delay implemented in t.sub.c 121 to cause the total delay through
the slaved lobe filter structure-up block 105 to that matching the
simultaneous delays through slaved lobe filter structure-flat block
103 and slaved lobe filter structure down block 104. The delayed
signal enters the finite impulse response (FIR) filter 120 of
multiple taps spaced at intervals based upon the same tuning
interval T but having a multiplier of (2o+1) and individually
weighted to generate the desired upward quadratic function at the
frequency of the signal of interest. The multiplier o will cause
the filter function to track the signal of interest as the
frequency changes and the circuit is tuned in response.
[0093] The outputs of the three slaved lobe filter shaping
structures 103, 104, 105 are each respectively weighted and summed
to generate the amplitude shaping necessary to match a received
signal which has been distorted in a dispersive multipath
propagation path from a co-located transmission antenna 2.
Quiescent values of W.sub.p1 113, W.sub.p2 119, and W.sub.p3 123
will be (1.0, 0.0, 0.0), providing minimal shaping are modified
thereafter by the adaptive amplitude control 225.
[0094] FIG. 9 provides the circuit structure of a tuning control
system, according to one embodiment, for continuously and
automatically tuning a quadratic amplitude matching filter (QAMF)
100 to a band center of a reference input signal for improved
interference cancellation system performance in a cosite
interference cancellation system.
[0095] An Adaptively Tuned Quadratic Control (ATQC) module 200
comprises two main elements; an inline quadratic amplitude matching
filter (QAMF) 100 and an offline Time Delay Tuning Control (TDTC)
element 202. The quadratic amplitude matching filter (QAMF) 100
further includes a variable lobe filter structure (VLFS) 201,
slaved lobe filter structure-down 104 and a slaved lobe filter
structure-up 105. It is noted that Variable lobe filter structure
(VLFS) 201 is a variation of slaved lobe filter structure-flat 103
of FIG. 1 with the time delay structure split into two blocks to
provide a tuning feed.
[0096] The variable lobe filter structure (VLFS) 201 implements the
functionality of a tuned and quiescent amplitude slope matched
filters of the prior art but is constructed in a novel manner as a
variation of a conventional in-line Lobe Filter Structure such that
the delay line forming the lobed filter is split into two blocks of
controlled variable time delay. Specifically, the delay line is
split into a first block 205 with delay T and a second block 208
with delay 2nT, yielding a total delay of (2n+1)T. The first block
205 is used for broadband tuning and the second block 208 is
implemented as a multiple of the first block 205, thus making the
nearly-flat path more narrowband. As discussed above, a resulting
nearly-flat path filter lobe formed by the (2n+1)T relationship is
centered in the lobe of an imaginary filter orthogonal to the null
of the broadband tuning filter lobe formed by the delay T but is
never actually formed. This establishes one path of the quadratic
amplitude matching filter (QAMF) 100. The other two paths are
slaved to the tuning value T and thus track the inserted coupled
transmitted signal. The weights on the three paths are then
adjusted for quadratic amplitude matching, as taught above and then
fed into port 8 of the ICS for improved interference cancellation,
advantageously requiring no control signals from the
transmitter.
[0097] The Timing Delay and Tuning Control (TDTC) module 202 uses
signal samples output from the variable lobe filter structure
(VLFS) 201 to control the first block, T 205 for tuning, which is
central to the teachings of the present disclosure, and the second
block, 2nT 208, to implement the lobed filter function-flat used by
the interference cancellation system, which is a pre-requisite for
implementing the amplitude slope adjustment to match the propagated
path of the interfering signal.
[0098] It should be understood that the tuning filter lobe is
referred to herein as imaginary in the sense that it is never
actually formed or used in actual operation but is instead
discussed herein to provide a more complete understanding of the
interrelationships of the various control signals and the
generation of the ASMF filter.
[0099] With continued reference to FIG. 9 reference signal 215 and
delayed signal 216 are sampled and fed into a differencing hybrid
221 to form a broadband RF filter, a first broadband RF filter
having a filter response 223. Thus the offline sine function filter
formed using the variable T 205 has a null orthogonal to the center
of the imaginary tuning filter lobe based upon this T, which is
never formed.
[0100] In accordance with a method for continuously and
automatically tuning a quadratic amplitude matching filter (QAMF)
100 to a band center for improved interference cancellation system
performance, the energy passing through the filter is measured by
controller 222. The controller can then dither the digital delay
control 232 by a value .DELTA.T, either positive or negative and
re-measure the energy passing through the filter. In this way, the
controller can determine direction to skew the tuning filter to
achieve a null on the input signal 203. This assumes increasing a
control voltage causes the delay T to be increased resulting in a
narrowing of the filter lobes while decreasing the control voltage
causes the delay T to be decreased resulting in a broadening of the
filter lobes. Signs can be easily changed for devices with opposite
control functions. There are a number of search algorithms common
in the art to perform this control that provide for desired
rejection of noise and timely convergence. These include but are
not limited to random searching, gradient searching, and
perturbation using orthogonal Walsh functions, each with advantages
and disadvantages. The selected control algorithm is not part of
this present disclosure.
[0101] FIG. 10 is a more detailed circuit diagram of FIG. 1 for
illustrating an improved interference cancellation circuit 20 for
elimination of interfering signals between radio transmitter 21,
and receiver 25 where system dynamics cause changes in the coupling
between a transmit and receive antenna on a platform, according to
one embodiment.
[0102] A time delayed, quadratic amplitude matched sample of
transmission signal 40 is output from adaptively tuned quadratic
control 200 as the delayed coupled signal 57 and supplied to
auxiliary port 8 of ACL 6. Interfering propagated reference signal,
contained in antenna signal 30 is fed into reference port 9 of ACL
6. A cancellation signal 65 is generated by ACL 6 via the processes
of autocorrelation 66, integration 67, and finally by applying a
complex weight 68 of phase and amplitude. The cancellation signal
65 is provided to summing junction 70. It is noted that when the
cancellation signal 65 is injected into summing junction 70 it has
substantially the same amplitude as the interfering propagated
reference signal, contained in antenna reference signal 71,
however, the cancellation signal 65 is manipulated so that it is
180.degree. out of phase with the interfering propagated reference
signal received by antenna 4 and included in antenna signal 30 so
as to substantially cancel the interfering signal. The adaptive
amplitude control 225 of prior art is still required to adjust the
quadratic amplitude matching filter (QAMF) 100 to the proper
weights to match the quadratic amplitude distortion of the sampled
transmission signal to that of the propagation path. Adaptive
amplitude control 225 implements this process by monitoring ICS
protected output 58 while dithering control lines W.sub.p1 113,
W.sub.p2 119, and W.sub.p3 123 under a sequence determined by its
algorithm and loop feedback. As a result, the signal remaining on
the protected output 58 is substantially the same as the received
antenna signal 30 provided by receiver antenna 4 without the
undesired contribution from interfering transmitter 1. ACL 6 is
configured as a Least Mean Square (LMS) analog control loop but
those familiar with the art will realize that many different
algorithms, implemented at RF and digital, can serve this
function.
[0103] The use of sum or difference hybrids in the off-line
processing and in-line processing may be switched to design the
system for a specific tuning band and slope control lobed filter
width. This embodiment is just one configuration.
[0104] If the tuning information is available from the transmitter,
it could be used for a table lookup of the starting point for the
value of T. Thus, when the transmitter switched frequency, tuning
would start at approximately the correct value. These stored values
may be a one-time set value at manufacture or may be updated every
time the frequency is visited.
[0105] Referring now to FIG. 11 there is shown four co-located
interfering transmitters 21a-21d, by way of example and not
limitation. Four are shown for ease of explanation. To counteract
the multiple interfering transmitters 21a-21d, and thus reduce or
minimize cosite interference, the improved cosite interference
cancellation system 20 includes a common adaptive amplitude control
225 of prior art operably coupled to a common summing junction for
four Interference Cancellation Systems (ICS) 6a-6d, and four
independent Adaptively Tuned Quadratic Control (ATQC) module
200a-200d comprised of two main elements; an inline quadratic
amplitude matching filter (QAMF) 100 and an offline Time Delay
Tuning Control (TDTC) element 202 operably coupled to a common
summing junction for four Interference Cancellation Systems (ICS)
6a-6d. Four of which are shown for ease of explanation and not
limitation. In this manner, the cosite interference cancellation
process described above with reference to FIG. 10 is independently
applied to each interfering transmitter 21a-21d to protect the
single receiver 25. This figure shows a preferred embodiment with
common, shared antenna signal 30, summing junction 70 and antenna
reference signal 71. The function of the Variable Lobe Filter
Structure (VLFS) 201a-201d are in-line and must be independent but
the function of the Time Delay Tuning Control (TDTC) element 202
and adaptive slope control 225 can be shared through multiplexing
techniques implemented in prior art of adaptive arrays where the
correlation and integration functions were shared. In other
embodiments, the ICS summing junctions are daisy-chained for the
use of a standard building block at the cost of additional
potential noise insertions and longer convergence times because of
signal interaction.
[0106] Referring now to FIG. 12 there is shown an improved cosite
interference cancellation system 20 for elimination of interfering
signals between a single co-located transceiver 21 and a plurality
of receivers to be protected. In the presently described
embodiment, it is desired to protect a multiplicity of receivers,
25a-25d, four of which are shown by way of example and not
limitation. To protect the plurality of receivers 25a-25d, each
receiver is coupled to a corresponding Interference Cancellation
Systems (ICS) 6a-6d operably coupled with associated independent
Adaptively tuned quadratic control 200a-200d each comprises three
main elements; a Variable Lobe Filter Structure (VLFS) 201, a Time
Delay Tuning Control (TDTC) element 202, and a common adaptive
amplitude control 225 of prior art.
[0107] The foregoing is construed as only being an illustrative
embodiment of this invention. Persons skilled in the art can easily
conceive of alternative arrangements providing a functionality
similar to this embodiment without any deviation from the
fundamental principles or the scope of the invention.
* * * * *