U.S. patent application number 12/901132 was filed with the patent office on 2011-10-13 for x-ray and gamma ray detector readout sytem.
This patent application is currently assigned to NOVA R&D, INC.. Invention is credited to Martin Clajus, Tumay O. Tumer, Gerard Visser.
Application Number | 20110248765 12/901132 |
Document ID | / |
Family ID | 26966670 |
Filed Date | 2011-10-13 |
United States Patent
Application |
20110248765 |
Kind Code |
A1 |
Tumer; Tumay O. ; et
al. |
October 13, 2011 |
X-Ray and Gamma Ray Detector Readout Sytem
Abstract
A readout electronics scheme is under development for high
resolution, compact PET (positron emission tomography) imagers
based on LSO (lutetium ortho-oxysilicate, Lu.sub.2SiO.sub.5)
scintillator and avalanche photodiode (APD) arrays. The key is to
obtain sufficient timing and energy resolution at a low power
level, less than about 30 mW per channel, including all required
functions. To this end, a simple leading edge level crossing
discriminator is used, in combination with a transimpedance
preamplifier. The APD used has a gain of order 1,000, and an output
noise current of several pA/ Hz, allowing bipolar technology to be
used instead of CMOS, for increased speed and power efficiency. A
prototype of the preamplifier and discriminator has been
constructed, achieving timing resolution of 1.5 ns FWHM, 2.7 ns
full width at one tenth maximum, relative to an LSO/PMT detector,
and an energy resolution of 13.6% FWHM at 511 keV, while operating
at a power level of 22 mW per channel. Work is in progress towards
integration of this preamplifier and discriminator with appropriate
coincidence logic and amplitude measurement circuits in an ASIC
suitable for a high resolution compact PET instrument. The detector
system and/or ASIC can also be used for many other applications for
medical to industrial imaging.
Inventors: |
Tumer; Tumay O.; (Beverly
Hills, CA) ; Clajus; Martin; (Los Angeles, CA)
; Visser; Gerard; (Bloomington, IN) |
Assignee: |
NOVA R&D, INC.
Riverside
CA
|
Family ID: |
26966670 |
Appl. No.: |
12/901132 |
Filed: |
October 8, 2010 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
10291251 |
Nov 8, 2002 |
7818047 |
|
|
12901132 |
|
|
|
|
60331161 |
Nov 9, 2001 |
|
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Current U.S.
Class: |
327/306 |
Current CPC
Class: |
G01T 1/2985 20130101;
A61B 6/037 20130101 |
Class at
Publication: |
327/306 |
International
Class: |
H03L 5/00 20060101
H03L005/00 |
Goverment Interests
GOVERNMENT RIGHTS NOTICE
[0002] There invention was made with U.S. Government support under
Contract Numbers DE-FG03-00ER83058 and DE-FG03-00ER83058/A002, both
awarded by Department of Energy. The U.S. Government has certain
rights in the invention.
Claims
1. A method of processing at least one signal with an integrated
circuit, comprising: receiving an input signal at an input of a
channel; connecting an amplifier to said channel input to amplify
said input signal; producing a peak of said amplified input signal;
generating an arrival time signal from said amplified input signal;
producing a coincidence information from said arrival time signal.
outputting said peak; outputting said arrival time signal; and
controlling said integrated circuit and communicating with external
circuits.
2. The method of claim 1, further comprising coupling a shaping
amplifier to said amplifier.
3. The method of claim 2, further comprising coupling a comparator
to said amplifier.
4. The method of claim 3, further comprising adjusting a threshold
of said comparator.
5. The method of claim 1, further comprising adjusting a gain of
said amplifier.
6. The method of claim 1, further comprising adjusting an offset of
said amplifier.
7. The method of claim 1, further comprising selecting a shaping
time of said amplifier.
8. The method of claim 1, wherein said amplifier is self
resetting.
9. The method of claim 1, further comprising outputting an address
of said channel.
10. The method of claim 1, further comprising detecting an event in
coincidence with another event.
11. The method of claim 1, further comprising measuring said
peak.
12. The method of claim 1, further comprising multiplexing said
output.
13. An integrated circuit for processing at least one signal,
comprising: a plurality of channels with an input that receives
said signal; an amplifier connected to said input to amplify said
signal; a peak hold circuit to produce a peak of said amplified
signal; a time measurement circuit to determine an arrival time of
said amplified signal; a coincidence circuit to determine a
coincidence information from said arrival time signal. an output
circuit to output said peak and said arrival time signal; and a
control circuit to control said integrated circuit.
14. The method of claim 13, further comprising connecting a shaping
amplifier to said amplifier.
15. The method of claim 14, further comprising connecting a
comparator to said amplifier.
16. The method of claim 15, further comprising adjusting a
threshold of said comparator.
17. The method of claim 13, further comprising adjusting a gain of
said amplifier.
18. The method of claim 13, further comprising adjusting an offset
of said amplifier.
19. The method of claim 14, further comprising selecting a shaping
time of said amplifier.
20. The method of claim 13, wherein said amplifier is self
resetting.
21. The method of claim 13, further comprising outputting an
address of one of said plurality of channels.
22. The method of claim 13, further comprising detecting an event
in coincidence with another event.
23. The method of claim 13, further comprising measuring said
peak.
24. The method of claim 13, further comprising multiplexing said
output.
Description
PRIORITY CLAIM
[0001] This application is a continuation of U.S. application Ser.
No. 10/291,251 filed Nov. 8, 2002 now issued U.S. Pat. No.
7,818,047 which claims the priority to U.S. Provisional Patent
Application No. 60/331,161 filed Nov. 9, 2001, both of which are
incorporated herein by reference in their entirety.
FIELD OF INVENTION
[0003] These detectors and readout electronics have been developed
for high resolution Positron Emission Tomography (PET) for
application to medical imaging. PET is an important new modality
for imaging metabolism of organic radiopharmaceuticals and
radiotracers. The detector system described here can also be used
for non-tomographic medical imaging of positron emitting compounds.
It can also be used for other gamma ray imaging medical
applications such as Gamma Camera and Single Photon Emission
Computed Tomography (SPECT).
[0004] The instrument described can also be used for many different
applications. In industrial imaging, for example, they can be used
for different gamma ray imaging applications such as
Non-Destructive Inspection (NDI) and Non-Destructive Evaluation
(NDE). In NDI and NDE it can be used to image objects for defects,
bubbles, cracks, etc. It may also be used to detect corrosion and
cracks on aircraft and other vehicles. In security applications it
can be used to scan baggage, parcel, container and vehicle. It can
also be used to scan people and search for radioactive material. In
military it can be used in the field in a different portable
embodiment to search and image radioactive material and/or objects
that contain radioactive materials.
[0005] The Application Specific Integrated Circuit (ASIC) is being
developed can be also used for other applications. It may be used
for astrophysics and nuclear physics. It may become an important
readout chip for instruments, which use Compton Scatter technique
to image gamma rays. The ASIC may also have other applications for
medical and industrial imaging markets.
BACKGROUND OF INVENTION
[0006] The American Cancer Society estimates more than 180,000 new
breast cancer diagnoses and more than 40,000 deaths from breast
cancer in the United States in about one year. Mammography is a
useful screening tool for detecting breast cancer, reducing
mortality by about 25%, but is limited by a large number of false
positive tests resulting in unnecessary biopsies and, more
importantly, a considerable number of false negative tests
resulting in missed diagnosis of cancer. In the last few years it
has become apparent that nuclear medicine techniques have the
potential to play an important role in the diagnosis and treatment
of patients with breast cancer. Positron emission tomography (PET),
using [.sup.18F] fluoro-2-deoxy-D-glucose (FDG) as a tracer of
tumor glucose metabolic activity, is an accurate, non-invasive
imaging technology, which probes tissue and organ function. This
provides information, which is complementary to the structural
image obtained from mammography. Whole body PET is a well
established technology, however it is expensive, and of limited
availability. Furthermore, the spatial resolution is 8-16 mm,
insufficient for accurate detection and imaging of smaller tumors.
The extension of PET to small, more widely available, higher
spatial resolution (<3 mm) systems optimized for breast cancer
imaging has the potential to save many lives. Therefore, we have
designed this PET system for dedicated breast imaging. However, it
can be used for full body PET and has many other uses as described
above, in section on Field of Invention.
[0007] For the reasons of cost and availability it is unrealistic
to expect nuclear medicine techniques to be used for mass
screening. There are, however, several important situations in
which the results from mammography can be unsatisfactory, and the
availability of a functional imaging technique to provide
additional diagnostic information would be extremely helpful. These
situations include: [0008] 1. Imaging of young women with very
dense breasts (where mammograms are often of poor quality and the
detection of early stage breast cancer is difficult and
inaccurate). [0009] 2. Imaging in women with silicone breast
implants (these have high radiodensity and breast displacement is
not always possible or effective). [0010] 3. Imaging in women with
widespread fibrocystic changes. [0011] 4. Screening for
post-lumpectomy tumor recurrence--the number of women opting for
breast conservation is increasing, and functional imaging
techniques, particularly the use of FDG with PET, have been shown
to be extremely good at differentiating recurrent tumor from scar
tissue or radiation necrosis.
[0012] Encouraging preliminary studies have already been carried
out using [.sup.99mTc]sestamibi with conventional gamma cameras and
2-[.sup.18F]-fluoro-2-deoxy-D-glucose (FDG) with whole-body PET
scanners. The role of functional imaging in breast cancer, however,
goes far beyond diagnosis. It is possible that PET techniques could
become fundamental in predicting and monitoring the effectiveness
of therapy, in particular chemotherapy and hormonal therapy.
Metabolic activity as measured by FDG PET has been shown to be a
more sensitive indicator of tumor response than anatomical
techniques. This would allow early response to treatment to be
identified and the chemotherapeutic regimen altered in the absence
of a response. In addition, PET can be used to assess the
concentration of estrogen receptors using the estrogen derivative
[.sup.18F]fluoroestradiol. The concentration of estrogen receptors
is an important predictor of the outcome of hormonal therapy.
[0013] In the future, chemotherapeutic agents could be directly
labeled with positron emitters and given in trace amounts to
predict response prior to the use of pharmacological levels. This
might allow tailoring of the drug regimen to the individual
patient, leading to a reduction in the costs and morbidity of
ineffective treatments. Further interesting possibilities involve
labeling monoclonal antibodies directed against breast tumor cells
with .sup.124I. This long-lived tracer would allow the distribution
of antibodies to be visualized prior to therapy
SUMMARY OF INVENTION
[0014] Mammography allows the detection of very small, non-palpable
lesions and has become the screening modality of choice in
postmenopausal women. However, this technique has a limited
diagnostic accuracy for detecting cancer, and image interpretation
is subject to considerable inter- and intra-observer variability.
Its sensitivity drops considerably in women with dense, fibrocystic
breasts. The incidence of positive biopsies performed after
mammographic findings ranges from 9% to 65%, with most
investigators reporting a 15 to 30% positive biopsy rate.
Microcalcifications, one of the classic signs of occult
malignancies, have a low predictive value of only 11.5% for the
presence of cancer. The predictive value of masses that are thought
to definitely represent malignancies is about 74%, but masses
thought to be possibly malignant turn out to be carcinoma in only
5.4% of the cases. Several studies have reported substantial
variability among radiologists in interpretation of mammographic
examinations. Observer agreement was two-times more likely for
examinations with less dense breasts. Other factors such as age,
ethnicity and estrogen replacement status affect mammographic
sensitivity. Sensitivity was only 54% in women younger than 40
years and 68% in women with dense breasts (vs. 85% for non-dense
breasts). In summary, mammography is a useful screening tool for
detecting cancer but is limited by a large number of false positive
tests resulting in unnecessary biopsies and, more importantly, a
considerable number of false negative tests resulting in missed
diagnosis of cancer, which results in unnecessary deaths. It will
be important if false negatives can be significantly reduced to
save lives.
[0015] In the last few years it has become apparent that nuclear
medicine techniques have the potential to play an important role in
the diagnosis and management of patients with breast cancer.
Positron Emission Tomography (PET), using [.sup.18F]
fluoro-2-deoxy-D-glucose (FDG) as a tracer of tumor glucose
metabolic activity, is an accurate, non-invasive imaging
technology, which probes tissue and organ function. This provides
information, which is complementary to the structural image
obtained from mammography. Whole body PET is used clinically to
diagnose and stage a variety of cancers. It detects breast cancer
with sensitivities between 70 and 90% and specificities of 84-97%.
The somewhat lower than desired sensitivity is due to relatively
poor accuracy for detecting tumors of less than 1 cm in size. A
high diagnostic accuracy of PET imaging for staging of axillary
lymph node involvement has also been reported. The detection of
malignant breast tumors with PET is limited by the spatial
resolution and sensitivity of whole body PET systems.
State-of-the-art whole body PET systems typically yield
reconstructed images with a resolution of 8-16 mm depending on the
injected dose, imaging time, and intrinsic resolution of the
scanner. Whole body PET is also an expensive technology, which is
generally only available in the larger medical facilities in the
U.S. Therefore, a dedicated compact higher resolution PET system
that improves the sensitivity, specificity, and availability of PET
imaging for breast cancer detection, which can also be used for
many other applications is discussed below.
[0016] A highly integrated multichannel mixed-signal (both analog
and digital) front-end electronics for the LSO-based PET
(positron-emission tomography) imager is developed. The LSO
(lutetium ortho-oxysilicate, Lu.sub.2SiO.sub.5) scintillator
crystals are read out at both ends by avalanche photodiodes (APDs)
supplied by RMD Inc. (Watertown, Mass.) Innovative front-end
electronics is essential for the development of commercial PET
systems. The small scintillator area (2.times.2 mm.sup.2) leads to
a large number of channels (in the range of 5,000-20,000) and
requires high-density electronics. Therefore, multichannel
front-end electronics integrated into a mixed signal ASIC
(Application Specific Integrated Circuit) is essential to build a
compact PET imager based on APD array readout. We have designed an
innovative, fast, low-noise multichannel mixed signal ASIC for the
LSO/APD arrays for application to breast cancer diagnosis. The
development of such an ASIC involves many challenges due to its
charge-sensitive nature and multichannel design, including
crosstalk, electromagnetic pickup, feedback from digital sections
into the highly sensitive front end, and fast trigger output for
the tight PET coincidence requirement Innovation also includes the
development of highly compact readout electronics so that the PET
instrument as a whole will be compact. The approach of placing APD
arrays on both front and back sides of the LSO crystals is also an
innovative concept that poses design challenges in ensuring that
the amount of absorber material in the photons' path is kept to a
minimum.
[0017] Several design options have been investigated and a
preliminary design for the ASIC is developed with particular
emphasis on the preamplifier and discriminator sections, which we
consider the most critical components for the project's success.
The design is based on the performance requirements identified for
the PET imager in general and the readout electronics in
particular. Specifically, good timing resolution on the order of 3
ns or better and low power consumption are critical for the
practical usefulness of the PET imager that will result from this
project. By building a transistor-level prototype of the critical
components of the circuit--the preamplifier and the
discriminator--and demonstrating that it meets or even exceeds the
design goals set forth in our project, the ASIC circuit is tested
and verified.
[0018] An ASIC-based readout electronics scheme is designed for
high resolution, compact PET imagers based on independent readout
of all channels of LSO scintillator and avalanche photodiode (APD)
arrays. Depth of interaction is obtained by readout of both ends of
the LSO crystals. A low power, highly integrated design is
critical. We report here on a discrete electronics prototype,
running at 22 mW per channel for the preamplifier and
discriminator. The measured timing resolution is 3.6 ns FWHM, 9.2
ns full width at one tenth maximum, relative to an LSO/PMT
detector, energy resolution is 13.3% FWHM at 511 keV, and depth of
interaction position resolution is 2.5 mm FWHM throughout the full
length of the crystal.
[0019] The preliminary ASIC design is completed by adding other
required circuitry, such as a shaper and peak detector and trigger
logic. The ASIC is instrumental in building two LSO/APD modules,
each consisting of a 4.times.4 crystal array read out at both ends.
The module is designed to achieve a full-fledged, commercially
viable breast cancer PET detector system.
[0020] Detector modules based on recently developed
planar-processed avalanche photodiode (APD) arrays from RMD
(Radiation Monitoring Devices, Inc.) and LSO scintillator crystals
are used. The APD arrays are available with a 2.48 mm or a 1.27 mm
pitch from RMD (Radiation Monitoring Devices, Inc.); the 2.48 mm
pitch array, which we work with here has a pixel active area of
2.times.2 mm.sup.2, a gain of order 1,000, and capacitance of 2.8
pF (excluding packaging). For room temperature operation, the
leakage current is around 100 nA and the current noise is several
pA/ Hz, when operated near maximum gain (for optimal timing
resolution). The quantum efficiency is >60% at 420 nm, the peak
emission wavelength of LSO. Our early measurements have been
performed with a single channel APD of the same 2.times.2 mm.sup.2
geometry and the same specifications.
[0021] The compact geometry and low mass of the APD arrays allow
for double-ended readout of the LSO crystals, to make depth of
interaction (DOI) measurements, with the added engineering
advantage of identical readout electronics for both sides of the
crystal array. DOI measurement is critical to achieving a uniform
spatial resolution in combination with high efficiency in an
affordable instrument, with a ring diameter of about 20 cm. Another
advantage of APDs is their relative insensitivity to magnetic
fields, possibly enabling co-imaging with PET and NMR techniques in
the future.
[0022] A complete, highly integrated, low power readout electronics
chain optimized for high resolution APD/LSO PET imaging is to date
not available, although encouraging results have been reported for
individual circuit blocks such as the preamplifier and
discriminator. A high resolution PET scanner with DOI for breast
cancer imaging will involve 5,000-20,000 channels, making power
dissipation a very critical parameter. Position sensitive readout
schemes (charge division) can be used to reduce the number of
electronics channels, but bring in additional uncertainty in the
position measurement and increase the electronics channel hit rate,
requiring lower dead time. Since the avalanche gain in an APD is
relatively low (compared to a typical PMT (photomultiplier tube)),
sophisticated low noise electronics must be placed in close
proximity to the APDs, further complicating the power dissipation
issue.
[0023] For minimum system power dissipation, a leading edge level
crossing discriminator is used, carefully designed to minimize time
walk. For pulses with amplitude greater than about 50% of the
photopeak, the time walk can be controlled within 1-2 ns. However,
the leading edge of the LSO scintillation light must be observed
with the maximum possible bandwidth. A high speed transimpedance
preamplifier is used, preserving the bandwidth of the APD. In
contrast to the case of a charge sensitive preamplifier, no fast
shaper or complex discriminator is required in the timing path,
offering considerable power savings. Furthermore, pole/zero
cancellation is irrelevant for a transimpedance preamplifier, an
important advantage for high rate operation. The pulse amplitude
measurement proceeds by using a low power slow shaper and peak
stretcher followed by an A/D converter.
[0024] The APD output in response to a typical 511 keV LSO
scintillation pulse is shown in FIG. 12. The APD bias is 1,824 V,
the crystal dimensions are 2.times.2.times.10 mm.sup.3, and the
measurement bandwidth is 240 MHz. The fit 10%-90% risetime is 10
ns, so allowing a 2 ns degradation the transimpedance amplifier
bandwidth should be set around 53 MHz. The APD risetime is limited
by diffusion of electrons in the low field region above the gain
region.
BRIEF DESCRIPTION OF THE DRAWINGS
[0025] FIG. 1 is an overall block diagram of the LSO/APD PET
Imaging ASIC.
[0026] FIG. 2 is a signal channel diagram of LSO/APD PET Imaging
ASIC.
[0027] FIG. 3 is a coincidence logic diagram of LSO/APD PET Imaging
ASIC.
[0028] FIG. 4 is a drawing of the PET detector module concept for
breast imaging using two APD arrays to read out an array of
scintillator elements, providing depth of interaction
information.
[0029] FIG. 5 is a photograph of a prototype 4.times.4 pixel
avalanche photodiode (APD) array. Individual elements are 2.times.2
mm.sup.2 in size. The gap between pixels is 0.4 mm.
[0030] FIG. 6 is a schematic diagram of the circuit A. The APD is
at the top left; the other diode is for protection against
breakdown in the APD.
[0031] FIG. 7 is a schematic diagram of APD and connection to the
12.1 mW, 100 MHz, 56 k.OMEGA. transimpedance amplifier of prototype
circuit B.
[0032] FIG. 8 is a schematic diagram of the discriminator (10.1 mW)
of prototype circuit B.
[0033] FIG. 9 is a photograph of the prototype circuit B fabricated
inside a shielding box showing all the connectors and wiring.
[0034] FIG. 10 is a diagram of the test setup for
depth-of-interaction (DOI) measurements
[0035] FIG. 11 is a diagram of the setup for timing and pulse
height measurements.
[0036] FIG. 12 is a graph of a typical APD current signal as
measured by circuit A, with a .sup.22Na source and LSO crystal. The
amplitude of this pulse is typical of the photopeak.
[0037] FIG. 13 is a graph of the distribution of the time
difference .DELTA.t=t.sub.APD-t.sub.PMT between the trigger times
of the APD and PMT signals. The curve represents a Gaussian fit to
the data.
[0038] FIG. 14 is a graph of the distribution of the time
difference .DELTA.t=t.sub.APD-t.sub.PMT between the trigger times
of the APD and PMT signals after a pulse height cut corresponding
to an energy deposit of at least 300 keV detected by the APD.
[0039] FIG. 15 is a graph of .sup.22Na spectrum from LSO read out
by a single-channel APD using prototype circuit B.
[0040] FIG. 16 is a graph of .sup.137Cs spectrum from LSO
scintillator read out by a single-channel APD.
[0041] FIG. 17 is an oscilloscope screen plot of typical
APD/preamplifier output pulses and also discriminator OR
output.
[0042] FIG. 18 is a graph of a DOI measurement: Front vs. back
pulse height scatter plot, aggregate of five different z positions
(0 mm, 4 mm, 8 mm, 12 mm, 16 mm). An approximate energy cut at 250
keV used in some of the analysis (F/620+B/525>1) is equivalent
to a line from 750 (vertical scale to 600 (horizontal) scale.
[0043] FIG. 19 is a graph of a DOI measurement:pulse height ratio
(A-B)/(A+B).
[0044] FIG. 20 is a graph of DOI calculated by the pulse height
ratio (A-B)/(A+B).
[0045] FIG. 21 is a graph of timing differences measured between
APD and PMT; APD vs. PMT, no energy cut.
[0046] FIG. 22 is a spectrum of .sup.137Cs using a
2.times.2.times.10 mm LSO crystal with single side readout.
[0047] FIG. 23 is a graph of energy resolution at 662 keV as a
function of shaping time (signals were peak-detected).
[0048] FIG. 24 is a graph of energy resolution at 662 keV as a
function of sampling time.
[0049] FIG. 25 is a graph of crystal identification measured with
an array of 2.times.2.times.10 mm.sup.3 LSO crystals coupled to the
APD array and read out by the RENA.TM. (Readout Electronics for
Nuclear Application) signal processor. The image on the left is a
flood source histogram of the array, and the plot on the right is a
profile across one row of the crystal array.
[0050] FIG. 26 is a graph of crystal identification measured with
an array of 2.times.2.times.10 mm.sup.3 LSO crystals coupled to the
APD array and read out by two HQV802-M hybrids with multiplexed
readout. The image on the left is a flood source histogram of the
array, and the plot on the right is a profile across one row of the
crystal array.
[0051] FIG. 27 is a graph of timing resolution, APD vs. PMT showing
the excellent timing that was obtained using APDs.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0052] There has been considerable interest in recent years in
developing dedicated high resolution positron emission tomography
(PET) systems for applications in breast cancer imaging and
small-animal imaging. The goal in these systems is to achieve much
higher spatial resolution and sensitivity for specific tasks than
is possible with whole-body PET scanners designed for general
purpose use. A second goal is to produce relatively inexpensive,
compact and easy to use systems that make PET more accessible.
Generally, these dedicated systems use small scintillator elements
read out by position-sensitive or multi-channel PMTs. In most
systems, some form of signal multiplexing is used to reduce the
number of channels to a manageable number. Since the predominant
mode of interaction at 511 keV in all scintillators currently used
for PET is Compton scatter, multiplexing can lead to significant
loss of position information. Furthermore, depth-of-interaction
(DOI) blurring or radial elongation error becomes a prominent
feature in these small diameter systems, and therefore several
groups have been exploring detector approaches that can measure
DOI.
[0053] Recently, avalanche photodiode (APD) arrays have become
available that when combined with Lu.sub.2SiO.sub.5 (LSO)
scintillator crystals offer new opportunities for high resolution
PET detectors. This work focuses on the high-gain APD arrays
developed by RMD, Inc. (Watertown, Mass.). The APD arrays are
available with a 2.48 mm (16 channels) or a 1.27 mm pitch (64
channels); the 2.48 mm pitch array, which we work with here has a
pixel active area of 2 mm.times.2 mm, a gain of order 1,000, and
capacitance per pixel of 2.8 pF. For room temperature operation,
the leakage current is around 100 nA and the current noise is
around 5 pA/ Hz, when operated near maximum gain (for optimal
timing resolution). The quantum efficiency is greater than 60% at
420 nm, the peak emission wavelength of LSO. The work reported here
has been performed with a single channel APD of the same 2
mm.times.2 mm geometry and the same specifications.
[0054] The compact geometry and low mass of the APD arrays allow
for double-ended readout of the LSO crystals, to make DOI
measurements, with the added engineering advantage of identical
readout electronics for both sides of the crystal array. DOI
measurement is critical to achieving uniformly high spatial
resolution in combination with high sensitivity in an affordable
instrument, with a ring diameter of about 20 cm. Furthermore, the
use of completely independent readout channels for each crystal of
the array, instead of a position sensitive readout scheme, may
enable the accurate analysis, or the unambiguous rejection, of some
events involving Compton interactions in the scintillator array.
Overall system deadtime can also be significantly reduced by using
independent readout channels for each crystal of the array.
[0055] Individual readout of each crystal of the array places a
high premium on cost, power dissipation, and size and mass of the
readout electronics. Most readout functions, including all
functions required on a per-channel basis, will have to be
integrated into an ASIC before such a system becomes viable. A
complete, highly integrated, low power readout ASIC optimized for
high resolution PET imaging with LSO/APD arrays is to date not
available, although encouraging results have been reported for
individual circuit blocks such as the preamplifier and
discriminator. We developed such readout electronics, specifically
optimized to meet the crystal identification, timing, energy, and
DOI requirements of high resolution PET while minimizing system
complexity and cost. The results, confirming that the architecture
and specifications of our readout electronics will deliver the
performance required for high resolution PET, are presented
here.
[0056] The APD gain is sufficiently high that the principal
electronics noise contribution in the system is the current noise
of the APD itself. Therefore, we use a transimpedance amplifier
input stage instead of a charge sensitive amplifier. The design
minimizes power while preserving the relatively short (5 to 10 ns)
risetime of the LSO/APD signal. Since the APD capacitance is only
about 2.8 pF, the transimpedance amplifier can have a wide
bandwidth with still relatively small noise contributions from the
voltage noise of the open loop amplifier and the current noise of
the feedback resistor.
[0057] For the timing pick-off, a leading edge discriminator is
used. This will lead to time walk, although--since the system noise
is low enough to allow a threshold around 50 keV or less--the time
walk for energies relevant to PET is under control. The long
crystals, with surfaces optimized for DOI measurement, add the
complication that the pulse height, even for the photopeak, may be
small for one of the APDs; to cope with this, we take the time
pick-off from either the front or back APD, whichever is the first
to cross threshold.
[0058] For the pulse height measurement a two-pole low pass filter
to shape the pulses with a peaking time of 180 ns, and capture the
pulse height in a sample and hold circuit timed from the
discriminator output is used. The pulse height is then digitized by
a 12-bit successive-approximation A/D converter. (The ASIC will
also include a sparse readout circuit to read the pulse height from
the front and back APDs of all and only those LSO crystals which
are over threshold for a given event.)
[0059] The developed system was studied for depth of interaction,
energy, and timing resolution. For all of these measurements the
APD bias voltages were 1,752.3 V and 1,737.3 V, with absolute
accuracy .+-.1.5%; stability and peak-to-peak noise is less than
100 mV. The bias voltages were tuned for the maximum reasonable
gain, beyond which the preamplifier output showed a significantly
increased noise level. The average photopeak pulse amplitudes seen
from the two APDs were within a factor of two of one another. The
measured temperature was 30.degree. C., but no active temperature
control system was used.
[0060] The discriminator thresholds were set at 177 nA and 99 nA,
respectively. Pulses from the 511 keV photopeak signal have an
amplitude around 1.9 .mu.A, by comparison, so that in energy terms
the timing threshold is set less than 47 keV.
[0061] The LSO crystal 102 dimensions were 2 mm.times.2 mm.times.20
mm; the long faces were plain saw-cut surfaces and both 2
mm.times.2 mm end faces were mechanically polished. The crystal was
wrapped in white teflon tape and coupled to the APDs with a small
amount of Bicron BC-630 optical grease.
[0062] The second detector 120 for all our measurements was
composed of a polished 2 mm.times.2 mm.times.10 mm LSO crystal,
coupled end-on to a Hamamatsu R1635 PMT 104, 124. A leading edge
discriminator 45 (constructed from a Motorola MC100LVEL16
integrated circuit) was used for timing from the PMT.
[0063] For the DOI tests, a 3.21 MBq .sup.22Na source 103, 122
(diameter 1 mm) was placed at a distance of 48 mm perpendicularly
from the side of the LSO crystal. The LSO/PMT detector was placed
at a distance of 72 mm on the far side of the source. Thus in
coincidence a spot size of order of 1.4 mm FWHM is illuminated on
the 2 mm.times.2 mm.times.20 mm LSO crystal 102, ignoring the
effects of the positron range and momentum. The PMT and the source
are fixed to a linear motion table parallel to the 2 mm.times.2
mm.times.20 mm LSO crystal 102. The position of the motion table,
and hence of the illuminated spot on the LSO crystal, is labeled
here by the coordinate `z`. No absolute position calibration was
used; z=0 mm is arbitrary, though it is near the back end of the
crystal. We recorded 5,000 events at each z position from z=0 mm to
21 mm by steps of 1 mm.
[0064] DOI measurement proceeds by a comparison of the
scintillation light detected at the front and back ends of the
crystal. FIG. 18 shows a scatter plot of pulse height measured on
the front APD (F) vs. pulse height measured on the back APD (B),
with the source located successively at four different z positions
separated by 6 mm. The energy and DOI capability can be quickly
appreciated from a consideration of this plot. The baseline level
digitized from the A/D converter with zero pulse input has been
subtracted from this data (and similarly for the remainder of this
paper). No other corrections have been applied to the pulse height
data as measured by the A/D converter.
[0065] For events at a given z position, the ratio of the front APD
pulse height to back APD pulse height is expected to be a constant,
and ideally there is a one-to-one correspondence between z and the
ratio F/B. It is convenient to use the ratio F/(F+B), or the angle
arctan(F/B), for analysis instead of the ratio F/B. FIG. 19 shows
histograms of the angle determined from the front to back pulse
height ratio, (F-B)/(F+B). Or it is possible to use the direct
ratio F/B.
[0066] DOI resolution is degraded for low-energy events, where the
angular separation in the front vs. back scatter plot evidently is
not as great. Typically, however, lower energy thresholds of
between 250 and 350 keV are used in a PET system. We therefore also
explore the effect of an energy cut (can be represented as a line
from 750 (vertical scale) to 600 (horizontal) scale on FIG. 18 for
250 keV cut) on the DOI resolution. Since the DOI resolution shown
here is good, it will be important to increase the crystal length
beyond 20 mm, which will boost the detection efficiency
further.
[0067] The position of the event may be inferred from the measured
front to back pulse height ratio. From our data we obtain the curve
shown in FIG. 20. Over the central 16 mm of the crystal, the
position resolution averages 2.86 mm FWHM if no energy cut is
applied, and 2.53 mm FWHM when the energy cut is used. The position
resolution degrades at the ends of the crystal, probably due to the
effects of direct interactions in the APDs and to the fact that
this data is taken at a constant number of events for each z
position, obviously increasing the relative contribution of Compton
scattered events when the main photon beam is past the end of the
crystal. However, if the physical constraint that the interaction
occurs inside the crystal is taken into account, then measured
position FWHM around 2.5 mm can be recovered.
[0068] Energy resolution is also measured extensively. To discuss
the energy resolution, it is useful to examine the front vs. back
pulse height scatter plot. As a first approximation, the energy may
be expressed as E=.alpha.F+.beta.B. The relative coefficients are
determined by a line fit to the photopeak region in the scatter
plot. The resulting energy spectrum is shown in FIG. 15. The energy
resolution is about 14% FWHM. The observed photopeak to Compton
ratio is about 0.7, which is roughly in agreement with the 0.521
expected for a small LSO scintillator. The difference probably can
be attributed to the nonzero energy threshold and to
multiple-interaction events.
[0069] In the front-back scatter plot two effects were clearly
visible which can limit the energy resolution, at least in
principle. The photopeak did not appear as a perfectly straight
line, but rather is bowed in slightly in the middle region,
indicating, as is to be expected, a lower light collection
efficiency for events near the middle of the crystal. Also the
photopeak is broadened and reduced in amplitude near the ends of
the crystal, which can probably be attributed to total internal
reflection from the end of the crystal. The critical angle between
LSO and BC-630 grease is 53.6.degree., so this only begins to occur
within 1.47 mm of the ends.
[0070] The depth of interaction information may be applied in an
attempt to improve the energy resolution, writing the energy as
E=(.alpha.F+.beta.B)f(F/B) and determining the coefficients and the
correction function f from a fit to the photopeak region in the
scatter plot. This technique may produce useful improvement.
[0071] We studied the coincidence time resolution (FIG. 13, FIG.
14, FIG. 21, FIG. 27) with the LSO/APD and LSO/PMT detectors
positioned in a line, 140 mm apart, and the source mid-way between
them (FIG. 11). The time difference was measured using a TDS7104
digital oscilloscope (bandwidth 1 GHz, sample interval 200 ps) in
delay measurement mode. With no explicit energy cut the coincidence
time resolution is 4.6 ns FWHM. Applying a cut for greater than 250
keV, coincidence time resolution is 3.6 ns FWHM, 9.2 ns full width
at one-tenth maximum (FWTM); time walk correction reduces this to
3.4 ns FWHM, 7.5 ns FWTM.
[0072] Some amount of time walk was evident at all energies,
although above 250 keV the effect of time walk is minimal. If
required, residual time walk correction could be applied by
programmable logic resources on the detector module board or on the
ASIC.
[0073] The resolution is limited by time walk in the LSO/PMT
readout (left side of the peak in FIG. 21), and possibly by noise
in the APD readout electronics. We addressed these issues by
optimizing the amplifier bandwidth and gain, and by making direct
APD/LSO-APD/LSO timing measurements.
[0074] The ASIC (application specific integrated circuit) is the
crucial part of this work for producing a compact and full function
PET system. FIG. 1 is an overall block diagram of the readout chip
(ASIC), of which only two signal channels, 5 and 17, are shown.
Additional signal channels are represented by the ellipsis 29. An
input signal 1, 13 from a high-gain avalanche photodiode is sent to
each signal channel. Additional inputs to each signal channel are
for user-supplied timing signals V.sub.U 2, 14 and V.sub.V 3, 15,
for a test signal 4, 16, and for a sample/hold control signal 22,
23 generated by the coincidence logic and sample/hold control
circuit 25. In another embodiment the sample/hold circuit may be
replaced by a peak/hold circuit. Each signal channel outputs
signals AOUT 6, 18, UOUT 7, 19, VOUT 8, 20, and HIT 9, 21. The
voltage output on AOUT 6, 18 indicates the size of the detected
signal. The voltages output on UOUT 7, 19 and VOUT 8, 20 are
representative of the time at which a signal was detected and may
be used in the analysis of the data acquired with the readout chip
to distinguish true detector signals from noise signals. These
signals may also, be used for timing the arrival time of the signal
or event, such as the arrival time of the x-ray or gamma ray
photon. These signals are input to an analog or digital multiplexer
or a shift register 10 and from there through a buffer circuit 11
to an analog-to-digital (A/D) converter 12. The HIT signal 9, 21
from signal channels 1 to n is input to the coincidence logic and
sample/hold control 25. Here 25, the HIT signals from all signal
channels are combined with the signal CORR_IN 26, to decide whether
the detector system has detected a valid positron emission event.
CORR_IN 26 may be generated inside the chip, also called ASIC, or
on the printed circuit board on which the chip is located, by
forming the logical OR of the CORR_OUT signals coming from other,
readout chips 28 that form part of the PET detector system. At the
same time, a signal CORR_OUT 27 is distributed to the other readout
chips, to be used to generate the CORR_IN signal for these chips'
coincidence logic, in the same manner as described above. In
another embodiment the CORR_OUT signals from different chips can be
routed to a logic circuit (not shown), which makes a decision if
two or more signals received by different chips are coincident or
not.
[0075] Coincident means that two or more gamma rays incident on two
or more separate detector arrays FIG. 431 arrive within a short
time depending on the geometry measured in nanoseconds, about 1 to
1,000 ns. Such events are well known for positron annihilation,
which produces two gamma rays back-to-back. These gamma rays,
therefore, travel in opposite direction at about 180.degree. from
the vertex of annihilation. Therefore, if both gamma rays are
detected then the position of the positron annihilation is
somewhere along the cord that connects the two detection sites or
pixels. This information is used in PET to image human body
especially metabolism in living tissue. There are other
applications where two or more gamma rays may be generated in
coincidence. The detector system developed and discussed here can
be used for detection and imaging of such applications and sources.
The developed system can also be used for applications where single
photons are emitted by a source or an object, detected and/or
imaged by the detector presented here. In such a case there is no
need for coincident detection and these sections of the detector
may not be used or deployed.
[0076] The ASIC has an onboard readout logic circuit (not shown)
which controls all the chip's functions and also outputs the
channel address(es) from which it has received a signal. This
circuit can do many functions besides controlling the ASIC
functions and outputting channel address(es). It can, for example,
supply information and control gain; offset and threshold
adjustment adjusting circuits such as digital to analog converters
(DACs); assist in controlling multiplexer(s) and shift register(s);
monitor chip temperature; controlling calibration and testing
modes; turn on and off certain sections of the chip such as channel
and test inputs; reset certain sections of the ASIC; monitor chip
functionality and status; and output chip status and problems that
may occur. Analog to digital (A/D) converter 12 may also be
designed to reside onboard the ASIC.
[0077] The data acquisition computer will use the information from
the ASICs reading out the APDs 30 on each end of the LSO detector
array 31 to determine if there is a signal from the two ends of a
single LSO crystal which is necessary do determine the DOI (depth
of interaction) in that crystal. FIG. 25 and FIG. 26 demonstrates
how the array is read out and what the imaging of gamma rays looks
like. FIG. 10 demonstrates the DOI principal and measurements
carried out at NOVA R&D, Inc. This information limits the
interaction point of the gamma ray to a small section of the
crystal 102, thus eliminating radial elongation error inherent in
present PET detectors which use either BGO or LSO or any other
scintillator material. The post analysis of this information after
the data is stored into the memory of the computer is slow and
delay image processing and also requires storing extra data, which
does not have signal from both sides of the LSO crystal. To solve
the slowness of the data analysis, in another embodiment, the ASIC
can be designed with required DOI determination circuitry on board
of the chip to analyze the signals received from the APD arrays at
the two ends 30 (100, 101, 102) of the LSO detector array 31. To
facilitate this function the two chips from each end of the LSO
array 31 can be designed to be daisy chained so that they will act
as if a single ASIC. This will allow the signals from both ends of
the LSO array 31 to be analyzed as if recorded inside the same
ASIC. This circuit (not shown) can determine if a signal is
received from the opposite ends of each LSO crystal in the array
and would not produce an event trigger if they did not, thus
reducing data rate, and calculating the DOI and outputting it as an
analog or digital signal. In another embodiment, the event trigger
and/or DOI will be output only if there is a coincident event
detection informed to the ASIC from other chips through the CORR_IN
26 signal.
[0078] To speed the readout time and rate the ASIC is designed to
have sparse readout capability. This capability will allow the
readout of the channels with signal (data) only. The other channels
will not be read out. There will be a mode which will allow readout
of all the channels for testing and calibration. There will be also
a test mode where the analog signals will be routed to the output
from the analog sections bypassing the S/H circuit and other
digital sections. There may be also a hit register circuit which
will output the channels which have signal. This information can be
used to readout channels adjacent to the channel with signal if
needed.
[0079] The signal channel 5, 17 is shown in detail in FIG. 2. The
detector signal 130 is input to a high-bandwidth transimpedance
amplifier 133, with feedback resistance 134. Alternatively, a test
signal 131 can be selected, via a switch 132, as input to the
amplifier 133. The high bandwidth enables the amplifier 133 to
faithfully reproduce the fast rise time of the detector signal 130
coming from the avalanche photodiode. Instead of APD other
sensitive and fast light detectors such as a fast photodiode, a
photomultiplier tube (PMT), solid state photomultiplier, or a multi
anode photomultiplier tube (MAPMT) may be used. The analog output
signal generated by the amplifier 133 is sent to a shaping
amplifier 135 and from there to a buffer 136. The signal from the
buffer 136 charges a capacitor 138 through 137 as long as the
sample/hold switch 139 is closed. When the switch 139 is open, the
charge is held on the capacitor 138 and is provided on the AOUT
output 141 via the buffer 140. This constitutes the analog signal
output. This output 141 may be digitized through an A/D converter
built on the ASIC or outside the ASIC.
[0080] The output from amplifier 133 is also sent to a
discriminator 143. Whenever the output voltage from amplifier 133
exceeds the voltage supplied to the threshold input 142, the HIT
signal 144 will be activated. The HIT signal 144 is provided to the
coincidence logic and sample/hold control 25. The HIT signal 144 is
also sent to a latch 155, whose output controls switches 146 and
151, which are used to sample and hold the timing voltages V.sub.U
145 and V.sub.V 150. may be supplied to the readout chip externally
or generated directly on the chip. V.sub.U 145 and V.sub.V 150
represent known time-variable (typically periodic) voltage signals.
When the latch 155 is activated, it opens the switches 146 and 151,
causing the momentary values of V.sub.U 145 and V.sub.V 150 to be
held on capacitors 147 and 152, respectively. These values are
output through buffers 148 and 153 as signals UOUT 149 and VOUT
154, which can thus be used to determine the time at which the HIT
signal 144 was activated. The latch 155 is reset after readout of
UOUT 149 and VOUT 154 is complete. Or latch 155 can be used to
sample the V.sub.U and V.sub.V with a short gate time. This allows
instantaneous sampling for some applications.
[0081] FIG. 3 shows details of the coincidence logic and
sample/hold control circuit 25. The HIT signals from all signal
channels on the readout chip are input to the coincidence logic.
FIG. 3 shows two HIT signals 160, 163; the signals from the other
channels, as well as the signal processing paths for these signals,
are represented by the ellipsis symbol 161. The coincidence logic
requires signals of a well-defined duration, which are generated
from the HIT signals 160, 163 by retriggerable one-shot circuits
162, 164. One-shot circuits produce a pulse whenever they are
triggered, where the duration of the pulse is adjustable. From the
outputs of these one-shot circuits, and those for the other
channels on the readout chip, represented by the ellipsis 170, a
logical OR 167 is formed. The resulting signal CORR_OUT 171 is
provided to the CORR_IN signal inputs 26 of other, similar readout
chips that form part of the PET detector system. CORR_IN 172 is
used to form coincidences 168, 173 with the output signals from the
one-shot circuits 162, 164. To compensate for the propagation delay
of the CORR_IN signal 172 coming from other, identical readout
chips that form part of the PET detector system, the output signals
from the one-shot circuits 162, 164 have to be sent through the
delay circuits 166, 165 before being input to the coincidence
circuits 168, 173. Whenever a coincidence circuit 168, 173 detects
a valid coincidence, a latch 177, 178 is triggered to activate the
corresponding sample/hold signal 169, 174. At the same time, a
readout trigger is generated and sent to the chip's readout logic
(not shown). For each channel for which a valid coincidence has
been detected, the readout logic then causes the multiplexer 10 to
send the AOUT 6, 18, UOUT 7, 19, and VOUT 8, 20 voltages, in turn,
to the output buffer 11 and from there to the A/D converter 12. The
digitized voltage data, together with the channel addresses for the
respective channels, are then sent to the data acquisition computer
(not shown). This function can be also incorporated into an
electronic circuit and implemented in real time to determine
automatically the wanted events. Such hardware circuit can increase
the data acquisition rate and reduce post-processing time as
superfluous data will be significantly reduced.
[0082] FIG. 4 shows a diagram of a scintillator array 31 viewed
from both ends by two two-dimensional APD arrays 30. The
scintillator can be made of LSO, BGO, or other high Z
scintillators. The APD array 30 (FIG. 5, 35) is optically connected
to the array of scintillator crystals 31. The connection can be
rigid using epoxies or similar compounds or can be non-rigid or
flexible. The individual detector crystals 31, 102 can be left as
they are or wrapped by reflecting, diffusing or nonreflecting
material to reduce crosstalk and/or improve light collection
capability to improve signal to noise ratio. In another embodiment
the crystals can be separated through separators placed in between
crystal layers. In another embodiment, the crystal array may be a
single uncut or partially cut crystal. In another embodiment the
detector arrays 31 are placed on each side of an object. Or a
single or multiple rings of detector arrays 31 can be placed around
the object to be imaged. Septas may be placed in between or inside
the detector arrays or a ring of detector arrays. Also hole or slot
collimators may be placed in front or around the detector array(s)
31.
[0083] FIG. 5 is showing a photograph of an APD array 35. This is a
4.times.4 array of APDs built as a single monolithic block. APDs
with a larger are or array can also be built. The wires coming out
of the array is used to connect the APDs to the amplifier inputs 1,
13, 130. FIGS. 6 and 7 also show the APD 41, 61 connected to BIAS
Voltage 40, 60 and Amplifier input 43, 63. A protection diode 42,
62 is used to protect the transimpedance amplifier 43, 63 if APD
41, 61 fails. The voltage is supplied through VAA 46, 67. A
standard fast transimpedance amplifier 43 is used to make
measurements using prototype circuit A (FIG. 6). A low power and
low noise amplifier 63 is developed using components for prototype
circuit B (FIG. 7). Differential outputs of amplifier 43 is
connected to the discriminator 45 through buffers 44. Differential
output 50 of discriminator is output. An amplifier 48 is used to
supply an offset to the output of the amplifier 43 to form a
threshold so that discriminator is not triggered if the its input
does not exceed a value determined by the threshold voltage 47. The
output of amplifier 48 is supplied to the discriminator input
through transistor 49. This will control the threshold voltage of
the discriminator. In some cases discriminator is called
comparator.
[0084] FIG. 7 shows a similar circuit for FIG. 6 but it is made
using individual electronic components so that it will be low
power. Output of transimpedance amplifier 63 goes to a buffer 64.
Buffer 64 feeds the amplified signal to an amplifier gain stage 65
which then goes into another buffer 66. The output of buffer 66 is
a differential signal and it is output from the chip 71. Circuit
has two power supplies VAA 67 and VAA2 68. Transistors 69 are
current sources. Amplifier 70 produces the offset voltage to the
analog output as discussed above using the threshold voltage VTHR
72. The outputs of the circuit shown in FIG. 7 goes into a
discriminator circuit shown in FIG. 8. This circuit is also made
using components so that it will be low power and fast. Low power
operation is important to achieve room temperature functionality
for a large multi-channel instrument. Inputs 80 to the
discriminator goes into gain stages 81, 84 and 86 one after the
other. In between there are buffers 83, 85 and 87. The output 91 is
differential and goes to the readout electronics. Transistors 90
are current sources. Power VAA 88 supplied to the circuit.
Resistors 92, 93, 94, 95 are for to produce hysteresis in the
discriminator circuit.
[0085] FIG. 9 is a photograph of the prototype circuit B built for
testing and demonstrating the capability of the instrument
developed.
[0086] FIG. 10 shows a drawing of the test setup and circuit for
the DOI measurements. The APDs 100, 101 on both ends of the
detector crystal 102 detect the light produced in the crystal 102
by gamma ray coming from .sup.22Na source 103. .sup.22Na source 103
is a positron emitter which annihilates in the material when comes
into contact with an electron and emits two gamma rays with energy
of 511 keV. The second gamma ray is detected by the second LSO
crystal 117 mounted on a PMT 104. The second crystal 117 defines a
direction for the gamma rays emitted back-to-back and therefore the
first gamma ray detected inside a small section of detector crystal
102 at a set depth. This depth is measured and first gamma ray
generated light is detected by both APDs 100, 101 simultaneously.
APD converts the light signal into electrons. The electron pulses
are amplified by preamps 106, 108 and turned into signals. The two
signals go to shapers 109, 111. Shaper circuits shape the signal
into inverted or non-inverted bell shape and output. Output of the
shaper circuits 109, 111 go to separate analog-to-digital
converters ADC 114, 116. the pulse height of these signals are
measured by the ADCs 114, 116. The output of the preamplifiers 106,
108 also go to two leading edge (LE) discriminators 107. The output
of the leading edge (LE) discriminator 107 goes to coincidence unit
112. The output of the PMT 104 also goes to a third LE
discriminator unit 105. Output of LE discriminator also goes to the
coincidence unit 112. Coincidence unit produces a trigger signal
115 which means that two gamma rays have been detected
simultaneously coming from .sup.22Na source 103 at the detector
crystals 102, 117. The ADC 114, 116 outputs from coincident events
as determined by the trigger signal 115 are used to calculate the
depth-of-interaction (DOI) for the gamma ray detected inside
detector crystal 102. The DOI results obtained from these
measurements are shown in FIG. 18, FIG. 19 and FIG. 20.
[0087] FIG. 11 shows a diagram of the setup used to carry out
timing and pulse height measurements. Two detector crystals 120,
123 are used. The detector 120 is mounted on an APD 121 and
detector 123 is mounted on a PMT 124. The output of APD 121 goes to
a preamplifier 126. Output of the preamplifier 126 goes both to a
discriminator 128 and an oscilloscope 127. Output of discriminator
128 goes to oscilloscope 127. The output of detector 124 goes to a
second discriminator 125. Output of the second discriminator goes
to the oscilloscope 127. The output of oscilloscope 127 goes to a
computer through a GPIB bus 129. This circuit is used to carry out
timing and pulse height measurements. The results of such
measurements are shown in FIG. 12, FIG. 13, FIG. 14 FIG. 15, FIG.
16, FIG. 17, FIG. 21, FIG. 22, FIG. 23, FIG. 24 and FIG. 27.
[0088] An ASIC design is developed which is optimized to read out
high-gain, fast APD arrays for use with LSO scintillator in PET
imaging, and to build a prototype PET module based on this ASIC.
Presently, no multichannel fast readout chips for APD arrays are
commercially available, and even those that were developed
non-commercially typically do not match the characteristics of the
high gain, low noise RMD APDs. Chips that were not specifically
developed for APD readout lack one or more of the required
features, such as a fast, low jitter trigger output or the input
capacitance or dynamic range to match the APD characteristics. The
developed chip will also have other applications for readout of APD
arrays and multianode PMTs wherever fast, accurate timing is
required.
[0089] In designing the readout electronics for an LSO/APD based
PET system, the main consideration was to obtain high-resolution
coincidence timing. This is required to achieve the combination of
high singles count rates and low accidental coincidence rates that
is needed for high-contrast PET imaging. Spurious coincidences
create an image background by yielding reconstructed photon
directions that have no correlation with the actual source
distribution. By a rough estimate, we expect a singles rate of
about 1.5 million counts/s in a PET system consisting of an 18 cm
diameter, 2 cm axial length detector ring (2,000 pixels) for a 10
mCi injection. Assuming that these counts are evenly distributed
over all pixels and that coincidences are formed between each
module and a 120.degree. ring section across from it (for a field
of view of half the ring diameter), this leads to an accidental
coincidence rate of 750 counts/s, per nanosecond timing resolution.
This has to be compared to an estimated true coincidence rate of
10-15 kcounts/s. Our measurements demonstrated that a coincidence
timing resolution of better than 2 ns FWHM is achievable for
coincidences between an APD and a (significantly faster) PMT. By
taking coincidences between two APDs instead, we expect the width
to increase by no more than 50%. Based on that, and the system
requirements outlined above require a coincidence timing resolution
<3 ns FWHM measured between two APD channels with a positron
annihilation source and 2.times.2.times.20 mm.sup.3 LSO
crystals.
[0090] To achieve the goal of high-resolution timing, we have
improved the preamplifier and the timing discriminator, FIG. 7 and
FIG. 8, respectively. As discussed above, a low-noise, fast
preamplifier will help improve timing resolution in two ways, by
reducing the (noise-induced) amplitude variations that invariably
translate into timing fluctuations, and by minimizing (by virtue of
a fast signal rise time) the direct slope-induced time walk, for
instance in a leading edge level crossing discriminator. Based on
our results, a leading edge discriminator design will be sufficient
for obtaining good timing resolution; this will keep chip's power
dissipation low. However, other discriminator or comparator designs
may be used such as constant fraction discriminator, which has
better timing accuracy and low timing jitter than the leading edge
discriminator for fast pulses. A constant fraction discriminator or
another discriminator, which takes into account the pulse height
difference between different pulses, may be used in this
application either integrated onto the ASIC our built outside.
[0091] In order to make a practical large-scale high-resolution
coincidence system, it is also necessary to address the issue of
controlling signal-independent systematic variations from channel
to channel in propagation delay (and therefore also in signal
baseline and discriminator threshold, at least with leading edge
discrimination). To do so, we minimized the variations and more
importantly their temperature coefficients, and also implemented
delay tuning circuitry on a channel-by-channel basis to line up the
discriminator outputs in the coincidence logic.
[0092] Energy measurement of each pulse is also important, in order
to reject the background from scatter within the imaged object or
other material. However, only a modest energy resolution is already
sufficient for this purpose. As discussed above, our work with this
APD has yielded an energy resolution of approximately 15% FWHM at
511 keV, and we expect at most minor changes to this value in the
ASIC, due, for example, to further optimization of the shaper
parameters or unexpected noise pickup.
[0093] For maximum sensitivity, and hence minimum total dose to the
patient, at high event rates, it is very important that the front
end electronics and readout system impose the minimum practical
dead time due to event processing. We took this into account in the
design of the readout circuitry, and expect to meet a deadtime
specification of no more than 200 to 300 ns per hit, and
furthermore that this deadtime will only apply to the channels
which are hit, not to an entire APD module or readout group.
[0094] Timing, measured relative to a second LSO crystal and a
Hamamatsu R1635 PMT, is shown in FIG. 14 and FIG. 27. An energy cut
at 300 keV is used. The resolution is 1.5 ns FWHM and 2.7 ns FW at
one tenth maximum. The energy spectrum is shown in FIG. 15, with a
resolution of 13.6% FWHM at the 511 keV.
[0095] Tables 1 to 4 summarize the achievements, energy
measurements, readout electronics specifications and ASIC
specifications.
TABLE-US-00001 TABLE 1 Achievements 1. Timing, energy, and
depth-of-interaction resolutions for readout of 2 .times. 2
mm.sup.2 cross section LSO crystals using avalanche photodiodes
achieved. 2. A discrete electronics prototype DOI LSO PET detector
element implementing the required readout functions efficiently,
from the point of view both of circuit complexity and power
dissipation is constructed. This prototype uses a single 2 .times.
2 .times. 20 mm.sup.3 LSO crystal and an APD on the front and back
ends. 3. Energy, depth-of-interaction, and timing resolution of the
prototype system is measured. 4. A readout ASIC to implement the
functions tested in the prototype system is designed.
TABLE-US-00002 TABLE 2 Energy Measurement And Shaper Design 1.
Because a transimpedance preamplifier is used, and the dominant
noise source is the APD, the preamplifier output noise is
essentially white, and optimal response may be expected with any
shaper with a peaking time greater than about 40 ns, the decay time
of LSO scintillation. This has been verified by studying a fixed
data set with varying shapers, as follows: 2. Feed APD preamplifier
and discriminator signals and PMT discriminator signal into
oscilloscope. 3. Trigger on coincidence between discriminator
signals. 4. For each trigger, acquire digitized waveform data into
computer. 5. For pulse-height analysis, apply an R-C filter to the
preamplifier data (in software); peak-detect the result or sample
at a fixed time after the trigger. 6. Determine energy resolution
as a function of shaping time and, for fixed .tau., as a function
of sampling time.
TABLE-US-00003 TABLE 3 Readout Electronics Specifications (For Asic
And For Prototype) 1. Transimpedance preamplifier: bandwidth 60 to
80 MHz, noise <30 nA rms @ 5 pF input loading, transimpedance 40
k.OMEGA.. 2. Leading edge discriminator: threshold .apprxeq. 40 keV
(adjustable), time walk <2.5 ns for pulses >250 keV. 3. Time
pick-off: from discriminator on front or back APDs (whichever fires
first); double-pulse resolution <100 ns. 4. Pulse height
measurement: 2-pole shaper amplifier (150-200 ns peaking time) and
sample/hold circuit. 5. Sparse readout system, using external
analog-to-digital converter; deadtime can be less than 300 ns
(depending on ADC speed) 6. Power dissipation: <20 mW/channel
for preamplifier and discriminator. 7. Also on the detector board,
external to ASIC: coincidence logic, delay tuning, list-mode data
buffering FIFO, LVDS data interface, count rate monitoring scalers,
local HV regulator (1,900 V input, 1,600-1,800 V output, 50 mV
stability), APD gain monitoring, temperature monitoring.
TABLE-US-00004 TABLE 4 LSO/APDP PET Imaging ASIC Preliminary
Specifications Input Transimpedance 40 k.OMEGA. Stage -3 dB
bandwidth 80 MHz Linear input signal range 0 to -20 .mu.A Overload
recovery time <500 ns (from -600 .mu.A, 10 ns pulse) Total
wideband output noise, <25 nA rms input-referred (with 5 pF
loading on input, no APD) Slew rate, input-referred >4 kA/s
Linearity (overall for amplitude 2% measurement, LSO pulse shape
only; direct pulses may violate this) Discrim- Sensitivity
(includes hysteresis) <15 mV inator Propagation delay dispersion
<1.5 ns (10 mV to 500 mV overdrive) Propagation delay <8 ns
Shaper Shaper filter time constant 100 ns General Power dissipation
<20 mW/channel Channel count 16-128
[0096] A prototype of the transimpedance preamplifier and the
discriminator has been constructed and used for measurements with
an LSO crystal and a single channel APD FIG. 11. The prototype is
implemented in bipolar technology, which has the advantage of high
speed/power efficiency; the much lower current noise capability of
CMOS is not important here, relative to the APD current noise
level. Furthermore, bipolar technology allows for easy prototyping
with discrete components. The input stage is a current amplifier
with a gain of 45. This is followed by a further gain stage and
then the leading edge level crossing discriminator 126, implemented
as a fully differential ECL-type circuit. The input referred
wideband noise is 25 nA rms (a signal to noise ratio of 160 for the
511 keV photopeak). The bandwidth is 67 MHz. The amplifier reported
here is similar to other designs for photodiode readout for
fiberoptic data communications and for wire chamber readout.
Transimpedance amplifiers used before specifically optimized for
APD PET applications, although with a 38 mW power dissipation, and
a bandwidth of 22 MHz, which would compromise the ability to get
sufficient timing resolution with a simple leading edge level
crossing discriminator. Therefore, the ASIC can be also designed
using small width CMOS or BiCMOS processes to achieve such high
speed and large bandwidth necessary for this application.
[0097] The crystal identification was studied with the same setup,
and is shown in FIG. 25, which is showing a graph of crystal
identification measured with an array of 2.times.2.times.10
mm.sup.3 LSO crystals coupled to the APD array and read out by the
RENA (Readout Electronics for Nuclear Application) signal processor
developed at NOVA R&D, Inc. The image on the left is a flood
source histogram of the array, and the plot on the right is a
profile across one row of the crystal array. All 16 crystals are
very well separated, the average peak-to-valley ratio is over 100:1
as measured from a profile across one crystal row in the image.
Although events were collected with a relatively high hardware
threshold, the results still demonstrate the minimal inter-channel
crosstalk and excellent crystal identification.
[0098] The flood source image acquired from the signal multiplexing
board based on the HQV802-M preamplifiers with the same crystal
array is shown in FIG. 26, which is showing a graph of crystal
identification measured with an array of 2.times.2.times.10
mm.sup.3 LSO crystals coupled to the APD array and read out by two
HQV802-M hybrids with multiplexed readout. The image on the left is
a flood source histogram of the array, and the plot on the right is
a profile across one row of the crystal array. This image is much
poorer than the previous one in FIG. 25. This is mainly due to the
fact that noise from all APD channels was added together in the
board to determine which crystal was hit. The lower energy
threshold (.about.150 keV) used in this experiment might contribute
to signal spreading and background in the flood histogram as well.
Nevertheless, all crystals are clearly identified with an average
peak-to-valley ratio about 12:1. FIG. 25 shows the superior images
obtained by the RENA chip based readout system. Therefore, this
measurement clearly shows that in order to take full advantage of
the APD array, integrated electronics with independent signal
processing must be used as seen here for the RENA results. In
systems with large numbers of channels, this will require
particular attention to cost considerations and issues related to
power dissipation.
[0099] Input amplifier is the most important part of the ASIC.
Input stage of the ASIC must be carefully designed to match the
characteristics of the APD in order to achieve minimum noise. Since
the primary objective is to maximize timing performance versus
power dissipation, the input stage will be based on an n-channel
MOSFET, which has higher transconductance and therefore lower
voltage noise due to channel thermal fluctuations. The greater 1/f
noise of an re-channel versus p-channel MOSFET will not be
detrimental to timing because it will be dominated by the APD
leakage current shot noise within the passband of the fast shaper
used in the timing signal path. The size of the input transistor is
carefully chosen, based on the expected APD (4 pF) and stray
capacitances, to optimize the voltage noise contribution to the
overall noise--too large a transistor will increase the equivalent
noise charge of the amplifier because of excessive capacitance,
whereas too small a transistor will have insufficient
transconductance and so a large voltage noise and therefore the
amplifier will have a large equivalent noise charge. Consideration
will also be given to the possibility of operating the input
transistor at the edge of weak inversion, where the
transconductance versus bias current is maximized. To achieve the
required timing spec, the input amplifier must also have a fast
risetime and good linearity, and the fast shaper amplifier must
also have good linearity. The preamplifier requirements for this
APD array are similar to (although not identical with) those which
have been reported. These authors have reported excellent timing
performance (although their chips do not yet integrate a timing
discriminator), and therefore it is certainly feasible to design
the required preamp in CMOS technology. Important to that will be
laboratory measurements, with a very wideband amplifier, of the APD
signal shape in combination with LSO, a positron source, and a fast
scintillator and photomultiplier tube. The optimum shaping time
depends on the variations in the APD charge collection, which this
measurement will determine, and also on the relative magnitude of
the APD leakage current shot noise and the input amplifier voltage
noise. The optimum shaping time (for timing measurement) is on the
order of 5 to 10 ns.
[0100] A selectable gain circuit will be included in the signal
path, so that different APD gains can be accommodated and so that
we can study the optimal operating point of this APD for LSO based
PET imaging.
[0101] Discriminator is also an important part of the ASIC. It is
well known that the use of a simple leading edge discriminator for
precise timing measurement in the presence of amplitude variations
will not lead to optimal results, simply because the time required
for the signal to rise from zero to the threshold level will depend
on the pulse amplitude and risetime. Several discriminator
architectures are available to address this issue. The most widely
known is of course the constant fraction discriminator (CFD), which
is a leading edge discriminator with a non-constant threshold which
looks forward in time, being set ideally to a fixed fraction of the
overall pulse height. This is traditionally implemented with a
delay line chosen carefully to match the input pulse
characteristics. Lumped-element filter circuits can be used as an
alternative, however, and are attractive because it is very
difficult to integrate a high-quality delay line in a monolithic
circuit, especially a modern submicron CMOS process which is
intended for digital applications. Several authors have recently
reported ASICs incorporating a CFD. Timing resolution of the order
of 1 ns FWHM is possible with these monolithic CMOS CFDs. For
optimum response, however, it is important to carefully design the
delay or filter network to match the characteristics of the input
pulse, which could lead to complications.
[0102] A second architecture for alleviating time walk effects is
the zero-crossing discriminator, which differentiates the input
signal and looks for a zero-crossing, which would be associated
with the peak of the original signal. The time of this
zero-crossing will be independent of the amplitude of the input
signal, if everything is linear and there is no slew rate limiting.
Recent implementations of zero-crossing discriminators show
excellent timing resolution and can be achieved in standard CMOS
technology. Both the CFD and the zero-crossing discriminator can
still suffer time walk due to non-ideal comparator response (slew
rate and overdrive dependence), but at least in the zero-crossing
case even this error can be cancelled by clever use of an analog
division circuit; performance as good as 0.2 ns FWHM is achievable
in standard CMOS.
[0103] The third approach to time walk is to use a leading edge
discriminator, or some combination of leading edge discriminators,
accept that time walk exists in its output, and compensate for it
either by sending the comparator output through a variable delay
device (analog or digital), or by altering the input signal or
threshold, based on the input signal amplitude. In a sense, the
standard CFD circuit falls into this category, but there are many
other possibilities. The extrapolated leading edge discriminator
uses two leading edge discriminators and delays the output of the
first one by a constant minus the time difference between the two,
to extrapolate back to zero threshold and so zero time walk.
Another approach uses a low-resolution flash A/D to control delays
applied to either the input or output of a leading edge
discriminator. That is rather complex for the present application,
however, using a digital delay with direct analog control from the
discriminator input signal is a viable alternative.
[0104] Many of these discriminators circuits can be used for the
LSO/APD readout chip for PET imaging. It is important to select the
best architecture, which minimizes circuit area and complexity,
power dissipation, sensitivity to process variations, and
temperature coefficient, while meeting the required time resolution
specification. It is very likely that a compensated leading edge
discriminator will provide the most efficient and robust
implementation with sufficient performance. Leading edge
discriminators 105, 107 have been already used successfully and the
results are presented here.
[0105] The circuit was tested with a single-pixel APD 121, type RMD
S0223, coupled with Bicron BC-630 optical grease to a
2.times.2.times.10 mm.sup.3 LSO crystal 120 wrapped in reflective
white teflon tape (FIG. 11). Except for the number of pixels, the
specifications of this APD type exactly match those of the
4.times.4 pixel array for which the readout ASIC will be developed.
The detector was irradiated by a .sup.22Na positron source. To
detect gamma-gamma coincidences from the positron annihilation, we
used a second LSO crystal 123 coupled to a 3/8'' photomultiplier
tube (Hamamatsu R1635) 124. To make the alignment of the
radioactive source and the two detectors less critical, this LSO
crystal was irradiated through one of its 2.times.10 mm.sup.2 wide
faces. The output signals from the APD preamplifier 126, the
discriminator 128, and the PMT were recorded on a high-bandwidth,
high-sample rate digital storage oscilloscope 127 (Tektronix TDS
7104, bandwidth 1 GHz, sample rate up to 10 GS/s, depending on the
number of traces recorded) and transferred to a desktop computer
via a GPIB connection. A typical APD signal(s) is shown in FIG. 12
and FIG. 17. The amplitude of this pulse is typical of the 511 keV
photopeak. Calibration of the current on the y axis is based on the
measured transimpedance of the amplifier circuit. The pulse fits to
e.sup.-(t-t.sup.0.sup.)/.tau..sup.1-e.sup.-(t-t.sup.0.sup.)/.tau..sup.2,
with .tau..sub.1=35 ns and .tau..sub.1=10.6 ns. The 10%-90%
risetime of such a pulse is 10.1 ns. This indicates that a
preamplifier bandwidth in the 100 to 150 MHz region is sufficient;
beyond this range the wideband noise will be increasing faster than
the signal slew rate, and timing resolution will suffer. Note also
that with a 10 to 15 ns risetime, if the threshold can be set at
10% or so, there can be no more than 1 to 1.5 ns time walk for a
simple leading edge discriminator, for signals reasonably near the
photopeak.
[0106] In order to verify the amplitude determination, we acquired
data for the 662 keV gamma line from .sup.137Cs. The spectrum is
shown in FIG. 16. When compared with sodium data FIG. 15 taken
under identical conditions, the two photopeak positions agree to
within 3% after correcting for the actual photon energies, less
than the width of either peak.
[0107] The coincidence time resolution was determined by recording
the APD and PMT analog signals in coincidence (with a wide enough
coincidence window to avoid affecting the subsequent analysis),
setting a threshold safely above the noise level for each of the
two signals, and determining the time at which each signal first
crossed the threshold. This algorithm simulates a simple
leading-edge discriminator. The distribution of time differences
between the two signals is plotted in FIG. 13. It has a width of
3.1 ns FWHM; corrections for pulse-height dependent time walk,
which have not been applied to the data shown in FIG. 13, reduce
this width by only a small amount. This suggests that more
complicated discriminator schemes, such as dual-level or constant
fraction discriminators may not significantly improve the timing
resolution of this circuit. Moreover, the image reconstruction will
have to involve a photon energy cut to reduce the background due to
gamma scattering in the sample, so those events which are most
affected by time walk will not be used for imaging purposes anyway.
Therefore, the coincidence window can be adjusted to reflect the
(narrower) width of the timing distribution for events that pass
the pulse height cut. It should be noted that we did not see any
significantly different result when we used the APD discriminator
output signal instead of the analog signal to determine the timing.
The discriminator worked properly.
[0108] The emphasis with prototype circuit B (FIG. 7) is on
achieving the required timing and energy resolution under the
constraint of low power dissipation. The bandwidth of the
preamplifier 63 and the propagation delay of the discriminator
(FIG. 8) are both considerably reduced from the levels of prototype
circuit A (FIG. 6). As a consequence, circuit B is perhaps less
suitable for detailed laboratory studies of the APD current
waveform in response to LSO scintillation light. This circuit was a
laboratory prototype for the functionality and specifications
considered for the ASIC. Of course, the ASIC includes further
circuitry besides the preamplifier and discriminator--there will be
shapers and sample/hold circuits and an analog output for the pulse
height measurement, and there will be some of the coincidence logic
and the readout control logic, and there may also be A/D
converter(s) and/or Constant fraction discriminators.
[0109] Schematic diagrams of prototype circuit B are shown in FIG.
7 and FIG. 8. It is implemented in a bipolar technology, owing to
the widespread availability of low noise high speed bipolar
transistors for wireless communications. The PET APD readout ASIC
may be developed either in bipolar, BiCMOS, or CMOS technology--all
should be capable of meeting the specifications, although there
will be specific advantages in terms of power efficiency, speed,
stability, and cost for these different technologies which will be
evaluated. All transistors are Philips BFT25A, chosen for its low
noise and for its high f.sub.T (over 3.5 GHz) and .beta. (about 75)
at a low (500 .mu.A) DC collector current, which makes it
especially suitable for low power, high speed amplifiers. The same
parameters are also crucial for a low current noise amplifier,
since the current noise at low to medium frequencies is dominated
by the base current shot noise, which in this amplifier is a low
1.72 pA/ Hz (the base bias current is 9.26 .mu.A).
[0110] The transimpedance amplifier 63 is a two-stage design; the
first stage provides an output signal current which is proportional
to the APD signal current. In the second stage this current is
applied to a load resistor, the resulting voltage is buffered, and
then it goes to a further gain stage 65 which has a differential
output; in this stage is also a baseline restorer, which also
serves for the introduction of an intentional offset voltage 70 to
lower the positive output (A+ on the schematic) 71 below the
negative output (A-) 71. This offset voltage is the discriminator
threshold level--the differential-input, differential-output
discriminator, which follows has its nominal threshold at zero
volts. A fully DC coupled circuit, replacing the baseline restorer
with a suitable bias voltage circuit, would also be possible in the
ASIC (and perhaps would be preferred for operation at high count
rates), but it was felt that for the discrete component prototype
this could not be successfully implemented, owing to device
matching problems and temperature differentials on the printed
circuit board. This gives another reason for preferring ASIC over a
system based on discrete components.
[0111] The noise of the transimpedance amplifier 63 of circuit B is
significantly lower than that of circuit A 43. Partially this is
due to the lower bandwidth, but in addition this circuit is more
optimized for the relatively high source capacitance of the APD
(and of the interconnect which will be necessary in a realistic
multichannel system), compared with the capacitance of a photodiode
for fiberoptic data receiver applications. Of course in an ASIC the
noise can be reduced further since the parasitic capacitances are
smaller and the transistor geometry can be tuned. With 5 pF source
capacitance, we measured 25 nA rms (a signal to noise ratio of 160
at typical signal levels). SPICE simulations indicate 33.7 nA rms.
The major noise contributions at low to medium frequencies are from
the input transistor base shot noise (1.72 pA/ Hz) and the feedback
resistor current noise (1.17 pA/ Hz), adding to 2.08 pA/ Hz; at
high frequencies the major noise contributions are from the input
transistor base resistance and collector shot noise. The effect of
the latter is dependent on the source capacitance. In addition to
these amplifier noise sources, there is of course the APD noise
current, about 3 to 5 pA/ Hz.
[0112] The discriminator of prototype circuit B is a low power,
reduced signal swing version of traditional differential ECL
buffers such as the MC100EL16 used in prototype circuit A. The
input signal swing is small, and the common-mode level is precisely
controlled; for this reason a more general purpose comparator input
stage would be an unnecessary effort. The small signal swing
internally and at the output helps to minimize propagation delay
dependence on the input signal slew rate and overdrive, and our
.sup.22Na coincidence measurements show that such propagation delay
variations are indeed under control.
[0113] The transimpedance amplifier 63, 71 outputs and also the
discriminator outputs 91 of prototype circuit B are buffered for
transmission to the oscilloscope with four AD8009 wideband op-amps
in an overall unity gain configuration. The large signal -3 dB
bandwidth of these buffers (2 V p-p, greater than our maximum
signal swing here) is 440 MHz, so they do not limit our
measurements. Similarly, the noise contribution of these buffers to
the transimpedance amplifier output signals is around 1.8 nA rms
(input referred), so again they are not limiting the measurements.
Power dissipation for prototype circuit B is 22.2 mW (measured); as
with circuit A this figure is for the indicated circuit only and
excludes the test point buffers.
[0114] A photograph of the prototype circuit B (FIG. 7 and FIG. 8)
is shown in FIG. 9. Note the APD and LSO crystal (wrapped in white
tape) installed on the 8-pin DIP header near the upper left.
[0115] Performance of the circuit was investigated using the same
setup as the one described for circuit A (FIG. 6). The energy
resolution and the timing resolution between the APD and PMT were
measured. The APD preamplifier signals were filtered with an
RC.sup.2 filter with a time constant of 50 ns. A sample .sup.22Na
source spectrum obtained with circuit B is shown in FIG. 15.
Compared to circuit B results, the width of the photopeak is
slightly improved, to 15.8% FWHM. Such an improvement is to be
expected from the lower noise level of circuit B.
[0116] For an alternative assessment of the energy resolution, we
used the oscilloscope's built-in mathematical capabilities to
integrate the APD signal over a period of 240 ns, starting 40 ns
before the trigger point. The resulting photopeak histogram had a
width of 13.6% FWHM.
[0117] The timing spectra is acquired with circuit B. Due to the
low preamplifier noise, however, it was possible to set the
discriminator threshold low enough that signals which were high
enough to be relevant for PET imaging were not significantly
affected by this time walk. With a pulse height cut corresponding
to 350 keV photon energy, a typical value for PET applications, a
timing resolution of 1.1 ns FWHM and 2.2 ns full width at one tenth
the maximum (FWTM) was obtained. The spectrum shown in FIG. 14 uses
a more conservative cut value of 300 keV and yields a resolution of
1.5 ns FWHM, 2.7 ns FWTM.
[0118] In designing the readout electronics for an LSO/APD based
PET system, the main consideration is to obtain high-resolution
coincidence timing. This is required to achieve the combination of
high singles count rates and low accidental coincidence rates that
is needed for high-contrast PET imaging. Spurious coincidences
create an image background by yielding reconstructed photon
directions that have no correlation with the actual source
distribution. By a rough estimate, we expect a singles rate of
about 1.5 million counts/s in a PET system consisting of an 18 cm
diameter, 2 cm axial length detector ring (2000 pixels) for a 10
mCi injection. Assuming that these counts are evenly distributed
over all pixels and that coincidences are formed between each
module and a 120.degree. ring section across from it (for a field
of view of half the ring diameter), this leads to an accidental
coincidence rate of 750 counts/s, per nanosecond timing resolution.
This has to be compared to an estimated true coincidence rate of
10-15 kcounts/s. Work discussed here has demonstrated that a
coincidence timing resolution of better than 2 ns FWHM is
achievable for coincidences between an APD and a (significantly
faster) PMT. By taking coincidences between two APDs instead, the
width is expected to increase by no more than 50%. Based on that,
and the system requirements outlined above, a design target for the
ASIC a coincidence timing resolution <3 ns FWHM measured between
two APD channels with a positron annihilation source and
2.times.2.times.20 mm.sup.3 LSO crystals is set.
[0119] To achieve the goal of high-resolution timing, we will
continue to focus our chip design efforts on two main areas, the
preamplifier and the timing discriminator. As discussed above, a
low-noise, fast preamplifier will help improve timing resolution in
two ways, by reducing the (noise-induced) amplitude variations that
invariably translate into timing fluctuations, and by minimizing
(by virtue of a fast signal rise time) the direct slope-induced
time walk, for instance in a leading edge level crossing
discriminator. Based on the results, a leading edge design may be
sufficient for obtaining good timing resolution; this will
considerably simplify the ASIC design and help us to keep the
chip's power dissipation low.
[0120] In order to make a practical large-scale high-resolution
coincidence system, it is also necessary to address the issue of
controlling signal-independent systematic variations from channel
to channel in propagation delay (and therefore also in signal
baseline and discriminator threshold, at least with leading edge
discrimination). To do so, we will investigate methods to minimize
the variations and more importantly their temperature coefficients,
and also will implement delay tuning circuitry on a
channel-by-channel basis to line up the discriminator outputs in
the coincidence logic.
[0121] Energy measurement of each pulse is also important, in order
to reject the background from scatter within the imaged object or
other material. However, only a modest energy resolution is already
sufficient for this purpose. As discussed above, the work with this
APD has yielded an energy resolution of approximately 15% FWHM at
511 keV, and at most minor changes to this value in the ASIC, due,
for example, to further optimization of the shaper parameters or
unexpected noise pickup is expected.
[0122] For maximum sensitivity, and hence minimum total dose to the
patient, at high event rates, it is very important that the front
end electronics and readout system impose the minimum practical
dead time due to event processing. We will take this into account
in the design of the readout circuitry, and expect to meet a
deadtime specification of no more than 200 to 300 ns per hit, and
furthermore that this deadtime will only apply to the channels
which are hit, not to an entire APD module or readout group.
Therefore a special pipeline technology can be implemented on the
ASIC to keep the readout working while the data is transferred to
the data acquisition computer. It is important to achieve fast data
rate, therefore, the ASIC is designed to be able to use innovative
techniques to achieve this. Other innovative features, besides the
pipeline data readout system, are; to have on board ADC circuitry
where no analog output is needed from the chip; fast analog
circuitry; fast discriminator circuits such as constant fraction
discriminator; large gain-bandwidth product; elimination of
sample-and-hold by using flash A/D converter combined to the
pipeline readout; on board coincidence detection circuit; on board
elimination or discrimination of events where only an APD on one
end of the LSO crystal detector produces a signal not in
coincidence with the APD on the other end; on board DOI
determination; a channel to channel time difference measuring
system using the signals V.sub.U 2, 14 and V.sub.V 3, 15 and a
completely digital circuit where the input signals are shaped or
without shaping are digitized immediately and time tagged and send
outside the chip to a very fast computer for realtime and/or post
processing. The chip may also have onboard adjustments for
amplifier gains and offsets, and discriminator or comparator
thresholds.
[0123] In designing the readout modules for the APD arrays, special
care will be taken to minimize the amount of inactive material in
front of and between the LSO crystals. This will help minimize
blurring caused by photon scattering in these materials and
maximize the detection efficiency of the system. Obviously, it will
not be possible to eliminate all material from the inside of the
ring formed by the detector arrays; most notably, the APDs and
readout chips required for DOI measurements will have to be right
on the inner surface. However, by moving all support circuitry
except the most essential bypass capacitors to the outside edge of
the ring, will reduce the impact of these components to a
minimum.
[0124] Another important consideration in designing a PET detector
system with a large number of elements is the design of the overall
trigger logic. We plan to implement a decentralized system, in
which each 16-element or 64-element detector module with ASICs on
each end sends its timing discriminator signals onto a bus that is
connected to as many modules on the opposite side of the ring as
are needed to achieve the desired field of view. Each module
autonomously determines whether there is a coincidence between its
own timing signals and any of the signals coming in from across the
ring; if this is the case, the module then initiates the readout of
its relevant timing and pulse height data. (A valid event then
requires that both of the modules involved recognize the
coincidence). Compared to a centralized design, this approach
considerably simplifies the trigger logic (no need for a
complicated coincidence matrix, nor to feed the coincidence outputs
back to the affected modules in order to initiate readout), albeit
at the slight expense of having to have trigger circuitry on each
module. In another embodiment, a processing system external to the
detector modules receive all the trigger and/or pulse height and/or
DOI information with channel addresses which have a hit and
determines which data to read, and/or calculates the gamma ray
directions or the event chord and sends processed data to the
imaging computer. In another embodiment all these functions are
carried out on the ASICs and the processed data goes to the imaging
computer. In another embodiment, microprocessors are placed on the
side of the ASICs which carry put the processing of the data and
the processed data goes to the imaging computer. In another
embodiment, chips send out raw digitized data very fast to a fast
data acquisition/analysis/imaging computer or computers, which
process the raw data and produce an image.
* * * * *