U.S. patent application number 13/014657 was filed with the patent office on 2011-09-01 for cascaded filter based noise and interference canceller.
This patent application is currently assigned to Intersil Americas Inc.. Invention is credited to Wei Chen, Wilhelm Steffen Hahn.
Application Number | 20110212692 13/014657 |
Document ID | / |
Family ID | 44505253 |
Filed Date | 2011-09-01 |
United States Patent
Application |
20110212692 |
Kind Code |
A1 |
Hahn; Wilhelm Steffen ; et
al. |
September 1, 2011 |
Cascaded Filter Based Noise and Interference Canceller
Abstract
Signals propagating from an aggressor communication channel can
cause detrimental interference in a victim communication channel. A
high input power cascaded filter canceller ("HIPCF") can obtain a
sample of the signal that imposes interference and process the
sampled signal to generate an interference compensation signal
that, when applied to the victim communication channel, cancels or
suppresses the detrimental interference. The HIPCF canceller can
include two or more cascaded filters, such as band-pass filters,
that block or reduce amplitude of a signal outside the
communication frequency band of a victim receiver. The HIPCF
canceller also includes an I/Q modulator that receives the filtered
signal and generates the interference compensation signal by
adjusting or updating one or more aspects of the filtered signal,
such as a gain, phase, or delay, based upon feedback from the
victim receiver or a power detector and algorithms executed by the
controller.
Inventors: |
Hahn; Wilhelm Steffen; (Los
Altos, CA) ; Chen; Wei; (Newark, CA) |
Assignee: |
Intersil Americas Inc.
Milpitas
CA
|
Family ID: |
44505253 |
Appl. No.: |
13/014657 |
Filed: |
January 26, 2011 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
61308697 |
Feb 26, 2010 |
|
|
|
61375491 |
Aug 20, 2010 |
|
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Current U.S.
Class: |
455/63.1 |
Current CPC
Class: |
H04B 1/52 20130101; H04B
1/44 20130101; H04B 1/525 20130101; H04B 15/00 20130101 |
Class at
Publication: |
455/63.1 |
International
Class: |
H04B 15/00 20060101
H04B015/00 |
Claims
1. A system for canceling interference on a first communication
path associated with a transmission on a second communication path,
comprising: an input operable to couple to the second communication
path to obtain a sample of the transmission; an output operable to
couple an interference compensation signal to the first
communication path; and a circuit disposed between the input and
the output, comprising: a plurality of filters arranged in a
cascaded manner for receiving the sample and outputting a filtered
signal; and a modulator coupled to an output of the cascaded
filters and operable to produce the interference compensation
signal in response to adjusting at least one of amplitude, phase,
and delay of the filtered signal.
2. The system of claim 1, wherein the system is an integrated
circuit.
3. The system of claim 1, wherein the first communication path and
the second communication path are disposed within a cellular
telephone system comprising the system.
4. The system of claim 1, wherein the each of the plurality of
filters comprises a band-pass filter.
5. The system of claim 1, wherein the plurality of filters
substantially reduce amplitude of components of the sample outside
of a frequency band associated with a receiver coupled to the first
communication path.
6. The system of claim 1, wherein the transmission is generated by
a transmitter comprising a power amplifier and wherein the input is
operable to couple to an output of the power amplifier.
7. The system of claim 1, wherein the input comprises at least one
of (a) a fixed capacitor and (b) a voltage-controlled variable
capacitor.
8. The system of claim 1, wherein the plurality of filters
comprises a first band-pass filter that receives the sample from
the input, the first band-pass filter comprising a high-Q inductor
and at least one switchable capacitor.
9. The system of claim 8, wherein the plurality of filters further
comprises a second band-pass filter coupled to an output of the
first band-pass filter, the second band-pass filter comprising a
high-Q inductor and at least one switchable capacitor, and wherein
the second band-pass filter couples to the output of the first
band-pass filter via a low noise amplifier.
10. The system of claim 9, wherein the plurality of filters further
comprises a third band-pass filter coupled to an output of the
second band-pass filter, the third band-pass filter comprising a
high-Q inductor and at least one switchable capacitor, and wherein
the third band-pass filter couples to the output of the second
band-pass filter via a variable gain amplifier.
11. The system of claim 1, wherein each of the plurality of filters
comprise a band-pass filter, and wherein the circuit further
comprises a controller logically coupled to each band-pass filter,
the controller being operable to adjust a resonant frequency of
each band-pass filter.
12. The system of claim 11, wherein each band-pass filter comprises
at least one switchable capacitor comprising a plurality of
selectable capacitors, and wherein the controller adjusts the
resonant frequency of one of the band-pass filters by selecting at
least one of the selectable capacitors of the one band-pass
filter.
13. The system of claim 11, wherein the controller adjusts the
resonant frequency of each band-pass filter in response to a change
in frequency associated with a receiver coupled to the first
communication path.
14. The system of claim 13, wherein the controller adjusts the
resonant frequency of each band-pass filter to match the frequency
associated with the receiver.
15. The system of claim 1, wherein the transmission is generated by
a transmitter and the first communication path is coupled to a
receiver.
16. The system of claim 15, wherein the transmitter comprises a
mobile telephone transmitter and the receiver comprises one of a
mobile TV tuner and a GPS receiver.
17. The system of claim 1, wherein the modulator comprises an I/Q
modulator and at least a portion of the components of each of the
plurality of band-pass filters and the I/Q modulator are fabricated
on an integrated circuit.
18. A method for suppressing interference on a first communication
path associated with a transmission on a second communication path,
the method comprising: obtaining a sample of the transmission;
filtering the sample by a plurality of cascaded filters; producing
an interference compensation signal by adjusting at least one of
amplitude, phase, and delay of the filtered signal; applying the
interference compensation signal to the first communication path;
and suppressing interference on the first communication path in
response to applying the interference compensation signal to the
first communication path.
19. The method of claim 18, wherein the interference compensation
signal comprises an amplitude substantially the same as amplitude
of the interference and a phase shifted approximately 180 degrees
with respect to the interference.
20. The method of claim 18, wherein the plurality of cascaded
filters reduce amplitude of a portion of the sample having a
frequency outside a frequency band for a receiver electrically
connected to the first communication path.
21. The method of claim 18, wherein the filtering comprises:
filtering the sample by a first band-pass filter having a first
quality factor; and further filtering the filtered sample by a
second band-pass filter having a second quality factor; and further
filtering the filtered sample by a third band pass filter having a
third quality factor.
22. The method of claim 18, further comprising: obtaining an
indication of an amount of interference imposed on a victim
receiver by the transmission; and adjusting the interference
compensation signal based on the amount of interference.
23. The method of claim 22, wherein the indication comprises a
receive signal quality indicator for the victim receiver.
24. The method of claim 18, further comprising: receiving a receive
signal quality indicator for a victim receiver; and adjusting one
or more settings of one or more of the plurality of filters and one
or more settings of an I/Q modulator that adjusts at least one of
amplitude, phase, and delay of the filtered signal based on the
receive quality indicator.
25. The method of claim 18, further comprising: detecting a change
in frequency for a victim receiver electrically connected to the
first communication path; and adjusting a parameter of each of the
cascaded filters based on the change in frequency.
26. The method of claim 25, wherein each of the cascaded filters
comprises a band-pass filter and wherein the parameter comprises a
resonant frequency.
27. The method of claim 18, further comprising adjusting at least
one of (a) a resonant frequency, (b) a resonance gain, and (c) a
quality factor of each of the plurality of filters in response to a
change in one or more environmental conditions.
28. An interference compensation device for suppressing interfering
signals introduced onto a receive path of a victim receiver by a
transmission from a transmitter, comprising: a first input for
receiving an indication of interference imposed on the victim
receiver; a first band pass filter comprising a first quality
factor for receiving and filtering a sample of the transmission; a
second band pass filter comprising a second quality factor and
disposed along an output signal path of the first band pass filter
for further filtering the sample; and an I/Q modulator disposed
along an output of the second band pass filter and operable to
produce an interference compensation signal in response to
adjusting at least one of amplitude, phase, and delay of the
filtered sample based on the indication, the interference
compensation signal operable to suppress the interfering
signals.
29. The interference compensation device of claim 28, wherein at
least a portion of the first band pass filter, at least a portion
of the second band pass filter, and the I/Q modulator are
fabricated in an integrated circuit.
30. The interference compensation device of claim 28, wherein the
first band pass filter and the second band pass filter each
comprise adjustable resonant frequencies and wherein the resonant
frequencies are adjusted based on a frequency of the victim
receiver.
31. The interference compensation device of claim 28, further
comprising a controller operable to adjust a resonant frequency of
the first band pass filter and a resonant frequency of the second
band pass filter.
32. The interference compensation device of claim 31, wherein the
controller is further operable to receive the indication and to
adjust in-phase and quadrature settings of the I/Q modulator based
on the indication.
33. The interference compensation device of claim 31, wherein the
controller is operable to control the first quality factor and the
second quality factor based upon an out of band signal and blocker
levels measured by a power detector connected to the signal
path.
34. The interference compensation device of claim 31, wherein the
controller is operable to control the first quality factor and the
second quality factor based upon a signal quality indicator
feedback from the victim receiver.
35. The interference compensation device of claim 28, further
comprising a third band pass filter disposed along a signal path
between the first band pass filter and the second band pass filter,
the third band pass filter comprising a third quality factor.
36. The interference compensation device of claim 35, further
comprising at least one amplifier disposed along the output signal
path of the first band pass filter, at least one variable gain
amplifier disposed along the output signal path of the second band
pass filter, and at least one buffer or amplifier disposed along
the output signal path of the third band pass filter.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This patent application claims to the benefit of U.S.
Provisional Patent Application No. 61/308,697, entitled "High Power
Cascaded Filter Based Noise Canceller" and filed Feb. 26, 2010.
This application also claims to the benefit of U.S. Provisional
Patent Application No. 61/375,491, entitled "Methods and Systems
for Noise and Interference Cancellation" and filed Aug. 20, 2010.
This application is related to U.S. patent application Ser. No.
______, [Attorney Docket No. 07982.105115], entitled "Methods and
Systems for Noise and Interference Cancellation," filed on the same
date as this application. The entire contents of each of the
foregoing priority and related applications are hereby fully
incorporated herein by reference.
BRIEF DESCRIPTION OF THE DRAWINGS
[0002] FIG. 1 is a functional block diagram of a communication
system, in accordance with certain exemplary embodiments.
[0003] FIG. 2 is a block schematic diagram of a high input power
cascaded filter (HIPCF) canceller, in accordance with certain
exemplary embodiments.
[0004] FIG. 3 is a block schematic diagram of certain components of
the HIPCF canceller of FIG. 2, in accordance with certain exemplary
embodiments.
[0005] FIG. 4 depicts a spectral diagram of signals received at a
victim receiver antenna, in accordance with certain exemplary
embodiments.
[0006] FIG. 5 depicts a spectral diagram of signals received at the
input of a victim receiver after cancellation of in-band unwanted
spectral components by an HIPCF canceller, in accordance with
certain exemplary embodiments.
[0007] FIG. 6 is a block schematic diagram of a Q-enhanced
band-pass filter (Q-Enhanced-BPF), in accordance with certain
exemplary embodiments
[0008] FIG. 7 is a block schematic diagram illustrating additional
components of the HIPCF canceller of FIG. 2, in accordance with
certain exemplary embodiments.
[0009] FIG. 8 depicts a functional block diagram of a communication
system, in accordance with certain exemplary embodiments.
[0010] FIG. 9 depicts a lookup table, in accordance with certain
exemplary embodiments.
[0011] FIG. 10 is a flow chart depicting a method for calibrating
certain components of the HIPCF canceller of FIG. 2, in accordance
with certain exemplary embodiments.
[0012] FIG. 11 is a flow chart depicting a method for configuring
the filters of the HIPCF canceller of FIG. 2 for a desired center
frequency, in accordance with certain exemplary embodiments.
[0013] FIG. 12 is a flow chart depicting a method for calibrating
an input band-pass filter (Input-BPF) of the HIPCF canceller of
FIG. 2, in accordance with certain exemplary embodiments.
[0014] FIG. 13 is a flow chart depicting a method for calibrating a
low noise amplifier band-pass filter (LNA-BPF) of the HIPCF
canceller of FIG. 2, in accordance with certain exemplary
embodiments.
[0015] FIGS. 14A and 14B, collectively FIG. 14, depict a flow chart
of a method for calibrating a Q-Enhanced-BPF of the HIPCF canceller
of FIG. 2, in accordance with certain exemplary embodiments.
[0016] FIG. 15 is a flow chart depicting a method for calibrating
the Input-BPF of the HIPCF canceller of FIG. 2, in accordance with
certain exemplary embodiments.
[0017] FIG. 16 is a flow chart depicting a method for determining
switch settings for a given frequency, in accordance with certain
exemplary embodiments.
[0018] FIG. 17 depicts implementation layers of noise and/or
interference cancellation algorithms, in accordance with certain
exemplary embodiments.
[0019] FIG. 18 is a diagram depicting receiver sensitivity plotted
versus coupled power amplifier phase noise, in accordance with
certain exemplary embodiments.
[0020] FIG. 19 is a diagram depicting an output signal to noise
ratio (SNR) of a mobile TV tuner plotted versus a received mobile
TV tuner signal strength, in accordance with certain exemplary
embodiments.
[0021] FIG. 20 is a flow chart depicting a fast binary algorithm
for canceling noise or interference, in accordance with certain
exemplary embodiments.
[0022] FIG. 21 depicts a graph of in-phase (I) and quadrature (Q)
values adjusted using binary algorithms, in accordance with certain
exemplary embodiments.
[0023] FIG. 22 is a flow chart depicting a minstep algorithm for
canceling noise and/or interference, in accordance with certain
exemplary embodiments.
[0024] FIG. 23 depicts an I-Q plane with pseudorandom feedback
values, in accordance with certain exemplary embodiments.
[0025] FIG. 24 is a graph depicting a receive quality indicator
plotted versus I or Q values resulting from an implementation of a
dual slope algorithm (DSA), in accordance with certain exemplary
embodiments.
[0026] FIG. 25 is a flow chart depicting a DSA for canceling noise
and/or interference, in accordance with certain exemplary
embodiments.
[0027] FIG. 26 is a graph depicting a receive quality indicator
plotted versus I or Q values resulting from an implementation of
the dual slope algorithm of FIG. 24, in accordance with certain
exemplary embodiments.
[0028] FIG. 27 is a flow chart depicting a track and search
algorithm (TSA) for canceling noise and/or interference, in
accordance with certain exemplary embodiments.
[0029] FIG. 28 is a graph depicting cancellation points along an
I-Q plane evaluated in an implementation of the TSA of FIG. 27, in
accordance with certain exemplary embodiments.
[0030] FIG. 29 is a flow chart depicting a method for finding a
preferred noise cancellation point for two noise cancellers
disposed in a communication system, in accordance with certain
exemplary embodiments.
[0031] FIG. 30 is a flow chart depicting a method for finding a
preferred noise cancellation point for two noise cancellers
disposed in a communication system, in accordance with certain
exemplary embodiments.
[0032] FIG. 31 is a flow chart depicting a method for finding a
preferred noise cancellation point for two noise cancellers
disposed in a communication system, in accordance with certain
exemplary embodiments.
[0033] Many aspects of the invention can be better understood with
reference to the above drawings. The drawings illustrate only
exemplary embodiments of the invention and are therefore not to be
considered limiting of its scope, as the invention may admit to
other equally effective embodiments. The elements and features
shown in the drawings are not necessarily to scale, emphasis
instead being placed upon clearly illustrating the principles of
exemplary embodiments of the present invention. Additionally,
certain dimensions may be exaggerated to help visually convey such
principles. In the drawings, reference numerals designate like or
corresponding, but not necessarily identical, elements.
DETAILED DESCRIPTION OF EXEMPLARY EMBODIMENTS
[0034] The present invention is directed to systems and methods for
compensating for signal interference occurring between two or more
communication channels or between two or more communication
elements in a communication system. Compensating for interference
can improve signal quality, enhance communication bandwidth or
information carrying capability, or improve receiver sensitivity. A
communication channel may comprise a transmission line, a printed
circuit board (PCB) trace, a flex circuit trace, an electrical
conductor, a waveguide, a bus, a communication antenna, a medium
that provides a signal path, or an active or passive circuit or
circuit element such as a filter, oscillator, diode, VCO, PLL,
amplifier, digital or mixed signal integrated circuit. Thus, a
channel can comprise a global system for mobile communications
(GSM) device, a processor, a detector, a source, a diode, an
inductor, an integrated circuit, a connector, a circuit trace, or a
digital signal processing (DSP) chip, to name only a few
possibilities.
[0035] Exemplary embodiments described herein can include a high
input power cascaded filter (HIPCF) noise and interference
canceling device. Exemplary HIPCF cancellers described herein can
support selectively canceling, correcting, addressing, or
compensating for interference, electromagnetic interference (EMI),
noise (e.g., phase noise, intermodulation products, and other
interfering noise), spurs, or other unwanted spectral components
associated with one or more communication paths of a communication
system, such as a high speed digital communication system in a
portable electronic device. For the purpose of this specification,
the term "high power" generally refers to signals having a power
ratio up to approximately +33 dBm (decibels relative to one
milliwatt) or more. For example, exemplary HIPCF cancellers
described herein can be coupled to the output of cellular telephone
power amplifiers having output power of this magnitude.
[0036] The HIPCF cancellers can obtain a sample of a communication
signal that imposes interference from a communication path of an
aggressor transmitting device and process the sampled signal to
produce an interference compensation signal. The HIPCF can deliver
the interference compensation signal into or onto a communication
path of a victim receiver that is a recipient of the interference,
to cancel, mitigate, suppress, or otherwise compensate for the
received interference.
[0037] Turning now to the drawings, in which like numerals indicate
like or corresponding (but not necessarily identical) elements
throughout the figures, exemplary embodiments of the invention are
described in detail. FIG. 1 is a functional block diagram of a
communication system 100, in accordance with certain exemplary
embodiments. Referring to FIG. 1, the communication system 100
includes a transmitter 105 that transmits electromagnetic signals
via a transmitting antenna 115. A transmit path 107, including one
or more electrical conductors, couples the transmitter 105 to the
transmitting antenna 115. In certain exemplary embodiments, the
transmitter 105 conveys data to a remote device using one or more
communications standards or methods, such as the Global System for
Mobile Communications (GSM), Code Division Multiple Access (CDMA),
Long Term Evolution (LTE), Wideband Code Division Multiple Access
(W-CDMA), Digital Cellular System (DCS), Personal Communication
Service (PCS), and Wireless Local Area Network (WLAN). One of
ordinary skill in the art having the benefit of the present
disclosure would appreciate that the communication system 100
described herein is not limited to the aforementioned communication
standards and methods, but instead can be used with many other
types of signal transmitting technologies.
[0038] Disposed along the transmit path 107 between the transmitter
105 and the transmitting antenna 115 is a power amplifier 110. The
power amplifier 110 adjusts the power level of the transmitter's
output signals prior to the signals being propagated by the antenna
115. When a power amplifier 110 adjusts the power level of a
signal, unwanted spectral components can be introduced onto the
signal. For example, the transmitter 105 can transmit signals
having a certain carrier frequency or a certain fundamental tone.
The power amplifier 110 can introduce intermodulation products
having a different frequency than that carrier frequency or
fundamental tone. Other components associated with the transmitter
105 also can cause noise or other unwanted spectral components to
be introduced onto the signal. For example, the transmitter 105 may
include a local oscillator and/or one or more up-conversion
mixer(s) that can cause unwanted spectral components, including
out-of-band noise (outside the frequency band of the transmitted
signal) sometimes referred to as out-of-band blockers, to be
introduced onto the signal.
[0039] The communication system 100 also includes a receiver 135
that receives signals via a receiving antenna 120 and a receive
path 133 that electrically couples the receiving antenna 120 to the
receiver 135. In certain exemplary embodiments, the receiver 135
receives signals within the same or a different frequency band than
that of the transmitter 105. For example, a mobile electronic
device, such as a mobile telephone, personal digital assistant
(PDA) or mobile computer, may include a transmitter 105 that
communicates via one of the communication protocols discussed above
and a receiver 135 that communicates in a different frequency band,
such as a mobile TV tuner, a Bluetooth receiver, a Worldwide
Interoperability for Microwave Access (WiMAX) receiver, or a Global
Positioning System (GPS) receiver. In the illustrated embodiment,
the receive path 133 includes an optional receive (RX) filter 140.
The optional receive filter 140 can include a band-pass filter or
other filter arrangement that allows communication signals received
by the antenna 120 within the frequency band of the receiver 135 to
pass to the receiver 135, while blocking signals outside the
frequency band of the receiver 135.
[0040] The frequency band of the receiver 135 may be near the
frequency band of the transmitter 105 such that phase noise or
other unwanted spectral components produced by the power amplifier
110 or another component disposed along the transmit path 107
degrades the sensitivity of the receiver 135. For example, the
communication system 100 may be embodied in a mobile device having
a CDMA, GSM or LTE transmitter 105 and a mobile TV tuner as
receiver 135. Certain types of CDMA and GSM transmitters 105
transmit signals within a frequency band of approximately 800 MHz
to 900 MHz and certain types of LTE transmitters 105 operate within
a frequency band of 698 MHz to 798 MHz. These transmitted signals
often include phase noise having a frequency between 450 MHz and
776 MHz, which falls within the receive band of some mobile TV
tuners and many other communication devices. If this in-band phase
noise is imposed onto the signal path of the receiver 135 (e.g.,
air coupled from the transmitting antenna 115 to the receiving
antenna 120), the phase noise can degrade the sensitivity of the
receiver 135. Generally, receive filters, such as receive filter
140, do not filter out in-band noise as the noise is within the
frequency band of the receiver 135 and thus, the pass-band of
receive filter 140. Therefore, the phase noise may pass through the
receive filter 140 and degrade the sensitivity of the receiver
135.
[0041] To prevent a degradation of the sensitivity of the receiver
135 caused by in-band noise (noise having a frequency within the
frequency band of the receiver 135) or nearby out-of-band noise
caused by the transmissions from the transmitting antenna 115, the
communication system 100 includes a HIPCF canceller 130. An input
of the HIPCF canceller 130 is coupled to the transmit path 107 at
the output of the power amplifier 110 by way of a sampling device
125. The sampling device 125 can include a capacitor, (e.g., a
sampling or tapping capacitor), a resistor, a coupler, a coil, a
transformer, a signal trace, or an antenna/detector. Sampling
devices having two ports or two terminals, such as a resistor,
capacitor, coil, transformer, or signal trace, can have a first
port electrically coupled to the transmit path 107 and a second
port electrically coupled to the input of the HIPCF canceller 130.
In an antenna/detector, having mostly one terminal, the second
terminal is formed by the electromagnetic field protruding from the
device, allowing for locating the device close to the transmit path
107.
[0042] In the illustrated embodiment, the sampling device 125 is
connected to the transmit path 107 at the output of the power
amplifier 110. The sampling device 125 obtains samples of the
signal ("sampled transmit signals") at the output of the power
amplifier 110 and provides the sampled transmit signals to the
HIPCF canceller 130. In certain exemplary embodiments, the sampling
device 125 may produce attenuation on the sampled transmit signals.
For example, the amplitude of the sampled transmit signal may be 20
dBc (decibels relative to carrier) lower than the signal at the
output of the power amplifier 110.
[0043] In certain exemplary embodiments, the sampling device 125
includes a voltage-controlled capacitor (varactor) for trimming
frequency dependent attenuation to a desired value and hence,
compensate for gain ripple. In one example, the sampling device 125
includes a voltage controlled varactor. The capacitance of the
varactor can be adjusted via a control voltage. This control
voltage can be generated by a controller 235 (FIG. 2) of the HIPCF
Canceller 130 and transmitted to the sampling device 125 via one or
more electrical conductors 127.
[0044] The HIPCF canceller 130 selectively suppresses or cancels
interfering signals (e.g., phase noise, intermodulation products,
unwanted spectral components, etc.) produced by the power amplifier
110 (or another component along the transmit path 107) having a
frequency within or near the receive frequency band of the receiver
135 that would otherwise interfere with the sensitivity of the
receiver 135. The HIPCF canceller 130 obtains samples of the
signals output by the power amplifier 110 and processes the sampled
transmit signals to produce an interference compensation signal
that, when applied to an input of the receiver 135, suppresses or
cancels the interfering signals. In certain exemplary embodiments,
the HIPCF canceller 130 tunes the interference compensation signal
using feedback, such as a receive signal quality indicator,
obtained from the receiver 135 via a feedback path 180 including
one or more electrical conductors. The exemplary HIPCF canceller
130 is described in further detail below in connection with FIGS.
2-31.
[0045] The interference compensation signal is applied to the
receive path 133 of the receiver 135 at a cancellation point 134.
In certain exemplary embodiments, the cancellation point 134 is
implemented by converging an electrical conductor of the receive
path 133 with an electrical conductor along the output path of the
HIPCF canceller 130 such that the electrical conductors make
electrical contact. For example, a flex circuit trace of the
receive path 134 may be connected to a flex circuit trace of the
HIPCF output. In certain exemplary embodiments, a component, such
as a coupler, a summation node, an adder, or another suitable
technology may be used to apply the interference compensation
signal to the receiver path 133 of the receiver 135.
[0046] The communication system 100 illustrated in FIG. 1 can
transmit electromagnetic signals having a frequency within a first
frequency range and receive electromagnetic signals having a
frequency within a second frequency range. The first frequency
range may be close to the second frequency range or even include
frequencies that overlap or are included in the second frequency
range. In operation, the transmitter 105 transmits signals along
the transmit path 107 to the power amplifier 110. The power
amplifier 110 adjusts the intensity of the signals received from
the transmitter 105 and outputs the intensity adjusted signal to
the transmitting antenna 115. The transmitting antenna 115
transmits the signals received from the power amplifier 110. A
portion of the signals transmitted by the transmitting antenna 115
is coupled to the receiving antenna 120 via air. If received by the
receiver 135, signals coupled onto the receiving antenna 120
originating from the transmitting antenna 105 may interfere with or
degrade the sensitivity of the receiver 135. For example, signals
transmitted by the transmitting antenna 115 having a frequency
within the frequency band or close to the frequency band of the
receiver 135 (e.g., intermodulation spectra appearing like phase
noise tails generated by the power amplifier 110) can degrade the
sensitivity of the receiver 135. To compensate for this
interference or sensitivity degradation, the HIPCF canceller 130
obtains samples of the signals output by the power amplifier 110
(via the sampling device 125) and processes the sampled transmit
signals to produce an interference compensation signal that, when
applied to an input of the receiver 135, compensates for the
interference imposed on the receiver 135 by the signals transmitted
by transmitting antenna 115.
[0047] FIG. 2 is a block schematic diagram of the HIPCF canceller
130 of FIG. 1, in accordance with certain exemplary embodiments.
The exemplary HIPCF canceller 130 includes a band-pass filter
(Input-BPF) 205 that receives signal samples from the sampling
device 125. In this exemplary embodiment, the Input-BPF 205
includes an inductor L1 and two switchable capacitors C1 and C2.
The resonant frequency of the Input-BPF 205 is tunable by adjusting
capacitance of one or both of the switchable capacitors C1 and C2.
The switchable capacitors C1 and C2 are described in further detail
below in connection with FIG. 3.
[0048] In certain exemplary embodiments, the inductor L1 is a
high-Q inductor. The use of a high-Q inductor can provide
performance advantages, such as providing additional attenuation to
signals outside of the pass band of the Input-BPF 205 and hence to
protect subsequent components in the HIPCF canceller 130, allowing
to trade linearity for a lower noise floor. In certain exemplary
embodiments, the inductor L1 is a low-Q inductor. In certain
exemplary embodiments, the Input-BPF 205 includes a Q-enhancement
circuit 290 to improve the quality factor (Q-factor) of the
inductor L1. However, some Q-enhancement circuits may introduce
noise or interference onto signals passed through the Input-BPF
205.
[0049] The resonant frequency of the Input-BPF 205 can be tuned to
(or near) the receive frequency of the receiver 135 in order to
pass interfering signals at that frequency that may be present on
the sampled transmit signals and block or filter out aggressor
signals, such as fundamental tones or carrier signals transmitted
by the transmitter 105 as well as other out-of-band blocker signals
(signals having a frequency outside of the receiver's frequency
band). If the receiver 135 includes a mobile TV tuner or other
frequency adjustable device, the resonant frequency of the
Input-BPF 205 may be adjusted to match the frequency of a current
channel to which the mobile TV tuner is set. For example, channel
50 of a mobile TV tuner may have a receive frequency within the
frequency band of 686 MHz to 692 MHz. While the mobile TV is tuned
to this frequency, the Input-BPF 205 also can be tuned to this
frequency automatically. If the mobile TV is subsequently tuned to
another channel having a different receive frequency, the resonant
frequency of the Input-BPF 205 can be adjusted to match the receive
frequency of the new channel. For example, the controller 235 may
communicate with the receiver 135 to obtain the current receive
frequency for the receiver 135. In response, the controller 235 may
adjust the switchable capacitors C1 and C2 such that the resonant
frequency of the Input-BPF 205 is close to or equal to the receive
frequency.
[0050] The Input-BPF 205 reduces the amplitude of signals having
frequencies differing from the resonant frequency of the Input-BPF
205. For example, if the receiver 135 and the transmitter 105 are
operating at different frequencies, the Input-BPF 205 can reduce
the amplitude of the fundamental tones of the sampled transmit
signal. In certain exemplary embodiments, the Input-BPF 205 may
reduce the amplitude of the fundamental tones of the sampled
transmit signal located at 824 MHz by approximately 13-18 dBc while
its center frequency is tuned to 749 MHz (corresponding to channel
60 of a mobile TV tuner). The output of the Input-BPF 205 is
electrically coupled to a low noise amplifier (LNA) 210. The LNA
210 amplifies the signal output by the Input-BPF 205 and passes
this amplified signal to a second band-pass filter, referred to
herein as LNA-BPF 215. In certain exemplary embodiments, the LNA
210 is a cascode LNA.
[0051] In this exemplary embodiment, the LNA-BPF 215 includes an
inductor L2 and a switchable capacitor C3. In certain exemplary
embodiments, the inductor L2 is a high-Q inductor. In certain
exemplary embodiments, the inductor L2 is a low-Q inductor. In
certain exemplary embodiments, the LNA-BPF 215 includes a
Q-enhancement circuit 291 to improve the Q-factor of the inductor
L2. In certain exemplary embodiments, the Q-factor of L2 is less
than the Q-factor of L1. In certain exemplary embodiments, the
Q-factor of L2 is greater than the Q-factor of L2.
[0052] Similar to the Input-BPF 205, the resonant frequency of the
LNA-BPF 215 can be set to the receive frequency of the receiver 135
to pass signals at that frequency and to further filter the
fundamental tones and out-of-band blockers from the sampled
transmit signal. In certain exemplary embodiments, the LNA-BPF 215
may further reduce the amplitude of the fundamental tones located
at 824 MHz by approximately 13-18 dBc while its center frequency is
tuned to 749 MHz.
[0053] The output of the LNA-BPF 215 is electrically coupled to a
variable gain amplifier (VGA) 220 that adjusts the amplitude of
signals output by the LNA-BPF 215. In certain exemplary
embodiments, the VGA 220 includes multiple variable gain amplifiers
for adjusting the amplitude of the signal received from the LNA-BPF
215. The amplitude adjusted signal output by the VGA 220 is then
passed to a third band-pass filter (Q-Enhanced-BPF) 225.
[0054] The Q-Enhanced-BPF 225 can include an inductor L3 and a
switchable capacitor 615 (FIG. 6) for tuning the Q-Enhanced-BPF 225
to the receive frequency of the receiver 135 to pass any signals at
that frequency and to further filter the fundamental tones and
out-of-band blockers of the sampled transmit signal. In certain
exemplary embodiments, the inductor L3 can be a high-Q inductor
(e.g., off-chip), or a low-Q on-chip spiral inductor. In certain
exemplary embodiments, the Q-Enhanced-BPF 225 also includes a
Q-enhancement circuit 292. In certain exemplary embodiments, the
Q-Enhanced-BPF 225 includes current switching (FIG. 6) to adjust
its Q-factor. In certain exemplary embodiments, the Q-Enhanced-BPF
225 can further reduce the amplitude of the fundamental tones
remaining in the signal received from the VGA 220 located at 824
MHz by up to 26 dBc or more while its center frequency is tuned to
749 MHz. The output of the Q-Enhanced BPF 225 is electrically
coupled to an I/Q modulator 230.
[0055] Although in the illustrated embodiment, a cascade of
band-pass filters 205, 215, and 225 are used to filter noise or
other signals having frequencies outside the frequency band of the
receiver 135, other types of filters may be utilized in addition to
or in place of one or more of the band-pass filters 205, 215, and
225. For example, one or more high-pass and/or low-pass filters can
be used in certain exemplary embodiments. The cascade of band-pass
filters 205, 215, 225 block or reduce the amplitude of signals
outside of the receive frequency band of the receiver 135 which
would normally not interfere with the receiver's sensitivity. The
signals within the receive frequency band of the receiver 135 are
passed through the band-pass filters 205, 215, and 225 to the I/Q
modulator 230. These in-band signals are also amplified by the LNA
210 and the VGA 220.
[0056] The I/Q modulator 230 adjusts at least one of the phase,
amplitude, and delay of the signal received from the Q-Enhanced-BPF
225 to produce an interference compensation signal that, when
applied to the receive path 133 of the receiver 135, reduces,
suppresses, cancels, or otherwise compensates for the noise and/or
interference present on the received path 133 of the receiver 135
imposed by signals transmitted by the transmitting antenna 115. In
certain exemplary embodiments, this interference compensation
signal has a 180 degree phase shift relative to that of the in-band
noisy signal and an amplitude close to or the same as that of the
in-band noisy signal. Thus, the interference compensation signal
reduces or cancels the in-band noisy signal.
[0057] In certain exemplary embodiments, the aforementioned
parameters of amplitude, phase, and delay are tuned based on a set
of instructions (e.g., algorithms) stored in a memory device 760
(FIG. 7) and executed by the controller 235 using feedback from the
victim receiver's receive signal quality indicator, such as
Bit-Error-Rate (BER), Packet-Error-Rate (PER), Receive Signal
Strength Indicator (RSSI), noise floor, Signal-Noise-Ratio (SNR),
Error Vector Magnitude (EVM), and Position Accuracy (for GPS) etc.
Exemplary algorithms for determining settings for adjusting the
amplitude, phase, and delay are described below with reference to
FIGS. 17-31.
[0058] As shown in FIG. 7, in certain exemplary embodiments, the
HIPCF canceller 130 includes a power detector 745, such as a peak
detector, coupled to the input of the I/Q modulator 230. The power
detector 745 senses or measures the power level of the signal at
the input of the I/Q modulator 230 and provides an indication of
the power level to the controller 235. The controller 235 uses this
power level value to trim the currents and hence Q.sub.max of the
Q-Enhanced-BPF 225 for maintaining an acceptable suppression of the
noise and/or interference imposed on the receiver 135 by signals
transmitted by the transmitting antenna 115. In certain exemplary
embodiments, the HIPCF canceller 130 includes an analog-to-digital
(A/D) converter 750 that receives the power level value from the
power detector 745 and provides a digital representation of the
power level value to the controller 235. The controller 235
executes a calibration routine to ensure an acceptable level of
suppression of the noise and/or interference imposed on the
receiver 135 by signals transmitted by the transmitting antenna
115. Exemplary calibration routines are described below with
reference to FIGS. 9-16.
[0059] The controller 235 can be implemented in the form of a
microcontroller, microprocessor, computer, state machine,
programmable device, control logic, analog and digital circuitry,
or other appropriate technology. The controller 235 can execute one
or more processes or programs for adjusting the settings of each of
the band-pass filters 205, 215, and 225 and for operating the
switchable capacitors C1-C3 and SCA 615 (FIG. 6). In one example,
the controller 235 automatically adjusts the resonant frequencies
of one or more of the band-pass filters 205, 215, 225 in response
to a change in frequency of the receiver 135. For example, if the
receiver 135 comprises a mobile TV tuner, the controller 235
adjusts the resonant frequency of the band-pass filters 205, 215,
225 to match or correspond to the receiver frequency. The
controller 235 adjusts the resonant frequencies of the band-pass
filters 205, 215, 225 by adjusting the capacitance of the
switchable capacitors C1-C3 and SCA 615, respectively, as discussed
below with reference to FIG. 3.
[0060] The controller 235 also can adjust or refine the settings of
the I/Q modulator 230, the band-pass filters 205, 215, and 225, and
the VGA 220 to account for environmental changes, such as changes
to temperature, supply voltage, and antenna coupling. In certain
exemplary embodiments, the controller 235 executes a calibration
routine (FIG. 16) to identify acceptable settings based on these
environmental changes and stores the identified optimal settings
for subsequent use. The algorithm(s) can be embodied as software
stored on the controller 235 or on a memory storage device 760.
Alternatively, the algorithm(s) can be implemented in one or more
hardware devices, such as discrete logic gates.
[0061] The HIPCF canceller 130 also includes auxiliary circuits
240. As shown in FIG. 7, the auxiliary circuits 240 include a
temperature sensor 755, a power detector 745, one or more analog to
digital converters 750, digital to analog converters, and other
types of circuits for use by the HIPCF canceller 130. The auxiliary
circuits 240 can also include one or more memory storage devices
760, such as RAM, ROM, and/or flash memory. Settings for each
band-pass filter 205, 215, and 225 may be stored on the memory
storage device 760. Additionally, settings for the I/Q modulator
230 may be stored on the memory storage device 760. For example,
settings for each channel of a mobile TV tuner may be stored on the
memory storage device 760.
[0062] Certain elements or functions of the HIPCF canceller 130 can
be embodied in an integrated circuit, for example as depicted by
the chip boundary 250 that FIG. 2 illustrates. For example, the
switchable capacitors C1-C3, the LNA 210, the VGA 220, the
Q-Enhanced-BPF 225, the I/Q modulator 230, the controller 235 and
one or more of the auxiliary circuits 240 can be embodied in a
single integrated circuit or multiple integrated circuits. Although
the inductors L1 and L2 are illustrated as off-chip inductors in
the illustrated exemplary embodiment, other exemplary embodiments
may employ on-chip inductors in the band-pass filters 205 and 215.
The integrated circuit(s) and/or the inductors L1 and L2 can be
installed on a mobile device, such as a mobile phone, as well as
other communication devices. The single or multiple integrated
circuits can be embodied in or on a complementary-metal-oxide
semiconductor (CMOS).
[0063] Referring to FIGS. 1 and 2, the HIPCF canceller 130
suppresses, cancels, or otherwise compensates for in-band or nearby
out-of-band (relative to the receive frequency of the receiver 135)
interfering signals imposed on the receiver 135 by signals
transmitted by the transmitter 105 via the transmitting antenna
115. That is, the HIPCF canceller 130 compensates for interfering
signals transmitted by the transmitting antenna 115 that has a
frequency within or near the frequency band of the receiver 135.
The HIPCF canceller 130 obtains samples of signals transmitted by
the transmitter 105 from the sampling device 125 and process the
samples to produce an interference compensation signal that, when
applied to an input of the receiver 135, compensates for the
imposed interfering signals.
[0064] The exemplary HIPCF 130 includes three band-pass filters
205, 215, and 225 that each filter, block, or reduce the intensity
of signal components of the sampled transmit signals received from
the sampling device 125 that are out-of-band with respect to the
receive frequency of the receiver 135. The components of the
sampled transmit signals in-band with respect to the receiver 135
are used to generate the interference compensation signal. At least
one of phase, amplitude, and delay of these components of the
sampled transmit signal are adjusted by the I/Q modulator 230 to
generate the interference compensation signal. The controller 235
can execute one or more calibration algorithms and/or one or more
tuning algorithms to improve the level of interference
compensation. The controller 235 can obtain feedback from the power
detector 745 or from the receiver 135 and use this feedback during
execution of the algorithms. These algorithms are discussed in
detail below with reference to FIGS. 9-31.
[0065] FIG. 3 is a block schematic diagram 300 of certain
components of the HIPCF canceller 130 of FIG. 2, in accordance with
certain exemplary embodiments. In particular, FIG. 3 is a
transistor level diagram of an exemplary Input-BPF 205, an
exemplary LNA-BPF 215, and an exemplary LNA 210. Referring to FIG.
3, the Input-BPF 205 includes a first switched capacitor array
(SCA) 305 and a second SCA 310. Each of the SCAs 305, 310 includes
an array having a number `n+1` of capacitors which typically
include 1 or 2 standard capacitor sizes (unity cap). Each capacitor
in the SCAs 305, 310 has a corresponding transistor switch (e.g., a
MOS transistor) for activating the capacitor. The resonant
frequency of the Input-BPF 205 can be adjusted by selecting one or
more of the capacitors from the SCAs 305 and 310. The capacitor(s)
can be selected by activating the switch associated with each
selected capacitor(s). For example, the capacitor C10 can be
selected by activating (or turning on) switch M10. Each capacitor
in the SCAs 305, 310 can have a different value of capacitance
corresponding to a different resonant frequency for the Input-BPF
205, or have a different weighted value to cover the frequency band
of the receiver 135. In certain exemplary embodiments, the
controller 235 can activate and deactivate the switches in the SCAs
305 and 310 to select the resonant frequency for the Input-BPF
205.
[0066] The SCAs 305 and 310 also can provide a voltage divider
function. This is particularly useful for mobile telephone
embodiments having a wide-band mobile TV tuner as a receiver 135.
In certain exemplary embodiments, the SCAs 305 and 310 have a
capacitor ratio (e.g., 1:5) that produces an additional 15 dBc
reduction of the amplitude of the sampled transmit signal, thus
either reducing the linearity requirements for the subsequent
stages or allowing for more gain when a smaller ratio (e.g., 1:1)
is selected. The ratio also may be varied depending on the channel
in order to flatten or adjust the overall gain over the entire
mobile TV band. To configure the Input-BPF 205 for high UHF (ultra
high frequency) channels (such as channel 50 for a mobile TV tuner)
that have frequencies close to that of a GSM, CDMA, or LTE
transmitter 105, switch M61 can be activated and switches M60 and
M62 can be deactivated. This provides a voltage divider between a
capacitor in the first SCA 305 and a capacitor in the second SCA
310.
[0067] To configure the Input-BPF 205 for low UHF channels, such as
channel 26 of a mobile TV tuner which may have a frequency between
542 MHz and 548 MHz, the second SCA 310 can be disconnected from
the Input-BPF 205 circuit by activating switches M60 and M62 and
deactivating switch M61. In certain exemplary embodiments, this
configuration reduces the attenuation of the sampled signal by 15
dB. This compensates for frequency dependent gain variations of the
three band-pass filters 205, 215, and 225.
[0068] In certain exemplary embodiments, the inductor L1 of the
Input-BPF 205 can be biased at half of Vdd for an integrated
circuit that the inductor L1 is coupled to in order to maximize the
input voltage swing without violating the integrated circuit's
specification while capacitor C4 provides a return path to ground.
In certain exemplary embodiments, the bias voltage for the inductor
L1 may be higher if adequate precautions are taken regarding the
maximum breakdown voltage of the circuit, for example by employing
zener diodes for ESD, cascoded input stages, larger channel
devices, LDD MOSFETs, etc.
[0069] The LNA-BPF 215 also includes an SCA 315 having `n+1` number
of capacitors. In this exemplary embodiment, each capacitor in the
SCA 315 includes a corresponding transistor switch (e.g., a MOS
transistor) for activating the capacitor. Similar to the Input-BPF
205, the resonant frequency of the LNA-BPF 215 can be adjusted by
selecting one or more of the capacitors of the SCA 315.
[0070] In this exemplary embodiment, the LNA 210 is a cascode LNA
having two transistors M4 and M5. The cascode LNA 210 can use a
frequency dependent degeneration that can be activated at high
frequencies by deactivating switch M7. This serves the purpose of
increasing input linearity at high frequencies as well as providing
sufficient gain at low frequencies for maintaining low noise figure
of the LNA 210 by activating switch M7.
[0071] The capacitors and the switches in each of the SCAs 305,
310, and 315 can be configured to avoid charge pumping due to their
single ended nature. As shown in FIG. 3, this can be accomplished
by inserting MOS switches M10 to M1n between capacitor C10 to C1n
and the chip input, MOS switches M20 to M2n between capacitor C20
to C2n and the AC coupling capacitor C4, and MOS switches M30 to
M3n between capacitor C30 to C3n and the output of LNA 210. High-Q
external inductors L1 and L2 may be used in the HIPCF canceller 130
instead of on-chip inductors to provide higher frequency
selectivity. The use of SCAs 305, 310, and 315 with sufficient
tuning range can compensate for the spread of the two off-chip
inductors L1 and L2 and parasitic capacitance associated with
printed circuit boards. The integrated circuit having components of
the HIPCF canceller 130 can have an input pin, an AC ground pin,
and an LNA pull-up pin, each having multiple ESD diodes arranged in
series to allow for a larger signal swing.
[0072] In certain exemplary embodiments, one or more of the
band-pass filters 205, 215, and 225 are implemented as parallel
resonance circuits. In certain exemplary embodiments, one or more
of the band-pass filters 205, 215, and 225 are implemented as
series resonance circuits. In certain exemplary embodiments, one or
more of the band-pass filters 205, 215, and 225 are implemented as
a low-pass filter rather than a band-pass filter. For example, each
of the band-pass filters 205, 215, and 225 may be replaced with a
low-pass filter if the main tone of the transmitter 105 has a
frequency greater than the frequency range for interference
suppression. In certain exemplary embodiments, one or more of the
band-pass filters 205, 215, and 225 are implemented as a high-pass
filter rather than a band-pass filter. For example, each of the
band-pass filters 205, 215, and 225 may be replaced with a
high-pass filter if the main tone of the transmitter 105 has a
frequency less than the frequency range for interference
suppression. In certain exemplary embodiments, a combination of
low-pass, high-pass, and band-pass filters may be used in place of
the band-pass filters 205, 215, and 225.
[0073] FIG. 4 depicts a spectral diagram 400 of signals received at
a victim receiver antenna, such as antenna 120 of FIG. 1, in
accordance with certain exemplary embodiments. Referring to FIGS. 1
and 4, the spectral diagram 400 shows the amplitude 403 of the
signals received at the antenna 120 plotted against signal
frequency 402. The spectral diagram 400 includes a first peak 404
corresponding to the carrier frequency F.sub.T of the aggressor
transmitter 105 and a second peak 405 corresponding to the channel
frequency F.sub.R of the victim receiver 135. The spectral diagram
400 also includes a noise sideband 406 corresponding to the phase
noise or other unwanted spectral components generated by the
aggressor transmitter 105. In certain exemplary embodiments, the
victim receiver's preferred signal-to-noise ratio (SNR) for proper
reception is not met by the amplitude difference between the second
peak 405 and the noise sideband 406.
[0074] FIG. 5 depicts a spectral diagram 500 of signals received at
the input of a victim receiver, such as receiver 135 of FIG. 1,
after cancellation of in-band unwanted spectral components by an
HIPCF canceller, such as the HIPCF canceller 130 of FIG. 1, in
accordance with certain exemplary embodiments. Referring to FIGS. 1
and 5, the spectral diagram, 500 shows the amplitude 403 of the
signals received at the receiver 135 plotted against signal
frequency 402. The spectral diagram 500 includes a noise sideband
506 corresponding to the phase noise or other unwanted spectral
components generated by the aggressor transmitter 105. This noise
sideband 506 differs from the noise sideband 406 of spectral
diagram 400 in that the noise sideband 506 includes a notch 507
centered at the channel frequency F.sub.R of the victim receiver
135. This notch 507 results from the compensation provided by the
interference compensation signal generated by the HIPCF canceller
130 and applied to the input of the receiver 135. In certain
exemplary embodiments, the SNR of the signal is improved by an
amount corresponding to the depth of the notch 507. Thus, the notch
507 improves the signal SNR, thus increasing the sensitivity of the
victim receiver 135. For example, improved cancellation of phase
noise or other unwanted spectral components by the HIPCF canceller
130 results in a deeper notch 507 and thus, better SNR for the
victim receiver 135.
[0075] FIG. 6 is a block schematic diagram of the Q-enhanced BPF
225 of FIG. 2, in accordance with certain exemplary embodiments. In
particular, FIG. 6 is a transistor level diagram of the Q-enhanced
BPF 225. The exemplary Q-enhanced BPF 225 includes an LC tank 610
having an inductor L3, a bypass switch 670, and an SCA 615. In
certain exemplary embodiments, the inductor L3 is a low-Q on-chip
spiral inductor. In certain exemplary embodiments, the inductor L3
is a high-Q off-chip inductor. Similar to the band-pass filters 205
and 215, the resonant frequency of the Q-Enhanced-BPF 225 can be
set (e.g., automatically by the controller 235) to the receive
frequency of the receiver 135 to pass in-band signal components and
to further filter, block, or reduce the intensity of the
fundamental tones and out-of-band blockers from the sampled
transmit signal.
[0076] The SCA 615 includes a number `n+1` of capacitors C40-C4n.
In the illustrated embodiment, each capacitor C40-C4n includes two
corresponding transistor switches (e.g., a MOS transistor) for
activating the capacitor. For example, the capacitor C40 includes
transistor switches M40 and M50. In addition, the Q-Enhanced-BPF
225 also includes two series-connected voltage controlled
capacitors VC1 and VC2 in parallel with the SCA 615. In certain
exemplary embodiments, the voltage controlled capacitors VC1 and
VC2 are varactors. Disposed between the two voltage controlled
capacitors VC1 and VC2 is a center tap 655 that electrically
couples the voltage controlled capacitors VC1 and VC2 to a
digital-to-analog (D/A) converter 650. The D/A converter 650 varies
the voltage level of the voltage control capacitors VC1 and VC2 in
response to a signal received from the controller 235. The
controller 235 can adjust the resonant frequency of the
Q-Enhanced-BPF 225 by activating one or more of the capacitors
C40-C4n (via switches M40-M4n and M50-M5n) and by controlling the
voltage level at the center tap 655 and thus, the capacitance of
the voltage controlled capacitors VC1 and VC2. The voltage
controlled capacitors VC1 and VC1 enable the controller 235 to
finely tune the resonant frequency of the Q-Enhanced-BPF 225.
[0077] The exemplary Q-Enhanced-BPF 225 also includes a
cross-coupled pair 620 of transistor switches M8 and M9 in parallel
with the SCA 615. The cross-coupled pair 620 provides a negative
resistance to reduce the resistance of an LC tank formed by
inductor L3, the SCA 615, and voltage controlled capacitors VC1 and
VC2.
[0078] The Q-Enhanced-BPF 225 includes a number `n+1` of current
sources M60-M6n (e.g., binary weighted), each having a gate
terminal electrically coupled together and with a reference current
(Ref_C). The Q-Enhanced-BPF 225 also includes a number `n+1` of
current switches M70-M7n. By selecting one or more of the current
sources M60-M6n via activating and deactivating (e.g., by the
controller 235) the corresponding current switch(es) M70-M7n, the
current in switches M8 and M9 can be adjusted which in turn adjusts
the resistance of the LC tank 610. Thus, the Q-factor of the
Q-Enhanced-BPF 225 can be adjusted. For example, the Q-factor of
the Q-Enhanced-BPF 225 can be adjusted to a desired level such that
filtering of out-of-band signals is improved or maximized without
the Q-Enhanced-BPF 225 oscillating.
[0079] The Q-Enhanced-BPF 225 also includes a bypass switch 670
having a resistor R8 electrically coupled to and disposed between
two transistor switches M80 and M81. As discussed in further detail
with reference to FIGS. 11-15, the switches M80 and M81 can be
activated or turned on during calibration of the Input-BPF 205 and
the LNA-BPF 215 while current sources M60-M6n are deactivated. When
the switches M80 and M81 are activated, the resistor R8 may detune
the LC tank. During normal operation, the switches M80 and M81 are
typically deactivated.
[0080] FIG. 7 is another block schematic diagram of the HIPCF
canceller 130 depicting additional components of the HIPCF 130, in
accordance with certain exemplary embodiments. As shown in FIG. 7,
the exemplary HIPCF 130 also includes bypass switches 720 and 725
for use during calibration of the HIPCF 130. In particular, the
Input-BPF 205 includes the bypass switch 720 and the LNA 215
includes the bypass switch 725. The bypass switch 720 includes a
transistor switch M82 and a resistor R2. Similarly, the bypass
switch 725 includes a transistor switch M83 and a resistor R3.
During the configuration of the HIPCF 130 (e.g., using automatic
test equipment (ATE), bench measurement, or in-site calibration),
each of the bypass switches 720, 725 and the bypass switch 670 of
the Q-Enhanced-BPF 225 can be activated and deactivated to
selectively tune the band-pass filters 205, 215, and 225.
[0081] The HIPCF canceller 130 also includes a buffer 770 disposed
between the VGA 220, Q-enhanced-BPF 225, and the I/Q Modulator 230.
The auxiliary circuits 240 include a power detector 745
electrically coupled to the output of the buffer 770. The power
detector 745 measures the power level of the sampled transmit
signal at the output of the buffer 770 and provides an indication
of the measurement to an A/D converter 750. The A/D converter 750
converts the indication to a digital signal and provides the
digital signal to the controller 235.
[0082] The auxiliary circuits 240 also include a temperature sensor
755 having an output electrically coupled to the controller 235.
The temperature sensor 755 is positioned on the chip (integrated
circuit) that the HIPCF canceller 130 is mounted or fabricated on
to measure the temperature of the chip. The controller 235 can
receive temperature measurements from the temperature sensor 755
and use these measurements for monitoring, calibration, and for
temperature compensation. In certain exemplary embodiments, the
output of the temperature sensor 755 is coupled to an A/D
converter, such as A/D converter 750 or a second A/D converter. In
exemplary embodiments having a shared A/D converter 750 for the
power detector 745 and the temperature sensor 755, the controller
235 can provide a signal to the A/D converter requesting which of
the two measurements (power or temperature) to obtain.
[0083] FIG. 8 depicts a functional block diagram of a communication
system 800, in accordance with certain exemplary embodiments. The
exemplary communication system 800 includes two communication
devices 805 and 850, each having a transmitter 810 and 855,
respectively, and a receiver 820 and 865, respectively. The
communication system 800 includes a first HIPCF canceller 880 for
compensating for noise and/or interference imposed onto an input of
the receiver 865 from signals transmitted by the transmitter 810
via a first antenna 825. The communication system 800 also includes
a second HIPCF canceller 885 for compensating for noise and/or
interference imposed onto an input of the receiver 820 from signals
transmitted by the transmitter 855 via a second antenna 870. Thus,
the communication system 800 includes interference compensation
circuits for protecting both communication devices 805 and 850. For
example, the communication device 805 may be a cellular radio and
the communication device 850 may be a WiFi radio. In this example,
the cellular radio would be protected from interference imposed on
the cellular radio receiver caused by signals transmitted by the
WiFi radio and, conversely, the WiFi radio would be protected from
interference imposed on the WiFi receiver from signals transmitted
by the cellular radio.
[0084] The HIPCF canceller 880 receives samples of signals
transmitted by the transmitter 810 via a sampling device 890
electrically coupled to the output of the transmitter's power
amplifier 815 and processes those samples to generate an
interference compensation signal. The HIPCF canceller 880 applies
the generated interference compensation signal to the input of the
receiver 865 at cancellation point 833 and, in turn, the
interference compensation signal cancels, suppresses, or otherwise
compensates for noise and/or interference imposed on the receiver
865. The HIPCF canceller 880 can include a controller similar to
controller 235 of FIG. 2 that executes one or more calibration and
one or more tuning algorithms to improve the noise and/or
interference compensation. The controller can receive feedback,
such as a "receive signal quality indicator," and use the feedback
during the execution of the algorithms to improve the noise and/or
interference compensation. Similar to the cancellation point 134,
the cancellation point 833 can be implemented as converging
electrical conductors, a coupler, a summation node, an adder, or
other suitable technology.
[0085] Similarly, the HIPCF canceller 885 receives samples of
signals transmitted by the transmitter 855 via a sampling device
895 electrically coupled to the output of the transmitter's power
amplifier 860 and processes those samples to generate an
interference compensation signal. The HIPCF canceller 885 applies
the generated interference compensation signal to the input of the
receiver 820 at cancellation point 834 and, in turn, the
interference compensation signal cancels, suppresses, or otherwise
compensates for noise and/or interference imposed on the receiver
820. The HIPCF canceller 885 can include a controller similar to
controller 235 of FIG. 2 that executes one or more calibration and
one or more tuning algorithms to improve the noise and/or
interference compensation. The controller can receive feedback,
such as a "receive signal quality indicator," and use the feedback
during the execution of the algorithms to improve the noise and/or
interference compensation. Similar to the cancellation point 134,
the cancellation point 834 can be implemented as converging
electrical conductors, a coupler, a summation node, an adder, or
other suitable technology.
[0086] FIG. 9 depicts a lookup table 900, in accordance with
certain exemplary embodiments. Referring to FIGS. 2, 7, and 9, the
lookup table 900 can be stored in the memory device 760 of the
HIPCF canceller 130. The exemplary lookup table 900 includes center
frequency settings 910 for the Input-BPF 205, center frequency
settings 920 for the LNA-BPF 215, and center frequency settings 930
for the Q-Enhanced-BPF 225. In this exemplary embodiment, the
Input-BPF center frequency settings 910 include three frequency
values (Freq1, Freq2, and Freq3) for which the band-pass filters
205, 215, and 225 have been characterized. For example, each of the
band-pass filters 205, 215, and 225 may be characterized at 450
MHz, 600 MHz, and 770 MHz in a mobile TV receiver 135 embodiment.
The Input-BPF center frequency settings 910 also include switched
capacitor array settings (SCA_Input_BPF1-SCA-Input_BPF3) for each
of the three frequency values (Freq1-Freq3), respectively. The
switched capacitor array settings (SCA_Input_BPF1-SCA-Input_BPF3)
control how the SCA 305 and the SCA 310 are controlled for each of
the frequencies (Freq1-Freq3) and thus, the resonant frequency of
the Input-BPF 205 for those frequencies. The Input-BPF center
frequency settings 910 also include temperature coefficient values
(Tempco1-Tempco3) for each frequency value (Freq1-Freq3),
respectively. The temperature coefficient values (Tempco1-Tempco3)
are used by the controller 235 to adjust the settings of the SCA
305 and 310 based on changes in temperature.
[0087] Similarly, the LNA-BPF center frequency settings 920
includes switched capacitor array settings
(SCA_LNA_BPF1-SCA-LNA_BPF3) for each of the three frequency values
(Freq1-Freq3), respectively. The switched capacitor array settings
(SCA_LNA_BPF1-SCA-LNA_BPF3) control how the SCA 315 is controlled
for each of the frequencies (Freq1-Freq3) and thus, the resonant
frequency of the LNA-BPF 215 for those frequencies. The LNA-BPF
center frequency settings 920 also include temperature coefficient
values (Tempco1-Tempco3) for each frequency value (Freq1-Freq3),
respectively. These temperature coefficient values
(Tempco1-Tempco3) are used by the controller 235 to adjust the
settings of the SCA 315 based on changes in temperature.
[0088] The Q-Enhanced-BPF center frequency settings 930 include
switched capacitor array settings (SCA_QE_BPF1-SCA-QE_BPF3) for
each of the three frequency values (Freq1-Freq3), respectively. The
switched capacitor array settings (SCA_QE_BPF1-SCA-QE_BPF3) control
how the SCA 615 is controlled for each of the frequencies
(Freq1-Freq3) and thus, the resonant frequency of the Input-BPF 205
for those frequencies. The Q-Enhanced-BPF center frequency settings
920 also include temperature coefficient values (Tempco1-Tempco3)
for each frequency value (Freq1-Freq3), respectively. These
temperature coefficient values (Tempco1-Tempco3) are used by the
controller 235 to adjust the settings of the SCA 615 based on
changes in temperature. The Q-Enhanced-BPF center frequency
settings 930 also include DAC settings (DAC1-DAC3) for the voltage
controlled capacitors VC1 and VC2 for each frequency (Freq1-Freq3),
respectively. The Q-Enhanced-BPF center frequency settings 920 also
include temperature coefficient values
(CurrentTempco1-CurrentTempco3) for each frequency value
(Freq1-Freq3), respectively. These temperature coefficient values
(CurrentTempco1-CurrentTempco3) are used by the controller 235 to
adjust the settings of the current switches M70-M7n, and thus, the
bias current in the Q-enhanced-BPF 225 based on changes in
temperature.
[0089] The exemplary lookup table 900 also includes seed values 940
for the I/Q modulator 230. The seed values 940 include in-phase and
quadrature (I, Q) settings ((I1, Q1)-(I3, Q3)) for the I/Q
modulator 230 at each frequency (Freq1-Freq3), respectively. The
lookup table 900 also includes miscellaneous settings 950. The
miscellaneous settings 950 include the temperature at which a
calibration of the HIPCF canceller 130 was performed, the process
parameters of the lot the HIPCF canceller 130 was fabricated in,
the temperature coefficient of the settings of the DAC 650, the
minimum current required to keep the transistor switches M8 and M9
of the Q-Enhanced-BPF 225 turned on, and the threshold of detecting
oscillation for the on-chip power detector 745.
[0090] The lookup table 900 is stored on the memory device 760 and
accessed by the controller 235 to adjust the settings of certain
components within the HIPCF canceller 130 during normal operation
and during calibration and tuning processes discussed below. Many
of the settings in the lookup table 900 are also populated during
these calibration and tuning processes, as discussed in further
detail below.
[0091] FIG. 10 is a flow chart depicting a method 1000 for
calibrating certain components of the HIPCF canceller 130, in
accordance with certain exemplary embodiments. After fabrication of
the HIPCF canceller 130, for example in an integrated circuit,
initial settings shown in the lookup table 900 of FIG. 9 are
populated during an ATE or bench characterization process in block
1005. In block 1010, in the application stage when the HIPCF
canceller 130 is powered on, the values for the settings in the
lookup table 900 are loaded into an internal register of the
controller 235. The controller 235 can access the lookup table 900
and control the components of the HIPCF canceller 130 using a
current temperature measurement from the temperature sensor 755 and
the channel frequency that the receiver 135 is tuned to. An
optional calibration routine may also be performed in block 1010 to
calibrate the band-pass filters 205, 215, and 225 and/or the I/Q
modulator 230.
[0092] In block 1015, if the channel of the receiver 135 changes,
the I/Q modulator 230 is recalibrated by the controller 235. This
recalibration can improve the noise and/or interference
cancellation based on the receiver's receive signal quality
indicator and cancellation algorithms described below. In block
1020, the controller 235 triggers the calibration of the band-pass
filters 205, 215, 225 and the I/Q modulator 230 in response to a
command from a user or in response to the temperature change
exceeding a preset threshold, for example 10 degrees C. During the
calibration process of the method 1000, the values in the lookup
table 900 are updated.
[0093] FIG. 11 is a flow chart depicting a method 1100 for
configuring the filters of the HIPCF canceller 130 for a desired
center frequency (e.g., 450 MHz, 600 MHz, or 770 MHz for a mobile
TV embodiment), in accordance with certain exemplary embodiments.
In block 1105, the Input-BPF 205 is calibrated. The LNA-BPF 215 and
the Q-Enhanced-BPF 225 are bypassed by activating bypass switches
725 and 670 and deactivating bypass switch 720. A pilot tone or
tuner signal is applied to the input of the HIPCF canceller 130 and
the power level of the pilot tone or tuner signal is measured at
the output of the HIPCF canceller 130. The settings of the SCA 305
and the SCA 310 are adjusted based on the measured power level
until the power level reaches an acceptable level. The settings of
the SCA 305 and the SCA 310 corresponding to the acceptable power
level are populated in the lookup table 900 for later use by the
controller 235. Block 1105 is discussed in further detail below
with reference to FIG. 12.
[0094] In block 1110, the LNA-BPF 215 is calibrated. The Input-BPF
205 and the Q-Enhanced-BPF 225 are bypassed by activating bypass
switches 720 and 670 and deactivating bypass switch 725. With the
pilot tone or tuner signal still applied to the input of the HIPCF
canceller 130, the settings of the SCA 315 are adjusted based on
the measured power level until the measured power level reaches an
acceptable level. The settings of the SCA 315 corresponding to the
acceptable power level are populated in the lookup table 900 for
later use by the controller 235. Block 1110 is discussed in further
detail below with reference to FIG. 13. In block 1115, the
Q-Enhanced-BPF 225 is calibrated. Block 1115 is discussed in
further detail below with reference to FIG. 14.
[0095] In block 1120, the temperature coefficients for the
band-pass filters 205, 215, and 225 are calculated. In certain
exemplary embodiments, the ATE (or bench measurement equipment)
calibrates the settings for each of the band-pass filters 205, 215,
and 225 for more than one temperature. For example, the band-pass
filters may be calibrated at room temperature (e.g., 27.degree.
C.), at 70.degree. C., and at 0.degree. C. The controller 235 can
calculate the temperature coefficients by taking the difference
between the settings for each band-pass-filter 205, 215, 225 at
each temperature. The temperature coefficients can be stored in the
lookup table 900 in the corresponding fields of fields 910, 920,
930, and 950.
[0096] In block 1125, the I and Q seed values for the I/Q modulator
230 are calibrated. In certain exemplary embodiments, the ATE (or
bench measurement equipment) can employ a setup similar to the
circuit 100 depicted in FIG. 1. The transmitter 105 can be
activated and one or more of the cancellation algorithms discussed
below can be executed to identify a preferred or acceptable
cancellation point for the desired center frequency. The (I, Q)
settings corresponding to the identified cancellation point can be
stored in field 940 of the lookup table 900.
[0097] After block 1125, the method 1100 ends. Of course, the
method 1100 could be executed more than one time. For example, the
method 1100 may be executed during ATE and then executed again
after the chip or system being placed into operation.
[0098] FIG. 12 is a flow chart depicting a method 1105 for
calibrating the Input-BPF 205 of the HIPCF canceller 130, in
accordance with certain exemplary embodiments, as referenced in
FIG. 11. In block 1205, the bypass switches 725 and 670 are
activated and bypass switch 720 is deactivated. This bypasses the
LNA-BPF 215 and the Q-Enhanced-BPF 225 for the calibration of the
Input-BPF 205. In certain exemplary embodiments, the controller 235
operates the bypass switches 670, 720, and 725 in response to a
command to configure the band-pass filters 205, 215, and 225.
[0099] In block 1210, a pilot tone or a tuner signal (e.g., a
mobile TV signal) with the desired center frequency (e.g., 450 MHz,
600 MHz, or 770 MHz) is applied to the input of the HIPCF canceller
130. In certain exemplary embodiments, the HIPCF canceller 130
generates the pilot tone or tuner like signal using an on-chip
phase locked loop. In certain exemplary embodiments, the pilot tone
or tuner signal is generated by re-using the phase locked loop of
the receiver 135, for example via one of the receiver's output
pins.
[0100] In block 1215, the power level of the pilot tone or tuner
signal is measured at the output of the HIPCF canceller 130. In
certain exemplary embodiments, the output power level of the pilot
tone or tuner signal is measured using ATE or bench
characterization equipment. For example, the ATE or bench
characterization equipment may include a spectrum analyzer. In
certain exemplary embodiments, the output power level of the pilot
tone or tuner signal is measured using a receive signal quality
indicator obtained from the receiver 135. In certain exemplary
embodiments, the output power level of the pilot tone or tuner
signal is measured using the power detector 745.
[0101] In block 1220, the controller 235 makes one or more
adjustments to the settings of the SCA 305 and the SCA 310 and
measures the output power level of the pilot tone or tuner signal
resulting from each adjustment. The controller 235 can continue to
make adjustments until the output power level of the pilot tone or
tuner signal reaches or exceeds an acceptable, preferred, or
maximum level. In addition or in the alternative, the controller
235 can make a certain number of adjustments and record the output
power level of the pilot tone or tuner signal (e.g., in memory
device 760) and identify the recorded output power level having the
best, preferred, or highest power level. In certain exemplary
embodiments, the controller 235 sweeps the setting values for the
SCA 305 and the SCA 310 in a monotonically increasing or decreasing
process (e.g., one least significant bit ("LSB") or multiple LSBs
at a time for digital SCAs). In certain exemplary embodiments, a
binary algorithm, such as the algorithm illustrated in FIG. 20 and
discussed below, could be used to find a preferred setting for the
SCA 305 and SCA 310.
[0102] In block 1225, the controller 235 stores the desired center
frequency and the settings for the SCA 305 and SCA 310
corresponding to the acceptable, preferred, or maximum level in the
lookup table 900 in the memory device 760. For example, the desired
center frequency may be stored in the field "Freq1" and the
settings for the SCA 305 and the SCA 310 may be stored in field
"SCA_Input_BPF1." After block 1225, the method 1105 proceeds to
block 1110, as referenced in FIG. 11.
[0103] FIG. 13 is a flow chart depicting a method 1110 for
calibrating the LNA-BPF 215 of the HIPCF canceller 130, in
accordance with certain exemplary embodiments, as referenced in
block 1110 of FIG. 11. In block 1305, the bypass switches 720 and
670 are activated and bypass switch 725 is deactivated. This
bypasses the Input-BPF 205 and the Q-Enhanced-BPF 225 for the
calibration of the LNA-BPF 215.
[0104] In block 1310, the controller 235 makes one or more
adjustments to the settings of the SCA 315 and measures the output
power level of the pilot tone or tuner signal resulting from each
adjustment. The controller 235 can continue to make adjustments
until the output power level of the pilot tone or tuner signal
reaches or exceeds an acceptable, preferred, or maximum level. In
addition or in the alternative, the controller 235 can make a
certain number of adjustments and record the output power level of
the pilot tone or tuner signal (e.g., in memory device 760) and
identify the recorded output power level having the best,
preferred, or highest power level. In certain exemplary
embodiments, the controller 235 sweeps the setting values for the
SCA 315 in a monotonically increasing or decreasing process (e.g.,
one LSB or multiple LSBs at a time for digital SCAs). In certain
exemplary embodiments, a binary algorithm, such as the algorithm
illustrated in FIG. 20 and discussed below, could be used to find a
preferred setting for the SCA 315.
[0105] In block 1315, the controller 235 stores the settings for
the SCA 315 corresponding to the acceptable, preferred, or maximum
level in the lookup table 900 in the memory device 760. For
example, the settings for the SCA 315 may be stored in field
"SCA_LNA_BPF1." After block 1315, the method 1110 proceeds to block
1115, as referenced in FIG. 11.
[0106] FIGS. 14A and 14B, collectively FIG. 14, depict a flow chart
of a method 1115 for calibrating the Q-Enhanced-BPF 225 of the
HIPCF canceller 130, in accordance with certain exemplary
embodiments, as referenced in block 1115 of FIG. 11. In block 1405,
the bypass switches 720 and 725 are activated and bypass switch 670
is deactivated. This bypasses the Input-BPF 205 and the LNA-BPF 215
for the calibration of the Q-Enhanced-BPF 225.
[0107] In block 1410, a bias current is applied (e.g., by the
controller 235) to the current switches M70-M7n for the purpose of
keeping the transistor switches M8 and M9 in the cross-coupled pair
620 and the current sources M60-M6n active or turned on, yet
avoiding oscillation of the Q-Enhanced-BPF 225. The amount of
current applied to the current switched M70-M7n may correspond to
the value of the "Minimum Current for QE" field of the lookup table
900.
[0108] In block 1415, the controller 235 makes one or more
adjustments to the settings of the SCA 615 and measures the output
power level of the pilot tone or tuner signal resulting from each
adjustment. The controller 235 can continue to make adjustments
until the output power level of the pilot tone or tuner signal
reaches or exceeds an acceptable, preferred, or maximum level. In
addition or in the alternative, the controller 235 can make a
certain number of adjustments and record the output power level of
the pilot tone or tuner signal (e.g., in memory device 760) and
identify the recorded output power level having the best,
preferred, or highest power level. In certain exemplary
embodiments, the controller 235 sweeps the setting values for the
SCA 615 in a monotonically increasing or decreasing process (e.g.,
one LSB or multiple LSBs at a time for digital SCAs). In certain
exemplary embodiments, a binary algorithm, such as the algorithm
illustrated in FIG. 20 and discussed below, could be used to find a
preferred setting for the SCA 615.
[0109] In block 1420, the controller 235 increases the amount of
current applied to the cross-coupled transistor switches M8, M9 by
increasing the settings of the current switches M70-M7n. In certain
exemplary embodiments, the amount of current is increased by a few
(e.g., 4) LSB. In block 1425, the pilot tone or tuner signal is
turned off. In block 1430, an inquiry is conducted by the
controller 235 as to whether there is any oscillation generated by
the Q-Enhanced-BPF 225. In certain exemplary embodiments, this
inquiry includes comparing the measured output power level of the
HIPCF canceller 130 with a predetermined threshold value at the ATE
or a threshold value "Power Detector Output Threshold for
Oscillation" stored in the miscellaneous values 950 of the lookup
table 900. If the measured output power level is below the
threshold, then the controller 235 determines that there is no
oscillation. If the controller 235 determines that there is no or
sufficiently low oscillation, the method 1115 proceeds to block
1435, where the controller 235 turns on the pilot tone or tuner
signal again and applies the pilot tone or tuner signal to the
input of the HIPCF canceller 130. After block 1435, the method 1115
returns to block 1415. If the controller 235 determines that there
is oscillation, the method 1115 proceeds to block 1440.
[0110] In block 1440, the controller 235 decreases the amount of
current applied to the current switches M70-M7n to the level prior
to oscillation. In block 1445, the controller 235 stores the
settings for the SCA 615 corresponding to current level prior to
oscillation in the lookup table 900 in the memory device 760. For
example, the settings for the SCA 615 may be stored in field
"SCA_QE_BPF1."
[0111] In block 1450, the pilot tone or tuner signal is reactivated
and applied to the input of the HIPCF canceller 130. The controller
235 makes one or more adjustments to the settings for the DAC 650
for biasing the voltage controlled capacitors VC1 and VC2 and
measures the output power level of the pilot tone or tuner signal
resulting from each adjustment. The adjustments to the DAC 650
adjust the voltage level at the voltage controlled capacitors VC1
and VC2. The controller 235 can continue to make adjustments until
the output power level of the pilot tone or tuner signal reaches or
exceeds an acceptable, preferred, or maximum level. In addition or
in the alternative, the controller 235 can make a certain number of
adjustments and record the output power level of the pilot tone or
tuner signal (e.g., in memory device 760) and identify the recorded
output power level having the best, preferred, or highest power
level. In certain exemplary embodiments, the controller 235 sweeps
the setting values for the DAC 650 in a monotonically increasing or
decreasing process (e.g., one LSB or multiple LSBs at a time). In
certain exemplary embodiments, a binary algorithm, such as the
algorithm illustrated in FIG. 20 and discussed below could be used
to find a preferred setting for the DAC 650.
[0112] In block 1455, the pilot tone or tuner signal is turned off.
In block 1460, the output power level of the HIPCF canceller 130 is
measured. In block 1465, an inquiry is conducted by the controller
235 as to whether there is any oscillation generated by the
Q-Enhanced-BPF 225, similar to block 1430. If the controller 235
determines that there is oscillation, the method 1115 proceed to
block 1470. If the controller 235 determines that there is no
oscillation, the method 1115 proceeds to block 1475.
[0113] In block 1470, the controller 235 lowers the current level
for biasing the cross-coupled transistors M8, M9 by decreasing the
settings of current switches M70-7n, for example by a few LSB.
After the current level is lowered, the method 1115 returns to
block 1450.
[0114] In block 1475, the controller 235 stores the settings for
the DAC 650 and the current switches M70-M7n in the lookup table
900 in the memory device 760. For example, the setting for the
current switches M70-M7n may be stored in the field "Currentl" and
the setting for the DAC 650 may be stored in field "DAC1." After
block 1475, the method 1115 ends. Of course, the method 1100 can be
repeated any number of times for any number of frequencies. For
example, the band-pass filters 205, 215, and 225 may be calibrated
for three frequencies (Freq1, Freq2, and Freq3).
[0115] FIG. 15 is a flow chart depicting a method 1500 for
calibrating the Input-BPF 205 of the HIPCF canceller 130 of FIG. 7,
in accordance with certain exemplary embodiments. This method 1500
is an alternative method to that of the method 1105 of FIG. 12. In
block 1505, the bypass switches 720, 725, and 670 are deactivated.
For example, the controller 235 may deactivate the bypass switches
720, 725, and 670.
[0116] In block 1510, a pilot tone or tuner signal with the desired
center frequency is applied to the input of the HIPCF canceller
130. In block 1515, a measurement of the reflected pilot tone or
tuner signal (e.g., reflection coefficient or return loss) is made
at the input of the HIPCF canceller 130. This measurement may be
made by a power detector or spectral analyzer, for example. In
block 1520, the controller 235 makes one or more adjustments to the
settings of the SCA 305 and the SCA 310 and measures the reflected
pilot tone or tuner signal. The controller 235 can continue to make
adjustments until the reflected pilot tone or tuner signal reaches
or exceeds an acceptable, preferred, or minimum level. In addition
or in the alternative, the controller 235 can make a certain number
of adjustments and record the reflected pilot tone or tuner signal
(e.g., in memory device 760) and identify the recorded reflected
pilot tone or tuner signal having the best, preferred, or lowest
level. In certain exemplary embodiments, the controller 235 sweeps
the setting values for the SCA 305 and the SCA 310 in a
monotonically increasing or decreasing process (e.g., one LSB or
multiple LSBs at a time for digital SCAs). In certain exemplary
embodiments, a binary algorithm, such as the algorithm illustrated
in FIG. 20 and discussed below could be used to find a preferred
setting for the SCA 305 and SCA 310.
[0117] In block 1525, the controller 235 stores the desired center
frequency and the settings for the SCA 305 and SCA 310
corresponding to the acceptable, preferred, or minimum level in the
lookup table 900 in the memory device 760. For example, the desired
center frequency may be stored in the field "Freq1" and the
settings for the SCA 305 and the SCA 310 may be stored in field
"SCA_Input_BPF1."
[0118] FIG. 16 is a flow chart depicting a method 1600 for
determining switch settings for a given frequency, in accordance
with certain exemplary embodiments. For example, the method 1600
may be performed in response to a user applying a channel change
for a mobile TV. In certain exemplary embodiments, the lookup table
900 includes the settings for each band-pass filter 205, 215, and
225 and the I and Q seed values for each channel the receiver 135
may tune to. In certain exemplary embodiments, the lookup table 900
includes the settings for each band-pass filter 205, 215, and 225
and the I and Q seed values for a predetermined number (e.g., 3) of
channel frequencies. For such embodiments that do not include
calibrated settings for each channel frequency, the method 1600
provides an exemplary process for computing the SCA switch settings
for each band-pass filter 205, 215, and 225 at any application
selectable channel frequency. The exemplary method 1600 takes into
account the calibration values in the lookup table 900 identified
for the predetermined number of channel frequencies and the actual
temperature measured by an on-chip temperature sensor, such as
temperature sensor 755.
[0119] In block 1605, the controller 235 conducts an inquiry to
determine whether to start determining switch settings for each
band-pass filter 205, 215, 225. In certain exemplary embodiments,
the controller 235 communicates with the receiver 135 to determine
whether the receive frequency for the receiver 135 has changed, for
example as a result of a change in channel for the receiver 135. If
the receive frequency for the receiver has changed, the controller
235 determines to start determining the switch settings for each
band-pass filter 205, 215, 225 and proceeds to block 1610.
Otherwise, the method 1600 remains in block 1605.
[0120] In certain exemplary embodiments, the controller 235
determines whether the temperature of the chip that the HIPCF
canceller 130 resides has changed. The controller 235 monitors the
temperature measurement received to determine whether the
temperature has changed by a certain threshold. If the controller
235 determines that the temperature has changed by an amount equal
to or exceeding the threshold, the controller 235 determines to
start determining the switch settings for each band-pass filter
205, 215, 225 and proceeds to block 1610. Otherwise, the method
1600 remains in block 1605.
[0121] In certain exemplary embodiments, the controller 235
determines whether the lookup table 900 has changed or whether a
setting or value in the lookup table 900 has been updated. If the
controller 235 determines that the lookup table has changed, the
controller 235 determines to start determining the switch settings
for each band-pass filter 205, 215, 225 and proceeds to block 1610.
Otherwise, the method 1600 remains in block 1605.
[0122] In block 1610, the controller 235 receives the receive
frequency for the receiver 135 ("target frequency"), the current
calibration values for the band-pass filters 205, 215, and 225 from
the lookup table 900, a real-time or near real-time temperature
measurement from the temperature sensor 755, and the temperature
value during calibration from the lookup table 900.
[0123] In block 1615, the controller 235 conducts an inquiry to
determine whether the target frequency is less than a frequency
threshold. For example in certain mobile TV embodiments, this
frequency threshold is set at 600 MHz which corresponds to the
middle of the receive band of certain mobile TV tuners. If the
target frequency is less than the frequency threshold, the method
1615 proceeds to block 1620. Otherwise, the method 1600 proceeds to
block 1625.
[0124] In block 1620, controller 235 computes a variable "DeltaF"
indicating the difference between the target frequency and the
frequency threshold. The controller 235 performs an interpolation
process, for example linear interpolation, using two or more of the
calibration values in the lookup table 900 to determine the
settings for the SCAs 305, 310, 315, and 615, the DAC settings for
the voltage controlled capacitors VC1 and VC2, and bias current
switch settings for the current switches M70-M7n. For example, for
each aforementioned component, the controller 235 uses the setting
stored for a first frequency, such as Freq1, and the setting stored
for a second frequency, such as Freq2, in a linear interpolation
calculation with DeltaF to determine the setting for that
component.
[0125] In block 1625, the controller computes the variable DeltaF
indicating the difference between the target frequency and a second
frequency value. In certain exemplary embodiments, if the frequency
threshold if 600 MHz, the second frequency value is 770 MHz. These
frequency values are exemplary, rather than limiting, and other
frequency values can be used without departing from the scope and
spirit of the present invention. Similar to block 1620, the
controller performs an interpolation process using two or more of
the calibration values in the lookup table 900 to determine the
settings for the SCAs 305, 310, 315, and 615, the DAC settings for
the voltage controlled capacitors VC1 and VC2, and bias current
switch settings for the current switches M70-M7n. For example, for
each aforementioned component, the controller 235 uses the setting
stored for a first frequency, such as Freq2, and the setting stored
for a second frequency, such as Freq3, in a linear interpolation
calculation with DeltaF to determine the setting for that
component.
[0126] As shown in blocks 1620 and 1625, the method 1600 uses two
different sets of calibrated settings to determine the settings for
the components of the HIPCF canceller 130 depending upon the target
frequency. This enables the controller 235 to use the calibrated
settings nearest the target frequency to determine the appropriate
settings for the components.
[0127] In block 1630, the controller 235 determines a temperature
compensation by computing a variable "DeltaTemp" which yields the
difference between actual temperature and the temperature at which
the last calibration was performed (stored in field 950 of the
lookup table 900). The controller 235 also computes offset values
caused by the temperature difference for the settings for each
component. The controller 235 uses the offset values to determine
final setting for the components at the target frequency. The
controller 235 stores the final settings in internal registers for
use in operating the components. Note that the I and Q settings for
the I/Q modulator 230 may not be temperature compensated in the
method 1600 as the I and Q settings may be calibrated using one of
the cancellation algorithms discussed below with reference to FIGS.
17-31.
[0128] FIG. 17 depicts implementation layers 1700 of noise and/or
interference cancellation algorithms, in accordance with certain
exemplary embodiments. These algorithms can use a feedback signal
from the victim receiver 135 to determine appropriate I and Q
settings for the HIPCF canceller 130. This feedback signal includes
a quality indicator (e.g., BER, PER, RSSI, noise floor, SNR, EVM,
and Position Accuracy, etc.) for the communication system 100. This
exemplary implementation of algorithms includes four layers, a link
control layer 1710, a signal processing layer 1720, an algorithm
control layer 1730, and an algorithm execution layer 1740. In
certain exemplary embodiments, each of the layers 1710-1740 may
reside in any of the following three components: 1) a baseband
integrated circuit of the victim receiver 135, 2) a stand alone
microcontroller, or 3) the on-chip controller 235 (or another
control device) of the HIPCF 130. For ease of discussion, the
layers 1710-1740 will be discussed hereinafter in terms of the
controller 235 performing the respective functions.
[0129] In the link control layer 1710, the feedback signal is
analyzed and tested for quality to determine whether cancellation
should be activated to improve the sensitivity of the victim
receiver 135. Due to the nature of the active noise and/or
interference cancellation which the HIPCF canceller 130 provides,
the HIPCF canceller 130 may also output its own noise floor while
canceling the noise and/or interference generated by the power
amplifier 110 (or another component) at the input of the victim
receiver 135. As a result, the overall noise floor seen by the
victim receiver 135 is the summation of the output noise floor of
the HIPCF canceller 130, the receiving antenna 120, the power
amplifier noise and/or interference received by the receiving
antenna 120, and the phase and gain adjusted noise floor of the
power amplifier 110 (via the HIPCF canceller 130), which in turn
can affect the sensitivity of the victim receiver 135. Thus, the
determination as to whether or not to activate the HIPCF canceller
130 to improve the sensitivity of the victim receiver 135 may be
decided based on the actual noise and/or interference of the power
amplifier 110 received by the victim receiver 135.
[0130] FIG. 18 depicts a diagram 1800 of receiver sensitivity
plotted versus coupled power amplifier noise for a mobile TV tuner
tuned at 746 MHz with a channel bandwidth of 8 MHz and a CDMA800
power amplifier, in accordance with certain exemplary embodiments.
Referring to FIG. 18, the diagram 1800 includes a first curve 1805
depicting the mobile TV tuner sensitivity with the HIPCF canceller
130 inactive and a second curve 1810 depicting the mobile TV tuner
sensitivity with the HIPCF canceller 130 canceling or suppressing
the power amplifier noise. As illustrated in the exemplary
implementation, there is no advantage to activating the HIPCF noise
canceller 130 for power amplifier noise below -174 dBm/Hz as the
output noise floor of the HIPCF canceller 130 would exceed the
benefit of canceling the coupled power amplifier noise. For power
amplifier noise above approximately -160 dBm/Hz, maximum
cancellation/sensitivity improvement (e.g., approximately 10 dB in
this exemplary implementation) would be achieved with the HIPCF
canceller 130 active as the received power amplifier noise is
typically much higher than the output noise floor of the HIPCF
canceller 130. Additionally, the link control layer 1710 detects in
multi-channel systems whether a particular channel has been
optimized in the past and passes the setting corresponding to prior
optimization from the memory to the HIPCF 130. An indication of
whether the channel has been optimized previously can be stored in
memory by the controller 235 at the conclusion of or during an
optimization.
[0131] The desired victim receive signal quality may be assessed
with respect to feedback (e.g., BER, PER, RSSI, noise floor, SNR,
EVM, and Position Accuracy, etc.) received from the receiver to
determine whether to activate the HIPCF canceller 130. For example,
the HIPCF 130 may be activated if the feedback indicates that the
receive signal is above the combined noise floor (i.e., the
summation of the output noise floor of the HIPCF canceller 130, the
receiving antenna 120, the power amplifier noise and/or
interference received by the receiving antenna 120, and the phase
and gain adjusted noise floor of the power amplifier 110 (via the
HIPCF canceller 130). This feature is illustrated in FIG. 19, which
depicts a diagram 1900 of an output SNR of a mobile TV tuner tuned
at 746 MHz with a channel bandwidth of 8 MHz versus received mobile
TV signal strength with a coupled CDMA800 power amplifier phase
noise at -161 dBm/Hz at the input of victim receiver 135, in
accordance with certain exemplary embodiments. Referring to FIG.
19, the diagram 1900 includes a first curve 1905 depicting the
mobile TV tuner output SNR with the HIPCF canceller 130 inactive,
and a second curve 1910 depicting the mobile TV tuner output SNR
with the HIPCF canceller 130 canceling or suppressing the power
amplifier phase noise. A third curve 1915 depicts a desired minimum
SNR output for the mobile TV tuner. As illustrated in the exemplary
implementation, there may not be an advantage in activating the
HIPCF noise canceller 130 when the received mobile TV signal is
below -90 dBm.
[0132] The exemplary link control layer 1710 includes several modes
of operation. The controller 235 can determine which mode of
operation to be active based on the signal quality of the signal
received by the receiver 135. In certain exemplary embodiments, the
link control layer 1710 includes four modes of operation, a maximum
cancellation mode, a limited cancellation mode, a wait for
acceptable signal mode, and a no signal mode. In this exemplary
embodiment, if the received signal strength is acceptable (e.g.,
above an acceptable threshold level which is -81 dBm in FIG. 19),
the controller 235 commences the maximum cancellation mode whereby
the noise and/or interference cancellation of the HIPCF canceller
130 is at a high level. If the received signal strength is low
(e.g., between an acceptable level threshold and a very low level
threshold, which is between -81 dBm and -90 dBm in FIG. 19), the
controller 235 commences the limited cancellation mode of the HIPCF
canceller 135 whereby the noise and/or interference cancellation
level is bounded by the noise floor. If the received signal is
lower than a threshold indicating, for example, a very low signal
(e.g. around -90 dBm in FIG. 19), the controller 235 will commence
the wait for acceptable signal mode whereby the controller 235
delays entering the subsequent layers 1720-1740 until the received
signal meets or exceeds the threshold. If the received signal meets
or exceeds the threshold while the controller 235 is in the wait
for acceptable signal mode, the controller 235 can enter the
subsequent layers 1720-1740. If there is no signal received at the
receiver 135, then the controller 235 commences the no signal mode
whereby the HIPCF canceller 130 is inactive.
[0133] The link control layer 1710 also may deduct information on
the time dependent passing of the thresholds, for example when both
threshold levels are passed in less than one second. A mobile
device may be transported into a tunnel or under a bridge and all
settings can be held constant until the passing of the acceptable
level threshold indicates that the mobile device has returned from
the tunnel or bridge. The operation of the link control layer 1710
can then resume using the held settings.
[0134] The signal processing layer 1720 includes several processes
that ensure stability and robustness of the feedback signal. A
first process includes averaging a predetermined number of feedback
values of the feedback signal before executing a noise cancellation
algorithm.
[0135] A second process includes correction for errors in feedback
signals during the execution of a noise cancellation algorithm. One
exemplary error correction process includes obtaining two feedback
values from the receiver 135 and computing the difference between
the two feedback values. If this difference is less than a
tolerance level, then the two feedback values are averaged.
Otherwise, a third feedback value is obtained from the receiver 135
and the difference between the third feedback value and the second
feedback value is determined. If this difference is less than the
tolerance level, then the second and third feedback values are
averaged. Otherwise, a fourth feedback value is obtained and a
similar process is performed for a predetermined number of
iterations. If no two feedback values are found that have a
difference less than the tolerance level, then an error may be
indicated and the HIPCF canceller 130 may be deactivated. A second
exemplary error correction process includes ranking a certain
number of feedback values and selecting a certain number of the
feedback values while a noise cancellation algorithm is running.
For example, the controller 235 may rank ten feedback values and
select the five feedback values ranked in the middle. The average
of the selected feedback signals are calculated and used in the
noise cancellation algorithms.
[0136] A third process of the signal processing layer 1720 includes
SNR averaging. This SNR averaging process includes computing the
average value of the SNRs for different satellites (SVs), for
example GPS systems, DARS (Digital Audio Radio Service), or
Iridium. The SNR averaging may be performed for satellites that
have a certain elevation level above an elevation threshold only,
to avoid an incorrect decision in the algorithm execution layer
1740.
[0137] In the algorithm control layer 1730, several user controls
can be implemented to control the algorithms described in the
algorithm execution layer 1740. One such user control is the
polarity of the logic used to compare two feedback values, for
example before and after a change of I and/or Q settings of the
HIPCF canceller 130. The polarity can either be positive (e.g.,
higher feedback value is better) or negative (e.g., lower feedback
value is better). Some exemplary feedback signals where a positive
polarity may be used are SNR, Carrier to Noise Ratio (C/N), and
Repeater Amplifier Gain. Some exemplary feedback signals where a
negative polarity may be used are PER, BER, Error Vector Magnitude,
Noise Floor Level, Adjacent Channel Power Ratio, and Adjacent
Channel Leakage Ratio.
[0138] The algorithm execution layer 1740 includes the execution of
one of several noise cancellation algorithms. These algorithms
include acts to adjust the I and Q values of the HIPCF canceller
130 and evaluate the feedback signal resulting from the adjustment
to find acceptable I and Q values for operating the HIPCF canceller
130. The algorithms include two types of binary algorithms (a fast
binary algorithm (FBA) and a binary correction algorithm (BCA)), a
minstep algorithm (MSA), a blind shot algorithm (BSA), a dual slope
algorithm (DSA), and a track and search algorithm (TSA).
[0139] FIG. 20 is a flow chart depicting a fast binary algorithm
2000 for canceling noise and/or interference, in accordance with
certain exemplary embodiments. In this exemplary FBA 2000, each bit
in I and Q values for the HIPCF canceller 130 are sequentially
reversed and tested for a better feedback value as determined by
the polarity defined in the algorithm control layer 1730. The FBA
2000 may start with a start bit and progress sequentially through
each of the bits of the I-value and Q-value until reaching a
pre-defined stop bit. In certain exemplary embodiments, the start
bit and stop bit may be user selected.
[0140] In block 2005, the controller 235 selects a first I-value
and a first Q-value for operating the HIPCF canceller 130. These
first values may be start values from the lookup table 900, seed
values, or middle of range values. In block 2010, the HIPCF
canceller 130 applies the first I-value and first Q-value to the
I/Q modulator 230.
[0141] In block 2015, the receiver 135 provides a feedback signal
having a feedback value to the controller 235. The feedback value
may be an SNR, an RSSI, a Carrier to Noise Ratio (C/N), RSSI, a
Repeater Amplifier Gain, a PER, a BER, an Error Vector Magnitude, a
Noise Floor Level, an Adjacent Channel Power Ratio, or an Adjacent
Channel Leakage Ratio. After obtaining the feedback value from the
receiver 135, the controller 235 stores the feedback value in
memory.
[0142] In block 2020, the controller 235 inverts a bit of the
I-value and transmits the updated I-value to the HIPCF canceller
130. In response, the HIPCF canceller 130 applies the updated
I-value to the I/Q modulator 230. For example, bit 2075 of I-value
2071 may be inverted from a value of "1" to a value of "0." In the
first iteration of this block 2020, the controller 235 may invert
the start bit of the I-value. In each subsequent iteration, the
next bit may be inverted until the stop bit is completed.
[0143] In block 2025, the controller 235 obtains an updated
feedback value from the receiver 135. In block 2030, the controller
235 compares the updated feedback value to the stored feedback
value to determine which of the two feedback values is better based
on the polarity defined in the algorithm control layer 1730. For
example, if the polarity is positive and the updated feedback value
is greater than the stored feedback value, then the controller 235
will determine that the updated feedback value is better. Likewise,
if the polarity is negative and the updated feedback value is
greater than the stored feedback value, then the controller 235
will determine that the stored feedback value is better. The
controller 235 stores the better feedback value and sets the
I-value to the I-value that resulted in the better feedback value.
The controller 235 also applies the I-value that resulted in the
better feedback value to the HIPCF canceller 130.
[0144] In block 2035, the controller 235 inverts a bit of the
Q-value and transmits the updated Q-value to the HIPCF canceller
130. In response, the HIPCF canceller 130 applies the updated
Q-value to the I/Q modulator 230. For example, bit 2085 of Q-value
2081 may be inverted from a value of "1" to a value of "0." In the
first iteration of this block 2035, the controller 235 may invert
the start bit of the Q-value. In each subsequent iteration, the
next bit may be inverted until the stop bit is completed.
[0145] In block 2040, the controller 235 obtains an updated
feedback value from the receiver 135. In block 2045, the controller
235 compares the updated feedback value to the stored feedback
value to determine which of the two feedback values is better based
on the polarity defined in the algorithm control layer 1730. The
controller 235 stores the better feedback value and sets the
Q-value to the Q-value that resulted in the better feedback value.
The controller 235 also applies the Q-value that resulted in the
better feedback value to the HIPCF canceller 130.
[0146] In block 2050, the controller 235 conducts an inquiry to
determine whether there are more bits in the I-value and Q-value to
test. For example, the controller 235 may determine whether the
previous iteration of blocks 2020-2050 evaluated the stop bit. If
there are more bits to test, the "Yes" branch is followed back to
block 2020 where another bit is inverted and evaluated for better
feedback. Otherwise, the "No" branch is followed to block 2055. In
block 2055, the controller 235 operates the HIPCF canceller 130
using the final stored I-value and Q-value.
[0147] In certain exemplary embodiments, the FBA 2000 illustrated
in FIG. 20 may not cover every condition and thus, it may be
improved by assigning a one or two bit start value (e.g., most
significant bit (MSB)) for both the I-value and the Q-value.
Another improvement to the FBA 2000 includes executing the BSA
described below prior to executing the FBA 2000 to obtain a start
value for the I-value and for the Q-value.
[0148] The BCA is a modification to the fast binary algorithm
illustrated in FIG. 20 and described above. In the BCA, each bit of
both the I and Q values are sequentially reversed as in the fast
binary algorithm, and either increased by a value of "1" if the
original value of the bit is "1" (and thus, causing a carry to its
immediate neighboring more significant bit) or decreased by "1" if
the original value is "0" (and thus, causing a borrow from its
immediate neighboring more significant bit). In both cases, the
controller 235 would evaluate the feedback value to determine which
value (I-value or Q-value depending on the block) resulted in
better feedback. Similar to the FBA, the I-value and Q-value that
resulted in the better feedback value is stored and used at the
completion of the algorithm to control the HIPCF canceller 130. The
BCA may begin with a start bit and proceed through each bit until
the stop bit is completed. In certain exemplary embodiment, the
start and stop bits may be user selected. In certain exemplary
embodiments, if no BSA is performed prior to the execution of the
binary correction algorithm, the binary correction algorithm may
start with the MSB and with bit reversal only for the MSB of the
I-value and the Q-value as there is not a more significant bit to
carry to or borrow from for the MSB. After the MSB for the I-value
and Q-value have been completed, the feature of increasing or
decreasing an evaluated bit by a value of "1" may begin with the
second MSB.
[0149] The motivation for implementing the BCA in place of the FBA
2000 can be discussed with reference to FIG. 21, which depicts a
graph 2100 of I and Q values adjusted using the binary algorithms.
Referring to FIG. 21, point X1 represents a plot of an initial
I-value and Q-value for the HIPCF canceller 130. As part of the
binary algorithms, the MSB of the I-value is inverted to proceed
from point X1 point X2. In this graph, the feedback value at point
X2 is determined to be better than the feedback value at point X1.
Thus, the binary algorithms would keep the I-value for point X2 and
invert the MSB of the Q-value to proceed to point X3.
[0150] At point X3, assuming the feedback value is determined to be
better at point X3 than at point X2, in the FBA 2000, the second
MSB of the I-value would be inverted. This bit inversion would
cause the algorithm to proceed from point X3 to point A, which is
further away from optimal point C and thus, would have an inferior
feedback value to that of point C. In the binary correction
algorithm, the feedback value would be tested at both points A and
B by increasing or decreasing the second MSB of the I-value by a
value of "1" and thus, affecting the MSB. Because point B is closer
to the optimal point C, point B would result in a better feedback
value than point A and the BCA would continue from point B rather
than point X3. Thus, the BCA can be more accurate than the fast
binary algorithm. However, the BCA may require more iterations and
more hardware in certain implementations.
[0151] FIG. 22 is a flow chart depicting a minstep algorithm 2200
for canceling noise and/or interference, in accordance with certain
exemplary embodiments. The exemplary MSA 2200 can provide fine
tuning to the noise cancellation, for example after one of the
binary algorithms has been executed. The MSA 2200 can follow
changes in the coupling channel between an interferer/noise source
(e.g., power amplifier 110) and the victim receiver 135. For a
given step size (e.g., 1 LSB to 7 LSB resolution), noise and/or
interference cancellation can be achieved by incrementing (plus
step size) or decrementing (minus step size) I-values and Q-values
sequentially. In certain exemplary embodiments, the incrementing or
decrementing stops at maximum or minimum values for the appropriate
I-value or Q-value (e.g., range criteria). In certain exemplary
embodiments, the MSA 2200 runs for a given number of iterations or
time period and can be interrupted by a user. I-values and Q-values
each oscillate around an desirable value or follow changes in
coupling channel.
[0152] Referring to FIGS. 1, 2, and 22, in block 2205, the
controller 235 selects a first I-value and a first Q-value for
operating the HIPCF canceller 130. In block 2210, the HIPCF
canceller 130 applies the first I-value and the first Q-value to
the I/Q modulator 230.
[0153] In block 2215, the receiver 135 provides a feedback signal
having a feedback value to the controller 235. The feedback value
may be a SNR, a RSSI, a Carrier to Noise Ratio (C/N), a Repeater
Amplifier Gain, a PER, a BER, an Error Vector Magnitude, a Noise
Floor Level, an Adjacent Channel Power Ratio, or an Adjacent
Channel Leakage Ratio, etc. After obtaining the feedback value from
the receiver 135, the controller 235 stores the feedback value in
memory.
[0154] In block 2220, the controller 235 increments the I-value by
a given step size (e.g., 1 LSB) and transmits the updated I-value
to the HIPCF canceller 130. In response, the HIPCF canceller 130
applies the updated I-value to the I/Q modulator 230. The
controller 235 also obtains an updated feedback value from the
receiver 135.
[0155] In block 2225, the controller 235 compares the updated
feedback value to the stored feedback value to determine which of
the two feedback values is better based on the polarity defined in
the algorithm control layer 1730. The controller 235 stores the
better feedback value and sets the I-value to the I-value that
resulted in the better feedback value. The controller 235 also
applies the I-value that resulted in the better feedback value to
the HIPCF canceller 130.
[0156] In block 2230, the controller 235 increments the Q-value by
a given step size (e.g., one LSB) and transmits the updated Q-value
to the HIPCF canceller 130. In response, the HIPCF canceller 130
applies the updated Q-value to the I/Q modulator 230. The
controller 235 also obtains an updated feedback value from the
receiver 135.
[0157] In block 2235, the controller 235 compares the updated
feedback value to the stored feedback value to determine which of
the two feedback values is better based on the polarity defined in
the algorithm control layer 1730. The controller 235 stores the
better feedback value and sets the Q-value to the Q-value that
resulted in the better feedback value. The controller 235 also
applies the Q-value that resulted in the better feedback value to
the HIPCF canceller 130.
[0158] In block 2240, the controller 235 decrements the I-value by
a given step size (e.g., one LSB) and transmits the updated I-value
to the HIPCF canceller 130. In response, the HIPCF canceller 130
applies the updated I-value to the I/Q modulator 230. The
controller 235 also obtains an updated feedback value from the
receiver 135.
[0159] In block 2245, the controller 235 compares the updated
feedback value to the stored feedback value to determine which of
the two feedback values is better based on the polarity defined in
the algorithm control layer 1730. The controller 235 stores the
better feedback value and sets the I-value to the I-value that
resulted in the better feedback value. The controller 235 also
applies the I-value that resulted in the better feedback value to
the HIPCF canceller 130.
[0160] In block 2250, the controller 235 decrements the Q-value by
a given step size (e.g., one LSB) and transmits the updated Q-value
to the HIPCF canceller 130. In response, the HIPCF canceller 130
applies the updated Q-value to the I/Q modulator 230. The
controller 235 also obtains an updated feedback value from the
receiver 135.
[0161] In block 2255, the controller 235 compares the updated
feedback value to the stored feedback value to determine which of
the two feedback values is better based on the polarity defined in
the algorithm control layer 1730. The controller 235 stores the
better feedback value and sets the Q-value to the Q-value that
resulted in the better feedback value. The controller 235 also
applies the Q-value that resulted in the better feedback value to
the HIPCF canceller 130.
[0162] In block 2260, the controller 235 conducts an inquiry to
determine whether to continue repeating blocks 2220 through 2255.
In certain exemplary embodiments, the determination is based on a
time period. If the time period has expired, then the controller
235 determines not to continue. In certain exemplary embodiments,
the determination is based on the sensitivity of the receiver 135
or based on the feedback value obtained in block 2255. In certain
exemplary embodiments, the determination is based on the number of
iterations executed. If the controller 235 determines to continue
repeating blocks 2220-2255, then the "Yes" branch is followed back
to block 2220. Otherwise, the "No" branch is followed to block
2265. In block 2265, the controller 235 operates the HIPCF
canceller 130 using the final selected I-value and Q-value.
[0163] In certain exemplary embodiments, the decision to change
from an increment in I-value or Q-value to a decrement is based
upon whether the previous iteration rejected the new feedback
value, i.e. the new feedback value was not preferred over the
previous feedback value.
[0164] Although the FBA 2000, the BCA 2100, and the MSA 2200 have
been discussed above in terms of a changing sequence of IQIQIQ, the
FBA 2000, the BCA 2100, and the MSA 2200 could also be implemented
using other sequences, including IIQQIIQQ, and IIIQQQIIIQQQ, for
example.
[0165] The BSA can be executed when signal conditions are poor
(e.g., acceptable start I- and Q-values are not available), or the
victim receiver baseband ICs have limited accuracy for BER or SNR
as the feedback value. In such an implementation, the BSA can be
executed to determine a start I-value and a start Q-value for the
algorithms discussed above (i.e., the FBA 2000, the BCA 2100, or
the MSA 2200. FIG. 23 depicts an I-Q plane 2300 having 16
sub-regions with feedback values that are pseudorandom. The BSA can
evaluate the feedback for multiple different I and Q pre-samples
(e.g., from a lookup table) and select the pre-sample having the
best feedback value. After the best pre-sample is determined, the
BSA can transition to either the FBA 2000, the BCA 2100, or the MSA
2200 and use the I-value and Q-value for the pre-sample as a
starting point for the algorithm.
[0166] There are several methods for implementing the BSA. In one
method, the I and Q values associated with the best feedback value
are selected from a number of samples (e.g., 4 or 16) with preset I
and Q values. In the case of 10-bit I and Q values, four samples of
feedback values may be taken from the following locations in the I
and Q plane:
[0167] I=(0xFF, 0x2FF, 0xFF, 0x2FF)
[0168] Q=(0x2FF, 0x2FF, 0xFF, 0xFF)
[0169] In the case of 10-bit I and Q values, sixteen samples of
feedback values may be taken from the following locations in the I
and Q plane:
[0170] I=(0x80, 0x80, 0x80, 0x80, 0x180, 0x180, 0x180, 0x180,
0x280, 0x280, 0x280, 0x280, 0x380, 0x380, 0x380, 0x380)
[0171] Q=0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380,
0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380)
[0172] The above locations are exemplary rather than limiting and
many other locations are feasible without departing from the scope
and spirit of the present invention.
[0173] A second method for implementing the BSA includes obtaining
feedback values at each of four (or other number) preset I and Q
points. The maximum and minimum feedback values of the obtained
feedback values can be identified. A feedback threshold is
determined by either a) averaging the minimum and maximum feedback
values, or b) adding a user selected offset value to the minimum
feedback value. After determining the feedback threshold, the BSA
can evaluate the feedback values for I and Q points proximal the
best field out of the four I and Q points. For example, the BSA can
use a user specified step size to explore I and Q points proximal
the best of the four I and Q points. The BSA can terminate when one
sample feedback meets or exceeds the feedback threshold. The BSA
can then transition to the MSA 2200.
[0174] The DSA uses an isosceles triangle approximation with two
equal and opposite slopes for approximating a noise funnel curve.
FIG. 24 is a graph 2400 depicting a receive signal quality
indicator 2405 plotted versus I or Q values resulting from an
implementation of a DSA, in accordance with certain exemplary
embodiments. The DSA can select four points (X1-X4) along a noise
funnel curve formed by the receive signal quality indicator 2405
and compute a vertex, which is close to cancellation point C. The
vertex can be computed using point-slope form of a linear equation.
Once the vertex is found, the DSA can transition to the MSA 2200,
using the vertex as starting I and Q values.
[0175] FIG. 25 is a flow chart depicting a DSA 2500 for canceling
noise and/or interference, in accordance with certain exemplary
embodiments. FIG. 26 is a graph 2600 depicting a curve 2605 of
receive signal quality indicator plotted versus either an I or Q
axis (against I and Q axes if FIG. 26 is plotted in three
dimensions) resulting from an implementation of the DSA 2500 of
FIG. 25, in accordance with certain exemplary embodiments. The
exemplary DSA 2500 uses an isosceles triangle approximation with
two equal and opposite slopes for a noise funnel curve formed by
the receiver signal quality indicator 2605. Referring to FIGS. 25
and 26, in block 2505, the controller 235 selects a number of
samples of I-values and/or Q-values along the I or Q axis. For
example, the controller 235 may select four samples. In certain
exemplary embodiments, the controller 235 uses a BSA to select the
location for the samples of I-values and/or Q-values.
[0176] In block 2510, the controller 235 communicates the samples
to the I/Q modulator 230 and the I/Q modulator applies each of the
samples one at a time. In block 2520, the controller 235 obtains a
feedback value, such as a "receive signal quality indicator," for
each of the applied samples and stores each feedback value and the
corresponding sample I and Q values in the memory device 760. In
certain exemplary embodiments, the controller 235 receives a
"receive signal quality indicator" for each sample from the
receiver 135.
[0177] In block 2520, the controller 235 compares the stored
feedback values and identifies the better feedback value. For
example, in FIG. 26, the controller 235 identifies point X1 as
resulting in the better feedback value. Let point X1 have an I and
Q value of (I.sub.1, Q.sub.1) and a feedback value of Y.sub.1.
[0178] In block 2525, with a preset step size, "STEP," (e.g.,
STEP=most significant bit (MSB) of the I-value or Q-value or the
MSB/2 or the MSB/4) the controller 235 selects another two points
around point X1 by varying the I-value. For example, the controller
235 may select points X2 (e.g., I.sub.1+STEP, Q.sub.1) and X3
(e.g., I.sub.1-STEP, Q.sub.1). The controller 235 communicates the
samples X2 and X3 to the I/Q modulator 230 and the I/Q modulator
230 applies the settings for the samples X2 and X3 one at a time.
For each sample, the controller 235 receives a feedback value, for
example from the receiver 135. Let the feedback value for X2 be
Y.sub.+ and the feedback value for X3 be Y.sub.-.
[0179] In block 2530, the controller 235 computes another sample
point based on dual slope. In certain exemplary embodiments, the
controller 235 computes another sample using
SLOPE=(Y.sub.+-Y.sub.1)/STEP. This equation represents the slope of
a straight line 2610 connecting points X2 and X1. Another straight
line 2615 is illustrated in FIG. 26 extending from point X3 and
having a slope opposite the line 2610. The lines 2610 and 2615
intersect at point 2620.
[0180] In block 2530, the controller computes the next I-value for
point 2620 using:
I.sub.2=I.sub.1-STEP*(Y.sub.+-Y.sub.-)/(Y.sub.+-Y.sub.1). In block
2535, the controller communicates I and Q values of (I.sub.2,
Q.sub.1) to the I/Q modulator 230 and the I/Q modulator 230 applies
the I and Q values. In block 2540, the controller 235 receives a
feedback value for (I.sub.2, Q.sub.1) and stores the feedback value
in the memory device 760. Let the feedback value for (I.sub.2,
Q.sub.1) be Y.sub.2.
[0181] In block 2545, with the preset step size, "STEP," the
controller 235 selects another two points around point X1 by
varying the Q-value from point (I.sub.2, Q.sub.1). For example, the
controller 235 may select points (I.sub.2, Q.sub.1+STEP) and
(I.sub.2, Q.sub.1-STEP). The controller 235 communicates the
samples to the I/Q modulator 230 and the I/Q modulator 230 applies
the settings for the samples one at a time. For each sample, the
controller 235 receives a feedback value. Let the feedback value
for (I.sub.2, Q.sub.1+STEP) be Y.sub.+ and the feedback value for
(I.sub.2, Q.sub.1-STEP) be Y.sub.-.
[0182] In block 2550, the controller 235 computes:
Q2=Q1-STEP*(Y.sub.+-Y.sub.-)/(Y.sub.+-Y.sub.2). In block 2555, the
controller 235 communicates I and Q values of (I.sub.2, Q.sub.2) to
the I/Q modulator 230 and the I/Q modulator 230 applies the I and Q
values. In block 2560, the controller 235 receives a feedback value
for (I.sub.2, Q.sub.2) and stores the feedback value in the memory
device 760. Let the feedback value for (I.sub.2, Q.sub.2) be
Y3.
[0183] In block 2565, the controller 235 reduces the size of STEP.
In this exemplary embodiment, the size of STEP is halved. However,
other (e.g., less conservative) reduction sizes are also feasible.
In block 2570, the controller 235 conducts an inquiry to determine
whether the size of STEP is less than a threshold, "STEP.sub.END."
If the size of STEP is less than STEP.sub.END, then the DSA 2500
proceeds to block 2580, where the controller 235 initiates an MSA
(e.g., MSA 2200) using (I.sub.2, Q.sub.2) as a starting point. If
the size of STEP is not less than STEP.sub.END, then the method
2500 proceeds to block 2575. In block 2575, the controller 235
assigns the I2, Q2, and Y2 values to I1, Q1, and Y1, respectively.
After block 2575, the DSA 2500 returns to block 2525.
[0184] The exemplary DSA 2500 can be particularly useful when there
are local preferred cancellation points with one global preferred
cancellation point. The local preferred cancellation points refer
to I and Q values where their feedback values are "locally"
preferred. For example, an MSA, such as MSA 2200, would not jump
outside the area proximal to the local preferred cancellation
point. Implementing the DSA 2500 for upper bits, the controller 235
could avoid getting stuck with those local preferred cancellation
points, while the MSA 2200 could finely tune to find the globally
preferred cancellation point.
[0185] FIG. 27 is a flow chart depicting a TSA 2700 for canceling
noise and/or interference, in accordance with certain exemplary
embodiments. FIG. 28 is a graph 2800 depicting cancellation points
along an I-Q plane 2801 evaluated in an implementation of the TSA
of FIG. 27, in accordance with certain exemplary embodiments.
Referring to FIGS. 27 and 28, in block 2705, the controller 235
selects a number (e.g., 4) of samples in the I-Q plane 2801. In
certain exemplary embodiments, the controller 235 uses the BSA to
select the location in the I-Q plane 2801 for the samples.
[0186] In block 2710, the controller 235 communicates the settings
for the selected samples to the I/Q modulator 230 and the I/Q
modulator 230 applies the settings for each sample one at a time.
In block 2715, the controller 235 receives, for each sample, a
feedback value (e.g., from the receiver 135) and stores the
feedback value and its corresponding setting in the memory device
760. In block 2720, the controller 235 compares the feedback value
and identifies the better or preferred feedback value. Let X1 in
FIG. 28 be the sample resulting in the preferred feedback
value.
[0187] In block 2725, with a predetermined step size, "STEP,"
(e.g., STEP=MSB/2 or MSB/4) the controller 235 selects another four
samples proximal to X1. For example, the controller 235 may select
(I.sub.1+STEP, Q.sub.1), (I.sub.1-STEP, Q.sub.1), (I.sub.1,
Q.sub.1+STEP), and (I.sub.1, Q.sub.1-STEP). The controller 235
communicates the four settings to the I/Q modulator 230 and the I/Q
modulator 230 applies the settings for each sample one at a time.
The controller 235 receives a feedback value for each sample and
stores the feedback value for each sample and the settings for each
sample in the memory device 760. The controller 235 compares the
feedback values for the four samples and identifies the preferred
feedback value. Let X2 in FIG. 28 be the sample resulting in the
preferred feedback value.
[0188] In block 2730, the controller 235 reduces the size of STEP.
In this exemplary embodiment, the size of STEP is halved. Other
size reductions are also feasible. In block 2735, the controller
235 conducts an inquiry to determine whether the size of STEP is
less than a threshold, "STEP.sub.END." If the size of STEP is less
than STEP.sub.END, then the TSA 2700 proceeds to block 2755, where
the controller 235 uses the setting (I.sub.n+1, Q.sub.n+1) to
control the I/Q modulator 230. If only one iteration of the TSA is
performed, then the controller 235 uses the settings for the sample
corresponding to the preferred feedback value in block 2725 to
control the I/Q modulator 230. If the size of STEP is not less than
STEP.sub.END, then the TSA 2700 proceeds to block 2740.
[0189] In block 2740, the controller 235 selects another four
samples proximal to the sample having the best stored feedback
value. If it is the first iteration, the sample is X2 with
(I.sub.2, Q.sub.2). This sample is designated as Xn as block 2740
may be executed multiple times. For example, the controller 235
selects samples (I.sub.n+STEP, Q.sub.n), (I.sub.n-STEP, Q.sub.n),
(I.sub.n, Q.sub.n+STEP), (I.sub.n, Q.sub.n-STEP). The controller
235 communicates the four settings to the I/Q modulator 230 and the
I/Q modulator 230 applies the settings for each sample one at a
time. The controller 235 receives a feedback value for each sample
and stores the feedback value for each sample and the settings for
each sample in the memory device 760. In block 2745, the controller
235 compares the feedback values for the four samples and
identifies the preferred feedback value. In block 2750, the
controller 235 reduces the size of STEP and the TSA returns to
block 2735. FIG. 28 illustrates the TSA 2700 employing four
iterations, represented by points X1-X4, that identifies a
preferred cancellation point 2815 in the I-Q plane 2801.
[0190] The exemplary TSA 2700 can be particularly useful when
searching for an improved cancellation point and corresponding I/Q
setting based on a previously preferred cancellation point, for
example in response to a change in temperature. In such scenarios,
the block 2705 can be adapted to use the previous preferred I/Q
setting rather than selecting four samples. The TSA 2700 can narrow
the field of search to the area in the I-Q plane 2801 near the
previously preferred cancellation point.
[0191] Individual algorithms (e.g., BSA, FBA, BCA, MSA, DSA, and
TSA) discussed above may be implemented as a standalone algorithm
to decide acceptable I and Q values. Or, multiple ones of the
algorithms can be employed together to increase the speed of the
evaluation and attain a desired accuracy. For example, the BSA can
be executed to determine either the first MSB or first MSB and
second MSB of both I and Q values. Following the BSA, the FBA or
BCA can be executed to determine the middle few bits of both I and
Q values. Finally, the MSA can be executed to finely tune both I
and Q values to achieve a better feedback value and thus, better
noise or interference cancellation.
[0192] Multiple iterations of the algorithms can be executed and/or
the algorithms can be executed for longer periods of time to
achieve better results. In certain exemplary embodiments,
algorithms used for fine tuning (e.g., MSA and TSA) are employed in
an always "on" mode where the controller 150 continues to execute
the algorithms while the noise canceller is in normal operation.
This enables the controller 150 to adjust the settings of the noise
canceller to account for environmental changes, such as changes in
temperature or operating conditions. In addition, noise cancellers
operating in parallel can each execute one or more of the
algorithms simultaneously or sequentially.
[0193] FIG. 29 is a flow chart depicting a method 2900 for finding
a preferred noise cancellation point for two noise cancellers
disposed in a communication system, such as the communication
system 100, in accordance with certain exemplary embodiments. For
example, the communication system 100 may include, in alternative
embodiments, two HIPCF cancellers 130 in parallel.
[0194] In block 2905, a control device, such as the controller 235
of one of two HIPCF canceller 130, arranges the (I, Q) settings for
the two cancellers in a sequence. For example, this sequence may be
arranged as: (IninQnqn . . . I0i0Q0q0) with In . . . I0 and Qn . .
. Q0 designating (I,Q) settings for a first canceller and in . . .
i0 and qn . . . q0 designating IQ settings for a second canceller.
The control device can then treat the two cancellers as a single
canceller having the arranged sequence.
[0195] In block 2910, the control device executes one or more of
the cancellation algorithms discussed above (e.g., BSA, FBA, BCA,
MSA, DSA, or TSA) using the sequence to determine a preferred
cancellation setting for the cancellers. In block 2915, the control
device stores the preferred cancellation settings in memory.
[0196] FIG. 30 is a flow chart depicting an alternative method 3000
for finding a preferred noise cancellation point for two noise
cancellers disposed in a communication system, in accordance with
certain exemplary embodiments. In block 3005, a control device,
such as the controller 235 of one of the two noise cancellers,
finds a preferred cancellation point for one of the two noise
cancellers while the settings for the second of the two noise
cancellers remain unchanged. One of the cancellation algorithms
discussed above (e.g., BSA, FBA, BCA, MSA, DSA, or TSA) can be used
to find the preferred noise cancellation point for the first noise
canceller.
[0197] In block 3010, the control device finds a preferred
cancellation point for the second noise canceller using one or more
of the cancellation algorithms (e.g., BSA, FBA, BCA, MSA, DSA, or
TSA) while the settings for the first noise canceller remain
unchanged at the preferred cancellation point found during
execution of block 3005. In block 3015, with both cancellers
operating using their respective preferred cancellation points, the
control device obtains a feedback value resulting from the two
noise cancellers. In block 3020, the control device compares the
obtained feedback value to a preset threshold value. If the
feedback value is better than the threshold or the method 3000 has
ran for more than a preset number of iterations, the method 3000
proceeds to block 3025. Otherwise, the method returns to block 3005
with the current (I, Q) settings for both cancellers as starting
values for the algorithm(s). In block 3025, the control device
stores the settings for the two noise cancellers and controls the
noise cancellers using the settings.
[0198] FIG. 31 is a flow chart depicting an alternative method 3100
for finding a preferred noise cancellation point for two noise
cancellers disposed in a communication system, in accordance with
certain exemplary embodiments. This method 3100 addresses
implantations where two noise cancellers are used to increase
cancellation bandwidth.
[0199] In block 3105, a control device, such as the controller 235
of one of the two noise cancellers, finds a preferred cancellation
setting (e.g., (I, Q) settings) for the first of the two noise
cancellers based on a feedback value for a lower portion of
bandwidth while the second noise canceller is turned off. In block
3110, the control device stores the preferred noise cancellation
setting for the first noise canceller.
[0200] In block 3115, the control device finds a preferred
cancellation setting (e.g., (I, Q) settings) for the second of the
noise cancellers based on a feedback value for an upper portion of
bandwidth while the first noise canceller is turned off. In block
3120, the control device stores the preferred noise cancellation
setting for the second noise canceller.
[0201] In block 3125, the control device turns both noise
cancellers on and applies the respective preferred cancellation
setting to each of the two noise cancellers. In block 3130, the
control device executes an MSA for one step on the first noise
canceller for the lower portion of the bandwidth. In block 3135,
the control device executes an MSA for one step on the second noise
canceller for the upper portion of the bandwidth.
[0202] In block 3140, the control device obtains a feedback value
for the noise cancellers and compares the feedback value to a
preset value. If the feedback value is greater than the preset
value or if blocks 3130 and 3135 have been executed for more than a
preset number of iterations, the method 3100 proceeds to block
3145. Otherwise, the method 3100 returns to block 3130. In block
3145, the control device stores the final settings in memory and
controls the noise cancellers using the final settings. Although
the methods 2900, 3000, 3100 are depicted and described in terms of
determining preferred cancellation points for two noise cancellers,
each method 2900, 3000, 3100 could also be employed to determine
preferred cancellation points for any number of noise cancellers.
For example, the methods 2900, 3000, 3100 could be employed to find
preferred cancellation points for three or more noise cancellers
arranged in parallel. Although the methods 2900, 3000, and 3100 are
depicted to find the preferred or improved cancellation points for
two noise cancellers, each method 2900, 3000, 3100 could also be
employed to find the preferred or improved cancellation points for
more than two noise cancellers, for example three or more noise
cancellers.
[0203] In summary, a communication system in accordance with
certain exemplary embodiments of the present invention can comprise
a transmitter that communicates information at a first frequency, a
receiver that receives communication signals at a second frequency
that may be the same or near the first frequency, and an
interference suppression device that cancels, corrects, addresses,
or compensates for interference, EMI, noise, spurs, or other
unwanted spectral components imposed onto the receiver by signals
transmitted by the transmitter. The interference suppression device
can be coupled to a transmit path of the transmitter (e.g., at the
output of the transmitter's power amplifier) to obtain a sample of
the transmitted signals. The interference compensation circuit can
include a plurality of filters, such as band-pass filters, that
block or suppress signals outside the frequency band of the
receiver while passing noise or other interference signals within
the frequency band of the receiver. The interference compensation
circuit also can include an I/Q modulator that generates an
interference compensation signal using the signal output by the
filters. This interference compensation signal can have an
amplitude the same as or close to the amplitude of the noise and a
phase shift of 180 degrees relative to interference. These
parameters are tuned in using a "receive signal quality indicator"
feedback from the victim receiver. The interference compensation
signal generated by the I/Q modulator is applied to a receive path
of the receiver to cancel or suppress the interference imposed on
the receiver by the transmitted signals.
[0204] The communication systems described herein can be embodied
in various communication devices, including cellular telephones,
mobile computers, PDAs, personal navigation devices (e.g., GPS
devices), or any other communication device comprising two or more
communication elements. For example, the communication system can
be embodied in a smartphone having a LTE/CDMA/GSM transceiver and a
mobile TV tuner. Another example is a smartphone having a
GSM/PCS/DCS/W-CDMA transceiver and a GPS receiver. Yet another
example includes a notebook computer having a WLAN transceiver and
a WiMAX or Bluetooth transceiver.
[0205] In a mobile device embodiment, the two or more communication
elements may communicate via two or more antennas with little
spatial separation. Thus, signals transmitted by the two or more
communication elements may impose interference on each other. To
suppress or cancel this interference, a HIPCF canceller as
described above can be employed in each communication direction.
That is, a first HIPCF canceller can cancel or suppress
interference imposed on a first of the two or more communication
elements by a second of the two or more communication elements,
while a second HIPCF canceller cancels or suppresses interference
imposed on the second of the two or more communication elements by
the first of the two or more communication elements. Certain
components of both HIPCF cancellers can be fabricated on a single
integrated circuit or on multiple integrated circuits, such as one
or more CMOS circuits.
[0206] Embodiments of the invention can be used with computer
hardware and software that perform the methods and processing
functions described above. As will be appreciated by those skilled
in the art, the systems, methods, and procedures described herein
can be embodied in a programmable computer, computer executable
software, or digital circuitry. The software can be stored on
computer readable media. For example, computer readable media can
include a floppy disk, RAM, ROM, hard disk, removable media, flash
memory, memory stick, optical media, magneto-optical media, CD-ROM,
etc. Digital circuitry can include integrated circuits, gate
arrays, building block logic, field programmable gate arrays
("FPGA"), etc.
[0207] Although specific embodiments of the invention have been
described above in detail, the description is merely for purposes
of illustration. It should be appreciated, therefore, that many
aspects of the invention were described above by way of example
only and are not intended as required or essential elements of the
invention unless explicitly stated otherwise. Various modifications
of, and equivalent acts corresponding to, the disclosed aspects of
the exemplary embodiments, in addition to those described above,
can be made by a person of ordinary skill in the art, having the
benefit of the present disclosure, without departing from the
spirit and scope of the invention defined in the following
claim(s), the scope of which is to be accorded the broadest
interpretation so as to encompass such modifications and equivalent
structures.
* * * * *