U.S. patent application number 12/966594 was filed with the patent office on 2011-04-07 for orthogonal spread-spectrum waveform generation with non-contiguous spectral occupancy for use in cdma communications.
This patent application is currently assigned to Telefonaktiebolaget LM Ericsson (publ). Invention is credited to Paul W. DENT.
Application Number | 20110080936 12/966594 |
Document ID | / |
Family ID | 40072283 |
Filed Date | 2011-04-07 |
United States Patent
Application |
20110080936 |
Kind Code |
A1 |
DENT; Paul W. |
April 7, 2011 |
ORTHOGONAL SPREAD-SPECTRUM WAVEFORM GENERATION WITH NON-CONTIGUOUS
SPECTRAL OCCUPANCY FOR USE IN CDMA COMMUNICATIONS
Abstract
The technology in this application spreads a signal over an
available discontinuous spectrum, such as a radio frequency band,
so that the spread signal only occupies the non-contiguous
spectrum. In this way, CDMA transmission and reception can be used
in a fragmented or non-contiguous spectrum that otherwise would not
be useable for direct sequence spreading. Spreading over
non-contiguous portions of spectrum is preferably performed without
producing unacceptable interference in portions of unavailable
spectrum located between the allowed spectrum. By avoiding
unacceptable interference in portions of unavailable spectrum
located between the allowed spectrum, the unavailable spectrum may
be used by other users or services.
Inventors: |
DENT; Paul W.; (Pittsboro,
NC) |
Assignee: |
Telefonaktiebolaget LM Ericsson
(publ)
Stockholm
SE
|
Family ID: |
40072283 |
Appl. No.: |
12/966594 |
Filed: |
December 13, 2010 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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11929268 |
Oct 30, 2007 |
7864663 |
|
|
12966594 |
|
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60940129 |
May 25, 2007 |
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Current U.S.
Class: |
375/143 ;
375/147; 375/152; 375/E1.002 |
Current CPC
Class: |
H04B 1/7097 20130101;
H04J 13/10 20130101; H04B 2201/709709 20130101 |
Class at
Publication: |
375/143 ;
375/147; 375/152; 375/E01.002 |
International
Class: |
H04B 1/707 20110101
H04B001/707 |
Claims
1. A method for use in a receiver for communicating information
using code division multiple access signals that occupy
predetermined non-contiguous regions of the electromagnetic
spectrum, comprising: receiving a signal waveform in the time
domain containing a non-contiguous spectrum signal; transforming
the one or more components of the time domain signal waveform
components to produce contiguous spectral components; selecting
from the contiguous spectral components those components
corresponding to the predetermined non-contiguous regions to obtain
non-contiguous spectral components; processing the non-contiguous
spectral components to produce one or more received symbol values;
and decoding received symbol values to reproduce the
information.
2. The method in claim 1, wherein the received signal includes
repeated, shaped signal blocks modulated with a data symbol, the
method further comprising: using a matched filter to weight each
signal block with a corresponding shaping function, adding
corresponding samples of the weighted blocks to generate a matched
filtered block corresponding to the non-contiguous spectrum
waveform time samples, and transforming the matched filtered block
to produce the contiguous spectral components.
3. The method in claim 2, wherein the processing includes weighting
and combining the non-contiguous spectral components to produce one
or more received symbol values.
4. The method in claim 3, wherein the weighting and combining
includes weighting the non-contiguous spectral components using a
complex conjugate of a weighting applied at a transmitter that
transmits the information, wherein the weighting applied at the
transmitter is a vector of weighting values selected from a set of
orthogonal codes, and wherein the method further comprises:
performing a first weighting and combining using the complex
conjugate of a first vector of weighting values to produce a first
data symbol value, and performing a second weighting and combining
using the complex conjugate of a second vector of weighting values
to produce a second data symbol value.
5. The method in claim 4, wherein the weighting further comprises
weighting non-contiguous spectral samples using an estimate of
phase changes of each sample induced by a propagation path from the
transmitter to the receiver.
6. Apparatus for use in a receiver for communicating information
using code division multiple access signals that occupy
predetermined non-contiguous regions of the electromagnetic
spectrum, comprising electronic circuitry configured to: receive a
signal waveform in the time domain containing a non-contiguous
spectrum signal; transform the one or more components of the time
domain signal waveform components to produce contiguous spectral
components; select from the contiguous spectral components those
components corresponding to the predetermined non-contiguous
regions to obtain non-contiguous spectral components; process the
non-contiguous spectral components to produce one or more received
symbol values; and decode received symbol values to reproduce the
information.
7. The apparatus in claim 6, wherein the received signal includes
repeated, shaped signal blocks modulated with a data symbol, and
wherein the electronic circuitry is configured to: use a matched
filter to weight each signal block with a corresponding shaping
function, add corresponding samples of the weighted blocks to
generate a matched filtered block corresponding to the
non-contiguous spectrum waveform time samples, and transform the
matched filtered block to produce the contiguous spectral
components.
8. The apparatus in claim 7, wherein the processing includes
weighting and combining the non-contiguous spectral components to
produce one or more received symbol values.
9. The apparatus in claim 8, wherein the weighting and combining
includes weighting the non-contiguous spectral components using a
complex conjugate of a weighting applied at a transmitter that
transmits the information.
10. The apparatus in claim 9, wherein the weighting applied at the
transmitter is a vector of weighting values selected from a set of
orthogonal codes, the electronic circuitry being further configured
to: perform a first weighting and combining using the complex
conjugate of a first vector of weighting values to produce a first
data symbol value, and perform a second weighting and combining
using the complex conjugate of a second vector of weighting values
to produce a second data symbol value.
11. The apparatus in claim 10, wherein the weighting further
comprises weighting non-contiguous spectral samples using an
estimate of phase changes of each sample induced by a propagation
path from the transmitter to the receiver
Description
PRIORITY APPLICATIONS
[0001] This application is a continuation of U.S. application Ser.
No. 11/11/929,268, filed on Oct. 30, 2007, which claims priority
from U.S. provisional application Ser. No. 60/940,129, filed on May
25, 2007, the contents of which are incorporated herein by
reference.
TECHNICAL FIELD
[0002] The technical field is communications. The technology
described below is related to Code Division Multiple Access (CDMA).
Non-limiting example applications include multiple-access
communications systems such as cellular and satellite communication
systems.
BACKGROUND
[0003] Direct sequence spread spectrum used in CDMA systems
generates a pseudorandom code and multiplicatively modulates a
narrowband, data-modulated signal with the pseudorandom code to
spread it over the radio frequency (RF) spectrum. Other signals
spread with different, preferably orthogonal, codes may overlap in
frequency and time and yet may be distinguished at a receiver
designed to demodulate or despread a particular signal using its
particular pseudorandom code. The frequency spectrum occupied by
the spread spectrum signal is contiguous meaning that the
transmitted signal occupies a particular frequency band of the
spectrum continuously across that frequency band; there are no
portions of the spectrum in that frequency band that are not
substantially occupied by the spread signal. That continuous
occupation across the spectrum is an unavoidable property of a
pseudorandom code because the spectrum is the Fourier transform of
the pseudorandom code's autocorrelation function. By definition,
pseudorandom code has an Dirac or unit impulse response function
autocorrelation, so its Fourier transform is flat and continuous in
the frequency domain.
[0004] Frequency hopping (FH) spread spectrum uses non-contiguous
spectrum with all of the signal energy appearing instantaneously in
one FH channel. But that channel moves rapidly from
instant-to-instant or hop-to-hop. The time during which the energy
appears in any one FH channel is called the "dwell time". The
interference characteristics of frequency hopping systems to
themselves and to or from other systems is therefore quite
different than Direct Sequence (DS) CDMA systems. Frequency hopping
transmissions tend to create, and be tolerant of high levels of
momentary, narrowband interference with a low probability of
incidence. Direct sequence transmissions create, and are tolerant
of, low levels of continuous, wideband interference. In scenarios
where there are large distance variations between wanted and
unwanted transmitters and receivers, called the "high near/far
ratio" scenario, frequency hopping generally provides superior
performance. In a scenario where wanted and unwanted radios are at
equal distance from each other, such as satellite systems that have
dedicated spectrum not shared with other terrestrial services,
direct sequence may provide better performance. Direct sequence
spread spectrum (DSSS) is also used in many modern wireless
communications systems, like cellular systems, where near/far
ratios may be managed by commanding a mobile terminal to
communicate via the nearest base station.
[0005] Until now, it has only been possible to use direct sequence
spread spectrum with a contiguous spectrum. For example, direct
sequence could not be used by licensees who, for historical or
other reasons, found themselves with a fragmented or non-contiguous
spectrum. Non-contiguous means that the transmitted signal
simultaneously occupies multiple portions of the spectrum across a
particular frequency band so that there are portions of the
spectrum in that band that are not occupied by the spread
signal.
[0006] Another problem concerns multi-path propagation. When signal
echoes are received via delayed paths, and the echo path delay is
more than a small fraction of the duration of a modulation symbol,
multi-path distortion is produced in which successive symbols
appear to be mixed. This multi-path distortion effect is also known
as Intersymbol Interference (ISI). Signals suffering from ISI may
be decoded using one of the various types of equalizers, such as
the Viterbi Maximum Likelihood Sequence Estimation algorithm, a
transversal equalizer, a decision feedback equalizer, or hybrids
thereof. A RAKE receiver for direct sequence spread spectrum
signals performs a similar function but in a different way.
[0007] Other solutions to multi-path distortion include splitting a
data stream into multiple, slower speed data streams, the symbol
duration of each now being much longer than the multi-path echo
delay, and transmitting the streams in parallel over separate
channels. The separate channels are contiguous, narrowband channels
within an allocated bandwidth, and this type of transmission was
variously known as "multi-tone modem modulation," "Kineplex," and
when the channel spacing of the narrowband channels bears a
particular relationship to the data rate on each channel,
Orthogonal Frequency Division Multiplexing (OFDM) of the separate
channels. OFDM is not spread spectrum because the total bandwidth
occupied is commensurate with the total underlying data rate. Two
OFDM signals should not overlap as they do not have a
distinguishing spread-spectrum code that would allow a receiver to
separately distinguish them.
[0008] In any event, it would be desirable to use Spread Spectrum
Code Division Multiple Access systems in non-contiguous spectrum
and to be able to reduce or minimize adverse multi-path effects on
such non-contiguous spectrum communications.
SUMMARY
[0009] The technology in this application solves these problems
(and others) and meets the desirable goals identified above (and
others). The technology spreads a signal over an available
discontinuous spectrum, such as a radio frequency band, so that the
spread signal substantially only occupies the non-contiguous
spectrum. As a result, the signal energy in unavailable spectrum
portions is generally below a predetermined maximum, which in some
cases might be a very low maximum. In this way, CDMA transmission
and reception can be used in a fragmented or non-contiguous
spectrum that otherwise would not be useable for direct sequence
spreading. In the domain of electromagnetic radiation, a spectrum
is a series of radiant energies arranged in order of wavelength or
of frequency. Although the technology applies to any type of
spectrum, for purposes of illustration and not limitation, a
frequency spectrum is used in the example embodiments below.
[0010] Spreading over non-contiguous portions of spectrum is
preferably performed without producing unacceptable interference in
portions of unavailable spectrum located between the allowed
spectrum. By avoiding unacceptable interference in portions of
unavailable spectrum located between the allowed spectrum, the
unavailable spectrum may be used by other users or services. Even
though multiple instances of such signals spread over the
non-contiguous spectrum may overlap, they can be distinguished
using different spread-spectrum codes. In contrast with OFDM, which
must perform a fast Fourier transform (FFT) for each new block of
data symbols, the non-contiguous spectrum spreading waveform
obtained by performing just one inverse Fourier transform on a
spreading codeword corresponding to a spread data symbol may be
repeated over and over again, each time being modulated with
different, successive, underlying data symbols. The inverse Fourier
transform of the non-contiguous spectrum therefore only needs to be
performed once at the transmitter and the resulting non-contiguous
spectrum spreading waveform stored.
[0011] In one example embodiment, a predetermined, non-contiguous
spectrum is generated having non-zero components in permitted or
available regions of the electromagnetic spectrum and zero
components in non-permitted or non-available regions of the
electromagnetic spectrum. A non-contiguous spectrum spreading
signal associated with the non-contiguous spectrum is produced. In
a preferred but example implementation, samples of the shaped
non-contiguous spectrum spreading signal may be stored in memory
for repeated use. The information is processed into a stream of
data symbols. The non-contiguous spectrum spreading signal and a
data symbol from the stream are combined to produce a data
modulated signal. The data modulated signal is then modulated onto
a carrier signal having a desired center frequency to produce a
signal for transmission and transmitted. Data modulated signals
produced by successive data symbols from the stream may be
overlapped to produce a transmission signal.
[0012] In one example implementation, producing the non-contiguous
spectrum spreading signal includes performing an inverseDiscrete
Fourier Transformation on the non-contiguous spectrum to produce a
sequence of time waveform samples. Combining the non-contiguous
spectrum spreading signal and a data symbol from the stream to
produce a data modulated signal includes multiplying the data
symbol value with each sample of the sequence of time waveform
samples. A block of signal samples corresponding to the
non-contiguous spectrum spreading signal is preferably repeated.
Successive samples from the repeated blocks are weighted using a
shaping function to produce a shaped, non-contiguous spectrum
spreading signal used in the combining. Samples of the resulting
shaped non-contiguous spectrum spreading signal are stored in
memory.
[0013] In another example implementation, producing the
non-contiguous spectrum spreading signal includes allocating a
complex number to represent the amplitude and phase of each
non-zero component of the non-contiguous spectrum and a zero value
to represent the zero spectral components. Combining the
non-contiguous spectrum spreading signal and a data symbol from the
stream to produce a data modulated signal includes multiplying the
data symbol value with each complex number to generate a set of
products and performing an inverse Discrete Fourier Transform on
the set of products to produce the data modulated signal. A block
of signal samples corresponding to the data modulated signal is
preferably repeated. Successive samples from the repeated blocks
are weighted using a shaping function to produce a shaped, data
modulated signal. Shaped data modulated signals produced by
successive data symbols from the stream are overlapped to produce a
transmission signal.
[0014] In another example implementation, a sequence or code from a
set of orthogonal codes is selected and used to generate the
non-contiguous spectrum and to determine a phase of successive
non-zero spectral components. A first data modulated signal is
produced using a first code from the set of orthogonal codes. At
least a second data modulated signal is produced using a second
code from the set of orthogonal codes. The first and second data
modulated signals are combined to produce a composite modulating
signal, which is used in modulating the data modulated signal on to
the carrier signal.
[0015] At the receiver, a signal waveform in the time domain
including a non-contiguous spectrum signal is received. One or more
components of the time domain signal waveform is transformed to
produce contiguous spectral components. From the contiguous
spectral components, those components that correspond to the
predetermined, non-contiguous regions of the spectrum are selected
to obtain non-contiguous spectral components. The non-contiguous
spectral components are processed to produce one or more received
symbol values. The symbol values are then decoded to reproduce the
information.
[0016] In one example implementation, the received signal includes
repeated, shaped signal blocks modulated with a data symbol. Each
signal block may be weighted with a corresponding shaping function.
Corresponding samples of the weighted blocks are added to generate
a matched filtered block corresponding to the non-contiguous
spectrum waveform time samples. The matched filter block is
transformed to produce the contiguous spectral components.
[0017] The processing function may include weighting and combining
the non-contiguous spectral components to produce one or more
received symbol values. The weighting and combining may include
weighting the non-contiguous spectral components using a complex
conjugate of a weighting applied at a transmitter that transmits
the information. The weighting can further include weighting
non-contiguous spectral samples using an estimate of phase changes
of each sample induced by a propagation path from the transmitter
to the receiver. Such weighting permits information decoding when
the propagation path includes interference from the same
transmitter and interference from a different transmitter.
[0018] The weighting applied at the transmitter may be, for
example, a vector of weighting values selected from a set of
orthogonal codes. A first weighting and combining is performed
using the complex conjugate of a first vector of weighting values
to produce a first data symbol value and a second weighting and
combining is performed using the complex conjugate of a second
vector of weighting values to produce a second data symbol
value.
BRIEF DESCRIPTION OF THE DRAWINGS
[0019] Various objects, features, and advantages will be apparent
from the following more particular description of the accompanying
drawings in which reference characters refer to the same parts
throughout the various views.
[0020] FIG. 1 is a flowchart diagram illustrating non-limiting
example procedures for generating and combining a non-contiguous
spectrum spreading waveform with data symbols for radio
transmission;
[0021] FIG. 2 is a flowchart diagram illustrating non-limiting
example procedures for receiving and despreading a signal spread
over a non-contiguous spectrum;
[0022] FIGS. 3A and 3B illustrate the spectral response resulting
from truncating the time domain signal using a shaping
function;
[0023] FIG. 4 illustrates in graphical form a receiver processing
using a matched filter with the same waveform shaping function used
in the transmitter;
[0024] FIGS. 5A-5E show an non-limiting example construction of a
non-contiguous spectrum spreading waveform;
[0025] FIG. 6 is a function block diagram of a non-limiting example
apparatus for combining a spreading waveform for a non-contiguous
spectrum with data symbols for radio transmission; and
[0026] FIG. 7 is a function block diagram of a non-limiting example
apparatus for implementing a correlation method at a radio
receiver; and
DETAILED DESCRIPTION
[0027] In the following description, for purposes of explanation
and not limitation, specific details are set forth such as
particular architectures, interfaces, techniques, etc. in order to
provide a thorough understanding. However, it will be apparent to
those skilled in the art that other embodiments that depart from
these specific details may be employed. That is, those skilled in
the art will be able to devise various arrangements which, although
not explicitly described or shown herein, embody the principles of
the invention and are included within its spirit and scope. In some
instances, detailed descriptions of well-known devices, circuits,
and methods are omitted so as not to obscure the description with
unnecessary detail. All statements herein reciting principles,
aspects, and embodiments, as well as specific examples thereof, are
intended to encompass both structural and functional equivalents
thereof. Additionally, it is intended that such equivalents include
both currently known equivalents as well as equivalents developed
in the future, i.e., any elements developed that perform the same
function, regardless of structure.
[0028] Thus, for example, it will be appreciated by those skilled
in the art that block diagrams herein can represent conceptual
views of illustrative circuitry embodying the principles of the
technology. Similarly, it will be appreciated that any flow charts,
state transition diagrams, pseudocode, and the like represent
various processes which may be substantially represented in
computer readable medium and so executed by a computer or
processor, whether or not such computer or processor is explicitly
shown.
[0029] The functions of the various elements including functional
blocks labeled or described as "processors" or "controllers" may be
provided through the use of dedicated hardware as well as hardware
capable of executing software in association with appropriate
software. When provided by a processor, the functions may be
provided by a single dedicated processor, by a single shared
processor, or by a plurality of individual processors, some of
which may be shared or distributed. Moreover, explicit use of the
term "processor" or "controller" should not be construed to refer
exclusively to hardware capable of executing software, and may
include, without limitation, digital signal processor (DSP)
hardware, read only memory (ROM) for storing software, random
access memory (RAM), and non-volatile storage.
[0030] A vector of values is selected from a limited alphabet of
values, e.g. the binary values +1 and -1, so that the vector is
orthogonal to other vectors in a set of mutually orthogonal
vectors. The values of the vector are regarded as the signs or
multiplier coefficients of corresponding sine or cosine waveforms.
Each sine or cosine waveform has a frequency centered on an allowed
portion of spectrum, such as a frequency channel, in an allocated
band of frequency spectrum. When the allowed portions of spectrum
are non-contiguous, zero coefficients are inserted between elements
of the vector corresponding to the locations of the non-allowed or
unavailable portions of spectrum. The resulting vector, expanded by
the insertion of zeros, is then inverse Fourier transformed to
produce a waveform, the spectrum of which is substantially
constrained to the allowed portions of spectrum. In one
non-limiting implementation, pulse-shaped OFDM may be used to limit
undesired truncation effects on the spread signal waveform in the
allowed portions of spectrum. In any event, the non-contiguous
spectrum spreading waveform is used to multiply narrowband
data-modulated waveforms to obtain a spread-spectrum signal that is
only spread over the allowed portions of the spectrum, thereby
producing a non-spectrally-contiguous CDMA signal.
[0031] Other vectors from the set may be selected to produce
orthogonal CDMA signals, while still being spectrally constrained
to the allowed portions of the spectrum. Non-limiting example
variations in the order of combining the spectral vector
sine/cosine waveforms, the orthogonal vectors, and the data streams
to be transmitted are described below that have advantages in
particular applications.
[0032] Table 1 below illustrates a non-limiting example
construction of a number of mutually orthogonal codewords in the
spectral domain, referred to as spectral vectors, that occupy
non-contiguous spectrum.
TABLE-US-00001 TABLE 1 +1 +1 +1 0 0 +1 +1 0 +1 +1 +1 +1 +1 +1 0 0
+1 -1 0 -1 -1 -1 +1 +1 -1 0 0 -1 +1 0 +1 -1 -1 +1 +1 -1 0 0 -1 -1 0
+1 +1 -1 +1 -1 +1 0 0 -1 +1 0 -1 +1 -1 +1 -1 +1 0 0 -1 -1 0 +1 -1
+1 +1 -1 -1 0 0 +1 +1 0 -1 -1 +1 +1 -1 -1 0 0 +1 -1 0 +1 +1 -1
[0033] The location of the zero values indicates spectrum that
should not be occupied by the spread signal, while the non-zero
values indicate spectral locations that will be occupied. Examining
only the non-zero values, it can be seen that each row is a code of
the Walsh-Hadamard set of order 8. Other orthogonal sets may be
used. For example, sets of orthogonal Fourier sequences may be
constructed for any desired number of non-zero points, while
Walsh-Hadamard codes are a length equal to a power of two, such as
the length 8 codes exemplified above. It is also possible to
scramble orthogonal code sequences. For example, Walsh-Hadamard
codes may be scrambled by multiplying corresponding elements of
each code by the same complex factor of magnitude unity. The order
of the elements of each code may also be scrambled without
destroying their mutual orthogonality provided that the same
scrambling order is used for each code. In this way, a variety of
orthogonal code sets may be produced for allocating to different
groups of users in adjacent service areas.
[0034] When an inverse Fourier Transform (IFT) is performed on one
of these 11-element spectral vectors, the result is a time domain
waveform that can be used for data transmission as described below.
The other spectral vectors, when similarly transformed, result in
other time domain waveforms that can be used to spread and transmit
other data at the same time over the same channel without mutual
interference. A receiver for such a non-contiguous spread spectrum
signal captures the received waveform in the time domain and
performs a Fourier Transform (FT) on it to reconstruct the spectral
vector. The reconstructed spectral vector is then correlated with a
pre-stored version of the actual spectral vector to determine the
transmitted data.
[0035] The 11-element example spectral vectors set forth above can
be "padded out" using leading and trailing zeros to obtain a
composite number of elements such as a power of two, so that an
inverse Fast Fourier Transform (IFFT) can be employed. For example,
the above 11-element spectral vector could be padded out to 16
elements. The resulting 16 time samples (after IFFTing the
16-element spectral vector) are then transmitted at a proper rate
to obtain the desired spectrum. The proper rate is that sample rate
that results in the spectral energy falling in the designated
spectral regions and that completes transmission of the block of
samples in a time "T" equal to the reciprocal of the spacing of
adjacent spectral lines (.DELTA.f), i.e., T=1/.DELTA.f (see FIG. 5
for an illustration). The receiver may sample the received waveform
at the same rate in order to decode the data. Data modulation in
the transmitter may be accomplished by multiplying the
non-contiguous spectrum spreading waveform obtained from the IFT or
IFFT by a complex data symbol, non-limiting examples of which
include a QPSK symbol or a 16-QAM symbol. The symbol value is
constant over the time period T containing the block of samples
corresponding to the non-contiguous spectrum spreading waveform.
Either or both of the transmitter and the receiver may use
oversampling to facilitate transition between discrete-time samples
and continuous waveforms. Namely, finer time-sampling may be used
so that less complicated and/or expensive smoothing or
band-limiting filters are needed.
[0036] FIG. 1 is a flowchart diagram illustrating non-limiting
example procedures for generating and combining a non-contiguous
spectrum spreading waveform with data symbols for radio
transmission. A non-contiguous spectrum is generated having
non-zero components in permitted regions of the electromagnetic
spectrum and zero components in non-permitted regions of the
electromagnetic spectrum (step S1). The non-contiguous spectrum is
transformed to produce a non-contiguous spectrum spreading signal
(step S2). One example transform is the Fourier transform. Other
transformations could be used, e.g., a FFT. The information is
processed into a stream of data symbols. The non-contiguous
spectrum spreading signal and a data symbol from the stream are
combined to produce a data modulated signal (step S3). The data
modulated signal is then modulated onto a carrier signal having a
desired center frequency (step S4) to produce a signal for
transmission and transmitted. Data modulated signals produced by
successive data symbols from the stream may be overlapped to
produce a transmission signal.
[0037] In one non-limiting example implementation, producing a
non-contiguous spectrum spreading signal includes performing an
inverse Discrete Fourier Transformation on the non-contiguous
spectrum to produce a sequence of time waveform samples. In this
case, combining the non-contiguous spectrum spreading signal and a
data symbol from the stream to produce the data modulated signal
includes multiplying the data symbol value with each sample of the
sequence of time waveform samples. Samples of the resulting shaped
non-contiguous spectrum spreading signal may be stored in
memory.
[0038] In another non-limiting example implementation, producing a
non-contiguous spectrum spreading signal includes allocating a
complex number to represent an amplitude and phase of each non-zero
component of the non-contiguous spectrum and a zero value to
represent the zero spectral components. Then, combining the
non-contiguous spectrum spreading signal and a data symbol from the
stream to produce a data modulated signal includes multiplying the
data symbol value with each of complex numbers to generate a set of
products. An inverse Discrete Fourier Transform is performed on the
set of products to produce the data modulated signal.
[0039] At the receiver, a reverse set of operations is performed.
FIG. 2 is a flowchart diagram illustrating non-limiting example
procedures for receiving and despreading a signal that occupies
predetermined regions of the electromagnetic spectrum. A signal
comprising a signal waveform in the time domain containing a
non-contiguous spectrum signal is received (step S10), and
downconverted to baseband (step S11). One or more components of the
time domain signal waveform are transformed to produce contiguous
spectral components (step S12). Contiguous spectral components
corresponding to the predetermined non-contiguous regions of the
spectrum are selected to obtain non-contiguous spectral components
(step S13). The non-contiguous spectral components are processed to
produce one or more received symbol values (step S14). The received
symbol values are then decoded to reproduce the information (step
S15).
[0040] The non-contiguous spectrum spreading waveform may be
modulated with successive complex data symbols to generate an
"unshaped" data transmission. But an unshaped transmission leaks
energy into unavailable areas of the spectrum where no energy was
intended, not licensed, etc. The leaking into the spectral
locations having zero values in the above spectral vectors is due
to the discontinuity between successive modulating symbols
imparting a sin(x)/x (sinc) shaped spread to each spectral point,
as depicted by FIGS. 3A and 3B. As is well known, the Fourier
transform assumes that the waveform is repetitive. In FIG. 3A, the
same block of time samples of the waveform, where the length of the
block corresponds to the frequency bandwidth B of the
non-contiguous spectrum (see an illustration in FIG. 5), is
repeated an infinite number of times. The corresponding spectrum in
this case are the thin spectral lines shown at the bottom of FIG.
3A. In contrast, FIG. 3B shows that if the repeated blocks of time
samples are truncated, the corresponding spectrum broadens
resulting in leakage into adjacent frequencies. The spectrum of
each broadened spectral line will be the same as that of the
shaping function. Fortunately, this leakage problem can be
significantly reduced using a "shaped" data transmission, analogous
to pulse-shaped OFDM, as described later.
[0041] For an example of unshaped data transmission, consider the
use of QPSK data symbols. Each QPSK data symbol can take on any of
the four values 1, j, -1, or -j representing respectively the
bit-pairs 00, 01, 11, and 10. Using the first of the above example
spectral codes from Table 1, the signal which would be transmitted
for the 4-symbol data sequence 1, 1, j, -1 would be formed as the
IFT of the following data-modulated spectral vectors: (+1 +1 +1 0 0
+1 +1 0 +1 +1 +1) for symbol "1"; (+1 +1 +1 0 0 +1 +1 0 +1 +1 +1)
for symbol "1"; (+j +j +j 0 0 +j +j 0 +j +j +j) for symbol "j"; and
(-1 -1 -1 0 0 -1 -1 0 -1 -1 -1) for symbol "-1". The data
modulation then corresponds to multiplying each symbol value by a
spectral vector (+1 +1 +1 0 0 +1 +1 0 +1 +1 +1). Each
data-modulated spectral vector is then inverse Fourier transformed
(IFT) into IFT values. The IFT values are concatenated to form a
time domain signal for transmission via an RF carrier.
Alternatively, the same time domain signal to be transmitted can be
formed by first performing an IFT on the unmodulated spectral
vector, which has the benefit of only requiring a single IFT type
operation the results of which can be stored and reused, and then
multiplying each data symbol by each successive element of the
transformed waveform vector.
[0042] As mentioned above, the Fourier transform assumes the
waveform transformed is repetitive/repeating. Consider as one
example, a 16-symbol block repeated three times so that there is a
cyclic prefix block of samples, a central block of samples, and a
cyclic postfix block of samples. There can be more cyclic prefix
blocks and postfix blocks. The cyclic prefix and postfix blocks are
preferably successively reduced in amplitude moving away from the
central block in order to smooth out truncation effects. The entire
train of blocks of one symbol may overlap the train of blocks used
for a neighboring symbol. A weighting function for the cyclic
prefix and postfix blocks is selected to have the Nyquist property
such that interference between successive symbols may be
avoided.
[0043] Subsequently, at the receiver, the prefix and postfix blocks
are weighted and added to the main block of received samples using
the same weighting function. Thus, the combined sample blocks will
have been weighted with the square of the shaping function. An
illustration of this is shown in FIG. 4 for five successive blocks.
The top waveform corresponds to the shaping function. The middle
waveform corresponds to the matched filter output at the receiver
in which the same shaping waveform is applied effectively squaring
the shaping waveform. The shaping function has the Nyquist property
in the frequency domain if, upon superimposing repeated blocks each
weighted with a section of the square of the shaping function, the
net weight applied to every sample is equal. The transition from
the middle graph to the bottom graph in FIG. 4 shows an example of
superpositioning. The result is a pulse (shown in bold) with a net
"flat" top.
[0044] The shaping function has the Nyquist property in the time
domain if the product of the shaping function with itself delayed
by one or more whole blocks has an integral (or mean value) of
zero. The latter property avoids a data symbol being corrupted by
intersymbol interference (ISI) from adjacent symbols even though
overlapped by the diminishing tails of their shaped prefix and
postfix blocks. Approximations to Nyquist shaping may also be used,
such as shaping with a Gaussian function, as a certain amount of
ISI between data symbols can often be tolerated and removed by
processing despread symbol values with an equalizer such as a
Viterbi Maximum Likelihood sequence estimator.
[0045] FIGS. 5A-5E show a non-limiting example construction of a
non-contiguous spectrum spreading waveform. FIG. 5A shows a
non-contiguous spectrum A with spectral lines at some frequencies
and no spectral lines at other frequencies in between. The
bandwidth of the non-contiguous spectrum A is designated as B. The
frequency spacing between adjacent spectral lines is designated
.DELTA.f. A time domain waveform B corresponding to the
non-contiguous frequency spectrum shown in FIG. 5A is shown in FIG.
5B. That time domain waveform may be obtained by performing a
Fourier Transform of the non-contiguous frequency spectrum shown in
FIG. 5A. The waveform in FIG. 5B is shown as a continuous curve,
but in practice it is more likely to consist of discrete points
depending on how many points are calculated by the Fourier
Transform. The spacing between the samples corresponds to 1/B. The
waveform "block" in the time domain corresponding to the
non-contiguous spectrum A in the frequency domain is shown as
lasting for a time period T, where T is the duration in which the
block of samples must be transmitted.
[0046] As explained above, the waveform B in FIG. 5B will only have
the exact frequency spectrum of the non-contiguous spectrum A in
FIG. 5A if the waveform B is repeated an infinite number of times,
as illustrated in FIG. 5C entitled "Spreading Waveform C," which
repeats forever as indicated by the dotted lines extending to the
right and left. But if waveform B is repeated only a finite number
of times, and those waveform B copies to the left and right of the
central waveform copy gradually taper off, the result is a
broadening of the spectral lines from a zero width, as depicted in
FIGS. 3A and 5A, to a shape which is the Fourier transform of the
shaping function, as depicted in FIG. 3B. An example of a
shaping/tapering function D is shown in FIG. 5D.
[0047] A finite tapering function may be used if the truncation is
chosen to meet a desired constraint on approximating the spectrum
in FIG. 5A. In this case, the desired constraint is that the
spectral energy in the unavailable regions of the spectrum is below
a desired maximum level. The desired maximum level determines the
extent of the tapering function. The shape of the tapering function
waveform is preferably chosen to have the Nyquist property, as
described above, such that when the receiver weights and adds
repeat blocks using the same shaping function, the sum will be
substantially zero for data symbols separated by one block duration
from the current symbol. Meeting this Nyquist property minimizes
intersymbol interference between data symbols.
[0048] The shaped spreading waveform E in FIG. 5E illustrates the
product of the waveforms C and D and has the same finite extent in
time as the shaping function D shown in FIG. 5D. Because it is
finite and does not at this stage depend on variable data symbols,
the shaped, non-contiguous spectrum spreading waveform E can, if
desired, be pre-computed and stored in memory. For example, a
non-contiguous spectrum spreading waveform for each spectral vector
may be pre-computed and stored.
[0049] Different data stream transmissions may be distinguished
using different non-contiguous spectrum spreading waveforms. The
different data streams may belong to different users, or may all
belong to the same user as a way of increasing data throughput
rate. The latter technique is known as "Multicode" operation.
[0050] The function block diagram in FIG. 6 illustrates how the
shaped, non-contiguous spectrum spreading waveform can be combined
with data symbols for transmission. The interconnections between
the functional units show signal flow paths, but it is understood
that other interconnections, such as control, power, clocks, and
timing signals may be present in a practical design. A data
processor 110 generates a sequence of addresses to read from a
non-contiguous spectrum spreading waveform memory 100 digital
values representing points on the shaped, non-contiguous spreading
waveform, such as waveform E shown in FIG. 5E. Of course, other
shaped, non-contiguous spreading waveforms may be used at different
times or simultaneously for multicode operation. The shaped,
non-contiguous spreading waveform may be complex, having a real
part and an imaginary part, and the values read from memory 100 are
the also complex values, having a real part and an imaginary part.
To simplify the description and illustration, however, only one
part is shown in FIGS. 5A-5E.
[0051] The non-contiguous spectrum spreading waveform values read
from the memory 100 are combined with successive data symbol values
in a multiplier 120 to generate a data modulated signal. The
multiplier 120 may be a real multiplier if both data and waveform
values are real, a half-complex multiplier if the waveform values
are complex and the data values are real as in binary bits, vice
versa if the waveform values are real and the data values are
complex as in 8-PSK or 16QAM values, or a full complex multiplier
if both the waveform values and the data values are complex.
[0052] New data values are combined with the shaped, non-contiguous
spreading waveform E shown in FIG. 5E every block period T, even
though the waveform duration is longer than T, for example 3T as
shown in FIGS. 5A-5E. As a result, the shaped tail of the postfix
of the previous data symbol waveform overlaps the center block of
the current data symbol waveform, and the shaped prefix tail of the
current data symbol waveform overlaps the center block of the
previous symbol. Likewise, the prefix and postfix tails of the
current and subsequent symbol overlap. If the combined length of
the prefix tail, center block, and postfix tail is for example
three block durations, i.e., 3T as is the case in FIGS. 5A-5E, then
each transmitted signal sample will be the sum of contributions
from the three overlapping data symbol waveforms. This summation
may be performed by accumulator 130. If a waveform block of
duration T is represented by N samples indexed by i=0 to N-1, and
the total length of the shaped, non-contiguous spectrum spreading
waveform shown in FIG. 5E is 3N samples corresponding to the
composite transmit waveform having an index that can vary from -N
to 2N-1, then the function performed (under the control of
processor 110) by the multiplier 120 and the accumulator 130 to
generate a k'th block of N output samples during the period of data
symbol Data(k) may be mathematically described by:
Output(k,i)=Data(k-1)*WaveformSample(i+N)+Data(k)*WaveformSample(i)+Data-
(k+1)*WaveformSample(i-N) (1)
By extension, if the non-contiguous spreading waveform shown in
FIG. 5E has a duration of 5T represented by 5N samples, then
equation (1) would have five terms instead of only three.
[0053] As mentioned previously, all the quantities in equation (1)
can be complex, so the accumulator output values can comprise real
and imaginary parts. The output values are fed to a
digital-to-analog (D-to-A) converter 140, which can comprise two
D-to-A converters to convert, respectively, the real accumulator
output value to a real analog signal I and the imaginary
accumulator output value to an imaginary or Quadrature analog
signal Q. The complex I, Q signals then drive a quadrature
modulator 150 which modulates a radio center frequency or carrier
signal with the I, Q waveforms to produce the radio frequency (RF)
signal for transmission.
[0054] When constructed according to the above, the transmitted
radio signal occupies non-contiguous areas of RF spectrum in a
particular RF band, but does not occupy other areas of RF spectrum
within that particular RF band that lie in between two or more of
the non-contiguous areas and which are unavailable to that
particular transmission. The unavailable spectrum may be allocated
to other services or signals. The suppression of energy in the
unavailable spectrum is improved the longer the pulse shaping
function shown in FIG. 5D. The non-contiguous spectrum may be
calculated using simulation techniques in order to optimize the
design. There are many suitable shaping functions, such as the
known IOTA pulse disclosed in U.S. Pat. No. 6,278,686, that have
the desirable Nyquist property providing good spectral
containment.
[0055] Radio signals constructed starting with a spectral vector
like that shown in FIG. 5A having different orthogonal sign
patterns, such as ++++++++++ and +-+-+++-, will be orthogonal to
each other, which means that their cross-correlations with each
other and with all the other spectral vectors in the set are zero
or are least minimal. In practical terms, orthogonality means that
a signal using one vector to generate its spreading waveform will
be substantially non-interfering with a signal using a different
vector. So using different code vectors allows different users to
communicate using the same spectrum without mutual interference.
Thus, the system described here has the characteristics of an
Orthogonal Code Division Multiple Access (OCDMA) system. Such
orthogonality is preferred when it can be achieved, but
non-orthogonal CDMA can also be used, in which case random vectors
could be selected.
[0056] Orthogonality requires that orthogonal signals be
time-aligned to an accuracy of some fraction of the reciprocal of
the total bandwidth B of the spectrum. This is readily achievable
for transmissions from the same base station, but transmissions
from different mobile terminals cannot be so readily synchronized.
Therefore, orthogonality may be used on the downlink
(base-to-mobile) but relinquished on the uplink (mobile-to-base).
Other interference reduction techniques may be used in the base
station to mitigate interference between different users on the
uplink, such as multi-user detection, successive detection with
interference subtraction, etc.
[0057] In the case where the number of non-zero spectral lines is a
power of two, a set of orthogonal sign patterns is given by the
Walsh functions, which are binary sequences of +1 or -1. In the
case where the number of non-zero spectral lines is not a power of
two, orthogonal sequences may be given by the Fourier sequences,
where w stands for the principal Nth root of unity, e.sup.j2.pi./N
[0058] 1 1 1 1 1 1 1 [0059] 1 w w.sup.2 w.sup.3 w.sup.4 w.sup.5
w.sup.6 [0060] 1 w.sup.2 w.sup.4 w.sup.6 w w.sup.3 w.sup.5 [0061] 1
w.sup.3 w.sup.6 w.sup.2 w.sup.5 w [0062] etc Because Fourier
sequences are complex, the resulting non-contiguous spreading
waveforms are also complex.
[0063] As in conventional, contiguous spectrum CDMA systems, the
orthogonality of different signals in a non-contiguous spectrum
CDMA system is degraded by multipath propagation. The manner in
which orthogonality is degraded by multipath for a conventional
contiguous spectrum CDMA system and a non-contiguous spectrum CDMA
system is described below.
[0064] In conventional CDMA using orthogonal codes, two
time-aligned signals using different codes are orthogonal and there
cross-correlation is theoretically zero. But due to multi-path
propagation, delayed reflections of one signal may be received that
are not time-aligned with the undelayed ray of another signal. The
codes that are not time-aligned are not orthogonal, but have a
non-zero cross correlation, which on average is 1/root(N), where N
is the code length in chips. Thus, non-orthogonality in
conventional CDMA is explained by the time-domain characteristics
of the propagation path. It is possible to restore orthogonality in
conventional CDMA by passing the signal through an inverse shaping
filter, which effectively undoes the multi-path propagation.
Unfortunately an inverse filter can be mathematically
ill-conditioned, resulting in magnification of noise and
interference from other transmitters, e.g., other cellular base
stations. Commonly-owned U.S. Pat. No. 5,572,552 to Dent et al,
incorporated here by reference, describes that a hybrid between an
inverse channel filter and a matched filter can provide optimum
performance when there is both same-cell and other-cell
interference.
[0065] Two generated non-contiguous spectrum spreading waveforms
are derived from corresponding spectral lines that are rotated or
sign-changed according to orthogonal sequences produce orthogonal
radio signals. Those signals remain orthogonal if the relative
amplitudes and phases of the spectral lines remain unchanged
through the propagation path. In the downlink from a cellular base
station transmitting multiple signals to a radio terminal,
corresponding spectral lines of each signal suffer the same phase
rotation, so this does not affect orthogonality provided that phase
rotation is compensated in a channel estimation used when
despreading a signal. In the uplink from multiple mobile terminals
to a base station, however, corresponding spectral lines do not
necessarily suffer the same phase shift, so it is much harder to
totally restore orthogonality by compensation. Accordingly, many
cellular systems are content to use non-orthogonal codes on the
uplink.
[0066] Returning to the downlink, multi-path propagation across the
occupied spectrum disrupts the constant, equal amplitudes of the
individual spectral components: As a result, when a composite
signal comprising the sum of many non-contiguous spectrum spreading
waveforms is received and correlated with one of the non-contiguous
spectrum spreading waveforms, the other non-contiguous spectrum
spreading waveforms will produce a non-zero correlation output.
Orthogonality may be restored by first flattening the received
spectrum so that all spectral lines of each signal are of equal
amplitude. The flattening process is equivalent to the inverse
channel filter that can be used in conventional CDMA to restore
downlink orthogonality. As in the case of conventional CDMA, a
compromise hybrid between a matched filter and an inverse channel
filter that does not quite flatten the spectrum is likely to
provide optimal performance when interference is partly same-cell
and partly other-cell interference and noise.
[0067] FIG. 7 is a function block diagram of a non-limiting example
apparatus for implementing a correlation method at a radio
receiver. The receiver correlation process is the inverse of the
waveform construction process, an example of which was shown in
FIGS. 5A-5E. That non-limiting example is continued here on the
receive side. Assuming the length of the shaping function shown in
FIG. 5D is 3T, the received composite waveform contains information
on a current symbol in 3 segments, labeled E1, E2 and E3 at the
bottom of FIG. 7. These three segments are copies of the time
waveform from FIG. 5B weighted by the corresponding three shaping
function segments D1, D2, and D3 of the shaping function D shown in
FIG. 5D.
[0068] As already explained, the prefix tail of the following
symbol and the postfix tail of the previous symbol overlap the
central signal block of the current symbol due to the extended
shaping function needed for spectral control. Accordingly, the
received composite signal is filtered by a matched filter which
weights the received signal segments E1, E2, and E3 with the same
shaping function as used at the transmitter, namely shaping
segments D1 (the prefix tail), D2 (the central block), and D3 (the
postfix tail). The weighted segments are summed together to produce
a received time domain waveform (similar to that shown in FIG. 5B)
equal to D1*E1+D2*E2+D3*E3.
[0069] When the shaping function has the Nyquist property, the
resulting weighted and combined signal sample block will have a
flat weighting across the block (illustrated by the flat top of the
pulse waveform shown in FIG. 4). This waveform relationship occurs
when D2(i-N)+D.sup.2(i)+D.sup.2(i+N))=1, where i is the index of a
sample of the shaping function D, which is a desirable constraint
on the design of the shaping function. Thus, when the reconstructed
received waveform B shown in FIG. 7 is Fourier transformed at block
210, the noncontiguous spectrum A shown in FIG. 5A for the
transmitter will be reproduced, albeit modified by the addition of
noise and interference and the frequency response shaping effect of
multi-path propagation. The spectral lines output from the Fourier
Transform function 210 are then combined in a weighted summer 200
to produce a received symbol output.
[0070] The weighting factors used for the spectral lines can be the
complex conjugate of the spectral line weights used in waveform
construction at the transmitter shown in FIG. 6. The weighting
factors can also correct for phase changes of each spectral line
due to multi-path propagation which are derived by forming channel
estimates. A channel estimate may be obtained using one or more
known pilot sequences modulated on to one of the orthogonal codes.
An example of this channel estimation technique is well known for
example in the IS-95 CDMA system. The weighting functions may
alternatively be calculated from an estimated channel and
estimations of same-cell and other-cell interference in such a way
as to provide optimum detection of symbols as described in U.S.
Pat. No. 5,572,552.
[0071] In the case of multicode operation, the receiver may
despread using multiple despreading waveforms to obtain multiple
despread complex values per data symbol period. These multiple
values can be regarded as a vector of despread values, and from the
sequence of despread vectors, a corresponding sequence of vectors
of data symbols is to be decoded. U.S. Pat. No. 6,963,532,
incorporated by reference, describes Block Decision Feedback
Equalizers (Block DFEs) for equalizing interference between
successive data symbol vectors, termed InterVector Interference
(IVI) therein.
[0072] None of the above description should be read as implying
that any particular element, step, range, or function is essential
such that it must be included in the claims scope. The scope of
patented subject matter is defined only by the claims. The extent
of legal protection is defined by the words recited in the allowed
claims and their equivalents. All structural and functional
equivalents to the elements of the above-described preferred
embodiment that are known to those of ordinary skill in the art are
expressly incorporated herein by reference and are intended to be
encompassed by the present claims. Moreover, it is not necessary
for a device or method to address each and every problem sought to
be solved by the present invention, for it to be encompassed by the
present claims. No claim is intended to invoke paragraph 6 of 35
USC .sctn.112 unless the words "means for" or "step for" are used.
Furthermore, no embodiment, feature, component, or step in this
specification is intended to be dedicated to the public regardless
of whether the embodiment, feature, component, or step is recited
in the claims.
* * * * *