U.S. patent application number 12/896783 was filed with the patent office on 2011-01-27 for method and apparatus for systematic and random variation and mismatch compensation for multilevel flash memory operation.
This patent application is currently assigned to Silicon Storage Technology, Inc.. Invention is credited to Hung Quoc Nguyen, Hieu Van Tran.
Application Number | 20110018634 12/896783 |
Document ID | / |
Family ID | 37910958 |
Filed Date | 2011-01-27 |
United States Patent
Application |
20110018634 |
Kind Code |
A1 |
Tran; Hieu Van ; et
al. |
January 27, 2011 |
Method and Apparatus for Systematic and Random Variation and
Mismatch Compensation for Multilevel Flash Memory Operation
Abstract
Method and means for random or systematic mismatch compensation
for a memory sensing system are disclosed. A sense amplifier
includes a bulk voltage source to set the bulk of the sensing
transistor to be a voltage different than the voltage driving the
sensing transistor. For an NMOS sensing transistor, a triple well
is used with the variable bulk voltage. Differential sense
amplifiers with various offset compensation are included.
Intentional offset creation for useful purpose is also
included.
Inventors: |
Tran; Hieu Van; (San Jose,
CA) ; Nguyen; Hung Quoc; (Fremont, CA) |
Correspondence
Address: |
DLA PIPER LLP (US )
2000 UNIVERSITY AVENUE
EAST PALO ALTO
CA
94303-2248
US
|
Assignee: |
Silicon Storage Technology,
Inc.
Sunnyvale
CA
|
Family ID: |
37910958 |
Appl. No.: |
12/896783 |
Filed: |
October 1, 2010 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
12131008 |
May 30, 2008 |
7825698 |
|
|
12896783 |
|
|
|
|
11235894 |
Sep 26, 2005 |
7405988 |
|
|
12131008 |
|
|
|
|
Current U.S.
Class: |
330/253 |
Current CPC
Class: |
G11C 7/02 20130101; G11C
7/08 20130101; G11C 11/5642 20130101; G11C 16/28 20130101; G11C
16/30 20130101; G11C 7/062 20130101 |
Class at
Publication: |
330/253 |
International
Class: |
H03F 3/45 20060101
H03F003/45 |
Claims
1. A differential amplifier comprising: a differential sensing
amplifier having an NMOS transistor pair input with well voltage
compensation.
2. The differential amplifier of claim 1 further comprising a
current bias trimming circuit to provide differential loading to
said differential sensing amplifier.
3. (canceled)
4. A differential amplifier comprising: an output transconductance
amplifier including MOS transistors, said output transconductance
amplifier comprising: a first PMOS transistor including first and
second terminals spaced apart with a channel therebetween and
including a gate for controlling current in said channel, said
first terminal being coupled to a first voltage terminal, said gate
being coupled to receive a bias voltage; a second PMOS transistor
including first and second terminals spaced apart with a channel
therebetween, including a gate for controlling current in said
channel, and including a bulk voltage terminal, said gate being
coupled to a first input terminal, said first terminal being
coupled to the second terminal of the first PMOS transistor, said
bulk voltage terminal being coupled to a second voltage terminal to
receive a voltage different than the voltage on the first voltage
terminal; a first NMOS transistor including first and second
terminals spaced apart with a channel therebetween, and including a
gate for controlling current in said channel, said second terminal
being coupled to a ground terminal, said first terminal being
coupled to said second terminal of said second PMOS transistor; a
third PMOS transistor including first and second terminals spaced
apart with a channel therebetween, including a gate for controlling
current in said channel, and including a bulk voltage terminal,
said gate being coupled to a second input terminal, said first
terminal being coupled to the second terminal of the first PMOS
transistor, said second terminal being coupled to a first output
terminal and to the gate of the first NMOS transistor, said bulk
voltage terminal being coupled to the second voltage terminal; a
second NMOS transistor including first and second terminals spaced
apart with a channel therebetween, and including a gate for
controlling current in said channel, said second terminal being
coupled to the ground terminal, said gate being coupled to the
second terminal of the second PMOS transistor, said first terminal
of the second NMOS transistor being coupled to the first output
terminal; a third NMOS transistor including first and second
terminals spaced apart with a channel therebetween, and including a
gate for controlling current in said channel, said second terminal
being coupled to the ground terminal, said gate being coupled to
receive a bias voltage, said first terminal being coupled to the
first terminal of the first NMOS transistor; and a fourth NMOS
transistor including first and second terminals spaced apart with a
channel therebetween, and including a gate for controlling current
in said channel, said second terminal being coupled to the ground
terminal, said gate being coupled to receive a bias voltage, said
first terminal being coupled to the first terminal of the second
NMOS transistor; and a bulk voltage generator coupled to at least
one of said MOS transistors to generate a bulk voltage different
than a supply voltage of the output transconductance amplifier.
5. (canceled)
6. (canceled)
7. (canceled)
8. (canceled)
9. (canceled)
10. (canceled)
11. (canceled)
12. (canceled)
13. (canceled)
14. (canceled)
15. (canceled)
16. The differential amplifier of claim 4 wherein the output
transconductance amplifier further comprises an autozero circuit
coupled between the gate and first terminal of the second NMOS
transistor.
17. The differential amplifier of claim 16 wherein the autozero
circuit further couples the gate and first terminal of the first
NMOS transistor.
Description
RELATED APPLICATIONS
[0001] This application is a divisional of U.S. patent application
Ser. No. 12/131,008 filed May 30, 2008, which is a divisional of
11/235,894 filed Sep. 26, 2005, both of which are incorporated
herein by this reference.
FIELD OF THE INVENTION
[0002] The present invention relates to sense amplifiers and, more
particularly, relates to sense amplifiers that compensate for
variations and mismatches in a memory circuit.
BACKGROUND OF THE INVENTION
[0003] As information technology progresses at an unprecedented
pace, the need for information storage increases proportionately.
Accordingly, the non volatile information in stationary or portable
communication demands higher capability and capacity storage. One
approach to increasing the amount of storage is by decreasing
physical dimensions of the stored bit (e.g., memory cell) to
smaller dimensions such as nanocell technology. Another approach is
to increase the storage density per bit. The second approach is
known as digital multilevel nonvolatile storage technology. A sense
amplifier reads the content of a memory cell by comparison to
reference levels. As more bits are stored in a multilevel memory
cell, the voltage separation of reference levels decreases.
Systematic and random variation and mismatch in a sense amplifier
may change data or reference levels to cause erroneous detection of
the content of a memory cell.
SUMMARY OF THE INVENTION
[0004] The present invention provides a sense amplifier that may
include well voltage compensation of transistors therein. It also
includes other compensation methods and means.
BRIEF DESCRIPTION OF THE DRAWINGS
[0005] FIG. 1 is a block diagram illustrating a digital multilevel
bit memory system.
[0006] FIG. 2 is a schematic diagram illustrating a conventional
sensing system.
[0007] FIG. 3 is a schematic diagram illustrating a first
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0008] FIG. 4 is a schematic diagram illustrating a second
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0009] FIG. 5 is a schematic diagram illustrating a third
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0010] FIG. 6 is a schematic diagram illustrating a first
embodiment of a bulk voltage generator of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0011] FIG. 7 is a schematic diagram illustrating a second
embodiment of a bulk voltage generator of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0012] FIG. 8 is a schematic diagram illustrating a fourth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0013] FIG. 9 is a schematic diagram illustrating a third
embodiment of a bulk voltage generator of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0014] FIG. 10 is a schematic diagram illustrating a fifth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0015] FIG. 11 is a schematic diagram illustrating a sixth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0016] FIG. 12 is a schematic diagram illustrating a seventh
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0017] FIG. 13 is a schematic diagram illustrating an eighth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0018] FIG. 14 is a schematic diagram illustrating a ninth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0019] FIG. 15 is a schematic diagram illustrating a tenth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0020] FIG. 16 is a schematic diagram illustrating an eleventh
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0021] FIG. 17 is a schematic diagram illustrating a twelfth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0022] FIG. 18 is a schematic diagram illustrating a fourth
embodiment of a bulk voltage generator of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0023] FIG. 19 is a schematic diagram illustrating a thirteenth
embodiment of a sensing system of the digital multilevel bit memory
system of FIG. 1.
[0024] FIG. 20 is a schematic diagram illustrating a first
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0025] FIG. 21 is a schematic diagram illustrating a fifth
embodiment of a bulk voltage generator of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0026] FIG. 22 is a schematic diagram illustrating a second
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0027] FIG. 23 is a schematic diagram illustrating a third
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0028] FIG. 24 is a schematic diagram illustrating a fourth
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0029] FIG. 25 is a schematic diagram illustrating a fifth
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0030] FIG. 26 is a schematic diagram illustrating a sixth
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0031] FIG. 27 is a schematic diagram illustrating a seventh
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0032] FIG. 28 is a schematic diagram illustrating an eighth
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0033] FIG. 29 is a schematic diagram illustrating a ninth
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0034] FIG. 30 is a schematic diagram illustrating a sixth
embodiment of a bulk voltage generator of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0035] FIG. 31 is a schematic diagram illustrating a tenth
embodiment of a differential amplifier of a sensing system of the
digital multilevel bit memory system of FIG. 1.
[0036] FIG. 32 is a schematic diagram illustrating a first
embodiment of a memory cell sensing system of the digital
multilevel bit memory system of FIG. 1.
[0037] FIG. 33 is a schematic diagram illustrating a second
embodiment of a memory cell sensing system of the digital
multilevel bit memory system of FIG. 1.
[0038] FIG. 34 is a schematic diagram illustrating a third
embodiment of a memory cell sensing system of the digital
multilevel bit memory system of FIG. 1.
[0039] FIG. 35 is a diagram illustrating voltages for memory levels
of the digital multilevel bit memory system of FIG. 1.
[0040] FIG. 36 is a block diagram illustrating a digital autozero
control system of the digital multilevel bit memory system of FIG.
1.
[0041] FIG. 37 is a flow chart illustrating the operation of the
control system of FIG. 36.
[0042] FIG. 38 is a block diagram illustrating an analog autozero
control system of the digital multilevel bit memory system of FIG.
1.
DETAILED DESCRIPTION
[0043] A memory system that compensates for systematic and random
variation and mismatch in a memory, such as flash memory, is
described. The compensation may minimize output variation between
output levels for reference and data cells for various cell levels
across a memory array. The compensation may include threshold
voltage modulation, data trimming, or voltage shifting, or
combinations thereof. Ratio tuning or margining may be achieved
using similar compensation. Further, the programming and erase
levels may be similarly compensated. The memory system may include
a differential amplifier with modulation of well voltage of an
input pair or well voltage of an output, and may include well
voltage tracking with common mode input voltage. An offset may be
created, such as offset addition or subtraction, for margining,
level speed up, supply voltage VDD and temperature compensation,
decoding compensation, or systematic compensation.
[0044] FIG. 1 is a block diagram illustrating a digital multilevel
bit memory array system 100.
[0045] The digital multilevel bit memory array system 100 includes
a memory array 101 that includes a plurality of memory cells (not
shown) and a reference array 106 that includes a plurality of
reference memory cells (not shown). An N bit digital multilevel
cell is defined as a memory cell capable of storing the 2.sup.N
levels. The reference array 106 is used as a reference system of
reference voltage levels to verify the contents of the memory array
101. In another embodiment, the memory array 101 may include
reference memory cells for storing the reference voltage
levels.
[0046] In one embodiment, the memory array 101 and the reference
array 106 include a source side injection flash technology, which
uses lower power in hot electron programming, and efficient
injector based Fowler-Nordheim tunneling erasure. The programming
may be done by applying a high voltage on the source of the memory
cell, a bias voltage on the control gate of the memory cell, and a
bias current on the drain of the memory cell. The programming in
effect places electrons on the floating gate of memory cell. The
erase is done by applying a high voltage on the control gate of the
memory cell and a low voltage on the source and/or drain of the
memory cell. The erase in effect removes electrons from the
floating gate of memory cell. The verify (sensing or reading) is
done by placing the memory cell in a voltage mode sensing, e.g., a
bias voltage on the source, a bias voltage on the gate, a bias
current coupled from the drain (bitline) to a low bias voltage such
as ground, and the voltage on the drain is the readout cell voltage
VCELL. The bias current may be independent of the data stored in
the memory cell. In another embodiment, the verify (sensing or
reading) is done by placing the memory cell in a current mode
sensing, e.g., a low voltage on the source, a bias voltage on the
gate, a load (resistor or transistor) coupled to the drain
(bitline) from a high voltage supply, and the voltage on the load
is the readout voltage. In one embodiment, the array architecture
and operating methods may be the ones disclosed in U.S. Pat. No.
6,282,145, entitled "Array Architecture and Operating Methods for
Digital Multilevel Nonvolatile Memory Integrated Circuit System" by
Tran et al., the subject matter of which is incorporated herein by
reference.
[0047] The multilevel memory cells of the memory array 101 may be
arranged in various ways, such as in rows and columns or in
segments. Various addressing schemes may be used which organize the
memory cells into bytes, pages or other arrangements.
[0048] The digital multilevel bit memory array system 100 further
includes an x decoder 120, a y decoder 110, an address controller
162, a sense amplifier circuit 111, and an intelligent input/output
interface 196. The y decoder 110 controls bitlines (not shown)
coupled to columns in memory cells and the reference voltage cells,
during a write, read (or verify), and erase operations. The sense
amplifier 111 senses the read data which is provided to the I/O
interface 196. The I/O interface 196 also buffers input into the
memory array system 100. The sense amplifier 111 also senses the
read data and verifies the read data against input data during
memory programming or erasing.
[0049] In response to an address signal 163 and other control
signals (not shown), the address controller 162 decodes the address
signal 163 and controls page, byte, segment or other addressing for
the x decoder 120 and the y decoder 110. The x decoder 120 selects
a row or a block of rows in the arrays 101 and 106 based on the
signals from the address controller 162 and provides precise
multilevel bias values over temperature, process, and power supply
used for consistent single level or multilevel memory operation for
the memory array 101. The system 100 includes power related
circuits (not shown), such as band gap voltage generators, charge
pumps, voltage regulators, and power management systems, and other
control circuits (not shown) such as voltage algorithm
controllers.
[0050] The system 100 may execute various operations on the memory
array 101. An erase operation may be done to erase all selected
multilevel cells by removing the charge on selected memory cells
according to the operating requirements of the non-volatile memory
technology used. A data load operation may be used to load in a
plurality of bytes of data to be programmed into the memory cells,
e.g., 0 to 512 bytes in a page. A read operation may be done to
read out in parallel a plurality of bytes of data if the data
(digital bits), e.g., 512 bytes within a page, stored in the
multilevel cells. A program operation may be done to store in
parallel a plurality of bytes of data in (digital bits) into the
multilevel cells by placing an appropriate charge on selected
multilevel cells depending on the operating requirements of the
non-volatile memory technology used. The operations on the memory
may be, for example, the operations described in U.S. Pat. No.
6,282,145, incorporated herein by reference above.
[0051] FIG. 2 is a schematic diagram illustrating a conventional
sensing system 200.
[0052] The conventional sensing system 200 comprises a reference
column 201, a plurality of data columns 202-0 through 202-N, and a
plurality of comparators 203-0 through 203-N. The reference column
201 comprises a reference memory cell 211, an NMOS transistor 212
and a PMOS transistor 215. A bitline resistor 213 is shown to
indicate resistance on the bitline. A bitline capacitor 214 is
shown to indicate capacitance on the bitline. The reference column
201 provides a voltage reference on the reference line 204 which is
applied to a first input of each of the comparators 203-0 through
203-N. Each data column 202 comprises a data memory cell 221, an
NMOS transistor 222 and a PMOS transistor 225. A bitline resistor
223 is shown to indicate resistance on the bitline. A bitline
capacitor 224 is shown to indicate capacitance on the bitline. Each
of the data columns 202-0 through 202-N provides a data output
voltage to a second input of a respective comparator 203-0 through
203-N so that the comparator 203 provides an output indicative of
the stored data in the corresponding data column 202.
[0053] The conventional sensing system 200 has mismatches within
the system because of differences in the PMOS transistors 215 and
225 that provide loads for the respective reference column 201 and
the data column 202. Further, the comparators 203 have a mismatch
in their inputs. These mismatches may lead to inaccurate reads of
the data cells 221. Moreover, the bitlines may have a mismatch in
capacitances that may lead to inaccurate reads, especially in
dynamic reads. Other mismatches may come from layout, such as
voltage drop along power lines or interconnect lines.
[0054] The mismatches may cause a difference dVo in voltage between
outputs of the comparators 203 due to the PMOS transistors mismatch
of the threshold voltage VT, beta mismatch, or voltage drop
mismatch, such as VDD, bias current Ibias, or voltage bias Vbias.
The difference voltage dVo is typically between 20 and 50
millivolts.
[0055] The sense amplifier 111 (FIG. 1) may include the sensing
systems of differential amplifiers, bulk voltage generators and
control systems of FIGS. 3-38.
[0056] FIG. 3 is a schematic diagram illustrating a sensing system
300.
[0057] The sensing system 300 uses selectable loading on bitlines
for sensing reference cells and data cells. The sensing system 300
comprises a reference column 301, a plurality of data columns
302-0-302-N, a plurality of comparators 303-0-303-N, and a
plurality of load circuits 305 and 306-0-306-N. The reference
column 301 comprises a reference memory cell 311, an NMOS
transistor 312 and a diode connected PMOS transistor 315. A bitline
resistor 313 is shown to indicate resistance on the bitline. A
bitline capacitor 314 is shown to indicate capacitance on the
bitline. The reference column 301 provides a voltage reference on a
reference line 304, which is applied to a first input of each of
the comparators 303-0-303-N. Each data column 302 comprises a data
memory cell 321, an NMOS transistor 322 and a diode connected PMOS
transistor 325. A bitline resistor 323 is shown to indicate
resistance on the bitline. A bitline capacitor 324 is shown to
indicate capacitance on the bitline. Each of the data columns
302-0-302-N provides a data output voltage to a second input of a
respective comparator 303-0-303-N so that the comparator 303
provides an output indicative of the stored data in the
corresponding data column 302 relative to the reference voltage
from the reference column 301. The load circuit 305 comprises a
plurality of diode connected PMOS transistors 331 and a plurality
of switches 332. The switches 332 selectively couple a
corresponding one of the diode connected PMOS transistors 331 to
the reference line 304 to further load the reference memory cell
311 during a sensing mode. Each of the load circuits 306 comprises
plurality of diode connected PMOS transistors 341 and a plurality
of switches 342 (only one is shown for clarity). The switch 342
selectively couples the diode connected PMOS transistor 341 to the
drain of the diode connected PMOS transistor 325 to load the data
memory cell 321 during sensing. Although one transistor 341 is
shown in the load circuit 306, other numbers of transistors may be
used for loading the data memory cell 321. Each load circuit 306
has its own individually selectable switches to compensate for its
own mismatches.
[0058] The load circuit 305 and 306 provide loads for offsetting
errors in the memory array, but the loading impacts the reading
speed and requires additional enabling lines. Further, the load
circuits may not provide perfect cancellation of mismatches, such
as for threshold voltage mismatch compensation.
[0059] FIG. 4 is a schematic diagram illustrating a sensing system
400.
[0060] The sensing system 400 uses auto zero to adjust offsets in
real time. (The sensing system 400 is shown in a voltage sensing
mode, but may be applied to a current sensing mode.) The sensing
system 400 comprises a reference column 401, a plurality of data
columns 402-0 through 402-N, and a plurality of comparators 403-0
through 403-N. The reference column 401 comprises a reference
memory cell 411, an NMOS transistor 412, and a biased NMOS
transistor 415 biased by a voltage VBIAS 430. A bitline resistor
413 is shown to indicate resistance on the bitline. A bitline
capacitor 414 is shown to indicate capacitance on the bitline. The
reference column 401 provides a voltage reference on a reference
line 404, which is applied to a first input of each of the
comparators 403-0 through 403-N. Each of the data columns 402-0
through 402-N provides a data output voltage to a second input of a
respective comparator 403-0 through 403-N so that the comparator
403 provides an output indicative of the stored data in the
corresponding data column 402 relative to the reference voltage
from the reference column 401. Each data column 402 comprises a
data memory cell 421, an NMOS transistor 422, and a biased NMOS
transistor 425. A bitline resistor 423 is shown to indicate
resistance on the bitline. A bitline capacitor 424 is shown to
indicate capacitance on the bitline. The NMOS transistor 425 is
biased by the bias voltage 430.
[0061] The comparators 403 include auto-zero. Although the sensing
system 400 provides auto zero offset in real time, the system may
use additional timing for offset settling, which may be in the
order of a few millivolts. Examples of auto zero circuits are
disclosed in co-pending published U.S. Patent Application No. US
2003/0103406 A1, published Jun. 5, 2003, the contents of which are
incorporated herein by reference.
[0062] FIG. 5 is a schematic diagram illustrating a sensing system
500.
[0063] The sensing system 500 comprises a reference column 501 and
a comparator 503. The reference column 501 comprises a reference
memory cell 511, a NMOS transistor 512 and a diode connected PMOS
transistor 515. The NMOS transistor 512 selectively couples the
reference memory cell 511 to a sense line 504, which is coupled to
the comparator 503. The NMOS transistor 512 may be a CMOS
transistor. A bitline resistor 513 is shown to indicate resistance
on the bitline. A bitline capacitor 514 is shown to indicate
capacitance on the bitline.
[0064] The source of the diode connected PMOS transistor 515 is
coupled to a supply voltage VSUPSA, which may be different than a
supply voltage applied to the memory system 100 (FIG. 1). The bulk
of the PMOS transistor 505 is biased by an adjustable voltage
applied to a bulk terminal 520, which may be a voltage that is
different or less than the sense amplifier supply voltage VSUPSA.
The bulk voltage may be provided by a voltage source such as
described in conjunction with FIGS. 6-7. In one embodiment, the
supply voltage minus the well voltage is less than the voltage of a
pn junction (VDD-VWELL<V-pn) to avoid forward biasing the
junction.
[0065] In illustrative embodiments of FIG. 5 and the following
FIGS. 8, 10 11, 15-17, and 19, described below, the bulk voltage
VBS may be used to change the threshold voltage VT, for example, at
a rate dVT/dVBS=0.1V/0.2V. The dVT Range may be approximately 0.3V
for a pn junction voltage V-pn of 0.6V.
[0066] In another embodiment, an NMOS pull-up load (such as its
source connected to ground) with a PWELL may be used, such as in a
triple well process. The PWELL voltage level (e.g., <VDD-VTN)
may be modulated to change the threshold voltage VT.
[0067] For an NWell embodiment, the voltage on the NWell may be set
greater than the supply voltage, e.g., by using a charge pump. The
voltage of the NWell may be set greater than the source voltage of
a pull-up PMOS transistor (for example, by regulating down source
voltage of the PMOS transistor).
[0068] FIG. 6 is a schematic diagram illustrating a bulk voltage
generator 600.
[0069] The bulk voltage generator 600 generates an adjustable bulk
voltage in response to selectable resistor tapping or bias current
modulation. The bulk voltage generator 600 comprises a plurality of
resistors 601-604, a current source 605, and a plurality of
switches 606-608. The resistor 601, 602, 603, 604 and the current
source 605 are coupled in series between a voltage supply VDD and
ground. Although four resistors are shown in FIG. 6, other numbers
of resistors may be used. The current source 605 generates a bias
current in response to digital to analog conversion of a digital
selection signal 611. The plurality of switches 606, 607, 608
selectively couple nodes between the resistors, which are arranged
as the voltage divider, to an output node 610. The output node 610
may be coupled to the bulk terminal 520 of the diode connected
transistor 515 (FIG. 5).
[0070] FIG. 7 is a schematic diagram illustrating a bulk voltage
generator 700.
[0071] The bulk voltage generator 700 generates an adjustable
well-bias voltage in response to selectable resistor tapping. The
voltage level may be set so that the voltage difference VDD minus
VWELL is less than the V-PN junction voltage to avoid forward
biasing the junction. The bulk voltage generator 700 comprises a
plurality of resistors 701 through 704, which are coupled in series
as a voltage divider between a voltage supply VDD and ground.
Although four resistors are shown in FIG. 7, other numbers of
resistors may be used. The bulk voltage generator 700 further
comprises a plurality of switches 706, 707 and 708, which
selectively couple nodes between the resistors 701, 702, 703 and
704, to an output node 710. The output node 710 may be coupled to
the bulk terminal 520 of the diode connected transistor 515 (FIG.
5).
[0072] FIG. 8 is a schematic diagram illustrating a sensing system
800.
[0073] The sensing system 800 comprises a reference column 501, a
plurality of PMOS transistors 801 and 802, and a comparator 803.
The PMOS transistors 801 and 802 are arranged as a buffer stage and
provide a load of a current mirror of the current sensed in the
reference column 501. A bulk terminal 820 provides a bulk voltage
to the PMOS transistor 802. The bulk voltage may be provided by a
voltage source, such as described below in conjunction with FIG.
9.
[0074] In illustrative embodiments of FIG. 8 and FIG. 10, described
below, the bulk substrate voltage VBS is used to change the PMOS
threshold voltage VTP to be greater than approximately 1.5
volts.
[0075] FIG. 9 is a schematic diagram illustrating a bulk voltage
generator 900.
[0076] The bulk voltage generator 900 generates an adjustable bulk
voltage in response to selectable resistor tapping or bias current
modulation. The bulk voltage generator 900 comprises a plurality of
current sources 901 and 902, a plurality of resistors 903 through
906, and a plurality of switches 907 through 909. The resistors 903
through 906 are coupled in series as a voltage divider between the
current sources 901 and 902. In one embodiment, the current sources
901 and 902 may generate fixed voltages. In another embodiment, one
or both of the current sources 901 and 902 may generate adjustable
current in response to a selection signal (not shown in FIG. 9).
Although four resistors are shown in FIG. 9, other numbers of
resistors may be used. The switches 907, 908, 909 selectively
couple nodes between the resistors 903, 904, 905 and 906 to an
output node 910. The output node 910 may be coupled to the bulk
terminal of the diode connected PMOS transistor 802 (FIG. 8).
[0077] FIG. 10 is a schematic diagram illustrating a sensing system
1000.
[0078] The sensing system 1000 comprises a reference column 501, a
plurality of PMOS transistors 1001, 1002 and 1003, a comparator
1005, and a plurality of switches 1006, 1007, 1008, and 1009. The
PMOS transistors 1001, 1002 and 1003 are arranged as a buffer and
provide a load of a current mirror of the current sensed in the
reference column 501. The switches 1106 and 1108 selectively couple
the bulk of the PMOS transistor 1002 to a voltage supply terminal
1111 and the source of the PMOS transistor 1002, respectively. The
switches 1007 and 1009 selectively couple the bulk of the PMOS
transistor 1003 to a voltage supply terminal 1112 and the source of
the PMOS transistor 1003, respectively. The bulk substrate voltage
VBS may be used to change the PMOS voltage threshold VTP to be
greater than about 1.5 volts. The switches 1006 and 1008 are used
to cause the bulk substrate voltage to switch the PMOS transistor
1002 on and off. The switches 1007 and 1009 are used to cause the
bulk substrate voltage to switch PMOS transistors 1003 on and
off.
[0079] FIG. 11 is a schematic diagram illustrating a sensing system
1100.
[0080] The sensing system 1100 comprises a reference column 1101
and a comparator 1103. The reference column 1101 comprises a
reference memory cell 1111, and an enable switch 1112 and a diode
connected NMOS transistor 1115. The switch 1112 selectively couples
the reference memory cell 1111 to a sense line 1104, which is
coupled to the comparator 1103. The switch 1112 may be an NMOS
transistor. The bitline resistor 1113 is shown to indicate
resistance on the bitline. A bitline capacitor 514 is shown to
indicate capacitance on the bitline. The bulk of the diode
connected NMOS transistor 1115 is biased by an adjustable voltage
applied to a bulk terminal 1120, which may be a voltage that is
different or less than the sense amplifier supply voltage. The bulk
voltage may be provided by a voltage source, such as described in
conjunction with FIGS. 6-7, 9 and 18.
[0081] The NMOS transistor 1115 may be formed in a separate PWELL
process. The p-well voltage V-PWELL may be trimmed from 0V to
(VDD+VTN).
[0082] FIG. 12 is a schematic diagram illustrating a sensing system
1200.
[0083] The sensing system 1200 comprises a reference column 501, a
NLZ NMOS transistor 1201, an NMOS transistor 1202, a comparator
1203, a plurality of resistors 1210, 1211, 1212, and 1213 and a
plurality of switches 1220, 1221, and 1222. The NMOS transistors
1201 and 1202 provide a buffer stage for the reference column 501.
The resistors 1210 through 1213 are coupled in series between the
source of the NMOS transistor 1201 and ground as a voltage divider.
The switches 1220, 1221 and 1222 selectively couple nodes of the
voltage divider to an input of the comparator 1203 for sensing. The
reference current is set equal to the data current. The reference
level is then trimmed using the switches 1220 through 1222 until
the comparator 1203 switches. In an illustrative embodiment, the
reference current and the data current are approximately 20
microamps.
[0084] FIG. 13 is a schematic diagram illustrating a sensing system
1300.
[0085] The sensing system 1300 comprises a reference column 501, an
NLZ NMOS transistor 1301, an NMOS transistor 1302, a comparator
1303, a plurality of resistors 1310, 1311, 1312, and 1313, and a
plurality of switches 1320, 1321, and 1322. The NMOS transistors
1301 and 1302 and the resistors 1310-1313 provide a buffer stage
for the reference column 501. The resistors 1310-1313 are coupled
in series between the source of the NMOS transistor 1301 and the
drain of the NMOS transistor 1302 to form a voltage divider between
the transistors 1301 and 1302. The switches 1320, 1321 and 1322
selectively couple nodes of the voltage divider to an input of the
comparator 1303 for sensing. The reference current is set equal to
the data current. The reference voltage is then trimmed using the
switches 1320-1322 until the comparator 1303 switches. In an
illustrative embodiment, the reference current and the data current
are approximately 20 microamps.
[0086] FIG. 14 is a schematic diagram illustrating a sensing system
1400.
[0087] The sensing system 1400 comprises a reference column 501, an
NLZ NMOS transistor 1401, a NMOS transistor 1402 and a comparator
1403. The transistors 1401 and 1402 are arranged as a buffer stage
to buffer the output of the reference column 501. The bias of the
NMOS transistor 1402 is adjusted until the comparator 1403
switches. In an illustrative embodiment, the reference current and
the data current are approximately 20 microamps.
[0088] FIG. 15 is a schematic diagram illustrating a sensing system
1500.
[0089] The sensing system 1500 comprises a reference column 501, a
PMOS transistor 1501, a diode connected NMOS transistor 1502, and a
comparator 1503. The PMOS transistor 1501 and the NMOS transistor
1502 provide a buffer stage for the reference column 501. The bulk
of the PMOS transistor 1501 is biased by an adjustable voltage
applied to the bulk terminal 1520, which may be at a voltage that
is different or less than the sense amplifier supply voltage. The
bulk voltage may be provided by a voltage source, such as described
above in conjunction with FIGS. 6-7.
[0090] FIG. 16 is a schematic diagram illustrating a sensing system
1600.
[0091] The sensing system 1600 comprises a reference column 501, a
PMOS transistor 1501, an NMOS transistor 1602, and a comparator
1503. The bulk of the diode connected NMOS transistor 1602 is
biased by an adjustable voltage applied to a bulk terminal 1620,
which may be at a voltage that is different or less than the sense
amplifier supply voltage. A NMOS transistor 1602 may be formed
using a triple well process, and the PWELL is isolated from the p
substrate. The bulk voltage may be provided by a voltage source,
such as described above in conjunction with FIGS. 6-7.
[0092] FIG. 17 is a schematic diagram illustrating a sensing system
1700.
[0093] The sensing system 1700 comprises a reference column 1701, a
plurality of PMOS transistors 1720 and 1721, a plurality of NMOS
transistors 1722 and 1723, and a comparator 1703. The reference
column 1701 comprises a reference memory cell 1711, an enable
switch 1712, and a diode connected PMOS transistor 1715. The PMOS
transistors 1720 and 1721 and the NMOS transistors 1722 and 1723
provide a two stage gain stage for the reference column 1701. The
NMOS transistors 1722 and 1723 may be formed using a triple well
process, and the PWELL is isolated from the p substrate. The bulk
of the PMOS transistors 1715, 1720, and 1721 are biased by an
adjustable voltage applied to a corresponding bulk terminal, which
may be at a voltage that is different or less than the sense
amplifier supply voltage. The bulk voltage may be provided by a
voltage source which is described above in conjunction with FIGS.
6-7. The bulk of the NMOS transistors 1722 and 1723 may be biased
by an adjustable voltage applied to a corresponding bulk terminal,
which may be at a voltage that is different or less than the sense
amplifier supply voltage. The bulk voltage may be provided by a
voltage source such as described below in conjunction with FIG.
18.
[0094] FIG. 18 is a schematic diagram illustrating a bulk voltage
generator 1800.
[0095] The bulk voltage generator 1800 comprises a current source
1801, a plurality of resistors 1803-1806, and a plurality of
switches 1807-1809. A current source 1801 and the resistors
1803-1806 are coupled in series between a supply voltage VDD and
ground. Although four resistors are shown in FIG. 18, other numbers
of resistors may be used. The current source 1801 generates a fixed
current for the voltage divider. The plurality of switches 1807,
1808, and 1809 selectively couple nodes between the resistors 1803,
1804, 1805, 1806, which are arranged as a voltage divider, to an
output node 1810. The output node 1810 may be coupled to the bulk
terminal of the NMOS transistors 1722 and 1723 (FIG. 17).
[0096] FIG. 19 is a schematic diagram illustrating a sensing system
1900.
[0097] The sensing system 1900 comprises a reference column 1701, a
plurality of PMOS transistors 1720 and 1721 and a plurality of NMOS
transistors 1722 and 1723 that are arranged in a manner similar to
the sensing system 1700 (FIG. 17). The sensing system 1900 further
comprises an NLZ NMOS transistor 1901 and an NMOS transistor 1902
arranged as a third buffer stage. The sensing system 1900 further
comprises a comparator 1703 coupled to the source of the NLZ NMOS
transistor 1901. The bulk of the NMOS transistors 1901 and 1902 may
be biased by an adjustable voltage applied to the bulk terminals.
The NMOS transistors 1722, 1723, and 1902 may be formed using a
triple well process, and the PWELL is isolated from the p
substrate.
[0098] FIG. 20 is a schematic diagram illustrating a differential
amplifier 2000.
[0099] The differential amplifiers described herein may be
implemented into operational amplifiers. The differential amplifier
2000 comprises a plurality of PMOS transistors 2001, 2002, and
2003, a plurality of NMOS transistors 2005 and 2006, and a
plurality of current sources 2010 and 2011. The current sources
2010 and 2011 are digitally programmable. The PMOS transistors
2001, 2002, and 2003 and the NMOS transistors 2005 and 2006 are
arranged as a differential amplifier in response to input signals
2020 and 2021 applied to the gates of the PMOS transistors 2002 and
2003, respectively. The current sources 2010 and 2011 are coupled
in parallel with the drain-source terminals of the NMOS transistors
2005 and 2006, respectively. The current sources 2010 and 2011
generate digital-to-analog conversion currents in response to
digital selection signals 2022 and 2023, respectively. The current
sources 2010 and 2011 provide an offset current to compensate for
the offset of the differential amplifier 2000. An n-well voltage
generator 2100 (see FIG. 21) may be used for the bulk of the PMOS
transistors 2001 through 2003. The MOS transistors 2005 and 2006
may be formed using a triple well process, and the PWELL is
isolated from the p substrate and its voltage can be trimmed. A
common mode node 2090 may be used to determine the bulk
voltage.
[0100] In illustrative embodiments of FIG. 20 and FIGS. 22-26,
described below, the bulk voltage VBS may be used to change the
threshold voltage VT, for example, at a rate dVT/dVBS=0.1V/0.2V.
The dVT Range may be approximately 0.3V for a pn junction voltage
V-pn of 0.6V.
[0101] FIG. 21 is a schematic diagram illustrating a bulk voltage
generator 2100.
[0102] The bulk voltage generator 2100 comprises a plurality of
resistors 2101, 2102, 2103, and 2104 and a current source 2105
coupled in series between a node 2120 and ground. The node 2120 may
be coupled to the node 2090 (FIG. 20). Although four resistors are
shown in FIG. 21, other numbers of resistors may be used. The bulk
voltage generator 2100 further comprises a plurality of switches
2106, 2107, 2108, which selectively couple nodes between the
resistors 2101, 2102, 2103, and 2104, to an output node 2110. The
output node 2110 may be coupled to the bulk of the NMOS transistors
2005 and 2006 (FIG. 20). The node 2110 may be coupled to the drain
of the PMOS transistor 2001 (FIG. 20). The current source 2105
generates a digital-to-analog conversion current in response to a
digital selection signal 2111.
[0103] FIG. 22 is a schematic diagram illustrating a differential
amplifier 2200.
[0104] The differential amplifier 2200 comprises a plurality of
PMOS transistors 2001, 2002 and 2003 and a plurality of NMOS
transistors 2005 and 2006 arranged in a similar manner as in the
differential amplifier 2000 (FIG. 20), and further includes a
buffer stage comprising a plurality of PMOS transistors 2212 and
2213 and a plurality of NMOS transistors 2215 and 2216. The MOS
transistors 2213 and 2216 are selectable (or trimmable) by digital
control bits to adjust for offset. The bulk of the MOS transistor
2213 and the NMOS transistor 2216 may be coupled to the bulk
voltage generator 2100 (FIG. 21). The MOS transistors 2215, 2216,
2005 and 2006 may be formed using a triple well process, and the
PWELL is isolated from the p substrate.
[0105] FIG. 23 is a schematic diagram illustrating a differential
amplifier 2300.
[0106] The differential amplifier 2300 comprises a plurality of
PMOS transistors 2301, 2302, and 2303, and a plurality of NMOS
transistors 2305 and 2306 arranged in a similar manner as the
differential amplifier 2000 (FIG. 20). The bulk of the PMOS
transistors 2302 and 2303 are coupled to a bulk voltage generator
2100 (FIG. 21), which is coupled between the common mode node
formed of the sources of the PMOS transistors 2302 and 2303 and
ground. The MOS transistors 2305 and 2306 may be formed using a
triple well process, and the PWELL is isolated from the p
substrate.
[0107] FIG. 24 is a schematic diagram illustrating a differential
amplifier 2400.
[0108] The differential amplifier 2400 comprises a plurality of
PMOS transistors 2401, 2402 and 2403, and a plurality of NMOS
transistors 2405 and 2406. The NMOS transistors 2405 and 2406 are
cross-coupled so that the gates of the transistors 2405 and 2406
are biased by the drain of the PMOS transistors 2403 and 2402,
respectively. The PMOS transistors 2402 and 2403 include a bulk
that is biased by the bulk voltage generator 2100 (FIG. 21) that is
coupled between the common mode node formed of the sources of the
PMOS transistors 2402 and 2403 and ground. The transistors 2405 and
2406 may be formed using a triple well process, and the PWELL is
isolated from the p substrate.
[0109] FIG. 25 is a schematic diagram illustrating a differential
amplifier 2500.
[0110] The differential amplifier 2500 comprises a differential
amplifier 2400 (FIG. 24) and a plurality of NMOS transistors 2510
and 2511. The NMOS transistors 2510 and 2511 provide current bias
and are coupled in parallel with the drain-source terminals of the
NMOS transistors 2405 and 2406, respectively.
[0111] FIG. 26 is a schematic diagram illustrating a differential
amplifier 2600.
[0112] The differential amplifier 2600 comprises an differential
amplifier 2300 (FIG. 23) and an autozero switch 1610. The autozero
switch 2610 autozeroes the drains of the PMOS transistors 2302 and
2303 before activation of the differential amplifier 2300.
[0113] FIG. 27 is a schematic diagram illustrating a differential
amplifier 2700.
[0114] The differential amplifier 2700 comprises a plurality of
PMOS transistors 2702 and 2703 and a plurality of NMOS transistors
2705 and 2706 that are arranged in a similar manner as the
differential amplifier 2300 (FIG. 23), but the PMOS transistors
2702 and 2703 are coupled to the supply voltage VDD instead of a
common mode node of a bias transistor. The bulk of the PMOS
transistors 2702 and 2703 may be coupled to the bulk voltage
generator 600 (FIG. 6). The voltage of the nwell is referenced to
the supply voltage VDD.
[0115] FIG. 28 is a schematic diagram illustrating a differential
amplifier 2800.
[0116] The differential amplifier 2800 comprises a differential
amplifier 2300 (FIG. 23), a PMOS transistor 2810, and an NMOS
transistor 2811. The transistors 2810 and 2811 are arranged as an
output stage to the operational amplifier 2300. The PMOS
transistors 2302 and 2303 of the differential amplifier 2300
include a bulk that is biased by the bulk voltage generator 2100
(FIG. 21), which is coupled between the common mode node formed of
the drain of the PMOS transistors 2302 and 2303 and ground. The
bulk of the PMOS transistor 2810 is biased by a voltage generator
600 (FIG. 6), which is referenced relative to the supply voltage
VDD.
[0117] FIG. 29 is a schematic diagram illustrating a differential
amplifier 2900.
[0118] The differential amplifier 2900 has an n-type differential
pair. The operational amplifier 2900 comprises plurality of PMOS
transistors 2902 and 2903 and a plurality of PMOS transistors 2905,
2906, and 2907 arranged as an differential amplifier. In response
to input signals 2920 and 2921 applied to the gates of the NMOS
transistors 2905 and 2906, respectively. The NMOS transistor 2907
provides bias to the differential amplifier 2900. The bulk of the
NMOS transistors 2905 and 2906 may be coupled to a bulk voltage
generator 3000 (FIG. 30). The voltage of an NWELL may be referenced
to the supply voltage VDD.
[0119] FIG. 30 is a schematic diagram illustrating a bulk voltage
generator 3000.
[0120] The voltage generator 3000 comprises a current source 3005
and a plurality of resistors 3001, 3002, 3003, and 3004 coupled in
series between a supply voltage and a node 3020, which may be
coupled to a common mode node 3020 or to ground. Although four
resistors are shown in FIG. 30, other numbers of resistors may be
used. The bulk voltage generator 3000 further comprises a plurality
of switches 3006, 3007, 3008, which selectively couple nodes
between the resistors 3001, 3002, 3003, 3004 to an output node
3010.
[0121] FIG. 31 is a schematic diagram illustrating a differential
amplifier 3100.
[0122] The differential amplifier 3100 comprises a differential
amplifier 2900 (FIG. 29) and a plurality of autozero switches 3101
and 3102. The autozero switch 3101 autozeroes the drain and gate of
the NMOS transistor 2906 before activation of the operational
amplifier 2900. The autozero switch 3102 autozeroes the drain and
gate of the NMOS transistor 2905 before activation of the
differential amplifier 2900.
[0123] FIG. 32 is a schematic diagram illustrating a memory cell
sensing system 3200.
[0124] The memory cell sensing system 3200 comprises a differential
amplifier 2300 (FIG. 23) and a sensing system 3201. The sensing
system 3201 comprises a memory cell column 3202 and a sensing stage
3203. The memory column 3202 comprises a reference memory cell 311,
an NMOS transistor 312 and a diode connected PMOS transistor 3215.
A bitline resistor 313 is shown to indicate resistance on the
bitline. A bitline capacitor 314 is shown to indicate capacitance
on a bitline. The data column 3202 provides an output voltage to
the sensing stage 3203 on the drain of the PMOS transistor 3215.
The bulk of the PMOS transistor 3215 may be adjustable. The sensing
stage 3203 comprises a PMOS transistor 3210 having a gate coupled
to the drain of the PMOS transistor 3215, and further comprises a
diode connected NMOS transistor 3211. The PMOS transistor 3210 and
the NMOS transistor 3211 may include a bulk that is coupled to an
adjustable voltage. The bulk of the transistors 3215, 3210, and
3211 may be coupled to a bulk voltage generator, such as the bulk
voltage generator 1800 (FIG. 18).
[0125] In illustrative embodiments of FIG. 32 and FIGS. 33-34,
described below, the bulk voltage VBS may be used to change the
threshold voltage VT, for example, at a rate dVT/dVBS=0.1V/0.2V.
The dVT Range may be approximately 0.3V for a pn junction voltage
V-pn of 0.6V.
[0126] FIG. 33 is a schematic diagram illustrating a memory cell
sensing system 3300.
[0127] The memory cell sensing system 3300 comprises a sensing
stage 3201 and an operational amplifier 2800.
[0128] FIG. 34 is a schematic diagram illustrating a memory cell
sensing system 3400.
[0129] The memory cell sensing system 3400 comprises a sensing
stage 3201 and an differential amplifier 3401. The differential
amplifier 3401 comprises a differential amplifier 2300 (FIG. 23), a
resistor 3410, and an NMOS transistor 3411. The resistor 3410 and
the drain-source terminals of the NMOS transistor 3411 are coupled
between the common mode node of the differential amplifier 2300 and
ground to adjust the voltage on the common mode node and to adjust
the bulk voltage of the PMOS transistors 2302 and 2303 of the
differential amplifier 2300. The resistor 3410 may be used to tap a
divided voltage for application to the bulk of the PMOS transistors
2302 and 2303. In an alternative embodiment, the bias current on
the gate of the NMOS transistor 3411 may be modulated to adjust the
tap voltage.
[0130] FIG. 35 is a diagram illustrating voltages for memory
levels.
[0131] As an illustrative embodiment, a two-bit memory cell system
is described. The voltage levels, Level 0, Level 1, and Level 2,
are used to divide the voltage range into two-bit data 00, 01, 10,
and 11. At low levels, the speed of sensing slows down which
implies that an offset addition may be used to speed up the
differential amp timing. As shown in FIG. 35, Level 0 has an offset
3501 that is greater than the offset 3502 for Level 1 and 3503 for
Level 2. The offset addition may be applied at the differential
amplifier or the load to compensate for the offset at a pull up
load for a differential amplifier or other systematic offset, for
example, from supply voltage VDD variation, interconnect mismatch,
current dependent speed mismatch, or decoding path mismatch. The
offset may be created by a combination, such as width/length
trimming of transistors, or well modulation. Different offset range
may be used for different levels. The offset may be used as a
margin check for each level.
[0132] FIG. 36 is a block diagram illustrating a digital autozero
control system 3600.
[0133] The autozero control system 3600 comprises a comparator 3601
and a control circuit 3602. The comparator 3601 may be, for
example, one of the differential amplifiers described above. The
control circuit 3602 provides a bias current in response to the
output of the comparator 3601. The control circuit 3602 comprises
an N-bit increment counter 3610 and an N-bit digital to current
converter 3611. The well voltage is started with a low offset and
gradually increased as the N-bit increment counter 3610 counts
until the comparator 3601 switches. The corresponding parameters,
such as the count in the counter 3610, are stored in volatile or
nonvolatile memory.
[0134] FIG. 37 is a flowchart illustrating the operation of the
control system 3600.
[0135] At autozero operation is commenced (block 3701). The voltage
compensation is compared to zero and if it is zero, the autozero is
completed (3704). Otherwise, the current IV is incremented (block
3703) and the voltage compensation is again analyzed (3702).
[0136] FIG. 38 is a flowchart illustrating an analog control system
3800.
[0137] The control system 3800 comprises a comparator 3601 and a
control circuit 3802. The control circuit 3802 is an analog
circuit. The control system 3800 operates in a similar manner as
the control system 3600, but the block 3703 of FIG. 31 is an
increasing bias current instead of an incremented bias current. The
control circuit 3802 operates as a voltage to current converter.
The control circuit 3802 comprises a current source 3810, a PMOS
transistor 3811, a plurality of NMOS transistors 3812 and 3813, a
capacitor 3814, and a resistor 3815. The output of the comparator
3601 is applied to the gate of the NMOS transistor 3812 which
controls the charging of the capacitor 3814, and generates a
voltage VH to bias the gate of the NMOS transistor 3813. The diode
connected PMOS transistor 3811 generates a bias current IB that
controls the comparator 3601. The voltage of the VWELL is started
from a low offset and gradually increased until the comparator 3601
switches to shut off the voltage to current conversion of the
control circuit 3802. The analog voltage VH is stored either as
volatile or non-volatile.
[0138] In the foregoing description, various methods and apparatus,
and specific embodiments are described. However, it should be
obvious to one conversant in the art, various alternatives,
modifications, and changes may be possible without departing from
the spirit and the scope of the invention which is defined by the
metes and bounds of the appended claims.
* * * * *