U.S. patent application number 12/666603 was filed with the patent office on 2010-12-23 for direct power converting apparatus.
Invention is credited to Kenichi Sakakibara.
Application Number | 20100321965 12/666603 |
Document ID | / |
Family ID | 40341384 |
Filed Date | 2010-12-23 |
United States Patent
Application |
20100321965 |
Kind Code |
A1 |
Sakakibara; Kenichi |
December 23, 2010 |
DIRECT POWER CONVERTING APPARATUS
Abstract
A transistor is brought into conduction when, for example, a
voltage between both ends of a second clamp capacitor exceeds a
predetermined reference voltage. A resistance value of a discharge
resistor is smaller than a value obtained by dividing the reference
voltage by the maximum value of a current flowing through the
discharge resistor. When the transistor is brought into conduction
as a result of a voltage between both ends of the second clamp
capacitor exceeding the predetermined reference voltage, a voltage
applied to the discharge resistor, which results from a
regenerative current, is larger one of the voltage between both
ends of the second clamp capacitor and a voltage drop of the
discharge resistor due to the regenerative current. The voltage
drop and the voltage between both ends are smaller than a voltage
between DC power supply lines, whereby it is possible to reduce an
electrostatic capacitance of the discharge resistor.
Inventors: |
Sakakibara; Kenichi; (Shiga,
JP) |
Correspondence
Address: |
BIRCH STEWART KOLASCH & BIRCH
PO BOX 747
FALLS CHURCH
VA
22040-0747
US
|
Family ID: |
40341384 |
Appl. No.: |
12/666603 |
Filed: |
August 6, 2008 |
PCT Filed: |
August 6, 2008 |
PCT NO: |
PCT/JP2008/064125 |
371 Date: |
December 23, 2009 |
Current U.S.
Class: |
363/37 |
Current CPC
Class: |
H02M 7/797 20130101;
H02M 5/297 20130101; H02M 2001/322 20130101; H02M 5/4585 20130101;
H02M 7/219 20130101 |
Class at
Publication: |
363/37 |
International
Class: |
H02M 5/458 20060101
H02M005/458 |
Foreign Application Data
Date |
Code |
Application Number |
Aug 7, 2007 |
JP |
2007-205602 |
Claims
1. A direct power converting apparatus, comprising: a positive-side
DC power supply line; a negative-side DC power supply line to which
a potential lower than a potential applied to said positive-side DC
power supply line is applied; input capacitors each connected
between ones of a plurality of input lines connected to a
multi-phase AC power supply and functioning as a power supply; a
current-source power converter converting a multi-phase AC voltage
input from said input lines into a square-wave-shape DC voltage
having two potentials, and supplying said positive-side DC power
supply line and said negative-side DC power supply line with said
DC voltage; a voltage-source power converter converting said DC
voltage having two potentials between said positive-side DC power
supply line and said negative-side DC power supply line into a
square-wave-shape AC voltage, and outputting said square-wave-shape
AC voltage to an inductive multi-phase load; a first capacitance
device having one end connected to said positive-side DC power
supply line; a second capacitance device connected to another end
of said first capacitance device and said negative-side DC power
supply line; a first diode device having, between said first
capacitance device and said second capacitance device, an anode
connected to said first capacitance device and a cathode connected
to said second capacitance device; a second diode device having an
anode connected to a point between said second capacitance device
and said first diode device and a cathode connected to said
positive-side DC power supply line; a third diode device having an
anode connected to said negative-side DC power supply line and a
cathode connected to a point between said first capacitance device
and said first diode; a discharge resistor connected between said
positive-side DC power supply line and said negative-side DC power
supply line; and a switching device connected in series with said
discharge resistor between said positive-side DC power supply line
and said negative-side DC power supply line, being brought into
conduction when a voltage between both ends of said first
capacitance device or said second capacitance device exceeds a
first predetermined value, and being interrupted when the voltage
falls below a second predetermined value.
2. The direct power converting apparatus according to claim 1,
wherein a resistance value of said discharge resistor is equal to
or smaller than a value obtained by dividing said first
predetermined value by a maximum value of a current flowing through
said discharge resistor.
3. The direct power converting apparatus according to claim 1,
further comprising: a third capacitance device connected between
said first diode device and said second capacitance device; a
fourth diode device having, between said second capacitance device
and said third capacitance device, an anode connected to said third
capacitance device and a cathode connected to said second
capacitance device and said second diode device; a fifth diode
device having an anode connected to a point between said first
diode device and said third capacitance device and a cathode
connected to said positive-side DC power supply line; and a sixth
diode device having an anode connected to said negative-side DC
power supply line and a cathode connected to a point between said
fourth diode device and said third capacitance device.
4. The direct power converting apparatus according to claim 2,
further comprising: a third capacitance device connected between
said first diode device and said second capacitance device; a
fourth diode device having, between said second capacitance device
and said third capacitance device, an anode connected to said third
capacitance device and a cathode connected to said second
capacitance device and said second diode device; a fifth diode
device having an anode connected to a point between said first
diode device and said third capacitance device and a cathode
connected to said positive-side DC power supply line; and a sixth
diode device having an anode connected to said negative-side DC
power supply line and a cathode connected to a point between said
fourth diode device and said third capacitance device.
5. The direct power converting apparatus according to claim 1,
wherein said voltage-source power converter and said switching
device are composed of a power integrated module.
6. The direct power converting apparatus according to claim 2,
wherein said voltage-source power converter and said switching
device are composed of a power integrated module.
7. The direct power converting apparatus according to claim 3,
wherein said voltage-source power converter and said switching
device are composed of a power integrated module.
8. The direct power converting apparatus according to claim 4,
wherein said voltage-source power converter and said switching
device are composed of a power integrated module.
9. The direct power converting apparatus according to claim 5,
wherein said voltage-source power converter and said switching
device are composed of a power integrated module.
Description
TECHNICAL FIELD
[0001] The present invention relates to a direct power converting
apparatus, and more particularly, to a direct power converting
apparatus including a clamp circuit in a DC link section.
BACKGROUND ART
[0002] Non-Patent Document 1, which will be described below,
discloses a direct AC power converting apparatus including a clamp
circuit. FIG. 9 shows the direct AC power converting apparatus
described in Non-Patent Document 1. It is assumed here that an IPM
motor is provided on an output side of this direct AC power
converting apparatus. When La represents an inductance per phase
which corresponds to an average value of effective inductances of
the IPM motor, i represents overload current which serves as a
reference for interrupting current supply to the IPM motor, Vc
represents voltage between both ends of a clamp capacitor, Cc
represents electrostatic capacitance of the clamp capacitor, and Vs
represents line voltage of a three-phase AC power supply, and when
all power stored in an inductor for three phases of the IPM motor
is regenerated to the clamp capacitor, the following relational
expression is satisfied.
[ Expression 1 ] 1 2 La ( i 2 + ( i 2 ) 2 + ( i 2 ) 2 ) = 1 2 Cc (
Vc 2 - ( 2 Vs ) 2 ) ( 1 ) ##EQU00001##
[0003] Therefore, the voltage between both ends of the clamp
capacitor is expressed by the following expression.
[ Expression 2 ] Vc = 3 2 La Cc i 2 + 2 Vs 2 ( 2 ) ##EQU00002##
[0004] FIG. 10 shows the relationship between voltage between both
ends and electrostatic capacitance of the clamp capacitor, which is
based on Expression (2). For example, if the power supply voltage
Vs is 400 V, the inductance La is 12 mH, the overload current i is
40 A, and the electrostatic capacitance of the clamp capacitor is
10 .mu.F, the voltage Vc between both ends of the clamp capacitor
is approximately 1,800 V. The voltage value exceeds device rating
1,200 V of a transistor and a diode with power supply voltage of
400 V class.
[0005] In order to keep the voltage Vc between both ends of the
clamp capacitor at approximately 750 V or lower, the electrostatic
capacitance of the clamp capacitor needs to be 200 .mu.F or larger
from Expression (2) and FIG. 10.
[0006] On the other hand, inrush current at power-on increases as
the electrostatic capacitance of the clamp capacitor is increased.
Here, a series circuit in which a power supply, a reactor, a
resistor and a capacitor are connected in series is taken as an
example of a series circuit for one phase, where L represents an
inductance of the reactor, R represents a resistance value of the
resistor, and C represents electrostatic capacitance of the clamp
capacitor. Then, a transfer characteristic of output (current) to
input (power supply voltage Vs) in the series circuit is expressed
by the following expression.
[ Expression 3 ] G ( s ) = ic Vs = sC 1 / LC s 2 + sR / L + 1 / LC
( 3 ) ##EQU00003##
[0007] The response to step input is expressed by the following
expression.
[ Expression 4 ] G ( s ) = sC 1 / LC s 2 + sR / L + 1 / LC 1 s = 1
/ L s 2 + sR / L + 1 / LC ( 4 ) ##EQU00004##
[0008] Here, Expression (4) is subjected to inverse Laplace
transform to obtain the response of current assuming that 1/L=D,
R/L=E and 1/LC=F, the following expression is derived.
[ Expression 5 ] i ( t ) = D .omega. - .alpha. sin .omega. t ( 5 )
[ Expression 6 ] .omega. = 4 F - E 2 2 , .sigma. = E 2 ( 6 )
##EQU00005##
[0009] F decreases as the electrostatic capacitance C of the
capacitor increases, and D and E remain constant irrespective of
the electrostatic capacitance C, and thus .omega. decreases as the
electrostatic capacitance C of the capacitor increases.
Accordingly, an amplitude term D/.omega. excluding attenuation
through time increases as the electrostatic capacitance C of the
capacitor increases. That is, inrush current increases along with
an increase in electrostatic capacitance C of the capacitor.
[0010] When the maximum value of current is obtained assuming that
a value obtained by differentiating i(t) with respect to time is 0
(i(t)'=0) from Expression (5), the following expression is
derived.
[ Expression 7 ] t = .pi. - .alpha. .omega. ( 7 ) ##EQU00006##
[0011] The maximum value is regarded as inrush current. FIG. 11
shows the relationship between inrush current
(i((.pi.-.alpha.)/.omega.)) and the electrostatic capacitance
C.
[0012] As described above, the voltage between both ends of the
clamp capacitor charged with the regenerative current is
approximately equal to or lower than 750 V, and accordingly if the
electrostatic capacitance of the clamp capacitor is 200 the maximum
value (inrush current) of current reaches 150 A from Expressions
(6) and (7) and FIG. 11.
[0013] In Non-Patent Document 1, for reducing the above-mentioned
inrush current and also reducing the voltage between both ends of
the clamp capacitor charged with the regenerative current, a
discharge circuit is provided in the clamp capacitor. More
specifically, the discharge circuit includes a discharge resistor
connected in parallel with the clamp capacitor. The inrush current
is reduced by reducing the electrostatic capacitance of the clamp
capacitor, and charges charged in the clamp capacitor are
discharged to the discharge resistor when the voltage between both
ends of the clamp capacitor exceeds a predetermined reference
voltage due to the regenerative current, whereby the voltage
between both ends is suppressed from increasing.
[0014] Note that Patent Documents 1 to 4 disclose the technologies
related to the present invention.
[0015] Non-Patent Document 1: J. Schoenberger, T. Friedli, S. D.
Round, J. W. Kolar, "An ultra sparse matrix converter with a novel
active clamp circuit", Proc. of the 4th power conversion conference
(PCC '07), pp. 784-791
[0016] Patent Document 1: U.S. Pat. No. 6,995,992
[0017] Patent Document 2: Japanese Patent Application Laid-Open No.
2006-54947
[0018] Patent Document 3: Japanese Patent Application Laid-Open No.
02-65667
[0019] Patent Document 4: Japanese Patent Publication No.
62-53918
DISCLOSURE OF INVENTION
Problem to be Solved by the Invention
[0020] However, in the technology described in Non-Patent Document
1, approximately same amount of voltage as the voltage between both
ends (=reference voltage) of the clamp capacitor is applied to the
discharge resistor, and hence the discharge resistor requires power
capacity equal to or more than (reference voltage).times.(reference
voltage)/(resistance value).
[0021] An object of the present invention is therefore to provide a
direct power converting apparatus capable of reducing power
capacity required by a discharge resistor.
Means to Solve the Problem
[0022] According to a first aspect of the present invention, a
direct power converting apparatus includes: a positive-side DC
power supply line (L1); a negative-side DC power supply line (L2)
to which a potential lower than a potential applied to the
positive-side DC power supply line is applied; input capacitors
(Cr, Cs, Ct) each connected between ones of a plurality of input
lines connected to a multi-phase AC power supply and functioning as
a power supply; a current-source power converter (1) converting a
multi-phase AC voltage input from the input lines into a
square-wave-shape DC voltage having two potentials, and supplying
the positive-side DC power supply line and the negative-side DC
power supply line with the DC voltage; a voltage-source power
converter (4) converting the DC voltage having two potentials
between the positive-side DC power supply line and the
negative-side DC power supply line into a square-wave-shape AC
voltage, and outputting the square-wave-shape AC voltage to an
inductive multi-phase load (5); a first capacitance device (C1)
having one end connected to the positive-side DC power supply line;
a second capacitance device (C2) connected to another end of the
first capacitance device and the negative-side DC power supply
line; a first diode device (D1) having, between the first
capacitance device and the second capacitance device, an anode
connected to the first capacitance device and a cathode connected
to the second capacitance device; a second diode device (D2) having
an anode connected to a point between the second capacitance device
and the first diode device and a cathode connected to the
positive-side DC power supply line; a third diode device (D3)
having an anode connected to the negative-side DC power supply line
and a cathode connected to a point between the first capacitance
device and the first diode; a discharge resistor (R1) connected
between the positive-side DC power supply line and the
negative-side DC power supply line; and a switching device (S1)
connected in series with the discharge resistor between the
positive-side DC power supply line and the negative-side DC power
supply line, being brought into conduction when a voltage (Vc1)
between both ends of the first capacitance device or the second
capacitance device exceeds a first predetermined value (Vref-h),
and being interrupted when the voltage falls below a second
predetermined value (Vref-L).
[0023] According to a second aspect of the direct power converting
apparatus of the present invention, in the direct power converting
apparatus according to the first aspect, a resistance value of the
discharge resistor (R1) is equal to or smaller than a value
obtained by dividing the predetermined value (Vref-h) by a maximum
value of a current flowing through the discharge resistor.
[0024] According to a third aspect of the direct power converting
apparatus of the present invention, in the direct power converting
apparatus according to the first or second aspect, which further
includes: a third capacitance device (C3) connected between the
first diode device (D1) and the second capacitance device (C2); a
fourth diode device (D6) having, between the second capacitance
device and the third capacitance device, an anode connected to the
third capacitance device and a cathode connected to the second
capacitance device and the second diode device (D2); a fifth diode
device (D7) having an anode connected to a point between the first
diode device and the third capacitance device and a cathode
connected to the positive-side DC power supply line; and a sixth
diode device (D8) having an anode connected to the negative-side DC
power supply line and a cathode connected to a point between the
fourth diode device and the third capacitance device.
[0025] According to a fourth aspect of the direct power converting
apparatus of the present invention, in the direct power converting
apparatus according to any one of the first to third aspects, the
voltage-source power converter and the switching device are
composed of a power integrated module (PIM).
EFFECTS OF THE INVENTION
[0026] According to the first aspect of the direct power converting
apparatus of the present invention, the first capacitance device
and the second capacitance device are charged with a regenerative
current from the inductive multi-phase load. In this case, the
first capacitance device and the second capacitance device are
charged in the state of being connected in series with each other
by rectifying functions of the first diode device to the third
diode device (see FIG. 3). The first capacitance device and the
second capacitance device divide a voltage between the
positive-side DC power supply line and the negative-side DC power
supply line, whereby it is possible to reduce the breakdown
voltages of the first capacitance device and the second capacitance
device.
[0027] Then, the switching device is brought into conduction when
the voltage between both ends of the first capacitance device or
the second capacitance device exceeds the first predetermined
value. On this occasion, the first capacitance device and the
second capacitance device are discharged in the state of being
connected in parallel with each other to the discharge resistor by
the rectifying functions of the first diode device to the third
diode device (see FIG. 4). The first capacitance device and the
second capacitance device are discharged in this manner, whereby it
is possible to suppress the voltage between both ends of the first
capacitance device and the voltage between both ends of the second
capacitance device from increasing due to the regenerative
current.
[0028] Further, it is possible to apply, to the discharge resistor,
the voltages between both ends of a pair of the first capacitance
device and the second capacitance device. Accordingly, compared
with a mode in which one clamp capacitor is provide between a
positive-side DC power supply line and a negative-side DC power
supply line, the electrostatic capacitance required by a discharge
resistor can be reduced.
[0029] According to the second aspect of the direct power
converting apparatus of the present invention, the voltage between
both ends of the discharging resistor when the switching device is
brought into conduction and thus the largest current flows through
the discharging resistor is smaller than the first predetermined
value. The voltage between both ends is equal to the voltages
between both ends of the first capacitance device and the second
capacitance device. Therefore, even in a case where the voltages
between both ends are the largest (the largest current flows
through the discharge resistor), the switching device can be
prevented from being in conduction for a long period of time, and
accordingly a time rating of the switching device can be
reduced.
[0030] According to the third aspect of the direct power converting
apparatus of the present invention, in conduction of the switching
device, the first capacitance device to the third capacitance
device are discharged to the discharge resistor in the state of
being connected in parallel with each other by the rectifying
functions of the first diode device to the sixth diode device.
Accordingly, compared with the case where the first capacitance
device to the third capacitance device are discharged in the state
of being connected in series with each other, the power capacity of
the discharge resistor can be reduced further.
[0031] According to the fourth aspect of the direct power
converting apparatus of the present invention, the voltage-source
power converter and the switching device can be manufactured
integrally, and thus are widely used in an indirect AC power
converting apparatus. Accordingly, the direct power converting
apparatus can be configured to be compact in size at low cost.
[0032] These and other objects, features, aspects and advantages of
the present invention will become more apparent from the following
detailed description of the present invention when taken in
conjunction with the accompanying drawings.
BRIEF DESCRIPTION OF DRAWINGS
[0033] FIG. 1 is a conceptual configuration diagram of a motor
driving device.
[0034] FIG. 2 is a conceptual configuration diagram of a circuit
for outputting a switch signal to a transistor included in a brake
circuit.
[0035] FIG. 3 is a diagram showing a state in which a clamp
capacitor is charged.
[0036] FIG. 4 is a diagram showing a state in which the clamp
capacitor is discharged.
[0037] FIG. 5 is a graph showing currents flowing through coils
included in a motor, currents flowing through DC power supply
lines, voltage between both ends of one clamp capacitor, voltages
between both ends of a pair of clamp capacitors, voltage between
the DC power supply lines, and current flowing through a discharge
resistor (transistor) in a motor driving device according to a
first embodiment.
[0038] FIG. 6 is a graph showing currents flowing through coils
included in a motor, currents flowing through DC power supply
lines, voltage between both ends of one clamp capacitor, voltages
between both ends of a pair of clamp capacitors, voltage between
the DC power supply lines, and current flowing through a discharge
resistor (transistor) in a motor driving device according to a
second embodiment.
[0039] FIG. 7 is another graph showing currents flowing through
coils included in a motor, currents flowing through DC power supply
lines, voltage between both ends of one clamp capacitor, voltages
between both ends of a pair of clamp capacitors, voltage between
the DC power supply lines, and current flowing through a discharge
resistor (transistor) in the motor driving device according to the
second embodiment.
[0040] FIG. 8 is a conceptual configuration diagram of a clamp
circuit included in a motor driving device according to a third
embodiment.
[0041] FIG. 9 is a diagram showing a power converting apparatus of
Non-Patent Document 1.
[0042] FIG. 10 is a graph showing a relationship between
electrostatic capacitance of clamp capacitor and a voltage between
both ends of the clamp capacitor.
[0043] FIG. 11 is a graph showing a relationship between the
electrostatic capacitance of the clamp capacitor and inrush current
of the clamp capacitor.
BEST MODE FOR CARRYING OUT THE INVENTION
First Embodiment
[0044] FIG. 1 is a conceptual configuration diagram of a motor
driving device as an example of a direct power converting apparatus
according to a first embodiment of the present invention. The motor
driving device includes a power supply E1, input lines ACLr, ACLs
and ACLt, reactors Lr, Ls and Lt, capacitors Cr, Cs and Ct, a
current-source converter 1, DC power supply lines L1 and L2, a
clamp circuit 2, a brake circuit 3, a voltage-source inverter 4 and
a motor 5.
[0045] The power supply E1 is a multi-phase AC power supply, which
is, for example, a three-phase AC power supply, and supplies the
input lines ACLr, ACLs and ACLt with three-phase AC current.
[0046] The reactors Lr, Ls and Lt are provided on the input lines
ACLr, ACLs and ACLt, respectively.
[0047] Each of the capacitors Cr, Cs and Ct is connected between
ones of the input lines ACLr, ACLs and ACLt. That is, the capacitor
Cr is provided between the input lines ACLr and ACLs, the capacitor
Cs is provided between the input lines ACLs and ACLt, and the
capacitor Ct is provided between the input lines ACLt and ACLr.
More specifically, one ends thereof are connected to the reactors
Lr, Ls and Lt on a side opposite to the power supply E1, and the
other ends thereof are connected to each other. Those are provided
on an input side of the current-source converter 1 and function as
a voltage source. The capacitors Cr, Cs and Ct constitute an LC
filter which suppresses switching current, together with the
reactors Lr, Ls and Lt.
[0048] The current-source converter 1 is connected to the power
supply E1 via the LC filter, and converts a multi-phase AC voltage
input from the input lines ACLr, ACLs and ACLt into a
square-wave-like DC voltage having two potentials to supply the DC
voltage between the DC power lines L1 and L2 (see voltage wave form
between the DC power supply lines L1 and L2 of FIGS. 5 to 7, which
will be described below).
[0049] More specifically, the current-source converter 1 includes
transistors Sip, Sm, Ssp, Ssn, Stp and Stn, and diodes Drp, Drn,
Dsp, Dsn, Dtp and Dtn.
[0050] Respective cathodes of the diodes Drp, Dsp and Dtp are
connected to the DC power supply line L1. Respective anodes of the
diodes Dm, Dsn and Dtn are connected to the DC power supply line
L2.
[0051] Emitters of the transistors Srp, Ssp and Stp are connected
to anodes of the diodes Drp, Dsp and Dtp, respectively. Collectors
of the transistors Sm, Ssn and Stn are connected to cathodes of the
diodes Dm, Dsn and Dtn, respectively. A collector of the transistor
Srp and an emitter of the transistor Sm, a collector of the
transistor Ssp and an emitter of the transistor Ssn, and a
collector of the transistor Stp and an emitter of the transistor
Stn are connected in common to the input lines ACLr, ACLs and ACLt,
respectively.
[0052] Respective bases of those transistors Srp, Sm, Ssp, Ssn, Stp
and Stn are supplied with a switch signal by a control section (not
shown) or the like, and the current-source converter 1 converts the
three-phase AC voltage into a square-wave-shape DC voltage having
two potentials. Note that the DC power supply line L1 is regarded
as a positive-side DC power supply line, and the DC power supply
line L2 is regarded as a negative-side DC power supply line to
which a potential lower than a potential applied to the DC power
supply line L1 is applied.
[0053] The clamp circuit 2 includes at least two clamp capacitors.
Those two clamp capacitors are charged in a state of being
connected in series with each other so as to have a voltage higher
than the higher potential of the square-wave-shape voltage, and are
discharged in a state of being connected in parallel with each
other so as to have a voltage lower than the lower potential of the
square-wave-shape voltage. In this manner, the clamp circuit acts
in a steady state, through the above-mentioned charging/discharging
operation, so as to balance a voltage when the discharge current is
higher than the charge current. More specifically, the clamp
circuit 2 includes clamp capacitors C1 and C2 and diodes D1 to
D3.
[0054] The clamp capacitor C1 has one end connected to the DC power
supply line L1. The clamp capacitor C2 is connected to the other
end of the clamp capacitor C1 and the DC power supply line L2. That
is, the clamp capacitors C1 and C2 are connected in series with
each other between the DC power supply lines L1 and L2.
[0055] Between the clamp capacitors C1 and C2, the diode D1 has an
anode and a cathode connected to the clamp capacitor C1 and the
clamp capacitor C2, respectively. The diode D2 has an anode
connected to a point between the clamp capacitor C2 and the diode
D1 and a cathode connected to the DC power supply line L1. The
diode D3 has an anode connected to the DC power supply line L2 and
a cathode connected to a point between the clamp capacitor C1 and
the diode D1.
[0056] The brake circuit 3 includes a discharge resistor R1, a
transistor S1 and diodes D4 and D5. The discharge resistor R1 is
connected between the DC power supply lines L1 and L2. The
transistor S1 is connected in series with the discharge resistor
R1. The diode D4 has an anode connected to a point between the
discharge resistor R1 and the transistor S1 and a cathode connected
to the DC power supply line L1. The diode D5 has an anode connected
to an emitter of the transistor S1 and a cathode connected to a
collector of the transistor S1.
[0057] The transistor S1 is brought into conduction when at least
any of the voltages between both ends of the clamp capacitors C1
and C2 exceeds a predetermined value. For example, FIG. 2 shows an
example of a circuit for outputting a switch signal to the
transistor S1. A differential amplifier 6 has a non-inverting input
terminal to which a voltage Vc1 between both ends of the clamp
capacitor 2 is applied and an inverting input terminal to which a
reference voltage Vref (though not shown herein, the differential
amplifier has hysteresis characteristics of reference voltages
Vref-h and Vref-L based on the reference voltage) serving as a
reference of the predetermined value is applied. An output of the
differential amplifier 6 is input to a base of the transistor S1 as
a switch signal.
[0058] A resistance value r1 of the discharge resistor R1 is
smaller than a value obtained by dividing a value, which is
obtained by multiplying the reference voltage Vref-h by the number
of the clamp capacitors C1 and C2, by the maximum value Imax of the
current flowing through the discharge resistor R1. That is,
r1<2Vref-h/Imax (hereinafter, referred to as Expression (8)) is
satisfied. This will be described below in detail.
[0059] The voltage-source inverter 4 converts the square-wave-shape
DC voltage having two potentials between the DC power supply lines
L1 and L2 into a square-wave-shape AC voltage and outputs the
square-wave-shape AC voltage to the motor 5. More specifically, the
voltage-source inverter 4 includes transistors Sup, Sun, Svp, Svn,
Swp and Swn and diodes Dup, Dun, Dvp, Dvn, Dwp and Dwn.
[0060] Respective collectors of the transistors Sup, Svp and Swp
and respective cathodes of the diodes Dup, Dvp and Dwp are
connected to the DC power supply line L1, and respective emitters
of the transistors Sun, Svn and Swn and respective anodes of the
diodes Dun, Dvn and Dwn are connected to the DC power supply line
L2.
[0061] An emitter of the transistor Sup, a collector of the
transistor Sun, an anode of the diode Dup and a cathode of the
diode Dun are connected in common to the motor 5, an emitter of the
transistor Svp, a collector of the transistor Svn, an anode of the
diode Dvp and a cathode of the diode Dvn are connected in common to
the motor 5, and an emitter of the transistor Swp, a collector of
the transistor Swn, an anode of the diode Dwp and a cathode of the
diode Dwn are connected in common to the motor 5.
[0062] Bases of those transistors Sup, Sun, Svp, Svn, Swp and Swn
are supplied with the switch signal by the control section (not
shown) or the like, and the voltage-source inverter 4 converts the
square-wave-shape DC voltage having two potentials between the DC
power supply lines L1 and L2 into a square-wave-shape AC voltage
and outputs the square-wave-shape AC voltage to the motor 5.
[0063] The motor 5 is, for example, a three-phase AC motor, and an
inductance component and a resistance component thereof are
represented by coils Lu, Lv and Lw, and resistors Ru, Rv and Rw,
respectively. The coils Lu, Lv and Lw are connected in series with
the resistors Ru, Rv and Rw, respectively. One ends of the coils
Lu, Lv and Lw on a side opposite to the resistors Ru, Rv and Rw are
connected to a point between the transistors Sup and Sun, between
the transistors Svp and Svn, and between the transistors Swp and
Swn, respectively. One ends of the resistors Ru, Rv and Rw on a
side opposite to the coils Lu, Lv and Lw are connected in common at
a neutral point P.
[0064] The motor 5 is supplied with the square-wave-shape AC
voltage from the voltage-source inverter 4. Thanks to the
inductance component of the motor 5, an AC current for driving the
motor 5 is smoothed. In other words, the motor 5 converts the
square-wave-shape AC voltage supplied from the voltage-source
inverter 4 into the AC current.
[0065] The capacitors Cr, Cs and Ct are charged with this AC
current flowing through the motor 5 via the voltage-source inverter
4 and the current-source converter 1, which is converted into the
AC voltage. In other words, the motor 5 is regarded also as a
current source for the current-source converter 1.
[0066] According to the clamp circuit 2 of the motor driving device
having the above-mentioned configuration, in a case where the
current flowing through the motor 5 delays with respect to the
voltage between the DC power supply lines L1 and L2 due to a load
power factor of the side of the voltage-source inverter 4, during a
predetermined period of time, a reflux current flows from the motor
5 to the DC power supply lines L1 and L2, whereby the clamp
capacitors C1 and C2 are charged in the state of being connected in
series with each other. The charging voltage (voltages between both
ends of a pair of the clamp capacitors C1 and C2) on this occasion
is also determined based on the load power factor. On the other
hand, when the voltages between both ends of the clamp capacitors
C1 and C2 rise to exceed the lower voltage of the square-wave-shape
voltage between the DC power supply lines L1 and L2, the clamp
capacitors C1 and C2 are discharged in the state of being connected
in parallel with each other. Note that the clamp capacitors C1 and
C2 are charged in the sate of being connected in series with each
other and discharged in the state of being connected in parallel
with each other, and thus the discharging voltage is a half of the
charging voltage.
[0067] Through the charging/discharging operation as described
above, the voltages of the clamp capacitors C1 and C2 are balanced
in a case where the discharging current is larger than the charging
current.
[0068] As described above, the reflux current of the motor 5 is
charged, and is discharged again to be supplied to the motor 5,
with the result that the motor 5 is driven efficiently. In
addition, the clamp circuit 2 does not require a so-called active
device such as a switching device, whereby power consumption and
manufacturing cost are reduced.
[0069] Further, in a case where an operating current to the motor 5
is reduced (the motor 5 is decelerated) or in a case where supply
of the operating current to the motor 5 is stopped, the
regenerative current from the motor 5 is supplied to the clamp
capacitors C1 and C2. Also in this case, the clamp capacitors C1
and C2 are charged in the state of being connected in series with
each other. FIG. 3 shows a state in which the clamp capacitors C1
and C2 are charged when the regenerative current flows. The clamp
capacitors C1 and C2 divide a voltage between the DC power supply
lines L1 and L2, which reduces breakdown voltages of the clamp
capacitors C1 and C2.
[0070] Further, as described above, the voltages between both ends
of the clamp capacitors C1 and C2 rise to exceed the lower
potential of the square-wave-shape voltage, the clamp capacitors C1
and C2 are discharged on, for example, the motor 5 side. In this
case, the clamp capacitors C1 and C2 are discharged in the state of
being connected in parallel with each other by rectifying functions
of the diodes D1 to D3.
[0071] Hereinafter, specific description will be given of a case
where the operation of the voltage-source inverter 4 is stopped for
protecting the motor 5 from overload to stop current supply to the
motor 5 when, for example, the operating current to be supplied to
the motor 5 exceeds a predetermined value.
[0072] As a specific operation example, a case where the power
supply voltage Vs of the power supply E1 is 400 V, the maximum
value Imax of the regenerative current is 40 A, the resistance
value r1 of the discharge resistor R1 is 15.OMEGA., and the
reference voltage Vref-h is 400 V will be described. Note that
those satisfy Expression (8). FIG. 4 shows a state in which the
clamp capacitors C1 and C2 are discharged. FIG. 5 shows currents
flowing through the coils Lu, Lv and Lw, currents flowing through
the DC power supply lines L1 and L2, a voltage between both ends of
the clamp capacitor C2, a sum of voltages between both ends of the
clamp capacitors C1 and C2, a voltage between the DC power supply
lines L1 and L2, and a current flowing through the discharge
resistor R1 (transistor S1).
[0073] For example, in a case where supply from the power supply E1
is stopped for stopping current supply to the motor 5 (see time 70
ms of FIG. 5), the regenerative current from the motor 5 flows
through the DC power supply lines L1 and L2 (see FIG. 5), and the
regenerative current is supplied to the clamp capacitors C1 and C2.
In this case, the clamp capacitors C1 and C2 are charged in the
state of being connected in series with each other, whereby the
voltages between both ends of the clamp capacitors C1 and C2 rise
(see FIG. 3 and FIG. 5).
[0074] Then, the transistor S1 is brought into conduction when, for
example, the voltage Vc1 between both ends of the clamp capacitor
C2 exceeds the reference voltage Vref-h. Note that the reference
voltage Vref-h (400 V) is set to a value larger than the voltage
Vc1 (approximately 350 V) between both ends of the clamp capacitor
C2 in driving the motor 5. If the transistor S1 is brought into
conduction, the clamp capacitors C1 and C2 are not discharged,
whereby all of the regenerative current flows through the brake
circuit 3. Specific description thereof will be given below. Note
that description will be given regardless of voltage drop of the
transistor S1 for the sake of simplicity.
[0075] A voltage drop Vr1 of the discharge resistor R1, which is
caused when the regenerative current flows through the discharge
resistor R1, is obtained by multiplying the resistance value r1 of
the discharge resistor R1 by the regenerative current. Assuming
that the regenerative current is almost the same as Imax at the
time when the transistor S1 is brought into conduction first,
Vr1=r1Imax=600 V.
[0076] On the other hand, the voltages between both ends of the
clamp capacitors C1 and C2 are each 400 V (equal to the reference
voltage Vref-h). The sum of the voltages between both ends of the
clamp capacitors C1 and C2 is 800 V, and the voltage drop Vr1 in
the case where all of the regenerative current flows through the
discharge resistor R1 is 600 V, and thus the relationship between
the regenerative current and the discharge resistor R1 becomes
dominant. More specifically, the regenerative current flowing
through the discharge resistor R1 does not flow through the clamp
capacitors C1 and C2 but flows into the discharge resistor R1. In
other words, the resistance value r1 satisfies Expression (8),
whereby it is possible to prevent the clamp capacitors C1 and C2 to
be charged with the regenerative current.
[0077] The clamp capacitors C1 and C2 are discharged in the state
of being connected in parallel with each other, and in this case,
the voltages between both ends (=reference voltage) of the clamp
capacitors C1 and C2 are smaller than the voltage drop Vr1 of the
discharge resistor R1. Accordingly, the clamp capacitors C1 and C2
are not discharged.
[0078] Then, the voltage drop Vr1 decreases along with a decrease
in regenerative current (see the voltage between the DC power
supply lines L1 and L2 of FIG. 5), and when the voltage drop Vr1
falls below the voltages between both ends of the clamp capacitors
C1 and C2, discharging of the clamp capacitors C1 and C2 to the
discharge resistor R1 is started (see FIG. 4 and FIG. 5).
[0079] After that, the transistor S1 is brought into non-conduction
when the voltage Vc1 between both ends of the clamp capacitor C2
falls below the reference voltage Vref-L, and the regenerative
current flows through the clamp capacitors C1 and C2, whereby these
are charged. Then, the transistor S1 is brought into conduction
when the voltage Vc1 between both ends of the clamp capacitor C2
again exceeds the reference voltage Vref-h, whereby the clamp
capacitors C1 and C2 are discharged.
[0080] As described above, it is possible to consume regenerative
energy due to the regenerative current while preventing the
voltages between both ends of the clamp capacitors C1 and C2 from
rising due to the regenerative current.
[0081] Further, in a mode in which one clamp capacitor is provided
between DC power supply lines, a voltage same as the voltage
between the DC power supply lines L1 and L2 (=voltage between both
ends of one clamp capacitor, which is 800 V under the
above-mentioned conditions) is applied to the discharge resistor R1
when the transistor S1 is brought into conduction. On the other
hand, in this motor driving device, the voltage drop Vr1 of the
discharge resistor R1 is lower than this voltage (for example, 800
V) as described above. Accordingly, it is possible to reduce the
power capacity required by the discharge resistor R1 with the same
resistance value.
Second Embodiment
[0082] A conceptual configuration diagram of a motor driving device
according to a second embodiment of the present invention is the
same as that of FIG. 1. In this motor driving device, the
transistor S1 is in conduction during a period of time in which the
voltage Vc1 between both ends of the clamp capacitor C2 exceeds the
reference voltage Vref-h. Accordingly, the longer this period of
time is, the larger time rating the transistor S1 requires.
Therefore, in the motor driving device according to the second
embodiment, the period of time in which the current keeps flowing
through the transistor S1 is reduced, whereby the time rating
required by the transistor S1 is reduced.
[0083] The resistance value r1 of the discharge resistor R1 is a
value equal to or smaller than a value obtained by dividing the
reference voltage Vref-h by the maximum value Imax of the current
flowing through the discharge resistor R1. That is,
r1.ltoreq.Vref-h/Imax (hereinafter, referred to as Expression (9))
is satisfied.
[0084] FIG. 6 shows the currents flowing through the coils Lu, Lv
and Lw, the current flowing through the DC power supply lines L1
and L2, the voltage between both ends of the clamp capacitor C2,
the sum of the voltages between both ends of the clamp capacitors
C1 and C2, the voltage between the DC power supply lines L1 and L2
and the current flowing through the discharge resistor R1
(transistor S1) when, for example, the operation of the
voltage-source inverter 4 is stopped for stopping current supply to
the motor 5.
[0085] Note that FIG. 6 shows the results in a case where the power
supply voltage Vs of the power supply E1 is 400 V, the maximum
value Imax of the regenerative current is 40 A, the resistance
value r1 of the discharge resistor R1 is 10.OMEGA., and the
reference voltage Vref-h is 400 V, which satisfy Expression
(9).
[0086] Description will be given in comparison with FIG. 5. In FIG.
5, the resistance value r1 is 15.OMEGA. and the maximum value Imax
of the regenerative current is 40 A, and thus the voltage drop Vr1
of the discharge resistor R1 when the transistor S1 is brought into
conduction first is 600 V, which is larger than 400 V (voltages
between both ends of the clamp capacitors C1 and C2) of the
reference voltage Vref-h. Therefore, the clamp capacitors C1 and C2
are not discharged until the voltage drop Vr1 falls below the
voltages between both ends of the clamp capacitors C1 and C2,
whereby the transistor S1 is in conduction for a long period of
time.
[0087] In the second embodiment, the voltage drop Vr1 of the
discharge resistor R1, which results from the regenerative current
at the time when the transistor S1 is brought into conduction
first, is 400 V (=10.OMEGA..times.40 A), which is the same as the
voltages between both ends of the clamp capacitors C1 and C2. The
voltage drop Vr1 decreases along with a decrease in regenerative
current, and thus the voltage drop Vr1 falls below the voltages
between both ends of the clamp capacitors C1 and C2 immediately
after the transistor S1 is brought into conduction first.
Accordingly, discharging of the clamp capacitors C1 and C2 to the
discharge resistor R1 is started. After that, as in the first
embodiment, the transistor S1 repeats conduction and non-conduction
based on, for example, the voltage Vc1 between both ends of the
clamp capacitor C2.
[0088] As described above, the resistance value r1 of the discharge
resistor R1 is equal to or smaller than the value obtained by
dividing the reference voltage Vref-h by the maximum value
T.sub.max of the regenerative current, and thus discharging of the
clamp capacitors C1 and C2 is started immediately after the
conduction of the transistor S1, which reduces the period of time
in which the transistor S1 is in conduction.
[0089] Further, the voltage drop Vr1 of the discharge resistor R1
is equal to or smaller than the reference voltage Vref-h, and thus
the power capacity required by the discharge resistor R1 can be
reduced further.
[0090] FIG. 7 shows the results of a case where the power supply
voltage Vs of the power supply E1 is 400 V, the maximum value Imax
of the regenerative current is 40 A, the resistance value r1 of the
discharge resistor R1 is 5.OMEGA., and the reference voltage Vref-h
is 400 V, which satisfy Expression (9).
[0091] For example, at time 70 ms, the regenerative current from
the motor 5 is supplied to the clamp capacitors C1 and C2, whereby
the voltages between both ends of the clamp capacitors C1 and C2
rise (see FIG. 7). Then, for example, the voltage Vc1 between both
ends of the clamp capacitor C2 exceeds the reference voltage
Vref-h, and thus the transistor S1 is brought into conduction.
[0092] On this occasion, the voltage drop Vr1 of the discharge
resistor R1, which results from only the regenerative current Imax,
is 200 V (=5.OMEGA..times.40 A), and thus the voltage drop Vr1 is
smaller than the voltage Vc1 (=reference voltage Vref-h=400 V)
between both ends of the clamp capacitor C2. In this case, a value
of the current flowing through the discharge resistor R1 is
determined from the relationship between the voltages between both
ends of the clamp capacitors C1 and C2 and the resistance value r1.
In other words, discharging of the clamp capacitors C1 and C2 to
the discharge resistor R1 is started. Upon discharging of the clamp
capacitors C1 and C2, the regenerative current and the discharging
currents from the clamp capacitors C1 and C2 flow through the
discharge resistor R1. Note that the current flowing through the
discharge resistor R1 on this occasion has a obtained by dividing
the voltages between both ends of the clamp capacitors C1 and C2 by
the resistance value R1 (=reference voltage Vref-h/resistance value
r1).
[0093] After that, the voltages between both ends of the clamp
capacitors C1 and C2 decrease to fall below the reference voltage
Vref-L, whereby the transistor S1 is brought into non-conduction.
Then, the regenerative current flows through the clamp capacitors
C1 and C2 to charge them, and the transistor S1 is brought into
conduction when the voltage Vc1 between both ends of the clamp
capacitor C2 again exceeds the reference voltage Vref-h, with the
result that the clamp capacitors C1 and C2 are discharged.
[0094] As described above, the clamp capacitors C1 and C2 can be
discharged almost at the same time with the conduction of the
transistor S1, and thus the period of time in which the transistor
S1 is in conduction can be reduced, which reduces the time rating
of the transistor S1.
[0095] Note that the voltage (voltage drop Vr1) applied to the
discharge resistor R1 is constant, which is almost the same as the
voltages between both ends (reference voltage Vref-h) of the clamp
capacitors C1 and C2, and hence larger loss is generated in the
discharge resistor R1 as the resistance value r1 decreases.
Therefore, the resistance value r1 is desirably as large as
possible. That is, the resistance value r1 is desirably a value
obtained by dividing the reference voltage Vref-h by the maximum
value of the current flowing through the discharge resistor R1.
Third Embodiment
[0096] A conceptual configuration diagram of a motor driving device
according to a third embodiment is the same as that of FIG. 1
except for the clamp circuit 2. FIG. 8 is a conceptual
configuration diagram of the clamp circuit 2 included in the motor
driving device according to the third embodiment.
[0097] Compared with the clamp circuit 2 shown in FIG. 1, the clamp
circuit 2 further includes a clamp capacitor C3 and diodes D6 to
D8. The clamp capacitor C3 is connected between the diode D1 and
the clamp capacitor C2. Between the clamp capacitors C2 and C3, the
diode D6 has an anode connected to the clamp capacitor C3 and a
cathode connected to the clamp capacitor C2 and the diode D2. The
diode D7 has an anode connected to a point between the diode D1 and
the clamp capacitor C3 and a cathode connected to the DC power
supply line L1. A diode D8 has an anode connected to the DC power
supply line L2 and a cathode connected to a point between the diode
D6 and the clamp capacitor C3.
[0098] According to the clamp circuit 2 having the above-mentioned
configuration, the clamp capacitors C1 to C3 are charged in the
state of being connected in series with each other and discharged
in the state of being connected in parallel with each other by the
rectifying functions of the diodes D1 to D3 and D6 to D8.
[0099] As a result, the voltage between the DC power supply lines
L1 and L2 is divided by the clamp capacitors C1 to C3, with the
result that the voltages applied to the clamp capacitors C1 to C3
are reduced further. Accordingly, the reference voltage Vref-h can
also be reduced.
[0100] Further, the resistance value r1 of the discharge resistor
R1 is set to a value equal to or smaller than the value obtained by
dividing the reference voltage Vref-h by the maximum value Imax of
the current flowing through the discharge resistor R1, whereby the
electrostatic capacitance (=reference voltage
Vref-h.times.reference voltage Vref-h/resistance value r1) required
by the discharge resistor R1 can be reduced further. This is
because the reference voltage Vref-h can be reduced further.
[0101] Note that in the motor driving devices described in the
first to third embodiments, the brake circuit 3 and the
voltage-source inverter 4 may be composed of a power integrated
module (PIM). In this case, those can be manufactured integrally
and are widely applied to an indirect AC power converting
apparatus, whereby a motor driving device can be configured to be
compact in size at inexpensive cost.
[0102] While the invention has been shown and described in detail,
the foregoing description is in all aspects illustrative and not
restrictive. It is therefore understood that numerous modifications
and variations can be devised without departing from the scope of
the invention.
* * * * *