U.S. patent application number 12/688305 was filed with the patent office on 2010-05-13 for wireless energy transfer using coupled resonators.
Invention is credited to John D. Joannopoulos, Aristeidis Karalis, Marin Soljacic.
Application Number | 20100117455 12/688305 |
Document ID | / |
Family ID | 37637764 |
Filed Date | 2010-05-13 |
United States Patent
Application |
20100117455 |
Kind Code |
A1 |
Joannopoulos; John D. ; et
al. |
May 13, 2010 |
WIRELESS ENERGY TRANSFER USING COUPLED RESONATORS
Abstract
Described herein are embodiments of transmitting power
wirelessly that includes driving a high-Q non-radiative resonator
at a value near its resonant frequency to produce a magnetic field
output, said non-radiative-resonator formed of a combination of
resonant parts, including at least an inductive part formed by a
wire loop, and a capacitor part that is separate from a material
forming the inductive part, and maintaining at least one
characteristic of said resonator such that its usable range has a
usable distance over which power can be received, which distance is
set by a detuning effect when a metallic structure gets too close
to said resonator.
Inventors: |
Joannopoulos; John D.;
(Belmont, MA) ; Karalis; Aristeidis; (Boston,
MA) ; Soljacic; Marin; (Belmont, MA) |
Correspondence
Address: |
GTC Law Group LLP & Affiliates
P.O. Box 113237
Pittsburgh
PA
15241
US
|
Family ID: |
37637764 |
Appl. No.: |
12/688305 |
Filed: |
January 15, 2010 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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12553957 |
Sep 3, 2009 |
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12688305 |
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11481077 |
Jul 5, 2006 |
7687939 |
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12553957 |
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60698442 |
Jul 12, 2005 |
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Current U.S.
Class: |
307/104 |
Current CPC
Class: |
H02J 50/12 20160201;
B60L 53/126 20190201; H02J 5/005 20130101; Y02T 90/14 20130101;
Y02T 10/70 20130101; Y02T 10/7072 20130101; Y02T 90/12 20130101;
H01F 38/14 20130101; H01Q 9/04 20130101 |
Class at
Publication: |
307/104 |
International
Class: |
H02J 17/00 20060101
H02J017/00 |
Goverment Interests
STATEMENT REGARDING GOVERNMENT FUNDING
[0002] This invention was made, in whole or in part by grant
DMR-0213282 from the National Science Foundation. Accordingly, the
Government may have certain rights in the invention.
Claims
1. A method of transmitting power wirelessly, comprising: driving a
high-Q non-radiative resonator at a value near its resonant
frequency to produce a magnetic field output, said
non-radiative-resonator formed of a combination of resonant parts,
including at least an inductive part formed by a wire loop, and a
capacitor part that is separate from a material forming the
inductive part; and maintaining at least one characteristic of said
resonator such that its usable range has a usable distance over
which power can be received, which distance is set by a detuning
effect when a metallic structure gets too close to said
resonator.
2. A method as in claim 1, further comprising receiving the
magnetic field in a second resonator, said second resonator also
having a series resonant part, and producing usable power from said
second resonator, and coupling said usable power to a load.
3. A method as in claim 1, wherein said useable distance is between
6 and 8 inches.
4. A method as in claim 1, wherein said usable distance is between
2 and 4 inches.
5. A method as in claim 1, further comprising setting the resonant
frequency to a value of approximately 13.56 MHz.
6. A method as in claim 1, wherein said non-radiative resonator has
a Q value of approximately 1400.
7. A method as in claim 1, further comprising using a signal
generator to produce said driving, and matching a characteristic of
the signal generator to a characteristic of the resonator at
resonance.
8. A method as in claim 1, wherein said resonator has a
substantially round outer form factor.
9. A method as in claim 1, wherein said resonator is a dipole that
includes a first high-Q inductive part, coupled to receive said
driving, and a second high-Q part, which is physically separated
from said inductive part, said second high-Q part formed of at
least one loop of wire in series with said capacitor.
10. A method as in claim 9, wherein said high-Q inductive part and
said second high-Q part have different outer form factors.
11. A method as in claim 9, wherein said high-Q inductive part and
said second high-Q part have substantially the same outer form
factors.
12. A method as in claim 9, wherein said second high-Q part is
electrically connected to said capacitor part.
13. A method as in claim 12, wherein said capacitor part is a
variable capacitor.
14. A method as in claim 1, wherein said effect is a detuning of
said resonant frequency of said resonator.
15. A method, comprising: generating a magnetic field using a first
high-Q resonator; receiving said magnetic field in a second high-Q
resonator; and in said second high-Q resonator, using power from
the magnetic field.
16. A method as in claim 15, further comprising using said power in
said second high-Q resonator to drive a load.
17. A method as in claim 15, wherein said generating comprises
using a capacitively loaded resonator for generating the magnetic
field.
18. A method, comprising: forming a magnetic field using a first
high-Q part; coupling said magnetic field to a second high-Q part
using near-field coupling with said first part, where said second
part is further than 6 inches from said first part; and in said
second part, recovering power from the coupled magnetic field.
19. A method as in claim 18, further comprising using said power in
said high-Q second part to drive a load.
20. A method as in claim 18, wherein said forming comprises using a
capacitively loaded resonator for forming the magnetic field.
21. A system comprising: a high-Q resonator; a driving part for
said resonator, driving said resonator at a value near its resonant
frequency to produce a magnetic field output, said resonator formed
of a combination of series resonant parts, including at least an
inductive part formed by a wire loop, and a capacitor part that is
separate from a material forming the inductive part, and wherein
said resonator has at least one characteristic such that its usable
range has a usable distance over which power can be received, which
distance is set by a detuning effect when a metallic structure gets
too close to said resonator.
22. A system as in claim 21, further comprising a second resonator,
also having a series resonant part that has a corresponding
resonance to said resonant frequency of said resonator, and has a
connection outputting usable power to a load.
23. A system as in claim 22, wherein said useable distance is
between 6 and 8 inches.
24. A system as in claim 21, wherein said usable distance is
between 2 and 4 inches.
25. A system as in claim 21, wherein the resonant frequency is set
to a value of approximately 13.56 MHz.
26. A system as in claim 21, wherein said resonator has a Q value
of approximately 1400.
27. A system as in claim 21, further comprising signal generator
that drives said resonator, said signal generator having a
characteristic of the signal generator that is matched to a
characteristic of the resonator at resonance.
28. A system as in claim 21, wherein said resonator has a
substantially round outer form factor.
29. A system as in claim 21, wherein said resonator includes a
first high-Q inductive part, coupled to receive said driving, and a
second high-Q part, which is physically separated from said
inductive part, said second part formed of at least one loop of
plural coils of wire in series with said capacitor.
30. A system as in claim 29, wherein said high-Q inductive part and
said second high-Q part have different outer form factors.
31. A system as in claim 29, wherein said high-Q inductive part and
said second high-Q part have substantially the same outer form
factors.
32. A system as in claim 29, wherein said second high-Q part is
electrically connected to said capacitor part.
33. A system as in claim 32, wherein said capacitor part is a
variable capacitor.
34. A system, comprising: a first high-Q resonator part formed of a
capacitively loaded resonator; a second high-Q resonator part,
tuned to have similar resonant characteristics to said first
resonator part, receiving a magnetic field therefrom, and producing
a power output from the magnetic field.
35. A system, comprising: a first high-Q LC circuit, connected to
receive a signal that forms a magnetic field; and a second high-Q
LC circuit that forms near-field coupling with said first high-Q LC
circuit, where said second part is further than 6 inches from said
first part, and has a connection for recovering power from the
coupled magnetic field.
36. A system as in claim 35, wherein said first LC circuit includes
a capacitively loaded resonator for generating the magnetic
field.
37. A method of transmitting power wirelessly, comprising: driving
a high-Q resonator at a value near a resonant frequency of said
resonator to produce a magnetic field output, said resonator
defining a distance between a metallic structure and said
resonator, and formed of a combination of resonant parts, including
at least an inductive part formed by a wire loop, and a capacitor
part that is separate from a material forming the inductive part;
and maintaining at least one characteristic of said resonator such
that its usable range has a usable distance over which power can be
received, which distance is set by an effect when a metallic
structure gets too close to said resonator.
38. A method as in claim 37, further comprising receiving the
magnetic field in a second resonator, said second resonator
producing usable power from said second resonator, and coupling
said usable power to a load.
39. A method as in claim 37, wherein said useable distance is
between 6 and 8 inches.
40. A method as in claim 37, wherein said usable distance is
between 2 and 4 inches.
41. A method as in claim 37, further comprising setting the
resonant frequency to a value of approximately 13.56 MHz.
42. A method as in claim 37, wherein said resonator has a Q value
of approximately 1400.
43. A method as in claim 37, wherein said resonator has a
substantially round outer form factor.
44. A method as in claim 37, wherein said resonator includes a
first high-Q inductive part, coupled to receive said driving, and a
second high-Q part, which is physically separated from said high-Q
inductive part, said second high-Q part formed of at least one loop
of wire in series with said capacitor.
45. A method as in claim 44, wherein said high-Q inductive part and
said second high-Q part have different outer form factors.
46. A method as in claim 44, wherein said high-Q inductive part and
said second high-Q part have substantially the same outer form
factors.
47. A method as in claim 44, wherein said second high-Q part is
electrically connected to said capacitor part.
48. A method as in claim 47, wherein said capacitor part is a
variable capacitor.
49. A method, comprising: generating a magnetic field using a first
high-Q resonator; receiving said magnetic field in a second high-Q
resonator; and in said second high-Q resonator, using power from
the magnetic field.
50. A method as in claim 49, further comprising using said power in
said second high-Q resonator to drive a load.
51. A method as in claim 49, wherein said generating comprises
using a capacitively loaded resonator for generating the magnetic
field.
52. A method, comprising: forming a magnetic field using a high-Q
first part; coupling said magnetic field to a second high-Q part
using near-field coupling with said first part, where said second
part is further than 6 inches from said first part; and in said
second part, recovering power from the coupled magnetic field.
53. A method as in claim 52, further comprising using said power in
said high-Q second part to drive a load.
54. A method as in claim 52, wherein said forming comprises using a
capacitively loaded resonator for forming the magnetic field.
55. A system comprising: a high-Q resonator; a driving part for
said resonator, driving said resonator at a value near a resonant
frequency of said resonator to produce a magnetic field output,
said resonator formed of a combination of parts, including at least
an inductive part formed by a wire loop, and a capacitor part that
is separate from a material forming the inductive part, and wherein
said resonator has at least one characteristic such that its usable
range has a usable distance over which power can be received, which
distance is set by a detuning effect that changes said resonant
frequency when a metallic structure gets too close to said
resonator.
56. A system as in claim 55, further comprising a second resonator,
also having part that has a corresponding resonance to said
resonant frequency of said resonator, and has a connection
outputting usable power to a load.
57. A system as in claim 56, wherein said useable distance is
between 6 and 8 inches.
58. A system as in claim 55, wherein said usable distance is
between 2 and 4 inches.
59. A system as in claim 55, wherein the resonant frequency is set
to a value of approximately 13.56 MHz.
60. A system as in claim 55, wherein said resonator has a Q value
of approximately 1400.
61. A system as in claim 55, wherein said resonator has a
substantially round outer form factor.
62. A system as in claim 55, wherein said resonator includes a
first high-Q inductive part, coupled to receive said driving, and a
second high-Q part, which is physically separated from said first
high-Q inductive part, said second high-Q part, formed of at least
one loop of plural coils of wire in series with said capacitor.
63. A system as in claim 62, wherein said first high-Q inductive
part and said second high-Q part have different outer form
factors.
64. A system as in claim 62, wherein said first high-Q inductive
part and said second high-Q part have substantially the same outer
form factors.
65. A system as in claim 62, wherein said second high-Q part is
electrically connected to said capacitor part.
66. A system as in claim 65, wherein said capacitor part is a
variable capacitor.
67. A system, comprising: a first high-Q resonator part formed of a
capacitively loaded resonator; and a second high-Q resonator part,
tuned to have similar resonant characteristics to said first
resonator part, receiving a magnetic field therefrom, and producing
a power output from the magnetic field.
68. A system, comprising: a first high-Q LC circuit, connected to
receive a signal that forms a magnetic field; and a second high-Q
LC circuit that forms near-field coupling with said first high-Q LC
circuit, where said second part is further than 6 inches from said
first part, and has a connection for recovering power from the
coupled magnetic field.
69. A method of wirelessly receiving power, comprising: receiving
power from a high-Q resonator, by interacting with said resonator
at a value near its resonant frequency to produce a power output
from a received magnetic field, said receiving comprising
connecting a receiving circuit directly to a first high-Q
resonator, and also receiving wireless power in a second high-Q
resonator loop formed of a combination of resonant parts, including
at least an inductive part and a capacitance, where said second
resonator loop is physically separated from said first high-Q
resonator.
70. A method as in claim 69, wherein said resonator has at least
one characteristic such that its usable range has a usable distance
over which power can be received, which distance is set by a
detuning effect that changes a resonant frequency when a metallic
structure gets too close to said resonator.
71. A method as in claim 70, wherein said useable distance is
between 6 and 8 inches.
72. A method as in claim 70, wherein said usable distance is
between 2 and 4 inches.
73. A method as in claim 69, wherein the resonant frequency is set
to a value of approximately 13.56 MHz.
74. A method of wirelessly transmitting power, comprising:
producing a magnetic power signal at a first frequency; coupling
said power signal to a first high-Q inductive loop by connecting to
said first inductive loop; and inducing said power signal into a
second high-Q inductive loop from the first inductive loop, said
second inductive loop having a resonant frequency substantially
matched to said first frequency.
75. A method as in claim 74, where said second high-Q inductive
loop is physically separated from said first high-Q inductive
loop.
76. A method of claim 74, wherein said high-Q inductive loop has at
least one characteristic such that its usable range has a usable
distance over which power can be received, which distance is set by
a detuning effect when a metallic structure gets too close to said
high-Q inductive loop.
77. A method as in claim 76, wherein said useable distance is
between 6 and 8 inches.
78. A method as in claim 76, wherein said useable distance is
between 2 and 4 inches.
79. A method as in claim 37, wherein said effect is a detuning of
said resonant frequency of said resonator.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of co-pending United
States Patent Application entitled WIRELESS NON-RADIATIVE ENERGY
TRANSFER filed on Sep. 3, 2009 having Ser. No. 12/553,957 ('957
application), the entirety of which is incorporated herein by
reference. The '957 application is a continuation of co-pending
U.S. patent application entitled WIRELESS NON-RADIATIVE ENERGY
TRANSFER filed on Jul. 5, 2006 and having Ser. No. 11/481,077 ('077
application), the entirety of which is incorporated herein by
reference. The '077 Application claims the benefit of provisional
application Ser. No. 60/698,442 filed Jul. 12, 2005 ('442
application), the entirety of which is incorporated herein by
reference.
BACKGROUND OF THE INVENTION
[0003] The invention relates to the field of oscillatory resonant
electromagnetic modes, and in particular to oscillatory resonant
electromagnetic modes, with localized slowly evanescent field
patterns, for wireless non-radiative energy transfer.
[0004] In the early days of electromagnetism, before the
electrical-wire grid was deployed, serious interest and effort was
devoted towards the development of schemes to transport energy over
long distances wirelessly, without any carrier medium. These
efforts appear to have met with little, if any, success. Radiative
modes of omni-directional antennas, which work very well for
information transfer, are not suitable for such energy transfer,
because a vast majority of energy is wasted into free space.
Directed radiation modes, using lasers or highly-directional
antennas, can be efficiently used for energy transfer, even for
long distances (transfer distance L.sub.TRANS<<L.sub.DEV,
where L.sub.DEV is the characteristic size of the device), but
require existence of an uninterruptible line-of-sight and a
complicated tracking system in the case of mobile objects.
[0005] Rapid development of autonomous electronics of recent years
(e.g. laptops, cell-phones, house-hold robots, that all typically
rely on chemical energy storage) justifies revisiting investigation
of this issue. Today, the existing electrical-wire grid carries
energy almost everywhere; even a medium-range wireless
non-radiative energy transfer would be quite useful. One scheme
currently used for some important applications relies on induction,
but it is restricted to very close-range
(L.sub.TRANS<<L.sub.DEV) energy transfers.
SUMMARY OF THE INVENTION
[0006] According to one aspect of the invention, there is provided
an electromagnetic energy transfer device. The electromagnetic
energy transfer device includes a first resonator structure
receiving energy from an external power supply. The first resonator
structure has a first Q-factor. A second resonator structure is
positioned distal from the first resonator structure, and supplies
useful working power to an external load. The second resonator
structure has a second Q-factor. The distance between the two
resonators can be larger than the characteristic size of each
resonator. Non-radiative energy transfer between the first
resonator structure and the second resonator structure is mediated
through coupling of their resonant-field evanescent tails.
[0007] According to another aspect of the invention, there is
provided a method of transferring electromagnetic energy. The
method includes providing a first resonator structure receiving
energy from an external power supply. The first resonator structure
has a first Q-factor. Also, the method includes a second resonator
structure being positioned distal from the first resonator
structure, and supplying useful working power to an external load.
The second resonator structure has a second Q-factor. The distance
between the two resonators can be larger than the characteristic
size of each resonator. Furthermore, the method includes
transferring non-radiative energy between the first resonator
structure and the second resonator structure through coupling of
their resonant-field evanescent tails.
[0008] In another aspect, a method of transferring energy is
disclosed including the steps of providing a first resonator
structure receiving energy from an external power supply, said
first resonator structure having a first resonant frequency
.omega..sub.1, and a first Q-factor Q.sub.1, and characteristic
size L.sub.1. Providing a second resonator structure being
positioned distal from said first resonator structure, at closest
distance D, said second resonator structure having a second
resonant frequency .omega..sub.2, and a second Q-factor Q.sub.2,
and characteristic size L.sub.2, where the two said frequencies
.omega..sub.1 and .omega..sub.2 are close to within the narrower of
the two resonance widths .GAMMA..sub.1, and .GAMMA..sub.2, and
transferring energy non-radiatively between said first resonator
structure and said second resonator structure, said energy transfer
being mediated through coupling of their resonant-field evanescent
tails, and the rate of energy transfer between said first resonator
and said second resonator being denoted by .kappa., where
non-radiative means D is smaller than each of the resonant
wavelengths .lamda..sub.1 and .lamda..sub.2, where c is the
propagation speed of radiation in the surrounding medium.
[0009] Embodiments of the method may include any of the following
features. In some embodiments, said resonators have Q.sub.1>100
and Q.sub.2>100, Q.sub.1>200 and Q.sub.2>200,
Q.sub.1>500 and Q.sub.2>500, or even Q.sub.1>1000 and
Q.sub.2>1000. In some such embodiments,
.kappa./sqrt(.GAMMA..sub.1*.GAMMA..sub.2) may be greater than 0.2,
greater than 0.5, greater than 1, greater than 2, or even grater
than 5. In some such embodiments D/L.sub.2 may be greater than 1,
greater than 2, greater than 3, greater than 5.
[0010] In another aspect, an energy transfer device is disclosed
which includes: a first resonator structure receiving energy from
an external power supply, said first resonator structure having a
first resonant frequency .omega..sub.1, and a first Q-factor
Q.sub.1, and characteristic size L.sub.1, and a second resonator
structure being positioned distal from said first resonator
structure, at closest distance D, said second resonator structure
having a second resonant frequency .omega..sub.2, and a second
Q-factor Q.sub.2, and characteristic size L.sub.2.
[0011] The two said frequencies .omega..sub.1 and .omega..sub.2 are
close to within the narrower of the two resonance widths
.GAMMA..sub.1, and .GAMMA..sub.2. The non-radiative energy transfer
between said first resonator structure and said second resonator
structure is mediated through coupling of their resonant-field
evanescent tails, and the rate of energy transfer between said
first resonator and said second resonator is denoted by .kappa..
The non-radiative means D is smaller than each of the resonant
wavelengths .lamda..sub.1 and .lamda..sub.2, where c is the
propagation speed of radiation in the surrounding medium.
[0012] Embodiments of the device may include any of the following
features. In some embodiments, said resonators have Q.sub.1>100
and Q.sub.2>100, Q.sub.1>200 and Q.sub.2>200,
Q.sub.1>500 and Q.sub.2>500, or even Q.sub.1>1000 and
Q.sub.2>1000. In some such embodiments,
.kappa./sqrt(.GAMMA..sub.1*.GAMMA..sub.2) may be greater than 0.2,
greater than 0.5, greater than 1, greater than 2, or even grater
than 5. In some such embodiments D/L.sub.2 may be greater than 1,
greater than 2, greater than 3, or even greater than 5.
[0013] In some embodiments, the resonant field in the device is
electromagnetic.
[0014] In some embodiments, the first resonator structure includes
a dielectric sphere, where the characteristic size L1 is the radius
of the sphere.
[0015] In some embodiments, the first resonator structure includes
a metallic sphere, where the characteristic size L1 is the radius
of the sphere.
[0016] In some embodiments, the first resonator structure includes
a metallodielectric sphere, where the characteristic size L1 is the
radius of the sphere.
[0017] In some embodiments, the first resonator structure includes
a plasmonic sphere, where the characteristic size L1 is the radius
of the sphere.
[0018] In some embodiments, the first resonator structure includes
a polaritonic sphere, where the characteristic size L1 is the
radius of the sphere.
[0019] In some embodiments, the first resonator structure includes
a capacitively-loaded conducting-wire loop, where the
characteristic size L1 is the radius of the loop.
[0020] In some embodiments, the second resonator structure includes
a dielectric sphere, where the characteristic size L2 is the radius
of the sphere.
[0021] In some embodiments, the second resonator structure includes
a metallic sphere where the characteristic size L2 is the radius of
the sphere.
[0022] In some embodiments, the second resonator structure includes
a metallodielectric sphere where the characteristic size L2 is the
radius of the sphere.
[0023] In some embodiments, the second resonator structure includes
a plasmonic sphere where the characteristic size L2 is the radius
of the sphere.
[0024] In some embodiments, the second resonator structure includes
a polaritonic sphere where the characteristic size L2 is the radius
of the sphere.
[0025] In some embodiments, the second resonator structure includes
a capacitively-loaded conducting-wire loop where the characteristic
size L2 is the radius of the loop.
[0026] In some embodiments, the resonant field in the device is
acoustic.
[0027] It is to be understood that embodiments of the above
described methods and devices may include any of the above listed
features, alone or in combination.
BRIEF DESCRIPTION OF THE DRAWINGS
[0028] FIG. 1 is a schematic diagram illustrating an exemplary
embodiment of the invention;
[0029] FIG. 2A is a numerical FDTD result for a high-index disk
cavity of radius r along with the electric field; FIG. 2B a
numerical FDTD result for a medium-distance coupling between two
resonant disk cavities: initially, all the energy is in one cavity
(left panel); after some time both cavities are equally excited
(right panel).
[0030] FIG. 3 is schematic diagram demonstrating two
capacitively-loaded conducting-wire loops;
[0031] FIGS. 4A-4B are numerical FDTD results for reduction in
radiation-Q of the resonant disk cavity due to scattering from
extraneous objects;
[0032] FIG. 5 is a numerical FDTD result for medium-distance
coupling between two resonant disk cavities in the presence of
extraneous objects; and
[0033] FIGS. 6A-6B are graphs demonstrating efficiencies of
converting the supplied power into useful work (raw), radiation and
ohmic loss at the device (.eta.d), and the source(.eta.s), and
dissipation inside a human (.eta.h), as a function of the
coupling-to-loss ratio .kappa./.GAMMA.d; in panel (a) .GAMMA.w is
chosen so as to minimize the energy stored in the device, while in
panel (b) .GAMMA.w is chosen so as to maximize the efficiency
.eta.w for each .kappa./.GAMMA.d.
DETAILED DESCRIPTION OF THE INVENTION
[0034] In contrast to the currently existing schemes, the invention
provides the feasibility of using long-lived oscillatory resonant
electromagnetic modes, with localized slowly evanescent field
patterns, for wireless non-radiative energy transfer. The basis of
this technique is that two same-frequency resonant objects tend to
couple, while interacting weakly with other off-resonant
environmental objects. The purpose of the invention is to quantify
this mechanism using specific examples, namely quantitatively
address the following questions: up to which distances can such a
scheme be efficient and how sensitive is it to external
perturbations. Detailed theoretical and numerical analysis show
that a mid-range (L.sub.TRANS.apprxeq.few*L.sub.DEV) wireless
energy-exchange can actually be achieved, while suffering only
modest transfer and dissipation of energy into other off-resonant
objects.
[0035] The omnidirectional but stationary (non-lossy) nature of the
near field makes this mechanism suitable for mobile wireless
receivers. It could therefore have a variety of possible
applications including for example, placing a source connected to
the wired electricity network on the ceiling of a factory room,
while devices, such as robots, vehicles, computers, or similar, are
roaming freely within the room. Other possible applications include
electric-engine buses, RFIDs, and perhaps even nano-robots.
[0036] The range and rate of the inventive wireless energy-transfer
scheme are the first subjects of examination, without considering
yet energy drainage from the system for use into work. An
appropriate analytical framework for modeling the exchange of
energy between resonant objects is a weak-coupling approach called
"coupled-mode theory". FIG. 1 is a schematic diagram illustrating a
general description of the invention. The invention uses a source
and device to perform energy transferring. Both the source 1 and
device 2 are resonator structures, and are separated a distance D
from each other. In this arrangement, the electromagnetic field of
the system of source 1 and device 2 is approximated by
F(r,t).apprxeq.a.sub.1(t)F.sub.1(r)+a.sub.2(t)F.sub.2(r), where
F.sub.1,2(r)=[E.sub.1,2(r) H.sub.1,2(r)] are the eigenmodes of
source 1 and device 2 alone, and then the field amplitudes
a.sub.1(t) and a.sub.2(t) can be shown to satisfy the "coupled-mode
theory":
a 1 t = - ( .omega. 1 - .GAMMA. 1 ) a 1 + .kappa. 11 a 1 + .kappa.
12 a 2 a 2 t = - ( .omega. 2 - .GAMMA. 2 ) a 2 + .kappa. 22 a 2 +
.kappa. 21 a 1 , ( 1 ) ##EQU00001##
where .omega..sub.1,2 are the individual eigen-frequencies,
.GAMMA..sub.1,2 are the resonance widths due to the objects'
intrinsic (absorption, radiation etc.) losses, .kappa..sub.12,21
are the coupling coefficients, and .kappa..sub.11,22 model the
shift in the complex frequency of each object due to the presence
of the other.
[0037] The approach of Eq. 1 has been shown, on numerous occasions,
to provide an excellent description of resonant phenomena for
objects of similar complex eigen-frequencies (namely
|.omega..sub.1-.omega..sub.2|<<|.kappa..sub.12,21| and
.GAMMA..sub.1.apprxeq..GAMMA..sub.2), whose resonances are
reasonably well defined (namely
.GAMMA..sub.1,2&Im{.kappa..sub.11,22}<<|.kappa..sub.12,21|)
and in the weak coupling limit (namely
|.kappa..sub.12,21|<<.omega..sub.1,2). Coincidentally, these
requirements also enable optimal operation for energy transfer.
Also, Eq. (1) show that the energy exchange can be nearly perfect
at exact resonance ((.omega..sub.1=.omega..sub.2 and
.GAMMA..sub.1=.GAMMA..sub.2), and that the losses are minimal when
the "coupling-time" is much shorter than all "loss-times".
Therefore, the invention requires resonant modes of high
Q=.omega./2.GAMMA.) for low intrinsic-loss rates .GAMMA..sub.1,2,
and with evanescent tails significantly longer than the
characteristic sizes L.sub.1 and L.sub.2 of the two objects for
strong coupling rate |.kappa..sub.12,21| over large distances D,
where D is the closest distance between the two objects. This is a
regime of operation that has not been studied extensively, since
one usually prefers short tails, to minimize interference with
nearby devices.
[0038] Objects of nearly infinite extent, such as dielectric
waveguides, can support guided modes whose evanescent tails are
decaying exponentially in the direction away from the object,
slowly if tuned close to cutoff, and can have nearly infinite Q. To
implement the inventive energy-transfer scheme, such geometries
might be suitable for certain applications, but usually finite
objects, namely ones that are topologically surrounded everywhere
by air, are more appropriate.
[0039] Unfortunately, objects of finite extent cannot support
electromagnetic states that are exponentially decaying in all
directions in air, since in free space: {right arrow over
(k)}.sup.2.omega..sup.2/c.sup.2. Because of this, one can show that
they cannot support states of infinite Q. However, very long-lived
(so-called "high-Q") states can be found, whose tails display the
needed exponential-like decay away from the resonant object over
long enough distances before they turn oscillatory (radiative). The
limiting surface, where this change in the field behavior happens,
is called the "radiation caustic", and, for the wireless
energy-transfer scheme to be based on the near field rather than
the far/radiation field, the distance between the coupled objects
must be such that one lies within the radiation caustic of the
other.
[0040] The invention is very general and any type of resonant
structure satisfying the above requirements can be used for its
implementation. As examples and for definiteness, one can choose to
work with two well-known, but quite different electromagnetic
resonant systems: dielectric disks and capacitively-loaded
conducting-wire loops. Even without optimization, and despite their
simplicity, both will be shown to exhibit fairly good performance.
Their difference lies mostly in the frequency range of
applicability due to practical considerations, for example, in the
optical regime dielectrics prevail, since conductive materials are
highly lossy.
[0041] Consider a 2D dielectric disk cavity of radius r and
permittivity c surrounded by air that supports high-Q
whispering-gallery modes, as shown in FIG. 2A. Such a cavity is
studied using both analytical modeling, such as separation of
variables in cylindrical coordinates and application of boundary
conditions, and detailed numerical finite-difference-time-domain
(FDTD) simulations with a resolution of 30 pts/r. Note that the
physics of the 3D case should not be significantly different, while
the analytical complexity and numerical requirements would be
immensely increased. The results of the two methods for the complex
eigen-frequencies and the field patterns of the so-called "leaky"
eigenmodes are in an excellent agreement with each other for a
variety of geometries and parameters of interest.
[0042] The radial modal decay length, which determines the coupling
strength .kappa..ident.|.kappa..sub.21|=|.kappa..sub.12|, is on the
order of the wavelength, therefore, for near-field coupling to take
place between cavities whose distance is much larger than their
size, one needs subwavelength-sized resonant objects
(r<<.lamda.). High-radiation-Q and long-tailed subwavelength
resonances can be achieved, when the dielectric permittivity
.epsilon. is as large as practically possible and the azimuthal
field variations (of principal number m) are slow (namely m is
small).
[0043] One such TE-polarized dielectric-cavity mode, which has the
favorable characteristics Q.sub.rad=1992 and .lamda./r=20 using
.epsilon.=147.7 and m=2, is shown in FIG. 2A, and will be the
"test" cavity 18 for all subsequent calculations for this class of
resonant objects. Another example of a suitable cavity has
Q.sub.rad=9100 and .lamda./r=10 using .epsilon.=65.61 and m=3.
These values of c might at first seem unrealistically large.
However, not only are there in the microwave regime (appropriate
for meter-range coupling applications) many materials that have
both reasonably high enough dielectric constants and low losses,
for example, Titania: .epsilon..apprxeq.96,
Im{.epsilon.}/.epsilon..apprxeq.10.sup.-3; Barium tetratitanate:
.epsilon..apprxeq.37, Im{.epsilon.}/.epsilon..apprxeq.10.sup.-4;
Lithium tantalite: .epsilon..apprxeq.40,
Im{.epsilon.}/.epsilon..apprxeq.10.sup.-4; etc.), but also
.epsilon. could instead signify the effective index of other known
subwavelength (.lamda./r<<1) surface-wave systems, such as
surface-plasmon modes on surfaces of metal-like
(negative-.epsilon.) materials or metallodielectric photonic
crystals.
[0044] With regards to material absorption, typical loss tangents
in the microwave (e.g. those listed for the materials above)
suggest that Q.sub.abs.about..epsilon./Im{.epsilon.}.about.10000.
Combining the effects of radiation and absorption, the above
analysis implies that for a properly designed resonant
device-object d a value of Q.sub.d-2000 should be achievable. Note
though, that the resonant source s will in practice often be
immobile, and the restrictions on its allowed geometry and size
will typically be much less stringent than the restrictions on the
design of the device; therefore, it is reasonable to assume that
the radiative losses can be designed to be negligible allowing for
Q.sub.s.about.10000, limited only by absorption.
[0045] To calculate now the achievable rate of energy transfer, one
can place two of the cavities 20, 22 at distance D between their
centers, as shown in FIG. 2B. The normal modes of the combined
system are then an even and an odd superposition of the initial
modes and their frequencies are split by the coupling coefficient
.kappa., which we want to calculate. Analytically, coupled-mode
theory gives for dielectric objects
.kappa..sub.12=.omega..sub.2/2.intg.d.sup.3rE.sub.1*(r)E.sub.2(r).epsilon-
..sub.1(r)/.intg.d.sup.3r|E.sub.1(r)|.sup.2.epsilon.(r), where
.epsilon..sub.1,2(r) denote the dielectric functions of only object
1 alone or 2 alone excluding the background dielectric (free space)
and .epsilon.(r) the dielectric function of the entire space with
both objects present. Numerically, one can find .kappa. using FDTD
simulations either by exciting one of the cavities and calculating
the energy-transfer time to the other or by determining the split
normal-mode frequencies. For the "test" disk cavity the radius
r.sub.C of the radiation caustic is r.sub.C.apprxeq.11r, and for
non-radiative coupling D<r.sub.C, therefore here one can choose
D/r=10, 7, 5, 3. Then, for the mode of FIG. 3, which is odd with
respect to the line that connects the two cavities, the analytical
predictions are .omega./2.kappa.=1602, 771, 298, 48, while the
numerical predictions are .omega./2.kappa.=1717, 770, 298, 47
respectively, so the two methods agree well. The radiation fields
of the two initial cavity modes interfere constructively or
destructively depending on their relative phases and amplitudes,
leading to increased or decreased net radiation loss respectively,
therefore for any cavity distance the even and odd normal modes
have Qs that are one larger and one smaller than the initial
single-cavity Q=1992 (a phenomenon not captured by coupled-mode
theory), but in a way that the average F is always approximately
.GAMMA..apprxeq..omega./2Q. Therefore, the corresponding
coupling-to-loss ratios are .kappa./T=1.16, 2.59, 6.68, 42.49, and
although they do not fall in the ideal operating regime
.kappa./.GAMMA.<<1, the achieved values are still large
enough to be useful for applications.
[0046] Consider a loop 10 or 12 of N coils of radius r of
conducting wire with circular cross-section of radius a surrounded
by air, as shown in FIG. 3. This wire has inductance
L=.mu..sub.0N.sup.2r[In(8r/a)-2], where .mu..sub.0 is the magnetic
permeability of free space, so connecting it to a capacitance C
will make the loop resonant at frequency .omega.=1/ {square root
over (LC)}. The nature of the resonance lies in the periodic
exchange of energy from the electric field inside the capacitor due
to the voltage across it to the magnetic field in free space due to
the current in the wire. Losses in this resonant system consist of
ohmic loss inside the wire and radiative loss into free space.
[0047] For non-radiative coupling one should use the near-field
region, whose extent is set roughly by the wavelength .lamda.,
therefore the preferable operating regime is that where the loop is
small (r<<.lamda.). In this limit, the resistances associated
with the two loss channels are respectively R.sub.ohm= {square root
over (.parallel..sub.0.rho..omega./2)}Nr/a and
R.sub.rad=.pi./6.eta..sub.0N.sup.2(.omega.r/c).sup.4, where .rho.
is the resistivity of the wire material and
.eta..sub.0.apprxeq.120.pi..OMEGA. is the impedance of free space.
The quality factor of such a resonance is then
Q=.omega.L/(R.sub.ohm+R.sub.rad) and is highest for some frequency
determined by the system parameters: at lower frequencies it is
dominated by ohmic loss and at higher frequencies by radiation.
[0048] To get a rough estimate in the microwave, one can use one
coil (N=1) of copper (p=1.6910.sup.-8 .OMEGA.m) wire and then for
r=1 cm and a=1 mm, appropriate for example for a cell phone, the
quality factor peaks to Q=1225 at f=380 MHz, for r=30 cm and a=2 mm
for a laptop or a household robot Q=1103 at f=17 MHz, while for r=1
m and a=4 mm (that could be a source loop on a room ceiling) Q=1315
at f=5 MHz. So in general, expected quality factors are
Q.apprxeq.1000-1500 at .lamda./r.apprxeq.50-80, namely suitable for
near-field coupling.
[0049] The rate for energy transfer between two loops 10 and 12 at
distance D between their centers, as shown in FIG. 3, is given by
.kappa..sub.12=.omega.M/2 {square root over (L.sub.1L.sub.2)},
where M is the mutual inductance of the two loops 10 and 12. In the
limit r<<D<<.lamda. one can use the quasi-static result
M=.pi./4.mu..sub.0N.sub.1N.sub.2(r.sub.1r.sub.2).sup.2/D.sup.3,
which means that .omega./2.kappa..about.(D/ {square root over
(r.sub.1r.sub.2)}).sup.3. For example, by choosing again D/r=10, 8,
6 one can get for two loops of r=1 cm, same as used before, that
.omega./2.kappa.=3033, 1553, 655 respectively, for the r=30 cm that
.omega./2.kappa.=7131, 3651, 1540, and for the r=1 m that
.omega./2.kappa.=6481, 3318, 1400. The corresponding
coupling-to-loss ratios peak at the frequency where peaks the
single-loop Q and are .kappa./.GAMMA.=0.4, 0.79, 1.97 and 0.15,
0.3, 0.72 and 0.2, 0.4, 0.94 for the three loop-kinds and
distances. An example of dissimilar loops is that of a r=1 m
(source on the ceiling) loop and a r=30 cm (household robot on the
floor) loop at a distance D=3 m (room height) apart, for which
.kappa./ {square root over (.GAMMA..sub.1.GAMMA..sub.2)}=0.88 peaks
at f=6.4 MHz, in between the peaks of the individual Q's. Again,
these values are not in the optimal regime
.kappa./.GAMMA.<<1, but will be shown to be sufficient.
[0050] It is important to appreciate the difference between this
inductive scheme and the already used close-range inductive schemes
for energy transfer in that those schemes are non-resonant. Using
coupled-mode theory it is easy to show that, keeping the geometry
and the energy stored at the source fixed, the presently proposed
resonant-coupling inductive mechanism allows for Q approximately
1000 times more power delivered for work at the device than the
traditional non-resonant mechanism, and this is why mid-range
energy transfer is now possible. Capacitively-loaded conductive
loops are actually being widely used as resonant antennas (for
example in cell phones), but those operate in the far-field regime
with r/.lamda..about.1, and the radiation Q's are intentionally
designed to be small to make the antenna efficient, so they are not
appropriate for energy transfer.
[0051] Clearly, the success of the inventive resonance-based
wireless energy-transfer scheme depends strongly on the robustness
of the objects' resonances. Therefore, their sensitivity to the
near presence of random non-resonant extraneous objects is another
aspect of the proposed scheme that requires analysis. The
interaction of an extraneous object with a resonant object can be
obtained by a modification of the coupled-mode-theory model in Eq.
(1), since the extraneous object either does not have a
well-defined resonance or is far-off-resonance, the energy exchange
between the resonant and extraneous objects is minimal, so the term
.kappa..sub.12 in Eq. (1) can be dropped. The appropriate
analytical model for the field amplitude in the resonant object
a.sub.1(t) becomes:
a 1 t = - ( .omega. 1 - .GAMMA. 1 ) a 1 + .kappa. 11 a 1 ( 2 )
##EQU00002##
[0052] Namely, the effect of the extraneous object is just a
perturbation on the resonance of the resonant object and it is
twofold: First, it shifts its resonant frequency through the real
part of .kappa..sub.11 thus detuning it from other resonant
objects. This is a problem that can be fixed rather easily by
applying a feedback mechanism to every device that corrects its
frequency, such as through small changes in geometry, and matches
it to that of the source. Second, it forces the resonant object to
lose modal energy due to scattering into radiation from the
extraneous object through the induced polarization or currents in
it, and due to material absorption in the extraneous object through
the imaginary part of .kappa..sub.11. This reduction in Q can be a
detrimental effect to the functionality of the energy-transfer
scheme, because it cannot be remedied, so its magnitude must be
quantified.
[0053] In the first example of resonant objects that have been
considered, the class of dielectric disks, small, low-index,
low-material-loss or far-away stray objects will induce small
scattering and absorption. To examine realistic cases that are more
dangerous for reduction in Q, one can therefore place the "test"
dielectric disk cavity 40 close to: a) another off-resonance object
42, such as a human being, of large Re{.epsilon.}=49 and
Im{.epsilon.}=16 and of same size but different shape, as shown in
FIG. 4A; and b) a roughened surface 46, such as a wall, of large
extent but of small Re{.epsilon.}=2.5 and Im{.epsilon.}=0.05, as
shown in FIG. 4B.
[0054] Analytically, for objects that interact with a small
perturbation the reduced value of radiation-Q due to scattering
could be estimated using the polarization
.intg.d.sup.3r|P.sub.x1(r)|.sup.2.varies..intg.d.sup.3r|E.sub.1(r)Re{.eps-
ilon..sub.X(r)}|.sup.2 induced by the resonant cavity 1 inside the
extraneous object X=42 or roughened surface X=46. Since in the
examined cases either the refractive index or the size of the
extraneous objects is large, these first-order perturbation-theory
results would not be accurate enough, thus one can only rely on
numerical FDTD simulations. The absorption-Q inside these objects
can be estimated through
Im{.kappa..sub.11}=.omega..sub.1/2.intg.d.sup.3r|E.sub.1(r)|.sup.2Im{.eps-
ilon..sub.X(r)}/.intg.d.sup.3r|E.sub.1(r)|.sup.2.epsilon.(r).
[0055] Using these methods, for distances D/r=10, 7, 5, 3 between
the cavity and extraneous-object centers one can find that
Q.sub.rad-1992 is respectively reduced to Q.sub.rad=1988, 1258,
702, 226, and that the absorption rate inside the object is
Q.sub.abs=312530, 86980, 21864, 1662, namely the resonance of the
cavity is not detrimentally disturbed from high-index and/or
high-loss extraneous objects, unless the (possibly mobile) object
comes very close to the cavity. For distances D/r=10, 7, 5, 3, 0 of
the cavity to the roughened surface we find respectively
Q.sub.rad=2101, 2257, 1760, 1110, 572, and Q.sub.abs>4000,
namely the influence on the initial resonant mode is acceptably
low, even in the extreme case when the cavity is embedded on the
surface. Note that a close proximity of metallic objects could also
significantly scatter the resonant field, but one can assume for
simplicity that such objects are not present.
[0056] Imagine now a combined system where a resonant source-object
s is used to wirelessly transfer energy to a resonant device-object
d but there is an off-resonance extraneous-object e present. One
can see that the strength of all extrinsic loss mechanisms from e
is determined by |E.sub.s(r.sub.c)|.sup.2, by the square of the
small amplitude of the tails of the resonant source, evaluated at
the position r.sub.e of the extraneous object. In contrast, the
coefficient of resonant coupling of energy from the source to the
device is determined by the same-order tail amplitude
|E.sub.s(r.sub.d)|, evaluated at the position r.sub.d of the
device, but this time it is not squared! Therefore, for equal
distances of the source to the device and to the extraneous object,
the coupling time for energy exchange with the device is much
shorter than the time needed for the losses inside the extraneous
object to accumulate, especially if the amplitude of the resonant
field has an exponential-like decay away from the source. One could
actually optimize the performance by designing the system so that
the desired coupling is achieved with smaller tails at the source
and longer at the device, so that interference to the source from
the other objects is minimal.
[0057] The above concepts can be verified in the case of dielectric
disk cavities by a simulation that combines FIGS. 2A-2B and 4A-4B,
namely that of two (source-device) "test" cavities 50 placed 10r
apart, in the presence of a same-size extraneous object 52 of
.epsilon.=49 between them, and at a distance 5r from a large
roughened surface 56 of .epsilon.=2.5, as shown in FIG. 5. Then,
the original values of Q=1992, .omega./2.kappa.=1717 (and thus
.kappa./.GAMMA.=1.16) deteriorate to Q=765, .omega./2.kappa.=965
(and thus .kappa./.GAMMA.=0.79). This change is acceptably small,
considering the extent of the considered external perturbation,
and, since the system design has not been optimized, the final
value of coupling-to-loss ratio is promising that this scheme can
be useful for energy transfer.
[0058] In the second example of resonant objects being considered,
the conducting-wire loops, the influence of extraneous objects on
the resonances is nearly absent. The reason for this is that, in
the quasi-static regime of operation (r<<.lamda.) that is
being considered, the near field in the air region surrounding the
loop is predominantly magnetic, since the electric field is
localized inside the capacitor. Therefore, extraneous objects that
could interact with this field and act as a perturbation to the
resonance are those having significant magnetic properties
(magnetic permeability Re{.mu.}>1 or magnetic loss
Im{.mu.}>0). Since almost all common materials are non-magnetic,
they respond to magnetic fields in the same way as free space, and
thus will not disturb the resonance of a conducting-wire loop. The
only perturbation that is expected to affect these resonances is a
close proximity of large metallic structures.
[0059] An extremely important implication of the above fact relates
to safety considerations for human beings. Humans are also
non-magnetic and can sustain strong magnetic fields without
undergoing any risk. This is clearly an advantage of this class of
resonant systems for many real-world applications. On the other
hand, dielectric systems of high (effective) index have the
advantages that their efficiencies seem to be higher, judging from
the larger achieved values of .kappa./.GAMMA., and that they are
also applicable to much smaller length-scales, as mentioned
before.
[0060] Consider now again the combined system of resonant source s
and device d in the presence of a human h and a wall, and now let
us study the efficiency of this resonance-based energy-transfer
scheme, when energy is being drained from the device for use into
operational work. One can use the parameters found before: for
dielectric disks, absorption-dominated loss at the source
Q.sub.s.about.10.sup.4, radiation-dominated loss at the device
Q.sub.d.about.10.sup.3 (which includes scattering from the human
and the wall), absorption of the source- and device-energy at the
human Q.sub.s-h, Q.sub.d-h.about.10.sup.4-10.sup.5 depending on
his/her not-very-close distance from the objects, and negligible
absorption loss in the wall; for conducting-wire loops,
Q.sub.s.about.Q.sub.d.about.10.sup.3, and perturbations from the
human and the wall are negligible. With corresponding loss-rates
.GAMMA.=.omega./2Q, distance-dependent coupling .kappa., and the
rate at which working power is extracted .GAMMA..sub.w, the
coupled-mode-theory equation for the device field-amplitude is
a d t = - ( .omega. - .GAMMA. d ) a d + .kappa. a s - .GAMMA. d - h
a d - .GAMMA. w a d . ( 3 ) ##EQU00003##
[0061] Different temporal schemes can be used to extract power from
the device and their efficiencies exhibit different dependence on
the combined system parameters. Here, one can assume steady state,
such that the field amplitude inside the source is maintained
constant, namely a.sub.s(t)=A.sub.se.sup.-i.omega.t, so then the
field amplitude inside the device is
a.sub.d(t)=A.sub.de.sup.-i.omega.t with
A.sub.d=i.kappa./(.GAMMA..sub.d+.GAMMA..sub.d-h+.GAMMA..sub.w)A.sub.s.
Therefore, the power lost at the source is
P.sub.s=2.GAMMA..sub.s|A.sub.s|.sup.2, at the device it is
P.sub.d=2.GAMMA..sub.d|A.sub.d|.sup.2, the power absorbed at the
human is
P.sub.h=2.GAMMA..sub.s-h|A.sub.s|.sup.2+2.GAMMA..sub.d-h|A.sub.d|.sup.2,
and the useful extracted power is
P.sub.w=2.GAMMA..sub.w|A.sub.d|.sup.2. From energy conservation,
the total power entering the system is
P.sub.total=P.sub.s+P.sub.d+P.sub.h+P.sub.w. Denote the total
loss-rates .GAMMA..sub.s.sup.tot=.GAMMA..sub.s+.GAMMA..sub.s-h and
.GAMMA..sub.d.sup.tot=.GAMMA..sub.d+.GAMMA..sub.d-h. Depending on
the targeted application, the work-drainage rate should be chosen
either .GAMMA..sub.w=.GAMMA..sub.d.sup.tot to minimize the required
energy stored in the resonant objects or
.GAMMA..sub.w=.GAMMA..sub.d.sup.tot {square root over
(1+.kappa..sup.2/.GAMMA..sub.s.sup.tot.GAMMA..sub.d.sup.tot)}>.GAMMA..-
sub.d.sup.tot such that the ratio of useful-to-lost powers, namely
the efficiency .eta..sub.w=P.sub.w/P.sub.total, is maximized for
some value of .kappa.. The efficiencies .eta. for the two different
choices are shown in FIGS. 6A and 6B respectively, as a function of
the .kappa./.GAMMA..sub.d figure-of-merit which in turn depends on
the source-device distance.
[0062] FIGS. 6A-6B show that for the system of dielectric disks and
the choice of optimized efficiency, the efficiency can be large,
e.g., at least 40%. The dissipation of energy inside the human is
small enough, less than 5%, for values .kappa./.GAMMA..sub.d>1
and Q.sub.h>10.sup.5, namely for medium-range source-device
distances (D.sub.d/r<10) and most human-source/device distances
(D.sub.h/r>8). For example, for D.sub.d/r=10 and D.sub.h/r=8, if
10W must be delivered to the load, then, from FIG. 6B, .about.0.4W
will be dissipated inside the human, .about.4W will be absorbed
inside the source, and .about.2.6W will be radiated to free space.
For the system of conducting-wire loops, the achieved efficiency is
smaller, .about.20% for .kappa./.GAMMA..sub.d.apprxeq.1, but the
significant advantage is that there is no dissipation of energy
inside the human, as explained earlier.
[0063] Even better performance should be achievable through
optimization of the resonant object designs. Also, by exploiting
the earlier mentioned interference effects between the radiation
fields of the coupled objects, such as continuous-wave operation at
the frequency of the normal mode that has the larger radiation-Q,
one could further improve the overall system functionality. Thus
the inventive wireless energy-transfer scheme is promising for many
modern applications. Although all considerations have been for a
static geometry, all the results can be applied directly for the
dynamic geometries of mobile objects, since the energy-transfer
time .kappa..sup.-1.about.1 .mu.s, which is much shorter than any
timescale associated with motions of macroscopic objects.
[0064] The invention provides a resonance-based scheme for
mid-range wireless non-radiative energy transfer. Analyses of very
simple implementation geometries provide encouraging performance
characteristics for the potential applicability of the proposed
mechanism. For example, in the macroscopic world, this scheme could
be used to deliver power to robots and/or computers in a factory
room, or electric buses on a highway (source-cavity would in this
case be a "pipe" running above the highway). In the microscopic
world, where much smaller wavelengths would be used and smaller
powers are needed, one could use it to implement optical
inter-connects for CMOS electronics or else to transfer energy to
autonomous nano-objects, without worrying much about the relative
alignment between the sources and the devices; energy-transfer
distance could be even longer compared to the objects' size, since
Im{.epsilon.(.omega.)} of dielectric materials can be much lower at
the required optical frequencies than it is at microwave
frequencies.
[0065] As a venue of future scientific research, different material
systems should be investigated for enhanced performance or
different range of applicability. For example, it might be possible
to significantly improve performance by exploring plasmonic
systems. These systems can often have spatial variations of fields
on their surface that are much shorter than the free-space
wavelength, and it is precisely this feature that enables the
required decoupling of the scales: the resonant object can be
significantly smaller than the exponential-like tails of its field.
Furthermore, one should also investigate using acoustic resonances
for applications in which source and device are connected via a
common condensed-matter object.
[0066] Although the present invention has been shown and described
with respect to several preferred embodiments thereof, various
changes, omissions and additions to the form and detail thereof,
may be made therein, without departing from the spirit and scope of
the invention.
* * * * *