U.S. patent application number 12/481385 was filed with the patent office on 2009-09-24 for apparatus for antenna diversity using joint maximal ratio combining.
This patent application is currently assigned to IPR LICENSING, INC.. Invention is credited to Gary L. Sugar, Yohannes Tesfai, Chandra Vaidyanathan.
Application Number | 20090239486 12/481385 |
Document ID | / |
Family ID | 30449285 |
Filed Date | 2009-09-24 |
United States Patent
Application |
20090239486 |
Kind Code |
A1 |
Sugar; Gary L. ; et
al. |
September 24, 2009 |
APPARATUS FOR ANTENNA DIVERSITY USING JOINT MAXIMAL RATIO
COMBINING
Abstract
A method for improving performance of radio frequency (RF)
communication of a station (STA) having an access point (AP) is
disclosed. The method includes using an arbitrary set of transmit
antenna weights, calculating a set of receive antenna weights, and
updating the transmit antenna weights based on the receive antenna
weights.
Inventors: |
Sugar; Gary L.; (Rockville,
MD) ; Vaidyanathan; Chandra; (Bethesda, MD) ;
Tesfai; Yohannes; (Falls Church, VA) |
Correspondence
Address: |
VOLPE AND KOENIG, P.C.;DEPT. ICC
UNITED PLAZA, SUITE 1600, 30 SOUTH 17TH STREET
PHILADELPHIA
PA
19103
US
|
Assignee: |
IPR LICENSING, INC.
Wilmington
DE
|
Family ID: |
30449285 |
Appl. No.: |
12/481385 |
Filed: |
June 9, 2009 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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11231161 |
Sep 20, 2005 |
7545778 |
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12481385 |
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10695229 |
Oct 28, 2003 |
6965762 |
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11231161 |
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10174728 |
Jun 19, 2002 |
6687492 |
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10695229 |
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60365797 |
Mar 21, 2002 |
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60361055 |
Mar 1, 2002 |
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Current U.S.
Class: |
455/101 |
Current CPC
Class: |
H04L 5/0007 20130101;
H04L 25/0204 20130101; H04L 25/0242 20130101; H04L 2025/03611
20130101; H04B 7/0669 20130101; H04L 25/03343 20130101; H04B 7/0845
20130101; H04L 25/0228 20130101; H04B 7/0854 20130101; H01Q 3/28
20130101; H04B 7/0417 20130101; H04L 27/2626 20130101; H04L 27/2602
20130101; H04B 7/0848 20130101; H04W 52/42 20130101; H04L
2025/03426 20130101; H04B 7/0617 20130101; H04B 7/0615 20130101;
H03G 3/3042 20130101; H04B 1/0483 20130101; H04L 1/06 20130101;
H04B 7/0857 20130101; H03G 3/3089 20130101; H04B 7/0671
20130101 |
Class at
Publication: |
455/101 |
International
Class: |
H04B 1/02 20060101
H04B001/02 |
Claims
1. A method for improving performance of radio frequency (RF)
communication of a station (STA) having an access point (AP), the
method comprising: transmitting a transmit antenna weight vector
using a complex transmit antenna weight vector for each of a
plurality of N antennas, wherein each complex transmit antenna
weight has a magnitude and a phase whose values may vary with
frequency; calculating a receive antenna weight vector comprising a
plurality of complex receive antenna weights for the N plurality of
antennas; and updating the transmit antenna weight vector for the
plurality of N antennas based on a computed conjugate of the
received antenna weight vector and a norm of the receive antenna
weight vector.
2. The method as in claim 1, wherein the transmit weight vector and
the receive antenna weight vector converge to values that optimize
the signal-to noise-ratio (SNR) of the STA.
3. The method as in claim 1, wherein the receive antenna weights
are computed from a matrix multiplication of an arbitrary set of
transmit antenna weights and a channel response matrix.
4. The method as in claim 2, further comprising storing in a memory
the optimized SNR value.
5. The method as in claim 4, wherein the optimized SNR value is
stored in a look-up-table.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of U.S. application Ser.
No. 11/231,161, filed Sep. 20, 2005, which is a continuation of
U.S. application Ser. No. 10/695,229, filed Oct. 28, 2003, which in
turn is a continuation of U.S. application Ser. No. 10/174,728,
filed Jun. 19, 2002, which issued on Feb. 3, 2004 as U.S. Pat. No.
6,687,492, which in turn claims priority to U.S. Provisional
Application No. 60/365,797, filed Mar. 21, 2002 and to U.S.
Provisional Application No. 60/361,055 filed Mar. 1, 2002, which
are incorporated by reference as if fully set forth.
FIELD OF THE INVENTION
[0002] The present application is directed to antenna (spatial)
signal processing useful in wireless communication applications,
such as short-range wireless applications.
[0003] Antenna diversity schemes are well known techniques to
improve the performance of radio frequency (RF) communication
between two RF devices. Types of antenna diversity schemes include
antenna selection diversity and maximal ratio combining. An antenna
selection diversity scheme selects one of two antennas for
transmission to a particular communication device based on which of
the two antennas best received a signal from the particular
communication device. On the other hand, maximal ratio combining
schemes involve beamforming a signal to be transmitted by two or
more antennas by scaling the signal with an antenna weight
associated with each antenna. A signal received by a plurality of
antennas can also be weighted by a plurality of receive antenna
weights. Selection of the antenna weights to optimize communication
between two communication devices is critical to the performance of
maximal ratio combining schemes.
[0004] There is room for improving the maximal ratio combining
antenna processing schemes to optimize the link margin between two
RF communication devices.
SUMMARY
[0005] An antenna signal processing scheme, hereinafter called
composite beamforming (CBF), is provided to optimize the range and
performance RF communication between two communication devices.
Composite beamforming (CBF) is a multiple-input multiple-output
(MIMO) antenna scheme that uses antenna signal processing at both
ends of the communication link to maximize the signal-to-noise
(SNR) and/or signal-to-noise-plus-interference (SNIR), thereby
improving the link margin between two communication devices, as
well as to provide for other advantages described herein.
[0006] Generally, a first communication device has a plurality of
antennas and the second communication has a plurality of antennas.
The first communication device transmits to the second
communication device using a transmit weight vector for
transmission by each the plurality of antennas and the transmit
signals are received by the plurality of antennas at the second
communication device. The second communication device determines
the receive weight vector for its antennas, and from that vector,
derives a suitable transmit weight vector for transmission on the
plurality of antennas back to the first communication device.
Several techniques are provided to determine the optimum frequency
dependent transmit weight vector and receive weight vector across
the bandwidth of a baseband signal transmitted between the first
and second communication devices so that there is effectively joint
or composite beamforming between the communication devices. The
link margin between communication devices is greatly improved using
the techniques described herein.
[0007] With the same antenna configuration, 2-antenna CBF (2-CBF)
provides an SNR improvement of up to 10 dB over transmit/selection
diversity when it is used at both ends of the link. A system design
using 4 antennas at a first communication device and 2 antennas at
a second communication device (hereinafter referred to as 4.times.2
CBF) provides nearly 14 dB of SNR improvement. In general, for a
fixed number of antennas, CBF outperforms the well-known space-time
block codes by up to 4 dB. Moreover, unlike space-time coding, CBF
does not require a change to an existing wireless standard.
[0008] The above and other objects and advantages will become more
readily apparent when reference is made to the following
description taken in conjunction with the accompanying
drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
[0009] FIG. 1 is a block diagram of two communication devices
performing an antenna diversity scheme using composite
beamforming.
[0010] FIG. 2 is a block diagram of a communication device that may
be used at either end of a composite beamforming communication
link.
[0011] FIG. 3 is a flow chart illustrating an adaptive method to
obtain antenna beamforming weights for communication between first
and second communication devices.
[0012] FIGS. 4 and 5 are graphical diagrams illustrating
convergence properties of the adaptive method shown in FIG. 3.
[0013] FIG. 6 is a diagram showing application of beamforming
weights to a baseband signal for transmission in a frequency
dependent communication channel.
[0014] FIG. 7 is a flow chart illustrating an adaptive algorithm to
obtain antenna beamforming weights in a frequency dependent
communication channel.
[0015] FIG. 8 is a block diagram of a composite beamforming
transmission process for a multi-carrier baseband modulation
scheme.
[0016] FIG. 9 is a block diagram of a composite beamforming
reception process for a multi-carrier baseband modulation
scheme.
[0017] FIG. 10 is a block diagram of a composite beamforming
transmission process for a single carrier baseband modulation
scheme.
[0018] FIG. 11 is a block diagram of a composite beamforming
reception process for a single carrier baseband modulation
scheme.
[0019] FIG. 12 is a flow chart of a beamforming training method
where one communication device of a communication link uses antenna
selection diversity.
[0020] FIG. 13 is a flow chart illustrating a collaborative method
to obtain antenna processing parameters for communication between
first and second communication devices.
DETAILED DESCRIPTION
[0021] Referring first to FIG. 1, a system 10 is shown in which a
first communication device 100 and a second communication device
200 communicate with each other using radio frequency (RF)
communication techniques. The techniques described herein are
useful in any radio frequency (RF) communication application, such
as short-range wireless applications. A wireless local area network
(WLAN) is only an example of an application. For example, device
100 may be an access point (AP) in a WLAN, and device 200 may be a
station (STA).
[0022] Generally, the device 100 has Nap antennas 110 and the
device 200 has Nsta antennas 210. FIG. 1 shows an example where the
device 100 has three antennas 110 and the device 200 has two
antennas 210. A complex transmit symbol at device 100 is scaled
(multiplied) using a set of complex transmit antenna weights
w.sub.tx,ap=[w.sub.0 . . . w.sub.Nap-1].sup.T before being
transmitted through respective ones of the antennas 110 (.sup.T
denotes the transpose operator). The received vector at device 200
is r.sub.sta=sH w.sub.tx,ap+n, where H is an Nsta by Nap channel
matrix of unity variance, complex Gaussian random variables (to
represent flat Rayleigh fading between each antenna), and n
represents noise and interference. At device 200, a combiner
C=(w.sub.rx,sta).sup.Hr is applied, C is passed to a detection
circuit (e.g., soft-decision QAM detector, etc.). If H is known at
both the transmitter and the receiver, w.sub.tx,ap and w.sub.rx,sta
may be selected to maximize the signal-to-noise ratio (SNR) at the
output of the combiner subject to a transmit power constraint,
i.e., (w.sub.tx,ap).sup.H w.sub.tx,ap=1.
[0023] The SNR for C is maximized over w.sub.tx,ap and w.sub.rx,sta
when w.sub.tx,ap is equal to e.sub.max, the unit norm eigenvector
for the maximum eigenvalue .lamda..sub.max of the matrix H.sup.HH,
and w.sub.rx,sta is a matched filter for He.sub.max, i.e.,
w.sub.rx,sta=k He.sub.max for some nonzero constant k. Under these
conditions, the SNR for C is equal to .lamda..sub.max. Since H is a
random matrix, .lamda..sub.max is a random variable. The
distribution on .lamda..sub.max is well known, and can be found in
M. Wennstrom, M. Helin, A. Rydberg, T. Oberg, "On the Optimality
and Performance of Transmit and Receive Space Diversity in MIMO
Channels", IEEE Technical Seminar on MIMO Systems, London,
December, 2001, which is incorporated herein by reference.
[0024] The transmit device and the receive device communicate using
time-division-duplexing at the same frequency. The channel matrix
for the reverse link is H.sub.r=H.sup.T, and the optimum transmit
weight vector w.sub.tx,ap is equal to the eigenvector for the
maximum eigenvalue of H.sub.r.sup.HH.sub.r=H*H.sup.T (* denotes the
conjugate operator). The maximum SNR at either end of the link is
the same (since it is a well known result that the nonzero
eigenvalues for both H*H.sup.T and H.sup.HH are the same). The
beamforming technique that results from this analysis is
hereinafter referred to as composite beamforming (CBF).
[0025] The communication devices at both ends of the link, i.e.,
devices 100 and 200 may have any known suitable architecture to
transmit, receive and process signals. An example of a
communication device block diagram is shown in FIG. 2. The
communication device 300 comprises an RF section 310, a baseband
section 320 and optionally a host 330. There are a plurality of
antennas, e.g., four antennas 302, 304, 306, 308 coupled to the RF
section 310 that are used for transmission and reception. The RF
section 310 has a transmitter (Tx) 312 that upconverts baseband
signals for transmission, and a receiver (Rx) 314 that downconverts
received RF signals for baseband processing. In the context of the
composite beamforming techniques described herein, the Tx 312
upconverts and supplies separately weighted signals to
corresponding ones of each of the plurality of antennas via
separate power amplifiers. Similarly, the Rx 314 downconverts and
supplies received signals from each of the plurality of antennas to
the baseband section 320. The baseband section 320 performs
processing of baseband signals to recover the information from a
received signal, and to convert information in preparation for
transmission. The baseband section 320 may implement any of a
variety of communication formats or standards, such as WLAN
standards IEEE 802.11x, Bluetooth.TM., as well as other protocol
standards, not necessarily used in a WLAN.
[0026] The intelligence to execute the computations for the
composite beamforming techniques described herein may be
implemented in a variety of ways. For example, a processor 322 in
the baseband section 320 may execute instructions encoded on a
processor readable memory 324 (RAM, ROM, EEPROM, etc.) that cause
the processor 322 to perform the composite beamforming steps
described herein. Alternatively, an application specific integrated
circuit (ASIC) configured with the appropriate firmware, e.g.,
field programmable gates that implement digital signal processing
instructions to perform the composite beamforming steps. This ASIC
may be part of, or the entirety of, the baseband section 320. Still
another alternative is for the beamforming computations to be
performed by a host processor 332 (in the host 330) by executing
instructions stored in (or encoded on) a processor readable memory
334. The RF section 310 may be embodied by one integrated circuit,
and the baseband section 320 may be embodied by another integrated
circuit. The communication device on each end of the communication
link need not have the same device architecture or
implementation.
[0027] Regardless of the specific implementation chosen, the
composite beamforming process is generally performed as follows. A
transmit weight vector (comprising a plurality of complex transmit
antenna weights corresponding to the number of transmit antennas)
is applied to, i.e., multiplied by, a baseband signal to be
transmitted, and each resulting weighted signal is coupled to a
transmitter where it is upconverted, amplified and coupled to a
corresponding one of the transmit antennas for simultaneous
transmission. At the communication device on the other end of the
link, the transmit signals are detected at each of the plurality of
antennas and downconverted to a baseband signal. Each baseband
signal is multiplied by a corresponding one of the complex receive
antenna weights and combined to form a resulting receive signal.
The architecture of the RF section necessary to accommodate the
beamforming techniques described herein may vary with a particular
RF design, and many are known in the art and thus is not described
herein.
[0028] Turning to FIG. 3, a process 400 is shown for achieving
optimum CBF between two communication devices. To restate the
results from the previous discussion, the optimum receive and
transmit weights at the AP are given by w.sub.rx,ap=e.sub.max(of
H.sup.HH), w.sub.tx,ap=w.sub.rx,ap*. The optimum receive and
transmit weights at the STA are given by w.sub.rx,sta=e.sub.max(of
H*H.sup.T), w.sub.tx,sta=w.sub.rx,sta*. Additionally,
w.sub.rx,sta=H w.sub.tx,ap, w.sub.rx,ap=H.sup.T w.sub.tx,ap. These
properties can be utilized to design an adaptive/iterative
algorithm that converges to the optimum eigenvector as follows.
[0029] Initially, in step 410, the AP uses an arbitrary set of
transmit antenna weights to transmit a signal to the STA. When the
STA receives the signal, the receive antenna weights at the STA are
matched to the receive signal such that w.sub.rx,sta(0)=H
w.sub.tx,ap(0). That is, the STA receive antenna weights are
computed from the received signals at each of the antennas by
matching to the received signals. In step 420, the STA computes the
conjugate of the receive weight vector made up of the receive
antenna weights for use as the transmit antenna weight vector for
transmitting on the STA's antennas back to the AP. The AP receives
the signal transmitted by the plurality of antennas of the STA and
matches the receive antenna weights to the received signal.
[0030] In step 430, the AP updates the new transmit antenna weights
by computing the conjugate of the receive weight vector (comprised
of the AP receive antenna weights) divided by the norm of the AP
receive weight vector. This process repeats in steps 440 through
460, ad infinitum. It can be shown that the weights converge to the
eigenvector corresponding to the maximum eigenvalue. See G. Golub,
C. V. Loan, "Matrix Computations", 2nd edition, pp. 351.
[0031] Within a few iterations, the transmit weight vector and
receive antenna weight vector of both devices will converge to
values that optimize the SNR at each of the devices. At such point,
the first communication device may store in a memory (in the
baseband section or host processor section) the current optimum
transmit antenna weights for a particular destination communication
device indexed against an identifier for that communication device.
The first communication device, such as an AP, may store in a
look-up-table optimum transmit antenna weights indexed against
corresponding identifiers (such as MAC addresses) for a plurality
of other communication devices it communicates with.
[0032] The adaptive process of FIG. 3 will converge to optimum
antenna weights even if one device has multiple antennas and can
weight signals supplied thereto, and the other device is a merely a
single antenna device. The device on the link with the multiple
antennas and combining capability can still converge to its optimum
transmit and receive weights for a single antenna device it
communicates with.
[0033] With reference to FIGS. 4 and 5, the convergence properties
of the adaptive algorithm were studied over 1000 randomly generated
channels. The average SNR at each receive antenna was set to 10 dB.
The normalized antenna array gain at the output of the receive
antenna array, |Hw.sub.tx|.sup.2/.lamda..sub.max(H.sup.HH) is used
to study the performance. In FIG. 4, (Nap=2, Nsta=2), it is shown
that the adaptive algorithm loss is less than a 1 dB at the 3rd
iteration with 95% probability. When the number of antenna elements
is increased to four, only one additional iteration was required
for the algorithm converge to less than 1 dB loss with 95%
probability.
[0034] An advantage of adaptive composite beamforming is that no
special training sequence is required for adaptation. In addition,
no changes to existing protocols are necessary, and there is no
impact on throughput. The antenna weights are updated when real
information or data is transmitted between devices. Transmit and
receive weight adaptation is the same regardless of whether CBF is
implemented at both ends of the link. However, if the destination
device uses selection diversity the performance can be improved by
estimating the channel response.
[0035] The indoor wireless channel is a frequency dependent
channel. Due to multi-path propagation the signal arrives at the
receiver with different delays. The different delays cause the
channel transfer function to be frequency selective. Therefore, to
account for these delays, the antenna weights need to be adjusted
according to the frequency dependent characteristics of the channel
transfer function between the transmitting device and the receiving
device.
[0036] Solutions for optimum antenna processing in a frequency
selective channel are described hereinafter. Between any two
communication devices, the communication channel will have a
frequency response depending on frequency selective fading
conditions, etc. The channel transfer function H(f) describes the
frequency response and is used to select the optimum antenna
transmit and receive weights for communication between those
terminals.
[0037] To understand the frequency selective situation, reference
is again made to FIG. 1, where the frequency dependent Nsta by Nap
transfer function between the first and communication device and
the second communication devices is denoted by the H(f). The Nap by
Nsta transfer function between the second communication device and
the first communication device is H.sup.T(f). The transmit weights
at the first and second communication devices are denoted by the
Nap.times.1 vector w.sub.tx,ap(f) and the Nsta.times.1 vector
w.sub.tx,sta(f), respectively.
w.sub.tx,ap(f)=[w.sub.tx,ap,1(f), w.sub.tx,ap,2(f), . . .
w.sub.tx,ap,Nap(f)].sup.T
w.sub.tx,sta(f)[w.sub.tx,sta,1(f), w.sub.tx,sta,2(f), . . .
w.sub.tx,sta,Nsta(f)].sup.T
[0038] The receive weights at the first and second communication
devices are denoted by the Nap.times.1 vector w.sub.rx,ap(f) and
the Nsta.times.1 vector w.sub.rx,sta(f), respectively
w.sub.rx,ap(f)=[w.sub.rx,ap,1(f), w.sub.rx,ap,2(f), . . .
w.sub.rx,ap,Nap(f)].sup.T
w.sub.rx,sta(f)=[w.sub.rx,sta,1(f), w.sub.rx,sta,2(f), . . .
w.sub.rx,sta,Nsta(f)].sup.T
[0039] The transmit and receive weights (only the first
communication device-second communication device link is described
below but the results apply in the reverse direction with
appropriate change in notation) are computed by optimizing a cost
function, C, with a constraint on the maximum transmit power. In a
communication system, the ultimate goal is to reduce bit-error rate
(BER).
[0040] However, optimization using the BER as a cost function is
not always analytically feasible. Therefore, cost functions that
implicitly reduce the BER are usually selected. The cost function
also depends on the receiver structure. Selection of the cost
function for different modulation schemes and receiver structures
is discussed.
min w tx , ap w rx , sta .intg. - 1 / 2 T 1 / 2 T C ( H ( f ) , w
tx , ap ( f ) , w rx , sta ( f ) ) f , subject to .intg. - 1 / 2 T
1 / 2 T | w tx , ap ( f ) | 2 f .ltoreq. P 0 ##EQU00001##
[0041] For a code division multiple access (CDMA) communication
system, such as IEEE 802.11b, the receiver is assumed to be a RAKE
receiver and the BER is a function of the SNIR (signal to
noise+interference ratio) at the output of the RAKE receiver.
Maximizing the SNIR at the output of the RAKE receiver minimizes
the BER.
max w tx , ap w rx , sta .intg. - 1 / 2 T 1 / 2 T SNIR ( H ( f ) ,
w tx , ap ( f ) , w rx , sta ( f ) ) f , subject to .intg. - 1 / 2
T 1 / 2 T | w tx , ap ( f ) | 2 f .ltoreq. P 0 ##EQU00002##
[0042] For an orthogonal frequency division multiplex (OFDM)
system, such as IEEE 802.11a, the receiver is a linear equalizer
followed by a Viterbi decoder. Since the Viterbi decoder is a
non-linear operator, optimizing the coded BER is very challenging.
An alternative is to minimize the mean square error (MSE) at the
output of the linear equalizer (note another possible approach is
to minimize the uncoded BER).
min w tx , ap w rx , sta .intg. - 1 / 2 T 1 / 2 T MSE ( H ( f ) , w
tx , ap ( f ) , w rx , sta ( f ) ) f , subject to .intg. - 1 / 2 T
1 / 2 T | w tx , ap ( f ) | 2 f .ltoreq. P 0 ##EQU00003##
[0043] A single carrier modulation scheme, such as IEEE 802.11b,
uses a decision feedback equalizer (DFE) at the receiver. The
receiver is a non-linear receiver. The transmit, receive and
feedback weights are computed jointly. This can be achieved by
minimizing the MSE at the output of the DFE.
min w tx , ap w rx , sta B .intg. - 1 / 2 T 1 / 2 T MSE ( H ( f ) ,
w tx , ap ( f ) , w rx , sta ( f ) ) f , subject to .intg. - 1 / 2
T 1 / 2 T | w tx , ap ( f ) | 2 f .ltoreq. P 0 ##EQU00004##
[0044] For all cases considered, the optimum transmit weights are
given by
w.sub.tx.sub.--.sub.ap(f)=p(f)e.sub.max(H.sup.H(f)H(f))
where e.sub.max is the eigenvector corresponding to the maximum
eigenvalue of the matrix H.sup.H(f) H(f), where p (f) is a
weighting function that weights each individual frequency bin and
is based on the cost function. Typically, the solution to p(f)
follows a waterpouring distribution.
[0045] For the linear equalizer case, the solution is given by
p 2 ( f ) = 1 .mu. .sigma. s 2 .sigma. n 2 .lamda. ma x ( f ) - 1
.sigma. s 2 .sigma. n 2 .lamda. ma x ( f ) ##EQU00005## SNR =
.sigma. s 2 .sigma. n 2 ##EQU00005.2##
[0046] For the DFE case, the solution is
p 2 ( f ) = 1 .mu. - 1 .sigma. s 2 .sigma. n 2 .lamda. max ( f )
##EQU00006##
where .mu. is selected to satisfy the power constraint
.intg. 1 / 2 T 1 / 2 T p 2 ( f ) f = P 0 ##EQU00007##
[0047] An optimal solution for p(f) requires knowledge of the
channel and SNR at the receiver. A suboptimal solution is obtained
by setting p(f) to a constant, p, across frequency.
w.sub.tx.sub.--.sub.ap(f)=pe.sub.max(H.sup.H(f)H(f))
[0048] This is referred to herein as a frequency shaping
constraint. To explain further, the frequency shaping constraint
requires that at each frequency of the baseband signal to be
transmitted (e.g., frequency sub-band or frequency sub-carrier k),
the sum of the power of signals across all of the transmit antennas
is equal to a constant value, P.sub.tx/K. This constraint is useful
to ensure that, in an iterative process between two communication
devices, the transmit weights of the two devices will converge to
optimal values. An additional benefit of this constraint is that
the transmitting device can easily satisfy spectral mask
requirements of a communication standard, such as IEEE 802.11x.
[0049] This solution does not require knowledge of the receiver SNR
and simulations have shown that the loss in performance over the
optimal solution is negligible. However, this solution requires
knowledge of the channel response at the transmitter.
[0050] For the cost functions maximizing the SNIR or minimizing the
MSE for a linear equalizer, the optimum receive weights are given
by
w.sub.rx,sta(f)=R.sub.ss.sup.-1(f)v.sub.mf,sta(f)
[0051] where v.sub.mf,sta(f) is matched to the received signal
v.sub.mf,sta(f)=H(f)w.sub.tx,ap(f)
and R.sub.ss(f) is the correlation matrix defined as
R ss ( f ) = .sigma. s 2 H ( f ) w tx , ap ( f ) w tx , ap H H H (
f ) + .sigma. n 2 I = .sigma. s 2 v mf , sta ( f ) v mf , sta H ( f
) + .sigma. n 2 I ##EQU00008##
[0052] When the MSE of the DFE is the minimized, the optimum
receive weights are given by
w.sub.rx,sta(f)=R.sub.ss.sup.-1(f)v.sub.mf,sta(f)(1+B(f))
where B(f) is the feedback filter.
[0053] The weights for the reverse link are similar to the forward
link and is summarized below. The optimum transmit weights at the
second communication are given by
w.sub.tx.sub.--.sub.sta(f)=p(f)e.sub.max(H*(f)H.sup.T(f))
and the suboptimal transmit weights are
w.sub.tx.sub.--.sub.sta(f)=pe.sub.max(H*(f)H.sup.T(f))
[0054] Similarly, the receive weights at the first communication
device are given by
w.sub.rx,ap(f)=R.sub.aa.sup.-1(f)v.sub.mf,ap(f)
where
v.sub.mf,ap(f)=H.sup.T(f)w.sub.tx,sta(f)
[0055] and for DFE case
R.sub.aa(f)=.sigma..sub.s.sup.2v.sub.mf,ap(f)+.sigma..sub.n.sup.2I
w.sub.rx,ap(f)=R.sub.xx.sup.-1(f)(1+B(f))
[0056] In the presence of co-channel interference R.sub.ss(f) is
given by
R ss ( f ) = .sigma. s 2 H ( f ) w tx , ap ( f ) w tx , ap H ( f )
H H ( f ) + k .noteq. 0 .sigma. k 2 H k ( f ) w k ( f ) w k H ( f )
H k H ( f ) + .sigma. n 2 I ##EQU00009##
where the terms in the summation are the contribution due to the
interferes. In this case, the optimum receive antenna weights
minimize the contribution of the interferes and the noise.
Therefore, in addition to diversity gain, optimum antenna combining
at the receiver also provides interference suppression
capability.
[0057] FIG. 6 illustrates how frequency selective beamforming
weights are applied to a baseband signal. The baseband signal may
be a single carrier signal or a multi-carrier signal. In either
case, the baseband signal will have a bandwidth or spectrum.
According to the composite beamforming (CBF) technique described
herein, when communication device 100 transmits a signal to
communication device 200, it applies (i.e., multiplies or scales) a
baseband signal s to be transmitted by a transmit weight vector
associated with a particular destination device, e.g.,
communication device 200, denoted w.sub.tx,1. Similarly, when
communication device 200 transmits a baseband signal s to
communication device 100, it multiplies the baseband signal s by a
transmit weight vector w.sub.tx,2, associated with destination
communication device 100. The (M.times.N) frequency dependent
channel matrix from the N plurality of antennas of the first
communication device 100 to M plurality of antennas of the second
communication device 200 is H(k), and the frequency dependent
communication channel (N.times.M) matrix between the M plurality of
antennas of the second communication device and the N plurality of
antennas of the first communication device is H.sup.T(k). The
variable k denotes the frequency dependent characteristic as
explained further hereinafter.
[0058] The transmit weight vectors w.sub.tx,1 and w.sub.tx,2 each
comprises a plurality of transmit weights corresponding to each of
the N and M antennas, respectively. Each transmit weight is a
complex quantity. Moreover, each transmit weight vector is
frequency dependent; it varies across the bandwidth of the baseband
signal s to be transmitted. For example, if the baseband signal s
is a multi-carrier signal of K sub-carriers, each transmit weight
for a corresponding antenna varies across the K sub-carriers.
Similarly, if the baseband signal s is a single-carrier signal
(that can be divided into K frequency sub-bands), each transmit
weight for a corresponding antenna varies across the bandwidth of
the baseband signal. Therefore, the transmit weight vector is
dependent on frequency, or frequency sub-band/sub-carrier k, such
that w.sub.tx becomes w.sub.tx(f), or more commonly referred to as
w.sub.tx(k), where k is the frequency sub-band/sub-carrier
index.
[0059] While the terms frequency sub-band/sub-carrier are used
herein in connection with beamforming in a frequency dependent
channel, it should be understood that the term "sub-band" is meant
to include a narrow bandwidth of spectrum forming a part of a
baseband signal. The sub-band may be a single discrete frequency
(within a suitable frequency resolution that a device can process)
or a narrow bandwidth of several frequencies.
[0060] The receiving communication device also weights the signals
received at its antennas with a frequency dependent receive antenna
weight vector w.sub.rx(k). Communication device 100 uses a receive
antenna weight vector w.sub.rx,1(k) when receiving a transmission
from communication device 200, and communication device 200 uses a
receive antenna weight vector w.sub.rx,2(k) when receiving a
transmission from communication device 100. The receive antenna
weights of each vector are matched to the received signals by the
receiving communication device.
[0061] Generally, transmit weight vector w.sub.tx,1 comprises a
plurality of transmit antenna weights
w.sub.tx,1,i=.beta..sub.1,i(k)e.sup.j.phi.1,i,(k), where
.beta..sub.1,i(k) is the magnitude of the antenna weight,
.phi.1,i,(k) is the phase of the antenna weight, i is the antenna
index (up to N), and k is the frequency sub-band or sub-carrier
index (up to K frequency sub-bands/sub-carriers). The subscripts
tx,1 denote that it is a vector that communication device 100 uses
to transmit to communication device 200. Similarly, the subscripts
tx,2 denote that it is a vector that communication device 200 uses
to transmit to communication device 100.
[0062] The frequency shaping constraint described above may be
imposed on the transmit weights for each antenna. As mentioned
above, the constraint requires that at each frequency of the
baseband signal to be transmitted (e.g., frequency sub-band or
frequency sub-carrier k), the sum of the power of signals across
all of the transmit antennas (|w.sub.tx,i(k)|.sup.2 for i=1 to N)
is equal to a constant value, P.sub.tx/K.
[0063] The relationship between transmit and receive weights are
summarized below: The optimum receive and transmit weights at the
first communication device are related as follows.
w.sub.tx,ap(f)=emax(H.sup.H(f)H(f)),v.sub.mf,sta(f)=H(f)w.sub.tx,ap(f)
[0064] Similarly at the second communication device, the optimum
receive and transmit weights are related as follows.
w.sub.tx,sta(f)=emax(H*(f)H.sup.T(f)),v.sub.mf,ap(f)=H.sup.T(f)w.sub.tx,-
sta(f)
Additionally,
v.sub.mf,ap(f)=w.sub.tx,ap*(f),v.sub.mf,sta(f)=w.sub.tx,ap*(f)
[0065] The properties outlined above can be utilized in an
adaptive/iterative process 480 shown in FIG. 7 that is similar to
the process shown in FIG. 3. The antenna weight parameters in FIG.
4 are written with indexes to reflect communication between an AP
and a STA, but without loss of generality, it should be understood
that this process is not limited to a WLAN application, and is
useful in any wireless application, such as a short-range
application. The AP has Nap antennas and the STA has Nsta antennas.
Assuming the AP begins with the first transmission to the STA, the
initial AP transmit weight vector w.sub.T,AP,0(k) is [1, 1, . . .
1], normalized by 1/(Nap).sup.1/2 for all antennas and all
frequency sub-bands/sub-carriers k. Phase for the transmit antenna
weights are also initially set to zero. The index T indicates it is
a transmit weight vector, index AP indicates it is an AP vector,
index 0 is the iteration of the vector, and (k) indicates that it
is frequency sub-band/sub-carrier dependent. In step 482, a
baseband signal is scaled by the initial AP transmit weight vector
w.sub.T,AP,0(k), upconverted and transmitted to the STA by the Nap
antennas. The transmitted signal is effectively altered by the
frequency selective channel matrix H(k) from AP-STA. The STA
receives the signal and matches its initial receive weight vector
w.sub.R,STA,0(k) to the signals received at its antennas. In step
484, the STA normalizes the receive weight vector w.sub.R,STA,0(k)
and computes the conjugate of normalized receive weight vector to
generate the STA's initial transmit weights for transmitting a
signal back to the AP. In step 486, the STA processes the signal to
be transmitted to the AP by the initial transmit weight vector,
upconverts that signal and transmits it to the AP. The transmitted
signal is effectively altered by the frequency selective channel
matrix H.sup.T(k). At the AP, the receive weight vector is matched
to the signals received at its antennas. The AP then computes the
conjugate of the gain-normalized receive weight vector as the next
transmit weight vector w.sub.T,AP,1(k) and transmits a signal to
the STA with that transmit weight vector. The STA receives the
signal transmitted from the AP with this next transmit weight
vector and matches to the received signals to compute a next
receive weight vector w.sub.R,STA,1(k). Again, the STA computes the
conjugate of the gain-normalized receive weight vector
w.sub.R,STA,1(k) as its next transmit weight vector
w.sub.T,STA,1(k) for transmitting a signal back to the AP. This
process repeats for several iterations as shown by steps 488 and
490, ultimately converging to transmit weight vectors that achieve
nearly the same performance as non-equal gain composite
beamforming. This adaptive process works even if one of the
devices, such as a STA, has a single antenna for transmission and
reception.
[0066] When storing the transmit weights of a frequency transmit
weight vector, in order to conserve memory space in the
communication device, the device may store, for each antenna,
weights for a subset or a portion of the total number of weights
that span the bandwidth of the baseband signal. For example, if
there are K weights for K frequency sub-bands or sub-carrier
frequencies, only a sampling of those weights are actually stored,
such as weights for every other, every third, every fourth, etc., k
sub-band or sub-carrier. Then, the stored subset of transmit
weights are retrieved from storage when a device is to commence
transmission of a signal, and the remaining weights are generated
by interpolation from the stored subset of weights. Any suitable
interpolation can be used, such as linear interpolation, to obtain
the complete set of weights across the K sub-bands or sub-carriers
for each antenna.
[0067] With reference to FIG. 8, a beamforming transmission process
500 is shown for a multi-carrier baseband modulation scheme. For an
orthogonal frequency division multiplexed (OFDM) system used, for
example, by the IEEE 802.11a standard, the data symbols are in the
frequency domain. K symbols are assigned to K sub-carriers (K=52
for 802.11a). For convenience, each of the transmit antenna weights
are described as a function of (k), the sub-carrier frequency. Each
of the N antennas has a transmit antenna weight w.sub.tx that is a
function of k, i.e., w.sub.tx(k) over k=1 to K. The transmit
antenna weights are computed by any of the processes described
above at each of the sub-carrier frequencies. There is a signal
processing path for each of the N antennas. In each signal
processing path, a multiplier 510 multiplies the frequency domain
symbol s(k) by the corresponding transmit antenna weight
w.sub.tx(k) and because w.sub.tx(k) has K values, there are K
results from the multiplication process. The results are stored in
a buffer 520 for k=1 to K. An inverse Fast Fourier Transform (IFFT)
530 is coupled to the buffer to convert the frequency domain
results stored in buffer 520 to a digital time domain signal for
each of the K sub-carriers. There is some adjustment made for
cyclic prefixes caused by the OFDM process. A filter 540 provides
lowpass filtering of the result of the IFFT process. The digital
results of the filtering process are converted to analog signals by
a D/A 550. The outputs of the D/A 550 are coupled to RF circuitry
560 that upconverts the analog signals to the appropriate RF signal
which is coupled via a power amplifier (PA) 570 to one of the N
antennas 580. In this manner, for each antenna 580, the signal s(k)
is multiplied by respective transmit antenna weights whose values
may vary as a function of the sub-carrier frequency k. The
frequency shaping constraint described above can also be applied to
the antenna weights.
[0068] FIG. 9 shows a beamforming reception process 600 that is
essentially the inverse of the transmission process shown in FIG.
8. There is a signal processing channel for each of the antennas
580. RF circuitry 610 downconverts the RF signals detected at each
antenna 580 for each of the sub-carriers. An A/D 620 converts the
analog signal to a digital signal. A lowpass filter 630 filters the
digital signal. There is some adjustment made for cyclic prefixes
caused by the OFDM process. A buffer 640 stores the time domain
digital signal in slots associated with each sub-carrier frequency
k. An FFT 650 converts the time domain digital signal in buffer 640
to a frequency domain signal corresponding to each sub-carrier
frequency k. The output of the FFT 650 is coupled to a multiplier
660 that multiplies the digital signal for each sub-carrier k by a
corresponding receive antenna weight w.sub.rx(k) for the
corresponding one of the N antennas. The outputs of each of the
multipliers 660 are combined by an adder 670 to recover the digital
frequency domain symbol s(k). The signal s(k) is then mapped back
to symbol b(k).
[0069] FIGS. 10 and 11 show transmission and reception processes,
respectively, for frequency dependent beamforming applied to a
single carrier baseband modulation scheme, such as that used by the
IEEE 802.11b standard. The data symbols in such a system are in the
time domain. FIG. 10 shows a beamforming transmission process 700
suitable for a single carrier modulation scheme. Since in a
frequency dependent channel, the transmit antenna weights are
frequency dependent, the passband of the baseband signal is
synthesized into frequency bins (K bins) and transmit beamforming
weights are computed for each frequency bin using any of the
processes described above. There are processing channels for each
antenna. In each processing channel, transmit filters 710 are
synthesized with the frequency response specified by the
beamforming weights. Thus, each transmit filter 710 has a frequency
response defined by the transmit antenna weight w.sub.tx(f)
associated with that antenna. The data symbol s(n) is passed
through the transmit filter 710 which in effect applies the
frequency selective antenna weight to the data symbol s(n). The D/A
720 converts the digital output of the transmit filter 710 to an
analog signal. The RF circuitry 730 upconverts the analog signal
and couples the upconverted analog signal to an antenna 750 via a
power amplifier 740. The frequency shaping constraint described
above can also be applied to the antenna weights.
[0070] FIG. 11 shows a reception process 800 suitable for a single
carrier modulation scheme. There is a processing channel for each
antenna 750. In each processing channel, RF circuitry 810
downconverts the received RF signal. An A/D 820 converts the
downconverted analog signal to a digital signal. Like the frequency
dependent transmit antenna weights, the receive antenna weights are
computed for several frequency sub-bands. Receive filters 830 are
synthesized with the frequency response specified by the receive
beamforming weights w.sub.rx(f) and the received digital signal is
passed through filters 830 for each antenna, effectively applying a
frequency dependent antenna weight to the received signal for each
antenna. The results of the filters 830 are combined in an adder
850, and then passed to a demodulator 860 for demodulation.
[0071] Referring next to FIG. 12, a procedure is shown for use when
only one of the two devices supports beamforming. For example,
N-CBF is supported at a first communication device (an AP) but not
at a second communication device (a STA). In this case, the STA is
likely to support 2-antenna Tx/Rx selection diversity as discussed
previously. If this is the case, it is possible for the AP to
achieve 3 dB better performance than Nth order maximal ratio
combining (MRC) at both ends of the link.
[0072] When the STA associates or whenever a significant change in
channel response is detected, the AP sends a special training
sequence to help the STA select the best of its two antennas. The
training sequence uses messages entirely supported by the
applicable media access control protocol, which in the following
example is IEEE 802.11x.
[0073] The sequence consists of 2 data units (such as an IEEE
802.11 MSDU ideally containing data that is actually meant for the
STA so as not to incur a loss in throughput). In step 900, the
first communication device sends the first data unit using the Tx
weight vector [1 0 . . . 0].sup.T. That is, the first communication
device sends the first data unit exclusively by one of its N
antennas. In step 910, the second communication device responds by
transmitting a message using one of its' two antennas. The first
device decodes the message from the second device, and obtains one
row of the H matrix (such as the first row h.sub.r1). In step 920,
the first device sends the second MSDU using a weight vector which
is orthogonal to the first row of H (determined in step 910). When
the second device receives the second MSDU, in step 930, standard
selection diversity logic forces it to transmit a response message
in step 930 using the other antenna, allowing the first device to
see the second row of the H matrix, h.sub.r2. Now the first device
knows the entire H matrix. The first device then decides which row
of the H matrix will provide "better" MRC at the second device by
computing a norm of each row, h.sub.r1 and h.sub.r2, of the H
matrix and, and selecting the row that has the greater norm as the
transmit weight vector for further transmissions to that device
until another change is detected in the channel.
[0074] For the frequency sensitive case, the process shown in FIG.
11 is repeated at each of a plurality of frequency sub-bands that
span the bandwidth of a single carrier baseband signal to be
communicated between the devices, or at each of the sub-carrier
frequencies of a multi-carrier baseband signal to be communication
between the devices.
[0075] Turning to FIG. 13 with continued reference to FIG. 1, a
method is described for a collaborative approach for maintaining
channel response information at one communication device for
transmission with another communication device. Initially, in step
1000, one communication device determines which other communication
devices are CBF-capable using a special CBF-capability request
message. For example, this message is sent by an AP whenever a new
STA associates with the AP. Non-CBF-capable devices will not
respond to the message since they will not recognize it without CBF
capability. Once it has confirmed CBF capability, whenever a
CBF-capable device (AP or STA) sends information to the other
device, in step 1010, a CBF training sequence is generated and
appended to a data unit. For example, in the context of IEEE
802.11x, when a directed media access control (MAC) Protocol Data
Unit (MPDU) to another CBF-capable terminal, it appends a small
(2*N orthogonal frequency division multiplexed (OFDM) symbols,
N=the number of antennas of the transmitting device) CBF training
sequence containing channel response information at the end of the
MPDU data segment. For example, the CBF training sequence may
comprise N consecutive 2-symbol long preamble sequences as defined
in 802.11a. These N sequences are multiplied by respective ones of
N linearly independent vectors that span the column matrix of the
channel response matrix. Such N linearly independent vectors may
be, for example, the transmitted using the transmit weight vectors
[1 0 . . . 0].sup.T, [0 1 0 . . . 0].sup.T, . . . , [0 0 . . .
1].sup.T. These vectors essentially cause individual sequences to
be transmitted exclusively on separate ones of the antennas, and
nevertheless produce a column vector of the channel response matrix
H at the receiving terminal. The CBF training sequence is appended
to the MPDU and transmitted to the destination communication device
in step 1020. The transmitting terminal uses the optimum transmit
weight vector when transmitting all other portions of the MPDU.
[0076] In step 1030, the destination device receives and decodes
the normal portion of the incoming MPDU using a matched filter
derived using the long preamble at the beginning of the incoming
burst to determine the optimum phase and gain relationships on each
receive antenna. Also, in step 1040, the destination device updates
the transmit weight vector to use when transmitting to the source
device (including the ACK to the incoming MPDU, for example) using
the channel response matrix H derived from the CBF training
sequence.
[0077] For example, suppose there are three antennas at the AP and
two antennas at the STA. The CBF training sequence that the AP
sends to the STA is transmitted using the transmit weight vectors
[1 0 0].sup.T, [0 1 0].sup.T and [0 0 1].sup.T. The channel
response H vector between these two devices is a 2.times.3 matrix
defined as [h.sub.11 h.sub.12].sup.T, [h.sub.21 h.sub.22].sup.T and
[h.sub.31 h.sub.32].sup.T. When these transmit weight vectors are
applied to the symbol s and transmitted, the result is s[h.sub.11
h.sub.12].sup.T, s[h.sub.21 h.sub.22].sup.T and s[h.sub.31
h.sub.32].sup.T. Therefore, the column vectors [h.sub.11
h.sub.12].sup.T, [h.sub.21 h.sub.22].sup.T and [h.sub.31
h.sub.32].sup.T of the H matrix can be computed by dividing each
receive vector ([r.sub.11 r.sub.12].sup.T, [r.sub.21
r.sub.22].sup.T and [r.sub.31 r.sub.32].sup.T, the receive output
of the antennas at the STA) by s since the transmit symbol s is
known at the STA because the STA will know the symbols used by the
AP for the training sequence.
[0078] Using the method described above, a communication device may
store the optimum transmit weight vectors for each of the other
communication devices it communicates with. For example, an AP
maintains a table mapping the MAC address for each STA to the
optimum Tx weight vector for that STA. CBF-capable STAs may also
store a table of such information when supporting communication in
a peer-peer or ad-hoc network. All transmit weight vectors may be
initially set to [1 0 . . . 0].sup.T.
[0079] For a 4-CBF scheme (4 antennas at the AP) using 1500 byte
packets at 54 Mbps, the loss in throughput for the above approach
is approximately 8%. The loss in throughput could be made smaller
using the following enhancements: one symbol long preambles instead
of 2 in the training sequence; use the channel response training
sequence only when it is needed; and/or transmitting the training
sequence during the IEEE 802.11 SIFS interval.
[0080] The training sequence scheme described above can be applied
to generate frequency dependent antenna weights. Steps 1010 through
1030 are repeated for each of a plurality of frequencies. For
example, in the multi-carrier signal case, steps 1010 through 1040
are repeated K times, for each sub-carrier frequency. Similarly,
for a single carrier modulation scheme, the training sequence would
be applied for each of a plurality of frequency sub-bands that span
the bandwidth of the baseband signal to be transmitted. In
addition, the transmit weights can be frequency shaped so that the
sum of the power across all of the antennas at a given frequency is
constant.
[0081] The antenna processing techniques described herein can be
incorporated into devices in a variety of ways. For example, an RF
chip can be built that supports 2 Tx/Rx antenna ports, and one
baseband chip that supports 2.times. to 4.times.CBF. One RF chip
together with one baseband chip can be used in a network interface
card, and two RF chips together with one baseband chip can be used
in an AP for a system that supports 4-CBF at an AP, and 2-CBF in a
STA. This system will perform up to 12 dB better than current
state-of-the-art system.
[0082] From simulations for 2-antenna selection diversity in an
indoor office environment w/50 ns RMS delay spread, 8 dB (4 dB) SNR
is required for 802.11a (802.11b) at the lowest data rate.
Including 6 dB of additional path loss for 802.11a at 5 GHz, a
total of 6+8-4=10 dB of additional received signal power is
required for 802.11a. For a path loss coefficient of 3.3 (indoor
environment), 10 dB of additional signal power corresponds to 1/2
the range.
[0083] In addition, the antenna processing schemes described herein
help reduce the performance degradation caused by interference. It
has been shown through simulations that the interference immunity
for a CBF-enhanced 802.11b network is approximately 2.2 times that
of a non-CBF network. In other words, a CBF enhanced communication
between two devices permits an interference source to be 2.2 times
close to a receiving device without degrading reception performance
at that device.
[0084] To again summarize, the antenna processing techniques
described above provide up to a 14 dB (25.times.) SNR improvement
over existing 802.11a/b implementations without requiring a change
to the communication protocol or standard. Moreover, compared to
current 2-antenna implementations, these techniques provide nearly
three times more range per AP; 7.3 times more coverage area; four
times less infrastructure cost at a fixed throughput per user; 7-10
times less infrastructure cost when optimized for coverage; 5 times
more throughput per user at a fixed infrastructure cost; normalized
and improved range for dual-mode 802.11a/b networks; and better
interference immunity and higher data rates. As much as 10 times
fewer APs are required to support a similar coverage area when
CBF-enhanced APs are used.
[0085] To summarize, a method is provided that accomplishes
communication between a first communication device and a second
communication device using radio frequency (RF) communication
techniques, comprising steps of applying a transmit antenna vector
to a baseband signal to be transmitted from the first communication
device to the second communication device, the transmit antenna
weight vector comprising a complex transmit antenna weight for each
of the N plurality of antennas, wherein each complex transmit
antenna weight has a magnitude and a phase whose values may vary
with frequency across a bandwidth of the baseband signal, thereby
generating N transmit signals each of which is weighted across the
bandwidth of the baseband signal; receiving at the N plurality of
antennas of the first communication device a signal that was
transmitted by the second communication device; determining a
receive weight vector comprising a plurality of complex receive
antenna weights for the N plurality of antennas of the first
communication device from one or more signals received by the N
plurality of antennas from the second communication device, wherein
each receive antenna weight has a magnitude and a phase whose
values may vary with frequency; and updating the transmit weight
vector for the plurality of antennas of the first communication
device for transmitting signals to the second communication device
by computing a conjugate of the receive weight vector of the first
communication device divided by a norm of the conjugate of the
receive weight vector. This same method may be embodied in the form
of instructions encoded on a medium or in a communication
device.
[0086] Also provided is a method that accomplishes communication
between a first communication device and a second communication
device, comprising steps of transmitting a first signal by one of N
plurality of antennas of the first communication device; receiving
a first response signal at the plurality of antennas of the first
communication device transmitted from a first of two antennas of
the second communication device; deriving a first row of a channel
response matrix that describes the channel response between the
first communication device and the second communication device;
transmitting a second signal by the plurality of antennas of the
first communication device using a transmit weight vector that is
orthogonal to the first row of the channel response matrix;
receiving a second response signal transmitted by a second of the
two antennas of the second communication device and deriving
therefrom a second row of the channel response matrix; and
selecting one of the first and second rows of the channel response
matrix that provides better signal-to-noise at the second
communication device as the transmit weight vector for further
transmission of signals to the second communication device. This
same method may be embodied in the form of instructions encoded on
a medium or in a communication device.
[0087] Still further provided is a method that accomplishes
communication between first and second communication devices
comprising steps of generating a training sequence comprising a
sequence of N consecutive symbols, where N is a number of antennas
of the first communication device, and the N symbols are multiplied
by respective ones of N linearly independent vectors that span
columns of a channel response matrix between the plurality of
antennas of the first communication device and a plurality of
antennas of the second communication device, thereby producing N
transmit signals; transmitting the N transmit signals from the
plurality of antennas of the first communication device; receiving
the N transmit signals at each of a plurality of antennas at the
second communication device; at the second communication device,
deriving from signals received by the plurality of antennas the
channel response matrix between the first communication device and
the second communication device; and at the second communication
device, generating a transmit weight vector from the channel
response matrix for transmitting a signal from the second
communication device to the first communication device using the
plurality of antennas of the second communication device. This same
method may be embodied in the form of instructions encoded on a
medium or in a communication device.
[0088] The above description is intended by way of example
only.
* * * * *