U.S. patent application number 12/207447 was filed with the patent office on 2009-03-12 for common mode transmission line termination.
This patent application is currently assigned to Robert D. Washburn. Invention is credited to Robert F. McClanahan, ROBERT D. WASHBURN.
Application Number | 20090067614 12/207447 |
Document ID | / |
Family ID | 34278683 |
Filed Date | 2009-03-12 |
United States Patent
Application |
20090067614 |
Kind Code |
A1 |
WASHBURN; ROBERT D. ; et
al. |
March 12, 2009 |
COMMON MODE TRANSMISSION LINE TERMINATION
Abstract
The present invention provides termination for transmission line
structures propagating common mode signals. Common mode signals
typically represent noise in systems wherein information is
transmitted as differential mode signals. The present invention
terminates the common mode signals in a dynamically matched
termination that prevents or significantly reduces reflection of
said common signals without interference with differential mode
transmission lines or their normal operation. Application is shown
for an unshielded, twisted pair transmission line as commonly used
in telephony-based systems for both voice and broadband data
communication. The methods for application of the present invention
to systems with large numbers of conductors are also shown.
Inventors: |
WASHBURN; ROBERT D.;
(Malibu, CA) ; McClanahan; Robert F.; (Valencia,
CA) |
Correspondence
Address: |
DLA PIPER US LLP
1999 AVENUE OF THE STARS, SUITE 400
LOS ANGELES
CA
90067-6023
US
|
Assignee: |
Washburn; Robert D.
Malibu
CA
McClanahan; Robert F.
Valencia
CA
|
Family ID: |
34278683 |
Appl. No.: |
12/207447 |
Filed: |
September 9, 2008 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
10933825 |
Sep 2, 2004 |
7430291 |
|
|
12207447 |
|
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|
|
60499824 |
Sep 3, 2003 |
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Current U.S.
Class: |
379/398 |
Current CPC
Class: |
H04L 25/0298
20130101 |
Class at
Publication: |
379/398 |
International
Class: |
H04M 9/00 20060101
H04M009/00 |
Claims
1.-3. (canceled)
4. An electronic circuit to provide termination for a plurality of
common mode signals propagating in a transmission line structure,
comprising: an active circuit; a transformer structure to
substantially block a plurality of differential mode signals and
transmit a plurality of common mode signals; a plurality of
variable components to optimize said plurality of common mode
signals and terminate a plurality of common mode transmission line
structures; a plurality of DSP controlled variable components;
means for matching a common mode termination of said plurality of
common mode transmission line structures over a communication
system operating frequency band; and means for balancing each of a
plurality of common mode transmission line terminations among a
plurality of common mode signal conductors within said plurality of
common mode transmission line structures.
5. The electronic circuit of claim 4 provides a complex impedance
matched termination for each of said plurality of common mode
transmission line structures.
6. The electronic circuit of claim 4 wherein said plurality of
common mode signal conductors is more than 2.
Description
CROSS-REFERENCE TO RELATED APPLICATION(S)
[0001] The present application claims the benefit of priority from
pending U.S. Provisional Patent Application No. 60/499,824,
entitled "Common Mode Transmission Line Termination", filed on Sep.
3, 2003, which is herein incorporated by reference in its
entirety.
BACKGROUND OF THE INVENTION
[0002] 1. Field of the Invention
[0003] The present invention relates to the field of dynamic,
impedance matched, transmission line terminations for common mode
signals.
[0004] 2. Background Art
[0005] Transmission lines play an important role in many fields of
electronics and are particularly important in communications.
Transmission line properties result from the geometrical
relationships among conductors and ground structures and the
properties of the conducting and insulating materials that form
them. As a result, transmission lines are realized in various forms
such as coaxial cables, twisted pairs of lines (both shielded and
unshielded), microstrip and stripline structures. The quality of a
transmission lines (the extent to which its performance approaches
that of an ideal transmission line) can vary considerably and the
choice is heavily influenced by the requirements of the specific
application.
[0006] Discussion of the present invention will focus on an
application with unshielded, twisted pair lines in a broadband
communication application (digital subscriber line or DSL). This
application reflects use of a "poor" quality line that is highly
susceptible to both pick-up and sourcing of common mode noise
signals. Of course these same twisted pair lines are "good" quality
for their original design application--as a telephone line to
transmit voice communication signals that are limited to under 4
KHz.
[0007] Referring to FIG. 1A, a typical transmission line is
symbolically illustrated. Transmission line TL100 has its source or
input port between nodes N101 and N102 and its terminal or output
port between nodes N103 and N104. TL100 is fully symmetric with
input and output ports. Voltage source V100 generates the input
signal, which may be of arbitrary form. ZSOURCE is the termination
impedance for the input port of TL100 and includes the source
resistance of V100. ZLOAD is the termination impedance for the
output port of TL100. The principal or defining electrical
characteristic of a transmission line is its characteristic
impedance (Z). For TL100 representing an ideal (lossless)
transmission line with characteristic impedance Z, under the
"matched" condition where ZSOURCE and ZLOAD are equal to Z, an
input signal generated V100 will propagate undistorted to ZLOAD.
Under this condition, half of the signal power will be dissipated
in ZLOAD and half in ZSOURCE.
[0008] For the condition where ZLOAD does not equal Z, a portion of
the propagating signal would be reflected back to the source where
a portion of the reflected signal would be further reflected back
toward the load if ZSOURCE also does not equal Z. FIG. 1B
illustrates the condition in which ZLOAD is an open circuit,
resulting in total reflection of the propagating signal power back
to the source.
[0009] Clearly, any physically realizable transmission line will
not be ideal and cannot be lossless. The propagating signal will
therefore be attenuated and distorted as a function of frequency to
at least some degree. Generally, these are characterized or modeled
on a per-unit-length basis, but will clearly become more
significant with increasing transmission line length.
[0010] The above discussion represents a very brief,
non-mathematical summary of classical transmission line theory.
What is not discussed in classical transmission line theory is the
fact that the transmission line system illustrated in FIG. 1A,
designed for transmission of differential mode signals produced by
source V100, also represents a transmission line system for common
mode signals that are picked up by the conductors. Since common
mode signals are easily converted to differential mode and will be
partially converted whenever they encounter an imbalance in
impedance-to-ground, they can represent a significant source of
signal distortion and noise, including crosstalk, to the desired
signal propagation through transmission line TL100.
[0011] The common mode transmission line is comprised of the same
physical structure as that for propagating the differential signals
from source V100 to termination impedance ZLOAD. However, it
represents one or more different type of transmission line
structure with different electrical characteristics including
characteristic impedance, signal velocity of propagation, and
frequency characteristics. In some ways it can appear to function
as a transmission line composed of a single conductor proximate to
a ground-plane. In other ways, it can appear to function as two of
these types of lines in parallel. Creation of common mode
transmission line models or mathematical descriptions of their
operation for any of the numerous transmission line topologies is
not the purpose of this teaching. The most significant fact is that
no "matched" termination is provided the common mode, allowing
these signals to reflect and bounce around until converted to
harmful differential noise by circuit non-linearity or impedance
imbalance. It would therefore be highly desirable to provide a
matched transmission line termination for common mode signals that
would not interfere with the normal operation of the transmission
line in propagating differential mode signals.
[0012] FIG. 1C illustrates the generation, coupling, and
transmission line propagation of common mode signals. Differential
signal source V100 and impedance ZSOURCE are not shown to simplify
the drawing. Source V101 is a ground referenced differential signal
source. Source V101 would typically be one of many which would be
distributed along the length of transmission line TL100. Sources
injecting common mode at points other than terminal ports of
transmission line TL100 cause common mode signals to propagate in
both directions in said transmission line, implying the
desirability of terminating common mode at both termination ports
of transmission line TL100 (and even at any significant
discontinuities existing between said ports). Source V101 typically
would represent a noise source including signals generated and
conducted on adjacent transmission lines, a noise source either
internal or external to the system(s) of which transmission line
TL100 is a part, and even signal source V100 which can source both
differential and common mode signals into transmission line
TL100.
[0013] Impedances ZCMSOURCE101 and ZCMSOURCE102 are the source
impedance for the noise source to each line of transmission line
TL100. They include the internal source impedance of source V101
that is common to both source impedances. Any mismatch between
impedances ZCMSOURCE101 and ZCMSOURCE102 will result in
proportionate conversion of the common mode signal to differential
that will then propagate along transmission line TL100 with the
desired signal from source V100. Since impedances ZCMSOURCE101 and
ZCMSOURCE102 are not short or open circuits, they will provide some
common mode signal termination at the point of noise injection but
this is not likely to be even close to providing a match
condition.
[0014] Finally, it should be noted that common mode signals present
at nodes N103 and N104 produce no current flow or power dissipation
within differential transmission line termination ZLOAD. For common
mode signals, ZLOAD can be replaced with the open circuit shown in
FIG. 1B without impact on common mode signals, again illustrating
the need (in many applications) to provide a common mode
termination for transmission line TL100.
[0015] FIG. 1D further illustrates the desirability to provide a
true common mode, matched, transmission line termination rather
than simply mitigate any imbalance in parasitic impedance from each
line to ground. Such mitigation might be accomplished by adding
precision, matched, "low" value resistors from each line to ground
to swamp out the existent impedances. With this approach, the added
resistors will also load the differential signal and must be
accounted for in the overall circuit design. Mitigation might also
be accomplished by adding a resistor to one line or the other to
directly reduce but not eliminate the imbalance between the
parasitic line impedances to ground.
[0016] In FIG. 1D, impedances Z103A and Z104A represent said
parasitic impedances from each line to ground. ZN represents the
impedance of a circuit branch coupling nodes N106 and N107 within
the electronic circuitry using or associated with the output port
signal of transmission line TL100. Impedances Z103B and Z104B are
parasitic impedances coupling node N107 to nodes N103 and N104
respectively. Impedances Z103C and Z104C are parasitic impedances
coupling node N106 to nodes N103 and N104 respectively. Impedances
Z106A and Z107A respectively couple nodes N106 and N107 to ground.
Impedances Z106A and Z107A provide the ground reference for common
mode signal conversion and can represent parasitic capacitance, or
actual components in the load circuit, or even zero if either node
N106 or N107 is a ground connection. For conditions where there is
an imbalance between impedances Z103B and Z104B or equivalently
between impedances Z103C and Z104C (as there always will be to some
extent), common mode signals are partially converted to
differential signals that then appear across ZN, injecting noise
directly into the circuitry. Presence of a common mode line
termination coupled with proper blocking of common mode signals and
isolation of the differential line termination can reduce this type
of noise problem to relative insignificance.
SUMMARY OF THE INVENTION
[0017] The present invention is an electronic circuit that provides
termination for common mode signals propagating in transmission
line structures. The common mode transmission line termination
provides a dynamically matched termination that prevents or
significantly reduces reflection of the common signals without
significant interference with differential mode transmission lines
or their normal operation. The present invention provides the
ability to reduce conversion of common mode noise to differential
mode, thereby improving the signal-to-noise ratio of the
communication system. In one or more embodiments, the present
invention comprises of a passive circuit, and a transformer
structure that substantially blocks differential mode signals and
transmits common mode signals. In other embodiments, the present
invention comprises of variable components for optimizing the
common mode signal match for termination of transmission line
structures.
[0018] In another embodiment, the present invention comprises an
active circuit. In one or more embodiments, the present invention
provides complex impedance matched termination for common mode
transmission line structures. In one or more embodiments, the
present invention comprises of DSP controlled variable components.
In one or more embodiments, the present invention comprises means
for matching the common mode termination of the common mode
transmission line structure over the communication system operating
frequency band. In one or more embodiments, the present invention
provides common mode transmission line termination where said
transmission lines comprise three or more common mode signal
conductors. In one or more embodiments, the present invention
comprises means for balancing the individual common mode
transmission line terminations among the common mode signal
conductors within the common mode transmission line structure.
BRIEF DESCRIPTION OF THE DRAWINGS
[0019] FIG. 1A is a circuit diagram of a prior art doubly
terminated, transmission line.
[0020] FIG. 1B is a circuit diagram of the prior art transmission
line of FIG. 1A with unterminated (open circuit) terminal port.
[0021] FIG. 1C is a circuit diagram of the prior art transmission
line of FIG. 1A with an unterminated (open circuit) input port in
the presence of a common mode signal source.
[0022] FIG. 1D is a circuit diagram of the prior art transmission
line of FIG. 1A with a doubly unterminated (open circuit) input
port in the presence of a common mode signal source.
[0023] FIG. 1E is a block diagram according to one embodiment of
the present invention.
[0024] FIG. 1F is a circuit diagram of the present invention for a
twisted pair transmission line, according to one embodiment of the
present invention.
[0025] FIG. 1G is a circuit diagram of the present invention for a
twisted pair transmission line with variable resistance to ground
balance adjustment, according to another embodiment of the present
invention.
[0026] FIG. 2A is a block diagram of another embodiment of the
present invention.
[0027] FIG. 2B is a block diagram showing functionality added to
the embodiment of FIG. 2A providing independent fine adjustment for
termination of each conductor in common mode transmission line
structure.
[0028] FIG. 3A is a circuit diagram of a preferred embodiment of
the present invention for a twisted pair transmission line.
[0029] FIG. 3B is a circuit diagram of the implementation of the
added functionality in the circuit shown in FIG. 3A.
[0030] FIG. 3C is a circuit diagram of the circuit of FIG. 3B with
added frequency dependent fine adjustment for termination of each
conductor in common mode transmission line structure.
DETAILED DESCRIPTION OF THE INVENTION
[0031] The present invention is directed to transmission line
terminations for common mode signals. In the following description,
numerous specific details are set forth to provide a more thorough
description of embodiments of the invention. It will be apparent,
however, to one skilled in the art, that the invention may be
practiced without these specific details. In other instances, well
known features have not been described in detail so as not to
obscure the invention. Except as noted herein, common components
and connections, identified by common reference designators
function in like manner in each circuit.
[0032] The present invention enables common mode signals
propagating in transmission line structures to be terminated,
thereby preventing their reflection back down the line. Typically,
common mode noise in systems is ignored unless the amplitude
becomes so large as to potentially damage system components. This
occurs because information is typically transmitted in differential
mode and differential mode transmission line terminations do not
draw current flow and therefore power from the common mode
signal.
[0033] The major problem with this approach is that common mode
signals are partially converted to differential by every nonlinear
circuit structure encountered (active circuitry is filled with
semiconductor junctions with their non-linear I-V and capacitance
characteristics) or by every imbalance in the impedance to ground
encountered. Similarly, differential noise is also easily converted
in part to common mode in which form it can propagate on the common
mode transmission line structure and then be partially converted
back to differential mode and interfere with system performance.
Common mode noise can be a major source of crosstalk among
transmission lines that are placed in close proximity, and
providing termination for these signals can be a major contributor
to crosstalk reduction and improved system performance.
[0034] In the prior art figures shown (FIG. 1A-1D), the active
devices are bipolar junction transistors because their relatively
constant voltage drop across junctions allows for formation of
simple current sources with resistor control of the current values
that are required to form the current mirrors. Current mirrors are
particularly useful in applications where signals are substantially
in the form of currents rather than voltages. This is not to imply
that other devices cannot be utilized in addition to the mirrors.
Field Effect Transistors (FETs) can be added in series with the
transistor collectors and biased to further increase the impedance
of the push-pull circuitry described below. FETs can also function
as switches in the value control of the various variable passive
components used.
Passive Embodiment of the Present Invention
[0035] FIG. 1F illustrates a passive embodiment of the present
invention for providing common mode, transmission line termination
for unshielded, twisted pair lines. Said termination network is
comprised of capacitors C100 and C101, transformer T150, and
resistor R100 and is coupled to the twisted pair transmission line
at nodes N108 and N109.
[0036] Capacitor C100 couples node N108 to the dotted end of
winding 1 of transformer T150 at node N110. Capacitor C101 couples
node N109 to the undotted end of winding 2 of transformer T150 at
node N111. Capacitors C100 and C101 provide DC blocks to prevent dc
loading of the twisted pair transmission line. Capacitance values
for capacitors C100 and C101 are dependent on the lowest signal
frequency (either common mode or differential mode) to be
terminated. An alternate topology can position DC blocking
capacitors C100 and C101 before the nodes (nodes N108 and N109 in
FIG. 1F) where connections to the common mode and differential mode
terminations separate. This configuration is meaningful only if the
transmission line termination does not draw dc power from the
transmission line. Any other system requirements such as operating
bias point shifts, contact establishment and maintenance,
communication of housekeeping information, and abnormal operation
including connection termination must also be accommodated with the
configuration.
[0037] Transformer T150 serves to pass common mode signals to line
termination resistor R100 while blocking differential signals. It
functions in a similar method to a "common mode choke" but the
oppositional connection effectively makes it a "differential mode
choke". A desirable configuration is to also have a complementary
common mode choke before the termination for the differential mode
signals.
[0038] Resistor R100 provides the common mode transmission line
termination. A major problem associated with the passive embodiment
of FIG. 1F is the determination of the value of resistor R100 that
provides matched termination for the common mode transmission line.
This results primarily because the nature of the ground is ill
defined and the ground structure plays a major role in the
determination of the impedance of a transmission line. The presence
of a shield as part of the transmission line structure such as in
coaxial cables, twisted shielded pair lines, and stripline
structures will generally not help as the shield will just provide
an additional path for common mode signal propagation.
[0039] One approach is to first disconnect termination resistor
R100 from the circuit at node N112 and measure the common mode
signal present at said node. Transmission line termination resistor
R100 is then reconnected to the circuit at node N112 and its value
adjusted so that the common mode signal present at node N112 is 50%
of the value measured in the first step. This process is simplified
if resistor R100 is variable, either as a single resistor or a more
complex network configuration.
[0040] If a resistor termination (purely real in mathematical
terms) is inadequate in specific applications, it can be replaced
with a passive network of varying size and complexity, having
complex impedance characteristics. Another alternative is to
replace resistor R100 with an active circuit that automatically
performs the matching function. This type of active network is in
reality the active embodiment described below. The optimum
configuration of the present invention effectively adds a
"differential mode choke" to the transmission line before the
circuitry comprising the active embodiment but after the
transmission line splits (or is tapped) for coupling in the common
mode termination of the present invention.
[0041] The embodiment shown in FIG. 1F is adequate for most
applications wherein the common mode transmission lines typically
can be matched within a few percent. This is particularly true with
closely coupled transmission line conductors such as in twisted
pair lines. It should be noted that some imbalances are present
that can not only produce deviation from an ideal match, but can
also actually convert common mode to differential noise. These
include variations in the values of capacitors C100 and C101,
imbalance in windings, coupling, and parasitic elements within
transformer T150, and the tolerance variation of resistor R100. In
addition, it should be noted that the match is actually a match for
the average impedance of the two lines. These values will be very
close for twisted pair lines but not identical. This averaging of
line impedances becomes more significant in multi-conductor systems
and is discussed in detail below.
[0042] For some applications, common mode transmission line
impedance mismatches of even a few percent are inadequate. For said
applications, the circuit shown in FIG. 1G provides added
capability to improve the common mode termination match. Capacitors
C100 and C101, transformer T150, and resistor R100 are common to
the embodiment of FIG. 1G. Variable resistors R101 and R102 are
provided to balance the resistance to ground on the output of
transformer T150 at nodes N113 and N114. By proper adjustment while
observing the output of the differential signal line (with no
differential signal present) and minimizing the converted common
mode signal present, an improved match can be obtained. Resistors
R101 and R102 can be augmented or replaces with complex, frequency
dependent networks that allow improvement of the termination across
the frequency range of interest. Resistors R103 and R104 are low
value balance resistors that should have the same value and tight
tolerances.
Active Embodiment of the Present Invention
[0043] FIG. 2A is a functional block diagram of an active
embodiment of the present invention for providing common mode,
transmission line termination for unshielded, twisted pair
lines.
[0044] The specific application illustrated in FIG. 2A is for a
digital subscriber line (DSL) communications system application. In
DSL applications, crosstalk is a major performance limiting noise
source and is comprised primarily of signals picked up from other
unshielded twisted pair transmission lines present in the same
cable bundle as the twisted pair transmission line referred to in
FIG. 2A. Although the crosstalk noise will have been converted to
differential mode to actually limit DSL system performance (both
range and data rate), a major portion of crosstalk is originally
common mode and it is the ease in which portions are converted to
differential mode that makes common mode transmission line
termination an essential part of the control of crosstalk in DSL
systems.
[0045] FIG. 3A is a circuit diagram of a preferred embodiment 300
of the present invention 20 conforming to the functional block
diagram shown in FIG. 2A. In the following discussions, FIG. 2A and
the corresponding circuit elements of FIG. 3A that constitute the
specific function will be discuss jointly.
[0046] Referring to FIG. 2A, common mode choke 100, isolation
transformer 110, and 2 to 4 wire hybrid 120 are existent components
of the analog front end (AFE) circuitry and an integral part of DSL
modems. This is the case whether the DSL modem is at the consumer
or at the Customer Premises Equipment (CPE) end of the twisted pair
transmission line or on a multi-modem plug-in card at the central
office (CO). In present DSL modems, common mode choke 100 is used
to block RF signals picked up from external sources and only
functions at frequencies that are well above any DSL signal
transmission frequency band. This is done primarily to save surface
area on circuit boards, particularly those located in the CO.
Isolation transformer 110 transmits differential mode signals and
has a ratio of secondary to primary winding turns that is typically
within the range of 1:1 to 2:1. The 2-wire to 4-wire hybrid is a
resistive bridge type network that separates transmitted and
received signals. Separation depends on the mismatch in actual
resistor values.
[0047] The circuit operates by sourcing or sinking common mode
current signals into the transmission line. Said current signals
from high impedance sources combine with the common mode signals
present on said transmission line such that the amplitude of the
resulting signal is half of the common signal present on the line
with no signals being injected. Within the gain-bandwidth
capability of the amplifiers in the circuit, the network will match
the common mode transmission line characteristics across the
operating band of interest. This is aided by controllable, passive
circuitry that shapes the frequency response to reduce the gain,
bandwidth, slewrate, and drive power requirements for the internal
amplifiers.
[0048] Common mode sense 200 functions to detect the common mode
signal present on the transmission line in close proximity to the
actual common mode termination point (point where the termination
correction signals described in the previous paragraph are
injected). Common mode sense 200 comprises resistors R311 and R312
that are connected in series. The series combination of said
resistors couples the two transmission line conductors together at
nodes N301 and N302. Since the transmission line impedance will
vary from around 50 to 200 ohms, values for R311 and R312 will
typically range from a few thousand to several tens of thousand
ohms. Node N314 is the connection point for the series combination
of resistors R311 and R312. The voltage at node N314 with respect
to ground is a representation of the average common mode signal
present on the twisted pair transmission line. Resistors R311 and
R312 typically will have the same value with very tight tolerances
although it is possible to weigh the inputs from the 2 lines by
using different values.
[0049] Clamp 205 is design to protect the circuitry against large
transient voltages. Clamp 205 is comprised of 2 separate clamp
circuits with the first being faster and the second oriented to
specific protection of the input to buffer amplifier 210. The first
clamp circuit limits the voltage at node N314 to approximately a
diode drop above positive bias voltage +V301 at node N303 or below
-V302 at node N304. The second clamp circuit limits the input
voltage at the positive input to buffer amplifier U305 to the same
values through separate circuitry. Differing clamp voltages can be
provided to the 2 clamp circuits if available. Likewise differing
clamp voltage levels can be provided for the positive and negative
levels.
[0050] The first clamp circuit is composed of capacitor C318,
diodes D301 and D302, and resistors R309 and R310. Capacitor C318
is intended to provide much lower impedance to fast transients than
resistor R340 so that the first clamp circuit provides the majority
of the protection. Capacitor C318 couples common mode voltage
detection node N314 to the anode of diode D301 and the cathode of
diode D302 at node N313. The cathode of diode D301 is coupled to
positive bias voltage V301 at node N303 and the anode of diode D302
is coupled to negative bias voltage V302 at node N304. Diodes D301
and D302 provide the actual clamping function. Resistors R309 and
R310 are connected in parallel with diodes D301 and D302
respectively, and provide 2 separate circuit functions. First,
resistors R309 and R310 provide a discharge path for capacitor C318
after a transient voltage charges it. Second, resistors R309 and
R310 bias node N313 at approximately the same level as node N314
with no common mode present, thereby maintaining substantially all
of the available dynamic operating range of the common mode
termination.
[0051] The second clamp circuit is composed of resistor R340 and
diodes D303 and D304. Resistor R340 couples common mode sense 200
at node N314 to buffer amplifier 210 at node N315, the
non-inverting input node for operational amplifier U305. Resistor
R340 limits the current into second clamp circuit diodes D303 and
D304 and the transient rise time at node N315 such that most of the
transient energy flows through capacitor C318 and the first clamp
circuit. The anode of diode D303 and the cathode of diode D304 are
coupled to buffer 210 at node N313. The cathode of diode D301 is
coupled to positive bias voltage V301 at node N303 and the anode of
diode D302 is coupled to negative bias voltage V302 at node N304.
Diodes D303 and D304 provide the actual clamping function.
[0052] Buffer 210 is comprised of operational amplifier U305,
capacitors C310 and C311, and resistors R313 and R314. Buffer 210
provides isolation from the input signal detection and protection
circuitry and a low impedance signal source to drive amplifier 220,
the primary gain control amplifier in the termination circuitry,
and possibly A/D converter 285 as well. Operational amplifier U305
is configured for unity gain although it could be configured to
provide some gain if said second configuration were desirable in a
specific application. The output and inverting inputs of U305 are
coupled to the input of amplifier 220 at node N318.
[0053] Capacitors C310 and C311 are power supply bypass capacitors
for operational amplifier U305. Capacitor C310 couples the positive
bias at node N316 to ground, and capacitor C311 couples the
negative bias at node N317 to ground. Resistors R313 and R314,
together with capacitors C310 and C311 form low pass filters for
the respective positive and negative bias power supply inputs.
These low pass filters, working in conjunction with similar filters
on other operational amplifiers, significantly reduces noise
coupling and unwanted feedback through the distribution means of
the bias power supplies. Typical values for resistors R313 and R314
are 5 to 10 ohms.
[0054] Digital Signal Processing (DSP) 280 and A/D converter 285
provide the principal digital functions for embodiment 300 of the
common mode transmission line termination. DSP 280 analyzes and
adjusts the frequency dependent performance of embodiment 300
through performance of mathematical calculations such as complex
FETs and its control of the gain of amplifier 220 and adjustment of
the values of passive elements comprising CM Z curve shaping
adjuster 225. For DSL type applications and given that the
transmission line impedances would not change rapidly, DSP 280
would typically be the modem DSP.
[0055] A/D converter 285 provides DSP 280 with a digital
representation of the common mode signal present on the output
buffer 210. In this configuration node N201 and node N318 would be
the same. An alternative input for A/D converter 285 would be node
N319 within amplifier 220. This would reduce potential below band
interference such as ring signals and power line harmonic noise but
would also reduce the ability of DSP 280 to detect and evaluate
impacts of below band signals on the performance of line
termination embodiment 300 as well as other elements of the DSL
system. In this alternate configuration, node N201 and node N319
would be the same point in the circuit. A/D 285 provides embodiment
300 with feedback capability necessary for closed loop operation
and performance optimization. In many applications, A/D converter
285 would be an integral part of DSP 280 and not a separate
integrated circuit. Since application of DSP 280 and A/D converter
285 is straightforward for one skilled in the art, the digital
circuits associated components and connections have been omitted
from FIG. 3A.
[0056] Amplifier 220 is the primary gain control stage for common
mode transmission line termination embodiment 300, and is comprised
of operational amplifier U306, capacitors C312, C313, and C314, and
resistors R315, R316, R317, R318, R319 and R320. The value of
resistor R319 is variable, controlled by DSP 280, and used to set
the gain of amplifier 220. Measuring the "open circuit" common mode
signal present on the transmission lines and then injecting the
match correcting error signal to reduce the signal amplitude 50% is
basic operating mode of the present invention. DSP 280 controls
this process by control of the gain of amplifier 220 and CM Z curve
shaping adjuster 225 described below. Resistor R319 is the primary
controlled element in amplifier 220.
[0057] Capacitors C313 and C314 together with their respective:
resistors R315 and R316, function as both bypass capacitors and low
pass filters on the dc bias power inputs to U306. They function
identically to the similar circuitry described previously as part
of buffer 210. Capacitor C313 couples the positive bias power input
for U306 at node N321 to ground. Resistor R315 couples the positive
bias power input for U306 at node N321 to the positive bias power
supply at node N303. Capacitor C314 couples the negative bias power
input for U306 at node N322 to ground. Resistor R316 couples the
negative bias power input for U306 at node N322 to the negative
bias power supply at node N304.
[0058] Capacitor C312 with resistors R317 and R318 form a simple
high pass filter designed to significantly reduce the amplitude of
below band signals and noise, which typically can include the
primary AC power frequency and its harmonics, telephone ring
signals, low frequency radio transmissions, and crosstalk pickup
from other telephone transmission lines. The lower cutoff frequency
is selected to provide adequate attenuation of below band noise
with insignificant impact on the system passband and communication
signals. For a DSL system as illustrated in FIG. 2A, having
transmission band low end at 25 KHz, approximately a 10 KHz filter
3 dB cutoff frequency is adequate. A more complex, higher order
filter may be substituted where required to meet requirements for a
specific design or application.
[0059] Resistor R317 and capacitor C312 are connected in series and
the series combination couples the output of buffer 220 at node
N318 to resistor R318 and the non-inverting input of U306 at node
N319. Resistor R318 couples node N319 to ground, providing the
ground reference for the high pass filter.
[0060] The values of resistors R319 and R320 jointly set the gain
of operational amplifier U306 and thereby substantially the gain of
amplifier 220. Resistor R319 is variable with its value typically
controlled and selected by DSP 280. As such, resistor R319 may be
implemented in a variety of ways most of which greatly exceed a
single resistor in complexity. Implementations can range from a
single, uncontrolled, selected test resistor or adjustable port to
a switched resistor network including multiple resistors and FETs.
Resistors R319 and R320 are coupled to the inverting input of
operational amplifier U306 at node N320. Resistor R319 couples node
N320 to ground. Resistor R320 couples node N320 to the output of
operational amplifier U306 and the output from amplifier 220 at
node N323.
[0061] The output of amplifier 200 is coupled to the
interconnection point, node N200, for reference circuit legs of
upper and lower current mirrors, comprising current mirror 240 and
current mirror 250 respectively by network function CM Z curve
shaping adjuster 225. As implemented in embodiment 300, CM Z curve
shaping adjuster 225 is a tee network formed by capacitor C317 and
resistors R333, R334, and R335. Resistor R333 couples the output of
amplifier 220 at node N323 to the common tie point or Tee-node of
CM Z curve shaping adjuster 225 at node N342. Resistor R335 couples
Tee-node N342 to current mirror reference leg at node N200. The
current injected into or removed from node N200 through resistor
R335 is the "common mode match correction error signal". Resistors
R333 and R335 are shown as fixed valued in FIG. 3A but can be
variable depending on the specific application. Resistor R334 and
capacitor C317 are connected in parallel and couple Tee-node N342
to ground. Resistor R334 and capacitor C317 are variable devices
used to shape the frequency response of the error signal driving
the current mirror reference leg at node N200. The values of
resistor R334 and capacitor C317 are typically determined and set
by DSP 280. Each component within the Tee network CM Z curve
shaping adjuster 225 can be replaced by multi-component, complex
impedance networks including multiple, variable, DSP controlled
devices.
[0062] The common mode transmission line impedance match correction
signal is applied to each individual conductor by a high impedance,
AC coupled, push-pull drive circuit formed by portions of upper and
lower current mirror circuits 240 and 250 of FIG. 2A. The reference
currents for upper and lower current mirror circuits 240 and 250
are coupled to node N200 by resistors R200 and R201 respectively.
In the absence of injection or removal of current, representing the
"common mode match correction error signal", at node N200 through
Tee network CM Z curve shaping adjuster 225, reference currents
flowing through resistors R200 and R201 are equal.
[0063] The reference leg of upper current mirror 240 is composed of
PNP transistor Q305 and resistors R337 and R339. Resistor R337
couples the emitter of transistor Q305 at node N344 to positive
bias voltage V301 at node N303. Resistor R339 couples the base of
transistor Q305 at node N348 to positive bias voltage V301 at node
N303. The collector of Q305 at node N345 is coupled to resistor 200
opposite node N200 and the base of transistor Q305 at node N348.
The base of transistor Q305 at node N348 is directly coupled to the
base of the PNP mirror transistor in each push-pull drive circuit,
comprising transistors Q301 and Q303 of FIG. 3A.
[0064] The reference leg of lower current mirror 250 is similarly
composed of NPN transistor Q306 and resistors R336 and R338.
Resistor R338 couples the emitter of transistor Q306 at node N347
to negative bias voltage V302 at node N304. Resistor R336 couples
the base of transistor Q306 at node N343 to negative bias voltage
V302 at node N304. The collector of Q306 at node N346 is coupled to
resistor 200 opposite node N200 and the base of transistor Q306 at
node N343. The base of transistor Q306 at node N343 is directly
coupled to the base of the NPN mirror transistor in each push-pull
drive circuit, comprising transistors Q302 and Q304 of FIG. 3A.
[0065] When the common mode signal on the transmission line
conductors is positive with respect to ground, current is injected
into node N200 from Tee network CM Z curve shaping adjuster 225.
The injected current flows through resistor R201, raising the
voltage of node N200 with respect to ground, increasing the current
flowing in the reference leg of lower current mirror 250, and
reducing the current flowing through resistor R200 and the current
mirror reference leg of upper current mirror 240. Reference leg
currents are mirrored in the respective push-pull drive circuit leg
transistors, resulting in a decrease in the voltage at nodes N334
and N339. Current is then extracted from the transmission line
conductors through coupling capacitors C315 and C316 that couple
nodes N334 and N339 to the 2 transmission line conductors at nodes
N301 and N302 respectively. This reduces the common mode signal
present on the transmission line conductors and the resulting
sensed common mode voltage at node N314. Node N200 then moves lower
toward its neutral or matched impedance operating point in typical
closed-loop, feedback circuit operation. The push-pull drive
circuits are required to be high impedance in order to prevent them
from loading the transmission lines with their circuitry rather
than the injected match error correction signal current.
[0066] Capacitors C315 and C316 are not simply DC blocking
capacitors. They will typically have nominal values that are
identical to maintain balanced operation of the circuit. For most
applications, capacitors C315 and C316 should have tolerances that
are reasonably tight but commonly available such as +/-5%. Values
for capacitors C315 and C316 should be chosen to present low
impedance with minimal phase shift both at the bottom of and across
the system signal transmission band of interest, and high impedance
for low frequency AC power distribution frequencies including
harmonics, as well as other low frequency system signals such as
telephone ring signals.
[0067] Referring to FIG. 3A, it is apparent that each of the high
impedance, AC coupled, push-pull drive circuits is substantially
identical. As a result, only one such circuit and its operation
will be described in detail. Circuit components in the second
network corresponding to those in the first network simply will be
identified by circuit designations without node or component
connection information available in FIG. 3A.
[0068] The first push-pull drive circuit, driving capacitor C315,
is comprised of upper current mirror PNP transistor Q301, lower
current mirror NPN transistor Q302, resistors R329 and R330, and
diodes D309 and D310. Corresponding components in the second
push-pull drive circuit are PNP transistor Q303, NPN transistor
Q304, resistors R331 and R332, and diodes D311 and D312. The first
push-pull drive circuit also contains a feedback circuit comprised
of circuit functions low pass filter and clamp protection 245,
buffer 260, and amplifier 265 shown in FIG. 2A. The purpose,
structure, and functioning of said feedback circuit will be
discussed below. Corresponding circuit functions for the second
push-pull drive circuit are low pass filter and clamp protection
255, buffer 270, and amplifier 275 shown in FIG. 2A.
[0069] In the first push-pull drive circuit, resistor R329 couples
the emitter of transistor Q301 at node N332 to positive bias
voltage V301 at node N303 and mirrors the current in resistor R337.
The base of transistor Q301 is coupled to the base of transistor
Q305 at node N348. The emitter of transistor Q301 at node N333 is
coupled to the anode of diode D309. The cathode of diode D309 is
coupled to the anode of diode D310, capacitor C315 and resistor
R327 (the input signal source for the feedback loop referred to in
the preceding paragraph). The cathode of diode D310 is coupled to
the collector of transistor Q302 at node N335. The base of
transistor Q302 is coupled to the base of transistor Q306 at node
N343. The emitter of transistor Q302 at node N336 is coupled to
resistors R328 (the output current signal for the feedback loop
referred to in the preceding paragraph) and R330. Resistor R330
couples node N336 to negative bias voltage V302 at node N304 and
mirrors the current in resistor R338.
[0070] The previously referenced feedback loops associated with
each push-pull drive circuit have specific characteristics that
enable or perform several functions. These are to set the nominal
operating point for the coupling node near the midpoint of the
push-pull circuit operating range, sense the location of the
coupling node during match correction and, return it to its nominal
operating point very slowly (very long time constant), and provide
gain for the feedback loop such that the coupling, node can be
maintained in it nominal operating condition.
[0071] The feedback loop for the first push-pull circuit includes
operational amplifiers U302 and U304, diodes D307 and D308,
capacitors C305, C306, C307, C308, and C309, and resistors R321,
R322, R323, R324, R325, R326, R327 and R328. Corresponding
components in the feedback loop for the second push-pull circuit
are operational amplifiers U301 and U303, diodes D305 and D306,
capacitors C301, C302, C303, C304, and C319, and resistors R301,
R302, R303, R304, R305, R306, R308 and R307.
[0072] Referring to the feedback loop for the first push-pull
circuit, diodes D307 and D308, capacitor C309 and resistor R327
comprise the feedback loop input circuit, specifically low pass
filter and clamp protection 245 of FIG. 2A. The input signal for
the feedback loop is the voltage to ground at node N334, which is
coupled to the input of buffer 260 at node N331 through a low pass
filter comprising resistor R327 and capacitor C309. Resistor R327
couples node N334 to node N331. Capacitor C309 couples node N331 to
ground and should be a large value (with R327) to produce a very
long time constant and response period for the feedback loop. Diode
D307 couples node N331 (diode anode) to positive bias voltage V301
at node N303 (diode cathode). Diode D308 couples node N331 (diode
cathode) to negative bias voltage V302 at node N304 (diode anode).
Under normal operating conditions, diodes D307 and D308 are reverse
biased and only serve to protect the buffer by limiting potential
transients to approximately the bias power supplies levels.
[0073] Continuing to refer to the feedback loop for the first
push-pull circuit, buffer 260 is composed of operational amplifier
U304, capacitors C307 and C308, and resistors R325 and R326.
Operational amplifier U304 is configured as a unity gain buffer
amplifier. The input signal to U304 is coupled to the non-inverting
input at node N331. The inverting input and output of U304 are
coupled together and to the input resistor R323 of amplifier 265 at
node N328. Capacitor C307 couples the positive bias power input of
U304 at node N329 to ground and functions as a bypass capacitor.
Similarly, capacitor C307 couples the negative bias power input of
U304 at node N330 to ground and also functions as a bypass
capacitor. Resistors R325 and R326 respectively couple the positive
and negative bias power inputs of U304 at nodes N329 and N330 to
the respective positive and negative bias power supplies V301 and
V302 at node N303 and N304. Resistors R325, R326 with capacitors
C307, C308 and comparable components associated with other
integrated circuits significantly reduce noise coupling between
integrated circuits through the power busses.
[0074] Continuing to refer to the feedback loop for the first
push-pull circuit, amplifier 265 is composed of operational
amplifier U302, capacitors C305 and C306, and resistors R321, R322,
R323, R324 and R328. The input signal for amplifier 265 is the
output of buffer 260, which is coupled to resistor R323 at node
N328. Resistor R323 couples node N328 to the inverting input of
U302 at node N324. The non-inverting input of U302 is coupled to
ground. Resistor R324 couples node N328 to the output of U302 at
node N327. Together resistors R323 and R324 set the gain of
operational amplifier U302 and thereby amplifier 265. Capacitors
C305 and C306 with resistors R321 and R322 perform the same bypass
and filter functions for U302 as capacitors C307 and C308 and
resistors R325 and R326 do for U304 in buffer 260. Resistor R328
couples the output of U302 at node N327 to the emitter of
transistor Q302 at node N336. The current through resistor R328
constitutes the output signal of the feedback loop.
[0075] Continuing consideration of the earlier example where the
common mode voltage sensed at node N314 moved positive producing a
shift to lower voltage at node N334, thus resulting in sinking,
through capacitor C315, of common mode signal from transmission
line conductor at node N301. The lower voltage at node N334
resulted in a slow reduction of the voltages at nodes N331 and
N328. Amplifier 265 must then sources current from its output at
node N327 through resistor R324 into virtual ground node N324 and
then through resistor R323 into node N328, which is at a negative
voltage. Increased current flow through resistor R324 from node
N327 to node N324 requires the voltage at node N327 to rise thereby
causing additional current to flow through resistor R328 into node
N336. The added current injected into node N336 reduces the current
flowing through the collector of transistor Q302, which in turn
causes the voltage at node N334 to creep slowly higher toward a
midrange, nominal operating point. Meanwhile, with the reduction of
the common mode signal on the transmission line through capacitor
C315 producing a condition closer to matched condition, the
detected common mode signal at node N314 should decline and the
voltage at node N334 quickly rise. This in turn may overshoot the
mark and require the slow feedback loop to lower the nominal
voltage at node N334.
[0076] The above discussion demonstrates termination of common mode
signal transmission lines for a classic 2 wire differential signal
transmission line that one skilled in the art can readily extend to
other forms. As previously discussed, however, common mode signals
are easily generated by coupling of signals to conductors within
circuits and between conductors and partial conversion of
differential signals to common mode by impedance imbalances within
circuits and transmission lines. As a result, common mode signal
transmission lines can occur that are not readily thought of as
transmission lines- and such systems are frequently multi-conductor
systems. A ribbon cable or printed data buss on a circuit board are
two common examples.
[0077] In general, there are two approaches for termination of such
multi-conductor systems. The first is to incorporate a 2-wire
termination previously described between each pair of conductors.
This approach has the advantage of providing near matched
conditions for each line. The source of crosstalk noise is a
distributed phenomena occurring along the entire length of the line
where it is in proximity to other lines that are the sources of
crosstalk noise, including out-of-phase re-coupling of the signal
being transmitted on the line of interest. The distributed nature
of the crosstalk noise source makes its best representation to be
an infinite number of independent sources distributed along each
wire in proximity to the transmission line of interest (this being
similar to lumped parameter model representations of transmission
lines). Though the coupling process is primarily linear so that
superposition of the coupled noise signals is generally applicable,
an infinite number of sources imply an infinite number of
propagating signals that can only be handled in aggregate. The
present invention effectively performs the aggregating function for
the lines being terminated.
[0078] Having a two wire termination between each set of conductors
is particularly useful because the coupling coefficient between any
of said point noise sources and each point on any other line will
vary significantly due to variation in distance of each line from
said point noise source. However, this approach will quickly
require massive amounts of circuitry as the number of lines
increases. The passive embodiment, with potentially large
components such as transformer T150 in FIG. 1F, is often
impractical to implement for even 2-wire applications, much less
for example, a 64-bit parallel data buss structure. The active
embodiment with its ability to be largely integrated into a single
integrated circuit is more often preferred.
[0079] For multi-conductor systems with a "large" number of common
mode transmission line conductors, it is generally preferred not to
attempt to provide an absolute match with zero reflection of common
mode signals on each line, but provide improved matching conditions
that significantly reduces reflected common mode signals, providing
improved but not absolute matching. In this approach, the average
of common mode signals present on all or a group of lines are
detected and individual lines matched against this average rather
than the open circuit common mode signal present on each specific
line.
[0080] Referring to FIG. 2A and specific embodiment 300 illustrated
in FIG. 3A, only 2 changes are required to implement said averaging
approach. First, common mode sense 200 requires modification to
provide an input from each transmission line conductor. This simply
converts the series resistor network comprised of resistors R311
and R312 into a star configuration with resistors coupling node
N314 to each transmission line conductor. All resistors in the star
will typically have the same value although they can be varied to
weight the contribution of individual lines to the composite common
mode signal representation. Another alternative is to specifically
match only one pair of lines and inject the same correction signal
into the other lines open loop. This again improves overall
matching conditions compared to the present open circuit, while
reducing the amount of circuitry required for implementation.
[0081] The second change for each additional transmission line
conductor is the addition of a high impedance, push-pull circuit
for sourcing or sinking the common mode match correction error
signal. A measure or estimate of the complexity of the drive stages
(amount of circuitry) for actual matching of each pair versus
simply providing improved matching can be easily determined. For a
transmission line system of n conductors, the complexity factor for
actual matching is approximately 2.sup.n-n instead of n for the
improved matching approach. Thus for a 5 wire system, the ratio is
27/5 or 5.5 times the output drive circuitry to achieve actual
matching. For a 64-bit buss, the ratio is approximately 2.8823e17
times larger, a total impossibility. The simpler implementation for
a 64-bit bus would be large and complex, but at least feasible. An
alternate approach that will frequently provide the best practical
implementation for a common mode transmission line system having a
large number of conductors is to provide actual matching for small
groupings of the total, say 4 conductors per group. The average or
weighting of each group can then be adjusted by the DSP based on
the relative common mode signal level present in each group.
[0082] In the previous discussion of the passive embodiment shown
in FIG. 1G, it was pointed out that for some applications, common
mode transmission line termination matching within a few percent
was inadequate. As discussed, resistors R101 and R102 provide
capability for a higher precision match in a passive embodiment.
Similar capability can be provided to the embodiment of FIG. 2A.
FIG. 2B is a block diagram illustrating the added functionality
required to provide higher precision matching. The individual
mirrored currents in each push-pull circuit leg within current
mirror 250 are adjusted by control signals from DSP 280. DSP 280
determines the amount and direction by performing a correlation
between the common mode crosstalk provided by A/D 280 of FIG. 2A
and the converted common mode signal that is present on a sample of
the differential output signal provided through A/D 290. DSP 280
adjusts the current balance to simultaneously maintain a best match
condition for the common mode transmission line pair and minimize
the differential noise that is correlated with it.
[0083] FIG. 3B shows another embodiment of the present invention
implemented in the circuit of FIG. 3A. Resistors R330 and R332 are
made variable with values controlled by DSP 280. FIG. 3C shows an
embodiment with addition of variable bypass capacitors C330 and
C332 in parallel with resistors R330 and R332. The addition of
reactive components provides an additional means for matching both
complex impedance of the line and improved matching across the
operating frequency band. In real world implementation, more
complex networks would likely replace the simple capacitors C330
and C332 shown.
[0084] Thus, a common mode transmission line termination is
described in conjunction with one or more specific embodiments. The
invention is defined by the following claims and their full scope
of equivalents.
* * * * *