U.S. patent application number 12/034633 was filed with the patent office on 2008-08-28 for high dynamic range tranceiver for cognitive radio.
Invention is credited to Stuart Rumley, Haiyun Tang.
Application Number | 20080207136 12/034633 |
Document ID | / |
Family ID | 39706644 |
Filed Date | 2008-08-28 |
United States Patent
Application |
20080207136 |
Kind Code |
A1 |
Tang; Haiyun ; et
al. |
August 28, 2008 |
High Dynamic Range Tranceiver for Cognitive Radio
Abstract
Embodiments of cognitive radio technology can recover and
utilize underutilized portions of statically-allocated
radio-frequency spectrum. A plurality of sensing methods can be
employed. Transmission power control can be responsive to adjacent
channel measurements. Digital pre-distortion techniques can enhance
performance. Embodiments of a high DNR transceiver architecture can
be employed.
Inventors: |
Tang; Haiyun; (Saratoga,
CA) ; Rumley; Stuart; (Redwood City, CA) |
Correspondence
Address: |
WEST & ASSOCIATES, A PC
2815 MITCHELL DRIVE, SUITE 209
WALNUT CREEK
CA
94598
US
|
Family ID: |
39706644 |
Appl. No.: |
12/034633 |
Filed: |
February 20, 2008 |
Related U.S. Patent Documents
|
|
|
|
|
|
Application
Number |
Filing Date |
Patent Number |
|
|
60890801 |
Feb 20, 2007 |
|
|
|
Current U.S.
Class: |
455/73 |
Current CPC
Class: |
H04B 17/354 20150115;
H04W 16/14 20130101; H04B 17/309 20150115; H04W 72/085 20130101;
H04W 72/02 20130101; H04N 5/50 20130101; H04B 17/327 20150115 |
Class at
Publication: |
455/73 |
International
Class: |
H04B 1/38 20060101
H04B001/38 |
Claims
1. A system for radio-frequency communications comprising: a
transceiver; and, a baseband processor comprising a sensing
processor element, a transmit power control element, wherein the
transceiver is coupled with the baseband processor.
Description
PRIORITY
[0001] This application is related to and claims priority under 35
U.S.C. 119(e) to U.S. Provisional Patent Application No. 60/890,801
filed on Feb. 20, 2007 entitled "SYSTEM AND METHOD FOR COGNITIVE
RADIO" by Haiyun Tang the complete content of which is hereby
incorporated by reference.
BACKGROUND
[0002] 1. Field of the Invention
[0003] The inventions herein described relate to systems and
methods for cognitive radio.
[0004] 2. Description of the Related Art
Spectrum Utilization Problems
[0005] A recent study by the FCC Spectrum Task Force [United
States' Federal Communications Commission (FCC), "Report of the
spectrum efficiency working group," November 2002,
http://www.fcc.gov/sptf/files/IPWGFinalReport.pdf] found that while
the available spectrum becomes increasingly scarce, the assigned
spectrum is significantly underutilized. This imbalance between
spectrum scarcity and spectrum underutilization is especially
inappropriate in this Information Age, when a significant amount of
spectrum is needed to provide ubiquitous wireless broadband
connectivity, which is increasingly becoming an indispensable part
of everyday life.
[0006] Static spectrum allocation over time can also result in
spectrum fragmentation. With lack of an overall plan, spectrum
allocations in the US and other countries over the past several
decades can appear to be random.
[0007] Despite some efforts to serve best interests at the time,
this leads to significant spectrum fragmentation over time. The
problem is exacerbated at a global level due to a lack of
coordinated regional spectrum assignments. In order to operate
under such spectrum conditions, a device can benefit from
operational flexibility in frequency and/or band shape; such
properties can help to maximally exploit local spectrum
availability.
[0008] To address the above problems, an improved radio technology
is needed that is capable of dynamically sensing and locating
unused spectrum segments, and, communicating using these spectrum
segments while essentially not causing harmful interference to
designated users of the spectrum. Such a radio is generally
referred to as a cognitive radio, although strictly speaking, it
may perform only spectrum cognition functions and therefore can be
a subtype of a broad-sense cognitive radio [J. M. III, "Cognitive
radio for flexible mobile multimedia communications," Mobile
Networks and Applications, vol. 6, September 2001.] that learns and
reacts to its operating environment. Key aspects of a cognitive
radio can include: [0009] Sensing: a capability to identify used
and/or unused segments of spectrum. [0010] Flexibility: a
capability to change operating frequency and/or band shape; this
can be employed to fit into unused spectrum segments.
[0011] Non-interference: a capability to avoid causing harmful
interference to designated users of the spectrum.
[0012] Such a cognitive radio technology can improve spectrum
efficiency by dynamically exploiting underutilized spectrum, and,
can operate at any geographic region without prior knowledge about
local spectrum assignments. It has been an active research area
recently.
FCC Spectrum Reform Initiatives
[0013] FCC has been at the forefront of promoting new spectrum
sharing technologies. In April 2002, the FCC issued an amendment to
Part 15 rules that allows ultra-wideband (UWB) underlay in the
existing spectrum [FCC, "FCC first report and order: Revision of
part 15 of the commission's rules regarding ultra-wideband
transmission systems," ET Docket No. 98-153, April 2002]. In June
2002, the FCC established a Spectrum Policy Task Force (SPTF) whose
study on the current spectrum usage concluded that "many portions
of the radio spectrum are not in use for significant periods of
time, and that spectrum use of these `white spaces` (both temporal
and geographic) can be increased significantly". SPTF recommended
policy changes to facilitate "opportunistic or dynamic use of
existing bands." In December 2003, FCC issued the notice of
proposed rule making on "Facilitating Opportunities for Flexible,
Efficient and Reliable Spectrum Use Employing Cognitive Radio
Technologies" [FCC, "Facilitating opportunities for flexible,
efficient, and reliable spectrum use employing cognitive radio
technologies," ET Docket No. 03-108, December 2003] stating that
"by initiating this proceeding, we recognize the importance of new
cognitive radio technologies, which are likely to become more
prevalent over the next few years and which hold tremendous promise
in helping to facilitate more effective and efficient access to
spectrum."
[0014] While both UWB and cognitive radio are considered as
spectrum sharing technologies, their approaches to spectrum sharing
are substantially different. UWB is an underlay (below noise floor)
spectrum sharing technology, while cognitive radio is an overlay
(above noise floor) and interlay (between primary user signals)
spectrum sharing technology as shown in FIG. 1. Through sensing
combined with operational flexibility, a cognitive radio can
identify and make use of spectral "white spaces" between primary
user signals. Because a cognitive user signal resides in such
"white spaces", high signal transmission power can be permitted as
long as signal power leakage into primary user bands does not
embody harmful interference.
Broadcast TV Bands.
[0015] Exemplary broadcast TV bands are shown in Graph 200 of FIG.
2. Each TV channel is 6 MHz wide. Between 0 and 800 MHz, there are
a total of 67 TV channels (Channels 2 to 69 excluding Channel 37
which is reserved for radio astronomy). The NPRM [FCC, May 2004,
op. cit.] excludes certain channels for unlicensed use: Channels
2-4, which are used by TV peripheral devices, and Channels 52-69,
which are considered for future auction. Among the channels
remaining, Channels 5-6, 7-13, 21-36, and 38-51 are available for
unlicensed use in all areas. Unlicensed use in Channels 14-20 is
allowed only in areas where they are not used by public safety
agencies [FCC, May 2004, op. cit.].
[0016] It can be appreciated that Channels 52-69 are currently used
by TV broadcasters and it is not clear if/when they will be
vacated. There is significant interference in the lower channels
5-6 and 7-13. Based on these considerations, the spectrum segment
470-806 MHz covering TV channels 14-69 can be of particular
interest.
Spectrum Opportunity in the TV Bands
[0017] Spectrum opportunity can be a direct result of incumbent
system inefficiency. In TV bands, a signal from a TV tower can
cover an area with a radius of tens of kilometers. TV receivers can
be sensitive to interference such that TV cell planning may be very
conservative to ensure there is essentially no co-channel
interference. This can leave a substantial amount of "white spaces"
between co-channel TV cells as illustrated in the the Map 300 of
FIG. 3. Those "white spaces" can constitute an opportunistic region
for cognitive users on a particular TV channel. Each TV channel may
have a differently shaped opportunistic region. The total spectrum
opportunity at any location can comprise the total number of
opportunistic regions covering the location. A measurement in one
locality shows an average spectrum opportunity in TV channels 14-69
of about 28 channels; that can be expressed as an equivalent
bandwidth of approximately 170 MHz.
BRIEF DESCRIPTION OF THE DRAWINGS
[0018] FIG. 1 graph of spectrum sharing technologies: UWB and
cognitive radio
[0019] FIG. 2 graph of exemplary television channel bands
[0020] FIG. 3 map of television co-channel coverage areas and
opportunistic region
[0021] FIG. 4 diagram: cognitive radio system
[0022] FIG. 5 diagram: amplification stages between antenna and
ADC
[0023] FIG. 6 diagram: heterodyne receiver
[0024] FIG. 7 diagram: heterodyne transceiver
[0025] FIG. 8 diagram: wideband direct-conversion receiver
[0026] FIG. 9 graph: frequency-domain non-linear effect
[0027] FIG. 10 diagram: double-ADC receiver architecture
[0028] FIG. 11 diagram: double-ADC receiver architecture,
detail
[0029] FIG. 12 graph: image problem, image rejection filter
[0030] FIG. 13 graph: solution for LO freq. with specified IF freq.
140 MHz
[0031] FIG. 14 graph: solution for LO freq. with specified IF freq.
70 MHz
[0032] FIG. 15 graph: solution for LO freq. with specified IF freq.
140 MHz and specified rejection margin
[0033] FIG. 16 graph: example SAW filter response
[0034] FIG. 17 graph: example SAW filter rejection mask
[0035] FIG. 18 graph: RF gain requirements
[0036] FIG. 19 diagram: heterodyne receiver, single-channel
[0037] FIG. 20 diagram: wideband direct-conversion transmitter
[0038] FIG. 21 graph: DTV transmission mask
[0039] FIG. 22 Diagram: wideband direct-conversion transmitter,
detail
[0040] FIG. 23 graph: simulated signal spectra for specified device
non-linearities.
DETAILED DESCRIPTION
[0041] FIG. 4 depicts an embodiment of a cognitive radio system in
block diagram. A transceiver 401 can be coupled with and/or in
communication with one or more antennae 402. Baseband signal
processing can be provided by elements of a baseband processor 403.
Elements of a baseband processor 403 can comprise a sensing
processor 404, a transmit power control element 405, and a
pre-distortion element 406. In some embodiments a pre-distortion
element 406 can be coupled with and/or in communication with a
transceiver 401. In some embodiments a transmit power control
element can be coupled with and/or in communication with a
transceiver 401. In some embodiments a collective sensing element
407 can be coupled with and/or in communication with a baseband
processor 403 and/or elements comprising a baseband processor.
[0042] In some embodiments transceiver 401 can comprise transceiver
and/or transmitter and/or receiver mechanisms disclosed herein. In
some embodiments sensing element 404 can comprise one or more
sensing mechanisms as described herein. By way of example and not
limitation these sensing mechanisms can include energy sensing,
NTSC signal sensing, and/or ATSC signal sensing. In some
embodiments a collective sensing element 407 can provide collective
sensing mechanisms as described herein.
[0043] In some embodiments transmit power control 405 can support
adaptive transmit power control mechanisms described herein. In
some embodiments pre-distortion element 406 can provide digital
pre-distortion mechanisms as described herein.
[0044] In some embodiments baseband processor 403 can support
additional processing mechanisms as described herein. By way of
example and not limitation these mechanisms can include filtering
and/or reconstruction.
RF System Analysis
Input Signal Dynamic Range
[0045] The diagram 200 of FIG. 2 depicts an embodiment of a
channel-based signal transmission scheme. Each of the channel
signals in an embodiment can be considered to be independent.
Hence, the total signal power over all channels considered (for
example, TV Channels 14-69) can be computed as the sum of the
individual signal powers of those channels.
[0046] Considering the wideband signal over all the channels in an
embodiment comprising TV channels, a total signal bandwidth can be
336 MHz and an antenna thermal noise floor over the signal
bandwidth can be calculated:
N 0 dB = 10 log 10 ( kTB ) = - 174 + 10 log 10 ( 336 .times. 10 6 )
.apprxeq. - 89 dBm ( 1 ) ##EQU00001##
[0047] In some embodiments, a maximum measured signal power can be
approximately -20 dBm.
[0048] For an individual TV channel in an embodiment, a thermal
noise floor can be
n.sub.0.sup.dB=-174+10 log.sub.10(6.times.10.sup.6).apprxeq.106 dBm
(2)
[0049] In some embodiments, a maximum single-channel power can have
a value of approximately -20 dBm. In an embodiment of a cognitive
radio system that operates close to the noise floor, a receiver can
see a channel power disparity of
-20-(-106+6).apprxeq.80 dB (3)
assuming a receiver noise figure of 6 dB.
Third-Order Intermodulation
[0050] In an ideal RF receive chain, all RF components can be
perfectly linear and there is no distortion on the received signal
after the signal has been processed by the RF receive chain.
Real-world RF components--especially active RF components like
amplifiers and mixers--can exhibit some degree of nonlinearity,
resulting in signal distortion. Small-signal nonlinearity of a
single RF component or cascaded RF components can be modeled by the
following input-output relationship
y(t)=.alpha..sub.0+.alpha..sub.1x(t)+.alpha..sub.2x.sup.2(t)+.alpha..sub-
.3x.sup.3(t)+ (4)
where x(t) is the input signal and y(t) is the output signal and in
some typical embodiments the nonlinearity can be dominated by the
low-order nonlinear terms.
[0051] RF components typically operate on passband signals. For
passband signals, even-order nonlinear terms can be discarded when
appropriate filtering is performed on the RF chain. The small
signal nonlinearity can then be approximated as:
y(t).apprxeq..alpha..sub.1x(t)+.alpha..sub.3x(t) (5)
retaining only the lowest odd order distortion term.
[0052] When a passband signal with baseband equivalent
representation s.sub.B(t) passes through an element with nonlinear
transfer function (3), the baseband equivalent representation of
the output signal can be expressed as
.alpha. 1 s B ( t ) Signal + 3 .alpha. 3 4 s B ( t ) 2 s B ( t ) 3
rd - order distortion ( 6 ) ##EQU00002##
[0053] At the output, the ratio of the distortion power to the
signal power, which is also the inverse of the dynamic range, can
be expressed as:
P DR - 1 = ( 3 .alpha. 3 4 ) 2 E [ s B ( t ) 6 ] .alpha. 1 2 E [ s
B ( t ) 2 ] = ( 3 .alpha. 3 4 ) 2 E [ s B ( t ) 6 ] .alpha. 1 2 E [
s B ( t ) 2 ] = 1 .alpha. 1 2 ( 3 .alpha. 3 4 ) 2 { E [ s B ( t ) 2
] } 2 .GAMMA. where ( 7 ) .GAMMA. = E [ s B ( t ) 6 ] { E [ s B ( t
) 2 ] } 3 ( 8 ) ##EQU00003##
is a factor that depends essentially only on the signal structure
of s.sub.B(t). For example, .GAMMA. is approximately 7.5 dB if
s.sub.B(t) is white noise. Suppose s.sub.B(t) is a combined signal
over all TV channels with power
P.sub.In=E[|s.sub.B(t)|.sup.2] (9)
[0054] The gain can be defined
g=.alpha..sub.1.sup.2 (10)
and output signal power
P.sub.Signal=gP.sub.In (11)
[0055] Using a two-tone IP3 relationship
1 .alpha. 1 2 ( 3 .alpha. 3 4 ) 2 = g 2 P IP 3 2 ( 12 )
##EQU00004##
[0056] It can be appreciated that a third-order intercept point
(IP3 or TOI) is the point at which a linear extrapolation (as a
function of input power) of linear output power and third-order
distortion power level meet.
[0057] Thus
P DR - 1 = g 2 P IP 3 2 P In 2 .GAMMA. = P Signal 2 P IP 3 2
.GAMMA. ( 13 ) ##EQU00005##
or in dB scale
P DR dB = 2 P IP 3 dB - 2 P Signal dB - .GAMMA. dB ( 14 )
##EQU00006##
Since the output 3rd-order distortion power is
P IM 3 = ( 3 .alpha. 3 4 ) 2 E [ s B ( t ) 6 ] ( 15 )
##EQU00007##
then, according to Equations (7) and (14)
P IM 3 dB = P Signal dB - P DR dB = 3 P Signal dB - 2 P IP 3 dB +
.GAMMA. dB ( 16 ) ##EQU00008##
[0058] Note that the term .GAMMA. in Equation (16) accounts for
added distortion that can result from a particular signal
structure. When an input signal s.sub.B(t) is essentially a
sinusoid (i.e. a single tone in frequency domain),
.GAMMA..sup.dB=0.
Overview of RF Receiver Functions
[0059] The functions of a RF receiver system can comprise: a)
Frequency translation and channel selection; and b) Signal
amplification.
Direct RF Sampling
[0060] An RF signal can reside in a particular frequency band
[f.sub.c-W,f.sub.c+W]
where f.sub.c is a carrier frequency and 2 W is a signal bandwidth.
In order to retrieve information content from the signal, the
signal can be digitized.
[0061] In theory, it is possible to directly sample the RF signal
at a carrier frequency. Such an approach, however, can be
prohibitively expensive in terms of hardware cost and power
consumption. For example, if a carrier frequency is 600 MHz, direct
Nyquist sampling of an associated RF signal can require a sampling
frequency at least 2(f.sub.c+W) or 1.2 GHz. In some embodiments an
overall RF signal can contain both strong and weak signal contents,
e.g. both TV signals and cognitive radio signals. A high-resolution
ADC can be advantageously specified for some such embodiments. By
way of non-limiting example, for a power difference between the
strong and weak signals of 70 dB, an ADC with a resolution of at
least 12 bits can be specified in some typical embodiments. Such
ADC requirements can present realization challenges, given that
some embodiments of current commercial ADCs can run at about 1 GHz
sampling frequency, with 8-bit resolution [National Semiconductor
Corporation, "ADC081000 High Performance, Low Power 8-bit, 1 GSPS
A/D Converter", DS200681, 2004], [Maxim Integrated Products,
"MAX108 Data Sheet: .+-.5V, 1.5 Gsps, 8-Bit ADC with On-Chip 2.2
GHz Track/Hold Amplifier", 19-1492; Rev 1; 10/01]. Direct RF
sampling embodiments may become a increasingly advantageous in the
future, as ADC and related technologies evolve.
Frequency Translation and Channel Selection
[0062] The high cost of RF direct sampling can be a result of the
sampling of unnecessary signal contents below f.sub.c-W. Given an
information bandwidth of 2 W, Nyquist sampling only requires a
sampling frequency of 2 W in the circumstance that the signal
center frequency can be shifted from the carrier frequency f.sub.c
to DC, i.e.
[f.sub.c-W,f.sub.c+W].fwdarw.[-W,W] (17)
[0063] Such frequency translation can typically be achieved in an
RF receiver through mixing. In addition to performing frequency
translation, a receiver can also perform channel selection in order
to acquire a signal in the desired 2 W-wide information band.
Signal Amplification
[0064] Another major function of an RF receiver can be signal
amplification. Consider an 8-bit ADC receiving an input signal with
peak-to-peak voltage of 600 mV [Nat'l Semi. Corp., DS200681, 2004,
op. cit.]. An associated quantization step can be 2.34 mV. The
quantization noise power assuming a 50-Ohm load can be
expressed
N q dB = 10 log 10 [ 2 .times. ( 2.34 .times. 10 - 3 ) 2 12 10 3 50
] .apprxeq. - 47 dBm ( 18 ) ##EQU00009##
where a factor of 2 results from considering the total quantization
noise power of the in-phase (I) and quadrature (Q) ADCs in the
system.
[0065] In some embodiments a received signal power level at the
antenna can be small, e.g. close to the exemplary thermal noise
level of -89 dBm in Equation (1). As illustrated in diagram 500
(FIG. 5), significant amplification through multiple amplification
stages along the RF chain can be provided in some embodiments to
ensure that a signal has enough power to overcome a quantization
noise floor when the signal reaches an ADC input. In some
embodiments, a specification can be employed to ensure that
quantization noise has a negligible impact on the system
performance; require that at the ADC input, the total thermal noise
(amplified thermal noise plus RF chain noise figure) is at least
X.sup.dB (e.g. 10 dB) above the quantization noise level. This
specification can translate into a requirement on the total RF
chain power gain g.sub.RF:
g.sub.RF.sup.dB-89 dBm+F.sub.RF.sup.dB.gtoreq.-47 dBm+X.sup.dB
(19)
where F.sub.RF is the RF chain noise figure. Alternatively, this
relationship can be expressed
g.sub.RF.sup.dB.gtoreq.42-F.sub.RF.sup.dB+X.sup.dB (20)
[0066] As an example, consider a receiver with a noise figure of 6
dB and X.sup.dB=10 dB. The total gain provided by the RF chain
needs to be at least 46 dB according to the above equation.
Accomplishing this gain can be a non-trivial task.
Receiver Architecture Choices Based on Channel Selection
Considerations
[0067] Since each exemplary 6 MHz TV channel can carry dissimilar
information content, in some embodiments channel selection can be
employed to decode the information content of a particular channel,
such as a TV channel. Channel selection can be performed at one or
more of an RF stage, IF stage, analog baseband, digital baseband,
and/or a combination of these stages.
RF Channel Selection:
[0068] In one design scenario, a channel selection filter can be
disposed in the RF stage immediately following the antenna in order
to select the desired channel. Several problems can attend this
approach. First, a high quality channel selection filter can
present challenges to realization at specified RF frequencies. A
quality metric for a filter can be defined as approximately its
3-dB bandwidth divided by its center frequency. For a specified
fixed channel width, a corresponding quality metric value increases
with increasing frequency. Hence, challenges to realizing such a
filter can increase with frequency. In some embodiments a receiver
can be specified to select any one of 55 TV channels from an
exemplary TV band. Thus in some embodiments, a tunable RF channel
selection filter can be employed, thereby further exacerbating
realization challenges. In some application embodiments, a
capability of simultaneous decoding multiple (eg., TV) channels can
be specified. In some such embodiments a complete RF chain after a
RF channel selection filter could be replicated for each additional
channel, and can thereby increase cost and/or complexity of a
realizable embodiment.
Heterodyne Receiver:
[0069] Diagram 600 depicts a block diagram embodiment of a
heterodyne receiver.
[0070] Channel selection in some embodiments of a conventional
heterodyne receiver can be achieved through a combination of
filtering stages along a RF (radio frequency) chain, which are
herein described:
[0071] An RF filter 604, also called a band selection filter. In
some embodiments this can be an RF frequency filter connected
directly to and/or coupled with an antenna 602. An RF filter 604
can select a frequency band of interest, such as an entire
exemplary TV band, and can reject signals outside the frequency
band of interest, e.g. 900 MHz cellular signals.
[0072] An Image rejection (IR) filter 612. In some embodiments this
filter can be disposed prior to a RF mixer 614 in order to reject
one or more image signals. In some embodiments an image signal can
otherwise fold into a desired signal band after mixing [B. Razavi,
RF Microelectronics. Pearson-Prentice Hall, 1998].
[0073] An IF filter 616, also called a channel selection filter. In
some embodiments this filter can be primarily responsible for
channel selection. In some embodiments an IF filter 616 can be
realized as a standalone component, e.g. a surface acoustic wave
(SAW) filter [C. Marshall and et al., "2.7v GSM transceiver ICs
with on-chip filtering," ISSCC Digest of Technical Papers, pp.
148-149, February 1995].
[0074] One or more baseband filters 624 634, also called
anti-aliasing filters. A baseband filter can be disposed prior to
an analog to digital converter (ADC) in order to reject alias
signals that can result from sampling. Diagram 600 depicts baseband
filter 624 employed in combination with ADC 628, and baseband
filter 634 employed in combination with ADC 638, corresponding
respectively to I and Q signal paths of a receiver embodiment.
[0075] In some embodiments, with the exception of a band selection
(RF) filter 604, each of the filters just described can provide a
degree of channel selection. In some embodiments a channel
selection (IF) filter 616 can be capable of providing the largest
contribution to selectivity. In some embodiments a heterodyne
receiver architecture can be relatively complex and/or costly if
multiple channels are to be decoded simultaneously. In some
embodiments, an RF chain comprising the elements after the IR
filter can be replicated for each additional channel in order to
support simultaneous decoding of multiple channels.
[0076] RF filter 604 can receive a signal from antenna 602. RF
filter 604 can provide a filtering function to a received signal.
Low noise amplifier LNA 610 can be coupled with and receive a
filtered signal from RF filter 601.
[0077] LNA 610 can provide a gain function with low noise to a
received signal. IR filter 612 can be coupled with and receive a
gain-modified signal from LNA 610. IR filter 612 can provide a
filtering function to a received signal. Oscillator LO.sub.1 608
can provide a signal that can be a tone signal at a specified
frequency. RF mixer 614 can be coupled with and receive a filtered
signal from IR filter 612. RF mixer 614 can be coupled with and
receive a signal that can be a tone signal at a specified frequency
from oscillator LO.sub.1 608. RF mixer 614 can provide a mixing
function, providing a signal responsive to a combination of a
signal received from IR filter 612 and a signal received from
oscillator LO.sub.1 608. IF filter 616 can be coupled with and
receive a signal from RF mixer 614. IF filter 616 can provide a
filtering function to a received signal. IF amp 618 can be coupled
with and receive a filtered signal from IF filter 616. IF amp 618
can provide a gain function to a received signal.
[0078] Oscillator LO.sub.2 609 can provide a signal that can be a
tone signal at a specified frequency. Quad splitter 623 can provide
a quadrature splitting function to a received signal, thereby
providing an in-phase (I) and a quadrature (Q) signal. Quad
splitter 623 can be coupled with and receive a signal from
Oscillator LO.sub.2 609. IF mixer 622 can be coupled with and
receive a signal of a first specified phase from Quad splitter 623.
IF mixer 622 can be coupled with and receive a gain-modified signal
from IF amp 618. IF mixer 622 can provide a mixing function,
providing a signal responsive to a signal received from Quad
splitter 623 and responsive to a signal received from IF amp 618.
Similarly, IF mixer 632 can provide a mixing function, providing a
signal responsive to a signal of a second specified phase received
from Quad splitter 623 and responsive to a signal received from IF
amp 618. Each of the baseband filters 624 634 can provide a
filtering function to a corresponding received signal. Baseband
filter 624 can be coupled with and receive a signal from IF mixer
622. Baseband filter 634 can be coupled with and receive a signal
from IF mixer 632. Each of the variable gain amplifiers (VGA) 626
636 can provide a variable gain to a corresponding received signal.
VGA 626 can be coupled with and receive a filtered signal from
baseband filter 624. VGA 636 can be coupled with and receive a
filtered signal from baseband filter 634.
[0079] Each of the analog to digital converters (ADC) 628 628 can
provide an analog to digital conversion function to a corresponding
received analog signal. ADC 628 can be coupled with and receive a
gain-modified signal from VGA 626. ADC 638 can be coupled with and
receive a gain-modified signal from VGA 636. ADC 628 can provide a
baseband digital output signal corresponding to the first specified
phase (I). ADC 638 can provide a baseband digital output signal
corresponding to the second specified phase (Q).
[0080] It can be appreciated that in alternative embodiments of a
heterodyne receiver 600 and in other receiver and transmitter
embodiments herein described, various gain elements can be omitted
and/or their functions realized by any known and/or convenient
method of providing signal gain.
Heterodyne Transceiver:
[0081] Diagram 700 depicts a block diagram embodiment of a
heterodyne transceiver. An upper portion of diagram 700 corresponds
directly to the heterodyne receiver 600 discussed herein. It can be
appreciated that upon coupling antenna 702 to the receiver
architecture through switch 706, there can be essentially a
one-to-one correspondence between elements of the receiver 600 and
elements of the receiver portion of the transceiver diagram
700.
[0082] The signal chain and function of the elements therein
correspond directly and respectively between [Antenna 602, RF
filter 604, LO.sub.1 608, LO.sub.2 609, LNA 610, IR filter 612, RF
mixer 614, IF filter 616, IF amp 618, IF mixer 622, Quad splitter
623, Baseband filter 624, VGA 626, ADC 628, IF mixer 632, Baseband
filter 634, VGA 636, ADC 638] and [Antenna 702, RF filter 704,
LO.sub.1 708, LO.sub.2 709, LNA 710, IR filter 712, RF mixer 714,
IF filter 716, IF amp 718, IF mixer 722, Quad splitter 723,
Baseband filter 724, VGA 726, ADC 728, IF mixer 732, Baseband
filter 734, VGA 736, ADC 738].
[0083] The receiver portion of diagram 700 further comprises a
Splitter 720 that couples elements with each other: IF amp 718, IF
mixer 722, and IF mixer 732. Corresponding elements IF amp 618, IF
mixer 622, and IF mixer 632 can be similarly coupled in the
embodiment of diagram 600.
[0084] In some embodiments the transmitter portion of diagram 700
can be advantageously realized using design analysis and/or
frequencies and/or element specifications and/or particular
elements in common with the receiver portion. In some embodiments
elements RF filter 704, LO.sub.1 708, and LO.sub.2 709 can be used
in common.
[0085] In some embodiments, elements of the transmitter [IR filter
712, RF mixer 714, IF filter 716, IF amp 718, Splitter 720, IF
mixer 722, Quad splitter 723, Baseband filter 724, VGA 726, IF
mixer 732, Baseband filter 734, VGA 736] can be substantially
similar to the corresponding and respective elements of the
receiver [IR filter 762, RF mixer 764, IF filter 766, IF amp 768,
Splitter 770, IF mixer 772, Quad splitter 773, Baseband filter 774,
VGA 776, IF mixer 782, Baseband filter 784, VGA 786].
[0086] Each of the digital to analog converters DAC 778 788 can
provide a digital to analog conversion function to a corresponding
received digital signal, thereby providing corresponding converted
corresponding analog signals. Baseband filters 774 784 can each
provide a filter function to a corresponding received signal.
Baseband filter 774 can be coupled with and receive an analog
signal from DAC 778. Baseband filter 784 can be coupled with and
receive an analog signal form DAC 788.
[0087] Oscillator LO.sub.2 709 can provide a signal that can be a
tone signal at a specified frequency. Quad splitter 773 can provide
a quadrature splitting function to a received signal, thereby
providing an in-phase (I) and a quadrature (Q) signal. Quad
splitter 773 can be coupled with and receive a signal from
Oscillator LO.sub.2 709. IF mixer 772 can be coupled with and
receive a signal of a first specified phase from Quad splitter 773.
IF mixer 772 can be coupled with and receive a filtered signal from
baseband filter 774. IF mixer 772 can provide a mixing function,
providing a signal responsive to a signal received from Quad
splitter 773 and responsive to a signal received from Baseband
filter 774. Similarly, IF mixer 782 can provide a mixing function,
providing a signal responsive to a signal of a second specified
phase received from Quad splitter 773 and responsive to a signal
received from Baseband filter 784.
[0088] Combiner 770 can provide a combining function, providing a
signal responsive to the combination of two received signals.
Combiner 770 can be coupled with and receive a signal corresponding
to a first specified phase from IF mixer 772. Combiner 770 can be
coupled with and receive a signal corresponding to a second
specified phase from IF mixer 782. IF amp 778 can provide a gain
function to a received signal. IF amp can be coupled with and
receive a combined signal from Combiner 770. IF filter can provide
a filter function to a received signal. IF filter can be coupled
with and receive a gain-modified signal from IF amp 778.
[0089] LO.sub.1 708 can provide a signal that can be a tone signal
at a specified frequency. RF mixer 764 can be coupled with and
receive a filtered signal from IF filter 766. RF mixer 764 can be
coupled with and receive a signal that can be a tone signal at a
specified frequency from LO.sub.1 708. RF mixer 764 can provide a
mixing function, providing a signal responsive to a combination of
a signal received from IF filter 766 and a signal received from
LO.sub.1 708. IR filter 762 can provide a filter function to a
received signal. IR filter 762 can be coupled with and receive a
mixed signal from RF mixer 764. PA 760 can provide a power
amplification function to a received signal. PA 760 can be coupled
with and receive a filtered signal from IR filter 762. RF filter
704 can provide a filter function to a received signal. RF filter
can be coupled with and receive a signal from PA 760 via Switch
706. Switch 706 can selectably couple PA 760 with RF filter 704.
Antenna 702 can provide an antenna transmission function to a power
amplified signal received from PA 760.
Wideband Direct-Conversion Receiver:
[0090] From the above discussion, it can be appreciated that as
long as the channel selection starts from a particular RF stage, in
some embodiments the RF chain from that stage onward can be
replicated for each additional channel. In some embodiments it can
be advantageous to defer channel selection all the way until the
digital baseband. Such an embodiment can comprise a receiver that
is capable of simultaneously decoding all of the channels in one or
more specified bands, such as all of the TV channels in depicted in
the graph 200. Two issues can be addressed in such a system.
[0091] First, there can be a need to have fast and high-resolution
sampling, because an ADC in such an embodiment sees an entire band
of interest, such as a TV band (Channels 14-69) with 336 MHz of
bandwidth.
[0092] Second, because before channel selection, the overall signal
consists of the signals from all the channels, some of which can be
strong while some of which can be weak, RF component nonlinearities
can cause signal intermodulations between one or more channels and
thus degrade system performance for the weak channels. Linearity
requirements on RF components constituting embodiments of such an
architecture can thus be relatively stringent, especially on
components disposed near to the ADC because such components can be
specified to operate on relatively high power signals and/or
amplified input signals.
[0093] Current technology trends of digital scaling along with
advances in high-speed ADCs can favor such an approach. An RF
system design for embodiments of such a wideband direct-conversion
receiver is herein described; diagram 800 depicts an embodiment.
Such an architecture may be considered wideband because the RF
receiver can operate on an entire band of interest, such as an
entire TV band of 336 MHz bandwidth. In some embodiments, a system
comprises a direct-conversion architecture wherein an RF signal can
be directly down-converted to a baseband.
[0094] RF filter 804 can receive a signal from antenna 802. RF
filter 804 can provide a filtering function to a received signal.
Low noise amplifier LNA 806 can be coupled with and receive a
filtered signal from RF filter 804. LNA 806 can provide a gain
function with low noise to a received signal.
[0095] Oscillator LO 810 can provide a signal that can be a tone
signal at a specified frequency. Quad splitter 808 can provide a
quadrature splitting function to a received signal, thereby
providing an in-phase (I) and a quadrature (Q) signal. Quad
splitter 808 can be coupled with and receive a signal from LO 810.
Mixer 820 can be coupled with and receive a signal of a first
specified phase from Quad splitter 808. Mixer 820 can be coupled
with and receive a gain-modified signal from LNA 806 . Mixer 820
can provide a mixing function, providing a signal responsive to a
signal received from Quad splitter 808 and responsive to a signal
received from LNA 806. Similarly, Mixer 830 can provide a mixing
function, providing a signal responsive to a signal of a second
specified phase received from Quad splitter 808 and responsive to a
signal received from LNA 806. Each of the Baseband filters 822 832
can provide a filtering function to a corresponding received
signal.
[0096] Baseband filter 822 can be coupled with and receive a signal
from Mixer 820. Baseband filter 830 can be coupled with and receive
a signal from Mixer 830. Each of the variable gain amplifiers (VGA)
824 834 can provide a variable gain to a corresponding received
signal. VGA 824 can be coupled with and receive a filtered signal
from Baseband filter 822. VGA 834 can be coupled with and receive a
filtered signal from baseband filter 832.
[0097] Each of the analog to digital converters (ADC) 826 836 can
provide an analog to digital conversion function to a corresponding
received analog signal. ADC 826 can be coupled with and receive a
gain-modified signal from VGA 824. ADC 638 can be coupled with and
receive a gain-modified signal from VGA 834. ADC 826 can provide a
baseband digital output signal corresponding to the first specified
phase (I). ADC 836 can provide a baseband digital output signal
corresponding to the second specified phase (Q).
Receiver Chain Frequency Planning
System Frequency Planning:
[0098] Referring to the TV band diagram 200 in FIG. 2, consider a
wideband direct-conversion receiver over the frequency range from
470 MHz to 806 MHz that can span TV channels 14-69. Since Channel
37 (608-614 MHz) is not used, the center frequency of Channel 37
can be employed as a direct-conversion carrier frequency, i.e.
f.sub.c=611 MHz (21)
[0099] A Nyquist bandwidth can be specified of
2 W=400 MHz (22)
covering the RF signal frequencies from 411 MHz to 811 MHz. A
number of alternative ADCs with 400 MHz sampling frequency and
above can be used in an embodiment [National Semiconductor
Corporation, ADC081000, 2004 op. cit.], [Maxim Integrated Products,
MAX108, 10/01, op. cit.], [Analog Devices, Inc. "AD12401 Data
Sheet, Rev A.", D05649-0-4/06(A), May 2006].
Frequency-Domain Effect of Second-Order Nonlinearity:
[0100] Referring to a signal path of the receiver block diagram
800: prior to the quadrature mixing stage comprising Mixer elements
820 830, there can typically be a plurality of amplification
stages, e.g. low noise amplifier (LNA) and/or amplification within
the mixers. In some embodiments, device nonlinearities in such
amplification stages can cause spectral contamination. In order to
ensure that frequency planning is adequate in the presence of such
spectral contamination, consider an RF signal
s.sub.c(t)=r(t)cos [2.pi.f.sub.ct+.theta.(t)] (23)
corresponding to a baseband signal
s.sub.B(t)=r(t)e.sup.j0(t) (24)
which is spectrally limited to [-W,W]. Taking into account device
nonlinearity, the signal after the amplification stages can be
expressed as
y ( t ) .apprxeq. .alpha. 0 + .alpha. 1 r ( t ) cos [ 2 .pi. f c t
+ .theta. ( t ) ] + .alpha. 2 r 2 ( t ) cos 2 [ 2 .pi. f c t +
.theta. ( t ) ] + .alpha. 3 r 3 ( t ) cos 3 [ 2 .pi. f c t +
.theta. ( t ) ] ( 25 ) ##EQU00010##
where under the small-signal condition, only the second-order and
third-order nonlinearities are retained. Third-order nonlinearity
is neglected since in-band third-order interference is inevitable.
However, to insure against in-band second-order interference,
consider the second-order nonlinearity term
.alpha. 2 r 2 ( t ) cos 2 [ 2 .pi. f c t + .theta. ( t ) ] =
.alpha. 2 r 2 ( t ) 1 4 { j [ 2 .pi. f c t + .theta. ( t ) ] + - j
[ 2 .pi. f c t + .theta. ( t ) ] } 2 = .alpha. 2 4 { [ s B ( t ) ]
2 j 2 .pi. ( 2 f c ) t Center = 2 f c + [ s B * ( t ) ] 2 j 2 .pi.
( - 2 f c ) t Center = - 2 f c + ce 2 s B ( t ) Center = D C 2 } (
26 ) ##EQU00011##
[0101] Consider Fourier transform pairs
s.sub.B(t)s.sub.B(t).revreaction.S.sub.B(f){circle around
(.times.)}S.sub.B(f)
s.sub.B*(t)s.sub.B*(t).revreaction.S.sub.B*(-f){circle around
(.times.)}S.sub.B*(-f)
s.sub.B(t)s.sub.B*(t).revreaction.S.sub.B(f){circle around
(.times.)}S.sub.B*(-f) (27)
and since S.sub.B(f) is spectrally limited to [-W,W], all the above
signal products (in the immediately preceding equations) can be
spectrally limited to [-2 W,2 W]. The graph 900 of FIG. 9
illustrates the above nonlinear effect. It is clear from the
illustration that as long as a carrier frequency f.sub.c
satisfies
f.sub.c.gtoreq.3 W (28)
a signal can be essentially free of second-order in-band
interference. In some embodiments this condition can be satisfied
by frequency planning, i.e.
611 MHz=f.sub.c>3 W=600 MHz (29)
Other Issues with Direct-Conversion Architecture:
[0102] Although some embodiments of a direct-conversion
architecture do not suffer an image problem as can some embodiments
of a heterodyne architecture, there can remain a number of
challenges to a practical implementation [B. Razavi, op. cit.]. In
some embodiments, LO self-mixing can create a DC offset. In some
embodiments, analog baseband circuitry can add considerable flicker
noise--also called 1/f noise, since noise power can be proportional
to 1/f. In some embodiments I/Q mismatch can occur if the I and Q
signal paths are not precisely balanced. Challenges of DC offset
and flicker noise--which can prominent around DC--can be addressed
in some embodiments of an improved receiver architecture by using
an empty 6 MHz signal channel, such as Channel 37 of an exemplary
TV band, at DC. In some embodiments, I/Q mismatch can be
compensated through digital calibration techniques.
Receiver Chain Gain Planning
[0103] In light of frequency planning as discussed above, an ADC
can be selected for an improved receiver embodiment. Consider using
National Semiconductor's ADC 081000, an 8-bit 1 GHz ADC [Nat'l
Semi. Corp., DS200681, 2004, op. cit.], as previously mentioned. A
receiver chain amplification calculation can be as discussed herein
regards Signal Amplification, and employed for each 6 MHz TV
channel. Assuming ADC operation at a 800 MHz sampling frequency, a
quantization noise per TV channel can be expressed
n.sub.q.sup.dB=N.sub.q.sup.dB-10 log.sub.10(800/6).apprxeq.-68 dBm
(30)
where N.sub.q is quantization noise power as calculated in Equation
(18). In order to scale noise contributions, RF chain gain g.sub.RF
can be specified such that thermal noise exceeds the quantization
noise at the ADC. In other words,
n.sub.0.sup.dB+F.sub.RF.sup.dB+g.sub.RF.sup.dB.gtoreq.n.sub.q.sup.dB+X.s-
up.dB (31)
Again a noise figure can be specified F.sub.RF.sup.dB=6 dB and a
margin X.sup.dB=10 dB so that
g.sub.RF.sup.dB.gtoreq.n.sub.q.sup.dB+4-n.sub.0.sup.dB=42 dB
(32)
[0104] An ADC can be operating at twice a specified sampling rate
of 400 MHz; this can account for the discrepancy between the result
shown here and that in discussion regards Signal Amplification.
[0105] In some embodiments a receiver chain can provides 42 dB of
amplification as just described. When operating with a maximum
received signal power of -20 dBm, an amplified signal at an ADC can
have a power level of
P.sub.Signal=22 dBm (33)
Amplified thermal noise at the ADC can have a power level of
-89+42+6=-41 dBm. In order to have third order intermodulation
(IM3) power remain below thermal noise power, according to Equation
(16), a required condition can be
3 P Signal dB - 2 P IP 3 dB + .GAMMA. dB = P IM 3 < - 41 P IP 3
dB > 1 2 ( 3 P Signal dB + .GAMMA. dB + 41 ) P IP 3 dB > 1 2
( 3 .times. 22 + 0 + 41 ) P IP 3 dB > 53.5 dBm ( 34 )
##EQU00012##
where for simplicity, it can be assumed that .GAMMA..sup.dB=0. Such
a high IP3 can be difficult to realize in an embodiment.
[0106] Another potentially complicating design consideration can be
that a specified ADC has an input digitizing range of input
(maximum) peak-to-peak 0.6V. A maximum input signal power for the I
and Q ADCs can be computed as
10 log 10 ( 2 .times. 0.3 2 50 .times. 10 3 ) = 5.6 dBm , ( 35 )
##EQU00013##
far smaller than the amplified signal power of 22 dBm.
A Novel Double-ADC Receiver Architecture
[0107] Diagram 1100 depicts an embodiment in some detail comprising
the double-ADC architecture of diagram 1000, and that can address
some issues discussed herein; particularly challenges to
realization of an embodiment. Some notable blocks are represented
in diagram 1000. An Amplification Stage 1 1004 can comprise an LNA
and/or optional additional amplifications. In one embodiment the
total gain provided by this stage can be 15 dB (after 1-to-2
splitting) and a receiver chain noise figure up to this point can
be 5 dB. Given an exemplary maximum receiver input signal power of
-20 dBm, signal power at the output of this amplification stage can
be -5 dBm.
[0108] Thermal noise power at the output of this amplification
stage can be -89+15+5=-69 dBm. In order to maintain an IM3 power
below the thermal noise floor, a specified IP3 of Amplification
Stage 1 must be larger than -5+-5-(-69)/2=27 dBm. In some
embodiments, a maximum component-wise IP3 in this amplification
stage can be somewhat higher than 27 dBm in order to take into
account losses through passive components, e.g. splitters and
filters, in the stage.
[0109] A signal arriving at analog to digital converter ADC1 1006
can be representative of an input signal received by antenna 1002.
A representative input signal can be expressed as
y ( t ) = k .di-elect cons. .OMEGA. x k ( t ) j 2 .pi. f k t + n (
t ) ( 36 ) ##EQU00014##
where x.sub.k(t) and f.sub.k are the baseband signal and frequency
of a kth channel respectively. The signal after ADC1 1006 sampling
can be expressed as
y ' ( t ) = k .di-elect cons. .OMEGA. x k ( t ) j 2 .pi. f k t + n
( t ) + q ( t ) = k .di-elect cons. .OMEGA. [ x k ( t ) + n k ( t )
+ q k ( t ) ] j 2 .pi. f k t .apprxeq. k .di-elect cons. .OMEGA. [
x k ( t ) + q k ( t ) ] j 2 .pi. f k t ( 37 ) ##EQU00015##
where q(t) is quantization noise; n.sub.k(t) and q.sub.k(t) are
baseband equivalent thermal noise and quantization noise on Channel
k; and in the approximation, thermal noise can be ignored because
thermal noise power per channel can be approximately -106+15+5=-86
dBm; this can be far smaller than quantization noise power per
channel, i.e. -68 dBm. The maximum input signal power to ADC1 1006,
i.e. EO|y(t)|.sup.2.right brkt-bot., can be approximately -20+15=-5
dBm, which can be smaller than a maximum allowable ADC input signal
power of 5.6 dBm.
[0110] In a baseband, digital filtering can be performed (by
Digital Filtering element 1008) to select one or more specified
channels. After filtering, a subset of the selected channels can be
selected .LAMBDA..OR right..OMEGA. whose SNRs exceed 25 dB. Element
Digital Filtering 1008 can be adapted to provide this capability. A
signal corresponding to the selected set of channels can be
expressed as
y .LAMBDA. ( t ) = k .di-elect cons. .LAMBDA. [ x k ( t ) + q k ( t
) ] j 2 .pi. f k t ( 38 ) ##EQU00016##
[0111] A signal y.sub..LAMBDA.(t) can be shown as y.sub.H(t) in
some figures herein; the "H" subscript indicating correspondence to
relatively high power channels of an input signal. Two operations
can be employed with this set of channels. First, this set of
channels can be sent to a digital baseband processing unit 1020 for
decoding, since they have adequate SNRs. Second, an analog waveform
can be reconstructed corresponding to the signal y.sub..LAMBDA.(t)
using a DAC 1010. A reconstructed analog waveform can be expressed
as
y .LAMBDA. ( t ) = k .di-elect cons. .LAMBDA. [ x k ( t ) + q k ( t
) ] j 2 .pi. f k t + p ( t ) ( 39 ) ##EQU00017##
where p(t) is quantization noise from the DAC 1010.
[0112] Subtracting a reconstructed waveform y.sub..LAMBDA.(t) from
an original signal y(t) can yield:
y ( t ) - y .LAMBDA. ( t ) = k .di-elect cons. ( .OMEGA. - .LAMBDA.
) x k ( t ) j 2 .pi. f k ( t ) - k .di-elect cons. .LAMBDA. q k ( t
) j 2 .pi. f k ( t ) - p ( t ) + n ( t ) ( 40 ) ##EQU00018##
and is depicted as a signal comprising y.sub.L(t) that can be
provided by summing node 1012 in FIG. 10, wherein y.sub.L(t)
corresponds to relatively low power channels of an input
signal.
[0113] Since the remaining channels belong to a set
.OMEGA.-.LAMBDA., and these channels can have signal powers less
than 25 dB above the an exemplary per channel quantization noise
floor of -68 dBm, a maximum signal power per channel can be
-68+25=-43 dBm. In an exemplary worst case, all of the channels can
have signal powers at -43 dBm and .OMEGA.-.LAMBDA. can comprise an
exemplary complete set of 55 TV channels. A worst-case power of the
signal y(t)-y.sub..LAMBDA.(t) then can be:
-43+10 log.sub.10(55).apprxeq.-25 dBm (41)
[0114] In order to provide a total RF chain amplification of 42 dB
with the first-stage amplification already providing 15 dB gain,
the second-stage amplification 1014 can be required to provide an
additional 27 dB gain. In the above worst case example, a signal
power at input of ADC2 1016 (after second-stage amplification 1014)
can be 2 dBm. Amplified thermal noise power at input of ADC2 1016
can be -89+42+6=-41 dBm. To maintain an IM3 below the thermal noise
floor, an IP3 of
2 + 2 - ( - 41 ) 2 = 23.5 dBm ##EQU00019##
for second amplification stage 1014 can be specified.
[0115] The summing node 1012 can provide a signal comprising
specified relatively low-power bands and/or channels of a
representative input signal but also comprising uncanceled residual
signal attributed to specified relatively high-power bands and/or
channels. Digital filtering 1018 can be adapted to substantially
remove undesirable energy corresponding to specified bands and/or
channels such as high-power channels corresponding to signal
y.sub.H(t). Digital filtering 1018 can provide an advantageously
filtered signal to digital baseband processing 1020. In some
embodiments digital baseband processing 1020 can further process
and/or decode such an advantageously filtered signal and can
provide one or more individual channel signals corresponding to
y.sub.L(t).
[0116] In order to prevent significant noise figure degradation,
quantization noise p(t) added by DAC 1010 can be kept small in
comparison to thermal noise n(t) in Equation (40). An exemplary DAC
can provide up to 16-bit resolution at 500 MHz with an output
peak-to-peak voltage swing of 1V. Examples of such DACs include
Analog Devices AD9726 [Analog Devices, Inc., "AD9726 Data Sheet,
Rev A", D04540-0-11/05(A), November 2005] and Maxim MAX5888 [Maxim
Integrated Products, "MAX5888 Data Sheet: 3.3V, 16-Bit, 500 Msps
High Dynamic Performance DAC with Differential LVDS Inputs",
19-2726; Rev 3; 12/03]. A quantization noise power for a 15-bit DAC
can be expressed
E [ p ( t ) 2 ] = 10 log 10 [ 2 .times. ( 1 / 2 15 ) 2 12 10 3 50 ]
.apprxeq. - 85 dBm , ( 42 ) ##EQU00020##
which is less than a specified thermal noise floor of -89+15+5=-69
dBm.
[0117] Diagram 1100 shows in some detail a RF block diagram of an
example direct-conversion double-ADC receiver. Many suitable
components for an exemplary embodiment are identified herein, by
way of non-limiting example.
[0118] The system of diagram 1100 comprises individual processing
elements well known in the art and/or described herein. Each of
these elements is generally identified herein with a name and/or
abbreviation that corresponds to its well known and/or herein
described function. Analog filters comprise BandPass 1108, LowPass1
1124 1174, and ReConstruction 1130 1180. Digital filtering and/or
other specified digital signal processing comprises Digital
Filtering 1127 1177. Gain modifying elements comprise low noise
amplifiers LNA1 1106 and LNA2 1110, automatic gain control AGC1
1122 1172 and AGC2 1134 1184. Analog to digital converters comprise
AD 1126 1176 1136 1186. Digital to analog converters comprise DA
1128 1178.
[0119] Splitters comprise elements 1112 and 1116. Mixers comprise
elements 1120 and 1170. Summing nodes comprise elements 1132 and
1182. Delay compensation elements comprise Delay Comp. 1125
1175.
[0120] Delay elements comprise Phase Shift 1118.
[0121] LNA1 1106 can be selectably coupled with Antenna 1102 via
switch 1104. When so coupled, LNA1 1106 can receive a signal from
Antenna 1102. BandPass 1108 can be coupled with and receive a
signal from LNA1 1106. LNA2 can be coupled with and receive a
signal from BandPass 1108.
[0122] Splitter 1112 can be coupled with and receive a signal from
LNA2 1110. Mixer 1120 can be coupled with and receive a signal from
Splitter 1112. Mixer 1120 can be coupled with and receive a signal
from Splitter 1116.
[0123] Mixer 1170 can be coupled with and receive a signal from
Splitter 1112. Mixer 1170 can be coupled with and receive a signal
from PhaseShift 1118. PhaseShift 1118 can be coupled with and
receive a signal from Splitter 1116. Splitter 1116 can be coupled
with and receive a signal from an oscillator LO 1114.
[0124] AGC1 1122 can be coupled with and receive a signal from
Mixer 1120.
[0125] LowPass1 1124 can be coupled with and receive a signal from
AGC1 1122.
[0126] Delay Comp. 1125 can be coupled with and receive a signal
from LowPass1 1124.
[0127] Summing node 1132 can be coupled with and receive a signal
from Delay Comp. 1125.
[0128] AD 1126 can be coupled with and receive a signal from
LowPass1 1124.
[0129] Digital Filtering 1127 can be coupled with and receive a
signal from AD 1126.
[0130] DA 1128 can be coupled with and receive a signal from
Digital Filtering 1127.
[0131] ReConstruction 1130 can be coupled with and receive a signal
from DA 1128.
[0132] Summing node 1132 can be coupled with and receive a signal
from ReConstruction 1130.
[0133] AGC2 1134 can be coupled with and receive a signal from
Summing node 1132.
[0134] AD 1136 can be coupled with and receive a signal from AGC2
1134.
[0135] AD 1136 can provide a baseband in-phase component
signal.
[0136] AGC1 1172 can be coupled with and receive a signal from
Mixer 1170.
[0137] LowPass1 1174 can be coupled with and receive a signal from
AGC1 1172.
[0138] Delay Comp. 1175 can be coupled with and receive a signal
from LowPass1 1174.
[0139] Summing node 1182 can be coupled with and receive a signal
from Delay Comp. 1175.
[0140] AD 1176 can be coupled with and receive a signal from
LowPass1 1174.
[0141] Digital Filtering 1177 can be coupled with and receive a
signal from AD 1176.
[0142] DA 1178 can be coupled with and receive a signal from
Digital Filtering 1177.
[0143] ReConstruction 1180 can be coupled with and receive a signal
from DA 1178.
[0144] Summing node 1182 can be coupled with and receive a signal
from ReConstruction 1180.
[0145] AGC2 1184 can be coupled with and receive a signal from
Summing node 1182.
[0146] AD 1186 can be coupled with and receive a signal from AGC2
1184.
[0147] AD 1186 can provide a baseband quadrature component
signal.
[0148] Exemplary digital-analog conversion devices can be
specified: National Semiconductor's ADC081000 [Nat'l Semi. Corp.,
DS200681, 2004, op. cit.], an 8-bit 1 GHz ADC, and Analog Devices'
AD9726 [Analog Devices, Inc., D04540-0-11/05(A), November 2005, op.
cit.], a 16-bit 600 MHz DAC. As shown in Diagram 1100, a first
amplification stage comprises LNAs, bandpass filters, splitters,
mixers, variable gain amplifiers, and lowpass filters, with a total
gain of 15 dB and a noise figure of approximately 5 dB. Exemplary
system components and a cascaded gain analysis are shown in the
following table.
[0149] ]Note that because of losses due to the passive components,
e.g. splitters and filters, in some embodiments one or more
amplifiers can be needed in a first amplification stage. In some
embodiments a second amplification stage can consist of variable
gain amplifiers. An IP3 calculation for a second amplification
stage can assume a maximum input signal power of -25 dBm, as
discussed herein.
TABLE-US-00001 Max. Output output Vendor: NF NF Gain power IP3 DR
Name Part (dB) (dB) (dB) (dBm) (dBm) (dB) LNA1 Mini- 3.5 3.5 12 -8
47 110 Circuits: HELA- 10B Bandpass TBD 3 3.6 -3 -11 .infin.
.infin. LNA2 Mini- 3.5 3.9 12 1 47 92 Circuits: HELA- 10B Splitter
Mini- 4 3.9 -4 -3 .infin. .infin. Circuits: ZFSC- 2-2 Mixer Mini- 8
4.1 -8 -11 30 82 Circuits: ZFY-2 AGC1 Linear 7 4.9 14 3 47 88 Tech:
LT5514 Lowpass1 TBD 8 4.9 -8 -5 .infin. .infin. AGC2 Linear 7 5.1
27 2 47 90 Tech: LT5514
[0150] The above discussions and analysis show a wideband
direct-conversion double-ADC receiver using exemplary hardware
components can provide enabling system performance levels for
embodiments of a TV-band cognitive radio system, and, can allow
simultaneous decoding of essentially all of the TV channels in a
designated spectrum.
[0151] A conventional single-channel heterodyne receiver can be
considered as a reference and a cost-effective alternative to the
embodiments above. A heterodyne receiver can use progressive
filtering in an analog domain in order to improve channel
selectivity. Although such a receiver may not have the capability
of simultaneous decoding of multiple channels, neither does it
require high-speed ADCs. It can also be instructive to compare the
single-channel performance of the heterodyne receiver with that of
the wideband receiver.
[0152] IP3 requirements for realizable embodiments of a double-ADC
architecture can be relatively stringent. In some embodiments, the
worst-case IM3 interference can be allowed to be higher than the
thermal noise floor.
[0153] Remaining interference can then be removed in a digital
domain through distortion compensation techniques.
A Reference Heterodyne Receiver Design
[0154] RF system design embodiments of a conventional
single-channel heterodyne receiver can serve as a reference point
and as an alternative to wideband direct-conversion receiver
embodiments discussed herein.
Heterodyne Frequency Planning
[0155] Frequency planning for a heterodyne receiver can present
further design challenges than that of a direct-conversion
receiver. For some embodiments of a heterodyne receiver, two
frequency translations can be required, i.e. from RF to IF and from
IF to baseband (although frequency translation between IF and
baseband can be achieved in some embodiments employing direct IF
sampling and/or digital frequency synthesis). One of the key design
issues of a heterodyne receiver embodiment can be specification of
an intermediate frequency (IF).
IF Filtering:
[0156] As discussed herein regards Receiver Architecture Choices, a
main purpose of an IF stage in a heterodyne receiver can be to
provide channel selection filtering, because effective filtering
can be more easily accomplished at a relatively low IF frequency
than at a relatively high RF frequency. Availability of
off-the-shelf IF filters can contribute to a practical selection
and/or specification of an IF frequency.
[0157] A surface acoustic wave (SAW) filter can be a typical choice
for IF channel selection. Some embodiments of exemplary
commercially available SAW filters can have specified center
frequencies of 40 MHz, 70 MHz, and 140 MHz [16,17].
Image Rejection:
[0158] Referring to Diagram 600 of FIG. 6: Mixer 614 can be a
second-order device, that is, a device that does not differentiate
between positive and negative frequencies. Consequently, after
mixing, a down-converted signal can contain both an intended signal
and an image signal as illustrated in Diagram 1200 of FIG. 12.
[0159] Mathematically, an intended signal can be represented as
R.sub.s(t)cos [2.pi.f.sub.ct+.phi..sub.s(t)] (43)
which can be band-limited to [f.sub.c-W,f.sub.c+W]; an image signal
can be represented as
R.sub.i(t)cos [2.pi.f.sub.it+.phi..sub.i(t)] (44)
and mixing can use a tone signal
cos(2.pi.f.sub.LOt)
[0160] A mixing operation can be expressed as
{ R s ( t ) cos [ 2 .pi. f c t + .phi. s ( t ) ] + R i ( t ) cos [
2 .pi. f i t + .phi. i ( t ) ] } .times. cos ( 2 .pi. f LO t ) = 1
2 R s ( t ) cos [ 2 .pi. ( f c - f LO ) t + .phi. s ( t ) ] 1 + 1 2
R s ( t ) cos [ 2 .pi. ( f c + f LO ) t + .phi. s ( t ) ] 2 + 1 2 R
i ( t ) cos [ 2 .pi. ( f i - f LO ) t + .phi. i ( t ) ] 3 + 1 2 R i
( t ) cos [ 2 .pi. ( f i + f LO ) t + .phi. i ( t ) ] 4 ( 45 )
##EQU00021##
[0161] A filtering operation
[f.sub.c-f.sub.LO-W,f.sub.c-f.sub.LO+W] can be applied to the
signal after mixing, whereupon the second and fourth term in the
above expression can essentially vanish. However, for an image
signal at
f.sub.i=2f.sub.LO-f.sub.c (46)
the third term above can become
1 2 R i ( t ) cos [ 2 .pi. ( f LO - f c ) t + .phi. i ( t ) ] = 1 2
R i ( t ) cos [ 2 .pi. ( f c - f LO ) t - .phi. i ( t ) ] ( 47 )
##EQU00022##
In other words, this signal can be in the same band, i.e.
[f.sub.c-f.sub.LO-W,f.sub.c-f.sub.LO+W], as an intended signal
after mixing (first term). One way to resolve the problem is to
apply an image rejection (IR) filter 1202 before mixing as shown in
the graph 1200 of FIG. 12 so that an image signal at
2f.sub.LO-f.sub.c can be rejected before a signal enters a
mixer.
Frequency Planning:
[0162] For some embodiments, an intended signal can be in a
specified band such as [470,806] MHz. An ideal image rejection
filter can be a brick-wall filter around a specified band. Suppose
such an ideal IR (image rejection) filter is used in an embodiment:
essentially full pass in [470,806] MHz and essentially infinite
rejection otherwise. f.sub.c, f.sub.LO, and f.sub.IF can be the
carrier, LO, and IF frequencies, respectively. To have image-free
mixing in some embodiments, the following conditions must be
essentially met
2f.sub.LO-f.sub.c<470 or 2f.sub.LO-f.sub.c>806 (48)
f.sub.c-f.sub.LO=+f.sub.IF or f.sub.c-f.sub.LO=-f.sub.IF (49)
Since 2f.sub.LO-f.sub.c is an image frequency, the first condition
above can suggest that the image frequency must stay in a rejection
band of an IR filter. The second condition can be expressed as
|f.sub.c-f.sub.LO|=f.sub.IF, where the absolute value is due to the
properties of a realizable signal mixer.
[0163] Given IF frequency candidates of 40 MHz, 70 MHz, and 140
MHz, the three possible IF frequencies can be substituted in the
above conditions and the systems solved for possible solutions.
Solutions can be advantageously perceived graphically, as shown in
graphs 1300, 1400, and 1500.
[0164] Graph 1300 corresponds to a condition (f.sub.IF=140 MHz).
Line A 1302 corresponds to (2f.sub.LO-f.sub.c=470). Line B 1304
corresponds to (2f.sub.LO-f.sub.c=806). Line C 1306 corresponds to
(f.sub.c-f.sub.LO=140). Line D 1308 corresponds to
(f.sub.c-f.sub.LO=-140).
[0165] A portion of line C 1306 shown in a region below line A 1302
(corresponding to (2f.sub.LO-f.sub.c<470)) can be part of a
solution, and, a portion of line D 1308 shown in a region above
line B 1304 (corresponding to (2f.sub.LO-f.sub.c>806)) can also
be part of a solution. By way of non-limiting example, f.sub.c=500
MHz is shown to be in Solution Region_1 1310 and with an
f.sub.LO=360 MHz, an image is thereby at 220 MHz and within a
rejection region of the IR filter. Since each solution region can
cover a part of the input signal frequency range (e.g. Solution
Region_1 1310 can cover 750 MHz and below and Solution Region_2
1312 can cover 543 MHz and above), both regions can be necessary
for an embodiment comprising an entire exemplary input frequency
range, i.e. [470,806] MHz. Thus the constraints of Graph 1300 can
lead to a practical realization for single-stage image-free IF
mixing in some embodiments.
[0166] Graph 1400 corresponds to a condition (f.sub.IF=70 MHz).
Line A 1402 corresponds to (2f.sub.LO-f.sub.c=470). Line B 1404
corresponds to (2f.sub.LO-f.sub.c=806). Line C 1406 corresponds to
(f.sub.c-f.sub.LO=70). Line D 1308 corresponds to
(f.sub.c-f.sub.LO=-70).
[0167] Graph 1400 shows a Gap 1414 between solution regions 1410
1412, corresponding to a region wherein image-free mixing can not
occur in some embodiments. For the constraints corresponding to
graph 1400, some embodiments employing 70 MHz IF filters for
single-stage image-free IF mixing can fail to provide a solution
for an entire exemplary TV band [470,806] MHz.
[0168] A similar analysis can show that some embodiments employing
40 MHz IF filters under such constraints can fail to provide a
solution covering an entire exemplary TV band [470,806] MHz.
[0169] The conditions for Graph 1300 and Graph 1400 correspond to
an ideal brick-wall IR filter over the signal band. In practice,
typical filter embodiments can have gradual edge roll-offs. Thus in
some embodiments margins can be employed at IR filter edges in
order to provide a specified level of image rejection. By way of
non-limiting example, a 100-MHz margin can be added to each side of
an IR filter in order to account for edge roll-offs. An image
rejection region can then be
2f.sub.LO-f.sub.c<370 and 2f.sub.LO-f.sub.c>906 (50)
[0170] Graph 1500 shows a solution for the conditions discussed.
Line A 1502 corresponds to (2f.sub.LO-f.sub.c=370). Line B 1504
corresponds to (2f.sub.LO-f.sub.c=906). Line C 1506 corresponds to
(f.sub.c-f.sub.LO=140). Line D 1508 corresponds to
(f.sub.c-f.sub.LO=-140).
[0171] An advantageous overlap between Solution Region_1 1510 and
Solution Region_2 1512 can be relatively smaller than the overlap
shown in Graph 1300. By way of non-limiting example, 650 MHz can be
a cutoff frequency. For exemplary TV channels 14 (center 473 MHz)
to 43 (center 647 MHz), an LO frequency can be
f.sub.LO=f.sub.c-140 (51)
and for exemplary TV channels 44 (center 653 MHz) to 69 (center 803
MHz), an LO frequency can be
f.sub.LO=f.sub.c+140 (52)
Gain Planning
[0172] Graph 1600 of FIG. 16 depicts the response of an exemplary
SAW filter [Vectron International, "Surface Acoustic Wave (SAW)
Products" http://www.vectron.com/products/saw/saw.htm]. The filter
has a specified passband of approximately 6 MHz. The specified
rejection for two 6 MHz channels adjacent to the pass band can be
specified as at least 15 dB (due to the finite roll-offs at filter
edges as shown in the figure). Specified rejection for the channels
not adjacent to the pass band can be at least 50 dB. The filter has
a specified insertion loss of 22.5 dB.
[0173] A SAW filter channel rejection mask as shown in FIG. 17 can
be assumed. A target channel k has 0 dB rejection. Rejection for
adjacent channels k.+-.1 is specified as 15 dB. Rejection for all
other channels is specified as 40 dB. Some embodiments of SAW
filters are able to essentially meet the specified requirements of
such a rejection mask.
[0174] As discussed regards Receiver Chain Gain Planning, a per
channel thermal noise floor n.sub.0.sup.dB of -106 dBm can be
specified, a per channel quantization noise floor n.sub.q.sup.dB of
-68 dBm can be specified, and a receiver chain noise figure
F.sub.thrmRF.sup.dB of 6 dB can be specified. A SNR degradation due
to the quantization noise can be required to be 0.46 dB,
corresponding to an X.sup.dB value of 10 dB. A total RF chain
amplification requirement can be obtained from the following SNR
equation
SNR Final dB = 10 log 10 g RF P k g RF n 0 F RF + n q = min { 30 dB
, P k dB - ( n 0 dB + F RF dB ) - 0.46 } ( 53 ) ##EQU00023##
where SNR.sub.Final.sup.dB is SNR measured at the baseband input;
g.sub.RF is total RF chain gain; and P.sub.k is input (received)
signal power of a target channel. It can be appreciated that the
SNR ceiling can be set to 30 dB in order to meet specified
performance levels. Graph 1800 of FIG. 18 depicts a graphical
solution to Equation (53). As shown in the figure, for high input
power levels the gain required can be reduced as a result of a SNR
ceiling at 30 dB.
[0175] A total input signal power can be expressed
P k + ( P k - 1 + P k + 1 ) + l .di-elect cons. .OMEGA. ' P l ( 54
) ##EQU00024##
where .OMEGA.' can be a whole channel set excluding channels k and
k.+-.1. Assuming a SAW filter rejection mask as shown in Diagram
1700, after SAW filtering, a total signal power can be
expressed:
P k + 10 - 15 10 ( P k - 1 + P k + 1 ) + 10 - 40 10 l .di-elect
cons. .OMEGA. ' P l ( 55 ) ##EQU00025##
and a total signal power at an ADC (after RF chain amplification)
can be expressed:
g RF [ P k + 10 - 15 10 ( P k - 1 + P k + 1 ) + 10 - 40 10 l
.di-elect cons. .OMEGA. ' P l ] ( 56 ) ##EQU00026##
[0176] A condition can be imposed that the signal powers of the two
adjacent channels satisfy
10 log 10 P k - 1 + P k + 1 P k < A dB ( 57 ) ##EQU00027##
where A.sup.dB can be a maximum specified adjacent channel power
differential, e.g. 40 dB. Without this condition, adjacent channel
leakage could overwhelm a signal in a desired channel (e.g.
referring to the DTV transmission mask in Diagram 2100). Assuming a
maximum total input signal power of -20 dBm, signal power at the
ADC can have an upper bound P.sub.Bound such that:
g RF [ P k + 10 - 15 10 ( P k - 1 + P k + 1 ) + 10 - 40 10 l
.di-elect cons. .OMEGA. ' P l ] < g RF [ P k + 10 - 15 10 AP k +
10 - 40 10 l .di-elect cons. .OMEGA. ' P l ] < g RF [ P k + 10 -
15 10 10 40 10 P k + 10 - 40 10 10 - 20 10 ] = P Bound ( 58 )
##EQU00028##
where in the second inequality A.sup.dB can be specified as 40 dB
and a constraint that .SIGMA..sub.l.epsilon..OMEGA.'P.sub.l is less
than -20 dBm can be employed. This upper bound is plotted in
Diagram 1800.
[0177] According to Diagram 1800, a maximum possible signal power
at an ADC can be less than -3 dBm. A thermal noise power, shown as
Final noise power in Diagram 1800, at this point can be -58 dBm. An
IP3 requirement for an amplifier in the signal chain just prior to
an ADC can then be expressed
- 3 + - 3 - ( - 58 ) 2 = 24.5 dBm ( 59 ) ##EQU00029##
[0178] In some embodiments, a 140 MHz IF signal can be
down-converted to a baseband using a conventional down-conversion
approach as shown in Diagram 600. Alternative embodiments can
employ direct IF sampling with digital down-conversion. In some
embodiments, ADCs with 400 MHz and/or greater sampling frequencies
[8,9,12] can be used to perform direct IF sampling.
[0179] In some embodiments, an LNA and a mixer can provide enough
gain to overcome a SAW filter insertion loss, which can have a
typical value of 20 dB. Exemplary low-loss SAW filters (with 10 dB
insertion loss) are available [Integrated Device Technology, Inc.,
"Saw Filter Products", http://www.idt.com/?id=3350]. Employing such
SAW filters in some embodiments can contribute to relaxing a
specified amplification requirement on an LNA and mixer. As shown
in Diagram 1800, an RF chain can be specified to provide an
adjustable gain range of 60 dB, i.e. from -20 dB to +40 dB. In some
embodiments an LNA and mixer can provide a switchable gain step of
20 dB. One or more amplifier(s) following a SAW filter can then
provide an adjustable gain of between 0 and 40 dB. This gain can be
combined with a 20 dB LNA-mixer gain step and can provide a
specified 60 dB dynamic range. Automatic gain control (AGC) can be
employed to ensure correct gain levels at an LNA and mixer and
gain-adjustable amplifier(s), under the condition of varying input
signal powers, in order to achieve optimal system performance.
Example System:
[0180] Diagram 1900 depicts a block diagram embodiment of an
example single-channel heterodyne receiver wherein exemplary
cascaded SAW filters can be used to achieve a desired level of
channel selectivity.
[0181] Many exemplary processing components are identified.
[0182] The system of diagram 1900 comprises individual processing
elements well known in the art and/or described herein. Each of
these elements is generally identified herein with a name and/or
abbreviation that corresponds to its well known and/or herein
described function. Analog filters comprise BandPass 1908 and
LowPass 1926. Exemplary SAW filters comprise IF Filter1 1920 and IF
Filter2 1922. Gain modifying elements comprise low noise amplifiers
LNA1 1906 and LNA2 1910, automatic gain control AGC1 1918 and AGC2
1924. Analog to digital converters comprise AD 1928. Mixers
comprise Mixer 1916. Attenuators comprise Attenuator 1912.
[0183] LNA1 1906 can be selectably coupled with Antenna 1902 via
switch 1904. When so coupled, LNA1 1906 can receive a signal from
Antenna 1902. BandPass 1908 can be coupled with and receive a
signal from LNA1 1906. LNA2 1910 can be coupled with and receive a
signal from BandPass 1908. Attenuator 1912 can be coupled with and
receive a signal from LNA2 1910. Mixer 1916 can be coupled with and
receive a signal from Attenuator 1912. Mixer 1916 can be coupled
with and receive a signal from Buffer 1914.
[0184] Buffer 1914 can provide an LO signal, as from an
oscillator.
[0185] AGC1 1918 can be coupled with and receive a signal from
Mixer 1916. IF Filter1 1920 can be coupled with and receive a
signal from AGC1 1918. IF Filter2 1922 can be coupled with and
receive a signal from IF Filter1 1920. AGC2 1924 can be coupled
with and receive a signal from IF Filter2 1922. LowPass 1926 can be
coupled with and receive a signal from AGC2 1924. AD 1928 can be
coupled with and receive a signal from LowPass 1926.
[0186] AD 1186 can provide a baseband component signal.
[0187] In some embodiments, an exemplary ADC, Analog Devices'
AD12401 [Analog Devices, Inc. AD12401, May 2006, op. cit.], a
12-bit 400 MHz ADC, can be used for direct IF sampling. The
following table shows a system gain analysis. An exemplary SAW
filter can have adjacent channel rejection of 8 dB and "Max. output
power" can be reduced accordingly at the output of each SAW filter.
Thus for some exemplary embodiments, a resulting overall system
noise figure can be computed to be about 5.2 dB.
TABLE-US-00002 Max. Output output Vendor: NF NF Gain power IP3 DR
Name Part (db) (dB) (dB) (dBm) (dBm) (dB) LNA1 Mini- 3.5 3.5 12 -8
47 110 Circuits: HELA- 10B Bandpass TBD 3 3.6 -3 -11 .infin.
.infin. LNA2 Mini- 3.5 3.9 12 1 47 92 Circuits: HELA- 10B
Attenuator TBD 4 3.9 -4 -3 .infin. .infin. Mixer Mini- 8 4.1 -8 -11
30 82 Circuits: ZFY-2 AGC1 Linear 7 4.9 17 6 47 82 Tech: LT5514 IF
Filter 1 Sawtek: 6 4.9 -6 -8 .infin. .infin. 854913 IF Filter 2
Sawtek: 6 4.9 -6 -22 .infin. .infin. 854913 AGC2 Linear 7 5.2 31 9
47 76 Tech: LT5514 Lowpass TBD 3 5.2 -3 6 .infin. .infin.
Transmitter Architecture
[0188] Diagram 2000 depicts an embodiment of a wideband
direct-conversion transmitter comprising a similar structure as
that of the wideband direct-conversion receiver of Diagram 800. ADC
elements 826 836 and DAC elements 2026 2036 have corresponding
positions within the depicted signal processing chains,
respectively. The position of LNA 806 corresponds to that of PA
2006. Essentially the same frequency planning approaches as
discussed regarding direct-conversion receiver embodiments can be
employed regarding direct-conversion transmitter embodiments. In
some embodiments, a mixing stage in diagram 2000 can perform an
up-conversion function; the mixing stage can comprise Mixer 2020
and Mixer 2030, and Quad splitter 2008.
[0189] Each of the digital to analog converters DAC 2026 2036 can
provide a digital to analog conversion function to a corresponding
received analog signal.
[0190] Each of the converters DAC 2026 2036 can be provided with a
baseband component signal (I and Q, respectively).
[0191] Each of the Baseband filters 2022 2032 can provide a
filtering function to a corresponding received signal.
[0192] Baseband filter 2022 can be coupled with and receive a
signal from DAC 2026. Baseband filter 2032 can be coupled with and
receive a signal from DAC 2036.
[0193] Oscillator LO 2010 can provide a signal that can be a tone
signal at a specified frequency.
[0194] Quad splitter 2008 can provide a quadrature splitting
function to a received signal, thereby providing an in-phase (I)
and a quadrature (Q) signal.
[0195] Quad splitter 2008 can be coupled with and receive a signal
from LO 2010.
[0196] Mixer 2020 can be coupled with and receive a signal of a
first specified phase from Quad splitter 2008.
[0197] Mixer 2020 can be coupled with and receive a filtered signal
from Baseband filter 2022.
[0198] Mixer 2030 can be coupled with and receive a signal of a
second specified phase from Quad splitter 2008.
[0199] Mixer 2030 can be coupled with and receive a filtered signal
from Baseband filter 2032.
[0200] Mixer 2020 can provide a mixing function, providing a signal
responsive to a signal received from Quad splitter 2008 and
responsive to a signal received from Baseband filter 2022.
Similarly,
[0201] Mixer 2030 can provide a mixing function, providing a signal
responsive to a signal of a second specified phase received from
Quad splitter 2008 and responsive to a signal received from
Baseband filter 2032.
[0202] Tx Power Control 2007 can provide a transmission power
control function to a received signal and/or received combination
of signals. A transmission power control function can comprise a
selectably adjustable gain and/or predistortion and/or any other
known and/or convenient transmission power control techniques.
[0203] Tx Power Control 2007 can be coupled with and receive a
combination of signals from Mixer 2020 and Mixer 2030. In some
embodiments, a combiner element can be employed to combine signals
from Mixer 2020 and Mixer 2030.
[0204] A power amplifier PA 2006 can provide a power amplification
function to a received signal.
[0205] PA 2006 can be coupled with and receive a signal from Tx
Power Control 2007.
[0206] RF filter 2004 can provide a filtering function to a
received signal.
[0207] RF filter 2004 can be coupled with and receive a
power-amplified signal from PA 2006.
[0208] Antenna 2002 can provide an antenna transmission function to
a received signal.
[0209] Antenna 2002 can be coupled with and receive a filtered
signal from RF filter 2004.
[0210] Antenna 2002 can provide transmission of a signal responsive
to a filtered signal received from RF filter 2004.
[0211] A maximum transmission power can be limited to 1 W or 30 dBm
according to the NPRM [FCC, May 2004, op. cit.]. Considering the
same exemplary 16-bit DAC as previously discussed, a maximum signal
power out of the DAC can be calculated
10 log 10 ( 2 .times. 0.5 2 50 .times. 10 3 ) = 10 dBm ( 60 )
##EQU00030##
[0212] Alternative modulation schemes can have varying backoff
requirements. For example, if OFDM is used, a backoff of 2.5 bits
translating into a power loss of 15 dB can be required. A maximum
signal power out of a DAC 2026 2036 can then be -5 dBm. A total
transmitter RF chain amplification of 35 dB can then be needed
before a signal reaches the antenna. A PA 2006 can typically
provide 20 dB to 30 dB of gain. Additional amplification stages can
then be needed between a PA 2006 and a DAC (2026 and/or 2036).
[0213] Transmitter power control (TPC) can be helpful in improving
wireless system capacity. TPC can be achieved using a variable gain
amplifier 2007 as shown in Diagram 2000. Alternatively, by
employing a DAC with an ample number of bits (16), transmission
power control can also be achieved using the DAC. For example, the
top 8 bits of a DAC output can be dedicated to TPC. This can
provide a total of 8.times.6=48 dB TPC range. In some embodiments,
the remaining 8 DAC bits can be used for OFDM modulation: 2.5 bits
for backoff and 5.5 bits for OFDM signal representation.
[0214] The FCC may adopt the same DTV transmit mask as shown in
Graph 200 for a TV-band cognitive radio. Given a modulation format,
using the spectrum mask, linearity requirements of RF components
can be derived.
[0215] Since a PA can provide a last amplification stage, transmit
chain nonlinearity can be dominated by that of the PA. Digital
pre-distortion can be used for PA linearization. Digital
pre-distortion techniques can be considered in a baseband system
design.
[0216] Diagram 2200 depicts a block diagram in some detail of an
example embodiment of a wideband direct-conversion transmitter
architecture essentially as depicted in Diagram 2000. In some
embodiments, an exemplary integrated wideband up-converter
HMC497LP4 from Hittite Microwave can be used for signal
up-conversion. In some embodiments, an exemplary Mini-Circuits
ZHL-3010 amplifier can be used as a PA driver. In some embodiments,
an Ophir 5303039A PA can have an output IP3 of 56 dBm and can
provide an output power of 36 dBm with out-of-band emission level
at -4 dBm. Notably, in some embodiments, every 1 dB reduction in
transmission power can result in a 2 dB reduction in out-of-band
emissions.
[0217] Transmission power control can be employed in some
embodiments to reduce out-of-band emissions.
[0218] The system of diagram 2200 comprises individual processing
elements well known in the art and/or described herein. Each of
these elements is generally identified herein with a name and/or
abbreviation that corresponds to its well known and/or herein
described function. Analog filters comprise BandPass 2204 and
LowPass 2222 2232. Gain modifying elements comprise Gain 2223 2233,
PA 2206, and VGA 2207. Digital to analog converters comprise DAC
2226 2236. An Upconverter 2209 can comprise splitter/combiners,
mixers, and a delay element. In some embodiments an Upconverter
2209 can be adapted to combine received (I) and (Q) baseband
component signals into a signal having a modulating or carrier
signal at the frequency of a received LO signal; hence
"upconversion". In some embodiments VGA 2207 can be adapted to
provide transmission power control.
[0219] Gain 2223 can be coupled with and receive a signal from DAC
2226. LowPass 2222 can be coupled with and receive a signal from
Gain 2223. Upconverter 2209 can be coupled with and receive a
baseband component signal from LowPass 2222. Gain 2233 can be
coupled with and receive a signal from DAC 2236. LowPass 2232 can
be coupled with and receive a signal from Gain 2233. Upconverter
2209 can receive an LO signal.
[0220] VGA 2207 can be coupled with and receive a modulated signal
from Upconverter 2209. PA 2206 can be coupled with and receive a
signal from VGA 2207. BandPass 2204 can be coupled with and receive
a signal from PA 2206. Antenna 2202 can be selectably coupled via
Switch 2203 with BandPass 2204. When so coupled, Antenna 2202 can
receive a signal from BandPass 2204 When so coupled, Antenna 2202
can provide transmission of a signal responsive to a filtered
signal received from BandPass 2204.
Baseband System Analysis:
[0221] A baseband system design is described herein.
[0222] FFT/IFFT-based digital filtering and reconstruction for
arbitrary channel rejection:
[0223] A double-ADC architecture for a wideband direct-conversion
TV-band cognitive radio receiver is herein described. An enabling
function for this architecture can be channel rejection through
digital filtering and reconstruction. Herein described is such a
channel rejection method from a baseband perspective.
[0224] Channel filtering can be accomplished using a common digital
filter, e.g. a raised-cosine filter. It can also be achieved using
an FFT and IFFT pair in combination. The latter approach can be
especially efficient in simultaneous filtering of multiple
channels, as required in some embodiments.
[0225] Herein described are derivations of a continuous-time
version of the operations of FFT/IFFT based filtering and
reconstruction. Equivalent discrete-time version of the operations
are subsequently described
Channel Rejection Analysis:
[0226] Referring to Equation (37), suppose a total signal is
y ( t ) = k .di-elect cons. .OMEGA. y k ( t ) = k .di-elect cons.
.OMEGA. [ x k ( t ) + q k ( t ) ] j2 .pi. f k t ( 61 )
##EQU00031##
from which a designated set of channels are to be rejected
l .di-elect cons. .LAMBDA. y l ( t ) ( 62 ) ##EQU00032##
An input signal can be truncated using a time-domain window
w(t):
y.sub.l(t)=w(t)y(t) (63)
which can then be "FFT'd" in order to generate a frequency-domain
signal representation
Y 1 ( f ) = F [ y 1 ( t ) ] = W ( f ) k .di-elect cons. .OMEGA. Y k
( f ) ( 64 ) ##EQU00033##
[0227] To retrieve the signal on a particular channel l .epsilon.
.LAMBDA., a frequency-domain rectangular window on Y.sub.1(f) can
be applied:
Y.sub.1(f)=.PI..sub.2C(f-f.sub.l)Y.sub.1(f) (65)
where .PI..sub.2C(f) is a rectangular window over the frequency
range [-C,C] with
C=3 MHz+.DELTA. (66)
and .DELTA. being the excess filter bandwidth. For all the channels
in .LAMBDA., then
Y ' ( f ) = l .di-elect cons. .LAMBDA. Y l ( f ) = [ l .di-elect
cons. .LAMBDA. .PI. 2 C ( f - f l ) ] Y 1 ( f ) ( 67 )
##EQU00034##
[0228] Note that for simplifying assumption that the channels in
.LAMBDA. are disjoint. In the case of contiguous channels, an
overall rectangular window can be applied to the contiguous
channels. The signal Y'(f) can then be transformed to time domain
in order to generate y'(t) as a reconstructed version of the
signals on the channels in .LAMBDA..
[0229] In order to evaluate how much rejection can be achieved, the
signal y'(t) can be subtracted from y.sub.1(t):
y 1 ( t ) - y ' ( t ) = w ( t ) y ( t ) - y ' ( t ) = w ( t ) [ k
.di-elect cons. ( .OMEGA. - .LAMBDA. ) y k ( t ) ] + [ w ( t ) l
.di-elect cons. .LAMBDA. y l ( t ) - y ' ( t ) ] ( 68 )
##EQU00035##
So the remaining signal power on the channels in .LAMBDA. can be
expressed:
.intg. - .infin. .infin. E [ w ( t ) l .di-elect cons. .LAMBDA. y l
( t ) - y ' ( t ) 2 ] t ( 69 ) ##EQU00036##
Since a similar amount of rejection can be applied to any
individual channel l .epsilon. .LAMBDA., consider that .LAMBDA.
only contains one channel l as a simplifying assumption. Using
Parseval's theorem
.intg. - .infin. .infin. E [ w ( t ) y l ( t ) - y l ( t ) 2 ] t =
.intg. - .infin. .infin. E [ W ( f ) Y l ( f ) - Y l ( f ) 2 ] f (
70 ) ##EQU00037##
Since the original signal power is
.intg. - .infin. .infin. E [ w ( t ) y l ( t ) 2 ] t = .intg. -
.infin. .infin. E [ W ( f ) Y l ( f ) 2 ] f ( 71 ) ##EQU00038##
rejection can be expressed as:
R dB = 10 log 10 .intg. - .infin. .infin. E [ W ( f ) Y l ( f ) 2 ]
f .intg. - .infin. .infin. E [ W ( f ) Y l ( f ) - Y l ( f ) 2 ] f
( 72 ) ##EQU00039##
Y.sub.1(f) can be assumed to be band-limited white Gaussian
noise--a justified assumption according to the central limit
theorem, if the signal x.sub.l(t) corresponds to filtered random
data samples at 6 MHz, e.g. the DTV signal. This can result in
E [ Y l ( f 1 ) Y l * ( f 2 ) ] = { N 0 .delta. ( f 1 - f 2 ) f 1 ,
f 2 .di-elect cons. [ f l - B , f l + B ] 0 Otherwise ( 73 )
##EQU00040##
where in some embodiments B=3 MHz. A spectral power of the original
signal, i.e. E[|W(f){circle around (.times.)}Y.sub.1(f)|.sup.2],
can be calculated as:
E [ W ( f ) Y l ( f ) 2 ] = E [ .intg. - .infin. .infin. W ( f - u
) Y l ( u ) u .intg. - .infin. .infin. W * ( f - v ) Y l * ( v ) v
] = E [ .intg. f l - B f l + B W ( f - u ) Y l ( u ) u .intg. f l -
B f l + B W * ( f - v ) Y l * ( v ) v ] = .intg. f l - B f l + B
.intg. f l - B f l + B u v W ( f - u ) W * ( f - v ) E [ Y l ( u )
Y l * ( v ) ] = .intg. f l - B f l + B .intg. f l - B f l + B u v W
( f - u ) W * ( f - v ) N 0 .delta. ( u - v ) = N 0 .intg. - B + B
W ( f - f l - u ) 2 u ( 74 ) ##EQU00041##
[0230] Now considering the spectral power after rejection, i.e.
E[|W(f){circle around (.times.)}Y.sub.1(f)-Y.sub.1(f)|.sup.2].
Inner terms can be expressed:
W ( f ) Y l ( f ) - Y l ( f ) = W ( f ) Y l ( f ) - .PI. 2 C ( f -
f l ) [ W ( f ) k .di-elect cons. .OMEGA. Y k ( f ) ] .apprxeq. W (
f ) Y l ( f ) - .PI. 2 C ( f - f l ) [ W ( f ) Y l ( f ) ] = [ 1 -
.PI. 2 C ( f - f l ) ] W ( f ) Y l ( f ) ( 75 ) ##EQU00042##
where an approximation can be taken because the signal Y.sub.1(f)
on channel l inside the rectangular window .PI..sub.2C(f-f.sub.l)
is far stronger (which is the reason it is being rejected) than the
signals on the other channels whose power leakages into the channel
are then negligible. From the above, it follows:
E [ W ( f ) Y l ( f ) - Y l ( f ) 2 ] = [ 1 - .PI. 2 C ( f - f l )
] 2 E [ W ( f ) Y l ( f ) 2 ] [ 1 - .PI. 2 C ( f - f l ) ] N 0
.intg. - B + B W ( f - f l - u ) 2 u ( 76 ) ##EQU00043##
Let
[0231] K(f)=.intg..sub.-B.sup.+B|W(f-u)|.sup.2 du (77)
The rejection can then be expressed as:
R = .intg. - .infin. .infin. N 0 K ( f - f l ) f .intg. - .infin.
.infin. [ 1 - .PI. 2 C ( f - f l ) ] N 0 K ( f - f l ) f = .intg. -
.infin. .infin. K ( f ) f .intg. - .infin. .infin. [ 1 - .PI. 2 C (
f ) ] K ( f ) f ( 78 ) ##EQU00044##
or
R dB = 10 log 10 [ .intg. - .infin. .infin. K ( f ) f .intg. -
.infin. .infin. [ 1 - .PI. 2 C ( f ) ] K ( f ) f ] ( 79 )
##EQU00045##
Assuming that the time-domain window is a raised-cosine window:
w ( t ) = { 1 2 + 1 2 cos { .pi. .beta. T w [ t + 1 2 ( 1 - .beta.
) T w ] } - 1 2 ( 1 + .beta. ) T w .ltoreq. t .ltoreq. - 1 2 ( 1 -
.beta. ) T w 1 - 1 2 ( 1 - .beta. ) T w .ltoreq. t .ltoreq. 1 2 ( 1
- .beta. ) T w 1 2 + 1 2 cos { .pi. .beta. T w [ t - 1 2 ( t -
.beta. ) T w ] } 1 2 ( 1 - .beta. ) T w .ltoreq. t .ltoreq. 1 2 ( 1
+ .beta. ) T w 0 Otherwise ( 80 ) ##EQU00046##
with frequency-domain representation:
W ( f ) = sin .pi. fT w .pi. f cos ( .pi..beta. fT w ) 1 - 4 .beta.
2 f 2 T w 2 ( 81 ) ##EQU00047##
[0232] In some embodiments a further assumption can be employed
that an FFT of size N is employed on input signal samples at 400
MHz such that
T w ( 1 + .beta. ) = N 1 400 T w = N / 400 ( 1 + .beta. ) ( 82 )
##EQU00048##
where T.sub.w is expressed in .mu.s. Note that in some embodiments
the subcarrier spacing (inverse of the FFT period) can be:
400 N MHz = 400 .times. 10 3 N kHz ( 83 ) ##EQU00049##
Channel Rejection Performance Simulation:
[0233] Computer simulation can be employed to compute the rejection
expression of Equation (79).
[0234] In some embodiments a 20-30 dB rejection can be sufficient
for a double-ADC architecture as discussed herein. The following
table shows three example configurations that can achieve 20 dB
rejection
TABLE-US-00003 N .beta. .DELTA. (kHz) 512 0.4 500 1024 0.3 160 2048
0.2 30
Equivalent Discrete-Time Operations for Filtering and
Reconstruction:
[0235] An embodiment utilizing equivalent discrete-time operations
can be described.
[0236] A windowing function can be applied
y.sub.l(n)=w(n)y(n) (84)
where w(n) is given by Equation (80) with T.sub.w given by Equation
(82) and a sampling time t can be replaced by a sampling index
n = t T s ( 85 ) ##EQU00050##
where
Y 1 ( k ) = 1 N n = - N / 2 N / 2 - 1 y 1 ( n ) - j2.pi. k N n ( 86
) ##EQU00051##
is the sampling period.
[0237] A FFT can be performed on the resulting signal
T s = 1 400 s ##EQU00052##
[0238] A rejection mask
.SIGMA..sub.l.epsilon..LAMBDA..PI..sub.2C(f-f.sub.i) can be
applied. This operation can comprise the steps of: finding
subcarriers whose indices are in the rejection mask; setting
Y'(k)=Y.sub.1(k) for those subcarriers; and, nullifying Y'(k) for
all other subcarriers.
[0239] An inverse Fourier transform can be applied
y ' ( n ) = 1 N k = - N / 2 N / 2 - 1 Y ' ( k ) j2.pi. k n N ( 87 )
##EQU00053##
Signal samples, i.e. y'(n)s, inside the flat portion of the window
w(t), i.e. t.epsilon.[-(1-.beta.)T.sub.w,(1-.beta.)T.sub.w], can be
sent to a DAC in order to construct a rejection signal y'(t).
[0240] In theory, the multiplication of two signals is only
equivalent in continuous-time and discrete-time domains if the
output signal is band-limited. Since w(t) is essentially
time-limited, it is essentially not frequency-limited. However,
because in an embodiment w(t) can have a bandwidth that is
significantly narrower than the sampling bandwidth, i.e. 400 MHz,
w(t) can be usefully approximated as a delta function in frequency
domain. Under these conditions the continuous- and discrete-time
multiplications can be essentially equivalent.
[0241] A FFT is of finite size can sample the input signal spectrum
at only certain frequencies. The rejection performance result
derived here for the continuous spectrum can represent an averaged
performance.
[0242] The operations just described above can construct a
rejection signal for the flat portion of a window. A signal in the
nonflat portion of the window can require additional compensation
that can introduce additional error. Constructing a rejection
signal for a non-flat portion of a window can require additional
FFT resources. That is, supporting a streaming operation can
require overlapping two FFT windows such that their flat portions
can be connected together.
[0243] The graph 2300 of FIG. 23 shows simulated multi-carrier
signal power spectrums at different IP3s (or different Ds).
Nonlinearity can cause spectrum "shoulders" in adjacent bands. The
decibel (dB) difference between the inband signal power and the
shoulder can be roughly 2D, or the system dynamic range
P.sub.DR.
[0244] The graph 2300 illustrates simulated signal power spectra
under varying device nonlinearities in a multi-carrier system with
subcarrier spacing 100 kHz, .beta.=0.16, number of guard band
subcarriers 8 (and number of valid data subcarriers 52). Individual
curves 2302 2304 2306 2308 are shown for IP3-related distance D
values of (respectively) 15 dB, 25 dB, 35 dB, and .infin..
[0245] In some embodiments with a fixed output power, a higher
device IP3 can be required in order to reduce adjacent channel
leakage. In some embodiments, an IP3 requirement can be reduced by
applying a digital predistortion technique and/or process.
[0246] In the foregoing specification, the embodiments have been
described with reference to specific elements thereof. It will,
however, be evident that various modifications and changes may be
made thereto without departing from the broader spirit and scope of
the embodiments. For example, the reader is to understand that the
specific ordering and combination of process actions shown in the
process flow diagrams described herein is merely illustrative, and
that using different or additional process actions, or a different
combination or ordering of process actions can be used to enact the
embodiments. For example, specific reference to NTSC and/or ATSC
and/or DTV embodiments are provided by way of non-limiting
examples. Systems and methods herein described can be applicable to
any other known and/or convenient channel-based communication
embodiments; these can comprise single and/or multiple carriers per
channel. The specification and drawings are, accordingly, to be
regarded in an illustrative rather than restrictive sense.
* * * * *
References