U.S. patent application number 11/630391 was filed with the patent office on 2008-05-08 for closed loop mimo systems and methods.
Invention is credited to Mo-Han Fong, Ming Jia, Jianglei Ma, Wen Tong, Hua Xu, Dong-Sheng Yu, Hang Zhang, Peiying Zhu.
Application Number | 20080108310 11/630391 |
Document ID | / |
Family ID | 35510066 |
Filed Date | 2008-05-08 |
United States Patent
Application |
20080108310 |
Kind Code |
A1 |
Tong; Wen ; et al. |
May 8, 2008 |
Closed Loop Mimo Systems and Methods
Abstract
Systems and methods for closed loop MIMO (multiple input and
multiple output) wireless communication are provided. Various
transmit formats including spatial multiplexing and STTD are
defined in which vector or matrix weighting is employed using
information fed back from receivers. The feedback information may
include channel matrix or SVD-based feedback.
Inventors: |
Tong; Wen; (Ottawa, CA)
; Jia; Ming; (Ottawa, CA) ; Ma; Jianglei;
(Kanata, CA) ; Zhu; Peiying; (Kanata, CA) ;
Xu; Hua; (Nepean, CA) ; Yu; Dong-Sheng;
(Ottawa, CA) ; Zhang; Hang; (Nepean, CA) ;
Fong; Mo-Han; (L'Original, CA) |
Correspondence
Address: |
SMART & BIGGAR;P.O. BOX 2999, STATION D
900-55 METCALFE STREET
OTTAWA
ON
K1P5Y6
CA
|
Family ID: |
35510066 |
Appl. No.: |
11/630391 |
Filed: |
June 22, 2005 |
PCT Filed: |
June 22, 2005 |
PCT NO: |
PCT/CA05/00958 |
371 Date: |
December 22, 2006 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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60581356 |
Jun 22, 2004 |
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60582298 |
Jun 24, 2004 |
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60601178 |
Aug 13, 2004 |
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60614621 |
Sep 30, 2004 |
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60619461 |
Oct 15, 2004 |
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60642697 |
Jan 10, 2005 |
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Current U.S.
Class: |
455/69 |
Current CPC
Class: |
H04W 72/042 20130101;
H04W 72/044 20130101; H04L 5/0007 20130101; H04B 7/0478 20130101;
H04L 5/023 20130101; H04B 7/0673 20130101; H04B 7/0619 20130101;
H04L 5/005 20130101; H04W 74/06 20130101; H04L 5/006 20130101; H04L
27/2601 20130101; H04B 7/0671 20130101; H04W 48/08 20130101; H04W
72/08 20130101; H04L 5/0028 20130101; H04B 7/0691 20130101; H04L
5/0023 20130101; H04B 7/0413 20130101; H04B 7/0617 20130101 |
Class at
Publication: |
455/069 |
International
Class: |
H04B 1/00 20060101
H04B001/00 |
Claims
1. A MIMO system comprising: a transmitter having multiple transmit
antennas; at least one receiver, each receiver having at least one
receive antenna; each receiver being adapted to transmit at least
one type of feedback information selected from a group consisting
of: information for use in performing beam-forming; antenna
selection/grouping information.
2. The system of claim 1 wherein a transmission format to each
receiver is selected from a group of transmission formats
consisting of: spatial multiplexing; vector weighted spatial
multiplexing; matrix weighted spatial multiplexing; K-stream
spatial multiplexing employing more than K transmit antennas;
single stream STTD; single stream STTD with proportional weighting
and antenna selection; multi-stream STTD; multi-stream STTD with
layer weighting; multi-stream STTD with a combination of layer
weighting and proportional weighting; and hybrid beam-forming and
spatial multiplexing.
3. The system of claim 2 wherein a defined sub-set of available
formats is made available for a given receiver, and wherein the
given receiver feeds back a selection of one of the defined sub-set
of available formats.
4. The system of claim 1 wherein: each receiver performs respective
channel measurements and feeds back information for use in
performing beam-forming based on the respective channel
measurements.
5. The system of claim 4 wherein the information for use in
performing beam-forming is selected from a group consisting of: a)
elements of a measured channel matrix; b) elements of a V matrix of
a SVD decomposed channel matrix; c) parameters of a Givens
decomposition of a V matrix of a SVD decomposed channel matrix; d)
parameters of a truncated Givens decomposition of a V matrix of a
SVD decomposed channel matrix, where one or more eigen-vectors are
discarded; e) differentially encoded elements of a measured channel
matrix; f) differentially encoded elements of a V matrix of a SVD
decomposed channel matrix; g) differentially encoded parameters of
a Givens decomposition or truncated Givens decomposition of a V
matrix of a SVD decomposed channel matrix; h) Householder
decomposition; i) full scalar quantization of any of the
information types of a) through h); j) partial scalar quantization
of any of the information types a) through g); k) scalar
quantization of any one of the information types a) through h)
where varying resolution is used to quantize parameters; l) vector
quantization of any of the information types of a) through h); m) a
combination of scalar quantization and differential quantization
for any of the information types a) through h); n) using a Delta
Sigma quantizer for any of the information types a) through h). o)
binary beam-forming weights; p) a differential index into a set of
vector quantizations; and q) pre-defined codebook.
6. The system of claim 1 wherein beam-forming feedback is performed
by each receiver as a function of receiver specific criteria.
7. The system of claim 6 wherein the receiver specific criteria is
selected from a group consisting of: b) Max SNR; b) Max Shannon
capacity; and c) True receiver operational process.
8. The system of claim 1 wherein antenna selection/grouping
information is at least one information type selected from a group
consisting of: a) selection between SM (spatial multiplexing) and
STTD (space time transmit diversity) transmission format; b)
selection of particular antennas for SM transmission; c) selection
and grouping of particular antennas for STTD transmission; and d)
eigen-mode selection information.
9. The system of claim 1 further comprising the receiver
determining the antenna selection/grouping information by
performing a step selected from a group of steps consisting of:
performing SVD decomposition and discarding weak eigen-modes;
selecting antennas using determinants of sub-MIMO channel
matrices.
10. The system of claim 1 wherein feed back and beam-forming and/or
antenna selection/grouping is performed for sub-carriers of a
multi-carrier system to a resolution selected from a group
consisting of: a) for every sub-carrier individually; b) for groups
of consecutive sub-carriers; c) for an entire set of sub-carriers;
d) for sets of groups of sub-carriers.
11. The system of claim 1 wherein transmission matrices and
feedback are in accordance with one of FIGS. 11 to 14.
12. The system of claim 1 wherein: the transmitter transmits pilots
on each transmit antenna for use in performing channel
estimation.
13. The system of claim 11 wherein at least some of the pilots are
punctured pilots.
14. The system of claim 11 wherein at least some of the pilots
comprise un-coded pilots for use by multiple receivers.
15. The system of claim 11 wherein the pilots comprise user
specific pre-coded pilots for use by particular receivers
receivers.
16. The system of claim 11 wherein the pilots comprise user
specific pre-coded pilots for use by particular receivers receivers
and un-coded pilots for use by multiple receivers.
17. The system of claim 11 wherein the pilot patterns are as shown
in any one of FIGS. 17-23 with generalizations as described.
18. The system of claim 11 wherein the pilot patterns are as shown
in one of FIGS. 26-31 with generalizations as described.
19. The system of claim 1 wherein feedback information is
transmitted using a feedback channel having the structure of one of
FIGS. 46 to 48 with generalizations as described.
20. The system of claim 1 wherein at least one receiver has a
plurality of receive antennas.
21. The system of claim 1 wherein the at least one receiver
comprises a plurality of receivers.
22. The system of claim 1 wherein sub-channels are defined using at
least one of: AMC sub-channels, where respective adaptive
modulation and coding is defined for each AMC sub-channel; PUSC
sub-channels.
23. A receiver adapted to implement receiver functionality of claim
1.
24. A transmitter adapted to implement transmitter functionality of
claim 1.
Description
RELATED APPLICATIONS
[0001] This application claims the benefit of U.S. Provisional
Patent Application No. 60/581,356 filed on Jun. 22, 2004, U.S.
Provisional patent Application No. 60/582,298 filed on Jun. 24,
2004, U.S. Provisional Patent Application No. 60/601,178 filed on
Aug. 13, 2004, Provisional Patent Application No. 60/514,621 filed
on Sep. 30, 2004, Provisional Patent Application No. 60/619,461
filed on Oct. 15, 2004 and Provisional Patent Application No.
60/642,697 filed on Jan. 10, 2005, all of which are hereby
incorporated by reference in their entirety.
FIELD OF THE INVENTION
[0002] The invention relates to MIMO (multiple input, multiple
output) systems and methods.
BACKGROUND OF THE INVENTION
[0003] In MIMO (multiple input multiple output) OFDM (orthogonal
frequency division multiplexing) systems, there are multiple
transmit antennas and multiple receive antennas and a plurality of
sub-carriers that are available for transmission between the
transmit antennas and the receive antennas for either one or
multiple users. New advances in MIMO OFDM systems are taught in
Applicant's co-pending application <attorney docket
71493-1320> entitled "Pilot Design For OFDM Systems With Four
transmit Antennas" filed Mar. 15, 2005, and in Applicant's
co-pending application <attorney docket 71493-1330) entitled
"Wireless Communication Methods, Systems, And Signal Structures"
filed Apr. 4, 2005, both hereby incorporated by reference in their
entirety. With open loop implementations, the transmitter transmits
on the multiple transmitter antennas and sub-carriers without the
benefit of channel information fed back from the receivers.
[0004] Efforts have been made to facilitate wireless closed-loop
MIMO communications including broadband closed-loop MIMO, which
might for example be based on OFDM modulation schemes, and
narrowband closed-loop MIMO. Broadband closed-loop MIMO includes
many sub-bands. Each of these sub-bands requires MTMO channel
feedback for a closed-loop implementation. As a result the feedback
resources required for broadband closed-loop MIMO can become quite
large. Narrowband closed-loop MIMO, by comparison, includes one or
a few sub-bands and requires a relatively smaller amount of
feedback resources. Broadband and narrowband MIMO, therefore, have
different applications.
SUMMARY OF THE INVENTION
[0005] According to one broad aspect, the invention provides a MIMO
system comprising: a transmitter having multiple transmit antennas;
at least one receiver, each receiver having at least one receive
antenna; each receiver being adapted to transmit at least one type
of feedback information selected from a group consisting of:
information for use in performing beam-forming; antenna
selection/grouping information.
[0006] In some embodiments, a transmission format to each receiver
is selected from a group of transmission formats consisting of:
spatial multiplexing; vector weighted spatial multiplexing; matrix
weighted spatial multiplexing; K-stream spatial multiplexing
employing more than K transmit antennas; single stream STTD; single
stream STTD with proportional weighting and antenna selection;
multi-stream STTD; multi-stream STUD with layer weighting;
multi-stream STTD with a combination of layer weighting and
proportional weighting; and hybrid beam-forming and spatial
multiplexing.
[0007] In some embodiments, a defined sub-set of available formats
is made available for a given receiver, and wherein the given
receiver feeds back a selection of one of the defined sub-set of
available formats.
[0008] In some embodiments, each receiver performs respective
channel measurements and feeds back information for use in
performing beam-forming based on the respective channel
measurements.
[0009] In some embodiments, the information for use in performing
beam-forming is selected from a group consisting of: a) elements of
a measured channel matrix; b) elements of a V matrix of a SVD
decomposed channel matrix; c) parameters of a Givens decomposition
of a V matrix of a SVD decomposed channel matrix; d) parameters of
a truncated Givens decomposition of a V matrix of a SVD decomposed
channel matrix, where one or more eigen-vectors are discarded; e)
differentially encoded elements of a measured channel matrix; f)
differentially encoded elements of a V matrix of a SVD decomposed
channel matrix; g) differentially encoded parameters of a Givens
decomposition or truncated Givens decomposition of a V matrix of a
SVD decomposed channel matrix; h) Householder decomposition; i)
full scalar quantization of any of the information types of a)
through h); j) partial scalar quantization of any of the
information types a) through g); k) scalar quantization of any one
of the information types a) through h) where varying resolution is
used to quantize parameters; l) vector quantization of any of the
information types of a) through h); m) a combination of scalar
quantization and differential quantization for any of the
information types a) through h); n) using a Delta Sigma quantizer
for any of the information types a) through h); o) binary
beam-forming weights; p) a differential index into a set of vector
quantizations; and q) pre-defined codebook.
[0010] In some embodiments beam-forming feedback is performed by
each receiver as a function of receiver specific criteria.
[0011] In some embodiments, the receiver specific criteria is
selected from a group consisting of: Max SNR; b) Max Shannon
capacity; and c) True receiver operational process.
[0012] In some embodiments, antenna selection/grouping information
is at least one information type selected from a group consisting
of: a) selection between SM (spatial multiplexing) and STTD (space
time transmit diversity) transmission format; b) selection of
particular antennas for SM transmission; c) selection and grouping
of particular antennas for STTD transmission; and d) eigen-mode
selection information.
[0013] In some embodiments, the system further comprises the
receiver determining the antenna selection/grouping information by
performing a step selected from a group of steps consisting of:
performing SVD decomposition and discarding weak eigen-modes;
selecting antennas using determinants of sub-MIMO channel
matrices.
[0014] In some embodiments, feed back and beam-forming and/or
antenna selection/grouping is performed for sub-carriers of a
multi-carrier system to a resolution selected from a group
consisting of: a) for every sub-carrier individually; b) for groups
of consecutive sub-carriers; c) for an entire set of sub-carriers;
d) for sets of groups of sub-carriers.
[0015] In some embodiments, transmission matrices and feedback are
in accordance with one of FIGS. 11 to 14.
[0016] In some embodiments, the transmitter transmits pilots on
each transmit antenna for use in performing channel estimation.
[0017] In some embodiments, at least some of the pilots are
punctured pilots.
[0018] In some embodiments, at least some of the pilots comprise
un-coded pilots for use by multiple receivers.
[0019] In some embodiments, the pilots comprise user specific
pre-coded pilots for use by particular receivers receivers.
[0020] In some embodiments, the pilots comprise user specific
pre-coded pilots for use by particular receivers receivers and
un-coded pilots for use by multiple receivers.
[0021] In some embodiments, the pilot patterns are as shown in any
one of FIGS. 17-23 with generalizations as described.
[0022] In some embodiments, the pilot patterns are as shown in one
of FIGS. 26-31 with generalizations as described.
[0023] In some embodiments, feedback information is transmitted
using a feedback channel having the structure of one of FIGS. 46 to
48 with generalizations as described.
[0024] In some embodiments, at least one receiver has a plurality
of receive antennas.
[0025] In some embodiments, the at least one receiver comprises a
plurality of receivers.
[0026] In some embodiments, sub-channels are defined using at least
one of; AMC sub-channels, where respective adaptive modulation and
coding is defined for each AMC sub-channel; PUSC sub-channels.
[0027] In another embodiment, a receiver is provided that is
adapted to implement receiver functionality as summarized
above.
[0028] In another embodiment, a transmitter is provided that is
adapted to implement transmitter functionality as summarized
above.
[0029] Other aspects and features of the present invention will
become apparent to those ordinarily skilled in the art upon review
of the following description of specific embodiments of the
invention in conjunction with the accompanying figures.
BRIEF DESCRIPTION OF THE DRAWINGS
[0030] Preferred embodiments of the invention will now be described
with reference to the attached drawings in which:
[0031] FIG. 1 is a schematic diagram representation of a cellular
communication system according to one embodiment of the present
invention;
[0032] FIG. 2 is a block diagram representation of a base station
according to one embodiment of the present invention;
[0033] FIG. 3 is a block diagram representation of a mobile
terminal according to one embodiment of the present invention;
[0034] FIG. 4 is a logical breakdown of an OFDM transmitter
architecture according to one embodiment of the present
invention;
[0035] FIG. 5 is a logical breakdown of an OFDM receiver
architecture according to one embodiment of the present
invention;
[0036] FIG. 6 is a first example schematic diagram for beam-forming
spatial multiplexing (SM) transmission using matrix or vector
weighting according to an embodiment of the invention;
[0037] FIG. 7 is a second example schematic diagram for
beam-forming SM transmission with matrix weighting according to an
embodiment of the invention;
[0038] FIG. 8 is a schematic diagram for use in describing
antenna/sub-channel selection criteria;
[0039] FIG. 9 is a graphical comparison of fixed D-STTD
(double-space-time time division) and antenna grouping D-STTD;
[0040] FIG. 10 is a schematic diagram of sub-channel allocation for
a 4-antenna transmitter and two 2-antenna receivers according to an
embodiment of the invention;
[0041] FIG. 11 is a closed loop STC/MIMO 3-transmit antenna
grouping arrangement in accordance with an embodiment of the
invention;
[0042] FIG. 12 is a closed loop STC/MIMO 3-transmit antenna
selection arrangement in accordance with an embodiment of the
invention;
[0043] FIG. 13 is a closed loop STC/MIMO 4-transmit antenna
arrangement in accordance with an embodiment of the invention;
[0044] FIG. 14 is a closed loop STC/MIMO 4-transmit antenna
arrangement in accordance with an embodiment of the invention;
[0045] FIGS. 15 and 16 show binary unitary beam-forming matrices in
accordance with embodiments of the invention;
[0046] FIG. 17 is pilot mapping for a pilot allocation for
4-antenna BS (base station) for the optional FUSC (full utilization
sub-channel) and Optional AMC (adaptive modulation and coding)
zones in 802.16d in accordance with an embodiment of the
invention;
[0047] FIGS. 18 and 19 are pilot mappings for a pilot allocation
for four transmit antennas in which there is no puncturing required
in accordance with an embodiment of the invention;
[0048] FIGS. 20 and 21 are pilot mappings for a pilot allocation
for eight transmit antennas in accordance with an embodiments of
the invention;
[0049] FIGS. 22 and 23 are pilot mappings for a pilot allocation
for twelve transmit antennas in accordance-with an embodiments of
the invention;
[0050] FIG. 24 is a schematic diagram showing an example of
pre-coding of MIMO pilots in accordance with an embodiment of the
invention;
[0051] FIG. 25 is a schematic diagram showing an example of
pre-coding of MIMO pilots in accordance with an embodiment of the
invention suitable for larger antenna arrays;
[0052] FIG. 26 is a pilot mapping showing a pre-coded pilot design
for a 2-antenna basestation (BS) for optional AMC in accordance
with an embodiment of the invention;
[0053] FIG. 27 is a second pre-coded pilot design for a 2-antenna
basestation (BS) for optional AMC in accordance with an embodiment
of the invention;
[0054] FIG. 28 is a pilot mapping showing a pre-coded pilot design
for a 3-antenna basestation (BS) for optional AMC in accordance
with an embodiment of the invention;
[0055] FIG. 29 is a pilot mapping showing a pilot design for a
4-antenna basestation (BS) for optional AMC in accordance with an
embodiment of the invention;
[0056] FIG. 30 is a pilot mapping showing a pre-coded pilot design
for a 2-antenna BS for PUSC (partial utilization sub-channel) zone
in accordance with an embodiment of the invention;
[0057] FIG. 31 is a pilot mapping showing a pre-coded pilot design
for a 4-antenna ES for PUSC zone in accordance with an embodiment
of the invention;
[0058] FIG. 32 is a schematic diagram of a set of closed loop
STC/MIMO arrangements with beam-former structures in accordance
with an embodiment of the invention;
[0059] FIGS. 33, 34, 35 and 36 present a comparison of SVD
(singular value decomposition) to antenna grouping;
[0060] FIG. 37 is a block diagram of a system employing a direct
differential encoding in accordance with an embodiment of the
invention in which MIMO channel and CQI (channel quality
indication) are separately fed back;
[0061] FIG. 38 is a block diagram of a system employing a direct
differential encoding in accordance with another embodiment of the
invention in which MIMO channel and CQI are jointly fed back;
[0062] FIG. 39 is a block diagram of a system employing a direct
differential encoding in accordance with an embodiment of the
invention featuring 1 bit DPCM;
[0063] FIG. 40 is a block diagram of a system employing a direct
differential encoding in accordance with an embodiment of the
invention and using a 1 bit AZ modulator;
[0064] FIG. 41 is a block diagram of a system employing a direct
differential encoding in accordance with an embodiment of the
invention for multiple users;
[0065] FIG. 42 contains a table of various direct differential
encoding feedback in accordance with embodiments of the
invention;
[0066] FIG. 43 is a block diagram of a system employing an SVD
based Givens transform feedback in accordance with an embodiment of
the invention;
[0067] FIG. 44 is another example of an SVD based Givens transform
in accordance with an embodiment of the invention in which a
further spherical code based quantization is performed;
[0068] FIG. 45 is a block diagram of a system employing a receiver
based Givens transform in accordance with an embodiment of the
invention;
[0069] FIG. 46 is an example of space-time coding for use on a
CQICH (channel quality indication channel) in accordance with an
embodiment of the invention suitable for a single input single
output application;
[0070] FIG. 47 is an example of space-time coding for CQICH in
accordance with an embodiment of the invention suitable for
supporting STUD;
[0071] FIG. 48 is an example of space-time coding for CQICH in
accordance with an embodiment of the invention suitable for SM
(spatial multiplexing);
[0072] FIGS. 49 and 50 are a set of tables for concatenation of STC
(space-time coding)/MIMO with a beam-former in accordance with an
embodiment of the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
General Background and Example System Overview The following
provides a glossary of some of the terms used in this
application:
AMC--Adaptive Coding and Modulation
BS or BTS--Base Station
CL_MIMO--Closed Loop MIMO
CQI--Channel Quality Indicator
CQICF--CQI channel
DFT--Discrete Fourier Transform
FB--Feedback
FDD--Frequency Duplex
FFT--Fast Fourier Transform
MIMO--Multiple Input Multiple Output
MLD--Maximum Likelihood Detector
MSE--Minimum square error
MSS--Mobile Subscriber Station
PUSC--Partially Utilized Sub-Channel
QoS--Quality of service
SISO--Single Input Single Output
SVD--Singular Value Decomposition
STTD--Space Time Transmit Diversity
SM--Spatial Multiplexing
SQ--Scalar Quantize
TDD--Time Duplex
VQ--Vector Quantize
[0073] For purposes of providing context for the embodiments
described below, an example OFDM system will now be described with
reference to FIGS. 1 to 5. FIG. 1 shows a base station controller
(BSC) 10 which controls wireless communications within multiple
cells 12, which cells are served by corresponding base stations
(BS) 14. In general, each base station 14 facilitates
communications using OFDM with mobile terminals 16, which are
within the cell 12 associated with the corresponding base station
14. The movement of the mobile terminals 16 in relation to the base
stations 14 results in significant fluctuation in channel
conditions. As illustrated, the base stations 14 and mobile
terminals 16 may include multiple antennas to provide spatial
diversity for communications.
[0074] An example of a high level overview of the mobile terminals
16 and base stations 14 that may be used with embodiments of the
present invention is provided prior to delving into the structural
and functional details of the preferred embodiments. Wish reference
to FIG. 2, a base station 14 configured according to one embodiment
of the present invention is illustrated. The base station 14
generally includes a control system 20, a baseband processor 22,
transmit circuitry 24, receive circuitry 26, multiple antennas 28,
and a network interface 30. The receive circuitry 26 receives radio
frequency signals bearing information from one or more remote
transmitters provided by mobile terminals 16 (illustrated in FIG.
3). Preferably, a low noise amplifier and a filter (not shown)
cooperate to amplify and remove broadband interference from the
signal for processing. Downconversion and digitization circuitry
(not shown) will then downconvert the filtered, received signal to
an intermediate or baseband frequency signal, which is then
digitized into one or more digital streams.
[0075] The baseband processor 22 processes the digitized received
signal to extract the information or data bits conveyed in the
received signal. This processing typically comprises demodulation,
decoding, and error correction operations. As such, the baseband
processor 22 is generally implemented in one or more digital signal
processors (DSPs) or application-specific integrated circuits
(ASICs). The received information is then sent across a wireless
network via the network interface 30 or transmitted to another
mobile terminal 16 serviced by the base station 14.
[0076] On the transmit side, the baseband processor 22 receives
digitized data, which may represent voice, data, or control
information, from the network interface 30 under the control of
control system 20, and encodes the data for transmission. The
encoded data is output to the transmit circuitry 24, where it is
modulated by a carrier signal having a desired transmit frequency
or frequencies. A power amplifier (not shown) will amplify the
modulated carrier signal to a level appropriate for transmission,
and deliver the modulated carrier signal to the antennas 28 through
a matching network (not shown). Modulation and processing details
are described in greater detail below.
[0077] With reference to FIG. 3, a mobile terminal 16 configured
according to one embodiment of the present invention is
illustrated. Similar to the base station 14, the mobile terminal 16
will include a control system 32, a baseband processor 34, transmit
circuitry 36, receive circuitry 38, multiple antennas 40, and user
interface circuitry 42. The receive circuitry 38 receives radio
frequency signals bearing information from one or more base
stations 14. Preferably, a low noise amplifier and a filter (not
shown) cooperate to amplify and remove broadband interference from
the signal for processing. Downconversion and digitization
circuitry (not shown) will then downconvert the filtered, received
signal to an intermediate or baseband frequency signal, which is
then digitized into one or more digital streams.
[0078] The baseband processor 34 processes the digitized received
signal to extract the information or data bits conveyed in the
received signal. This processing typically comprises demodulation,
decoding, and error correction operations, as will be discussed on
greater detail below. The baseband processor 34 is generally
implemented in one or more digital signal processors (DSPs) and
application specific integrated circuits (ASICs).
[0079] For transmissions the baseband processor 34 receives
digitized data, which may represent voice, data, or control
information, from the control system 32, which it encodes for
transmission. The encoded data is output to the transmit circuitry
36, where it is used by a modulator to modulate a carrier signal
that is at a desired transmit frequency or frequencies. A power
amplifier (not shown) will amplify the modulated carrier signal to
a level appropriate for transmission, and deliver the modulated
carrier signal to the antennas 40 through a matching network (not
shown). Various modulation and processing techniques available to
those skilled in the art are applicable to the present
invention.
[0080] In OFDM modulation, the transmission band is divided into
multiple, orthogonal carrier waves. Each carrier wave is modulated
according to the digital data to be transmitted. Because OFDM
divides the transmission band into multiple carriers, the bandwidth
per carrier decreases and the modulation time per carrier
increases. Since the multiple carriers are transmitted in parallel,
the transmission rate for the digital data, or symbols, on any
given carrier is lower than when a single carrier is used.
[0081] OFDM modulation typically employs the performance of an
Inverse Fast Fourier Transform (IFFT) on the information to be
transmitted. For demodulations a Fast Fourier Transform (FFlT) is
typically performed on the received signal to recover the
transmitted information. In practice, the IFFT and FFT are provided
by digital signal processing carrying out an Inverse Discrete
Fourier Transform (IDFT) and Discrete Fourier Transform (DFT),
respectively. Accordingly, the characterizing feature of OFDM
modulation is that orthogonal carrier wave are generated for
multiple bands within a transmission channel. The modulated signals
are digital signals having a relatively low transmission rate and
capable of staying within their respective bands. The individual
carrier waves are not modulated directly by the digital signals.
Instead, all carrier waves are modulated at once by IFFT
processing.
[0082] In the preferred embodiment, OFDM is used for at least the
downlink transmission from the base stations 14 to the mobile
terminals 16. Each base station 14 is equipped with N transmit
antennas 28, and each mobile terminal 16 is equipped with M receive
antennas 40. Notably, the respective antennas can be used for
reception and transmission using appropriate duplexers or
switches.
[0083] With reference to FIG. 4, a logical OFDM transmission
architecture is provided according to one embodiment. Initially,
the base station controller 10 will send data to be transmitted to
various mobile terminals 16 to the base station 14. The base
station 14 may use the CQIs associated with the mobile terminals to
schedule the data for transmission as well as select appropriate
coding and modulation for transmitting the scheduled data. The CQIs
may be directly from the mobile terminals 16 or determined at the
base station 14 based on information provided by the mobile
terminals 16. In either case, the CQI for each mobile terminal 16
is a function of the degree to which the channel amplitude (or
response) varies across the OFDM frequency band.
[0084] The scheduled data 44, which is a stream of bits, is
scrambled in a manner reducing the peak-to-average power ratio
associated with the data using data scrambling logic 46. A cyclic
redundancy check (CPC) for the scrambled data is determined and
appended to the scrambled data using CAC adding logic 48. Next,
channel coding is performed using channel encoder logic 50 to
effectively add redundancy to the data to facilitate recovery and
error correction at the mobile terminal 16. The channel coding for
a particular mobile terminal 16 may be based on the CQI. The
channel encoder logic 50 uses known Turbo encoding techniques in
one embodiment. The encoded data is then processed by rate matching
logic 52 to compensate for the data expansion associated with
encoding.
[0085] Bit interleaver logic 54 systematically reorders the bits in
the encoded data to minimize the loss of consecutive data bits. The
resultant data bits are systematically mapped into corresponding
symbols depending on the chosen baseband modulation by mapping
logic 56. Preferably, Quadrature Amplitude Modulation (QAM) or
Quadrature Phase Shift Key (QPSK) modulation is used. The degree of
modulation is preferably chosen based on the CQI for the particular
mobile terminal. The symbols may be systematically reordered to
further bolster the immunity of the transmitted signal to periodic
data loss caused by frequency selective fading using symbol
interleaver logic 58.
[0086] At this point, groups of bits have been mapped into symbols
representing locations in an amplitude and phase constellation.
Then spatial diversity is desired, blocks of symbols are then
processed by space-time block code (STC) encoder logic 60, which
modifies the symbols in a fashion making the transmitted signals
more resistant to interference and more readily decoded at a mobile
terminal 16. The STC encoder logic 60 will process the incoming
symbols and provide N outputs corresponding to the number of
transmit antennas 28 for the base station 14. The control system 20
and/or baseband processor 22 will provide a mapping control signal
to control STC encoding. At this point, assume the symbols for the
N outputs are representative of the data to be transmitted and
capable of being recovered by the mobile terminal 16. See A. F.
Naguib, N. Seshadri, and A. R. Calderbank, "Applications of
space-time codes and interference suppression for high capacity and
high data rate wireless systems," Thirty-Second Asilomar Conference
on Signals, Systems & Computers, Volume 2, pp. 1803-1810, 1998,
which is incorporated herein by reference in its entirety.
[0087] For the present example, assume the base station 14 has two
antennas 2a (N=2) and the STC encoder logic 60 provides two output
streams of symbols. Accordingly, each of the symbol streams output
by the STC encoder logic 60 is sent to a corresponding IFFT
processor 62, illustrated separately for ease of understanding.
Those skilled in the art will recognize that one or more processors
may be used to provide such digital signal processing, alone or in
combination with other processing described herein. The IFFT
processors 62 will preferably operate on the respective symbols to
provide an-inverse Fourier Transform. The output of the IFFT
processors 62 provides symbols in the time domain. The time domain
symbols are grouped into frames, which are associated with a prefix
by like insertion logic 64. Each of the resultant signals is
up-converted in the digital domain to an intermediate frequency and
converted to an analog signal via the corresponding digital
up-conversion (DUC) and digital-to-analog (D/A) conversion
circuitry 66. The resultant (analog) signals are then
simultaneously modulated at the desired RF frequency, amplified,
and transmitted via the RF circuitry 68 and antennas 28. Notably,
pilot signals known by the intended mobile terminal 11 are
scattered among the sub-carriers. The mobile terminal 16, which is
discussed in detail below, will use the pilot signals for channel
estimation.
[0088] Reference is now made to FIG. 5 to illustrate reception of
the transmitted signals by a mobile terminal 16. Upon arrival of
the transmitted signals at each of the antennas 40 of the mobile
terminal 16, the respective signals are demodulated and amplified
by corresponding RF circuitry 70. For the sake of conciseness and
clarity, only one of the two receive paths is described and
illustrated in detail. Analog-to-digital (A/D) converter and
down-conversion circuitry 72 digitizes and downconverts the analog
signal for digital processing. The resultant digitized signal may
be used by automatic gain control circuitry (AGC) 74 to control the
gain of the amplifiers in the RF circuitry 70 based on the received
signal level.
[0089] Initially, the digitized signal is provided to
synchronization logic 76, which includes coarse synchronization
logic 78, which buffers several OFDM symbols and calculates an
auto-correlation between the two successive OFDM symbols. A
resultant time index corresponding to the maximum of the
correlation result determines a fine synchronization search window,
which is used by fine synchronization logic 80 to determine a
precise framing starting position based on the headers. The output
of the fine synchronization logic 80 facilitates frame acquisition
by frame alignment logic 84. Proper framing alignment is important
so that subsequent FFT processing provides an accurate conversion
from the time to the frequency domain. The fine synchronization
algorithm is based on the correlation between the received pilot
signals carried by the headers and a local copy of the known pilot
data. Once frame alignment acquisition occurs, the prefix of the
OFDM symbol is removed with prefix removal logic 86 and resultant
samples are sent to frequency offset correction logic 88, which
compensates for the system frequency offset caused by the unmatched
local oscillators in the transmitter and the receiver. Preferably,
the synchronization logic 76 includes frequency offset and clock
estimation logic 82, which is based on the headers to help estimate
such effects on the transmitted signal and provide those
estimations to the correction logic 88 to properly process OFDM
symbols.
[0090] At this point, the OFDM symbols in the time domain are ready
for conversion to the frequency domain using FFT processing logic
90. The results are frequency domain symbols, which are sent to
processing logic 92. The processing logic 92 extracts the scattered
pilot signal using scattered pilot extraction logic 94, determines
a channel estimate based on the extracted pilot signal using
channel estimation logic 96, and provides channel responses for all
sub-carriers using channel reconstruction logic 98. In order to
determine a channel response for each of the sub-carriers, the
pilot signal is multiple pilot symbols that are scattered among the
data symbols throughout the OFDM sub-carriers in a known pattern in
both time and frequency. Continuing with FIG. 5, the processing
logic compares the received pilot symbols with the pilot symbols
that are expected in certain sub-carriers at certain times to
determine a channel response for the sub-carriers in which pilot
symbols were transmitted. The results are interpolated to estimate
a channel response for most, if not all, of the remaining
sub-carriers for which pilot symbols were not provided. The actual
and interpolated channel responses are used to estimate an overall
channel response, which includes the channel responses for most, if
not all, of the sub-carriers in the OFDM channel.
[0091] The frequency domain symbols and channel reconstruction
information, which are derived from the channel responses for each
receive path are provided to an STC decoder 100, which provides STC
decoding on both received paths to recover the transmitted symbols.
The channel reconstruction information provides equalization
information to the STC decoder 100 sufficient to remove the effects
of the transmission channel when processing the respective
frequency domain symbols
[0092] The recovered symbols are placed back in order using symbol
de-interleaver logic 102, which corresponds to the symbol
interleaver logic 58 of the transmitter. The de-interleaved symbols
are then demodulated or de-mapped to a corresponding bitstream
using de-mapping logic 104. The bits are then de-interleaved using
bit de-interleaver logic 106, which corresponds to the bit
interleaver logic 54 of the transmitter architecture. The
de-interleaved bits are then processed by rate de-matching logic
108 and presented to channel decoder logic 10 to recover the
initially scrambled data and the CRC checksum. Accordingly, CRC
logic 112 removes the CRC checksum, checks the scrambled data in
traditional fashion, and provides it to the de-scrambling logic 114
for de-scrambling using the known base station de-scrambling code
to recover the originally transmitted data 116.
[0093] In parallel to recovering the data 116, a CQI, or at least
information sufficient to create a CQI at the base station 14, is
determined and transmitted to the base station 14. As noted above,
the CQI in a preferred embodiment is a function of the
carrier-to-interference ratio (CR), as well as the degree to which
the channel response varies across the various sub-carriers in the
OFDM frequency band. For this embodiment, the channel gain for each
sub-carrier in the OFDM frequency band being used to transmit
information are compared relative to one another to determine the
degree to which the channel gain varies across the OFDM frequency
band. Although numerous techniques are available to measure the
degree of variation, one technique is to calculate the standard
deviation of the channel gain for each sub-carrier throughout the
OFDM frequency band being used to transmit data.
[0094] It is to be understood that the FIGS. 1 through 5 are for
the purpose of providing an example system within which the closed
loop MIMO embodiments described below can be applied. More
generally, they can be applied in any system with multiple transmit
antennas and multiple receive antennas, the particular number of
transmit antennas and receive antennas being an implementation
specific decision. Most of the embodiments are applicable to
wide-band multi-carrier systems such as OFDM. However, they may
also be applied to narrow band closed loop MIMO and even to single
carrier closed loop MIMO. It is clear from the description that
follows which embodiments would be applicable to a single carrier
versus a multi-carrier context exclusively.
[0095] The embodiments set forth below represent the necessary
information to enable those skilled in the art to practice the
invention and illustrate the best mode of practicing the invention.
Upon reading the following description in light of the accompanying
drawing figures, those skilled in the art will understand the
concepts of the invention and will recognize applications of these
concepts not particularly addressed herein. It should be understood
that these concepts and applications fall within the scope of the
disclosure and the accompanying claims. While the embodiments can
be applied in a system such as exemplified in FIGS. 1 to 5
described above, embodiments are in no way limited to such
application,
Transmission Formats for Closed Loop MIMO
[0096] Various Transmission formats are provided for closed loop
MIMO air-interface designs. MIMO transmission formats and
signalling apparatus are generalized to allow a variety MIMO
schemes to operate using the same air-interface design. According
to one aspect of the invention, basic transmission formats include:
(1) spatial multiplexing (SM) and (2) space-time transmit diversity
(STTD), with vector or matrix weighted full MIMO or sub-MIMO
transmission based on various transmit antennas configurations.
These formats can be used in single or multi-carrier
configurations. The schemes can also be generalized to cover
multiple base station transmission.
[0097] Some embodiments feature a receiver selecting between
various available transmit formats. The particular formats to
include in a given system are implementation specific.
Spatial Multiplexing (BLAST)
[0098] BLAST is also called spatial multiplexing (SM). This
transmission format allows maximum parallel transmission of data
streams and can achieve a theoretical capacity limit.
[0099] For an open loop M.times.N MIMO SM transmission (M transmit
antennas and N receive antennas), basic transmission matrices are
presented below in Table 1 for up to four transmit antennas. It is
readily apparent how similar transmission matrices can be
constructed for numbers of transmit antennas larger than four.
TABLE-US-00001 TABLE 1 Basic SM matrices 2-transmit 3-transmit
4-transmit matrix-A matrix-B matrix-C Time k Time k Time k Tx.sub.1
[ s 1 s 2 ] ##EQU1## [ s 1 s 2 s 3 ] ##EQU2## [ s 1 s 2 s 3 s 4 ]
##EQU3## Tx.sub.2 Tx.sub.3 Tx.sub.4
[0100] For M transmit antenna SM, there can be N receiver antennas
with N.gtoreq.M. Where N<M for a given receiver, antenna
selection can be performed to identify a subset of the M transmit
antennas for use in BLAST transmission to the receiver. Mechanisms
for performing antenna selection are described below.
Vector Weighted Closed Loop SM Transmission
[0101] For M.times.N MIMO vector weighted SM transmission according
to an embodiment of the invention, the transmission matrices are
presented below in Table 2 for up to four transmit antennas. It is
readily apparent how vector weighted SM matrices can be extended to
handle more than four transmit antennas. With vector weighted
closed loop transmission, each symbol is transmitted on a single
transmit antenna and a single weight is applied to each such
symbol. Other users can then be supported using the remaining
antenna(s). TABLE-US-00002 TABLE 2 Vector weighted SM matrices
2-transmit 3-transmit 4-transmit matrix-A matrix-B matrix-C Time k
Time k Time k Tx.sub.1 [ w 1 .times. s 1 w 2 .times. s 2 ] ##EQU4##
[ w 1 .times. s 1 w 2 .times. s 2 w 3 .times. s 3 ] ##EQU5## [ w 1
.times. s 1 w 2 .times. s 2 w 3 .times. s 3 w 4 .times. s 4 ]
##EQU6## Tx.sub.2 Tx.sub.3 Tx.sub.4
[0102] Vector weighted SM allows, for example, per antenna power
allocation and balancing, SINR optimization from the receiver, and
joint transmit-receive closed loop optimization as a Weiner
solution. For M transmit antenna vector weighted SM, there can be N
receiver antennas with N.gtoreq.M, Where N<M for a given
receiver, antenna selection can be performed to identify a subset
of the M transmit antennas for use in BLAST transmission to the
receiver. Mechanisms for performing antenna selection are described
below. Other users can then be supported using the remaining
antenna(s)
Matrix Weighted Closed Loop SM Transmission
[0103] M.times.N MIMO matrix weighted SM transmission according to
an embodiment of the invention has associated transmission matrices
as shown in Table 3, where 2, 3 and 4 transmit antenna matrices are
shown. As in the previous examples, it is readily apparent how
matrix weighted SM transmission matrices can be extended to handle
more than four transmit antennas. With matrix weighted spatial
multiplexing, each symbol is represented on multiple transmit
antennas with a respective weight. However, depending upon the
weight selected, a given symbol may not necessarily be represented
on all transmit antennas. Different examples of this are presented
below. TABLE-US-00003 TABLE 3 Matrix weighted SM matrices
2-transmit 3-transmit 4-transmit matrix- matrix-A matrix-B C Time k
Time k Time k Tx.sub.1 [ w 1 .times. s 1 + w 2 .times. s 2 w 3
.times. s 1 + w 4 .times. s 2 ] ##EQU7## [ w 1 .times. s 1 + w 2
.times. s 2 + w 3 .times. s 3 w 4 .times. s 1 + w 5 .times. s 2 + w
6 .times. s 3 w 7 .times. s 1 + w 8 .times. s 2 + w 9 .times. s 3 ]
##EQU8## [ w 1 .times. s 1 + w 2 .times. s 2 + w 3 .times. s 3 + w
4 .times. s 4 w 5 .times. s 1 + w 6 .times. s 2 + w 7 .times. s 3 +
w 8 .times. s 4 w 9 .times. s 1 + w 10 .times. s 2 + w 11 .times. s
3 + w 12 .times. s 4 w 13 .times. s 1 + w 14 .times. s 2 + w 15
.times. s 3 + w 16 .times. s 4 ] ##EQU9## Tx.sub.2 Tx.sub.3
Tx.sub.4
[0104] The matrix weighted M.times.N SM transmission can be applied
to a single user reception application where N.gtoreq.M or to the
multi-user concurrent transmission cases, such as
2.times.2.times.1, 4.times.4.times.1, 4.times.2.times.2, for
example. In the above examples, 2.times.2.times.1 for example,
means two transmit antennas, two users, and each user having one
receive antenna. Where N<M for a given receiver, antenna
selection can be performed to identify a subset of the M transmit
antennas for use in BLAST transmission to the receiver. Mechanisms
for performing antenna selection are described below. Other users
can then be supported using the remaining antenna(s).
[0105] Tables 4 to 6 and FIGS. 6 and 7 below provide illustrative
examples of single and multi-user configurations of vector and
matrix weighted SM beam-formers. Referring first to Table 4, this
is a set of example configurations for the two transmit antenna
case, i.e., using the 2-transmit matrix-A of Table 3 describe
above. Two different transmissions formats are shown with format-1
shown in the first row and format-2 shown in the second row. With
format-1 a single user is assigned to the entire stream mapping.
With format-2, a first user is assigned to symbol s.sub.1 in the
stream mapping and a second user is assigned to a symbol 52 in the
stream mapping. Thus, either one user is given both symbols of each
pair, or two users are each given a single symbol in each pair. The
transmit beam-forming that is performed is the same in both cases,
however, the weights/feedback information will be different for one
user as opposed to two users. The three right hand columns in Table
4 show various options for feeding back channel information. These
will be expanded upon in much more detail below. For this example
and the other examples below, the feedback options shown consist of
feeding back the channel matrix or a portion thereof for a given
user, with the form of the channel matrix or portion thereof being
shown in the table, or feeding back unitary matrix information,
vector information that may be quantized. The symbol "x" simply
indicates that this option is available for each of the formats.
With the example of Table 4, for format-1 the entire channel matrix
can be fed back from the single receiver to the transmitter.
Format-2, a first user would only transmit back a portion of the
channel matrix, namely h.sub.11, h.sub.12. The other user would
similarly transmit a portion of the channel matrix back, namely
h.sub.21, h.sub.22. In the table, the values w.sub.1, w.sub.2,
w.sub.3 and w.sub.4 are the weights that are applied in transmit
beam-forming. These weights are determined as a function of the
feedback information received. These weights are described in much
more detail below. Note that in a typical implementation, only one
of the various feedback options would be implemented. While the
table shows three different feedback options, a larger number of
feedback options are described below and any of these can be used
in a given implementation. TABLE-US-00004 TABLE 4 Configuration of
Matrix weighted SM matrix-A User Stream Transmit Channel Unitary
Compressed Assignment Mapping Beam-forming Matrix Matrix/Vecter
Quantizer FORMAT-1 User-1 [ s 1 s 2 ] ##EQU10## [ w 1 .times. s 1 +
w 2 .times. s 2 w 3 .times. s 1 + w 4 .times. s 2 ] ##EQU11## [ h
11 h 12 h 21 h 22 ] ##EQU12## x x FORMAT-2 User-1 user-2 [ s 1 s 2
] ##EQU13## [ h 11 h 12 ] ##EQU14## [ h 21 h 22 ] ##EQU15## x x x
x
[0106] FIG. 6 provides an example of beam-forming SM transmission
of matrix weighted matrix-A, wherein a first configuration
generally indicated by 100 corresponds to a single user format with
matrix weighting and a second configuration generally indicated by
110 corresponds to a multi-user format with matrix weighting.
[0107] The first configuration 100 is for a single user. The user
input is indicated at 102, this consisting of two symbols x.sub.1,
x.sub.2 at a given instant. These two inputs are then multiplied
beam-forming weights in a beam-former 104 that implements
beam-forming matrix G.sub.1 104. The output of the beam-former 104
is then fed to a pair of transmit antennas 106. In the event of a
matrix weighting, the beam-forming matrix corresponds to that of
matrix-A for Table 3 above. A similar structure exists in the event
of vector weighting (not shown), in which case the beam-forming
matrix would correspond to matrix-A of Table 2 described above.
[0108] For the second configuration, generally indicated at 110,
there are two separate user inputs indicated at 112, 114
respectively. Each of the two user inputs are multiplied by a
respective vector with the first input 112 being multiplied by
V.sub.1 113, and the second input being multiplied by V.sub.2 115.
The outputs of these vector multiplications are summed on a per
element basis in adders 116,118. The output of the first adder 116
is fed to the first transmit antenna 120 and the output of the
second adder 118 is fed to the second transmit antenna 122. The two
vector multiplications V.sub.1 113, V.sub.2 115 collectively
implement the beam-forming operation of matrix-A of Table 3. Given
that it is a multi-user format, it also corresponds to the
transmission format-2 of Table 4 above.
[0109] Table 5 below provides similar information to that described
in Table 4 above, but for a three transmit antenna case. In other
words, this is elaborating the details of transmit formats
available for the matrix weighted SM matrix-B described above in
Table 3. The options available with a three transmit case are to
use all three streams for a single user as indicated at format-1 in
the first row, to use one of the three symbol streams for a first
user and the remaining two symbol streams for a second user as
shown in format-2 in the second row, and to assign a different user
to each of the three streams as shown in format-3 in the third row.
In all cases, the transmit beam-forming matrix has the same
structure. The feedback options are again listed in the right hand
three columns of Table 5. TABLE-US-00005 TABLE 5 Configuration of
Matrix weighted SM matrix-B FEEADBACK OPTIONS TRANSMISSION FORMATS
Unitary User Stream Transmit Beam- Channel Matrix/ Compressed
Assignment Mapping forming Matrix Vector Quantizer FORMAT-1 User-1
[ s 1 s 2 s 3 ] ##EQU16## [ h 11 h 12 h 13 h 21 h 22 h 23 h 31 h 32
h 33 ] ##EQU17## [ h 11 h 12 h 13 h 21 h 22 h 23 h 31 h 32 h 33 ]
##EQU18## x x FORMAT-2 User-1 [ s 1 s 2 s 3 ] ##EQU19## [ h 11 h 12
h 13 ] ##EQU20## x x User-2 [ h 11 h 12 h 13 h 21 h 22 h 23 ]
##EQU21## x x FORMAT-3 User-1 [ s 1 s 2 s 3 ] ##EQU22## [ h 11 h 12
h 13 ] ##EQU23## x x User-2 [ h 21 h 22 h 23 ] ##EQU24## User-3 [ h
31 h 32 h 33 ] ##EQU25##
[0110] Similarly, the configurations available for a four transmit
antenna configuration with matrix weighting are shown in Table 6
below. A single user format is shown in the first row; two user
formats are shown in the second and fourth rows, a three user
format is shown in the third row, and a four user format is shown
in the fifth row. It is to be understood that not all of the
potential formats have been exhaustively shown in Table 6. In all
cases, the transmit beam-forming matrix has the same structure, and
the feedback options for the different formats are shown in the
right hand three columns of Table 6. TABLE-US-00006 TABLE 6
Configuration of Matrix weighted SM matrix-C FEEDBACK OPTIONS
TRANSMISSION FORMATS Unitary User Stream Matrix/ Compressed
Assignment Mapping Transmit Beam-forming Channel Matrix Vector
Quantizer FORMAT-1 User-1 [ s 1 s 2 s 3 s 4 ] ##EQU26## [ w 1
.times. s 1 + w 2 .times. s 2 + w 3 .times. s 3 + w 4 .times. s 4 w
5 .times. s 1 + w 6 .times. s 2 + w 7 .times. s 3 + w 8 .times. s 4
w 9 .times. s 1 + w 10 .times. s 2 + w 11 .times. s 3 + w 12
.times. s 4 w 13 .times. s 1 + w 14 .times. s 2 + w 15 .times. s 3
+ w 16 .times. s 4 ] ##EQU27## [ h 11 h 12 h 13 h 14 h 21 h 22 h 23
h 24 h 31 h 32 h 33 h 34 .times. h 41 h 42 h 43 h 44 ] ##EQU28## x
x FORMAT-2 User-1 [ s 1 s 2 s 3 s 4 ] ##EQU29## [ h 11 h 12 h 13 h
14 h 21 h 22 h 23 h 24 ] ##EQU30## x x User-2 [ h 31 h 32 h 33 h 34
.times. h 41 h 42 h 43 h 44 ] ##EQU31## x x FORMAT-3 User-1 [ s 1 s
2 s 3 s 4 ] ##EQU32## [ h 11 h 12 h 13 h 14 ] ##EQU33## x x User-2
[ h 21 h 22 h 23 h 24 ] ##EQU34## x x User-3 [ h 31 h 32 h 33 h 34
.times. h 41 h 42 h 43 h 44 ] ##EQU35## x x FORMAT-4 User-1 [ s 1 s
2 s 3 s 4 ] ##EQU36## [ h 21 h 22 h 23 h 24 ] ##EQU37## x x User-2
x x FORMAT-5 User-1 [ s 1 s 2 s 3 s 4 ] ##EQU38## [ h 11 h 12 h 13
h 14 ] ##EQU39## x x User-2 [ h 21 h 22 h 23 h 24 ] ##EQU40## x x
User-3 [ h 31 h 32 h 33 h 34 .times. ] ##EQU41## x x User-4 [ h 41
h 42 h 43 h 44 ] ##EQU42## x x
[0111] FIG. 7 shows two examples of beam-forming SM transmission of
Matrix weighted matrix-C. A first configuration, generally
indicated at 130, corresponds to format-1 of Table 6 above, namely
a format in which a single user is transmitting on all four
antennas. The single user's data inputs are indicated at 132; these
are processed by beam-forming matrix G.sub.1 134 the output of
which is transmitted on four antennas 136.
[0112] The second configuration shown in FIG. 2, generally
indicated at 140, corresponds to format-2 of Table 6 described
above, namely a format in which there are two users each of which
occupy two of the streams. The first user's input is indicated at
142, and the second user's input is indicated at 144. The entire
arrangement indicated at 146 performs a beam-forming operation on
the two user inputs. This involves multiplying user 1 input by
matrix G.sub.1 147 and user 2 input 144 by matrix G.sub.2 148. The
output to these two multiplication operations are added in adders
150,152,154,156 and then output on antennas 158,160,162,164. It
should be readily apparent how similar the structure can be created
for each of the other available formats described above.
[0113] The above described BLAST vector and weighted transmission
configurations are generally applicable for closed loop MIMO
systems with 2, 3 and 4 antennas. It should be readily apparent how
these can be extended to handle a larger number of antennas.
[0114] In a multi-carrier environment, these transmission
configurations can be applied either on a per-carrier basis, for
blocks of contiguous carriers, sets of non- or partially-contiguous
carriers, or for an entire transmit bandwidth. Thus, for the OFDM
particular implementations, in some implementations, a different
beam-forming matrix/user assignment can be applied on a per-carrier
basis; in other implementations, the available OFDM bandwidth is
divided into sub-bands, and the same beam-forming matrix/user
assignment is used on each sub-band; and yet a further available
option, the entire OFDM bandwidth is used to transmit with the same
beam-forming configuration. In yet another option, the entire
available OFDM bandwidth divided into small sub-channels, with each
user being assigned a respective set of such sub-channels.
Spatial Multiplexing when there are More Transmit Antennas than
Receive Antennas
[0115] When there are extra transmit antennas, antenna selection
and signal pre-processing can be used to significantly enhance link
level performances. The SNR relation is: where
.gamma..sub.pp>.gamma..sub.as>>.gamma..sub.fa, where
.gamma..sub.pp, .gamma..sub.as, and .gamma..sub.fa are the SNRs of
pre-processing, antenna selection, and fixed antenna configuration
(using M fixed transmit antennas from a pool of N). The gain of
signal pre-processing over antenna selection is mainly from the use
of beam-forming. Details of how feedback can be performed for
beam-forming are given below. The gain of antenna selection over
fixed antenna configuration is from two areas: diversity gain and
average SNR gain (over STC). Preferably, the water-filling
principle (the principle of using the most energy where conditions
are the best) is applied to both antenna selection and
beam-forming, Antenna selection is the most simple and effective
way to achieve diversity and SNR gain.
Antenna Selection for Spatial Multiplexing
[0116] With antenna selection, the receiver analyzes conditions on
the channel and selects antennas that are best for that receiver.
This antenna selection information is then fed back to the
transmitter for use in performing the actual transmissions.
Antenna Selection for a Single User With Single Antenna
[0117] A single stream selective transmit diversity technique is
represented in Table 7 below. This example assumes that there are
four transmit antennas available to be selected for a single
receive antenna user. The four different transmit options are shown
in the table to include: the user's symbol s.sub.1 being
transmitted either using the first transmit antenna Tx.sub.1, or
the second, third or fourth transmit antenna similarly labelled. In
order for the transmitter to know which antenna to use, preferably
a feedback signal from the receiver is employed. In the particular
example shown, a two bit feedback signal is shown in the first
column that can be used to identify one of four different possible
transmit antennas. Other formats for the feedback information can
be employed. When a single antenna is used, there is no spatial
multiplexing.
[0118] Once again for a multi-carrier implementation, antenna
selection can take place to any appropriate sub-carrier resolution.
The particular feedback signalling is one particular example of how
to signal back an identity of one of four antennas. An appropriate
signalling mechanism can be employed. Furthermore, the antenna
selection mechanism can be extended to differing numbers of
transmit antennas. TABLE-US-00007 TABLE 7 SISO selective transmit
diversity Tx.sub.1 Tx.sub.2 Tx.sub.3 Tx.sub.4 b.sub.0b.sub.1 = 00
Time t s.sub.1 0 0 0 b.sub.0b.sub.1 = 01 Time t 0 s.sub.1 0 0
b.sub.0b.sub.1 = 10 Time t 0 0 s.sub.1 0 b.sub.0b.sub.1 = 11 Time t
0 0 0 s.sub.1
2-Stream Spatial Multiplexing for Single User
[0119] An embodiment of the intention proposes closed loop sub-MIMO
selection for SM. The term "sub-MIMO" is used to describe the case
where there are multiple transmit antennas and a fewer number of
receive antennas, and antenna selection is used to select a subset
of the transmit antennas for use by the given receiver with the
remaining antennas being available for other receivers. This will
be described in the context of a 4.times.2 sub-MIMO system, in
which the channel matrix to a given receiver has the following
format: H = [ h 11 h 12 h 13 h 14 h 21 h 22 h 23 h 24 ] . ##EQU43##
Selecting two of the four antennas results in a smaller 2.times.2
channel matrix for the given receiver. There are six different
possible channel matrices depending upon which of the two of the
four transmit antennas are selected. The six possible channel
matrices are as follows: H 12 = [ h 11 h 12 h 21 h 22 ] ##EQU44## H
13 = [ h 11 h 13 h 21 h 23 ] ##EQU44.2## H 23 = [ h 12 h 13 h 22 h
23 ] ##EQU44.3## H 14 = [ h 11 h 14 h 21 h 24 ] ##EQU44.4## H 24 =
[ h 12 h 14 h 22 h 24 ] ##EQU44.5## H 34 = [ h 13 h 14 h 23 h 24 ]
##EQU44.6## These six potential channel matrices can then be
analyzed to determine which sub-MIMO channel is best, i.e., which
of the two four transmit antennas are best for the particular user.
Any appropriate way of selecting the two antennas can be
implemented. In the particular example that follows, antenna
selection is performed on the basis of determinants calculated for
the different channel matrices. The particular pair of antennas
that is best for a given receiver is determined by selecting the
sub-system H.sub.ij that satisfies
|det(H.sub.ij)|=max{|det(H.sub.12)|,|det(H.sub.13)|,|det(H.sub.23)|,|det(-
H.sub.14)|,|det(H.sub.24)|,|det(H.sub.34)|}. This is depicted in
FIG. 8 where a transmitter has four transmit antennas 190, and a
receiver has two receive antennas 192. The determinant method is
employed to select two of the four transmit antennas 190 for
transmitting to the receiver.
[0120] More generally, in a K receive antenna system, K columns
from channel matrix II can be selected, where K is the size of the
subset antenna group. There are a total of C.sup.K.sub.M choices.
For each possible selection, calculate det(H'H). Finally, select
the antenna group with max(det(H'H)).
[0121] As in previous discussions, for multi-carrier applications,
antenna selection can be used at whatever frequency resolution may
be determined for a given application. The best antenna group may
differ for different sub-carriers. If no other users are
simultaneously being transmitted to, null sub-carriers are
preferably fed into non-selected antennas. This antenna selection
process is typically performed at the receiver. Once the best two
transmit antennas have been selected at the receiver, the selection
is fed back to the transmitter. An example of how this feedback can
be performed is shown in Table 8 below. The transmit contents that
are possible are shown for each of the four transmit antennas. It
can be seen that there are six different permutations depending
upon which two transmit antennas are selected. The first column is
used to show how three bits can be used to transmit a selection of
two of the six antennas. Other feedback formats are possible. In
some implementations feed back is for every sub-carrier. If three
bits are fed back on a per sub-carrier basis, then for L
sub-carriers, L.times.3 bits would be required, in some
implementations, feed back is for a sub-channel of contiguous
sub-carriers. If three bits are fed back for a contiguous block of
sub-carriers, under the assumption that channel conditions are
substantially constant across the block, obviously fewer feedback
bits are required. TABLE-US-00008 TABLE 8 2-Layer SM with
Beam-forming with antenna selection Tx.sub.1 Tx.sub.2 Tx.sub.3
Tx.sub.4 b.sub.0b.sub.1b.sub.2 = 000 Time t w.sub.1s.sub.1
w.sub.2s.sub.2 0 0 b.sub.0b.sub.1b.sub.2 = 001 Time t
w.sub.1s.sub.1 0 w.sub.2s.sub.2 0 b.sub.0b.sub.1b.sub.2 = 010 Time
t 0 w.sub.1s.sub.1 w.sub.2s.sub.2 0 b.sub.0b.sub.1b.sub.2 = 011
Time t w.sub.1s.sub.1 0 0 w.sub.2s.sub.2 b.sub.0b.sub.1b.sub.2 =
100 Time t 0 w.sub.1s.sub.1 0 w.sub.2s.sub.2 b.sub.0b.sub.1b.sub.2
= 101 Time t 0 0 w.sub.1s.sub.1 w.sub.2s.sub.2
[0122] In this case, the sub-MIMO channel selection is based on the
sub-MIMO channel condition and the antenna power allocation for the
SM data can be adjusted by the feedback information pertaining to
the weights.
[0123] Antenna group index (illustratively 3 bits), and
proportional weighting vector (illustratively 2 real elements (or
alternatively complex elements), each element being of 4-bits,
6-bits, or 8-bits) are preferably provided as feedback, signals. It
should be appreciated, however, that weights may be derived from
feedback information and need not necessarily be directly provided
in feedback information.
Closed Loop Sub-MIMO Selection with Beam-Forming
[0124] For a DL 4.times.2 MIMO system, consider six sub-MIMO
systems H.sub.1,2, H.sub.1,3, M.sub.1,4, H.sub.2,3, H.sub.2,4, and
H.sub.3,4. Assuming that H.sub.ij, H.sub.ik and H.sub.il are the
sub-MIMO systems that satisfy:
|det(H.sub.ij)|+|det(H.sub.ik)|+|det(H.sub.il)|=max{|det(H.sub.li)|+|det(-
H.sub.lk)|+|det(H.sub.ll)|,|det(H.sub.lj)|+|det(H.sub.jk)|+|det(H.sub.jl)|-
} then by beam-forming with the j.sup.th and k.sup.th columns of H,
and setting the weights to w j = det * .function. ( H ij ) det *
.function. ( H il ) 2 + det * .function. ( H ik ) 2 + det *
.function. ( H il ) 2 ##EQU45## w k = det * .function. ( H ik ) det
* .function. ( H il ) 2 + det * .function. ( H ik ) 2 + det *
.function. ( H il ) 2 ##EQU45.2## w l = det * .function. ( H il )
det * .function. ( H il ) 2 + det * .function. ( H ik ) 2 + det *
.function. ( H il ) 2 ##EQU45.3## respectively, where the
particular selection of the three determinants maximizes the
denominator in the above-weight calculation we have
det(H.sub.ij.sup.(jkl))= {square root over
(|det*(H.sub.ij)|.sup.2+|det*(H.sub.lk)|.sup.2+|det*(H.sub.il)|.sup.2)}.
[0125] Table 9 is an illustrative example of this. A single antenna
is used to transmit a stream without any weighting; another antenna
is used to transmit the same stream with weighting; and the
remaining two antennas are used to transmit the other second
stream, both with weighting. The particular antenna used to
transmit either the un-weighted symbol or the weighted symbols is
information that is preferably fed back to the transmitter. The
weighting vector or information from which the weights can be
derived is also preferably fed back from the receiver to the
transmitter. Thus, the illustrated example, this would require the
equivalent of three bits to identify one of the six permutations,
and additional feedback for the three weights w.sub.1, w.sub.2 and
w.sub.3. This format is a hybrid of beam-forming and spatial
multiplexing. In this example, there are two data streams mapped
onto four antennas. Each symbol is transmitted on two different
antennas with the weights as shown in the table below. Antenna
selection is performed to select which antennas go together to
transmit the same symbol, and beam forming is used to apply weights
to three of the four antennas. Four weights could be applied
alternatively. It should be readily apparent how this concept can
be extended to handle larger numbers of antennas overall and/or
larger numbers of antennas per stream and/or larger numbers of
streams. This approach will be referred to as K-layer SM employing
more than K transmit antennas. In the example, we have 2-layer SM
with four transmit antennas, TABLE-US-00009 TABLE 9 2-Layer SM with
antenna selection and beam-forming Tx.sub.1 Tx.sub.2 Tx.sub.3
Tx.sub.4 b.sub.0b.sub.1b.sub.2 = 000 Time t s.sub.1 w.sub.1s.sub.2
w.sub.2s.sub.1 w.sub.3s.sub.2 b.sub.0b.sub.1b.sub.2 = 001 Time t
s.sub.1 w.sub.2s.sub.1 w.sub.1s.sub.2 w.sub.3s.sub.2
b.sub.0b.sub.1b.sub.2 = 010 Time t s.sub.1 w.sub.1s.sub.1
w.sub.2s.sub.2 w.sub.3s.sub.2 b.sub.0b.sub.1b.sub.2 = 011 Time t
w.sub.3s.sub.2 s.sub.1 w.sub.1s.sub.2 w.sub.2s.sub.1
b.sub.0b.sub.1b.sub.2 = 100 Time t w.sub.3s.sub.2 s.sub.1
w.sub.2s.sub.1 w.sub.1s.sub.2 b.sub.0b.sub.1b.sub.2 = 101 Time t
w.sub.3s.sub.2 w.sub.2s.sub.1 s.sub.1 w.sub.1s.sub.2
STTD Transmission Formats
[0126] The transmission formats described thus far have all been
based on the BLAST approach in which there is a single transmitted
symbol for each information symbol, or on the BLAST variant
described by way of example with reference to Table 9 above in
which each K-stream blast is employed on more than K antennas, so
at least some of the symbols are transmitted twice. Additional
transmission formats are based on STTD (space time transmit
diversity) in which there are at least two transmitted symbols
based on each information symbol, and typically sets of two
information symbols are transmitted using 2.times.2 Alamouti block.
However, other STTD formats are also contemplated.
Single Stream Weighted STTD (SSTD) for a Single User with Antenna
Selection and Proportional Weighting
[0127] In another transmission format, for a single data steam,
transmit diversity is "proportionally weighted" STTD. This is a
middle approach between conventional STTD (no weighting) and
antenna switching. One advantage of this scheme is to safeguard
against possible feedback error, feedback delay, and possible
channel change between feedback updates for the antenna selection.
The channel matrix condition (CMC) does not affect STTD when
orthogonal codes are used. For quasi-orthogonal codes, CMC does
affect signal quality.
[0128] In the case of single stream transmit diversity, we also
have the two options of beam-forming and antenna selection that can
be performed similarly to that described above for the BLAST
transmission formats. Beam-forming involves applying weights to the
elements of the STTD groups, and antenna selection involves
selecting a sub-set of a set of available transmit antennas for
transmitting the stream.
[0129] For an SSTD system with four transmit antennas, a hybrid
proportionally weighted STTD approach can be used. In one
embodiment, the two antennas with the strongest combined SNR are
selected, and proportional weighting on these two selected antennas
is then applied. The advantages of this scheme include; diversity
and SNR gain due to antenna selection, increased robustness
provided by proportional weighting, and the employment of efficient
Alamouti codes. An example of SSTD with antenna selection and
proportional weighting is given in Table 10, where two antennas out
of four are selected and weighted, TABLE-US-00010 TABLE 10 SSTD
with antenna selection and proportional weighting Tx.sub.1 Tx.sub.2
Tx.sub.3 Tx.sub.4 b.sub.0b.sub.1b.sub.2 = 000 Time t w.sub.1s.sub.1
w.sub.2s.sub.2 0 0 Time (t + T) -w.sub.1s.sub.2* w.sub.2s.sub.1* 0
0 b.sub.0b.sub.1b.sub.2 = 001 Time t w.sub.1s.sub.1 0
w.sub.2s.sub.2 0 Time (t + T) -w.sub.1s.sub.2* 0 w.sub.2s.sub.1* 0
b.sub.0b.sub.1b.sub.2 = 010 Time t 0 w.sub.1s.sub.1 w.sub.2s.sub.2
0 Time (t + T) 0 -w.sub.1s.sub.2* w.sub.2s.sub.1* 0
b.sub.0b.sub.1b.sub.2 = 011 Time t w.sub.1s.sub.1 0 0
w.sub.2s.sub.2 Time (t + T) -w.sub.1s.sub.2* 0 0 w.sub.2s.sub.1*
b.sub.0b.sub.1b.sub.2 = 100 Time t 0 w.sub.1s.sub.1 0
w.sub.2s.sub.2 Time (t + T) 0 -w.sub.1s.sub.2* 0 w.sub.2s.sub.1*
b.sub.0b.sub.1b.sub.2 = 101 Time t 0 0 w.sub.1s.sub.1
w.sub.2s.sub.2 Time (t + T) 0 0 -w.sub.1s.sub.2*
w.sub.2s.sub.1*
[0130] In this case, feedback signals preferably include antenna
group index (illustratively 3 bits), proportional weighting vector
or other information allowing the determination of the weighting
vector (illustratively 2 real elements, each element might for
example be of 4-bits, 6-bits, or 8-bits). It should be readily
apparent how the single stream case can be extended to larger
numbers of transmit antennas.
Double Stream Transmit Diversity (D-STTD) for a Single User
[0131] With double stream transmit diversity, two streams are
transmitted to a single user with each stream being sent using a
respective weighted Alamouti code structure (or some other STTD
structure). More generally, with an M transmit antenna system, M/2
STTD sub-groups can be formed and transmitted to a different
receiver, although multiple or all of the sub-groups might be sent
to one receiver. This is a hybrid case of STTD and
space-multiplexing (SM). Because of inter-code interference,
antenna grouping becomes more important. Preferably, layer based
water-filling is employed to provide additional gain to the system.
With layer-based water-filling, more power is transmitted to the
layer having the better channel. Thus, with a double STTD system, a
respective pair of antennas is assigned to each stream, and the
respective "layer" is transmitted on that pair of antennas.
Preferably, more power is given to the layer having the better
channel conditions.
[0132] An example of a set of possible layer weighting permutations
is shown in Table 11 for the four transmit antenna case. In the
first row, transmit antennas 1 and 2 are grouped together, and
antenna 3 and 4 are grouped together, to form two constituent STTD
codes that can be assigned to respective receivers or to a single
receiver. In the example of Table 11 below, symbols s.sub.1 and
s.sub.2 are of one layer, while symbols s.sub.3 and s.sub.4 are of
a second layer. TABLE-US-00011 TABLE 11 Double STTD with layer
weighting Tx.sub.1 Tx.sub.2 Tx.sub.3 Tx.sub.4 b.sub.0b.sub.1 = 00
Time t w.sub.1s.sub.1 w.sub.1s.sub.2 w.sub.2s.sub.3 w.sub.2s.sub.4
Time (t + T) -w.sub.1s.sub.2* w.sub.1s.sub.1* -w.sub.2s.sub.4*
w.sub.2s.sub.3* b.sub.0b.sub.1 = 01 Time t w.sub.1s.sub.1
w.sub.2s.sub.3 w.sub.1s.sub.2 w.sub.2s.sub.4 Time (t + T)
-w.sub.1s.sub.2* -w.sub.2s.sub.4* w.sub.1s.sub.1* w.sub.2s.sub.3*
b.sub.0b.sub.1 = 10 Time t w.sub.1s.sub.1 w.sub.2s.sub.3
w.sub.2s.sub.4 w.sub.1s.sub.2 Time (t + T) -w.sub.1s.sub.2*
-w.sub.2s.sub.4* w.sub.2s.sub.3* w.sub.1s.sub.1*
[0133] The feedback signals in this case are used to select one of
the three different permutations, and this can take the form of an
antenna group index (2 bits in the example shown), and weights or
information relating to weights. In this implementation, there are
two STTD water-filling weights and each can be fed back to an
implementation specific resolution, for example, 4-bits, 6-bits, or
8-bits. The water-filling weights determine the relative amount of
power used on the two STTD groups.
[0134] In summary, the options for STTD transmissions for a single
user include a pair of antennas to transmit a single layer, or
multiple pairs of transmit antennas each used to transmit a
respective layer, in each of these cases each layer being
transmitted using STTD. Preferably, for multi-carrier applications,
antenna selection/grouping is performed on a per sub-carrier basis.
Alternatively, it can be performed for some other sub-carrier
breakdown. In the event an entire OFDM band, this would be referred
to as "fixed" STTD, while in the event of antenna grouping on a per
sub-carrier basis, this is referred to as antenna grouping D-STTD,
FIG. 9 shows a comparison of the performance of fixed D-STTD and
antenna grouping D-STTD for an OFDM system. It can be seen that
over a range of sub-carriers, the fixed D-STTD will perform
similarly to the antenna grouping D-STTD on sub-carriers where the
particular fixed antennas perform the best. Elsewhere, the antenna
grouping D-STTD will perform better. In the example of FIG. 9, on a
per sub-carrier basis, an antenna grouping has been performed. The
feedback required for this implementation would be two bits per
sub-carrier to choose the antennas. Preferably, weights are also
fed back if there is available bandwidth for this.
[0135] In addition to antenna grouping, antenna selection can also
be used to eliminate some of the antennas and only use a single
antenna per STTD block. This will make a 4.times.2 double STTD
system degenerate to a 2.times.2 BLAST system. In fact, this can be
viewed as applying antenna selection to each STTD sub-code--after
the rule of antenna grouping has been applied.
[0136] For example, if antenna selection is also used with the
W-STTD format in the second row of Table 11 to eliminate the first
and fourth transmit antennas, the result is something that is
effectively equivalent to a 2.times.2 BLAST system as shown in
Table 12 below. TABLE-US-00012 TABLE 12 Weighted BLAST degenerated
from double STTD Tx.sub.1 Tx.sub.2 Tx.sub.3 Tx.sub.4 Time t 0
w.sub.2s.sub.3 w.sub.1s.sub.2 0 Time (t + T) 0 -w.sub.2s.sub.4*
w.sub.1s.sub.1* 0
[0137] In another embodiment, proportional weighting is applied to
each constituent STTD code after antenna grouping, rather than or
in addition to for the STTD codes as a whole as was the case in
Table 11 above. An example of weighted double STTD with antenna
grouping and proportional weighting is given in Table 13 below.
There are two levels of weighting in this design: water-filling is
applied across layers (i.e. across constituent STTD codes), and
proportional weighting is applied within each constituent STTD
code. In other words, weighting is sued to get the best allocation
of the Alamouti pairs among the antennas, and weighting is used to
weight the antennas of a given pair differently. In the table
below, both of these weights are included within the single weight
w.sub.i applied to each symbol. In the example below, antennas 1
and 3 have been grouped to form a first STTD group, and antennas 2
and 4 have been grouped to form a second STTD group. TABLE-US-00013
TABLE 13 Weighted double STTD with antenna grouping and
proportional weighting of each element Tx.sub.1 Tx.sub.2 Tx.sub.3
Tx.sub.4 Time t w.sub.1s.sub.1 w.sub.2s.sub.3 w.sub.3s.sub.2
w.sub.4s.sub.4 Time (t + T) -w.sub.1s.sub.2* -w.sub.2s.sub.4*
w.sub.3s.sub.1* w.sub.4s.sub.3*
[0138] Referring now to FIG. 10, shown is a very specific example
of how any of the above described transmission formats can be
allocated to a pair of users occupying an OFDM band. In this case,
there are four transmit antennas 170,172,174,176, and there are two
receivers indicated at 178 and 180. Each receiver has a respective
pair of receive antennas. With the particular example shown,
antenna selection is used to select the first and fourth antennas
170,176 for transmitting to a second receiver 180 and to select the
second and third antennas 172,174 for transmission to the first
receiver 178. Furthermore, the OFDM sub-carriers that are allocated
to the two receivers are indicated at 182 for the first receiver
and 184 for the second receiver. It can be seen that for this
example, the sub-carriers are allocated in a non-overlapping
fashion such that a different set of sub-carriers are transmitted
to each of the two receivers 178,180. In this example, the
sub-carriers are allocated to antennas in groups of six. However,
more generally, sub-carrier allocation to particular antennas, and
antenna assignment to particular users can be performed on a per
sub-carrier basis, or to any other desired resolution.
[0139] Only a subset of antennas/sub-channels are selected for
transmitting signals to a given receiver. Receivers only need to
feedback antennas/sub-channels group indexes to the transmitter. In
the event weighting is also employed, then this case, weighting
information might be fed back for each sub-carrier, or for groups
of six sub-carriers.
[0140] In multi-user applications, antenna/sub-channel selection is
based on multi-user diversity; different users may select different
antenna groups. Quasi-water-filling over is preferably employed in
selecting antennas/sub-channels: transmitting power on good
channels and antennas, while avoiding poor channels. Power
balancing can be achieved through multi-user diversity. Since
different receivers experience different channel environments,
their antenna group selection will not necessarily the same. When
different receivers select different antenna groups, the average
power transmitted by each transmit antenna can be somewhat
balanced.
Antenna Selection for STTD Formats and SM Formats
[0141] The above STTD and SM options have all featured weighting,
either in the form of water-filling across different STTD groups/SM
antennas and/or proportional weighting within STTD groups. Antenna
selection/antenna grouping can be combined with these weighting
options. In another embodiment, antenna selection/antenna grouping
alone is employed. Several examples of pure antenna selection will
now be described.
Antenna Selection/Grouping for three Transmit Antennas
[0142] Example STTD formats for three transmit antenna applications
will now be described. In the example of FIG. 11, each 4.times.3
matrix has three rows and each row is transmitted on a respective
transmit antenna. In the first example matrix A.sub.1, the first
two symbols s.sub.1, s.sub.2 are transmitted using STC coding on
the first two antennas and during two transmit intervals, and the
next two symbols s.sub.3, s.sub.4 are transmitted using space time
coding on the first and third antennas during the next two transmit
intervals. The matrices A.sub.2 and A.sub.3 show other permutations
for selecting two of the antennas for each STC block. The matrices
A.sub.1, A.sub.2 and A.sub.3 are single stream transmission
matrices. During four transmission intervals, four symbols s.sub.1,
s.sub.2, s.sub.3, s.sub.4 are transmitted.
[0143] Another antenna grouping example is shown in FIG. 11 for
matrices B.sub.1, B.sub.2, B.sub.3. This is a two stream example.
It can be seen that one of the streams consists of s.sub.1,
s.sub.2, s.sub.3, s.sub.4 transmitted over the four transmission
intervals, and the other stream consists of symbols s.sub.5,
s.sub.6, s.sub.7, s.sub.8 transmitted over the four symbol
transmission intervals. In this case, space time coding is applied
for both streams, but diversity is applied for one of the streams
but not for the other. The three matrices B.sub.1, B.sub.2, B.sub.3
show different ways of assigning the antennas to the various
streams.
[0144] FIG. 12 shows examples of antenna groupings and antenna
selections for closed loop STC/MIMO 3-transmit antenna
arrangements. Also shown in FIG. 12 is a set of examples of
antennas selection feedback for a three antenna system. For a
single stream application, the receiver simply selects one of three
antennas, and feeds back an identification of the selected antenna.
Thus, C.sub.1 represents the selection of the first antenna,
C.sub.2 represents the selection of the second antenna, and C.sub.3
represents the selection of the third antenna. The transmission
matrix also shows a term "c" which represents the amount of power
boosting that can be implemented. In this case, such a single
antenna is used for the particular user, the entire power for that
user can be applied to that single antenna and no power boosting
need be employed. An example of CQICH bits that might be fed back
to identify one of the three permutations is also shown.
[0145] Also shown is a two stream example where the receiver must
select two of the three possible antennas, and feedback to the
transmitter which two of the three antennas it would like to
receive its data on. In this case, a power boosting factor of 1/
{square root over (2)} such that the total amount of power is
divided across the two antennas for the given user.
Antenna Selection/Grouping for Four Transmit Antennas
[0146] FIG. 13 shows a set of example antenna groupings for
4-transmit closed loop STC/MIMO. Shown are examples for one stream,
two streams and four streams. The possible matrices for one stream
are indicated at A.sub.1, A.sub.2, A.sub.3. In each of these
examples, a stream consisting of s.sub.1, s.sub.2, s.sub.3, s.sub.4
is transmitted using all four antennas, with space time coding
being applied between s.sub.1 and s.sub.2, on two antennas and
between s.sub.3, and s.sub.4 over two antennas. The three different
antenna groupings represent different selections of antennas for
the two STC groups.
[0147] With the two stream example, the matrices B.sub.1, B.sub.2,
B.sub.3, B.sub.4, B.sub.5, B.sub.6 show how two streams consisting
of s.sub.1, s.sub.2, s.sub.3, s.sub.4 and s.sub.5, s.sub.6,
s.sub.7, s.sub.8 can be spread across four transmit antennas in six
different ways. Space time coding is again employed between sets of
two symbols within each stream.
[0148] FIG. 14 shows antenna selection options for a closed loop
STC/MIMO 4-transmit antenna arrangement in accordance with an
embodiment of the invention. Shown are 1, 2 and 3 stream antenna
selection options. With one stream antenna selection option, a
single of the four antenna as selection for transmitting a single
symbol. The transmission matrices are indicated at C.sub.1,
C.sub.2, C.sub.3, C.sub.4. A power boosting of c=1 is applied since
only a single transmit antenna is employed.
[0149] For two streams, two of the four antennas are selected, and
there are six different options as set out in the table. A power
boosting factor of 1/ {square root over (2)} is applied for this
case. For a three stream example, there are four different ways to
select three of the four antennas as set out in the table. A power
boosting factor of 1/ {square root over (3)} is applied. Also shown
in the table is an example set of bits that can used to feedback on
the channel quality indicator channel selection of an appropriate
one of the potential transmission matrices depending upon whether
there are 1, 2 or 3 streams.
Binary Beam-Forming
[0150] It is noted that while the above examples show antenna
selection feedback, binary beam-forming can be used to similar
effect. With binary beam-forming, the beam-forming weights are
binary, meaning that a given antenna is either on or off. This is
analogous to selection. Two specific examples of using beam-forming
to perform antenna selection are provided in FIGS. 15 and 16 for
three and four antenna systems respectively. For the four antenna
system, assuming the symbol stream can be defined in the form of
x=diag[s1, s2, s3, s4] and the transmitted beam-formed matrix is
C=xW where is listed in FIG. 16. This is the same result as the
antenna selection examples of FIG. 14. This allows the unification
of antenna selection with beam-forming, where W consists of binary
weights in this particular case.
MIMO Downlink Pilot Designs
[0151] Referring now to FIGS. 17 through 23, a number of different
pilot transmission schemes will now be described for MIMO
applications of OFDM. In these examples, it is assumed that a
common pilot structure is used for any receivers that might need to
rely on it. In order that multiple receivers can make use of the
pilots, no receiver specific pre-coding is performed. In other
examples described further below, pre-coding of pilots for specific
receivers is performed. In all of the Figures showing pilot
transmission schemes, the horizontal axis represents OFDM
sub-carriers separated in frequency, and the vertical axis is the
time dimension, with each row representing on OFDM symbol
transmission interval. In all of the examples, the receiver
recovers the transmitted pilots, compares them to known content of
the pilots, and determines channel estimates on the basis of this
comparison. Channel estimates can be interpolated for the non-pilot
sub-carrier locations.
[0152] Referring first to FIG. 17, shown is a pilot transmission
scheme in which pilots are allocated for four antennas as indicated
at 400,402,406,408. The pilots are allocated in blocks of 2.times.2
sub-carrier locations, with two consecutive sub-carriers over two
consecutive OFDM symbols. In this example, the location of the
pilot for each antenna is the same with each group of four
sub-carrier locations. Furthermore, note that the pilots 406,408
for antennas 1 and 3 are "punctured pilots", meaning that these
pilots are transmitted in place of data sub-carriers such that the
receiver must rely on error correction to demodulate and recover
correctly the transmitted information in the absence of the
punctured data locations. In other words, where in a 2-antenna
system, data would be transmitted on sub-carriers indicated at 409,
with the 4-antenna system no data is transmitted on the sub-carrier
locations used for the punctured pilot locations. Each pilot is
transmitted only by the respective antenna with null sub-carriers
inserted in the other antennas. In the illustrated example, pilots
are inserted in every pair of two consecutive OFDM symbols, but in
some implementations, not every OFDM symbol contains pilots. The
location of the 2.times.2 blocks of pilots is staggered between
adjacent pairs of OFDM symbols. In the illustrated example, the
staggering is not uniform, but it could be uniform in other
implementations.
[0153] Referring now to FIG. 18, another example of a 4-antenna
pilot transmission scheme is shown. In this example, four pilots
are inserted on a single sub-carrier for the four antennas as shown
in 410,412,414,416 over four OFDM symbol transmission intervals.
Multiple such groups of four pilots are inserted in sub-carrier
locations that are separated by eight data sub-carriers in the
example shown. Furthermore, the particular sub-carriers used are
offset in consecutive sets of four OFDM symbols. It can be seen
that half as many pilots are inserted with the example of FIG. 18
as are inserted in the example of FIG. 17. The result is that for a
given sub-carrier and a given antenna, an accurate channel estimate
can be obtained at the receiver half as often. The interpolated
channel estimates may therefore be less accurate. This has the
potential of effecting the accuracy of data recovery at the
receiver.
[0154] While a particular staggering arrangement has been shown in
FIGS. 17 and 18, it is to be understood that other arrangements are
possible.
[0155] Referring now to FIG. 19, shown is another example of a
pilot transmission scheme for a 4-antenna system. In this system,
for groups of four consecutive OFDM symbols, pilots are inserted
only for two of the antennas. The group of four OFDM symbols
indicated at 420, pilots are inserted for antennas 0 and 1, while
for the group of four OFDM symbols 422 pilots are inserted for
antennas 1 and 3. For subsequent groups, the set of antennas
alternates. Within a given set, pilot pairs are inserted in
selected OFDM sub-carrier locations over two OFDM symbols. For
example, one such pair of pilot symbols is indicated at 424,426,
this including one pilot for antenna 0 and one pilot for antenna 2.
The sub-carriers used for the pilots are offset between the first
two OFDM symbols and the second OFDM symbols of the set of four
OFDM symbols 420. A similar arrangement is shown for the pilots of
antennas 1 and 3 during the second set of four OFDM symbols 422.
While a particular offsetting arrangement has been shown, other
offsetting arrangements can be implemented.
[0156] Referring now to FIG. 20, shown is an example of a pilot
transmission scheme for an 8-antenna system. Groups of eight pilots
are inserted in a staggered fashion within the time frequency
plane. Each group is two sub-carriers by four OFDM symbols, and
includes one pilot for each of the eight antennas. One such group
of eight is indicated at 430.
[0157] Referring now to FIG. 21, shown is another example of pilot
transmission scheme for an 8-antenna system. In this example,
alternating sets of four OFDM symbols are used to transmit pilots
for respective sets of four antennas. Thus, the first set of four
OFDM symbols 440 is used for the first four antennas and the next
set of four OFDM symbols 442 is used for the next set of four
antennas. For each set of four antennas, the pilots are inserted in
blocks of two sub-carriers by two OFDM symbols each containing one
pilot for each of the four antennas, and the location of the block
is offset between consecutive pairs of OFDM symbols.
[0158] Referring now to FIG. 22, shown is a pilot transmission
scheme for a 12-antenna system. This example is basically an
extension of the example of FIG. 20 but with each group of pilots
being inserted having a dimension of two OFDM sub-carriers by six
OFDM symbols, and each block of OFDM pilots containing one pilot
for each of 12 different antennas. One example block is indicated
at 450.
[0159] Referring now to FIG. 23, shown is another example of a
pilot transmission scheme for a 12-antenna system. In this case,
the overall pilot pattern is 12 OFDM symbols in duration. The first
four OFDM symbols 452 are used for first four antennas; the next
four OFDM symbols 454 are used for the next four antennas, and the
last four OFDM symbols 456 are used for the last four antennas.
Each set of four OFDM symbols is shown arranged similarly to the
sets of four OFDM symbols described previously with reference to
FIG. 28 for the eight antenna design.
[0160] As indicated above, for the example pilot patterns of FIGS.
17 through 23, as described, the pilots are not pre-coded, and can
therefore be used by any receiver to obtain channel estimations for
the particular antenna. In another embodiment, for any one of the
patterns described thus far, the pilots for at least one antenna
are pre-coded for a particular user. As described above, pre-coding
of pilots for a particular user may enable pilots to be inserted at
a lower frequency or for the particular user and thereby save
transmission bandwidth. Further examples are given below that
involve pre-coding of pilots for closed-loop MIMO sub-channel
transmissions.
Pilot Transmission Schemes with Pilot Pre-Coding
[0161] Systems and methods employing pre-coding of MIMO pilots in
accordance with an embodiment of the invention will now be
described. In the above described embodiments, it has been assumed
that pilot channel information is transmitted without any weights
being applied. In this manner, any receiver can look at the
transmitted pilot information and recover a respective channel for
the particular receiver. In some embodiments, a pre-coding of the
pilot is performed.
[0162] For a pre-coded pilot, the equivalent channel measured at
the receiver becomes G=HW, where H is the actual channel, and W are
the weights applied to the pilots. The feedback beam-former W, or
information from which it can be derived, may be computed from the
channel H for a particular user, Many examples of how this feedback
can be performed are detailed below.
[0163] In the event pre-coding of the pilot is performed, the
channel can be recovered according to: H=GW'(WW').sup.-1 However,
the pilots need to be user-specific for such a case because
beam-forming weights are receiver-specific. This is in contrast to
the case where the pilots are not pre-coded and the channel can be
recovered by each user.
[0164] FIGS. 24 and 25 compare and contrast the use of
non-pre-coded pilots as opposed to pre-coded pilots. In FIG. 24, it
is assumed that a set of non-pre-coded pilots were transmitted. A
distribution of data sub-carrier locations 451 and pilot
sub-carrier locations 453 are shown for the purpose of example
only, In this case, the channel H can be used for:
[0165] 1. current H analysis and future feedback; and
[0166] 2. multiple users (possibly including non-MIMO) to assist in
channel interpolation.
[0167] The non-pre-coded channel H can be re-encoded to coherently
demodulate current data for each user.
[0168] In contrast, with the example of FIG. 25, some of the pilots
are dedicated to a particular user, and as such can be pre-coded
while other pilots are for multiple use, and as such are not
pre-coded. The use of pre-coding for pilots can be constant, or can
vary over different time. A distribution of data sub-carrier
locations 451 and pilot sub-carrier locations 453 are shown for the
purpose of example only. During time t.sub.1, the pilot
sub-carriers 461 are pre-coded pilots for a first receiver 465, and
during a second time t.sub.2, pre-coded pilots are inserted for a
different receiver 467. There may also be pilots that are
transmitted for use by both users (not shown).
[0169] Advantageously, by using pre-coded pilots, a smaller number
of pilots can be used. The number of pilots can be reduced in
either the time or frequency dimension. Preferably, for cases in
which the receiver is mobile, the density in time is kept the same
and the density is reduced in the frequency domain. Furthermore,
for band adaptive modulation and coding, it is preferable to select
the most flat part of the band or use larger FFT size, and/or to
use an interlaced antenna pilot mapping.
[0170] For the case where a receiver is nomadic, i.e., it is not
highly mobile, but does move periodically, it is advantageous to
keep the frequency density the same, and to reduce the density in
the time domain. In this case, preferably a block antenna pilot
mapping is employed.
[0171] The location of the receiver-specific pilots is an
implementation specific decision. For the purpose of completeness,
several examples are presented below.
[0172] FIG. 26 shows an example of pre-coded pilot design for a
2-antenna basestation (BS) in accordance with an embodiment of the
invention. In this examples sub-channels are defined that are nine
sub-carriers wide for a given receiver, each sub-channel being
assigned to a respective receiver. Thus, two sub-channels are
indicated at 500,502. These sub-channels can be continuously
allocated for a given mobile station, or they can be allocated in a
time division manner. Pilot carriers are inserted as indicated at
504,506 for antenna 0 and antenna 1 respectively and for the first
mobile station. These are inserted in the sub-channel 500 for the
first mobile station. Similarly, pilot carriers 508,510 are
inserted in the sub-channel for the second receiver for antenna 0
and antenna 1 respectively. Because the pilot sub-carriers are
dedicated to a particular receiver, preferably they are pre-coded
as discussed above. Pre-coded pilot sub-carriers axe coded for the
particular receiver using feedback information received from each
receiver.
[0173] The example of FIG. 26 shows sub-channels that consist of a
set of sub-carriers being allocated to respective mobile stations.
This approach can be extended to multiple sub-channels, more than
simply the two shown. Furthermore, the particular location of the
sub-carriers used for pilots can vary on an implementation specific
basis. Preferably the pilots are somewhat scattered within each
sub-channel for a given receiver so that better channel information
can be interpolated for the remaining data sub-carrier
locations.
[0174] Referring now to FIG. 27, shown is another example in which
sub-channels are divided up in a time-wise manner. The sub-channel
for a first mobile station is transmitted on the first two OFDM
symbols as indicated at 520, and the sub-channel for a second
mobile station is transmitted on the second set of two OFDM symbols
as indicated at 522, and then this pattern repeats. The sub-channel
for a given mobile station may contain fewer or greater number of
OFDM symbols than the two shown in FIG. 27. Furthermore, there can
be any appropriate number of sub-channels defined. In this example,
receiver specific pilot carriers are inserted during each of the
two sub-channels 520,522. The pilots for the first sub-channel are
indicated at 524,526 and the pilot carriers for the second receiver
are indicated at 528,530. In this example, the pilots for antenna 0
are inserted in one OFDM symbol, and the pilots for antenna 1 are
inserted in the next OFDM symbol within each sub-channel. A
particular spacing for the pilot carriers is shown, but of course
different spacings and frequencies can alternatively be employed.
Once again, since the pilots are dedicated to a particular users
they can be pre-coded for each user.
[0175] The example of FIG. 26 can be extended to differing numbers
of antennas. An example is shown in FIG. 28 where there are three
antennas and as such each sub-channel has pilot carriers for each
of the three antennas. With the example of FIG. 28, the pilots for
one of the antennas, in the illustrated example antenna 1, are
punctured pilot carriers. This means that data locations would
normally have been transmitted there, and as such the receiver must
rely on error correction to correctly recover an entire transmitted
data stream. Alternatively, pilot locations could be allocated for
all three antennas that do no rely on puncturing. Furthermore, it
may be that only some of the pilots for a given antenna are
puncture locations. A very similar example is shown in FIG. 29 for
a four transmit antenna example in which the pilot carriers for two
of the antennas, namely antennas 1 and 3 are punctured pilot
carriers. In this example and the other examples, zero, one, or
multiple pilots may rely on puncturing.
[0176] In accordance with another embodiment of the invention a
pre-coded pilot design for the PUSC (partial use sub-carrier
applications) is provided. For PUSC the symbol is first divided
into basic clusters of consecutive sub-carriers. Pilots and data
carriers are allocated within each cluster of sub-carriers.
According to IEEE 802.16-2004, a permutation is used when
allocating the data carriers to a sub-channel, a permutation
consisting of a remapping of the a cluster sub-carriers into other
allocations (that can be almost random). The data carriers in each
cluster may be assigned to different MSSs, and therefore, pilots
may be used by multiple MSSs. To support the dedicated pilots to
closed-loop MSSs operating in PUSC mode, the permutation procedure
to partition the sub-carriers into sub-channels is disabled or is
allowed to apply to the PUSC sub-channel within a single user. Each
sub-channel includes 48 data carriers from two clusters.
[0177] FIG. 30 shows a pre-coded pilot design for a 2-antenna BS
for a PUSC zone in accordance with an embodiment of the invention,
where the pilot design shows the form of the cluster prior to
remapping/permutation. In other words, the pattern of FIG. 30 would
actually be scrambled across an OFDM bandwidth. In this example,
the first four OFDM symbols are used for a sub-channel for a first
mobile station, and the next four OFDM symbols 542 form a
sub-channel for another mobile station. In the frequency domain,
there may be additional sub-channels for other mobile stations, not
shown, or the entire bandwidth may be applied for a single wireless
station at a time. In this case, pilots are inserted for two
antennas, and these can be inserted in a receiver specific manner.
Pilot carriers for the first receiver on antenna 0 are indicated at
544, and pilot carriers for the first receiver on the next antenna
are indicated at 546. These are inserted in pairs on consecutive
sub-carrier locations and in every other OFDM symbol at
illustrated. The pattern continues with pilots carriers 545,547
inserted during the sub-channel 542 for the second receiver, and
the pilots are pre-coded for the respective receivers.
[0178] Referring now to FIG. 31, shown is another example similar
to that of FIG. 30, but in which pilot carriers are inserted for
four antennas. In this case, the pilot carrier insertions for
antenna 0 and 1 are identical to those of FIG. 30. In addition to
this, pilots are inserted in the first OFDM symbol of each
sub-channel for antennas 2 and 3 as indicated at 550,552. In this
case, it is assumed that the pilots for antennas 2 and 3 are
inserted in punctured pilot locations. Furthermore, the pilots are
inserted half as frequently for the antennas 2 and 3, namely for
only one OFDM symbol of each set of four.
Closed Loop Feedback Options for MIMO Applications
[0179] The transmission formats discussed above rely on antenna
selection/grouping feed back and/or beam-forming weights. FIG. 32
shows a summary of available closed loop STC/MIMO arrangements with
beam-former structures in accordance with embodiments of the
invention.
[0180] The first option indicated at 600 is to perform MIMO/STC set
up based on closed loop feedback. This may involve selection
between SM and STTD, and/or antenna selection and/or antenna
grouping, as defined for a given application. Given a set of
options that are to be made available to a given receiver, the
feedback information allows the receiver to select between the
options and inform the transmitter of the selection.
[0181] The next option indicated at 602 is to perform beams forming
based on feedback information. The feedback information may come in
a variety of forms described in detail below. The information that
is fed back allows a transmitter to generate weights for the
particular receiver. The feedback information can be the weights
per se, or other information.
[0182] The next option indicated at 604 involves performing
MIMO/STC setup based on feedback and feeding the output of this
into a beam-former that might be static in nature.
[0183] In another embodiment, indicated at 606, the output of a
MIMO/STC matrix is input into a beam-former that undergoes
beam-forming as a function of feedback.
[0184] Finally, the next option indicated at 608 involves
performing MIMO/STC set up based on feedback, and feeding the
output of this into a beam-former that also performing beam-forming
a function of feedback.
[0185] In summary, the options are to apply feedback for MIMO/STC
set up and/or applying feedback to the beam-former The beam-former
feedback information may be for a unitary beam-former structure, or
some other beam-former. Detailed examples have been given
previously as to how MIMO/STC set up information can be fed back,
Many options for beam-former feedback are given in the details
below.
[0186] In accordance with various embodiments of the invention
there is provided methods of facilitating closed loop MIMO
pre-coding and feedback in a communications network that might, for
example be operating in accordance with the IEEE 802.16(e) and IEEE
802.11(n) standards. As will be apparent to one of skill in the art
the various embodiments can be implemented in software, firmware or
an ASIC (application specific integrated circuit) or any suitable
combination of these or other components.
Feedback for Antenna Selection/Grouping and Eigen-mode
Selection
SVD Approach
[0187] SVD (singular value decomposition) is theoretically optimal
in terms of the best channel matching transmission and achieving
the link level Shannon capacity. However, SVD typically requires a
large amount of computing and a large amount of CSI (channel state
information) feedback in the FDD (frequency division duplexing)
case. With SVD, the Channel matrix H is decomposed according to
H=U.LAMBDA.V where U and V are unitary, and .LAMBDA. is a diagonal
matrix containing the eigenvalues. Typically, information defining
V is fed back (although H could be fed back and the decomposition
re-executed at the transmitter). Then the transmitter sends
V.sup.TS, where S is the information vector. The receiver receives
this, as modified by the channel, i.e.,
S=HV.sup.TS=U.LAMBDA.VV.sup.TS(+noise). The receiver multiplies
this by U.sup.1 (conjugate transpose of U, known at the receiver)
to yield U.sup.1.LAMBDA.VV.sub.S.sup.T'=.LAMBDA.S due to the
properties of unitary matrices. The diagonal matrix .LAMBDA. is
also known at the receiver and S can then be recovered. Various
methods of efficiently feeding back V or information allowing the
transmitter to reconstruct V are given below.
[0188] H can be further expressed as follows:
H=.lamda..sub.1u.sub.1v.sub.1+.lamda..sub.2u.sub.2v.sub.2+.lamda..sub.3u.-
sub.3v.sub.3+.lamda..sub.4u.sub.4v.sub.4 the U matrix consists of
columns equalling the vectors u1,u2,u3,u4 and the V matrix consists
of rows equalling the vectors v1,v2,v3,v4 all the vectors being
unitary.
[0189] Thus, the matrix V.sup.T used in the transmitter consists of
a set of unitary vectors as columns, each associated with a
respective eigenvalue. For example, a 4.times.4 matrix is
[v1,v2,v3,v4] where v1,v2,v3,v4 are unitary vectors in columns of
the matrix. Eigen-mode selection involves setting one or more of
the vectors in V to zero, typically the vectors associated with the
weak eigenvalue(s). In some embodiments, eigen-mode selection
information is fed back to the receiver.
Antenna Selection/Grouping
[0190] Antenna selection/grouping is based on the selection of
sub-set of antennae from a larger set of available antennas by the
terminal based on a simple criterion. The terminal generally needs
to feedback very little information to the base station, Mechanisms
for transmitting this feedback information from the receiver to the
transmitter have been presented above.
[0191] Antenna grouping criterion can be based on determinant as
described previously, Antenna grouping can also be based on an
eigenvalue approach. This approach may avoid selection being
dominated by weak layers. Typically degradation due to antenna
selection is smaller when the group size is larger.
[0192] For a MIMO system, the eigenvalues of the channel matrix are
often unevenly distributed. Removing the layers associated with
small eigenvalues will have little effects on channel capacity.
With a reduced number of layers, more sophisticated MEMO decoders
can be used to improve link performance. The power saved from
layers that are not used for this a particular receiver can be used
by other users on other sub-bands. C = 10 .times. i = 1 M .times.
lg ( 1 + P M .times. .times. .beta. l ) .beta. l = .sigma. 2
.times. i = 1 M .times. ( v ij 2 .lamda. j ) C = 10 .times. i = 1 M
- 1 .times. lg .function. ( 1 + P M .times. .lamda. i .sigma. 2 )
##EQU46## This expression is for the channel capacity. It can be
seen that larger lambdas make a larger contribution to the channel
capacity. While a default would be to assign equal power to all
eigenvalues, in another embodiment, water filling according to
eigenvalue is performed, and/or some eigenvalues are not used at
all.
[0193] The V matrix of an SVD decomposed channel matrix consists of
a set of horizontal vectors each of which are multiplied by a
respective eigenvalue in the product UDV. In the transmitter,
through the use of a beamforming matrix V.sup.T a respective stream
can be transmitted in association with each eigenvalue/column of
V.sup.T=row of V. Thus, if V.sup.T=[v1,v2,v3,v4], then what is
transmitted can be expressed as
v1.times.s1+v2.times.s2+v3.times.s3.times.v4.times.s4 for a
4.times.4 system. Transmission using such a vector will be referred
to as an "eigen-mode". Different AMC can be applied in the
transmitter for each such eigen-mode. For example, in a 4.times.4
system, there will be 4 eigenvalues, and four beamforming vectors
each for use with a respective stream. In the event water filling
is employed, the power allocated to the streams is not equal, and
in some cases one or more of the layers is simply not used at all,
and the power that would have been used for that stream is
available for the other layers. Feedback in such systems can
consist of an index of the eigen-modes to employ, with the
remaining eigen-modes shutdown, and/or with channel quality
readings for some or all of the selected eigen-modes. In some
embodiments, the channel quality defines an associated AMC for use
on the eigen-mode at the transmitter.
[0194] These two approaches are summarized in FIG. 33 for the SVD
approach showing the discarding of weak eigen-modes, and FIG. 34
showing de-selected antennas turned off/not used for the particular
user. The simulation results of FIGS. 35 and 36 show that antenna
grouping can achieve very close to SVD performance with
significantly lower complexity.
[0195] SVD generally provides optimal performance when adaptive
modulation and coding (AMC) can be performed on each individual
layer. When AMC is not layer based, the performance difference is
generally smaller due to imbalanced layer SNR distribution.
Preferably, therefore, AMC is implemented on per layer basis.
[0196] In some embodiments, when SVD is used the 4-th layer can
generally be eliminated meaning only 3 columns of the V-matrix
generally need to be fed back to the transmitter. In systems with
larger numbers of antennas, it may be possible to eliminate more
layers.
Antenna Grouping Algorithms
[0197] Various antenna grouping algorithms will now be described.
Any of these can be employed on a per sub-carrier, per block of
sub-carriers, per group of blocks of sub-carriers or for an entire
OFDM band. They can be applied for a single or multiple user
system.
Selection Between STTD and SM
[0198] An antenna grouping algorithm in accordance with an
embodiment of the invention will now be described. With channel
estimates from the pilots: the receiver computes criterion for use
in selecting between STTD and SM transmission modes as follows,
where if the STTD criteria is greater than the SM criteria, then
STTD transmission is employed and alternatively, SM is employed.
Other criteria can be used. STTD SM ##EQU47## i = 1 N .times. j = 1
M .times. h ij 2 > .gamma. 0 M - 1 2 i = 1 M .times. g i 2
.times. .times. where .times. .times. .gamma. 0 = P M .times.
.times. .sigma. 2 ##EQU47.2## and g.sub.i is the row of matrix G,
where G=(H'H).sup.-1 H', and H is the channel matrix. For a MISO
arrangement the weights may be provided as follows: Beam .times. -
.times. forming .gamma. = 1 .sigma. 2 .times. i = 1 N .times. h l 2
Antenna .times. .times. selection .gamma. = 1 .sigma. 2 .times. max
.times. { h 1 2 , h 2 2 , .times. , h N 2 } STTD .gamma. = 1 N
.times. .times. .sigma. 2 .times. i = 1 N .times. h l 2
##EQU48##
[0199] According to another embodiment one may exploit the benefits
of both. For example, when a channel changes slowly, beam-forming
provides a SNR gain, while STTD provides diversity gain (on the
symbol level) which improves FEC performance (especially for higher
code rates). Then, when a channel changes quickly one can
degenerate to an STTD system which will preserve protection against
fading.
Antenna Selection for STTD
[0200] An antenna grouping algorithm in accordance with an
embodiment of the invention will now be described for an STTD mode.
The antenna group selection can be based on criterion including the
following: SS STTD = arg .times. max SS .times. l = 1 N .times. j =
1 M SS .times. h ij 2 ##EQU49## Antenna Selection for SM
[0201] For the SM mode, the antenna group selection can be based on
criterion including the following: SS V .times. - .times. BLAST =
arg .times. min SS .times. l = 1 M SS .times. g i 2 ##EQU50##
Antenna Selection for STTD, SM, and Switching Criterion
[0202] An antenna grouping algorithm in accordance with an
embodiment of the invention will now be described. For the STTD
mode, in accordance with an embodiment of the invention, define
.xi. j = l = 1 N .times. h ij 2 ##EQU51## and select a set of
antennas according to: L K = arg .times. min j .di-elect cons. { M
} .times. ( .xi. j ) ##EQU52## And wherein transmission on an
antenna is stopped when: L.sub.l<.alpha.L.sub.K In accordance
with embodiment of the invention, for the SM mode, if Q = .lamda.
max .lamda. min .gtoreq. .beta. ##EQU53## then a new subset of
antennas then can be selected according to criterion including
maximum determinant criterion: S.sub.a=arg
max(det(H.sub.a'H.sub.a)) where H.sub.a is the channel matrix of
the new sub-MIMO size, and .alpha. and .beta. are predefined
thresholds, and where .lamda..sub.max and .lamda..sub.min are the
largest and smallest eigenvalues respectively.
[0203] The following is a particular example of an antenna grouping
procedure:
Step 1: Decide antenna group size for STTD and SM mode;
Step 2: Select-transmit antenna sub-sets based on selected
size;
Step 3: Decide transmit rode: STTD vs. SM;
Step 4: If SM is selected and the antenna group size is larger than
SM layers, then use Matrix B, otherwise, use Matrix C.
Multi-User Pre-Coding
[0204] A multi-user pre-coding algorithm in accordance with an
embodiment of the invention will now be described. Consider a MIMO
broadcast channel transmit antennas to transmit to K users each
with N receive antennas. The base band model of this system is
defined by: y.sub.k=H.sub.ks+w.sub.h 1.ltoreq.k.ltoreq.K where
H.sub.k.epsilon.C.sup.N.times.M denotes the channel matrix from the
base station to the k.sup.th user, s.epsilon.C.sup.M.times.1
represents the transmitted vector, and
y.sub.k.epsilon.C.sup.N.times.1 signifies received vector by the
k.sup.th user. The vector w.sub.k.epsilon.C.sup.N.times.1 is white
Gaussian noise with zero-mean and unit-variance.
[0205] In the proposed methods the transmitted vector a carries
information for M users, defined as follows. s = j = 1 M .times.
.sigma. .pi. .function. ( j ) .times. w .pi. .function. ( j )
##EQU54## where .pi.(j), .pi.(j), j=1 . . . M are the indexes of a
subset of users so-called active user
v.sub..pi.(j).epsilon.C.sup.M.times.1, j=1 . . . M are a set of
orthogonal vectors, and d.sub..pi.(j) includes information for the
user .pi.(j). In addition, dirty-paper pre-coding is used on top of
the system such that if i>j, the interference of the user
.pi.(i) over user .pi.(j) is zero.
[0206] For demodulation, the user .pi.(j) multiplies the received
vector to a normal vector u'.sub..pi.(j) where (.)' denotes
transpose conjugate operation. In the next section, we specify a
method to select the set of active users, modulation vectors
v.sub..pi.(j), and demodulation vectors u.sub..pi.(j) for j=1 . . .
M.
Determination of the Active Users, Modulation, and Demodulation
Vectors
[0207] In this part, it is assumed that the channel state
information is available at the transmitter. Later, this algorithm
is modified such that only partial channel state information is
required.
[0208] Each stage of the algorithm includes two optimizing
operations. First, for each user, finding a direction in which that
user has maximum gain. Second, selecting the best user in terms of
the larger gain. This optimization is performed in the null space
of the former selected coordinates. In the following, the proposed
algorithm is presented.
1) Set j=1 and the condition matrix G.sub.eq=O.sub.M.times.M.
2) Find .sigma..sup.2.sub..pi.(j) where
.sigma..sup.2.sub..pi.(j)=max.sub.rmax.sub.xx'H.sub.rH'.sub.rx
x'x=1 G.sub.eqx=0 set .pi.(j) and v.sub..pi.(j) to be equal to the
optimizing parameter r and x, respectively. 3) Set u .pi.
.function. ( j ) = 1 .sigma. .pi. .function. ( j ) .times. H .pi.
.function. ( j ) .times. v .pi. .function. ( j ) ##EQU55## 4) Set
g.sub.j=v.sub..pi.(j), where g.sub.j is the j.sup.th column of the
matrix G.sub.eq. 5) j.rarw.j+1. If j.ltoreq.M go to step two,
otherwise stop. Modified Algorithm
[0209] As it has been mentioned before, in this algorithm, a major
part of the processing can be accomplished at the receivers.
Therefore the perfect channel state information is not required at
the transmitter which results a significant decreasing in the rate
of feedback.
1) Set j=1 and G.sub.eq=Q.sub.M.times.M.
2) each user calculates .sigma..sup.2.sub.r(j), defined as follows:
.sigma..sup.2.sub.r(j)max.sub.xx'H.sub.rH'.sub.rx x'x=1 G.sub.eqx=0
v.sub.r(j) represents the optimizing parameter x. 3) Each user
calculates u r .function. ( j ) = 1 .sigma. r .function. ( j )
.times. H r .times. v r .function. ( j ) ##EQU56## 4) Each user
sends .sigma..sup.2.sub.r(j) and V.sub.r(j) to the base station, if
.sigma..sub.r(j).sup.2.gtoreq.th(j). th(j) is a threshold which is
predetermined by the base station. 5) Base station selects the user
with the largest .sigma..sup.2.sub.r(j). Let .pi.(j) be the index
of the selected user. The corresponding gain and coordinate of that
user. The corresponding gain and coordinate of that user are
.sigma..sup.2.sub..pi.(j) and v.sub..pi.(j), respectively. 6) The
.pi.(j).sup.th user sends U.sub..pi.(j)H.sub..pi.(j)w.sub..pi.(j)
i=1 . . . i-1 to the base station. 7) Base station send
v.sub..pi.(j) to all users. All include v.sub..pi.(j) in G.sub.eq
as the j.sup.th column. 8) j.rarw.j+1. If j.ltoreq.M go to step
two, otherwise stop. Feedback MIMO Channel and CQI Separately
[0210] Referring now to FIG. 37, shown is a block diagram of a
system in which MIMO channel information for use in performing
beam-forming, and MIMO/STC setup information are separately fed
back in accordance with an embodiment of the invention. In the
illustrated example, the MIMO/STC setup information is fed back on
a CQICH (channel quality indication) feedback channel, various
permutations of which are available in existing standards
definitions, but it is to be understood that in this and the other
examples that follow, any appropriate physical/logical channel can
be employed to feed back the MIMO/STC setup information as
required.
[0211] At the transmitter, there is a transmit beam-former 202 that
uses matrix V.sup.T this being the transpose of the matrix v in the
singular value decomposition of the channel H. The output of the
beam-former is transmitted on M transmit antennas 204. This is
received at the receiver by N receive antennas 206, and the MIMO
channel H is measured as indicated at 208. There are many existing
methods of measuring the channel. This may for example be done
using pilot symbols transmitted at known locations in time and/or
frequency, possibly using one of the formats introduced previously
and performing interpolation if necessary. This may include
receiver specific pilots and or generic pilots. Next, block 212
performs a singular value decomposition of the channel matrix H
into the matrix product UDV where U and V are unitary matrices and
D is a diagonal matrix containing the eigenvalues.
[0212] At 220, AMC/eigen assignment according to according to
eigenvalue is performed for any appropriate frequency resolution,
for example for a given sub-channel. A sub-channel may for example
be a contiguous block of sub-carriers that have substantially the
same eigenvectors/eigenvalues, As discussed previously, weak
eigen-modes can be discarded. In some embodiments, all that needs
to be fed back is an index of the eigen-modes that are to be used.
Then, the receiver, after it reconstructs the H or V matrix from
the fed back channel information, can re-construct the eigen-modes,
and use each eigen-mode for a respective stream. This information
is then sent back to the transmitter on the CQI feedback channel
222 or using some other mechanism.
[0213] For the channel information, preferably the difference
between the new channel matrix H and the previous channel matrix E
is measured and a differential is determined between the two most
recent channel matrices. This amount is encoded at 210 and
transmitted back to the receiver on the MIMO feedback channel 214.
Advantageously, the differential of two consecutive channel
matrices will have a significantly smaller dynamic range than would
the actual channel matrix H or V matrix. This enables the MIMO
feedback channel 214 to use fewer bits to transmit back effectively
the same amount of channel information.
[0214] At the transmitter, the channel matrix H is reconstructed at
216 using the fed back channel differential and the previous
channel matrix H. Singular value decomposition is performed at 218
to recover the V matrix component of the channel matrix H, and the
transpose of this matrix is then used in the beam-former 202.
Furthermore, the CQI feedback channel 222 feeds into the per stream
AMC function 200 that performs adaptive modulation of coding on
each stream using the channel quality information as sent back from
the receiver. In the above-described embodiment, a differential of
the channel matrix H is fed back. Equivalently, the unitary matrix
V or a differential of the unitary matrix V could be fed back.
Various detailed methods of feeding back the channel matrix or the
unitary matrix V are provided below.
[0215] The embodiment of FIG. 37 and the other embodiments
described generalize to an arbitrary number of transmit antennas
and receive antennas. While a particular arrangement of components
are shown for each of the various embodiments, it is to be
understood that the functionality of these components can be
combined or separated differently than shown on an implementation
specific basis. Furthermore, FIG. 37 can be thought of representing
either a system or a method.
Feedback MIMO Channel and CQI Jointly
[0216] Referring now to FIG. 3B, shown is a system example in which
the MIMO channel and the CQI are jointly fed back in accordance
with an embodiment of the invention. Shown in the transmit side is
a per stream AMC and beam-former V.sup.T 240 that has outputs
connected to M transmit antennas 242. On the receive side, there
are N receive antennas 244 and there is a function that measures
the MIMO channel H at 246. The MIMO channel H is differentially
encoded at 248. The receiver has knowledge of a set of possible
differentially encoded H (or v in which case SVD decomposition is
performed at 250) matrices, for example 64 different differential
matrices. The receiver determines the candidate differential H
matrix that is closest to the actual differential. Each possible H
matrix has an associated predetermined adaptive modulation and
coding and an associated index. Therefore, by simply feeding back
the index from the receiver to the transmitter, the transmitter can
determine the proper beam-former V.sup.T, and the appropriate per
stream adaptive modulation and coding to perform. The transmitter
then reconstructs N at 254 and performs singular value
decomposition at 256 to recover the unitary matrix V which is then
used in the beam-former step. Note that while the illustrated
embodiment shows H being differentially encoded and fed back,
equivalently the unitary matrix could be differentially encoded and
fed back.
Differential Scalar Quantizer--Element by Element Quantization
[0217] The two above-described embodiments employ the feedback of
differentials in either the channel matrix H or the unitary matrix
V. FIG. 39 shows a block diagram of an example of a differential
scalar quantizer that can be used to feedback quantized
differentials. In FIG. 39, the input 260 is the new value that
needs to be quantized. This can be element in a channel matrix or a
unitary matrix for example. Every k frames, this amount is passed
through the scalar quantizer 262 and on to the feedback channel
264. This means that every k frames, the receiver feeds back the
actual value as opposed to the differential value. This avoids the
potential problem of drift that can occur over time due to ongoing
differential quantization. For the remaining frames, quantization
is performed. In the illustrated example, this consists of +1, -1
quantization 266. The quantization is performed on the basis of a
difference between the current input 260 and the "previous input"
268 by difference function 270. The previous input is maintained by
updating the value with the actual value every k frames in
accumulator 272 and then applying any plus or minus quantizations
as determined at the output of the +1, -1 quantizer 266 and storing
the result in a delay element 274.
[0218] While the illustrated example uses +1, -1 quantization, this
requiring one bit per element to be transmitted on the feedback
channel 264, higher resolution quantization could alternatively be
employed. Furthermore, some embodiments may not necessarily feature
the scalar quantization step. Note that for a 4.times.4 channel
matrix, each channel element is complex and therefore has an I and
Q component. Therefore, to feedback quantized information for a
2.times.2 channel matrix requires eight bits, two bits per channel
element, one bit per I and Q components,
.DELTA..SIGMA. Modulation 1-bit Quantizer
[0219] In accordance with another referred embodiment of the
invention a .DELTA..SIGMA. modulator 1-bit quantizer is employed to
determine the information to be fed back. An example of a
1.sup.st-order .DELTA..SIGMA. modulation encoder is shown in FIG.
40. Referring to FIG. 40, a new channel measurement is input at
280. This might for example be a C/I measurement for a particular
channel element. An offset is applied at 282 so as to move the
dynamic range of the channel measurement within the range of the
.DELTA..SIGMA. signal modulator that follows. A limiter 286 is also
shown that limits the dynamic range of the input to prevent
.DELTA..SIGMA. overflow. A given implementation may not include
either of the offset and limiter features or may include just one
of them or both of them as illustrated. The input level is often
less than the quantizer step size .DELTA. and hence is denoted by
.alpha..DELTA. where .alpha.<1 dependent on .DELTA..SIGMA. type.
A dither signal may also be applied 289 to the 1.sup.st-order
.DELTA..SIGMA. modulator input to eliminate the limit cycle
inherent in the 1.sup.st-order modulators and in this instance is a
sequence (.DELTA./8, -.DELTA./8). The remainder of the encoder is a
classical .DELTA..SIGMA. modulators with the exception there are no
delays in the signal paths, Once again, the output per I or Q
component of a channel matrix is a plus or minus one. Again,
different resolutions can be implemented if a given application
needs the higher resolution. Furthermore, .DELTA..SIGMA. modulation
can more generally be applied to any parameter for which feedback
is to be implemented.
Multi-User Differential Feedback
[0220] FIG. 41 shows an example of a system that employs
differential encoding of the channel matrix H from multiple
receivers. This is simply an example to show that for the most part
any of the examples given herein can be adapted for use with
multiple users. With this example, there are two users each having
a respective pair of antennas 300,302. Each user performs a channel
measurement 304,306 respectively and differential encoding of the
channel matrix 308,310. The channel information is then fed back to
the transmitter where a per stream AMC and beam-forming is
performed as indicated at 314. It is noted that the feedback
channel from the users typically has limited capacity. This may be
a dedicated channel for each user, or may be a channel that is
shared across users in a scheduled fashion.
[0221] The differential encoder output may be grouped together to
map to the existing CQI feedback channel or MIMO feedback
channel.
Givens Feedback
[0222] In another embodiment of the invention, the V matrix
determined through singular value decomposition is fed back from
the receiver to the transmitter using Givens feedback. Unitary
matrices have the property that they can be decomposed into a
product of Givens matrices. In particular, an n.times.n V matrix
can be decomposed into Givens matrices containing n.sup.2-n complex
parameters. The parameters can be fed back directly, or
differentially and/or quantized to potentially reduce the amount of
feedback required. Then, at the transmitter, the Givens matrices
can be reconstructed, and then the V matrix reconstructed as the
product of the reconstructed Givens matrices.
[0223] An example of a system employing Givens feedback is shown
FIG. 43. Shown is a transmit beam-former 320 with transmit antennas
322. At the receiver, receive antennas 324 receive signals, and the
channel measurement is performed at 326. This produces the channel
matrix which is then SVD decomposed at 328. Next, the V matrix is
decomposed by the Givens transform 330 to produce a series of
matrices G.sub.1, G.sub.2, . . . , the number of matrices being
required being a function of the size of the channel matrix. Each
such Givens matrix can then be uniquely represented by two
parameters .theta. and C. These parameters are selected at 332 and
differentially quantized at 334.
[0224] Then, the quantized parameters .theta. and C are fed back
over the MIMO feedback channel 336, this requiring two parameters
to be sent back for each Givens matrix. At the transmitter, the fed
back parameters .theta. and C are reconstructed at 338, and then
the beam-forming matrix V is reconstructed by determining the
product of the Givens matrices generated using the reconstructed
parameters, at 340. Finally, the transpose of the matrix V is used
by the beam-former 320. The parameters may be directly fed back or
differentially fed back.
[0225] Preferably, at the same time AMC/eigen assignment is
performed on the basis of eigenvalues of the channel matrix of the
SVD composition performed at 328 as described previously. This step
is indicated at 332. The AMC/eigen assignment information is fed
back over the CQI channel at 342. This information is fed back to
the transmitter where per stream AMC is performed is indicated at
344.
[0226] Advantageously, for V matrix feedback, the base station is
able to verify the integrity of the received matrix V by exploiting
the orthogonality of the matrix.
[0227] By decomposing the SVD-based unitary V matrix into Givens
matrices, the V matrix can be represented by n.sup.2-n independent
complex parameters.
[0228] In accordance with an embodiment of the invention,
2(n.sup.2-n) bits (one bit for I and Q of each parameter) are used
for the differential quantization of the unitary V matrix, although
other methods of feeding back the parameters can be used.
[0229] In accordance with this embodiment an n stream CQI channel
may also be used, preferably 4 bits each. Combining the Given's
feedback and the CQI yields the total amount for the unitary matrix
feedback of 2n.sup.2+2n bits. However, for the direct differential
encoding approach the total feedback used is 2n.sup.2 including the
CQI information, since it is embedded in H. FIG. 42 shows an
example set of feedback requirements for N by N MIMO systems, in
which a comparison is made between the number of bits fed back for
the unitary matrix, fed back using Givens feedback and CQI channel
information, as opposed to the direct H matrix feedback. It can be
seen that the complexity becomes increasingly comparable as a
number of transmit antennas increases.
2-Antenna Givens Transform
[0230] The following is an SVD based Givens transform algorithm in
accordance with an embodiment of the invention suitable for
2-Transmit Antennas. With 2-antenna Givens transformation,
2.sup.2-2=2 complex parameters are required to feedback the V
matrix. There are multiple known formats for Given transformations
any of which can be used. The Givens rotation for 2 transmit
antennas (Format-A) includes: G = [ c ^ s ^ .times. e j .times.
.theta. ^ - s ^ .times. e - j .times. .theta. ^ c ^ ] ##EQU57## The
parameter space includes .theta.={-.pi.+.pi.} and c={0,1}
s-(1-c.sup.2).sup.1/2.
[0231] An alternative format is the following Givens rotation for 2
transmit antenna (Format-B) that takes the form: G = [ cos
.function. ( .eta. ) e j.theta. .times. sin .function. ( .eta. ) -
e - j.theta. .times. sin .function. ( .eta. ) cos .function. (
.eta. ) ] ##EQU58##
[0232] The parameter space includes .pi.={-.pi.+.pi.} and
.eta.={-.pi.+.pi.}
3-Antenna Givens Transform
[0233] An SVD based Givens transform algorithm in accordance with
an embodiment of the invention will now be described for 3-Transmit
Antennas. With 3-antenna Givens transforms, n.sup.2-n=6 complex
parameters can be used to completely represent the V matrix. These
can be represented in the following Givens rotation for 3 transmit
antenna (Format-A) includes G=G.sub.1G.sub.2G.sub.3 represented as
follows where each of the three matrices has a respective parameter
c (c.sub.1, c.sub.2, c.sub.3) and s (.theta..sub.1, .theta..sub.2,
.theta..sub.3) which differs between the three matrices although
this is not shown. G = [ c s 0 - s * c 0 0 0 1 ] .function. [ c 0 s
0 1 0 - s * 0 c ] .function. [ 1 0 0 0 c s 0 - s * c ]
##EQU59##
[0234] The parameter space includes .theta..sub.1{-.pi.,+.pi.} and
c.sub.1={0,1} for G.sub.1 .theta..sub.2{-.pi.,+.pi.} and
c.sub.2={0,1} for G.sub.2 .theta..sub.3{-.pi.,+.pi.} and
c.sub.3={0,1} for G.sub.3 Preferably, these are quantized as
follows: .theta..sub.1.fwdarw.2 bit and c.sub.1.fwdarw.1 bit
.theta..sub.2.fwdarw.2 bit and c.sub.2.fwdarw.1 bit
.theta..sub.3.fwdarw.2 bit and c.sub.3.fwdarw.1 bit
[0235] It can be seen in the above example, the quantization
performed for the parameter .theta. is more precise than that
performed for the parameter C. It is generally the case that
different quantization accuracies may be performed for the
different parameters being fed back using given speed back. Through
experimentation, it has been found that acceptable performance can
be achieved with a less accurate C parameter. In some embodiments
described below, certain C parameters can be ignored.
4-Antenna Givens Transform
[0236] An SVD based Givens transform SQ algorithm in accordance
with an embodiment of the invention will now be described for
4-Transmit Antennas. 4-antenna Givens transformation requires
4.sup.2-4-12 complex parameters. These can be represented in the
Givens rotation for 4 transmit antenna (Format-A) as
G=G.sub.1G.sub.2G.sub.3G.sub.4G.sub.5G.sub.6 represented as follows
where each of the six matrices has a respective parameter c
(c.sub.1, c.sub.2, c.sub.3, c.sub.4, c.sub.5, c.sub.6) and s
(.theta..sub.1, .theta..sub.2, .theta..sub.3, .theta..sub.4,
.theta..sub.5, .theta..sub.6): G = [ c s 0 0 - s * c 0 0 0 0 1 0 0
0 0 1 ] .function. [ c 0 s 0 0 1 0 0 - s * 0 c 0 0 0 0 1 ]
.function. [ c 0 0 s 0 1 0 0 0 0 1 0 - s * 0 0 c ] .function. [ 1 0
0 0 0 c s 0 0 - s * c 0 0 0 0 1 ] .times. [ 1 0 0 0 0 c 0 s 0 0 1 0
0 - s * 0 c ] .function. [ 1 0 0 0 0 1 0 0 0 0 c s 0 0 - s * c ]
##EQU60## The parameter space includes .theta..sub.1={-.pi.+.pi.}
and c.sub.1={0,1} for G.sub.1 .theta..sub.2={-.pi.+.pi.} and
c.sub.2={0,1} for G.sub.2 .theta..sub.3={-.pi.+.pi.} and
C.sub.3={0,1} for G.sub.3 .theta..sub.4={-.pi.+.pi.} and
C.sub.4={0,1} for G.sub.1 .theta..sub.5={-.pi.+.pi.} and
C.sub.5={0,1} for G.sub.2 .theta..sub.6={-.pi.+.pi.} and
C.sub.6={0,1} for G.sub.3 The scalar quantizers can be implemented
as follows; .theta..sub.1.fwdarw.2 bit and c.sub.1.fwdarw.1 bit
.theta..sub.2.fwdarw.2 bit and c.sub.2.fwdarw.1 bit
.theta..sub.3.fwdarw.2 bit and c.sub.3.fwdarw.1 bit
.theta..sub.4.fwdarw.2 bit and c.sub.4.fwdarw.1 bit
.theta..sub.5.fwdarw.2 bit and c.sub.5.fwdarw.1 bit
.theta..sub.6.fwdarw.2 bit and c.sub.6.fwdarw.1 bit Givens
Transform--General Case
[0237] An SVD based Givens transform algorithm in accordance with
an embodiment of the invention will now be described for n-Transmit
Antennas. The Givens rotation for n transmit antennas (Format-A)
includes: V = k = 1 n - 1 .times. i = 1 n - k .times. G .function.
( k , i ) ##EQU61## G .function. ( k , i ) = [ 1 0 0 0 0 c s 0 0 -
s * c 0 0 0 0 1 ] ##EQU62## Preferably, scalar quantization is
performed for each of the parameters as discussed previously.
Truncation of Givens Expansion
[0238] An SVD based Givens transform algorithm in accordance with
an embodiment of the invention will now be described that employs
truncation of the Givens expansion. Preferably, weak eigen-modes
may be discarded with application of water filling in the eigen
domain. Each Given matrix is associated with one or more
eigen-modes and respective eigenvalue. As such, if a given weak
eigenvalue is to be discarded, then the associated Givens matrices
can be discarded. In the receiver, the V matrix is re-generated,
and then only the selected eigen-modes used to transmit streams as
in the above example for eigen-modes selection.
[0239] For example, where the full Givens expansion for the
3.times.3 case is as follows: G = [ c s 0 - s * c 0 0 0 1 ]
.function. [ c 0 s 0 1 0 - s * 0 c ] .function. [ 1 0 0 0 c s 0 - s
* c ] ##EQU63## this can be truncated by dropping the "weakest"
eigen-mode. For example if the third matrix corresponds to the
weakest eigen-mode, the truncated expansion is: G = [ c s 0 - s * c
0 0 0 1 ] .function. [ c 0 s 0 1 0 - s * 0 c ] ##EQU64## The full
expansion above corresponds to all three eigenvectors of the
channel. In contrast, the truncated expansion corresponds to only a
single eigenvector. The correspondence of eigenvector to Givens
matrices is a known math property of Givens expansions.
[0240] The following is another example of truncation of Givens
expansion for a 4-antenna case. The Givens expansion for
eigenvector #1 of a 4.times.4 channel matrix is as follows: G = [ c
s 0 0 - s * c 0 0 0 0 1 0 0 0 0 1 ] .function. [ c 0 s 0 0 1 0 0 -
s * 0 c 0 0 0 0 1 ] .function. [ c 0 0 s 0 1 0 0 0 0 1 0 - s * 0 0
c ] ##EQU65## The Givens expansion for eigenvectors 1 and 2 for a
4.times.4 system is as follows: G = [ c s 0 0 - s * c 0 0 0 0 1 0 0
0 0 1 ] .function. [ c 0 s 0 0 1 0 0 - s * 0 c 0 0 0 0 1 ]
.function. [ c 0 0 s 0 1 0 0 0 0 1 0 - s * 0 0 c ] .function. [ 1 0
0 0 0 c s 0 0 - s * c 0 0 0 0 1 ] .times. [ 1 0 0 0 0 c 0 s 0 0 1 0
0 - s * 0 c ] ##EQU66## The Givens expansion for all four
eigenvectors of the 4.times.4 system is as follows: G = [ c s 0 0 -
s * c 0 0 0 0 1 0 0 0 0 1 ] .function. [ c 0 s 0 0 1 0 0 - s * 0 c
0 0 0 0 1 ] .function. [ c 0 0 s 0 1 0 0 0 0 1 0 - s * 0 0 c ]
.function. [ 1 0 0 0 0 c s 0 0 - s * c 0 0 0 0 1 ] .times. [ 1 0 0
0 0 c 0 s 0 0 1 0 0 - s * 0 c ] .function. [ 1 0 0 0 0 1 0 0 0 0 c
s 0 0 - s * c ] ##EQU67##
[0241] It can be seen that discarding two eigen-modes reduces the
Givens expansion by one of the six matrices while discarding three
of the four eigen-modes will reduce the Givens expansion by three
of the six matrices. Given that each matrix requires two parameters
to be fed back, a corresponding reduction in the quantity of
feedback information that needs to be transmitted is realized.
Generally, it can be seen that in an arbitrary n.times.n case, a
Givens expansion can be employed, and the amount of feedback can be
optionally reduced by discarding one or more eigen-modes.
Bit Allocation for Givens Feedback--Full Scalar Quantization
[0242] An SVD based Givens transform algorithm in accordance with
an embodiment of the invention will now be described in which full
scalar quantization is performed. As discussed above, it can be
advantageous to use different quantizations for the various
parameters being fed back using Givens feedback. Based on the
pre-coding QoS requirement, the bit allocation may provide a
tradeoff between performance and feedback penalty.
[0243] For the Givens parameter in Format-A, .theta. is uniformly
distributed resulting in more bits being needed for accurate
feedback, whereas C is non-uniformly distributed resulting in less
bits being needed. In some implementations, some C have only
minimum impact on performance and can be treated as constant
without a substantial performance penalty.
[0244] Using a Lloyd-MAX quantizer the following bit allocations
can be employed to realize a full scalar quantizer. TABLE-US-00014
# of Transmit Antennas 2 3 4 Givens matrix G1 G1 G2 G3 G1 G2 G3 G4
G5 G6 Bits to represent 2 2 2 2 2 2 2 2 2 2 theta Bits to represent
C 2 1 1 1 1 1 1 1 1 1 Total bit allocation 4 3 3 3 3 3 3 3 3 3 to
Givens Bit Allocation for Givens Feedback - Partial Scalar
Quantization
[0245] An SVD based Givens transform algorithm in accordance with
an embodiment of the invention will now be described in which
partial scalar quantization is performed. With partial scalar
quantization, a differential is not fed back for each and every
parameter. In an example solution, a 1 bit or 0 bit is used for
each parameter c and a differential 1-bit is used for each
parameter theta. Another approach includes using less bits or zero
bits for the non-significant Givens parameters.
[0246] The following is an example of bit allocation with partial
scalar quantization for 2, 3 and 4 antennas. It can be seen that in
a particular example, for the 3-antenna case for example, the C
parameter is only fed back for the first of three Givens matrices,
and a three bit value is sent back for theta for the first of the
three matrices, and a one bit differential is used for theta for
the other two matrices. This has the potential of reducing the
number of feedback bits required from nine bits from the full
quantization example given above to five bits for the example in
the table. TABLE-US-00015 # of transmit 2 3 4 Givens matrix G1 G1
G2 G3 G1 G2 G3 G4 G5 G6 Theta 2 2 1 1 2 2 1 1 1 1 C 2 1 0 0 1 1 0 0
0 0 Bit allocation to 4 3 1 1 3 3 1 1 1 1 Givens
[0247] The following table shows an example set of feedback
allocations for different MIMO configurations using Givens feedback
with full scalar quantization versus partial scalar quantization
with a 1-bit differential for theta and 1-bit for c. It can be seen
that the savings in the number of bits required for feedback
increase as the complexity of the MIMO configuration increases.
Also shown in the table are the number of bits required for a
Hausholder based feedback mechanism. TABLE-US-00016 MIMO
CONFIGURATION 2x2 3x1 3x2 3x3 4x1 4x2 4x3 4x4 #of spatial 2 1 2 3 1
2 3 4 streams #of parameters 1 2 3 3 3 5 6 6 Full Scalar 4 3 9 9 9
15 18 18 1-bit 2 2 6 6 6 10 12 12 differential theta 1-bit for c
Hausholder 4 4 9 9 11 11 15 15
[0248] The table below shows examples of the feedback allocations
for various MIMO configurations comparing full scalar feedback to
partial feedback in which a 1-bit differential for theta is
employed, and 1-bit or zero bits are used for each c parameter.
Again, the requirements for Hausholder feedback are also shown for
the sake of comparison. TABLE-US-00017 MIMO Configuration 2x2 3x1
3x2 3x3 4x1 4x2 4x3 4x4 #of spatial 2 1 2 3 1 2 3 4 streams #of
parameters 1 2 3 3 3 5 6 6 Full Scalar 4 3 9 9 9 15 18 18 1-bit 2 2
4 4 5 7 9 9 differential theta 1-bit or zero bit for c Hausholder 4
4 9 9 11 11 15 15
Grassman Subspace Packing--Vector Quantization
[0249] With scalar quantization, be it full scalar or partial
scalar, each feedback element pertains to one parameter of
interest. This parameter can be a Givens parameter, or a channel
matrix parameter from H or V for example, or some other value. In
another embodiment, vector quantization is performed, in which each
element fed back represents a vector, either in absolute terms or
differential terms, and in respect of a set of parameters of
interest. Each potential state for the output of the vector
quantization operation is represented by a code word
identifier/index which is fed back, and used by the transmitter to
look up the vector. The vectors thus looked up are then used to
generate the beam former in the transmitter. The vector might be a
vector of a channel matrix, a vector of a V matrix (this would be a
unitary vector), or a set of Givens parameters for example.
[0250] In one embodiment, an SVD based Givens transform vector
quantization algorithm is provided that employs Grassmann Subspace
Packing. However, other forms of vector quantization may
alternatively be employed.
[0251] Each individual Givens matrix is a unitary matrix. The
product of Givens matrices is a unitary matrix. The partial product
of two or more Givens matrices is a unitary matrix. After
determining a truncated or full Givens expansion, the resulting of
unitary matrix may be quantized by using Grassmann sub-space
packing (see for example D. J. Love, R. W. Heath Jr., and T.
Strohmer, "Grassmannian Beamforming for Multiple-Input
Multiple-Output Wireless Systems," IEEE Transactions on Information
Theory, vol. 49, pp. 2735-2747, October 2003). In this case, the
code book for quantization of a unitary matrix may be considered as
sub-space packing in a Grassmann manifold.
[0252] Two codebooks that might be used include the uniform
distributed Grassmann space, and Block circulant DFT, but other
codebooks may alternatively be employed.
[0253] An example of encoding using shape and gain quantizer
according to an embodiment of the invention is shown in FIG. 44.
The input to the shape and gain quantizer is a vector to be
quantized. Preferably, this vector is a column of a unitary V
matrix, but it may not necessarily be so. For example, it may be a
set of Givens parameters that are not necessarily unitary, or a
vector of the channel matrix H. The vector is normalized at 352.
Please note however that this step is only required if the input is
not a unit length vector. V should consist of unitary columns and
rows and as such this step can be skipped if the input is a
column/row of the V matrix. At step 356, shape quantization is
performed. This involves determining from a codebook the closest
codeword/vector in the codebook to the input vector. In the event
the input vector is not unitary, the magnitude can be taken at 354
and this quantized with gain quantizer 358. Then, the shape and
gain can be together fed back at 360. Once again, the gain
quantization step is not needed if the input is unitary. For
example, if a four element input vector is used, then a "codebook"
of different four element vectors is developed to define the
overall feedback space. The shape quantizer then determines which
of the defined potential four element feedback vectors is closest
to the input, and then all that needs to be fed back is the
identity of this vector. This can then be looked up at the
transmitter and used to re-generate the vector. In effect each
vector is rounded off to the nearest vector in the code book. In a
system in which there are 16 different codewords, the codeword can
be uniquely identified by 4-bits. Thus, each column of the V matrix
could be represented by 4-bits with this approach.
Spherical Codebook Quantizer
[0254] An SVD based Givens transform vector quantization algorithm
in accordance with an embodiment of the invention will now be
described that employs a Spherical Code Based Quantizer. This
method begins with putting the Givens parameters into a vector and
then encoding the vector using a spherical code based quantizer.
The following steps are performed: [0255] 1. Given k element vector
form the vector X.rarw.R.sup.k [0256] 2. Compute g=|X| and S=X/g
[0257] 3. Use the gain codebook to quantize g as g [0258] 4. Find i
such that .alpha..sub.i=<sin-.sup.1x.sub.k<.alpha..sub.i+1
and compute
h.sub.i(s)=X/.parallel.X.parallel.(.parallel.X.sub.L.sup.j.parallel.-.par-
allel.X.sub.L-X.parallel.) [0259] 5. Find the nearest neighborhood
h.sub.i(S) to h.sub.i(S) [0260] 6. Compute
h.sub.i.sup.-1(h.sub.i(S)) to identify the quantized shape S [0261]
7. Compute the index gS and transmit In accordance with an
embodiment, Leech Lattice is used as a codebook. Alternatively, a
trellis-coded quantizer can also be employed. Feedback Setup Using
Receiver Criteria
[0262] A receiver based Givens transform in accordance with an
embodiment of the invention will now be described in which search
criteria axe established in the receiver. [0263] In accordance with
this embodiment of the invention channel filling is based on
criteria including the receiver criteria. Based on the QoS
requirements, the receiver determines the minimum feedback needed
adaptively. The process may include the following:
[0264] determining the Givens truncation level;
[0265] based on the Scalar quantizer structure assigning parameter
values; and/or
[0266] performing combinatorial searches of combinations. One
example search criterion may include receiver MSE. Others may
include Max SNR, Max Shannon capacity, or True receiver operational
process defining as follows: l opt = arg .times. min l .di-elect
cons. { 1 , 2 , .times. .times. , L } .times. MSE .times. { E s N o
.times. tr .function. ( I M + E s N o .times. N r .times. P l H
.times. H H .times. HP l ) - 1 } ##EQU68##
[0267] Based on the receiver criterion, exhaustive search the code
book and maximize the given received based criterion to determine
the best pre-code matrix. These equations represent exhaustively
computing beam-forming weights to see which one is the best for a
given criteria. SNR n , i r = .times. ( h n , i r ) H .times. ( h n
, l r .function. ( h n , l r ) H + .sigma. 2 .times. I ) - l
.times. h n , i r .times. i , j = 2 , i .noteq. j V n opt = .times.
arg .times. .times. max l .times. ( max i .times. ( SNR n , i r ) )
##EQU69##
[0268] Referring now to FIG. 45, shown is an example of a system in
which the receiver sets up criteria for searching. Much of this
example is similar to the Givens feedback system of described
above, and this material in common will not be repeated. After
constructing the Givens matrices at 328, the set up receiver
criteria 360 are used to search for a minimum. Examples of receiver
criteria have been set out above, but other examples can
alternatively be employed. The searching involves performing
hypotheses testing against a set or dictionary of theta, c
permutations 362 using the search minimum step 364. Once the
minimum has been found, the parameters representing the minimum are
selected at 366 and fed back on the MTMO channel 336 as before.
[0269] In any of the examples above, the Givens decomposition can
be computed using any appropriate method. Iterative approaches may
for example be employed such as Cholsky factorization and reverse
order multiplication. These methods are known in the art and will
not be described further here. It is noted that Givens
decomposition is about 10% of the complexity of SVD computing
complexity. This can be seen from the following table comparing the
complexity of SVD versus Givens as a function of the number of
transmit antennas, and the complexity is measured in terms of
multiply and add operations. TABLE-US-00018 COMPLEXITY ANTENNAS SVD
GIVENS GIVENS/SVD 1 21 0 2 168 12 7.14% 3 567 54 9.52% 4 1344 14
10.71% 5 2625 300 11.43% 6 4536 540 11.90% 7 7203 882 12.24% 8
10752 1344 12.50%
[0270] The following table shows a comparison of the complexity of
Givens based feedback as opposed to Hausholder based feedback as a
function of the number of the transmit antennas. It can be see that
the Hausholder approach is significantly more complex.
TABLE-US-00019 # of Givens Hausholder transmit Complex Complex
Complex Complex Complex Comple antennas Division Multiply Add
Comparison Multiply Add Storag 3 3 30 30 32 259 160 32 4 4 56 56 64
771 672 64 6 6 132 132 256 3843 3744 256 8 8 240 240 1024 20227
20128 1024 12 12 552 552 16384 413443 413344 16384
Example of Codebook Construction for Vector Pre-Coding
[0271] An example of Vector Pre-coding (code-book construction) for
purposes of context and comparison will now be described. The
cross-correlation of the codeword in this example has a block
circulant structure. The diagonal rotation matrix Q is defined as:
Q = [ e j .times. 2 .times. .pi. L .times. u 1 0 0 e j .times. 2
.times. .pi. L .times. u L ] .times. u 1 U .function. [ u 1 u L ]
##EQU70## Matrix P, is selected from sub-matrixes of DET matrix as
P 1 = [ d c 1 d c N 1 ] ##EQU71## ( D N l ) m .times. n = [ e j
.times. 2 .times. .pi. N l .times. ( m - 1 ) .times. ( n - 1 ) ] m
.times. n D = [ d 1 d N l ] ##EQU71.2## The codebook can be
constructed as: P.sub.l=Q.sup.lP.sub.l l=2,3 . . . L
[0272] The codebook can be optimized by choosing rotation matrix Q
indexes c=.left brkt-bot.c.sub.1 . . . c.sub.M.sub.1.right
brkt-bot. The code book can be optimized by choosing rotation
matrix Q indexes u=.left brkt-bot.u.sub.1 . . . c.sub.N.sub.1.right
brkt-bot. By exhaustive search of the codebook we have l opt = arg
.times. min l .di-elect cons. { 1 , 2 , .times. .times. , L }
.times. MSE .times. { E s N o .times. tr .function. ( I M + E s N o
.times. N r .times. P l H .times. H H .times. HP l ) - 1 }
##EQU72## Examples of Codebook Construction for Matrix
Pre-Coding
[0273] An overview of Matrix Pre-coding (column by column vector
quantize channel-1) for purposes of context and comparison will now
be described. Preferably, the V matrix is quantized column by
column and recursively. STEP-0: Denote the beam-forming matrix as V
= [ v 11 v 12 v 12 v 21 v 22 v 23 v 31 v 32 v 33 v 41 v 42 v 43 ]
##EQU73## STEP-1: Quantize the first column of V denoted as v.sub.1
as follows. {circumflex over (v)}=arg
max.sub.u.epsilon..sub.1.parallel.u.sup.Hv.sub.1.parallel. where
C.sub.1 is a codebook containing unit 4-vectors for quantization.
{circumflex over (v)}.sub.1 has the maximum inner product among all
unit vectors in the codebook. STEP-2: Compute Hausholder reflection
matrix as follows F 1 = I - 2 w 1 2 .times. w 1 .times. w 1 H
##EQU74## where .phi..sub.1 is the phase of v.sub.11 F 1 .times. V
= [ e j.PHI. 1 0.0 0.0 0.0 v ^ 11 v ^ 12 0.0 v ^ 21 v ^ 22 0.0 v ^
31 v ^ 32 ] .times. .times. where .times. .times. v 2 = [ v ^ 11 v
^ 12 v ^ 21 v ^ 22 v ^ 31 v ^ 32 ] ##EQU75## where two properties
are employed to get the result, i.e. {circumflex over (v)}.sub.11
is real and the unitary property of V Since both F.sub.1 and V are
unitary, V.sub.2 is unitary. From STEP-2, we see that the size of
V.sub.2 is 3 by 2 and it is reduced from that of V.sub.1 by one on
both the row and column dimensions. STEP-3: Quantize the first
column of V.sub.2 denoted as v.sub.2, using another codebook of
unit 3-vectors, whose first element of each codeword is real.
STEP-4: Construct a Hausholder reflection matrix F.sub.2 STEP-5:
Multiply F.sub.2 with V.sub.2 as follows. F 2 .times. V 2 = [ e
j.PHI. 2 0.0 0.0 v ~ 11 0.0 v ~ 21 ] .times. .times. where .times.
.times. v 3 = [ v ~ 11 v ~ 21 ] ##EQU76## The reconstruction of the
beam-forming matrix V is as follows: STEP-0; Two vectors, v.sub.3
and v.sub.2, are reconstructed using the feedback quantization
indexes and the corresponding 2-vector and 3-vector codebooks.
STEP-1: Compute a Hausholder matrix using the reconstructed V.sub.2
as F 2 = I - 2 w 2 .times. ww H ##EQU77## where w={circumflex over
(v)}.sub.2-e.sub.1 and {circumflex over (v)}.sub.2 is the
reconstructed 3-vector; F.sub.2 can be stored beforehand to reduce
computation. STEP-2: V.sub.2 can be reconstructed as V ^ 2 = F 2
.function. [ 1 0 0 0 v ^ 3 ] ##EQU78## STEP-3; we reconstruct the
first column of V using the quantization index and compute a
Hausholder matrix as F 1 = I - 2 w 2 .times. ww H ##EQU79## where
w={circumflex over (v)}.sub.1-e.sub.1 and {circumflex over
(v)}.sub.1 is the reconstructed first column of V. STEP-4 the
beam-forming matrix V is given by V ^ = F 1 .function. [ 1 0 0 0 0
V ^ 2 0 ] ##EQU80##
[0274] The codebook is constructed such that the codeword vectors
distribute on the n-dimension complex unit sphere uniformly.
Additionally, the firs: element of each codeword is set to be real
for the next step.
[0275] The Hausholder matrix can be computed and stored beforehand
for small codebooks. Even in the case that there is no quantization
error, the reconstructed matrix could be different from the
original V by a global phase on each column and this is fine with
closed loop MIMO.
Example of 2-Transmit Antenna Codebook for Givens Feedback
[0276] Pre-design the rotation matrix for 2 transmit antennas, and
then the rotation matrix is parameterized. A set of parameterized
rotation matrixes serves as codebook. V n 1 , n 2 l = [ e j.PHI. n
2 .times. cos .times. .times. .theta. n 1 - e j.PHI. n 2 .times.
sin .times. .times. .theta. n 1 sin .times. .times. .theta. n 1 cos
.times. .times. .theta. n 1 ] ##EQU81## .PHI. n 2 = 2 .times. .pi.
.times. .times. n 2 N 2 , n 2 = 0 , 1 .times. .times. .times.
.times. N 2 - 1 ##EQU81.2## .theta. n 1 = 2 .times. .pi. .times.
.times. n 1 N 1 , n 1 = 0 , 1 .times. .times. .times. .times. N 1 -
1 ##EQU81.3## Feedback Differential Codebook Index
[0277] According to an embodiment of the invention a differential
index feedback is provided as illustrated pictorially in FIG. 48.
The differential index is representative of the sub-space
searching. According to this embodiment the channel will not change
very fast. Therefore, the indices of consecutive feedbacks will not
be far from the previous feed-back.
Avoiding the Impact of Ageing
[0278] A MIMO feedback channel ageing algorithm in accordance with
an embodiment of the invention will now be described, in which
receiver ageing beam-former correction is utilized.
[0279] For a mobile MIMO channel the channel matrix may be
time-varying: Time-0.fwdarw.H.sub.0=U.sub.0D.sub.0V.sub.0' Time
1.fwdarw.H.sub.1=U.sub.1D.sub.1V.sub.1' The beam forming matrix
V.sub.0 sent to the transmitter may already be old by the time it
is utilized for transmission. However, the receiver may still be
able to compute the latest receiver beam forming matrix U.sub.0 as
follows:
U.sub.1'y=U.sub.1'U.sub.1D.sub.1V.sub.1'V.sub.0's+U.sub.1'n=D.sub.1V.sub.-
1'V.sub.0s+n This may prevent the ageing impact at receiver side,
the ageing impact potentially causing the inter-antenna
interference. In the above example, at the receiver the received
sequence y has been multiplied by the latest beam-forming matrix
U.sub.1, rather than use U.sub.0 that would have been used with the
channel information at time 0. In the result, the product
V.sub.1'V.sub.0 or likely be close to the identity matrix and can
be ignored; alternatively given that both V.sub.1' and V.sub.0 are
known at the receiver, the effect of this can also be divided out
at the receiver. This would increase the complexity however.
Physical Layer Design
[0280] A CQICH Support of Differential Encoding Algorithm in
accordance with an embodiment of the invention will now be
described. For OFDM systems, preferably differential encoding is
used to cross multiple sub-carriers to feedback the vector index or
other feedback information. The following is an example of a
mini-tile modulation scheme for sending back a vector index of zero
or one. Preferably this would be transmitted on two sub-carriers.
In this case, two different phases are transmitted for vector index
zero, and a different arrangement of the same phases is sent for
vector index one. TABLE-US-00020 Vector index M.sub.n,8,
M.sub.n,.theta.m+1 0 P0, P1 1 P1 ,P0 P .times. .times. 0 = exp
.times. .times. j .times. .times. .pi. 4 ##EQU82## P .times.
.times. 1 = exp .times. .times. j .times. .times. 3 .times. .pi. 4
##EQU83##
[0281] Another example is shown in FIG. 46, this being a feedback
approach that allows the transmission of one of eight different
vector indexes can be used to transmit up to 3-bits of information.
The vector index might for example be an index of codebook, or
alternatively the raw bits might be used to indicated feedback
Information, be it channel feedback or Givens feedback or unitary
matrix feedback. In the example shown, a tile generally indicated
at 370 is four sub-carriers by three symbols, and the four corner
sub-carrier locations in the tile are used for transmitting pilot
information. The remaining eight sub-carrier locations over the
three symbols are used to transmit the vector index. The encoding
used for the index is indicated in the table generally indicated at
372. In this example, four different phases are used, and a
codeword consisting of eight phases is used to represent each of
the potential vector indices. The receiver of the feedback
information can then recover the phase information and perform a
correlation to identify the transmitted vector index and thereby
recover the feedback information.
[0282] Another example is shown in FIG. 47 for the case where a
receiver has two antennas. In this case, pilot symbols are sent in
different locations for the two antennas, but identical feedback
information is sent on the remaining eight sub-carriers, the same
as in FIG. 46. For a given antennae is transmitting its pilot
symbols.
[0283] In yet another example, shown in FIG. 48, two different
users are feeding back the two patterns indicated at 380,382
respectively. Similar to the two antenna case of FIG. 47, each of
the two users transmits pilots in different locations within the
tile so as to not interfere with the other users pilots. In this
case, each user transmits eight sub-carrier locations containing
the vector index. The data transmitted by the two users will be
different, and the receiver of the feedback information will need
to distinguish between the two. According to an embodiment of the
invention STTD antenna assignment with power weighting may be used.
This reduces inter-code interference and reduces the feedback
bandwidth required.
[0284] FIGS. 49 and 50 shows a concatenation of STC/MIMO with a
beam-former in accordance with an embodiment of the invention.
[0285] Each embodiment is generalizable to an arbitrary number of
sub-carriers and/or an arbitrary number of transmit
antennas/receive antennas as will be apparent to one skilled in the
art. Embodiments provide transmitters adapted to generate signals
containing the disclosed transmit code-sets/sub-carrier allocations
methods of transmitting such signals, receivers adapted to receive
such transmissions, and methods of receiving and decoding such
signals.
[0286] Numerous modifications and variations of the present
invention are possible in light of the above teachings. It is
therefore to be understood that within the scope of the appended
claims, the invention may be practised otherwise than as
specifically described herein.
* * * * *