U.S. patent application number 11/975607 was filed with the patent office on 2008-02-21 for systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels.
Invention is credited to Ismail Adnan Lakkis.
Application Number | 20080043653 11/975607 |
Document ID | / |
Family ID | 34215349 |
Filed Date | 2008-02-21 |
United States Patent
Application |
20080043653 |
Kind Code |
A1 |
Lakkis; Ismail Adnan |
February 21, 2008 |
Systems and methods for wireless communication over a wide
bandwidth channel using a plurality of sub-channels
Abstract
A method of communicating over a wideband communication channel
divided into a plurality of sub-channels comprises dividing a
single serial message intended for one of the plurality of
communication devices into a plurality of parallel messages,
encoding each of the plurality of parallel messages onto at least
some of the plurality of sub-channels, and transmitting the encoded
plurality of parallel messages to the communication device over the
wideband communication channel. This Abstract is provided for the
sole purpose of complying with the Abstract requirement rules that
allow a reader to quickly ascertain the subject matter of the
disclosure contained herein. This Abstract is submitted with the
explicit understanding that it will not be used to interpret or to
limit the scope or the meaning of the claims.
Inventors: |
Lakkis; Ismail Adnan; (San
Diego, CA) |
Correspondence
Address: |
PULSE-LINK, INC.
1969 KELLOGG AVENUE
CARLSBAD
CA
92008
US
|
Family ID: |
34215349 |
Appl. No.: |
11/975607 |
Filed: |
October 18, 2007 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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10010601 |
Dec 6, 2001 |
7289494 |
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11975607 |
Oct 18, 2007 |
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Current U.S.
Class: |
370/310 |
Current CPC
Class: |
H04B 1/71632 20130101;
H04B 1/71635 20130101 |
Class at
Publication: |
370/310 |
International
Class: |
H04B 7/00 20060101
H04B007/00 |
Claims
1. A transmitter, comprising: a serial-to-parallel transformer
configured to transform a single serial bit stream comprising
messages for a plurality of communication device into a plurality
of bit streams; a modulator configured to perform time division
modulation or frequency division modulation on each of the
plurality of bit streams; and a summer configured to sum the
plurality of bit streams into a single transmit signal.
2. The transmitter of claim 1, further comprising: a scrambler
configured to scramble each of the plurality of bit streams; an
encoder configured to encode each of the plurality of bit streams;
and an interleaver configured to interleave each of the plurality
of bit streams.
3. The transmitter of claim 2, further comprising a symbol mapper
configured to perform symbol mapping on each of the plurality of
bit streams.
4. The transmitter of claim 3, wherein the type of symbol mapping
used on each of the plurality of bit streams is programmable based
on signal to interference measurements provided by each of the
plurality of communication devices.
5. The transmitter of claim 1, wherein the modulator is further
configured to filter each of the plurality of bit streams to
provide a required pulse shaping for each of the plurality of bit
streams.
6. The transmitter of claim 1, further comprising a RF transmitter
configured to transmit the single transmit signal over a
communication channel comprising sub-channels corresponding to each
of the plurality of bit streams.
7. A transmitter, comprising: a serial-to-parallel transformer
configured to transform a single serial bit stream with an overall
data rate into a plurality of bit streams; a modulator configured
to perform time division modulation or frequency division
modulation on each of the plurality of bit streams; a rate
controller configured to adjust the overall data rate; and a summer
configured to sum the plurality of bit streams into a single
transmit signal.
8. The transmitter of claim 7, wherein the rate controller is
configured to adjust the overall data rate by spreading the
plurality of bit streams in the frequency domain.
9. The transmitter of claim 7, wherein the rate controller is
configured to adjust the overall data rate down by encoding n data
streams onto m channels, where m>n.
10. The transmitter of claim 9, wherein the rate controller is
further configured to scramble the duplicate bit streams.
11. The transmitter of claim 7, wherein the rate controller is
configured to adjust the overall data rate by: splitting each of
the plurality of bit streams into a first bit stream and a second
bit stream, and delaying each second bit stream by half a symbol
period relative to each first bit stream; and wherein the modulator
is further configured to: filter each first and second bit stream,
and sum each filtered first bit stream and each filtered second bit
stream together.
12. The transmitter of claim 7, wherein the rate controller is
configured so that it can be programmed to increase the overall
data rate or decrease the overall data rate.
13. A transmitter, comprising: a serial-to-parallel transformer
configured to transform a single serial bit stream into a plurality
of bit streams; a modulator configured to perform time division
modulation or frequency division modulation on each of the
plurality of bit streams; a frequency encoder configured to encode
information from more than one of the plurality of bit streams onto
each of a plurality of frequencies corresponding to a plurality of
sub-channels; and a summer configured to sum the plurality of bit
streams into a single transmit signal.
14. The transmitter of claim 13, further comprising a frequency
encoder that is configured so that a type of frequency encoding
performed is programmable.
15. A transmitter, comprising: a serial-to-parallel transformer
configured to transform a single serial bit stream with an overall
data rate into a plurality of bit streams; a modulator configured
to perform time division modulation or frequency division
modulation on each of the plurality of bit streams; a rate
controller configured to adjust the overall data rate, and a
frequency encoder configured to encode information from more than
one of the plurality of bit streams onto each of a plurality of
frequencies corresponding to a plurality of sub-channels; and a
summer configured to sum the plurality of bit streams into a single
transmit signal.
16. The transmitter of claim 15, wherein the rate controller and
frequency encoder are configured to be turned on and of as
required.
17. The transmitter of claim 15, wherein the rate controller is
configured so that it can be programmed to increase the overall
data rate or decrease the overall data rate.
18. The transmitter of claim 15, wherein the frequency encoder is
configured so that the type of frequency encoding performed is
programmable.
19. A receiver, comprising: a RF receiver configured to receive a
transmit signal comprising a plurality of bit streams, each bit
stream comprising a frequency offset; a frequency shifter
configured to shift the frequency of each of the plurality of bit
streams so as to remove the corresponding frequency offset; a
filter configured to filter each of plurality of bit streams so as
to apply a correct pulse shaping to each of the plurality of bit
streams; an equalizer configured to perform equalization on each of
the plurality of bit streams; and a parallel-to-serial transformer
configured transform the plurality of bit streams into a single
serial bit stream.
20. The receiver of claim 19, wherein the received transmit signal
comprises an overall bandwidth divided into a plurality of
sub-channels, and wherein the frequency offsets of each of the
plurality of bit streams correspond to a particular
sub-channel.
21. The receiver of claim 20, wherein the frequency shifter is
programmed to only shift the frequency for those of the plurality
of bit streams that correspond to sub-channels that have been
assigned to a communication device comprising the receiver.
22. The receiver of claim 19, further comprising: a descrambler for
descrambling each of the plurality of bit streams; a decoder for
decoding each of the plurality of bit streams; and a deinterleaver
for deinterleaving each of the plurality of bit streams.
23. The receiver of claim 19, wherein the equalizer is a time
domain equalizer or a frequency domain equalizer.
24. The receiver of claim 19, wherein the received transmit signal
comprises an overall bandwidth that is divided into a plurality of
sub-channels, and wherein the receiver further comprises an
estimator configured to estimate a signal to noise and interference
ratio for each subchannel.
25. The receiver of claim 19, wherein the received transmit signal
comprises an overall bandwidth that is divided into a plurality of
sub-channels, and wherein the receiver further comprises an
estimator configured to estimate the channel impulse response for
each sub-channel.
Description
[0001] This application claims priority under 35 U.S.C. .sctn. 120
as a continuation of U.S. patent application Ser. No. 10/010,601,
filed Dec. 6, 2001, entitled "SYSTEMS AND METHODS FOR WIRELESS
COMMUNICATION OVER A WIDE BANDWIDTH CHANNEL USING A PLURALITY OF
SUB-CHANNELS."
[0002] This application may be related to the following U.S. patent
applications: Ser. No. 10/961,592, filed Oct. 8, 2004; Ser. No.
10/961,614, filed Oct. 8, 2004; Ser. No. 10/963,026, filed Oct. 12,
2004; Ser. No. 10/962,935, filed Oct. 12, 2004; Ser. No.
10/964,482, filed Oct. 13, 2004; Ser. No. 10/964,336, filed Oct.
13, 2004; Ser. No. 10/984,436, filed Nov. 8, 2004; Ser. No.
10/985,977, filed Nov. 9, 2004; Ser. No. 10/985,861, filed Nov. 10,
2004; Ser. No. 10/988,373, filed Nov. 12, 2004; Ser. No.
11/055,525, filed Feb. 9, 2005; Ser. No. 10/120,456, filed Apr. 9,
2002; Ser. No. 10/810,948, filed Mar. 25, 2004; Ser. No.
10/811,223, filed Mar. 26, 2004; Ser. No. 10/934,316, filed Sep. 3,
2004; Ser. No. 10/948,099, filed Sep. 23, 2004; Ser. No.
10/810,410, filed Mar. 26, 2004; Ser. No. 10/952,458, filed Sep.
27, 2004; and Ser. No. 11/332,946, filed Jan. 17, 2006.
BACKGROUND OF THE INVENTION
[0003] 1. Field of the Invention
[0004] The invention relates generally to communications, and more
particularly to systems and methods for high data rate
communications.
[0005] 2. Background
[0006] Wireless communication systems are proliferating at the Wide
Area Network (WAN), Local Area Network (LAN), and Personal Area
Network (PAN) levels. These wireless communication systems use a
variety of techniques to allow simultaneous access to multiple
users. The most common of these techniques are Frequency Division
Multiple Access (FDMA), which assigns specific frequencies to each
user, Time Division Multiple Access (TDMA), which assigns
particular time slots to each user, and Code Division Multiple
Access (CDMA), which assigns specific codes to each user. But these
wireless communication systems and various modulation techniques
are afflicted by a host of problems that limit the capacity and the
quality of service provided to the users. The following paragraphs
briefly describe a few of these problems for the purpose of
illustration.
[0007] One problem that can exist in a wireless communication
system is multipath interference. Multipath interference, or
multipath, occurs because some of the energy in a transmitted
wireless signal bounces off of obstacles, such as buildings or
mountains, as it travels from source to destination. The obstacles
in effect create reflections of the transmitted signal and the more
obstacles there are, the more reflections they generate. The
reflections then travel along their own transmission paths to the
destination (or receiver). The reflections will contain the same
information as the original signal; however, because of the
differing transmission path lengths, the reflected signals will be
out of phase with the original signal. As a result, they will often
combine destructively with the original signal in the receiver.
This is referred to as fading. To combat fading, current systems
typically try to estimate the multipath effects and then compensate
for them in the receiver using an equalizer. In practice, however,
it is very difficult to achieve effective multipath
compensation.
[0008] A second problem that can affect the operation of wireless
communication systems is interference from adjacent communication
cells within the system. In FDMA/TDMA systems, this type of
interference is prevent through a frequency reuse plan. Under a
frequency reuse plan, available communication frequencies are
allocated to communication cells within the communication system
such that the same frequency will not be used in adjacent cells.
Essentially, the available frequencies are split into groups. The
number of groups is termed the reuse factor. Then the communication
cells are grouped into clusters, each cluster containing the same
number of cells as there are frequency groups. Each frequency group
is then assigned to a cell in each cluster. Thus, if a frequency
reuse factor of 7 is used, for example, then a particular
communication frequency will be used only once in every seven
communication cells. Thus, in any group of seven communication
cells, each cell can only use 1/7.sup.th of the available
frequencies, i.e., each cell is only able to use 1/7.sup.th of the
available bandwidth.
[0009] In a CDMA communication system, each cell uses the same
wideband communication channel. In order to avoid interference with
adjacent cells, each communication cell uses a particular set of
spread spectrum codes to differentiate communications within the
cell from those originating outside of the cell. Thus, CDMA systems
preserve the bandwidth in the sense that they avoid reuse planning.
But as will be discussed, there are other issues that limit the
bandwidth in CDMA systems as well.
[0010] Thus, in overcoming interference, system bandwidth is often
sacrificed. Bandwidth is becoming a very valuable commodity as
wireless communication systems continue to expand by adding more
and more users. Therefore, trading off bandwidth for system
performance is a costly, albeit necessary, proposition that is
inherent in all wireless communication systems.
[0011] The foregoing are just two examples of the types of problems
that can affect conventional wireless communication systems. The
examples also illustrate that there are many aspects of wireless
communication system performance that can be improved through
systems and methods that, for example, reduce interference,
increase bandwidth, or both.
[0012] Not only are conventional wireless communication systems
effected by problems, such as those described in the preceding
paragraphs, but also different types of systems are effected in
different ways and to different degrees. Wireless communication
systems can be split into three types: 1) line-of-sight systems,
which can include point-to-point or point-to-multipoint systems; 2)
indoor non-line of sight systems; and 3) outdoor systems such as
wireless WANs. Line-of-sight systems are least affected by the
problems described above, while indoor systems are more affected,
due for example to signals bouncing off of building walls. Outdoor
systems are by far the most affected of the three systems. Because
these types of problems are limiting factors in the design of
wireless transmitters and receivers, such designs must be tailored
to the specific types of system in which it will operate. In
practice, each type of system implements unique communication
standards that address the issues unique to the particular type of
system. Even if an indoor system used the same communication
protocols and modulation techniques as an outdoor system, for
example, the receiver designs would still be different because
multipath and other problems are unique to a given type of system
and must be addressed with unique solutions. This would not
necessarily be the case if cost efficient and effective
methodologies can be developed to combat such problems as described
above that build in programmability so that a device can be
reconfigured for different types of systems and still maintain
superior performance.
SUMMARY OF THE INVENTION
[0013] In order to combat the above problems, systems and methods
of ultra-wideband communication are provided. In one ultra-wideband
communication method, a communication channel is divided into a
plurality of non-overlapping sub-channels by dividing a single
serial message intended for an ultra-wideband communication device
into a plurality of parallel messages. Each of the plurality of
parallel messages are then encoded onto at least some of the
plurality of sub-channels, and then transmitted.
[0014] Other aspects, advantages, and novel features of the
invention will become apparent from the following Detailed
Description of Preferred Embodiments, when considered in
conjunction with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
[0015] Preferred embodiments of the present inventions taught
herein are illustrated by way of example, and not by way of
limitation, in the figures of the accompanying drawings, in
which:
[0016] FIG. 1 is a diagram illustrating an example embodiment of a
wideband channel divided into a plurality of sub-channels in
accordance with the invention;
[0017] FIG. 2 is a diagram illustrating the effects of multipath in
a wireless communication system;
[0018] FIG. 3 is a diagram illustrating another example embodiment
of a wideband communication channel divided into a plurality of
sub-channels in accordance with the invention;
[0019] FIG. 4 is a diagram illustrating the application of a
roll-off factor to the sub-channels of FIGS. 1 and 2;
[0020] FIG. 5A is a diagram illustrating the assignment of
sub-channels for a wideband communication channel in accordance
with the invention;
[0021] FIG. 5B is a diagram illustrating the assignment of time
slots for a wideband communication channel in accordance with the
invention;
[0022] FIG. 6 is a diagram illustrating an example embodiment of a
wireless communication in accordance with the invention;
[0023] FIG. 7 is a diagram illustrating the use of synchronization
codes in the wireless communication system of FIG. 5 in accordance
with the invention;
[0024] FIG. 8 is a diagram illustrating a correlator that can be
used to correlate synchronization codes in the wireless
communication system of FIG. 5;
[0025] FIG. 9 is a diagram illustrating synchronization code
correlation in accordance with the invention;
[0026] FIG. 10 is a diagram illustrating the cross-correlation
properties of synchronization codes configured in accordance with
the invention;
[0027] FIG. 11 is a diagram illustrating another example embodiment
of a wireless communication system in accordance with the
invention;
[0028] FIG. 12A is a diagram illustrating how sub-channels of a
wideband communication channel according to the present invention
can be grouped in accordance with the present invention;
[0029] FIG. 12B is a diagram illustrating the assignment of the
groups of sub-channels of FIG. 12A in accordance with the
invention;
[0030] FIG. 13 is a diagram illustrating the group assignments of
FIG. 12B in the time domain;
[0031] FIG. 14 is a flow chart illustrating the assignment of
sub-channels based on SIR measurements in the wireless
communication system of FIG. 11 in accordance with the
invention;
[0032] FIG. 15 is a logical block diagram of an example embodiment
of transmitter configured in accordance with the invention;
[0033] FIG. 16 is a logical block diagram of an example embodiment
of a modulator configured in accordance with the present invention
for use in the transmitter of FIG. 15;
[0034] FIG. 17 is a diagram illustrating an example embodiment of a
rate controller configured in accordance with the invention for use
in the modulator of FIG. 16;
[0035] FIG. 18 is a diagram illustrating another example embodiment
of a rate controller configured in accordance with the invention
for use in the modulator of FIG. 16;
[0036] FIG. 19 is a diagram illustrating an example embodiment of a
frequency encoder configured in accordance with the invention for
use in the modulator of FIG. 16;
[0037] FIG. 20 is a logical block diagram of an example embodiment
of a TDM/FDM block configured in accordance with the invention for
use in the modulator of FIG. 16;
[0038] FIG. 21 is a logical block diagram of another example
embodiment of a TDM/FDM block configured in accordance with the
invention for use in the modulator of FIG. 16;
[0039] FIG. 22 is a logical block diagram of an example embodiment
of a frequency shifter configured in accordance with the invention
for use in the modulator of FIG. 16;
[0040] FIG. 23 is a logical block diagram of a receiver configured
in accordance with the invention;
[0041] FIG. 24 is a logical block diagram of an example embodiment
of a demodulator configured in accordance with the invention for
use in the receiver of FIG. 23;
[0042] FIG. 25 is a logical block diagram of an example embodiment
of an equalizer configured in accordance with the present invention
for use in the demodulator of FIG. 24; and
[0043] FIG. 26 is a logical block diagram of an example embodiment
of a wireless communication device configured in accordance with
the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
1. Introduction
[0044] In the following paragraphs, the present invention will be
described in detail by way of example with reference to the
attached drawings. While this invention is capable of embodiment in
many different forms, there is shown in the drawings and will
herein be described in detail specific embodiments, with the
understanding that the present disclosure is to be considered as an
example of the principles of the invention and not intended to
limit the invention to the specific embodiments shown and
described. That is, throughout this description, the embodiments
and examples shown should be considered as exemplars, rather than
as limitations on the present invention. As used herein, the
"present invention" refers to any one of the embodiments of the
invention described herein, and any equivalents. Furthermore,
reference to various feature(s) of the "present invention"
throughout this document does not mean that all claimed embodiments
or methods must include the referenced feature(s).
[0045] In order to improve wireless communication system
performance and allow a single device to move from one type of
system to another, while still maintaining superior performance,
the systems and methods described herein provide various
communication methodologies that enhance performance of
transmitters and receivers with regard to various common problems
that afflict such systems and that allow the transmitters and/or
receivers to be reconfigured for optimal performance in a variety
of systems. Accordingly, the systems and methods described herein
define a channel access protocol that uses a common wideband
communication channel for all communication cells. The wideband
channel, however, is then divided into a plurality of sub-channels.
Different sub-channels are then assigned to one or more users
within each cell. But the base station, or service access point,
within each cell transmits one message that occupies the entire
bandwidth of the wideband channel. Each user's communication device
receives the entire message, but only decodes those portions of the
message that reside in sub-channels assigned to the user. For a
point-to-point system, for example, a single user may be assigned
all sub-channels and, therefore, has the full wide band channel
available to them. In a wireless WAN, on the other hand, the
sub-channels may be divided among a plurality of users.
[0046] In the descriptions of example embodiments that follow,
implementation differences, or unique concerns, relating to
different types of systems will be pointed out to the extent
possible. But it should be understood that the systems and methods
described herein are applicable to any type of communication
systems. In addition, terms such as communication cell, base
station, service access point, etc. are used interchangeably to
refer to the common aspects of networks at these different
levels.
[0047] To begin illustrating the advantages of the systems and
methods described herein, one can start by looking at the multipath
effects for a single wideband communication channel 100 of
bandwidth B as shown in FIG. 1. Communications sent over channel
100 in a traditional wireless communication system will comprise
digital data bits, or symbols, that are encoded and modulated onto
a RF carrier that is centered at frequency f.sub.c and occupies
bandwidth B. Generally, the width of the symbols (or the symbol
duration) T is defined as 1/B. Thus, if the bandwidth B is equal to
100 MHz, then the symbol duration T is defined by the following
equation: T=1/B= 1/100 megahertz(MHZ)=10 nanoseconds(ns). (1)
[0048] When a receiver receives the communication, demodulates it,
and then decodes it, it will recreate a stream 104 of data symbols
106 as illustrated in FIG. 2. But the receiver will also receive
multipath versions 108 of the same data stream. Because multipath
data streams 108 are delayed in time relative to the data stream
104 by delays d1, d2, d3, and d4, for example, they may combine
destructively with data stream 104.
[0049] A delay spread d.sub.s is defined as the delay from
reception of data stream 104 to the reception of the last multipath
data stream 108 that interferes with the reception of data stream
104. Thus, in the example illustrated in FIG. 2, the delay spread
d.sub.s is equal to delay d4. The delay spread d.sub.s will vary
for different environments. An environment with a lot of obstacles
will create a lot of multipath reflections. Thus, the delay spread
d.sub.s will be longer. Experiments have shown that for outdoor WAN
type environments, the delay spread d.sub.s can be as long as 20
microseconds. Using the 10 ns symbol duration of equation (1), this
translates to 2000 symbols. Thus, with a very large bandwidth, such
as 100 MHz, multipath interference can cause a significant amount
of interference at the symbol level for which adequate compensation
is difficult to achieve. This is true even for indoor environments.
For indoor LAN type systems, the delay spread d.sub.s is
significantly shorter, typically about 1 microsecond. For a 10 ns
symbol duration, this is equivalent to 100 symbols, which is more
manageable but still significant.
[0050] By segmenting the bandwidth B into a plurality of
sub-channels 202, as illustrated in FIG. 2, and generating a
distinct data stream for each sub-channel, the multipath effect can
be reduced to a much more manageable level. For example, if the
bandwidth b of each sub-channel 202 is 500 KHz, then the symbol
duration is 2 microseconds. Thus, the delay spread d.sub.s for each
sub-channel is equivalent to only 10 symbols (outdoor) or half a
symbol (indoor). Thus, by breaking up a message that occupies the
entire bandwidth B into discrete messages, each occupying the
bandwidth b of sub-channels 202, a very wideband signal that
suffers from relatively minor multipath effects is created.
[0051] Before discussing further features and advantages of using a
wideband communication channel segmented into a plurality of
sub-channels as described, certain aspects of the sub-channels will
be explained in more detail. Referring back to FIG. 3, the overall
bandwidth B is segmented into N sub-channels center at frequencies
f.sub.o to f.sub.N-1. Thus, the sub-channel 202 that is immediately
to the right of f.sub.c is offset from f.sub.c by b/2, where b is
the bandwidth of each sub-channel 202. The next sub-channel 202 is
offset by 3b/2, the next by 5b/2, and so on. To the left of
f.sub.c, each sub-channel 202 is offset by -b/2, -3b/2, -5b/2,
etc.
[0052] Preferably, sub-channels 202 are non-overlapping as this
allows each sub-channel to be processed independently in the
receiver. To accomplish this, a roll-off factor is preferably
applied to the signals in each sub-channel in a pulse-shaping step.
The effect of such a pulse-shaping step is illustrated in FIG. 2 by
the non-rectangular shape of the pulses in each sub-channel 202.
Thus, the bandwidth b of each sub-channel can be represented by an
equation such as the following: b=(1+r)/T; (2) [0053] Where r=the
roll-off factor; and [0054] T=the symbol duration.
[0055] Without the roll-off factor, i.e., b=1/T, the pulse shape
would be rectangular in the frequency domain, which corresponds to
a (sin x)/x function in the time domain. The time domain signal for
a (sin x)/x signal 400 is shown in FIG. 4 in order to illustrate
the problems associated with a rectangular pulse shape and the need
to use a roll-off factor.
[0056] As can be seen, main lobe 402 comprises almost all of signal
400. But some of the signal also resides in side lobes 404, which
stretch out indefinitely in both directions from main lobe 402.
Side lobes 404 make processing signal 400 much more difficult,
which increases the complexity of the receiver. Applying a roll-off
factor r, as in equation (2), causes signal 400 to decay faster,
reducing the number of side lobes 404. Thus, increasing the
roll-off factor decreases the length of signal 400, i.e., signal
400 becomes shorter in time. But including the roll-off factor also
decreases the available bandwidth in each sub-channel 202.
Therefore, r must be selected so as to reduce the number of side
lobes 404 to a sufficient number, e.g., 15, while still maximizing
the available bandwidth in each sub-channel 202.
[0057] Thus, the overall bandwidth B for communication channel 200
is given by the following equation: B=N(1+r)/T; (3) or B=M/T, (4)
Where M=(1+r)N. (5)
[0058] For efficiency purposes related to transmitter design, it is
preferable that r is chosen so that M in equation (5) is an
integer. Choosing r so that M is an integer allows for more
efficient transmitters designs using, for example, Inverse Fast
Fourier Transform (IFFT) techniques. Since M=N+N(r), and N is
always an integer, this means that r must be chosen so that N(r) is
an integer. Generally, it is preferable for r to be between 0.1 and
0.5. Therefore, if N is 16, for example, then 0.5 could be selected
for r so that N(r) is an integer. Alternatively, if a value for r
is chosen in the above example so that N(r) is not an integer, B
can be made slightly wider than M/T to compensate. In this case, it
is still preferable that r be chosen so that N(r) is approximately
an integer.
2. Example Embodiment of a Wireless Communication System
[0059] With the above in mind, FIG. 6 illustrates an example
communication system 600 comprising a plurality of cells 602 that
each use a common wideband communication channel to communicate
with communication devices 604 within each cell 602. The common
communication channel is a wideband communication channel as
described above. Each communication cell 602 is defined as the
coverage area of a base station, or service access point, 606
within the cell. One such base station 606 is shown for
illustration in FIG. 6. For purposes of this specification and the
claims that follow, the term base station will be used generically
to refer to a device that provides wireless access to the wireless
communication system for a plurality of communication devices,
whether the system is a line of sight, indoor, or outdoor
system.
[0060] Because each cell 602 uses the same communication channel,
signals in one cell 602 must be distinguishable from signals in
adjacent cells 602. To differentiate signals from one cell 602 to
another, adjacent base stations 606 use different synchronization
codes according to a code reuse plan. In FIG. 6, system 600 uses a
synchronization code reuse factor of 4, although the reuse factor
can vary depending on the application.
[0061] Preferably, the synchronization code is periodically
inserted into a communication from a base station 606 to a
communication device 604 as illustrated in FIG. 6. After a
predetermined number of data packets 702, in this case two, the
particular synchronization code 704 is inserted into the
information being transmitted by each base station 606. A
synchronization code is a sequence of data bits known to both the
base station 606 and any communication devices 604 with which it is
communicating. The synchronization code allows such a communication
device 604 to synchronize its timing to that of base station 606,
which, in turn, allows device 604 to decode the data properly.
Thus, in cell 1 (see lightly shaded cells 602 in FIG. 6), for
example, synchronization code 1 (SYNC1) is inserted into data
stream 706, which is generated by base station 606 in cell 1, after
every two packets 702; in cell 2 SYNC2 is inserted after every two
packets 702; in cell 3 SYNC3 is inserted; and in cell 4 SYNC4 is
inserted. Use of the synchronization codes is discussed in more
detail below.
[0062] In FIG. 5A, an example wideband communication channel 500
for use in communication system 600 is divided into 16 sub-channels
502, centered at frequencies f.sub.o to f.sub.15. A base station
606 at the center of each communication cell 602 transmits a single
packet occupying the whole bandwidth B of wideband channel 500.
Such a packet is illustrated by packet 504 in FIG. 5B. Packet 504
comprises sub-packets 506 that are encoded with a frequency offset
corresponding to one of sub-channels 502. Sub-packets 506 in effect
define available time slots in packet 504. Similarly, sub-channels
502 can be said to define available frequency bins in communication
channel 500. Therefore, the resources available in communication
cell 602 are time slots 506 and frequency bins 502, which can be
assigned to different communication devices 604 within each cell
602.
[0063] Thus, for example, frequency bins 502 and time slots 506 can
be assigned to 4 different communication devices 604 within a cell
602 as shown in FIG. 5. Each communication device 604 receives the
entire packet 504, but only processes those frequency bins 502
and/or timeslots 506 that are assigned to it. Preferably, each
device 604 is assigned non-adjacent frequency bins 502, as in FIG.
5A. This way, if interference corrupts the information in a portion
of communication channel 500, then the effects are spread across
all devices 604 within a cell 602. Hopefully, by spreading out the
effects of interference in this manner the effects are minimized
and the entire information sent to each device 604 can still be
recreated from the unaffected information received in other
frequency bins. For example, if interference, such as fading,
corrupted the information in bins f.sub.o-f.sub.4, then each user
1-4 loses one packet of data. But each user potentially receives
three unaffected packets from the other bins assigned to them.
Hopefully, the unaffected data in the other three bins provides
enough information to recreate the entire message for each user.
Thus, frequency diversity can be achieved by assigning non-adjacent
bins to each of multiple users.
[0064] Ensuring that the bins assigned to one user are separated by
more than the coherence bandwidth ensures frequency diversity. As
discussed above, the coherence bandwidth is approximately equal to
1/d.sub.s. For outdoor systems, where ds is typically 1
microsecond, 1/d.sub.s=1/1 microsecond=1 Mega Hertz (MHz). Thus,
the non-adjacent frequency bands assigned to a user are preferably
separated by at least 1 MHz. It is even more preferable, however,
if the coherence bandwidth plus some guard band to ensure
sufficient frequency diversity separate the non-adjacent bins
assigned to each user. For example, it is preferable in certain
implementations to ensure that at least 5 times the coherence
bandwidth, or 5 MHz in the above example, separates the
non-adjacent bins.
[0065] Another way to provide frequency diversity is to repeat
blocks of data in frequency bins assigned to a particular user that
are separated by more than the coherence bandwidth. In other words,
if 4 sub-channels 202 are assigned to a user, then data block a can
be repeated in the first and third sub-channels 202 and data block
b can be repeated in the second and fourth sub-channels 202,
provided the sub-channels are sufficiently separated in frequency.
In this case, the system can be said to be using a diversity length
factor of 2. The system can similarly be configured to implement
other diversity lengths, e.g., 3, 4, . . . , l.
[0066] It should be noted that spatial diversity can also be
included depending on the embodiment. Spatial diversity can
comprise transmit spatial diversity, receive spatial diversity, or
both. In transmit spatial diversity, the transmitter uses a
plurality of separate transmitters and a plurality of separate
antennas to transmit each message. In other words, each transmitter
transmits the same message in parallel. The messages are then
received from the transmitters and combined in the receiver.
Because the parallel transmissions travel different paths, if one
is affected by fading, the others will likely not be affected.
Thus, when they are combined in the receiver, the message should be
recoverable even if one or more of the other transmission paths
experienced severe fading.
[0067] Receive spatial diversity uses a plurality of separate
receivers and a plurality of separate antennas to receive a single
message. If an adequate distance separates the antennas, then the
transmission path for the signals received by the antennas will be
different. Again, this difference in the transmission paths will
provide imperviousness to fading when the signals from the
receivers are combined.
[0068] Transmit and receive spatial diversity can also be combined
within a system such as system 600 so that two antennas are used to
transmit and two antennas are used to receive. Thus, each base
station 606 transmitter can include two antennas, for transmit
spatial diversity, and each communication device 604 receiver can
include two antennas, for receive spatial diversity. If only
transmit spatial diversity is implemented in system 600, then it
can be implemented in base stations 606 or in communication devices
604. Similarly, if only receive spatial diversity is included in
system 600, then it can be implemented in base stations 606 or
communication devices 604.
[0069] The number of communication devices 604 assigned frequency
bins 502 and/or time slots 506 in each cell 602 is preferably
programmable in real time. In other words, the resource allocation
within a communication cell 602 is preferably programmable in the
face of varying external conditions, i.e., multipath or adjacent
cell interference, and varying requirements, i.e., bandwidth
requirements for various users within the cell. Thus, if user 1
requires the whole bandwidth to download a large video file, for
example, then the allocation of bins 502 can be adjust to provide
user 1 with more, or even all, of bins 502. Once user 1 no longer
requires such large amounts of bandwidth, the allocation of bins
502 can be readjusted among all of users 1-4.
[0070] It should also be noted that all of the bins assigned to a
particular user can be used for both the forward and reverse link.
Alternatively, some bins 502 can be assigned as the forward link
and some can be assigned for use on the reverse link, depending on
the implementation.
[0071] To increase capacity, the entire bandwidth B is preferably
reused in each communication cell 602, with each cell 602 being
differentiated by a unique synchronization code (see discussion
below). Thus, system 600 provides increased immunity to multipath
and fading as well as increased bandwidth due to the elimination of
frequency reuse requirements.
3. Synchronization
[0072] FIG. 8 illustrates an example embodiment of a
synchronization code correlator 800. When a device 604 in cell 1
(see FIG. 6), for example, receives an incoming communication from
the cell 1 base station 606, it compares the incoming data with
SYNC1 in correlator 800. Essentially, the device scans the incoming
data trying to correlate the data with the known synchronization
code, in this case SYNC1. Once correlator 800 matches the incoming
data to SYNC1 it generates a correlation peak 804 at the output.
Multipath versions of the data will also generate correlation peaks
806, although these peaks 806 are generally smaller than
correlation peak 804. The device can then use the correlation peaks
to perform channel estimation, which allows the device to adjust
for the multipath using an equalizer. Thus, in cell 1, if
correlator 800 receives a data stream comprising SYNC1, it will
generate correlation peaks 804 and 806. If, on the other hand, the
data stream comprises SYNC2, for example, then no peaks will be
generated and the device will essentially ignore the incoming
communication.
[0073] Even though a data stream that comprises SYNC2 will not
create any correlation peaks, it can create noise in correlator 800
that can prevent detection of correlation peaks 804 and 806.
Several steps can be taken to prevent this from occurring. One way
to minimize the noise created in correlator 800 by signals from
adjacent cells 602, is to configure system 600 so that each base
station 606 transmits at the same time. This way, the
synchronization codes can preferably be generated in such a manner
that only the synchronization codes 704 of adjacent cell data
streams, e.g., streams 708, 710, and 712, as opposed to packets 702
within those streams, will interfere with detection of the correct
synchronization code 704, e.g., SYNC1. The synchronization codes
can then be further configured to eliminate or reduce the
interference.
[0074] For example, the noise or interference caused by an
incorrect synchronization code is a function of the cross
correlation of that synchronization code with respect to the
correct code. The better the cross correlation between the two, the
lower the noise level. When the cross correlation is ideal, then
the noise level will be virtually zero as illustrated in FIG. 9 by
noise level 902. Therefore, a preferred embodiment of system 600
uses synchronization codes that exhibit ideal cross correlation,
i.e., zero. Preferably, the ideal cross correlation of the
synchronization codes covers a period 1 that is sufficient to allow
accurate detection of multipath 906 as well as multipath
correlation peaks 904. This is important so that accurate channel
estimation and equalization can take place. Outside of period 1,
the noise level 908 goes up, because the data in packets 702 is
random and will exhibit low cross correlation with the
synchronization code, e.g., SYNC1. Preferably, period l is actually
slightly longer then the multipath length in order to ensure that
the multipath can be detected.
[0075] a. Synchronization Code Generation
[0076] Conventional systems use orthogonal codes to achieve cross
correlation in correlator 800. In system 600 for example, SYNC1,
SYNC2, SYNC3, and SYNC4, corresponding to cells 1-4 (see lightly
shaded cells 602 of FIG. 5) respectively, will all need to be
generated in such a manner that they will have ideal cross
correlation with each other. In one embodiment, if the data streams
involved comprise high and low data bits, then the value "1" can be
assigned to the high data bits and "-1" to the low data bits.
Orthogonal data sequences are then those that produce a "0" output
when they are exclusively ORed (XORed) together in correlator 800.
The following example illustrates this point for orthogonal
sequences 1 and 2: sequence .times. .times. 1 .times. : 1 1 - 1 1
sequence .times. .times. 2 .times. : 1 1 1 - 1 .times. 1 1 - 1 - 1
= 0 ##EQU1##
[0077] Thus, when the results of XORing each bit pair are added,
the result is "0".
[0078] But in system 600, for example, each code must have ideal,
or zero, cross correlation with each of the other codes used in
adjacent cells 602. Therefore, in one example embodiment of a
method for generating synchronization codes exhibiting the
properties described above, the process begins by selecting a
"perfect sequence" to be used as the basis for the codes. A perfect
sequence is one that when correlated with itself produces a number
equal to the number of bits in the sequence. For example: Perfect
.times. .times. sequence .times. .times. 1 .times. : .times.
.times. 1 .times. .times. 1 - 1 .times. .times. 1 ##EQU2## .times.
1 .times. .times. 1 - 1 .times. .times. 1 _ ##EQU2.2## .times. 1
.times. .times. 1 .times. .times. 1 .times. .times. 1 .times. = 4
##EQU2.3##
[0079] But each time a perfect sequence is cyclically shifted by
one bit, the new sequence is orthogonal with the original sequence.
Thus, for example, if perfect sequence 1 is cyclically shifted by
one bit and then correlated with the original, the correlation
produces a "0" as in the following example; Perfect .times. .times.
sequence .times. .times. 1 .times. : .times. .times. 1 .times.
.times. 1 - 1 .times. .times. 1 ##EQU3## .times. 1 .times. .times.
1 - 1 .times. .times. 1 _ ##EQU3.2## .times. 1 .times. .times. 1 -
1 - 1 .times. = 0 ##EQU3.3##
[0080] If the perfect sequence 1 is again cyclically shifted by one
bit, and again correlated with the original, then it will produce a
"0". In general, you can cyclically shift a perfect sequence by any
number of bits up to its length and correlate the shifted sequence
with the original to obtain a "0".
[0081] Once a perfect sequence of the correct length is selected,
the first synchronization code is preferably generated in one
embodiment by repeating the sequence 4 times. Thus, if perfect
sequence 1 is being used, then a first synchronization code y would
be the following: y=1 1-11 11-11 11-11 11-11.
[0082] Or in generic form:
y=x(0)x(1)x(2)x(3)x(0)x(1)x(2)x(3)x(0)x(1)x(2)x(3)x(0)x(1)x(2)x(3).
[0083] For a sequence of length L: y=x(0)x(1) . . . x(L)x(0)x(1) .
. . x(L)x(0)x(1) . . . x(L)x(0)x(1) . . . x(L)
[0084] Repeating the perfect sequence allows correlator 800 a
better opportunity to detect the synchronization code and allows
generation of other uncorrelated frequencies as well. Repeating has
the effect of sampling in the frequency domain. This effect is
illustrated by the graphs in FIG. 10. Thus, in TRACE 1, which
corresponds to synchronization code y, a sample 1002 is generated
every fourth sample bin 1000. Each sample bin is separated by
1/(4L.times.T), where T is the symbol duration. Thus, in the above
example, where L=4, each sample bin is separated by 1/(16.times.T)
in the frequency domain. TRACES 2-4 illustrate the next three
synchronization codes. As can be seen, the samples for each
subsequent synchronization code are shifted by one sample bin
relative to the samples for the previous sequence. Therefore, none
of the sequences interfere with each other.
[0085] To generate the subsequent sequences, corresponding to
TRACES 2-4, sequence y must be shifted in frequency. This can be
accomplished using the following equation:
z.sup.r(m)=y(m)*exp(j*2*.pi.*r*m/(n*L)), (5)
[0086] for r=1 to L (# of sequences) and m=0 to 4*L-1 (time);
and
[0087] where: z.sup.r(m)=each subsequent sequence; [0088] y(m)=the
first sequence; and [0089] n=the number of times the sequence is
repeated.
[0090] It will be understood that multiplying by an
exp(j2.pi.(r*m/N)) factor, where N is equal to the number of times
the sequence is repeated n multiplied by the length of the
underlying perfect sequence L, in the time domain results in a
shift in the frequency domain. Equation (5) results in the desired
shift as illustrated in FIG. 10 for each of synchronization codes
2-4, relative to synchronization code 1. The final step in
generating each synchronization code is to append the copies of the
last M samples, where M is the length of the multipath, to the
front of each code. This is done to make the convolution with the
multipath cyclic and to allow easier detection of the
multipath.
[0091] It should be noted that synchronization codes can be
generated from more than one perfect sequence using the same
methodology. For example, a perfect sequence can be generated and
repeated four times and then a second perfect sequence can be
generated and repeated four times to get a n factor equal to eight.
The resulting sequence can then be shifted as described above to
create the synchronization codes.
[0092] b. Signal Measurements Using Synchronization Codes
[0093] Therefore, when a communication device is at the edge of a
cell, it will receive signals from multiple base stations and,
therefore, will be decoding several synchronization codes at the
same time. This can be illustrated with the help of FIG. 11, which
illustrates another example embodiment of a wireless communication
system 1100 comprising communication cells 1102, 1104, and 1106 as
well as communication device 1108, which is in communication with
base station 1110 of cell 1102 but also receiving communication
from base stations 1112 and 1114 of cells 1104 and 1106,
respectively.
[0094] If communications from base station 1110 comprise
synchronization code SYNC1 and communications from base station
1112 and 1114 comprise SYNC2 and SYNC3 respectively, then device
1108 will effectively receive the sum of these three
synchronization codes. This is because, as explained above, base
stations 1110, 1112, and 1114 are configured to transmit at the
same time. Also, the synchronization codes arrive at device 1108 at
almost the same time because they are generated in accordance with
the description above.
[0095] Again as described above, the synchronization codes SYNC1,
SYNC2, and SYNC3 exhibit ideal cross correlation. Therefore, when
device 1108 correlates the sum x of codes SYNC1, SYNC2, and SYNC3,
the latter two will not interfere with proper detection of SYNC1 by
device 1108. Importantly, the sum x can also be used to determine
important signal characteristics, because the sum x is equal to the
sum of the synchronization code signal in accordance with the
following equation: x=SYNC1+SYNC2+SYNC3. (6)
[0096] Therefore, when SYNC1 is removed, the sum of SYNC2 and SYNC3
is left, as shown in the following: x-SYNC1=SYNC2+SYNC3. (7)
[0097] The energy computed from the sum (SYNC2+SYNC3) is equal to
the noise or interference seen by device 1108. Since the purpose of
correlating the synchronization code in device 1106 is to extract
the energy in SYNC 1, device 1108 also has the energy in the signal
from base station 1110, i.e., the energy represented by SYNC1.
Therefore, device 1106 can use the energy of SYNC1 and of
(SYNC2+SYNC3) to perform a signal-to-interference measurement for
the communication channel over which it is communicating with base
station 1110. The result of the measurement is preferably a
signal-to-interference ratio (SIR). The SIR measurement can then be
communicated back to base station 1110 for purposes that will be
discussed below.
[0098] The ideal cross correlation of the synchronization codes,
also allows device 1108 to perform extremely accurate
determinations of the Channel Impulse Response (CIR), or channel
estimation, from the correlation produced by correlator 800. This
allows for highly accurate equalization using low cost, low
complexity equalizers, thus overcoming a significant draw back of
conventional systems.
4. Sub-Channel Assignments
[0099] As mentioned, the SIR as determined by device 1108 can be
communicated back to base station 1110 for use in the assignment of
channels 502. In one embodiment, due to the fact that each
sub-channel 502 is processed independently, the SIR for each
sub-channel 502 can be measured and communicated back to base
station 1110. In such an embodiment, therefore, sub-channels 502
can be divided into groups and a SIR measurement for each group can
be sent to base station 1110. This is illustrated in FIG. 12A,
which shows a wideband communication channel 1200 segmented into
sub-channels fo to f.sub.15. Sub-channels fo to f.sub.15 are then
grouped into 8 groups G1 to G8. Thus, in one embodiment, device
1108 and base station 1110 communicate over a channel such as
channel 1200.
[0100] Sub-channels in the same group are preferably separated by
as many sub-channels as possible to ensure diversity. In FIG. 12A
for example, sub-channels within the same group are 7 sub-channels
apart, e.g., group G1 comprises f.sub.0 and f.sub.8. Device 1102
reports a SIR measurement for each of the groups G1 to G8. These
SIR measurements are preferably compared with a threshold value to
determine which sub-channels groups are useable by device 1108.
This comparison can occur in device 1108 or base station 1110. If
it occurs in device 1108, then device 1108 can simply report to
base station 1110 which sub-channels groups are useable by device
1108.
[0101] SIR reporting will be simultaneously occurring for a
plurality of devices within cell 1102. Thus, FIG. 12B illustrates
the situation where two communication devices corresponding to User
1 and User 2 report SIR levels above the threshold for groups G1,
G3, G5, and G7. Base station 1110 preferably then assigns
sub-channel groups to User 1 and User 2 based on the SIR reporting
as illustrated in FIG. 12B. When assigning the "good" sub-channel
groups to User 1 and User 2, base station 1110 also preferably
assigns them based on the principles of frequency diversity. In
FIG. 12B, therefore, User 1 and User 2 are alternately assigned
every other "good" sub-channel.
[0102] The assignment of sub-channels in the frequency domain is
equivalent to the assignment of time slots in the time domain.
Therefore, as illustrated in FIG. 13, two users, User 1 and User 2,
receive packet 1302 transmitted over communication channel 1200.
FIG. 13 also illustrated the sub-channel assignment of FIG. 12B.
While FIGS. 12 and 13 illustrate sub-channel/time slot assignment
based on SIR for two users, the principles illustrated can be
extended for any number of users. Thus, a packet within cell 1102
can be received by 3 or more users. Although, as the number of
available sub-channels is reduced due to high SIR, so is the
available bandwidth. In other words, as available channels are
reduced, the number of users that can gain access to communication
channel 1200 is also reduced.
[0103] Poor SIR can be caused for a variety of reasons, but
frequently it results from a device at the edge of a cell receiving
communication signals from adjacent cells. Because each cell is
using the same bandwidth B, the adjacent cell signals will
eventually raise the noise level and degrade SIR for certain
sub-channels. In certain embodiments, therefore, sub-channel
assignment can be coordinated between cells, such as cells 1102,
1104, and 1106 in FIG. 10, in order to prevent interference from
adjacent cells.
[0104] Thus, if communication device 1108 is near the edge of cell
1102, and device 1118 is near the edge of cell 1106, then the two
can interfere with each other. As a result, the SIR measurements
that device 1108 and 1118 report back to base stations 1110 and
1114, respectively, will indicate that the interference level is
too high. Base station 1110 can then be configured to assign only
the odd groups, i.e., G1, G3, G5, etc., to device 1108, while base
station 1114 can be configured to assign the even groups to device
1118. The two devices 1108 and 1118 will then not interfere with
each other due to the coordinated assignment of sub-channel
groups.
[0105] Assigning the sub-channels in this manner reduces the
overall bandwidth available to devices 1108 and 1118, respectively.
In this case the bandwidth is reduced by a factor of two. But it
should be remembered that devices operating closer to each base
station 1110 and 1114, respectively, will still be able to use all
channels if needed. Thus, it is only devices, such as device 1108,
that are near the edge of a cell that will have the available
bandwidth reduced. Contrast this with a CDMA system, for example,
in which the bandwidth for all users is reduced, due to the
spreading techniques used in such systems, by approximately a
factor of 10 at all times. It can be seen, therefore, that the
systems and methods for wireless communication over a wide
bandwidth channel using a plurality of sub-channels not only
improves the quality of service, but can also increase the
available bandwidth significantly.
[0106] When there are three devices 1108, 1118, and 1116 near the
edge of their respective adjacent cells 1102, 1104, and 1106, the
sub-channels can be divided by three. Thus, device 1108, for
example, can be assigned groups G1, G4, etc., device 1118 can be
assigned groups G2, G5, etc., and device 1116 can be assigned
groups G3, G6, etc. In this case the available bandwidth for these
devices, i.e., devices near the edges of cells 1102, 1104, and
1106, is reduced by a factor of 3, but this is still better than a
CDMA system, for example.
[0107] The manner in which such a coordinated assignment of
sub-channels can work is illustrated by the flow chart in FIG. 14.
First in step 1402, a communication device, such as device 1108,
reports the SIR for all sub-channel groups G1 to G8. The SIRs
reported are then compared, in step 1404, to a threshold to
determine if the SIR is sufficiently low for each group.
Alternatively, device 1108 can make the determination and simply
report which groups are above or below the SIR threshold. If the
SIR levels are good for each group, then base station 1110 can make
each group available to device 1108, in step 1406. Periodically,
device 1108 preferably measures the SIR level and updates base
station 1110 in case the SIR as deteriorated. For example, device
1108 may move from near the center of cell 1102 toward the edge,
where interference from an adjacent cell may affect the SIR for
device 1108.
[0108] If the comparison in step 1404 reveals that the SIR levels
are not good, then base station 1110 can be preprogrammed to assign
either the odd groups or the even groups only to device 1108, which
it will do in step 1408. Device 1108 then reports the SIR
measurements for the odd or even groups it is assigned in step
1410, and they are again compared to a SIR threshold in step
1412.
[0109] It is assumed that the poor SIR level is due to the fact
that device 1108 is operating at the edge of cell 1102 and is
therefore being interfered with by a device such as device 1118.
But device 1108 will be interfering with device 1118 at the same
time. Therefore, the assignment of odd or even groups in step 1408
preferably corresponds with the assignment of the opposite groups
to device 1118, by base station 1114. Accordingly, when device 1108
reports the SIR measurements for whichever groups, odd or even, are
assigned to it, the comparison in step 1410 should reveal that the
SIR levels are now below the threshold level. Thus, base station
1110 makes the assigned groups available to device 1108 in step
1414. Again, device 1108 preferably periodically updates the SIR
measurements by returning to step 1402.
[0110] It is possible for the comparison of step 1410 to reveal
that the SIR levels are still above the threshold, which should
indicate that a third device, e.g., device 1116 is still
interfering with device 1108. In this case, base station 1110 can
be preprogrammed to assign every third group to device 1108 in step
1416. This should correspond with the corresponding assignments of
non-interfering channels to devices 1118 and 1116 by base stations
1114 and 1112, respectively. Thus, device 1108 should be able to
operate on the sub-channel groups assigned, i.e., G1, G4, etc.,
without undue interference. Again, device 1108 preferably
periodically updates the SIR measurements by returning to step
1402. Optionally, a third comparison step (not shown) can be
implemented after step 1416, to ensure that the groups assigned to
device 1408 posses an adequate SIR level for proper operation.
Moreover, if there are more adjacent cells, i.e., if it is possible
for devices in a 4.sup.th or even a 5.sup.th adjacent cell to
interfere with device 1108, then the process of FIG. 14 would
continue and the sub-channel groups would be divided even further
to ensure adequate SIR levels on the sub-channels assigned to
device 1108.
[0111] Even though the process of FIG. 14 reduces the bandwidth
available to devices at the edge of cells 1102, 1104, and 1106, the
SIR measurements can be used in such a manner as to increase the
data rate and therefore restore or even increase bandwidth. To
accomplish this, the transmitters and receivers used in base
stations 1102, 1104, and 1106, and in devices in communication
therewith, e.g., devices 1108, 1114, and 1116 respectively, must be
capable of dynamically changing the symbol mapping schemes used for
some or all of the sub-channel. For example, in some embodiments,
the symbol mapping scheme can be dynamically changed among BPSK,
QPSK, 8PSK, 16QAM, 32QAM, etc. As the symbol mapping scheme moves
higher, i.e., toward 32QAM, the SIR level required for proper
operation moves higher, i.e., less and less interference can be
withstood. Therefore, once the SIR levels are determined for each
group, the base station, e.g., base station 1110, can then
determine what symbol mapping scheme can be supported for each
sub-channel group and can change the modulation scheme accordingly.
Device 1108 must also change the symbol mapping scheme to
correspond to that of the base stations. The change can be effected
for all groups uniformly, or it can be effected for individual
groups. Moreover, the symbol mapping scheme can be changed on just
the forward link, just the reverse link, or both, depending on the
embodiment.
[0112] Thus, by maintaining the capability to dynamically assign
sub-channels and to dynamically change the symbol mapping scheme
used for assigned sub-channels, the systems and methods described
herein provide the ability to maintain higher available bandwidths
with higher performance levels than conventional systems. To fully
realize the benefits described, however, the systems and methods
described thus far must be capable of implementation in a cost
effect and convenient manner. Moreover, the implementation must
include reconfigurability so that a single device can move between
different types of communication systems and still maintain optimum
performance in accordance with the systems and methods described
herein. The following descriptions detail example high level
embodiments of hardware implementations configured to operate in
accordance with the systems and methods described herein in such a
manner as to provide the capability just described above.
5. Sample Transmitter Embodiments
[0113] FIG. 15 is logical block diagram illustrating an example
embodiment of a transmitter 1500 configured for wireless
communication in accordance with the systems and methods described
above. The transmitter could, for example be within a base station,
e.g., base station 606, or within a communication device, such as
device 604. Transmitter 1500 is provided to illustrate logical
components that can be included in a transmitter configured in
accordance with the systems and methods described herein. It is not
intended to limit the systems and methods for wireless
communication over a wide bandwidth channel using a plurality of
sub-channels to any particular transmitter configuration or any
particular wireless communication system.
[0114] With this in mind, it can be seen that transmitter 1500
comprises a serial-to-parallel converter 1504 configured to receive
a serial data stream 1502 comprising a data rate R.
Serial-to-parallel converter 1504 converts data stream 1502 into N
parallel data streams 1504, where N is the number of sub-channels
202. It should be noted that while the discussion that follows
assumes that a single serial data stream is used, more than one
serial data stream can also be used if required or desired. In any
case, the data rate of each parallel data stream 1504 is then R/N.
Each data stream 1504 is then sent to a scrambler, encoder, and
interleaver block 1506. Scrambling, encoding, and interleaving are
common techniques implemented in many wireless communication
transmitters and help to provide robust, secure communication.
Examples of these techniques will be briefly explained for
illustrative purposes.
[0115] Scrambling breaks up the data to be transmitted in an effort
to smooth out the spectral density of the transmitted data. For
example, if the data comprises a long string of "1"s, there will be
a spike in the spectral density. This spike can cause greater
interference within the wireless communication system. By breaking
up the data, the spectral density can be smoothed out to avoid any
such peaks. Often, scrambling is achieved by XORing the data with a
random sequence.
[0116] Encoding, or coding, the parallel bit streams 1504 can, for
example, provide Forward Error Correction (FEC). The purpose of FEC
is to improve the capacity of a communication channel by adding
some carefully designed redundant information to the data being
transmitted through the channel. The process of adding this
redundant information is known as channel coding. Convolutional
coding and block coding are the two major forms of channel coding.
Convolutional codes operate on serial data, one or a few bits at a
time. Block codes operate on relatively large (typically, up to a
couple of hundred bytes) message blocks. There are a variety of
useful convolutional and block codes, and a variety of algorithms
for decoding the received coded information sequences to recover
the original data. For example, convolutional encoding or turbo
coding with Viterbi decoding is a FEC technique that is
particularly suited to a channel in which the transmitted signal is
corrupted mainly by additive white gaussian noise (AWGN) or even a
channel that simply experiences fading.
[0117] Convolutional codes are usually described using two
parameters: the code rate and the constraint length. The code rate,
k/n, is expressed as a ratio of the number of bits into the
convolutional encoder (k) to the number of channel symbols (n)
output by the convolutional encoder in a given encoder cycle. A
common code rate is 1/2, which means that 2 symbols are produced
for every 1-bit input into the coder. The constraint length
parameter, K, denotes the "length" of the convolutional encoder,
i.e. how many k-bit stages are available to feed the combinatorial
logic that produces the output symbols. Closely related to K is the
parameter m, which indicates how many encoder cycles an input bit
is retained and used for encoding after it first appears at the
input to the convolutional encoder. The m parameter can be thought
of as the memory length of the encoder.
[0118] Interleaving is used to reduce the effects of fading.
Interleaving mixes up the order of the data so that if a fade
interferes with a portion of the transmitted signal, the overall
message will not be affected. This is because once the message is
de-interleaved and decoded in the receiver, the data lost will
comprise non-contiguous portions of the overall message. In other
words, the fade will interfere with a contiguous portion of the
interleaved message, but when the message is de-interleaved, the
interfered with portion is spread throughout the overall message.
Using techniques such as FEC, the missing information can then be
filled in, or the impact of the lost data may just be
negligible.
[0119] After blocks 1506, each parallel data stream 1504 is sent to
symbol mappers 1508. Symbol mappers 1508 apply the requisite symbol
mapping, e.g., BPSK, QPSK, etc., to each parallel data stream 1504.
Symbol mappers 1508 are preferably programmable so that the
modulation applied to parallel data streams can be changed, for
example, in response to the SIR reported for each sub-chaimel 202.
It is also preferable, that each symbol mapper 1508 be separately
programmable so that the optimum symbol mapping scheme for each
sub-channel can be selected and applied to each parallel data
stream 1504.
[0120] After symbol mappers 1508, parallel data streams 1504 are
sent to modulators 1510. Important aspects and features of example
embodiments of modulators 1510 are described below. After
modulators 1510, parallel data streams 1504 are sent to summer
1512, which is configured to sum the parallel data streams and
thereby generate a single serial data stream 1518 comprising each
of the individually processed parallel data streams 1504. Serial
data stream 1518 is then sent to radio module 1512, where it is
modulated with an RF carrier, amplified, and transmitted via
antenna 1516 according to known techniques.
[0121] The transmitted signal occupies the entire bandwidth B of
communication channel 100 and comprises each of the discrete
parallel data streams 1504 encoded onto their respective
sub-channels 102 within bandwidth B. Encoding parallel data streams
1504 onto the appropriate sub-channels 102 requires that each
parallel data stream 1504 be shifted in frequency by an appropriate
offset. This is achieved in modulator 1510.
[0122] FIG. 16 is a logical block diagram of an example embodiment
of a modulator 1600 in accordance with the systems and methods
described herein. Importantly, modulator 1600 takes parallel data
streams 1602 performs Time Division Modulation (TDM) or Frequency
Division Modulation (FDM) on each data stream 1602, filters them
using filters 1612, and then shifts each data stream in frequency
using frequency shifter 1614 so that they occupy the appropriate
sub-channel. Filters 1612 apply the required pulse shaping, i.e.,
they apply the roll-off factor described in section 1. The
frequency shifted parallel data streams 1602 are then summed and
transmitted. Modulator 1600 can also include rate controller 1604,
frequency encoder 1606, and interpolators 1610. All of the
components shown in FIG. 15 are described in more detail in the
following paragraphs and in conjunction with FIGS. 16-22.
[0123] FIG. 17 illustrates one example embodiment of a rate
controller 1700 in accordance with the systems and methods
described herein. Rate control 1700 is used to control the data
rate of each parallel data stream 1602. In rate controller 1700,
the data rate is halved by repeating data streams d(0) to d(7), for
example, producing streams a(0) to a(15) in which a(0) is the same
as a(8), a(1) is the same as a(9), etc. FIG. 17 also illustrates
that the effect of repeating the data streams in this manner is to
take the data streams that are encoded onto the first 8
sub-channels 1702, and duplicate them on the next 8 sub-channels
1702. As can be seen, 7 sub-channels separate sub-channels 1702
comprising the same, or duplicate, data streams. Thus, if fading
effects one sub-channel 1702, for example, the other sub-channels
1702 carrying the same data will likely not be effected, i.e.,
there is frequency diversity between the duplicate data streams. So
by sacrificing data rate, in this case half the data rate, more
robust transmission is achieved. Moreover, the robustness provided
by duplicating the data streams d(0) to d(7) can be further
enhanced by applying scrambling to the duplicated data streams via
scramblers 1708.
[0124] It should be noted that the data rate can be reduced by more
than half, e.g., by four or more. Alternatively, the data rate can
also be reduced by an amount other than half. For example if
information from n data stream is encoded onto m sub-channels,
where m>n. Thus, to decrease the rate by 2/3, information from
one data stream can be encoded on a first sub-channel, information
from a second data stream can be encoded on a second data channel,
and the sum or difference of the two data streams can be encoded on
a third channel. In which case, proper scaling will need to be
applied to the power in the third channel. Otherwise, for example,
the power in the third channel can be twice the power in the first
two.
[0125] Preferably, rate controller 1700 is programmable so that the
data rate can be changed responsive to certain operational factors.
For example, if the SIR reported for sub-channels 1702 is low, then
rate controller 1700 can be programmed to provide more robust
transmission via repetition to ensure that no data is lost due to
interference. Additionally, different types of wireless
communication system, e.g., indoor, outdoor, line-of-sight, may
require varying degrees of robustness. Thus, rate controller 1700
can be adjusted to provide the minimum required robustness for the
particular type of communication system. This type of
programmability not only ensures robust communication, it can also
be used to allow a single device to move between communication
systems and maintain superior performance.
[0126] FIG. 18 illustrates an alternative example embodiment of a
rate controller 1800 in accordance with the systems and methods
described. In rate controller 1800 the data rate is increased
instead of decreased. This is accomplished using serial-to-parallel
converters 1802 to convert each data streams d(0) to d(15), for
example, into two data streams. Delay circuits 1804 then delay one
of the two data streams generated by each serial-to-parallel
converter 1802 by 1/2 a symbol. Thus, data streams d(0) to d(15)
are transformed into data streams a(0) to a(31). The data streams
generated by a particular serial-to-parallel converter 1802 and
associate delay circuit 1804 must then be summed and encoded onto
the appropriate sub-channel. For example, data streams a(0) and
a(1) must be summed and encoded onto the first sub-channel.
Preferably, the data streams are summed subsequent to each data
stream being pulsed shaped by a filter 1612.
[0127] Thus, rate controller 1604 is preferably programmable so
that the data rate can be increased, as in rate controller 1800, or
decreased, as in rate controller 1700, as required by a particular
type of wireless communication system, or as required by the
communication channel conditions or sub-channel conditions. In the
event that the data rate is increased, filters 1612 are also
preferably programmable so that they can be configured to apply
pulse shapping to data streams a(0) to a(31), for example, and then
sum the appropriate streams to generate the appropriate number of
parallel data streams to send to frequency shifter 1614.
[0128] The advantage of increasing the data rate in the manner
illustrated in FIG. 18 is that higher symbol mapping rates can
essentially be achieved, without changing the symbol mapping used
in symbol mappers 1508. Once the data streams are summed, the
summed streams are shifted in frequency so that they reside in the
appropriate sub-channel. But because the number of bits per each
symbol has been doubled, the symbol mapping rate has been doubled.
Thus, for example, a 4QAM symbol mapping can be converted to a
16QAM symbol mapping, even if the SIR is too high for 16QAM symbol
mapping to otherwise be applied. In other words, programming rate
controller 1800 to increase the data rate in the manner illustrated
in FIG. 18 can increase the symbol mapping even when channel
conditions would otherwise not allow it, which in turn can allow a
communication device to maintain adequate or even superior
performance regardless of the type of communication system.
[0129] The draw back to increasing the data rate as illustrated in
FIG. 18 is that interference is increased, as is receiver
complexity. The former is due to the increased amount of data. The
latter is due to the fact that each symbol cannot be processed
independently because of the 1/2 symbol overlap. Thus, these
concerns must be balanced against the increase symbol mapping
ability when implementing a rate controller such as rate controller
1800.
[0130] FIG. 19 illustrates one example embodiment of a frequency
encoder 1900 in accordance with the systems and methods described
herein. Similar to rate encoding, frequency encoding is preferably
used to provide increased communication robustness. In frequency
encoder 1900 the sum or difference of multiple data streams are
encoded onto each sub-channel. This is accomplished using adders
1902 to sum data streams d(0) to d(7) with data streams d(8) to
d(15), respectively, while adders 1904 subtract data streams d(0)
to d(7) from data streams d(8) to d(15), respectively, as shown.
Thus, data streams a(0) to a(15) generated by adders 1902 and 1904
comprise information related to more than one data streams d(0) to
d(15). For example, a(0) comprises the sum of d(0) and d(8), i.e.,
d(0)+d(8), while a(8) comprises d(8)-d(0). Therefore, if either
a(0) or a(8) is not received due to fading, for example, then both
of data streams d(0) and d(8) can still be retrieved from data
stream a(8).
[0131] Essentially, the relationship between data stream d(0) to
d(15) and a(0) to a(15) is a matrix relationship. Thus, if the
receiver knows the correct matrix to apply, it can recover the sums
and differences of d(0) to d(15) from a(0) to a(15). Preferably,
frequency encoder 1900 is programmable, so that it can be enabled
and disabled in order to provided robustness when required.
Preferable, adders 1902 and 1904 are programmable also so that
different matrices can be applied to d(0) to d(15).
[0132] After frequency encoding, if it is included, data streams
1602 are sent to TDM/FDM blocks 1608. TDM/FDM blocks 1608 perform
TDM or FDM on the data streams as required by the particular
embodiment. FIG. 20 illustrates an example embodiment of a TDM/FDM
block 2000 configured to perform TDM on a data stream. TDM/FDM
block 2000 is provided to illustrate the logical components that
can be included in a TDM/FDM block configured to perform TDM on a
data stream. Depending on the actual implementation, some of the
logical components may or may not be included. TDM/FDM block 2000
comprises a sub-block repeater 2002, a sub-block scrambler 2004, a
sub-block terminator 2006, a sub-block repeater 2008, and a sync
inserter 2010.
[0133] Sub-block repeater 2002 is configured to receive a sub-block
of data, such as block 2012 comprising bits a(0) to a(3) for
example. Sub-block repeater is then configured to repeat block 2012
to provide repetition, which in turn leads to more robust
communication. Thus, sub-block repeater 2002 generates block 2014,
which comprises 2 blocks 2012. Sub-block scrambler 2004 is then
configured to receive block 2014 and to scramble it, thus
generating block 2016. One method of scrambling can be to invert
half of block 2014 as illustrated in block 2016. But other
scrambling methods can also be implemented depending on the
embodiment.
[0134] Sub-block terminator 2006 takes block 2016 generated by
sub-block scrambler 2004 and adds a termination block 2034 to the
front of block 2016 to form block 2018. Termination block 2034
ensures that each block can be processed independently in the
receiver. Without termination block 2034, some blocks may be
delayed due to multipath, for example, and they would therefore
overlap part of the next block of data. But by including
termination block 2034, the delayed block can be prevented from
overlapping any of the actual data in the next block.
[0135] Termination block 2034 can be a cyclic prefix termination
2036. A cyclic prefix termination 2036 simply repeats the last few
symbols of block 2018. Thus, for example, if cyclic prefix
termination 2036 is three symbols long, then it would simply repeat
the last three symbols of block 2018. Alternatively, termination
block 2034 can comprise a sequence of symbols that are known to
both the transmitter and receiver. The selection of what type of
block termination 2034 to use can impact what type of equalizer is
used in the receiver. Therefore, receiver complexity and choice of
equalizers must be considered when determining what type of
termination block 2034 to use in TDM/FDM block 2000.
[0136] After sub-block terminator 2006, TDM/FDM block 2000 can
include a sub-block repeater 2008 configured to perform a second
block repetition step in which block 2018 is repeated to form block
2020. In certain embodiments, sub-block repeater can be configured
to perform a second block scrambling step as well. After sub-block
repeater 2008, if included, TDM/FDM block 2000 comprises a sync
inserter 210 configured to periodically insert an appropriate
synchronization code 2032 after a predetermined number of blocks
2020 and/or to insert known symbols into each block. The purpose of
synchronization code 2032 is discussed in section 3.
[0137] FIG. 21, on the other hand, illustrates an example
embodiment of a TDM/FDM block 2100 configured for FDM, which
comprises sub-block repeater 2102, sub-block scrambler 2104, block
coder 2106, sub-block transformer 2108, sub-block terminator 2110,
and sync inserter 2112. As with TDM/FDM block 2000, sub-block
repeater 2102 repeats block 2114 and generates block 2116.
Sub-block scrambler then scrambles block 2116, generating block
2118. Sub-block coder 2106 takes block 2118 and codes it,
generating block 2120. Coding block correlates the data symbols
together and generates symbols b. This requires joint demodulation
in the receiver, which is more robust but also more complex.
Sub-block transformer 2108 then performs a transformation on block
2120, generating block 2122. Preferably, the transformation is an
IFFT of block 2120, which allows for more efficient equalizers to
be used in the receiver. Next, sub-block terminator 2110 terminates
block 2122, generating block 2124 and sync inserter 2112
periodically inserts a synchronization code 2126 after a certain
number of blocks 2124 and/or insert known symbols into each block.
Preferably, sub-block terminator 2110 only uses cyclic prefix
termination as described above. Again this allows for more
efficient receiver designs.
[0138] TDM/FDM block 2100 is provided to illustrate the logical
components that can be included in a TDM/FDM block configured to
perform FDM on a data stream. Depending on the actual
implementation, some of the logical components may or may not be
included. Moreover, TDM/FDM block 2000 and 2100 are preferably
programmable so that the appropriate logical components can be
included as required by a particular implementation. This allows a
device that incorporates one of blocks 2000 or 2100 to move between
different systems with different requirements. Further, it is
preferable that TDM/FDM block 1608 in FIG. 16 be programmable so
that it can be programmed to perform TDM, such as described in
conjunction with block 2000, or FDM, such as described in
conjunction with block 2100, as required by a particular
communication system.
[0139] After TDM/FDM blocks 1608, in FIG. 16, the parallel data
streams are preferably passed to interpolators 1610.
[0140] After Interpolators 1610, the parallel data streams are
passed to filters 1612, which apply the pulse shaping described in
conjunction with the roll-off factor of equation (2) in section 1.
Then the parallel data streams are sent to frequency shifter 1614,
which is configured to shift each parallel data stream by the
frequency offset associated with the sub-channel to which the
particular parallel data stream is associated.
[0141] FIG. 22 illustrates an example embodiment of a frequency
shifter 2200 in accordance with the systems and methods described
herein. As can be seen, frequency shifter 2200 comprises
multipliers 2202 configured to multiply each parallel data stream
by the appropriate exponential to achieve the required frequency
shift. Each exponential is of the form: exp(j2.pi.f.sub.cnT/rM),
where c is the corresponding sub-channel, e.g., c=0 to N-1, and n
is time. Preferably, frequency shifter 1614 in FIG. 16 is
programmable so that various channel/sub-channel configurations can
be accommodated for various different systems. Alternatively, an
IFFT block can replace shifter 1614 and filtering can be done after
the IFFT block. This type of implementation can be more efficient
depending on the implementation.
[0142] After the parallel data streams are shifted, they are
summed, e.g., in summer 1512 of FIG. 15. The summed data stream is
then transmitted using the entire bandwidth B of the communication
channel being used. But the transmitted data stream also comprises
each of the parallel data streams shifted in frequency such that
they occupy the appropriate sub-channel. Thus, each sub-channel may
be assigned to one user, or each sub-channel may carry a data
stream intended for different users. The assignment of sub-channels
is described in section 3b. Regardless of how the sub-channels are
assigned, however, each user will receive the entire bandwidth,
comprising all the sub-channels, but will only decode those
sub-channels assigned to the user.
6. Sample Receiver Embodiments
[0143] FIG. 23 illustrates an example embodiment of a receiver 2300
that can be configured in accordance with the present invention.
Receiver 2300 comprises an antenna 2302 configured to receive a
message transmitted by a transmitter, such as transmitter 1500.
Thus, antenna 2302 is configured to receive a wide band message
comprising the entire bandwidth B of a wide band channel that is
divided into sub-channels of bandwidth b. As described above, the
wide band message comprises a plurality of messages each encoded
onto each of a corresponding sub-channel. All of the sub-channels
may or may not be assigned to a device that includes receiver 2300;
Therefore, receiver 2300 may or may not be required to decode all
of the sub-channels.
[0144] After the message is received by antenna 2300, it is sent to
radio receiver 2304, which is configured to remove the carrier
associated with the wide band communication channel and extract a
baseband signal comprising the data stream transmitted by the
transmitter. The baseband signal is then sent to correlator 2306
and demodulator 2308. Correlator 2306 is configured to correlated
with a synchronization code inserted in the data stream as
described in section 3. It is also preferably configured to perform
SIR and multipath estimations as described in section 3(b).
Demodulator 2308 is configured to extract the parallel data streams
from each sub-channel assigned to the device comprising receiver
2300 and to generate a single data stream therefrom.
[0145] FIG. 24 illustrates an example embodiment of a demodulator
2400 in accordance with the systems and methods described herein.
Demodulator 2402 comprises a frequency shifter 2402, which is
configured to apply a frequency offset to the baseband data stream
so that parallel data streams comprising the baseband data stream
can be independently processed in receiver 2400. Thus, the output
of frequency shifter 2402 is a plurality of parallel data streams,
which are then preferably filtered by filters 2404. Filters 2404
apply a filter to each parallel data stream that corresponds to the
pulse shape applied in the transmitter, e.g., transmitter 1500.
Alternatively, an IFFT block can replace shifter 1614 and filtering
can be done after the IFFT block. This type of implementation can
be more efficient depending on the implementation.
[0146] Next, receiver 2400 preferably includes decimators 2406
configured to decimate the data rate of the parallel bit streams.
Sampling at higher rates helps to ensure accurate recreation of the
data. But the higher the data rate, the larger and more complex
equalizer 2408 becomes. Thus, the sampling rate, and therefore the
number of samples, can be reduced by decimators 2406 to an adequate
level that allows for a smaller and less costly equalizer 2408.
[0147] Equalizer 2408 is configured to reduce the effects of
multipath in receiver 2300. Its operation will be discussed more
fully below. After equalizer 2408, the parallel data streams are
sent to de-scrambler, decoder, and de-interleaver 2410, which
perform the opposite operations of scrambler, encoder, and
interleaver 1506 so as to reproduce the original data generated in
the transmitter. The parallel data streams are then sent to
parallel to serial converter 2412, which generates a single serial
data stream from the parallel data streams.
[0148] Equalizer 2408 uses the multipath estimates provided by
correlator 2306 to equalize the effects of multipath in receiver
2300. In one embodiment, equalizer 2408 comprises Single-In
Single-Out (SISO) equalizers operating on each parallel data stream
in demodulator 2400. In this case, each SISO equalizer comprising
equalizer 2408 receives a single input and generates a single
equalized output. Alternatively, each equalizer can be a
Multiple-In Multiple-Out (MIMO) or a Multiple-In Single-Out (MISO)
equalizer. Multiple inputs can be required for example, when a
frequency encoder or rate controller, such as frequency encoder
1900, is included in the transmitter. Because frequency encoder
1900 encodes information from more than one parallel data stream
onto each sub-channel, each equalizers comprising equalizer 2408
need to equalize more than one sub-channel. Thus, for example, if a
parallel data stream in demodulator 2400 comprises d(1)+d(8), then
equalizer 2408 will need to equalize both d(1) and d(8) together.
Equalizer 2408 can then generate a single output corresponding to
d(1) or d(8) (MISO) or it can generate both d(1) and d(8)
(MIMO).
[0149] Equalizer 2408 can also be a time domain equalizer (TDE) or
a frequency domain equalizer (FDE) depending on the embodiment.
Generally, equalizer 2408 is a TDE if the modulator in the
transmitter performs TDM on the parallel data streams, and a FDE if
the modulator performs FDM. But equalizer 2408 can be an FDE even
if TDM is used in the transmitter. Therefore, the preferred
equalizer type should be taken into consideration when deciding
what type of block termination to use in the transmitter. Because
of power requirements, it is often preferable to use FDM on the
forward link and TDM on the reverse link in a wireless
communication system.
[0150] As with transmitter 1500, the various components comprising
demodulator 2400 are preferably programmable, so that a single
device can operate in a plurality of different systems and still
maintain superior performance, which is a primary advantage of the
systems and methods described herein. Accordingly, the above
discussion provides systems and methods for implementing a channel
access protocol that allows the transmitter and receiver hardware
to be reprogrammed slightly depending on the communication
system.
[0151] Thus, when a device moves from one system to another, it
preferably reconfigures the hardware, i.e. transmitter and
receiver, as required and switches to a protocol stack
corresponding to the new system. An important part of reconfiguring
the receiver is reconfiguring, or programming, the equalizer
because multipath is a main problem for each type of system. The
multipath, however, varies depending on the type of system, which
previously has meant that a different equalizer is required for
different types of communication systems. The channel access
protocol described in the preceding sections, however, allows for
equalizers to be used that need only be reconfigured slightly for
operation in various systems.
[0152] a. Sample Equalizer Embodiment
[0153] FIG. 25 illustrates an example embodiment of a receiver 2500
illustrating one way to configure equalizers 2506 in accordance
with the systems and methods described herein. Before discussing
the configuration of receiver 2500, it should be noted that one way
to configure equalizers 2506 is to simply include one equalizer per
channel (for the systems and methods described herein, a channel is
the equivalent of a sub-channel as described above). A correlator,
such as correlator 2306 (FIG. 23), can then provide equalizers 2506
with an estimate of the number, amplitude, and phase of any
multipaths present, up to some maximum number. This is also known
as the Channel Impulse Response (CIR). The maximum number of
multipaths is determined based on design criteria for a particular
implementation. The more multipaths included in the CIR the more
path diversity the receiver has and the more robust communication
in the system will be. Path diversity is discussed a little more
fully below.
[0154] If there is one equalizer 2506 per channel, the CIR is
preferably provided directly to equalizers 2506 from the correlator
(not shown). If such a correlator configuration is used, then
equalizers 2506 can be run at a slow rate, but the overall
equalization process is relatively fast. For systems with a
relatively small number of channels, such a configuration is
therefore preferable. The problem, however, is that there is large
variances in the number of channels used in different types of
communication systems. For example, an outdoor system can have has
many as 256 channels. This would require 256 equalizers 2506, which
would make the receiver design too complex and costly. Thus, for
systems with a lot of channels, the configuration illustrated in
FIG. 25 is preferable. In receiver 2500, multiple channels share
each equalizer 2506. For example, each equalizer can be shared by 4
channels, e.g., Ch1-Ch4, Ch5-Ch8, etc., as illustrated in FIG. 25.
In which case, receiver 2500 preferably comprises a memory 2502
configured to store information arriving on each channel.
[0155] Memory 2502 is preferably divided into sub-sections 2504,
which are each configured to store information for a particular
subset of channels. Information for each channel in each subset is
then alternately sent to the appropriate equalizer 2506, which
equalizes the information based on the CIR provided for that
channel. In this case, each equalizer must run much faster than it
would if there was simply one equalizer per channel. For example,
equalizers 2506 would need to run 4 or more times as fast in order
to effectively equalize 4 channels as opposed to 1. In addition,
extra memory 2502 is required to buffer the channel information.
But overall, the complexity of receiver 2500 is reduced, because
there are fewer equalizers. This should also lower the overall cost
to implement receiver 2500.
[0156] Preferably, memory 2502 and the number of channels that are
sent to a particular equalizer is programmable. In this way,
receiver 2500 can be reconfigured for the most optimum operation
for a given system. Thus, if receiver 2500 were moved from an
outdoor system to an indoor system with fewer channels, then
receiver 2500 can preferably be reconfigured so that there are
fewer, even as few as 1, channel per equalizer. The rate at which
equalizers 2506 are run is also preferably programmable such that
equalizers 2506 can be run at the optimum rate for the number of
channels being equalized.
[0157] In addition, if each equalizer 2506 is equalizing multiple
channels, then the CIR for those multiple paths must alternately be
provided to each equalizer 2506. Preferably, therefore, a memory
(not shown) is also included to buffer the CIR information for each
channel. The appropriate CIR information is then sent to each
equalizer from the CIR memory (not shown) when the corresponding
channel information is being equalized. The CIR memory (not shown)
is also preferably programmable to ensure optimum operation
regardless of what type of system receiver 2500 is operating
in.
[0158] Returning to the issue of path diversity, the number of
paths used by equalizers 2506 must account for the delay spread
d.sub.s in the system. For example, if the system is an outdoor
system operating in the 5 Giga Hertz (GHz) range, the communication
channel can comprise a bandwidth of 125 Mega Hertz (MHz), e.g., the
channel can extend from 5.725 GHz to 5.85 GHz. If the channel is
divided into 512 sub-channels with a roll-off factor r of 0.125,
then each subchannel will have a bandwidth of approximately 215
kilohertz (KHz), which provides approximately a 4.6 microsecond
symbol duration. Since the worstcase delay spread d.sub.s is 20
microseconds, the number of paths used by equalizers 2504 can be
set to a maximum of 5. Thus, there would be a first path P1 at zero
microseconds, a second path P2 at 4.6 microseconds, a third path P3
at 9.2 microseconds, a fourth path P4 at 13.8 microseconds, and
fifth path P5 at 18.4 microseconds, which is close to the delay
spread d.sub.s. In another embodiment, a sixth path can be included
so as to completely cover the delay spread d.sub.s; however, 20
microseconds is the worst case. In fact, a delay spread d.sub.s of
3 microseconds is a more typical value. In most instances,
therefore, the delay spread d.sub.s will actually be shorter and an
extra path is not needed. Alternatively, fewer sub-channels can be
used, thus providing a larger symbol duration, instead of using an
extra path. But again, this would typically not be needed.
[0159] As explained above, equalizers 2506 are preferably
configurable so that they can be reconfigured for various
communication systems. Thus, for example, the number of paths used
must be sufficient regardless of the type of communication system.
But this is also dependent on the number of sub-channels used. If,
for example, receiver 2500 went from operating in the above
described outdoor system to an indoor system, where the delay
spread d.sub.s is on the order of 1 microsecond, then receiver 2500
can preferably be reconfigured for 32 sub-channels and 5 paths.
Assuming the same overall bandwidth of 125 MHz, the bandwidth of
each sub-channel is approximately 4 MHz and the symbol duration is
approximately 250 nanoseconds.
[0160] Therefore, there will be a first path P1 at zero
microseconds and subsequent paths P2 to P5 at 250 ns, 500 ns, 750
ns, and 1 microsecond, respectively. Thus, the delay spread ds
should be covered for the indoor environment. Again, the 1
microsecond delay spread d.sub.s is worst case so the 1 microsecond
delay spread d.sub.s provided in the above example will often be
more than is actually required. This is preferable, however, for
indoor systems, because it can allow operation to extend outside of
the inside environment, e.g., just outside the building in which
the inside environment operates. For campus style environments,
where a user is likely to be traveling between buildings, this can
be advantageous.
7. Sample Embodiment of a Wireless Communication device
[0161] FIG. 26 illustrates an example embodiment of a wireless
communication device in accordance with the systems and methods
described herein. Device 2600 is, for example, a portable
communication device configured for operation in a plurality of
indoor and outdoor communication systems. Thus, device 2600
comprises an antenna 2602 for transmitting and receiving wireless
communication signals over a wireless communication channel 2618.
Duplexor 2604, or switch, can be included so that transmitter 2606
and receiver 2608 can both use antenna 2602, while being isolated
from each other. Duplexors, or switches used for this purpose, are
well known and will not be explained herein.
[0162] Transmitter 2606 is a configurable transmitter configured to
implement the channel access protocol described above. Thus,
transmitter 2606 is capable of transmitting and encoding a wideband
communication signal comprising a plurality of sub-channels.
Moreover, transmitter 2606 is configured such that the various
sub-components that comprise transmitter 2606 can be reconfigured,
or programmed, as described in section 5. Similarly, receiver 2608
is configured to implement the channel access protocol described
above and is, therefore, also configured such that the various
sub-components comprising receiver 2608 can be reconfigured, or
reprogrammed, as described in section 6.
[0163] Transmitter 2606 and receiver 2608 are interfaced with
processor 2610, which can comprise various processing, controller,
and/or Digital Signal Processing (DSP) circuits. Processor 2610
controls the operation of device 2600 including encoding signals to
be transmitted by transmitter 2606 and decoding signals received by
receiver 2608. Device 2610 can also include memory 2612, which can
be configured to store operating instructions, e.g.,
firmware/software, used by processor 2610 to control the operation
of device 2600.
[0164] Processor 2610 is also preferably configured to reprogram
transmitter 2606 and receiver 2608 via control interfaces 2614 and
2616, respectively, as required by the wireless communication
system in which device 2600 is operating. Thus, for example, device
2600 can be configured to periodically ascertain the availability
is a preferred communication system. If the system is detected,
then processor 2610 can be configured to load the corresponding
operating instruction from memory 2612 and reconfigure transmitter
2606 and receiver 2608 for operation in the preferred system.
[0165] For example, it may preferable for device 2600 to switch to
an indoor wireless LAN if it is available. So device 2600 may be
operating in a wireless WAN where no wireless LAN is available,
while periodically searching for the availability of an appropriate
wireless LAN. Once the wireless LAN is detected, processor 2610
will load the operating instructions, e.g., the appropriate
protocol stack, for the wireless LAN environment and will reprogram
transmitter 2606 and receiver 2608 accordingly. In this manner,
device 2600 can move from one type of communication system to
another, while maintaining superior performance.
[0166] It should be noted that a base station configured in
accordance with the systems and methods herein will operate in a
similar manner as device 2600; however, because the base station
does not move from one type of system to another, there is
generally no need to configure processor 2610 to reconfigure
transmitter 2606 and receiver 2608 for operation in accordance with
the operating instruction for a different type of system. But
processor 2610 can still be configured to reconfigure, or reprogram
the sub-components of transmitter 2606 and/or receiver 2608 as
required by the operating conditions within the system as reported
by communication devices in communication with the base station.
Moreover, such a base station can be configured in accordance with
the systems and methods described herein to implement more than one
mode of operation. In which case, controller 2610 can be configured
to reprogram transmitter 2606 and receiver 2608 to implement the
appropriate mode of operation.
[0167] While embodiments and implementations of the invention have
been shown and described, it should be apparent that many more
embodiments and implementations are within the scope of the
invention. Accordingly, the invention is not to be restricted,
except in light of the claims and their equivalents.
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