U.S. patent application number 11/618922 was filed with the patent office on 2007-05-17 for integrated crosspoint switch with band translation.
This patent application is currently assigned to RF Magic, Inc.. Invention is credited to Keith P. Bargroff, Bert L. Fransis, Dale Hancock.
Application Number | 20070111661 11/618922 |
Document ID | / |
Family ID | 32512584 |
Filed Date | 2007-05-17 |
United States Patent
Application |
20070111661 |
Kind Code |
A1 |
Bargroff; Keith P. ; et
al. |
May 17, 2007 |
Integrated Crosspoint Switch with Band Translation
Abstract
Multiple input signal sources in predetermined frequency bands
are each applied to block frequency converters. Each block
frequency converter frequency converts an input signal to one of a
plurality of predetermined frequency bands. A crosspoint switch is
configured to route the frequency converted input signals at one of
the plurality of predetermined frequency bands to any one of a
plurality of available band translation devices. Each of the band
translation devices is configured to frequency convert the signal
from a first predetermined frequency band to a second predetermined
frequency band. Output signals from one or more band translation
devices can be combined into a composite signal.
Inventors: |
Bargroff; Keith P.; (San
Diego, CA) ; Fransis; Bert L.; (San Diego, CA)
; Hancock; Dale; (San Diego, CA) |
Correspondence
Address: |
CLIFFORD B. PERRY
132 N. EL CAMINO REAL, #347
ENCINITAS
CA
92024-2801
US
|
Assignee: |
RF Magic, Inc.
San Diego
CA
|
Family ID: |
32512584 |
Appl. No.: |
11/618922 |
Filed: |
January 1, 2007 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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10735521 |
Dec 11, 2003 |
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11618922 |
Jan 1, 2007 |
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60433061 |
Dec 11, 2002 |
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60433063 |
Dec 11, 2002 |
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60433066 |
Dec 11, 2002 |
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60433067 |
Dec 11, 2002 |
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Current U.S.
Class: |
455/13.3 ;
455/131; 455/140 |
Current CPC
Class: |
H04N 7/20 20130101; H04Q
2213/1319 20130101; H03F 2200/171 20130101; H04Q 2213/13322
20130101; H03G 3/3036 20130101; H04Q 2213/13034 20130101; H03D
2200/0043 20130101; H03F 2200/294 20130101; H03D 7/1425 20130101;
H03D 2200/0025 20130101; H03D 7/00 20130101; H04Q 3/521 20130101;
H03D 7/1433 20130101; H03D 7/1458 20130101; H04Q 2213/1304
20130101; H04H 40/90 20130101; H04Q 2213/1302 20130101; H03F
2200/451 20130101; H04B 1/126 20130101; H03F 3/19 20130101; H04N
7/102 20130101 |
Class at
Publication: |
455/013.3 ;
455/131; 455/140 |
International
Class: |
H04B 7/185 20060101
H04B007/185; H03D 7/16 20060101 H03D007/16 |
Claims
1-50. (canceled)
51. An apparatus for distributing one or more channels included
within a plurality of N signals to one or more output devices, the
apparatus including a crosspoint switch and a plurality of
frequency translation devices, the crosspoint switch including a
plurality of N crosspoint switch inputs and a plurality of
crosspoint switch outputs, each of the N crosspoint switch inputs
coupled to receive one of the N signals, the crosspoint switch
operable to switchably couple any of the plurality of N crosspoint
switch inputs to any one or more of the plurality of crosspoint
switch outputs, and wherein each of the plurality of frequency
translation devices includes an input coupled to a respective one
of the crosspoint switch outputs and an output configured to couple
to one or more output devices, each of the plurality of frequency
translation devices operable to frequency translate one or more of
the channels as supplied to respective one or more channels,
wherein the outputs of at least two of the plurality of frequency
translation devices are coupled together, the apparatus comprising:
at least a portion of the signal path within the crosspoint switch
or one or more of the plurality of frequency translation devices
configured as a differential signal path operable to support the
propagation of a differential mode signal.
52. The apparatus of claim 51, wherein the one or more channels
included within one or more of the N signals comprises at least one
multiplexed channel, the multiplexed channel operating at a
predetermined carrier frequency and comprising the content of two
or more channels.
53. The apparatus of claim 52, wherein the content of the two or
more channels comprises digital content, and wherein the
multiplexed channel comprises a multiplexed digital channel.
54. The apparatus of claim 51, wherein at least one of the N
signals comprises a plurality of frequency bands.
55. The apparatus of claim 51, wherein two or more frequency
translation devices are coupled to different variable local
oscillator sources.
56. The apparatus of claim 51, wherein the output of each one of
the frequency translation devices is configured to couple to a
single output device.
57. The apparatus of claim 51, wherein the coupled output of the
two or more frequency translation devices is configured to couple
to a single output device.
58. The apparatus of claim 51, further comprising a signal combiner
having (i) a plurality of inputs coupled to respective plurality of
frequency translation device outputs, and (ii) an output coupled to
one or more output devices, said output comprising the coupled
output of the two or more frequency translation devices.
59. The apparatus of claim 58, wherein the output of the signal
combiner is coupled to a first subset of the one or more output
devices, the apparatus further comprising a second signal combiner
having a plurality of inputs coupled to respective plurality of
frequency translation device outputs, and an output coupled to a
second subset of one or more output devices.
60. The apparatus of claim 58, further comprising a respective
plurality of filters, each respective filter coupled between a
frequency translation device output and a signal combiner
input.
61. The apparatus of claim 51, further comprising a plurality of
variable gain amplifiers coupled to the crosspoint switch, each of
the plurality of variable gain amplifiers operable to apply gain or
attenuation to a signal input thereto.
62. The apparatus of claim 51, wherein said crosspoint switch
comprises a first crosspoint switch, said plurality of frequency
translation devices comprises a plurality of first frequency
translation devices, and said output devices comprise first output
devices, the apparatus further including a plurality of second
frequency translation devices, each having an input coupled to a
respective one of the second crosspoint switch outputs and an
output configured to couple to one or more second output devices,
each of the one or more second frequency translation devices
operable to frequency translate one or more of the channels as
supplied to respective one or more channels, the apparatus further
comprising: a second crosspoint switch having a plurality of N
crosspoint switch inputs and a plurality of crosspoint switch
outputs, each of the N second crosspoint switch inputs coupled to
either: (i) a respective one of the first crosspoint switch inputs,
or (ii) a respective one of the first crosspoint switch outputs,
each of the second crosspoint switch outputs configured to couple
to one or more second output devices, the second crosspoint switch
operable to switchably couple any of the plurality of N second
crosspoint switch inputs to any one or more of the plurality of
second crosspoint switch outputs; wherein the outputs of at least
two of the plurality of second frequency translation devices are
coupled together, and at least a portion of the second crosspoint
switch or at least a portion of one or more of the plurality of
second frequency translation devices comprises a differential
signal path.
63. The apparatus of claim 62, wherein the output of at least one
of the first frequency translation devices is coupled to at least
one output of the second frequency translation devices.
64. In a signal distribution system including (i) a crosspoint
switch having N inputs operable to receive a respective plurality
of N signals and a plurality of crosspoint switch outputs, (ii) a
respective plurality of frequency translation devices coupled to
the crosspoint switch outputs, and (iii) one or more output devices
coupled to at least one of the plurality of frequency translation
devices, the crosspoint switch being selectively switchable, such
that any of the plurality of N received signals is coupled, via one
or more signal paths, to any one or more of the crosspoint switch
outputs, wherein one or more crosspoint switch output signals are
supplied to respective frequency translation devices, each of the
one or more crosspoint switch output signals including one or more
channels, and further wherein one or more of the plurality of the
frequency translation devices includes a signal path, each
frequency translation device operable to frequency translate one or
more of the channels as supplied to respective one or more
channels, the outputs of at least two of the plurality of frequency
translation devices being coupled together; and wherein the
frequency-translated channels are output to one or more output
devices operable to render the one or more channels supplied
thereto, a method for distributing one or more channels included
within any one of the plurality of N received signals to the one or
more output devices, the method comprising: propagating the one or
more channels from the respective crosspoint switch input to at
least one frequency translation device output along one or more of
said signal paths, wherein at least a portion of said one or more
signal paths comprises a differential signal path operable to
support the propagation of a differential mode signal.
65. The method of claim 64, wherein said crosspoint switch
comprises the only crosspoint switch operable to distribute, to the
one or more coupled output devices, one or more channels included
within any of the plurality of N received signals.
66. The method of claim 64, wherein controlling comprises:
downconverting at least one of the channels from the first
frequency to an intermediate frequency; and upconverting the at
least one of the channels from the intermediate frequency to the
second frequency.
67. The method of claim 66, wherein downconverting comprises
downconverting the at least one channel to baseband.
68. The method of claim 64, wherein frequency translating comprises
either (i) downconverting at least one of the channels from a first
frequency to the second frequency, or (ii) up converting at least
one of the channels from a first frequency to a second
frequency.
69. The method of claim 64, further comprising variably adjusting a
power level of one or more of the N signals.
70. The method of claim 64, further comprising: receiving the
plurality of the N signals into a second crosspoint switch, the
second crosspoint switch input having a respective plurality of N
inputs; selectively switching the second crosspoint switch, whereby
any one of the plurality of N received signals is coupled to any
one or more of the second crosspoint switch outputs; supplying one
or more second crosspoint switch output signals to respective
second frequency translation devices, each of the one or more
second crosspoint switch output signals including one or more
channels, wherein the outputs of at least two of the plurality of
second frequency translation devices being coupled together;
controlling one or more of the plurality of the second frequency
translation devices to frequency translate one or more of the
channels as supplied to respective one or more channels; and
outputting the frequency-translated channels to one or more second
output devices operable to render the one or more channels supplied
thereto, wherein at least a portion of the second crosspoint switch
or at least a portion of one or more of the plurality of second
frequency translation devices comprises a differential signal
path.
71. An apparatus for distributing one or more channels included
within each of a plurality of N satellite signals to one or more
output devices, the apparatus comprising: a first LNB unit operable
to receive a plurality of the N satellite signals, the first LNB
unit comprising: a first LNB converter coupled to receive a
plurality of the N satellite signals and operable to produce a
plurality of first satellite IF signals; a first crosspoint switch
having a plurality of first crosspoint switch inputs coupled to
receive respective first satellite IF signals and a plurality of
first crosspoint switch outputs, the first crosspoint switch
operable to switchably couple any of the first crosspoint switch
inputs to any one or more of the first crosspoint switch outputs;
and respective plurality of first frequency translation devices,
each first frequency translation device having an input coupled to
a respective one of the first crosspoint switch outputs and an
output configured to couple to one or more output devices, each of
the first frequency translation devices operable to frequency
translate one or more of the channels as supplied to respective one
or more channels, wherein the outputs of the first frequency
translation devices are coupled together to provide a first LNB
unit output, wherein at least a portion of the first crosspoint
switch or at least a portion of one or more of the plurality of
first frequency translation devices comprises a differential signal
path; a second LNB unit operable to receive a plurality of the N
satellite signals, the second LNB unit comprising: a second LNB
converter coupled to receive a plurality of the N satellite signals
and operable to produce a plurality of second satellite IF signals;
a second crosspoint switch (140) having a plurality of second
crosspoint switch inputs coupled to receive respective second
satellite IF signals and a plurality of second crosspoint switch
outputs, the second crosspoint switch operable to switchably couple
any of the second crosspoint switch inputs to any one or more of
the second crosspoint switch outputs; and respective plurality of
second frequency translation devices, each second frequency
translation device having an input coupled to a respective one of
the second crosspoint switch outputs and an output configured to
couple to one or more output devices, each of the second frequency
translation devices operable to frequency translate one or more of
the channels as supplied to respective one or more channels,
wherein the outputs of the second frequency translation devices are
coupled together to provide a second LNB unit output; wherein at
least a portion of the second crosspoint switch or at least a
portion of one or more of the plurality of second frequency
translation devices comprises a differential signal path, and a
signal combiner having inputs coupled to receive the first and
second LNB unit outputs and an output coupled to one or more output
devices.
Description
RELATED APPLICATIONS
[0001] This application claims priority to, and hereby incorporates
by reference in their entirety, the following patent
applications:
[0002] U.S. Provisional Patent Application No. 60/433,066, filed on
Dec. 11, 2002, entitled INTEGRATED CROSSPOINT SWITCH WITH BAND
TRANSLATION;
[0003] U.S. Provisional Patent Application No. 60/433,061, filed on
Dec. 11, 2002, entitled IN-LINE CASCADABLE DEVICE IN SIGNAL
DISTRIBUTION SYSTEM WITH AGC FUNCTION;
[0004] U.S. Provisional Patent Application No. 60/43,067, filed on
Dec. 11, 2002, entitled N.times.M CROSSPOINT SWITCH WITH BAND
TRANSLATION;
[0005] U.S. Provisional Patent Application No. 60/433,063, filed on
Dec. 11, 2002, entitled MIXER WITH PASS-THROUGH MODE WITH CONSTANT
EVEN ORDER GENERATION.
[0006] This application is related to, and hereby incorporates by
reference in their entirety, the following patent applications:
[0007] U.S. patent application Ser. No. 10/xxx,xxx, filed
concurrently herewith, entitled IN-LINE CASCADABLE DEVICE IN SIGNAL
DISTRIBUTION SYSTEM WITH AGC FUNCTION;
[0008] U.S. patent application Ser. No. 10/xxx,xxx, filed
concurrently herewith, entitled N.times.M CROSSPOINT SWITCH WITH
BAND TRANSLATION;
[0009] U.S. patent application Ser. No. 10/xxx,xxx, filed
concurrently herewith, entitled MIXER CIRCUIT WITH BYPASS AND
MIXING MODES HAVING CONSTANT EVEN ORDER GENERATION AND METHOD OF
OPERATION.
BACKGROUND OF THE INVENTION
[0010] 1. Field of the Invention
[0011] The invention relates to electronic signal processing. More
particularly, the invention relates to Radio Frequency (RF) signal
distribution and frequency conversion.
[0012] 2. Description of the Related Art
[0013] Communication systems are designed to provide information
from a source to a destination using a communication channel.
Electronic communication channels can be implemented in a variety
of ways. An electronic communication system can implement a
communication channel as point to point channels, broadcast
channels, or a combination of the two. The communication channels
can be implemented using optical communication links or electronic
communication links. Optical communication links include, but are
not limited to, free space optical links and fiber optic links.
Electronic communication links similarly include, but are not
limited to, wireless links and wired links.
[0014] The amount of information that can be carried on a
particular link is limited by the usable bandwidth in the link.
Some communication links can be band limited due to a physical
characteristic, while other communication links can be band limited
due to artificial factors, which can include system design
limitations and regulatory restrictions. Regulatory restrictions
are particularly prevalent in wireless communication systems in
order to minimize interference that can result if the radio
spectrum was unregulated. A signal that occupies the same bandwidth
as another uncorrelated signal, whether occurring in a wired system
or a wireless system, appears as interference to the uncorrelated
signal. Similarly, the uncorrelated signal appears as interference
to the first signal. Wireless communication systems commonly
operate in regulated bands because of the difficulty in isolating a
signal from interfering signals in a wireless environment.
[0015] Common communication systems that are band limited by
regulations, such as those imposed by the Federal Communications
Commission (FCC) in the United States, or the International
Telecommunications Union (ITU), include broadcast radio and
television systems. Regulation and standardization of the frequency
bands used in broadcast radio and television enable consumer
electronics to standardize their frequency bands of operation.
[0016] Numerous techniques are known and available to more
efficiently utilize a particular frequency band. The more common
techniques employ some form of signal multiplexing. Common
multiplexing techniques include Time Division Multiplexing (TDM),
Frequency Division Multiplexing (FDM), Code Division Multiplexing
(CDM), Orthogonal Frequency Division Multiplexing (OFDM), and the
like.
[0017] In TDM systems, independent users are allocated time slots
that do not overlap. Thus, at any particular time, one user is
allowed to occupy the entire allocated bandwidth. Similarly, in FDM
systems, users are allocated portions of the entire allocated
frequency band and the portion of the frequency band allocated to
one user does not overlap those portions allocated to another user.
In CDM systems each user is allocated the entire frequency band for
the entire time, but each user communicates in the frequency band
using an orthogonal code. Because the user codes are orthogonal,
each user can recover their respective signal from the other
signals occupying the frequency band. However, in a CDM system,
each user appears as an interfering signal to every other user in
the frequency band. In order to minimize the interference
contributed by any one user to all other users in the frequency
band, the power transmitted by each user in a CDM system is tightly
controlled to the minimum power that is required to achieve a
particular signal quality. In an OFDM system, the frequency band is
divided into a number of distinct frequency bands. Each user can
communicate over the entire band, but the information from each
user is divided up into a number of parallel streams that are
broadcast as orthogonal waveforms in each of a predetermined number
of sub-bands. Of course, this discussion of multiplexing techniques
is not exhaustive and some communication systems can implement
other multiplexing techniques or a combination of more than one
multiplexing technique.
[0018] One consequence of regulated frequency bands and
communication channels within the regulated frequency bands is
compatibility. Communication systems operating in a first
communication band can desire to provide information to second
communication system operating in second communication band. One
manner of interfacing two otherwise compatible signals at different
frequency bands is frequency translation. A signal from a first
communication system at a first frequency can be translated to a
second frequency in order to make the signal available in the
second communication system. Communication systems can incorporate
a number of frequency translations in providing a signal from a
source to its ultimate destination.
[0019] Additionally, signals from a first communication system can
be frequency translated to be compatible with a second
communication system. One example of the use of frequency
translation is in cable television systems. Televisions are
typically produced to be compatible with a particular television
standard. In the United States, most televisions are compatible
with the National Television System Committee (NTSC) standard and
the frequency allocation for television channels regulated by the
FCC. However, cable television providers are, to some extent, able
to provide the same content on practically any channel.
Additionally, the number of signals to which a cable operator has
can be far in excess of the number of channels that a typical
television receiver can tune. Thus, at a head end or cable
television signal source, a cable operator can receive a desired
number of signals and frequency translate each signal to a desired
television channel.
[0020] The system design implemented by the cable operator
determines frequency translation and the majority of the
contributors to inter-channel interference such that these issues
are typically of no concern to the end user. However, with the
increased availability of content from different sources, receivers
that are local to the end user are incorporating the ability to
interface to multiple sources and combine those multiple sources
into a signal that is presented to the end user.
[0021] For example, a single household can have access to
over-the-air television broadcast channels, cable television
channels, satellite television channels, microwave television
channels, closed circuit television channels, and television
channels from other sources. These multiple channels from multiple
signal sources can easily provide more channels to which a typical
television may be tuned. Additionally, some of the channels from
one or more of the signal sources can appear on the same channel
such that direct signal combination can not be feasible. For
example, signals from a satellite transponder can be downconverted
to a particular block of channels that can coincide with channels
provided by the cable television source. The channels from the two
signal sources cannot be directly combined without channels from
the satellite television source interfering with the coincident
channels from the cable television source. Thus, typically, a
switch can be provided to allow an end user to choose one signal
source and eliminate interfering signals from all other
sources.
[0022] To further compound the problems associated with
distributing signals from multiple sources to an end user, signals
from a source such as a satellite television source, do not arrive
from a single source but rather from multiple sources. A satellite
television receiver is typically capable of simultaneously
receiving signals from multiple satellite transponders positioned
on one or more satellites. The signals received from the multiple
satellite transponders can exceed the channel tuning capability of
a typical television tuner. Thus, only a subset of all available
satellite television signals can be simultaneously made available
to a particular television receiver.
[0023] Although a source switch can be a viable solution for a
single receiver, the use of a source switch quickly becomes
unfeasible when multiple receivers, such as televisions, interface
to the same sources via a common distribution mechanism, such as a
shared coaxial cable. It is typically unfeasible to provide every
signal source to each television location and provide a source
switch at the television. However, it is desirable for each
television receiver to independently have the ability to control
the signal source. Additionally, where signals from a particular
source, such as satellite television, can be provided on the same
channels, it is desirable to have the ability to frequency
translate some of the signals such that they are provided at
different channels.
SUMMARY OF THE INVENTION
[0024] Signal distribution systems, apparatus, and methods are
disclosed herein. In broad overview the system allows signals from
any one of a first number of input signals to be directed to any
one of a number of destinations. Additionally, each of the first
number of input signals can comprise signals in at least one
frequency band from a number of predetermined input frequency
bands. The system also allows signals from each of the number of
predetermined input frequency bands to be frequency translated to
one of a number of predetermined output frequency bands. The
predetermined input frequency bands can coincide with the
predetermined output frequency bands. Additionally, an input signal
in an input frequency band can be output to an output frequency
band that is substantially the same as the input frequency band.
Amplifiers within the system allow a signal from one signal source
to be provided to any number of independent frequency translation
blocks. Some or all of the system can be implemented in a single
integrated circuit.
[0025] Desired signals can be received from multiple distinct
signal sources. The communication link from the signal sources to
local receivers can be wired or wireless. Each of the signal
sources provides one or more source signals that can be frequency
translated to one of a number of predetermined input frequency
bands, such as, for example, by a low noise block converter.
Alternatively, one or more of the source signals can be provided in
one or more of the predetermined input frequency bands. The input
frequency bands are typically distinct and define a contiguous
composite frequency band. However, the input frequency bands can
overlap or define a disjoint composite frequency band. The input
and output frequency bands may carry data for one or more channels
originating on one or more transponders.
BRIEF DESCRIPTION OF THE DRAWINGS
[0026] The features, objects, and advantages of the invention will
become more apparent from the detailed description set forth below
when taken in conjunction with the drawings in which like reference
characters identify correspondingly throughout and wherein:
[0027] FIG. 1 is a functional block diagram of a satellite
communication system configured to provide signals from multiple
satellites to multiple user devices.
[0028] FIG. 2 is a functional block diagram of cascaded band
translation switches.
[0029] FIG. 3 is a functional block diagram of a band translation
switch having one band translation device configured to provide
pass through and one band translation device configured to provide
translation.
[0030] FIG. 4A illustrates a simplified switch diagram of a doubly
balanced mixer as known in the art.
[0031] FIG. 4B illustrates the doubly balanced mixer of FIG. 4A as
a Gilbert cell mixer circuit.
[0032] FIG. 5 illustrates a common oscillator, multiple mixer
system in which the mixer circuit could be employed.
[0033] FIG. 6A illustrates a simplified switch diagram of a mixer
circuit in accordance with one embodiment of the invention.
[0034] FIG. 6B illustrates a method of operating the mixer circuit
shown in FIG. 6A.
[0035] FIG. 6C illustrates the mixer circuit of FIG. 6A as a
modified Gilbert cell mixer circuit.
[0036] FIG. 7 illustrates a second embodiment of the mixer
circuit.
[0037] FIGS. 8A and 8B illustrates common oscillator, multiple
mixer systems utilizing the mixer circuits of FIGS. 3A and 4 in
accordance with embodiments of the invention.
[0038] FIGS. 9A-9C are functional block diagrams of cascadable
buffer amplifiers with switches.
[0039] FIG. 10 is a functional block diagram of an integrated
crosspoint switch with band translation.
[0040] FIGS. 11A-11D are functional block diagrams of switches.
[0041] FIG. 12 is a functional block diagram of an integrated
crosspoint switch with band translation.
[0042] FIG. 13 is a functional block diagram of an integrated
crosspoint switch with band translation.
[0043] FIG. 14 is a functional block diagram of an integrated
crosspoint switch with band translation.
[0044] FIG. 15 is a functional block diagram of an integrated band
translation switch interfacing with additional components to
provide two signal outputs.
[0045] FIG. 16 is a functional block diagram of a signal
distribution system configured to receive signals from satellites
and distribute them to multiple user devices.
[0046] FIGS. 17A-17D are functional block diagrams of AGC
amplifiers.
[0047] FIGS. 18A-18B are functional block diagrams of cascaded
amplifier configurations.
[0048] FIG. 19 is a functional block diagram of cascaded integrated
band translation switches.
[0049] FIG. 20 is a flowchart of a method of distributing signals
using cascadable AGC amplifiers.
[0050] FIG. 21 is a functional block diagram of a band translation
switch configured to frequency convert an input signal having a
predetermined bandwidth.
[0051] FIG. 22 is a functional block diagram of a band translation
switch configured to switch and frequency translate signals from
two sources to a single output.
[0052] FIG. 23 is a flowchart of a method of frequency translating
input frequency bands using an integrated band translation
switch.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
[0053] FIG. 1 is a functional block diagram of one embodiment of a
satellite based communication system, such as a satellite
television system 100. However, the invention is not limited to
application in a satellite based communication system, nor is the
invention limited to use in a television system. The invention is
applicable to any communication system where multiple signals in
one or more input frequency bands can be distributed as signals in
one or more output frequency bands to one or more receivers.
[0054] The satellite television system 100 includes one or more
satellites 110a-110c that are set at various different orbital
slots. Although three satellites 110a-110c are shown in FIG. 1, any
number of satellites can exist in a particular satellite television
system 100. The satellites can operate at different carrier
frequencies and polarizations. The different carrier frequencies
and polarizations that can be used by the satellites 110a-110c
provide a degree of isolation of one satellite transmission from
another. Additionally, the satellites 110a-110c can implement a
directional antenna to provide further signal selectivity. Thus, a
receiver can select the signals from a desired satellite, for
example 110a, by receiving the broadcast signals with a
corresponding polarized antenna oriented in the general direction
of the desired satellite 110a and tuning to the desired satellite
frequency. Because each satellite 110a-110c is configured in a
similar manner, a more detailed description is provided for only
one of the satellites 110a.
[0055] A satellite 110a in a satellite television system 100 can
include a single transponder (not shown), but typically includes
multiple transponders. Each of the transponders typically transmits
at a different frequency and has an associated polarization. Two
different transponders on the same satellite 110a can transmit on
the same frequency but with different polarities if the selectivity
provided by the difference in polarities is sufficient for the
system. If each transponder transmits at a different frequency, the
different transponders on a single satellite 110a can all transmit
with the same polarity, or can use different polarities.
[0056] Additionally, some transponders can be configured with
multiple carrier frequencies having various channel offsets. Other
transponders may multiplex numerous digital channels on a single
carrier. The integrated crosspoint switch with band translation
described below can be configured to operate over one or more
frequency bands with any transponder modulation type.
[0057] For example, a satellite 110a can include a first
transponder that provides information on multiple carrier
frequencies, with the carrier frequency spacing corresponding to a
channel spacing for a television receiver. The transponders in a
satellite 110a are typically arranged as transponder groups. For
example, the transponder group can be configured to provide a
contiguous group of channels. Alternatively, the signals in a
particular transponder group can have varied channel offsets, with
one or more channels having different carrier bandwidths or symbol
rates. Additionally, the transponders in a satellite group can be
configured to all transmit using the same polarization. A typical
satellite 110a configured for a satellite television system 100 can
include two transponder groups having sixteen transponders in each
transponder group, with each group having a different polarity. Of
course, the satellite 110a is not limited to any particular
transponder configuration, nor are transponder groups necessarily
limited to sixteen transponders.
[0058] A satellite 110a configured to operate in a satellite
television system 100 typically transmits downlink signals in one
of two frequency bands. Each frequency band can include one or more
channels corresponding to one or more transponders. A first
downlink frequency band is in the C-band and typically spans
3.6-4.2 GHz. A second downlink frequency band is in the Ku-band and
typically spans 10.7-12.75 GHz. Of course, each satellite or some
other signal source may transmit signals over one or more frequency
bands. The frequency bands are not limited to the two listed
frequency bands, and may be any suitable frequency bands, including
one or more frequency bands that have yet to be defined and
allocated by regulating bodies.
[0059] Of course, the upper and lower band edges for the one or
more downlink frequency bands are not absolutes because of the
practical limitations on constructing a brick wall filter. Rather,
the frequency bands typically represent passbands and the operating
transponder downlink frequency band typically comprises a frequency
band that includes a frequency band having the upper and lower band
edges within the passband. Alternatively, the band edges can define
stop band edges and the transponder can transmit a substantially
diminished energy outside of the band edge frequencies. Thus,
practically, the downlink frequency bands can span about, or
substantially, 3.6-4.2 GHz and 10.7-12.75 GHz. Additionally, while
a satellite 110a can be configured to use a particular downlink
frequency band, the satellite 110a may not actually transmit
signals at all frequencies within the downlink frequency band. A
satellite 110a is not limited to transmitting a downlink signal in
these two frequency bands, and there can be additional downlink
frequency bands implemented by the satellite 110a. These additional
downlink frequency bands can be distinct from the previously
described downlink frequency bands or can overlap some or all of
the previously described downlink frequency bands.
[0060] The downlink signals transmitted by the satellites 110-110c
can be received by a terrestrial television system and displayed to
one or more televisions 170a-170c. An antenna 120 is typically used
to receive the signals from the satellites 110a-110c. The antenna
120 is shown in FIG. 1 as a dish antenna but other antenna 120
configurations can also be used. In the embodiment implementing a
dish antenna 120, a reflector can direct the downlink signals to an
antenna feed 122. Although the antenna 120 is shown with only one
antenna feed 122, one or more antenna feeds 122 can be implemented
on a single antenna 120. Some antenna configurations suitable for
operation within the system can not include an antenna feed 122.
The antenna 120 or antenna feed 122 can be configured to receive
signals from a particular downlink frequency band or a particular
polarization. For example, the antenna 120 and antenna feed 122 can
be configured to receive the 10.7-12.75 GHz frequency band having a
left hand circular polarization. Another antenna feed (not shown)
included as part of the antenna 120 can be configured to receive
another downlink frequency band having the same or different
polarization. Additionally, although one antenna 120 is shown in
FIG. 1, multiple antennae can be implemented in a location or
multiple locations as part of a single system.
[0061] The output from the antenna 120 is connected to a receiver
180 that is used to process the received signals. In a typical
satellite television system 100 the receiver 180 includes low noise
amplifiers that amplify the signals while minimizing the associated
noise contribution. Additionally, the signals received at the
satellite downlink frequencies are typically frequency translated
to one or more predetermined frequency bands, or common
Intermediate Frequency (IF) bands. The received downlink signals
can also be filtered to remove out of band signals that can
contribute interference.
[0062] In one embodiment the carrier frequency spacing of the
downlink signals transmitted by the satellites 110a-110c typically
corresponds to a channel spacing used by a television receiver or a
set top box. In this embodiment, it can be advantageous to
frequency convert the entire received downlink frequency band to
one of the predetermined frequency bands used by television
receivers or set top boxes. Alternatively, the received downlink
frequency band can be frequency converted to predetermined
frequency bands at intermediate frequencies for further processing
prior to conversion to frequencies compatible with television
receivers or set top boxes. In another embodiment, several channels
may be multiplexed using a single carrier. In this embodiment, one
or more multiplexed carriers can be frequency converted to input
frequencies of a set top box.
[0063] The process of low noise amplification, filtering, and
initial frequency conversion can be performed by low noise block
converters (LNB) 130a-130c. Three LNB's are shown in FIG. 1, though
fewer or more can be used. A LNB, for example 130a, can be
configured to receive signals from one or more antennae, for
example 120, amplify, filter, and block frequency convert the
signals to a common IF band. A first set of downlink signals, such
as those from a first transponder group, can be block converted to
a first common IF band and a second set of downlink signals, such
as those from a second transponder group, can be block converted to
a second common IF band. For example, the LNB 130a can receive
downlink signals from two transponder groups. The multiple signals
from two transponder groups can be received at one or more antennae
120, or one or more antenna feeds 122. Additionally, the downlink
signals can originate from one satellite, for example 10a, or more
than one satellite 10a-110c.
[0064] For example, the LNB 130a can block convert the signals from
the first transponder group to a common IF band of 950-1450 MHz.
Similarly, the LNB 130a can simultaneously block convert the
signals from the second transponder group to a common IF band of
1650-2150 MHz. The block converted signals at the two common IF
bands can be combined prior to being output from the LNB 130a. This
process of block converting two transponder groups to different
predetermined frequency bands and then combining the signals from
the predetermined frequency bands is commonly referred to as
band-stacking. In the previous example, the band stacked output
from the LNB 130 comprises block converted transponder signals in a
first common IF band at 950-1450 MHz and block converted
transponder signals in a second common IF band at 1650-2150 MHz.
Conceivably, based on the channel spacing and carrier bandwidths
employed in particular transponder groups, signals from two
transponder groups can be block converted to the same common IF
band and combined without having two channels assigned to the same
carrier frequency. Typically, two independent signals would not be
combined at the same IF carrier frequency because each would appear
as an interference source for the other, potentially making both
signals unresolvable. In systems such as TDM or CDM systems, two
signals can occupy the same frequency space and still be
independently resolvable provided they occupy different spaces in
other dimensions, such as time or code.
[0065] If the number of transponder groups exceeds the number of
predetermined frequency bands, or common IF bands, it may not be
possible to band-stack the signals from all of the transponder
groups. In such a situation, the band-stacked output from a
particular LNB 130a may constitute only a subset of all available
transponder groups. Additional LNB's 130b-130c can be used to
ensure that signals from all of the transponder groups are
represented in one of the band-stacked outputs of the LNB's
130a-130c. However, the band-stacked outputs of the LNB's 130a-130c
are not limited to having signals from distinct transponder groups.
Thus, one or more of the band-stacked LNB outputs can have signals
in common with another of the band-stacked LNB outputs. In other
embodiments, band-stacking is not used, and each transponder group
is outputted from the LNB independently.
[0066] The outputs from the LNB's 130a-130c are connected to the
input of a switch configuration, referred to herein as an N.times.M
crosspoint switch 140. The N.times.M crosspoint switch 140 includes
N inputs and M outputs. Signals from each of the N inputs can be
selectively routed to any of the M outputs. Thus, the band-stacked
output from a first LNB 130a can be connected to a first input of
the crosspoint switch 140 and can be selectively routed to any of
the outputs of the crosspoint switch 140.
[0067] The crosspoint switch 140 can be configured such that only
one input can be selectively routed to an output. Alternatively,
the crosspoint switch 140 can be configured to selectively route
more than one input to the same output. Additionally, the
crosspoint switch 140 can also be configured such that an input
signal can be selectively routed to only one output. Alternatively,
the crosspoint switch 140 can be configured to selectively route an
input signal to more than one output. Typically, the crosspoint
switch 140 is configured to selectively route an input to a single
output and only one input can be routed to the particular output.
Where the crosspoint switch 140 configuration limits one output to
one input, there can be some inputs that cannot be routed to
outputs if the number if inputs, N, is greater than the number of
outputs, M. Similarly, some input signals can not be able to be
routed to an output if the crosspoint switch 140 configuration
limits an output to a signal from only one input, and one input can
be routed to multiple outputs.
[0068] Conversely, some outputs can not have any signals routed to
them if the crosspoint switch 140 configuration only allows one
input to be routed to one output and the number of inputs, N, is
less than the number of outputs, M. Similarly, some outputs may not
have any signals routed to them if multiple inputs can be routed to
the same output and an input can only be routed to one output. The
crosspoint switches in each of the embodiments can be configured in
the various alternatives discussed above.
[0069] Each of the outputs of the crosspoint switch 140 is coupled
to a corresponding input to a band translation section 150. The
band translation section 150 can represent an integrated device
that is configured to independently provide frequency band
translation to signals at each of its inputs. Alternatively, the
band translation section 150 can represent a collection of one or
more band translation devices that are configured to frequency band
translate signals at each of the inputs. In one embodiment, the
band translation section 150 can include one or more band
translation devices configured to frequency band translate one or
more signals using a common local oscillator. In another
embodiment, the band translation section can include one or more
band translation devices configured to independently frequency band
translate each of the input signals.
[0070] In one embodiment, a band translation device within the band
translation section 150 has an input connected to an output of the
crosspoint switch 140. An output of the band translation device
represents an output of the band translation section 150. The band
translation device can be configured to selectively couple an input
signal directly to the output with no frequency translation, or
alternatively to frequency translate the input signal to an output
signal at a frequency band that differs from the input frequency
band. The frequency translation device can further be configured,
such that when frequency translation is selected, to selectively
frequency translate the input signal from a first one of the
predetermined frequency bands to a second one of the predetermined
frequency bands.
[0071] In the satellite television embodiment described above,
there are two predetermined frequency bands. A first predetermined
frequency band spans 950-1450 MHz and the second predetermined
frequency band spans 1650-2150 MHz. In this embodiment, a band
translation device can frequency translate an input signal at one
of the two predetermined frequency bands to an output signal at one
of the same two predetermined frequency bands. It can be seen that
there are four distinct possibilities. An input signal in the lower
of the two predetermined frequency bands, 950-1450 MHz, can be
frequency translated by the band translation device to either the
lower, or the upper, of the two predetermined frequency bands.
Thus, in the example, the signal output from the band translation
device can be in the lower predetermined frequency band, 950-1450
MHz, or the upper predetermined frequency band, 1650-2150 MHz. Of
course, in one of the conditions, there is no frequency
translation, but rather, the input signal is coupled directly from
the input of the band translation device to the output of the band
translation device. The direct coupling from input to output
without frequency translation can be referred to as a pass through
state.
[0072] Similarly, an input signal provided to the band translation
device at the upper frequency band can be output from the band
translation device at the upper frequency band or the lower
frequency band. In one state the band translation device is
configured in pass through and in the other state the frequency
translation device is configured to frequency translate the input
signal.
[0073] The band translation section 150 can be configured to
combine the outputs from one or more band translation section.
Alternatively, external components (not shown) can combine one or
more band translation device outputs.
[0074] Thus, a receiver 180 can implement the LNB's 130a-130c, the
crosspoint switch 140, and the band translation section 150. The
receiver 180 can implement all of these elements in a single
integrated circuit or can implement one or more of the elements on
separate integrated circuits or stand-alone devices. For example,
the LNB's 130a-130c can each be implemented as stand-alone devices
and the crosspoint switch 140 with the band translation section 150
can be implemented on a single integrated circuit. The LNB's
130a-130c, crosspoint switch 140 and band translation section 150
can be implemented in a single housing. This arrangement can be
particularly advantageous where size of the components is of
concern. Additionally, combining the crosspoint switch 140 with the
band translation section 150 onto a single integrated circuit can
greatly reduce the power requirements over a discrete
configuration. Reducing the power requirements can result in
additional advantages. For example, an integrated circuit
crosspoint switch 140 and band translation section 150 having
reduced power requirements may allow a system with a smaller power
supply. Additionally, reduced power consumption typically
corresponds to reduced heat dissipation. A system having reduced
heat dissipation requirements can often use smaller heatsinks or
may eliminate heatsinks. The use of smaller heatsinks can further
reduce the size of the system. Additionally, an integrated circuit
embodiment can advantageously have reduced cost as compared to a
discrete system. The cost savings can be attributable to savings in
components and materials that can be minimized or eliminated when
the crosspoint switch 140 and band translation section 150 are
configured as an integrated circuit.
[0075] In another receiver 180 embodiment, portions of the
crosspoint switch 140 and portions of the band translation section
150 can be implemented on separate integrated circuits and one of
the integrated circuits can be packaged within a LNB, for example
130a. In still another receiver 180 embodiment, the LNBs 130a-130c
can be housed in a device that is remote from the crosspoint switch
140 and band translation section 150.
[0076] The outputs of the band translation section 150, and thus,
the outputs of the receiver 180, are coupled to corresponding
inputs of set top boxes 160a-160c. In the embodiment described, the
predetermined frequency bands do not correspond to typical
television receiver bands. Thus, the set top boxes 160a-160c can
further frequency translate the signals to television receiver
operating bands. Additionally, the output signals from the band
translation section 150 can be in a format that is not compatible
with standard television receivers 170a-170c. The set top boxes
160a-160c can then function as signal processing stages. For
example, the satellite downlink signals can be digitally modulated
in a format that is not compatible with a typical television
receiver 170a-170c. The set top boxes 160a-160c can be configured
to demodulate the digitally modulated signals, process the
demodulated signals, and then modulate a television channel carrier
frequencies with the signals for delivery to television receivers
170a-170c.
[0077] Alternatively, if the signals output from the band
translation section 150 are in a format and are at a frequency band
that is compatible with television receivers 170a-170c, the set top
boxes 160a-160c may not be required. In still another alternative,
one or more of the functions performed by the set top boxes
160a-160c can be integrated into the television receivers
170a-170c.
[0078] In the embodiment described in FIG. 1 and in the embodiments
described in the other figures, each of the television receivers
170a-170c can be connected to an output from one of the set top
boxes 160a-160c. Each of the set top boxes 160a-160c can have one
or more individually programmable outputs. However, more than one
television receiver 170a-170c can be connected to an output from a
single set top box, for example 160a. Alternatively, outputs from
more than one set top box 160a-160c, or multiple outputs from one
set top box such as 160a, can be combined or otherwise connected to
a single television receiver, for example 170a, although such a
configuration is not typical. A television receiver, for example
170a, can be configured to tune to a particular channel within the
one or more frequency bands provided by the set top box, such as
160a. The television receiver 170a can process the signal from the
selected channel to present some media content, such as video or
audio, to the user.
[0079] A user is typically provided control, such as through a
remote control for the television 170a or set top box 160a, to
selectively configure the crosspoint switch 140 or band translation
section 150. For example, a user can be allowed to select, using a
remote control configured to operate with the set top box 160a, to
receive signals from two distinct satellite transponder groups.
[0080] One of the satellite transponder groups can be received and
frequency converted to a common IF band using the first LNB 130a.
The first LNB 130a can be configured to frequency convert the
signals to the upper IF band, 1650-2150 MHz. The second of the
satellite transponder groups can be received and frequency
converted to a common IF band using the Nth LNB 130c. The Nth LNB
130c can also be configured to frequency convert the signals to the
upper IF band, 1650-2150 MHz. The LNB's of the other embodiments
can be similarly configured. Thus, the block converted signals from
the two transponder groups would ordinarily not be combinable if
any two channels in the two transponder groups share signal
bandwidths in the common IF bands.
[0081] However, in this example, the crosspoint switch 140 can be
configured by control signals to output the signals from the first
LNB 130a to a first crosspoint switch output and to output the
signals from the Nth LNB 130c to a second crosspoint switch output.
The band translation section 150 can then be configured, using the
control signals provided by the set top box 160a, to pass frequency
translate the signals from the first switch output from the upper
IF band to the lower IF band. The band translation section 150 can
also be configured to pass through the signals from the second
switch output without frequency translation. A combiner within the
band translation section can be configured to combine the output
signals from the first and second band translation outputs. The
composite signal then includes the signals from the first
transponder group, located at the upper common IF band, and the
signals from the second transponder group, located at the lower
common IF band.
[0082] Thus, the example can be generalized to allow signals from
any N signal sources, which can be satellite transponder groups, to
be combined to M distinct band stacked signals. The band stacked
signals can each include from one to M distinct frequency bands.
Each of the band stacked signals can then be delivered to a set top
box, multiple set top boxes, or one or more other receivers for
presentation to one or more users.
[0083] For example, an output from a first output of the receiver
180 can be coupled to one or more set top boxes, for example 160a
and 160b. Alternatively, multiple receiver 180 outputs that have
information in mutually exclusive bands can be power combined and
coupled to a single cable or distribution system for delivering the
signal to one or more set top boxes or receivers. In still another
embodiment, the crosspoint switch 140 may direct the same input
signal to two separate inputs of the band translation section 150.
The band translation section 150 may then frequency translate a
portion of the input to a first frequency band and may also
frequency translate a second portion of the input signal to a
second frequency band. The two frequency bands can be combined into
a signal that is directed to a single cable or distribution system.
In still other embodiments, two separate LNB's with their own
crosspoint switch with band translation section 150 having output
signals in separate frequency bands can have their signals power
combined at the LNB outside the house. In some embodiments, the
LNBs 130a-130c, crosspoint switch 140 and band translation section
150 are implemented as a single device that may be placed, for
example, at the antenna 120. In other embodiments, the LNBs
130a-130c may be implemented in a first device and the crosspoint
switch 140 and band translation section can be implemented as one
or more devices that can be located locally or remotely from the
LNBs.
[0084] The LNB's 130a-130c, crosspoint switch 140, band translation
section 150, and set top boxes 160a-160c can be assembled in many
different configurations. In each configuration, multiple
independent users can each select different channels from one or
more independent signals without affecting other users or
devices.
[0085] Other receiver embodiments can be assembled with multiple
crosspoint switches connected to the same LNB's. FIG. 2 is a
functional block diagram of an embodiment of a receiver 200 having
cascaded band translation switches 220, 221 that can be implemented
as integrated circuits (IC). In this embodiment of a receiver 200,
each of the band translation switches 220, 221 is configured with
two inputs and a single output. As discussed earlier with respect
to FIG. 1, the band translation switches 220, 221 are not limited
to any particular input and output configuration, but rather, can
implement any number of inputs and outputs. Additionally, the first
band translation switch 220 need not be configured the same as a
second band translation switch 221, although the similar
configurations are shown in FIG. 2.
[0086] Two LNB's 210a, 210b are configured to provide the input
signals to the first band translation switch 220. Each of the LNB's
210a, 210b can provide one or more signals in one or more frequency
bands. A first LNB 210a is connected to a first set of buffer
amplifiers 222a and 222b. The two buffer amplifiers 222a and 222b
are configured in parallel. Various alternative buffer amplifier
configurations can be used, as will be discussed in greater detail
below. A first internal buffer amplifier 222a connects the first
input of the first band translation switch 220 to an input of a
crosspoint switch 226. A first cascading buffer amplifier 222b
connects the first input of the first band translation switch 220
to a first cascaded output. The first internal buffer amplifier
222a and the first cascading buffer amplifier 222b can be
configured similarly or can be configured differently. Each of the
first set of buffer amplifiers 222a, 222b, can be configured to
provide gain or attenuation. The buffer amplifiers 222a, 222b can
be unity gain amplifiers or can provide significant signal gain.
Alternatively, the buffer amplifiers 222a, 222b can provide
attenuation rather than gain.
[0087] The buffer amplifiers 222a, 222b can be configured such that
changes at the output of one of the amplifiers, for example 222a,
do not affect the other amplifier 222b. For example, the output of
the internal buffer amplifier 222a is connected to an input of the
crosspoint switch 226. The load that the input of the crosspoint
switch 226 provides to the output of the internal buffer amplifier
222a can vary depending on various factors. For example, the
impedance of the crosspoint switch 226 input can vary as a function
of frequency. Alternatively, the impedance of the crosspoint switch
226 input can vary depending on the switch output selected or on
the configuration of devices that can be connected to the selected
crosspoint switch 226 output. Additionally, signals, such as local
oscillator signals can be present at the output of the buffer
amplifiers. Theoretically, the input of the crosspoint switch 226
can present any load from a short circuit to an open circuit,
although in an actual configuration the actual variation in the
load will not likely span the entire range. Ideally, the internal
buffer amplifier 222a is configured such that its effect on the
input of the band translation switch 220 does not vary for any load
or signal presented at its output. Typically, the internal buffer
amplifier 222a provides a level of signal isolation that is not
infinite, but is great enough such that effects experienced at its
input are minimal. Similarly, input load variation is not zero, but
is minimal.
[0088] Similarly, the cascading buffer amplifier 222b can
experience a load variation spanning from an open circuit to a
short circuit because the output of the cascading buffer amplifier
222b is connected to an output of the first band translation switch
220. The cascading buffer amplifier 222b can likewise be configured
to minimize the effect of load variations on the input of the
amplifier.
[0089] The second input to the first band translation switch 220 is
configured similarly to the first input. A second set of buffer
amplifiers 224a, 224b have inputs connected to the second input. A
second internal buffer amplifier 224a connects the second input to
a second input of the crosspoint switch 226. A second cascading
buffer amplifier 224b connects the second input to a second
cascaded output of the first band translation switch 220. The
second set of buffer amplifiers 224a, 224b are also configured to
be insensitive to load variations and signals present at their
outputs.
[0090] The crosspoint switch 226 in the first band translation
switch 220 is configured to selectively couple each of the two
internal buffer amplifier, 222a, 224a, outputs to one of two switch
outputs. Although only two switch outputs are shown in the first
band translation switch 220, any number of switch outputs can be
incorporated into the crosspoint switch 226.
[0091] A first switch output is connected to a first band
translation device 228a and a second switch output is similarly
connected to a second band translation device 228b. As discussed
earlier with respect to FIG. 1, each of the band translation
devices 228a-228b can be configured to frequency translate a signal
or pass the signal to its output without frequency translation.
[0092] The outputs of the band translation devices 228a-228b are
connected to inputs of a signal combiner 230, here represented as a
signal summer. The output of the first band translation device 228a
is connected to a first input of the signal combiner 230 and the
output of the second band translation device 228b is connected to a
second input of the signal combiner 230.
[0093] The signal combiner 230 is configured to combine the signals
provided at its inputs and output a combined composite signal. The
signal combiner 230 is shown as a signal summer, which is
configured to sum the input signals and provide the composite
signal at its output. Typically, the signal provided by the first
band translation device 228a occupies a frequency band that is
distinct from frequency band occupied by the signal provided by the
second band translation device 228b. Because the signals from the
two band translation devices 228a-228b are effectively frequency
multiplexed, the input signals can be combined to provide a
composite signal without experiencing destructive interference. Of
course, the input signals are not required to occupy distinct
frequency bands. The first input signal can occupy a band that
overlaps some or all of the frequency band occupied by the second
input signal. The resulting combined composite signal can
experience some destructive signal interference if desired signals
in the input signals occupy the same signal space. Alternatively,
no signal interference can occur in the combined signal if the
signal components in the input signals do not occupy the same
signal space, for example frequency, time, or code space.
[0094] The composite output from the signal combiner 230 is
connected to a first set top box 240a. Alternatively, the output
from the signal combiner 230 can be coupled to more than one set
top box, or to a signal distribution system (not shown) that can be
coupled to one or more set top boxes and one or more receivers. As
explained earlier with respect to FIG. 1, the set top box 240a can
be configured to further process the composite signal in order for
the signals to be compatible with an end user device (not shown),
such as a television.
[0095] A second band translation switch 221 is configured similarly
to the first band translation switch 220. A first set of buffer
amplifiers 223a, 223b receives a first input signal and a second
set of buffer amplifiers 225a, 225b receives a second input signal.
The first set of input buffers 223a, 223b includes an internal
input buffer amplifier 223a that amplifies the first input signal
and provides the amplified signal to an input of a crosspoint
switch 227. The first set of input buffers 223a, 223b also includes
a cascading buffer amplifier 223b that amplifies the first input
signal and provides the amplified signal to an output of the second
band translation switch 221.
[0096] The second set of input buffers 225a, 225b is similarly
configured. The crosspoint switch 227, band translation devices
229a, 229b, and signal combiner 231 for the second band translation
switch 221 are configured similarly to the corresponding element
from the first band translation switch 220. The output of the
second band translation switch 221 similarly is connected to a
second set top box 240b and can be coupled to more than one set top
box.
[0097] However, the input signals provided to the second band
translation switch 221 are provided from the cascading buffer
amplifiers 222b, 224b of the first band translation switch 220.
Thus, by providing a cascading buffer amplifiers on the band
translation switches 220, 221 the signals from LNB's 210a-210b can
be provided to any number of band translation switches 220, 221 and
ultimately to any number of set top boxes 240a-240b.
[0098] The LNB 210a-210b outputs can provide signals to multiple
set top boxes 240a-240b without the number of set top boxes
240a-240b significantly affecting the quality of the signals to any
other set top box, for example 240a. Thus, the signal quality from
a particular LNB, for example 210a, at a particular set top box,
for example 240a, is not significantly affected by the number of
set top boxes 240a-240b that are ultimately connected to the LNB
210a. Additionally, connections from an LNB 210a to set top boxes
240a-240b can be added or subtracted, either through changes in the
number of band translation switches 220, 221 cascaded or through
selection at one of the crosspoint switches 226, 227, without
substantially affecting the signal quality at a particular set top
box 210a. The ability of the end user device to present the
information, and the end users' ability to perceive differences in
signal quality are factors that contribute to the amount of signal
degradation that can occur in a signal to a set top box without
there being a substantial affect on the signal quality at the set
top box.
[0099] In addition to providing cascade outputs, a band translation
switch can be configured to provide a pass through signal path in
the band translation devices. The band translation switch FIG. 3
shows a functional block diagram of an embodiment of a band
translation switch 300 having band translation devices 330a-330b
configured to selectively provide frequency translation or pass
through. The band translation switch 300 can be implemented as a
single integrated circuit.
[0100] A series of signal inputs are each coupled to the input of a
corresponding buffer amplifier 310a-310d. In one embodiment, each
of the signal inputs can receive a frequency multiplexed signal,
such as an output from a LNB (not shown). The frequency multiplexed
signal can be a band-stacked signal including signals from one or
more predetermined frequency bands. In another embodiment, one or
more of the signal inputs can receive information corresponding to
a single channel. In still another embodiment, one or more of the
signal inputs can receive a multiplexed signal containing several
channels. The multiplexed signal can be, for example, a digitally
modulated signal that is multiplexed to carry several channels.
[0101] Each of the buffer amplifiers 310a-310d amplifies the
received signal and couples the amplified output to a corresponding
input of a crosspoint switch 320. The buffer amplifiers 310a-310d
can be configured to couple the amplified signals solely to the
respective crosspoint switch 320 inputs. Alternatively, one or more
of the buffer amplifiers 310a-310d can be configured to also couple
the amplified signals to corresponding cascade outputs (not shown)
as will be disclosed in further detail below with respect to FIGS.
4A-4C.
[0102] As previously disclosed, the crosspoint switch 320 can be
configured to selectively couple any one of its inputs to any one
of its outputs. Thus, the amplified signals output from the buffer
amplifiers 310a-310d and provided to the inputs of the crosspoint
switch 320 can be selectively coupled to any of the outputs of the
crosspoint switch 320. In the embodiment shown in FIG. 3, the
crosspoint switch 320 has two outputs, although the crosspoint
switch 320 can typically implement any number of outputs.
[0103] A first output from the crosspoint switch 320 is coupled to
a first band translation device 330a. Similarly, a second output of
the crosspoint switch 320 is coupled to a second band translation
device 330b. Although the two band translation devices 330a, 330b
are typically similar structures, this is not a requirement. Each
of the band translation devices 330a-330b can, for example, have a
different structure.
[0104] The first band translation device 330a is configured with an
input switch 336a which selectively routes an input signal to one
of a frequency translation path or a pass through path in the band
translation device 330a. The input switch 336a for the first band
translation device 330a is shown in the frequency translation
position.
[0105] When the input switch 336a is configured to route an input
signal to a frequency translation path, the input signal is routed
to an input of a frequency translation device, such as a mixer
332a. A signal from a local oscillator (LO) is coupled to a LO port
on the mixer 332a. The LO can be integrated onto the same
integrated circuit as other components of the band translation
switch 300 or can be implemented externally to an integrated
circuit band translation switch 300.
[0106] The band translated output frequency from the mixer 332a is
a function of the input signal to the mixer and the LO frequency.
As discussed above, the signals provided to the buffer amplifiers
310a-310d, and thus to the band translation devices 330a-330b, can
be signals in predetermined frequency bands. The LO frequency can
be programmable, or otherwise tunable, in order to allow an any one
of the predetermined input frequency bands to be frequency
translated to any one of the predetermined output frequency bands.
Typically, the mixer 332a is configured such that the frequency
translated output from the mixer 332a is optimized for ideal
multiplier products. The desirable multiplier products can include
the sum and difference frequency components or only one of the sum
or difference frequency components. Typically, the input signal and
the LO signal are undesirable signals at the output of the mixer.
The mixer 332a can be configured such that higher order mixer
products are negligible, that is, the higher order mixer products
are below the desired mixer products by a predetermined amount,
such as 40 dB. Alternatively, higher order mixer products can be
lower than 10 dB, 15 dB, 20 dB, 30 dB, 50 dB, 60 dB or some other
level. Alternatively, the mixer 332a can be configured such that
significant higher order mixer products appear at the output of the
mixer 332a. Some or all of the higher order mixer products can be
determined to not adversely affect the system. Alternatively, some
or all of the higher order mixer products can be substantially
reduced through filtering in subsequent stages (not shown).
Filtering can be performed by implementing filters or can be
accomplished as a result of the frequency response of elements that
appear after the mixer 332a output. The filters can be implemented
on the same substrate as the buffers 310a-310d, crosspoint switch
320, and mixers 332a and 332b. Alternatively, the filters can be
implemented separate from the buffers 310a-310d, crosspoint switch
320, and mixers 332a and 332b and need not even be implemented
locally to the other devices. The filters may have a passband that
is greater than, equal to, or less than an input frequency
bandwidth.
[0107] Some or all of the predetermined input frequency bands can
be the same as, or different from, some or all of the predetermined
output frequency bands. In one embodiment, the predetermined input
frequency bands are distinct from one another and the predetermined
output frequency bands are substantially identical to the
predetermined input frequency bands. For example, the predetermined
input frequency bands can include a first input frequency band of
950-1450 MHz and a second input frequency band of 1650-2150 MHz.
The predetermined output frequency bands can then include a first
output frequency band at 950-1450 MHz and a second output frequency
band of 1650-2150 MHz, such that the predetermined input and output
frequency bands are the same.
[0108] The frequency translated signal is output from the mixer
332a and coupled to an output switch 338a. The output switch 338a
is configured to couple the signal output from the selected signal
path to an output of the band translation switch 300. The output
switch 338a for the first band translation device 330a is shown as
selecting the frequency translation path.
[0109] The second band translation device 330b is configured
similarly to the first band translation device 336a. An input
switch 336b is configured to select a frequency translation signal
path or a pass through signal path. The frequency translation path
uses a mixer 332b driven by a LO to frequency translate the input
signal. An output switch 338b couples the signal from the selected
signal path to an output of the band translation switch 300.
[0110] The input switch 336b of the second band translation device
336b is shown as selecting the pass through signal path. Similarly,
the output switch 338b or the second band translation device 330b
is shown as selecting the pass through signal path. The pass
through signal path is shown as a direct connection from the input
switch 336b to the output switch 338b. However, any number of
elements can be present in the pass through signal path. The only
limitations are that the elements in the pass through signal path
do not result in a frequency translation of the desired input
signal to a different frequency at the output of the band
translation device 330b. One or more filters or frequency selective
devices placed after the band translation device 330b can be
configured to filter the output of the band translation device
330b. The filters may have a passband that is greater than, equal
to, or less than an input frequency bandwidth. Thus, even if the
band translation device, for example 330a or 330b, is configured in
pass through mode, the output band can be narrower than the input
frequency band.
[0111] A band translation device, for example 330a, having an input
switch 336a and an output switch 338a configured to select a signal
path, typically controls the input switch 336a and the output
switch 338a to select the same signal path. It can be advantageous
to implement a band translation device 330a with both an input
switch 336a and an output switch 338a in order to limit the effects
the non-selected signal path has on the performance of the selected
signal path. Of course, a band translation device, for example
330a, is not limited to any particular configuration, and need not
incorporate two switches. Additionally, the band translation
device, for example 330a, need not incorporate a pass through path,
but instead, can incorporate two or more frequency translation
paths.
[0112] FIG. 4A illustrates a simplified switch diagram of a doubly
balanced mixer as known in the art. The mixer 400 includes RF and
IF ports 410 and 430, respectively, each of which is shown
differentially, but may be single-ended in another embodiment. The
differential RF signal 410 is supplied to the input of SPDT
switches 422, the states of which are switched at a rate determined
by an LO signal 425 supplied thereto. The outputs of the switches
422 are coupled to differential IF ports 430 operable to provide
the differential IF signal 430.
[0113] FIG. 4B illustrates the doubly balanced mixer of FIG. 4A as
a Gilbert cell multiplier or mixer circuit known in the art. The
mixer circuit includes two cross-coupled differential transistor
pairs 422 whose base terminals are coupled to the LO source 425,
collector terminals are coupled to the IF loads 430, and emitter
terminals are coupled to buffer transistors 417. Responsive to the
differential RF signal applied at terminals 410a and 410b, a
voltage difference is established across resistor 415, resulting in
the corresponding modulation of the quiescently-supplied current
driving the transistor pairs 422 that comprise the mixer core.
Those skilled in the art will appreciate that illustrated mixer
circuit is only exemplary, and numerous variations of the circuit
are also widely used.
[0114] While doubly balanced mixers provide a high level of even
order mixing product suppression, circuit imperfections lead to
degradation in that suppression. For example, a relatively low
impedance parasitic 412 (e.g., capacitance) can load the emitter
nodes of the mixers, the impedance operating to convert the
rectified LO voltage into a common mode even order LO interferer
current. The LO interferer can then pass through the mixer core and
to the output loads.
[0115] Reduced mixer even-order suppression can be especially
problematic when the mixer is integrated with other circuitry. FIG.
5 illustrates one example of such an instance where multiple mixers
are supplied by a single VCO. Each mixer is configured to operate
in either a mixing mode, whereby the synthesized signal 505 and
input signals 510a and 510b are provided to respective mixers 520a
and 520b to produce respective mixed signals 530a and 530b, or in a
bypass mode, whereby the synthesized signal 505 is not supplied to
the mixer 520c and the input signal 510c is routed such that it
bypasses the mixer 520c.
[0116] FIG. 6A illustrates a simplified switch diagram of a mixer
circuit 600 in accordance with one embodiment of the invention. The
mixer circuit 600 includes a mixer core 620 and a mode select
circuit 640. The mixer core 620 includes two input switches 624 and
628, each switch having an input 624a, 628a, and two outputs 624b,
624c, and 628b, 328c, respectively. Switches are depicted to convey
the component's general function, and those skilled in the art will
readily appreciate that each may be realized using an variety of
circuit elements, including transistors (BJT and FET types),
diodes, and the like. Accordingly, as used herein, the term
"switch" or "switches" shall denote any of these circuit elements,
or equivalents thereof.
[0117] Input switches 624 and 628 are operable to accept a signal
at a first frequency f.sub.1 in either a differential or
single-sided form. In a differential form, the first frequency
signal f.sub.1 will consist of a differential signal, wherein
separate polarities of the first frequency signal f.sub.1 are
supplied to separate switch inputs 410a and 410b, respectively.
During single-sided operation, only one of the switches' inputs
(e.g., 410a) is needed to receive the first frequency signal
f.sub.1. In this embodiment, the input of the other switch (e.g.,
410b) is coupled to an ac ground.
[0118] The first and second switches 624 and 628 are further
configured to receive a switching signal 425, which operates to
switch the first and second switches between their respective
output states at a second frequency f.sub.2, as will be further
described below. In a particular embodiment shown, the first and
second switches 624 and 628 are configured such that both, upon
receiving the switching signal 425, switch to the opposite states
(i.e., one to its first output, and the other to its second
output). In such an embodiment, the switching signal 425 may be
supplied in anti-phase to configure the first and second switches
in opposite output states.
[0119] The mixer circuit 600 further includes a mode select circuit
640, implemented in one embodiment as third and fourth switches 644
and 648. Third switch 640 includes an input 644a switchable to two
outputs 644b and 644c, and fourth switch 648 includes an input 648a
switchable to two outputs 648b and 648c. As shown, the third
switch's input 644a is coupled to the second output 624c of the
first switch. The third switch's first output 644b is coupled to
the first switch's first output 624b, and the third switch's second
output 644c is coupled to the second switch's second output 628c.
The fourth switch's input 648a is coupled to the second switch's
first output 628b. The fourth switch's first output 648b is coupled
to the first switch's first output 624b, and a second output 648c
coupled to the second switch's second output 628c.
[0120] The third and fourth switches are further configured to
receive a mode select signal 650 operable to select the output
state of the third and fourth switches 644 and 648. In a specific
embodiment, the third and fourth switches 644 and 648 are
collectively configured to operate in one of two states: a bypass
state or a mixing state. The bypass state is exemplified in FIG. 6A
with the third switch 644 coupled to its first output 644b and the
fourth switch 648 coupled to its second output 648c as shown in
FIG. 6A. The mixing state could be alternatively realized by
switching both states of the third and fourth switches, as will be
further illustrated below. The resulting bypass or mixed signal is
provided in differential form at output ports 430a and 430b. IF
loads 665a and 665b are each coupled to a respective output port
and an ac ground, as described below. In an alternative embodiment,
a single IF load may be coupled between ports 665a and 665b. The
term "IF" load shall not infer that the loads' frequency of
operation is limited to those frequencies below the input signals,
and in fact may be a higher operational frequency when the desired
mixing product is an upconverted signal. Further, the IF load may
comprise active or passive components as known in the art.
[0121] The switches (or their corresponding implementation in
transistors, diodes, or other components) may be discretely or
integrally formed using a variety of fabrication techniques known
in the art, including monolithic fabrication in a Bipolar
Complementary Metal Oxide Semiconductor (Bi-CMOS) process.
Additional circuitry described herein, such as IF loads and
oscillator circuitry, as well as other components may be
monolithically formed onto an integrated circuit device in
accordance with the invention.
[0122] FIG. 6B illustrates a method for operating the mixer circuit
shown in FIG. 6A in accordance with the invention. Initially at
672, a first frequency signal is supplied to one or both of the
input switches 624 and 628. As explained above, the first frequency
signal may be in the form of a single-sided signal, in which case
the first frequency signal is applied to one of the inputs 410a or
410b, the other switch input being coupled to an ac ground. In the
case of an applied differential signal, oppositely polarized
signals are supplied to respective signal inputs 410a and 410b.
[0123] Next at 674, the input terminal of the first and second
switches is switched between each switch's first and second output
at a second frequency f.sub.2. In a specific embodiment, this
process is performed by using the second frequency signal as a
switch control signal. In such an embodiment, an oscillator or
other frequency source used to generate the second frequency signal
may be local/integrated with one or more of the switches, or may be
externally located and supplied to the first and second switches
via a transmission medium.
[0124] Next at 676, the first and second outputs of each input
switch are coupled to either: (i) a node common to the switch's
first and second outputs, or (ii) nodes of opposite polarity.
Connecting the two outputs to a common node results in the first
frequency signal being output at 430 (bypass mode), and connecting
the two outputs to opposite polarity nodes results in a mixing
operation of the first and second frequency signals, and
accordingly the generation of one or more mixing products
therefrom. The process of 676 is performed by the mode select
circuit 640 illustrated in FIG. 6A, whereby in the bypass mode of
operation the third switch is configured to connect the first and
second outputs of the first switch 624 to a common node 430a, and
the fourth switch 648 is configured to connect the first and second
outputs of the second switch 628 to a common node 430b. To operate
in the alternate mixing mode, the process is performed by switching
the states of the third and fourth switches. In this state, the
first and second outputs of the first switch 624 will alternately
connect to opposite polarity nodes 430a and 430b, and the first and
second outputs of the second switch 628 will also alternately
connect to opposite polarity nodes 430b and 430a. In this manner,
the output states of the third and fourth switches 644 and 648 are
configurable either in a bypass mode where the outputs of the first
and second switches are connected to a common node, or in a mixing
mode where the outputs of the first and second switches are
connected to opposite polarity nodes. Further preferably, the
connection between the mixer core (first and second switches) and
the IF loads 665a and 665b is maintained during operation in either
the bypass mode or mixing mode. Additionally, the first and second
switches are controlled to continuously switch between their
respective outputs at the second frequency during operation in
either the bypass or mixing modes. As noted above, a single IF load
may be coupled between nodes 430a and 430b in an alternative
embodiment under the invention.
[0125] FIG. 6C illustrates the mixer circuit of FIG. 6A as a
modified Gilbert cell mixer circuit 680 in accordance with the
invention, with previously identified components retaining their
reference numerals. As depicted, each of the switches 624, 628,
644, and 648 are implemented as a differential pair of npn bipolar
junction transistors Q1-Q8. A mixer/bypass control circuit produces
a control signal 650 which is supplied differentially to the base
terminals of the mode select switches 644 and 648. The polarity of
control signal may be reversed to switch circuit operation between
bypass and mixing modes. A signal source (e.g., a LO source) is
operable to provide the second frequency signal 425 to the mixer
core switches 624 and 628. The input signal f.sub.1 is applied to
the input terminals 410a and 410b of buffer transistors 417, or
alternatively may be provided as a single-ended signal, in which
case one of the input terminals 410a or 410b is coupled to an ac
ground, as described above.
[0126] In a specific embodiment of the mixer circuit 680,
transistors Q1-Q8 are npn bipolar transistors 20 um.times.0.4 um,
IF loads 665a and 665b are 200 ohms, resistor 415 is 200 ohms, the
first frequency signal f.sub.1 operates at 950-2150 MHz, the second
frequency signal f.sub.2 operates at 3100 MHz, and the mode select
signal 650 is 500 mV DC. The circuit's supply V.sub.cc operates at
+6 VDC. Further specifically, the illustrated components are
integrally formed using a 0.35 um Bi-CMOS photolithographic
process. Skilled practitioners will appreciate that that the
circuit 680 can be alternatively realized using various
modifications, e.g., pnp-type bipolar transistors, n or p-type
field effect transistors, or other components such as diodes, and
the like.
[0127] FIG. 7 illustrates a second embodiment of the mixer circuit
in accordance with the invention. The mixer circuit 700 includes a
mode select switch 710 having an input 710a to receive the first
frequency signal f.sub.1, a first output 710b coupled to a bypass
circuit 720, and a second output 710c coupled to a mixing core 730.
The bypass circuit 720 may be any transmission medium operable to
support the propagation of the first frequency signal therealong,
including printed/integrated circuit traces (including ungrounded
lines or grounded lines such as microstrip, stripline, coplanar
waveguide and the like), wire, twisted pair line, coaxial cable,
conductive or dielectric waveguide, and the like. The mixer core
730 has an input coupled to the switch's second output 710c, a
second input operable to receive the second frequency signal 425,
and an output. The mixer core 730 can be of any conventional type
(i.e., single-ended, singly-balanced, doubly balanced, etc.) and
realized in any of the known forms, such as a Gilbert cell
mixer.
[0128] The mixer circuit further comprises a signal combiner 740
coupled to the bypass circuit 720 and the mixer output. The signal
combiner 740 has an output coupled to a common load 750. The signal
combiner may be realized as a commonly connected port, power
combiner (active or passive), or similar circuits. Selection of the
bypass or mixing mode is provided by a mode select signal 650,
which is supplied by a control circuit. The mixer core 730 is
supplied the second frequency signal 425 via a local oscillator.
One or more of the described components may be integrally
fabricated into a monolithic circuit using semiconductor processing
techniques appropriate for the particular material. Preferably, the
operation of the mixer core continues and circuit connections
between the mixer core 730 and the common load 750 is maintained
during operation within or switching between the bypass and mixing
modes.
[0129] FIGS. 8A and 8B illustrate exemplary systems using the
improved mixer circuit of FIG. 6A or 7 in accordance with the
invention, the exemplary systems comprising a common oscillator,
multiple mixer system. Referring first to FIG. 8A, the system 820
includes a frequency synthesizer 822 and a VCO 824 coupled to three
mixer circuits 680.sub.1-3, the detailed architecture of each being
described in FIGS. 6A and 6C, above. The frequency synthesizer 822
and VCO 824 operate to produce the second signal frequency f.sub.2,
which is supplied to each of the three mixer circuits 680.sub.1-3.
Each mixer circuit 680.sub.1-3 is additionally supplied with a
first frequency signal 612.sub.1-3 in differential form. Each first
frequency signal 612.sub.1-3 is identified as f.sub.1 for
convenience, and the reader will appreciate that each of these
frequencies may be different.
[0130] The mixer circuits 680.sub.1-3 are supplied respective mode
select signals 650.sub.1-3 to configure each corresponding mixer
circuit to their desired output. In the shown embodiment, the first
mixer circuit 680.sub.1 is supplied a "bypass" mode signal
650.sub.1, resulting in the (differential) output at the first
signal frequency f.sub.1. The second mixer circuit 680.sub.2 is
supplied a "mix" mode signal 650.sub.2, resulting in the output of
the mixing product described above. Similarly, the third mixer
circuit 680.sub.3 receives the first and second frequency signals
f.sub.1, f.sub.2, and the "mix" mode signal 650.sub.3, resulting in
the mixing mode of operation. Preferably, the first and second
switches of each mixer circuit 680.sub.1-3 continues to switch at
the second frequency f.sub.2 regardless of whether the supplied
control signal 650.sub.1-3 sets the mixer circuit to a bypass mode
or mixing mode. Additionally, the mixer circuit's coupling to the
IF loads 665a and 665b (or to a single IF load coupled between
differential nodes 430a and 430b) is maintained during and
switching between the bypass and mixing modes.
[0131] FIG. 8B illustrates another embodiment of a common
oscillator, multiple mixer system 850 in which the mixer circuit of
FIG. 7 is employed. The system 850 similarly employs a frequency
synthesizer 822 and a VCO 824 for generating the second frequency
signal f.sub.2 which is commonly supplied to the mixer circuits
700.sub.1-3. In this system, the first frequency signal 612.sub.1-3
(identified as f.sub.1 for convenience only) is supplied to the
input of each mixer circuit. A corresponding mode select signal
650.sub.1-3 is also supplied to set the switches in the desired
bypass or mixing mode enabling the independent control of each
mixer circuit in either a bypass or mixing mode. As described above
with respect to the mixer embodiment of FIG. 7, the operation of
the mixer continues and circuit connections between the mixer core
730 and the common load 750 can be maintained during operation
within or switching between the bypass and mixing modes in order to
maintain a substantially constant level of LO even-order
interference.
[0132] Just as there are various embodiments for the band
translation devices, there are also various embodiments for
cascadeable buffer amplifiers. FIGS. 9A-9C show functional block
diagrams of various embodiments of cascadable buffer amplifiers
with switches. FIG. 9A shows a functional block diagram of an
embodiment having a single buffer amplifier 910 configured to drive
both the input to the crosspoint switch 912 as well as the cascade
output. This configuration uses fewer elements than do the
embodiments shown in FIGS. 9B or 9C. However, the design of the
buffer amplifier 910 can be more complicated because the buffer
amplifier 910 is configured to drive two potentially independent
loads. Ideally, changes in the load placed on the cascade output
should not affect the signal provided to the crosspoint switch 912.
Similarly, changes in the configuration of the crosspoint switch
912 and devices (not shown) connected to the output of the
crosspoint switch 912 should not affect the signal provided to the
cascade output. Thus, the embodiment shown in FIG. 9A can show the
fewest elements, but can require a more complex buffer amplifier
910 design.
[0133] The embodiment of FIG. 9B is equivalent to the buffer
amplifier embodiment disclosed in FIG. 2. The buffer amplifier
embodiment uses two buffer amplifiers 920, 924 connected in
parallel. The input signal provided, for example, by an LNB is
coupled to an internal buffer amplifier 920 and a cascading buffer
amplifier 924. The internal buffer amplifier 920 amplifies the
input signal and drives the input to the crosspoint switch 928. The
cascading buffer amplifier 924 amplifies the input signal and
drives the cascade output. The cascade output can represent an
output from an IC which includes the buffer amplifier embodiment.
As discussed previously with respect to FIG. 2, the internal buffer
amplifier isolates load changes and interference at its output from
its input. The internal buffer amplifier 920 thus provides signal
isolation to the buffer amplifier input and the cascade output.
Similarly, the cascading buffer amplifier 924 isolates load changes
and interference at its output from its input, thereby providing
signal isolation to the buffer amplifier input and the cascade
output.
[0134] The two amplifier embodiment has the advantages of high
isolation between the outputs of the buffer amplifiers and between
the outputs of the buffer amplifiers to the input of the buffer
amplifiers. However, the multiple amplifier embodiment can require
more active devices, which consume power, generate heat, generate
interference, and increase cost. Thus, the multiple amplifier
configuration may not be the most desirable solution for all
applications.
[0135] Another buffer amplifier embodiment is shown in the
functional block diagram of FIG. 9C. In the embodiment of FIG. 9C,
the input signal, for example the output from an LNB, is provided
to the input of a power divider 930. The power divider 930 can for
example, be a passive power divider or an active power divider.
Additionally, the power divider 930 can be configured to provide
substantially an equal power split or can be configured to provide
a majority of the power to one of the output ports. The signals
provided at each of the output ports of the power divider 930 can
be in-phase or can be out of phase.
[0136] One of the output signals from the power divider 930 is
coupled to a buffer amplifier 934 that is configured to amplify the
signal and couple the amplified signal to an input of the
crosspoint switch 938. The other output from the power divider 930
is coupled to the cascade output.
[0137] The embodiment of FIG. 9C can be advantageous because the
power divider 930 can be implemented as a passive device that
consumes no power. Additionally, the power divider 930 provides a
degree of isolation between the cascade output and the buffer
amplifier output. There is also signal isolation between the output
ports of the power divider 930. However, if the power divider 930
is a passive device, the signal isolation between the output ports
is typically not as great as is obtainable from the two amplifier
configuration. Additionally, a passive device can require
considerable area on an integrated circuit or can not be feasible
to implement on an integrated circuit. Also, the signal quality can
be severely degraded if multiple passive power divider stages are
cascaded.
[0138] Thus, various buffer amplifier embodiments have been shown.
However, each of the embodiments has advantages and disadvantages
and the optimal configuration will depend on the requirements of
the system. Additionally, the embodiments disclosed are only
examples and are not meant to be an exhaustive list of available
configurations. Other configurations can be implemented.
[0139] FIG. 10 is a functional block diagram of a crosspoint switch
with band translation 1000. A two input and two output version of
the receiver 180 of FIG. 1 can be implemented with the crosspoint
switch with band translation 1000 of FIG. 10 in combination with
two LNB's. For example, the receiver of FIG. 1 can include LNB
modules connected to an integrated circuit implementation of the
crosspoint switch with band translation 1000. This configuration of
a receiver allows signal routing and band translation to be
performed at a location physically close to the LNBs. The physical
proximity of LNBs to the crosspoint switch with band translation
200 minimizes the loss and induced noise experienced by the
received signals.
[0140] The crosspoint switch with band translation 1000 is not
limited to having only two inputs and two outputs. Other
embodiments of the crosspoint switch with band translation 1000 can
include additional inputs and outputs. The number of inputs can be
generalized to any number, N. The number of inputs, N, can be, for
example, 2, 3, 4, 5, 6, 7, 8, 9, 10, 11, 12, 13, 14, 15, 16, or
some other number. Similarly, the number of outputs can be
generalized to any number, M. The number of outputs, M, can be, for
example, 1, 2, 4, 6, 8, 10, 12, 14, 16, 18, 20, 22, 24, 26, 28, 30,
32, or some other number.
[0141] Additionally, the crosspoint switch with band translation
1000 can be located remote from a signal source, such as an antenna
or LNB modules. For example, one or more coaxial cables can couple
the outputs from LNB modules to inputs of the crosspoint switch
with band translation 1000. In an example environment such as
signal distribution within a residence, the LNB modules can be a
distance of more than 250 feet away from the crosspoint switch and
can couple to the LNB modules with coaxial cables.
[0142] The crosspoint switch with band translation 1000 can be
configured using differential signal interconnections to improve
signal isolation. The device can be implemented with single ended
signal interconnections but differential signal interconnections
typically provide greater isolation. Signal isolation is of greater
concern when the device is implemented in a single integrated
circuit.
[0143] The crosspoint switch with band translation 1000 has a first
signal path and a second signal path. The first signal path
includes a first low noise amplifier (LNA) 1010a connected to an
arrangement of switches, 1022a, 1024a, 1026a, and 1028a, that can
selectively route a signal at the output 1014a of the LNA 1010a to
a first band translation device 1030a or a second band translation
device 1030b. The crosspoint switch with band translation 1000 of
FIG. 10 is configured to provide voltage-mode switching of the
signals.
[0144] The first LNA 1010a is configured with a differential input
1012a and a differential output 1014a. The differential input 1012a
of the first LNA 1010a can be, for example, matched to 75 ohm
differential. The differential output 1014a of the first LNA 1010a
is configured to have a low impedance. The crosspoint switch with
band translation 1000 maximizes signal isolation and minimizes
switching transients by connecting a high isolation switch
configuration to the output of the first LNA 1010a. Band
translation devices 1030a, 1030b having high input impedances are
connected to the outputs of the switch configuration.
[0145] In one embodiment, a low output impedance refers to a
typical magnitude less than 10 ohms differential. In other
embodiments, low impedances may refer to other impedance magnitudes
that may be higher or lower than 10 ohms, and need not be defined
differentially. For example, a low impedance can refer to a
magnitude of substantially less than 33 ohms. In another
embodiment, a high impedance refers to a magnitude of typically
greater than 1 kohm differential. In other embodiments, high
impedances may refer to other impedance magnitudes that may be
higher or lower than 1 kohm, and need not be defined
differentially. For example, in another embodiment, high impedance
can refer to a magnitude of typically greater than 330 ohms. In
general the terms low impedance and high impedance are defined
relative to one another. That is, high impedance is defined to be
greater than or equal to approximately ten times the low impedance
value. Thus, for a low impedance value of 33 ohms, a high impedance
value is greater than approximately 330 ohms.
[0146] The in-phase output of the first LNA 1010a is connected to
switches 1022a and 1024a that selectively switch the signal to the
in-phase inputs of the band translation devices 1030a, 1030b based
on switch control signals provided by, for example, the controller
in the set top box 160a of FIG. 1. In an alternative embodiment, a
microprocessor local to, or integrated with the crosspoint switch
with band translation 1000 can process signals, such as one or more
control messages, from an associated set top box or receiver. The
inverted phase output of the first LNA 1010a is connected to
switches 1026a, 1028a that selectively switch the signal to the
inverted inputs of the band translation devices 1030a, 1030b. A
switch connected to the in-phase output, for example 1022a, is
typically paired with a switch on the inverted output, for example
1026a, such that a differential signal is selectively connected by
the switch pair 1022a, 1026a.
[0147] Thus, the controller in the set top box can direct a first
switch pair 1026a, 1026a to selectively connect the differential
output of the first LNA 1010a to the differential input of the
first band translation device 1030a. A second switch pair 1024a,
1028a selectively connects the differential output of the first LNA
1010a to the second band translation device 1030b.
[0148] The first band translation device 1030a can selectively
frequency translate the signal at its input to an output frequency
band. The first band translation device 1030a uses a signal from a
first Local Oscillator (LO) 1040a to perform the frequency
translation.
[0149] A second signal path is configured similar to the first
signal path. A second LNA 1010b has a differential input 1012b and
a differential output 1014b. The signal at the differential output
1014b of the second LNA 1010b is selectively connected to the first
band translation device 1030a using a third switch pair 1022b,
1026b. The signal at the differential output 1014b of the second
LNA 1010b is selectively connected to the second band translation
device 1030b using a fourth switch pair 1024b, 1028b.
[0150] Typically, the signals from the first LNA 1010a and the
second LNA 1010b are not switched to the same band translation
device, for example 1030a. The output of a single LNA 1010a can be
switched to both band translation devices 1030a, 1030b while the
other LNA signal is not provided to any of the band translation
devices 1030a, 1030b.
[0151] The crosspoint switch with band translation 1000 is
configured to provide high signal isolation between the input
signals and the output signals from the LNA's 1010a and 1010b, and
high isolation through the crosspoint switch section 1022a-1028b.
Additionally, the crosspoint switch with band translation 1000
provides high signal isolation at the input and output of the band
translation devices 1030a and 1030b. Additionally, the crosspoint
switch with band translation 1000 has high signal isolation and low
switching transients. Low switching transients are achieved through
the use of low impedance at the LNA outputs combined with high
impedance inputs at the band translation devices 1030a, 1030b. High
signal isolation is achieved using differential signal
configuration and is also achieved through the use of high
isolation switches.
[0152] High signal isolation typically refers to greater than 30 dB
of isolation. It may be advantageous to achieve a high signal
isolation that is greater than approximately 40 dB. In general,
high signal isolation can refer to greater than 20 dB, 25 dB, 30
dB, 35 dB, 40 dB, 45 dB, 50 dB or some other greater level of
isolation.
[0153] FIGS. 11A-11D are embodiments of high isolation switches.
Each of the switch embodiments of FIGS. 11A-11D are single-ended
configurations. The switch embodiments can be duplicated to allow
switching of in-phase and inverted signals of differential signals.
Thus, a pair of switches from FIGS. 11A-11D can be used as the
switch pairs of FIG. 10.
[0154] FIG. 11A is a first switch embodiment having a single
transistor 1102 controlled to selectively connect a signal from its
input to its output based on the signal applied to the control
input. The transistor 1102 can be controlled to selectively isolate
a signal at its input from its output based on the signal applied
to its control input. Signal isolation is controlled by the ability
of the transistor 1102 to isolate the input from the output. A pair
of transistors 1102 can be used to switch differential signals.
[0155] FIG. 11B is a second switch embodiment. A signal is input at
the base of a first transistor 1110 configured as an emitter
follower. Additionally, a bias voltage, which is typically a DC
bias voltage, is applied to the base of the first transistor 1110.
The emitter of the first transistor 1110 is selectively biased with
a controllable current source 1112. The first transistor 1110
selectively couples a signal from its base to its emitter when the
controllable current source 1112 conducts. Conversely, a signal at
the base of the first transistor 110 is isolated from the emitter
when the controllable current source 1112 is off. A pull up device
1114 connects the emitter of the first transistor 1110 to a voltage
that is greater than the bias voltage, for example (V.sub.b+1V) to
ensure the first transistor 1110 is cut off when the controllable
current source 1112 is off.
[0156] FIG. 11C is a third switch embodiment having multiple
transistors configured to provide increased signal isolation. A
signal is provided to a first transistor 1120. The output of the
first transistor 1120 is connected to an input of a second
transistor 1122. The output of the second transistor 1122 is the
output of the switch. A third transistor 1124 is connected to the
output of the first transistor 1120 and is configured to
selectively couple the output of the first transistor 1120 and
input of the second transistor 1122 to ground or signal return.
[0157] A differential control signal is used to control the third
switch embodiment. An in-phase control signal controls the first
transistor 1120 and second transistor 1122. An inverted control
signal controls the third transistor 1124. Thus, when the first and
second transistors 1120, 1122 are controlled to be conducting, the
third transistor 1124 is controlled to be cut off. Conversely, when
the first and second transistors 1120, 1122 are controlled to be
cut off, the third transistor 1124 is controlled to be
conducting.
[0158] FIG. 11D is a fourth switch embodiment. The fourth switch
embodiment is similar to the second switch embodiment with
additional transistors configured to provide additional signal
isolation.
[0159] A signal is input at the base of a first transistor 1130
configured as an emitter follower. Additionally, a bias voltage,
Vb, which is typically a DC bias voltage, is applied to the base of
the first transistor 1130. The emitter of the first transistor 1130
is selectively biased with a controllable current source 1132. The
first transistor 1130 selectively couples a signal from its base to
its emitter when the controllable current source 1132 conducts.
Conversely, a signal at the base of the first transistor 1130 is
isolated from the emitter when the controllable current source 1132
is off.
[0160] A second transistor 1134 is configured to selectively pull
up the emitter of the first transistor 1130 to a voltage that is
greater than the bias voltage, for example (V.sub.b+1V), to ensure
the first transistor 1130 is cut off when the controllable current
source 1132 is off. Additionally, the second transistor 1134 can
also shunt any signal leakage at the emitter node to AC ground via
the bias point, thus improving signal isolation. A third transistor
1136 has an input connected to the emitter of the first transistor
1130 and an output that is the output of the switch. The third
transistor 1136 is selectively controlled to couple the signal from
the emitter of the first transistor 1130 to the switch output when
the controllable current source 1132 is conducting. The third
transistor 1136 is selectively controlled to isolate the signal
from the emitter of the first transistor 1130 when the controllable
current source is off.
[0161] FIG. 12 is a functional block diagram of a crosspoint switch
with band translation 1200 that can also be integrated as a portion
of the receiver 180 of FIG. 1. A two input and two output version
of the receiver 180 of FIG. 1 can be implemented with the
crosspoint switch with band translation 1200 of FIG. 12 in
combination with two LNB's.
[0162] The crosspoint switch with band translation 1200 is similar
to the crosspoint switch with band translation 1000 of FIG. 10 with
the exception that the device of FIG. 12 uses current mode
switching while the device of FIG. 10 uses voltage mode switching.
Thus, the crosspoint switch with band translation 1200 can be used
interchangeably with the device of FIG. 10. However, in some
instances, current mode switching can be advantageous because of
the ability to sum currents into a common node.
[0163] The crosspoint switch with band translation 1200 has a first
signal path and a second signal path. The first signal path
includes a first LNA 1210a connected to a pair of transconductance
devices, 1222a and 1224a that can selectively route a signal at the
output 1214a of the LNA 1210a to a first band translation device
1230a or a second band translation device 1230b. The crosspoint
switch with band translation 1200 uses the transconductance
devices, for example 1222a and 1222b, to provide current-mode
switching of the signals.
[0164] The first LNA 1210a is configured with a differential input
1212a and a differential output 1214a. The differential input 1212a
of the first LNA 1210a can be matched to 75 ohm differential. The
differential output 1214a of the first LNA 1210a is configured to
have a low impedance. The crosspoint switch with band translation
1200 maximizes signal isolation and minimizes switching transients
by connecting high isolation transconductance devices, 1222a and
1224a, to the output of the first LNA 1210a. Band translation
devices 1230a, 1230b having low input impedances are connected to
the outputs of the transconductance devices 1222a and 1224a.
[0165] The differential output 1214a of the first LNA 1210a is
connected to the high impedance differential inputs of the
transconductance devices 1222a and 1224a. The first LNA 1210a can
drive both transconductance devices 1222a and 1224a because the
differential inputs of the transconductance devices 1222a and 1224a
are high impedance.
[0166] Each of the transconductance devices 1222a and 1224a
includes a control input, 1223a and 1225a respectively, that is
used to switch the transconductance device 1222a and 1224a on or
off. When the signal from the first LNA 1210a is to be routed to
the first band translation device 1230a, the first transconductance
device 1222a is controlled to provide a current output to the input
of the first and translation device 1230a. Similarly, the second
transconductance device 1224a can be controlled to provide a
current output to the input of the second band translation device
1230b. One or more transconductance devices, for example 1222a and
1224a connected to an LNA 1210a can simultaneously be enabled such
that one input, for example a signal at 1212a, can be routed to all
band translation devices 1230a and 1230b.
[0167] The first band translation device 1230a can selectively
frequency translate the signal at its input to an output frequency
band. The first band translation device 1230a uses a signal from a
first LO 1240a to perform the frequency translation. The first band
translation device 1230a has a low impedance input and thus,
operates as a current summing node for the currents from the
transconductance devices 1222a and 1222b to which its input is
connected.
[0168] A second signal path is configured similar to the first
signal path. A second LNA 1210b has a differential input 1212b and
a differential output 1214b. The signal at the differential output
1214b of the second LNA 1210b is selectively connected to the first
band translation device 1230a using a third transconductance device
1222b. The signal at the differential output 1214b of the second
LNA 1210b is selectively connected to the second band translation
device 1230b using a fourth transconductance device 1224b. The
second band translation device 1230b operates in conjunction with a
second LO 1240b.
[0169] The transconductance devices 1222a, 1222b, 1224a, and 1224b
can be any type of transconductance devices, such as transistors,
FETs, and the like. The transconductance devices 1222a, 1222b,
1224a, and 1224b have a high output impedance. Thus, multiple
transconductance devices, for example 1222a and 1222b can
selectively provide a signal to the same band translation device
1230a without the output impedance of the first transconductance
device 1222a affecting the performance of the other
transconductance device 1222b. The low input impedance band
translation device 1230a operates as a current summing node.
[0170] In an alternative embodiment of the crosspoint switch with
band translation 1200, the LNA's 1210a and 1210b are omitted and
the input signals are directly coupled to the inputs of the
transconductance devices 1222a, 1222b, 1224a, and 1224b. The inputs
to the first and second signal paths can be matched to a
predetermined impedance using a matching circuit (not shown) which
can be as simple as a resistor placed across the differential
inputs.
[0171] FIG. 13 is a functional block diagram of a crosspoint switch
with band translation 1300 having LNA/band translation device pairs
for each input/output combination and summing the outputs of the
band translation devices in the current domain. As with the
crosspoint switch with band translation devices of FIGS. 10 and 12,
the crosspoint switch with band translation 1300 can be combined
with LNBs in the receiver 180 of FIG. 1. The devices in the
crosspoint switch with band translation 1300 utilize differential
signals to minimize noise, but single-ended devices can be used in
other embodiments.
[0172] Each LNA/band translation pair can selectively provide a
signal to an output or be controlled to isolate the signal at the
input from the output. The LNA can be selectively controlled to
isolate the signal by removing the bias, or by reversing the bias
on the amplifier. For example, the controller in the set top box
160a of FIG. 1 can receive user input and control the bias control
pins, labeled A, B, C, and D, to selectively enable or disable the
bias to the LNAs 1310a-b, 1320a-b.
[0173] A first LNA/band translation device pair includes a first
LNA 1310a connected to a first input 1312a. The first LNA 1310a is
controlled to selectively amplify or isolate the input signal based
on a signal provided to its control input 1314a. The output of the
first LNA 1310a is connected to a first band translation device
1332 having a high output impedance. The output of the first band
translation device 1332 is connected to a first signal output
1340a.
[0174] A second LNA/band translation device pair includes a second
LNA 1320a having an input connected to the first input 1312a. The
controller in the set top box can control the control input 1324a
of the second LNA 1320a to selectively amplify or isolate the input
signal. The output of the second LNA 1320a is connected to a second
band translation device 1334 having a high output impedance. The
output of the second band translation device 1334 is connected to a
second signal output 1340b.
[0175] Thus, in order to selectively route a signal from the first
input 1312a to the first signal output 1340a, the controller in the
set top box selectively controls the first LNA 1310a to amplify the
input signal by providing an enable signal to the control input,
1314a, on the first LNA 1310a. In order to isolate a signal at the
first input 1312a from the first output 1340a, the first LNA 1310a
is selectively controlled to isolate the signal.
[0176] A second differential input 1312b is connected to the inputs
of a third LNA 1310b and a fourth LNA 1320b. The third LNA 1310b is
controlled to selectively amplify or isolate the input signal based
on a signal provided to its control input 1314b. The output of the
third LNA 1310b is connected to a third band translation device
1336 having a high output impedance. The output of the third band
translation device 1336 is connected to a first signal output
1340a.
[0177] Similarly, the fourth LNA 1320b is controlled to selectively
amplify or isolate the input signal based on a signal provided to
its control input 1324b. The output of the fourth LNA 1320b is
connected to a fourth band translation device 1338 having a high
output impedance. The output of the fourth band translation device
1338 is connected to a first signal output 1340b.
[0178] Thus, a signal provided to the second differential input
312b can selectively be routed to the first or second signal
outputs, 1340a or 1340b or simultaneously to both signal outputs.
In order to route the signal from the second input 1312b to the
first signal output 1340a, a control signal is provided to the
control input 1314b of the third LNA 1310b to enable the third LNA
1310b to amplify the second input signal. In order to route the
signal from the second input 1312b to the second signal output
1340b, a control signal is provided to the control input 1324b of
the fourth LNA 1320b to enable the fourth LNA 1320b to amplify the
second input signal.
[0179] The outputs of the first and third band translation devices
1332, 1336 can be summed at the load if both signals are routed to
the first signal output 1340a. Similarly, the outputs of the second
and fourth band translation devices 1334 and 1338 can be summed at
the load if both provide signals to the second signal output 1340b.
Thus, by using current outputs from high impedance devices driving
matched impedance loads, multiple signals can be summed in a common
node.
[0180] FIG. 14 is another embodiment of a 2.times.2 crosspoint
switch with band translation 1400. The specific embodiment is
optimized for implementation within a single integrated circuit
having impedance matched inputs and outputs. It is evident that the
number of inputs or outputs can be expanded to any other number.
The embodiment uses current mode switching. LNA's having a matched
input, variable gain, and a low impedance output are used. Signals
at a first input 1412a can be routed, using first and second
transconductance devices, to one or both outputs 1470a and 1470b.
Similarly, signals at a second input 1412b can be routed, using
third and fourth transconductance devices, to one or both outputs
1470a and 1470b.
[0181] The 2.times.2 crosspoint switch with band translation 1400
receives the input signal at a matched signal input of the low
noise amplifiers. The low noise amplifiers generate intermediate
signals at their low impedance outputs. The intermediate signals
are provided to high impedance inputs of current sources configured
as transconductance devices. A controller can selectively control
the transconductance devices to provide an output current based in
part on the intermediate signal. Additionally, the controller can
selectively enable or disable each of the transconductance devices.
For example, the bias to each of the transconductance device may be
controllable to selectively enable or disable the device.
Alternatively, the bias current may be varied linearly to control
the gain of the transconductance devices. Alternatively, the gain
may be varied via other means and the transconductor may be enabled
and disabled by other means.
[0182] The current output of the transconductance devices can then
be received at low impedance inputs of band translation devices
that can frequency translate the current signals from a first
frequency band to a second frequency band. The band translation
devices can have matched impedance outputs.
[0183] A first signal path is configured to amplify, band
translate, and route a first signal to one of two outputs. A first
LNA 1410a has a differential input 1412a configured to accept the
first signal. The input 1412a of the first LNA 1410a can be a
differential input that is matched to a predetermined impedance,
such as 75 .OMEGA. or 50 .OMEGA.. The differential output of the
first LNA 1410a has an in-phase output 1414a and an inverted output
1416a. The differential output of the first LNA 1410a can be a low
output impedance, a matched output impedance, or a high output
impedance. The output impedance of the first LNA 1410a can be, for
example, 200 ohms differential.
[0184] The in-phase output 1414a of the first LNA 1410a is
connected to a first emitter follower 1422a that has a low output
impedance. The in-phase output 1414a of the first LNA is connected
to the base of the first emitter follower 1422a. The emitter of the
first emitter follower 1424a is connected to a current source 1424a
that biases the first emitter follower 1424a. The output of the
first emitter follower 1424a is connected to the in-phase inputs of
the differential inputs to first and second transconductance
devices. The transconductance devices have high input impedances.
The transconductance devices can be bipolar devices that can be
selectively enabled or disabled by controlling the bias
currents.
[0185] Similarly, the inverted output 1416a of the first LNA is
connected to the input of a second emitter follower 1426a. The
second emitter follower 1426a is biased using a current source
1428a connected to its emitter. The output of the second emitter
follower 1426a is connected to the inverted inputs of the first and
second transconductance devices.
[0186] Alternatively, the first and second emitter followers, 1422a
and 1426a, with their associated current sources, 1424a and 1428a,
can be considered the low impedance output stage of the first LNA
1410a.
[0187] The first transconductance device includes a first
transistor 1432a with the base of the first transistor 1432a
serving as the in-phase input of the first transconductance device.
A first resistor 1433a connects the emitter of the first transistor
1432a to a controllable current source 1438a. The base of a second
transistor 1434a is used as the inverted input of the first
transconductance device. A second resistor 1435a connects the
emitter of the second transistor 1434a to the controllable current
source 1438a.
[0188] The controllable current source 1438a provides the bias for
the transistors, 1432a and 1434a of the first transconductance
device. The controllable current source 1438a can be selectively
enabled or disabled based on a control signal. The first
transconductance device isolates a signal at its input from its
output when the controllable current source 1438a is disabled, and
conversely, provides a current output that can be proportional to
the input signal when the controllable current source 1438a is
enabled.
[0189] A first differential buffer amplifier having two transistors
1452a and 1454a is used to sum the currents from multiple
transconductance devices and provide a differential signal to the
first band translation device 1460a.
[0190] The first band translation device 1460a is configured with a
low input impedance and an output impedance matched to a
predetermined impedance. For example, the output of the first band
translation device 1460a can be matched to 75 .OMEGA.. The
differential output of the first band translation device 1460a is
connected to the first signal output 1470a. The first band
translation device 1460a is driven with a first LO 1462a. The first
LO 1462a frequency can be tunable to allow the frequency
translation of the first band translation device 1462a to be tuned.
Alternatively the output frequency of the first LO 1462a can be
fixed. The first band translation device 1462a can be configured to
frequency translate the signal or to pass the signal without
frequency translation.
[0191] The first LNA 1410a also provides a signal that can be
selectively routed to a second output 1470b. The differential
outputs from the first and second emitter followers, 1422a and
1426a are connected to the differential inputs of a second
transconductance device.
[0192] The base of a first transistor 1442a in the second
transconductance device is connected to the in-phase output from
the first emitter follower 1422a. The base of a second transistor
1444a in the second transconductance device is connected to the
inverted output from the second emitter follower 1426a. Resistors
1443a and 1445a connect the emitters of the first and second
transistors 1442a and 1444a to a controllable current source 1448a
that selectively provides bias to the first and second transistors
1442a and 1444a. The second transconductance device provides an
output current when the controllable current source 1448a is
enabled. Conversely, the second transconductance device does not
provide an output current when the controllable current source
1448a is disabled.
[0193] The differential output from the second transconductance
device is connected to the differential input of a second
differential buffer amplifier. The second differential buffer
amplifier includes two transistors 1452b and 1454b and is used to
sum the currents from multiple transconductance devices and provide
a differential signal to the second band translation device
1460b.
[0194] The output of the second differential buffer amplifier is
connected to the differential input of a second band translation
device 1460b. The second band translation device 1460b has with a
low input impedance and an output impedance matched to a
predetermined impedance such as 75 .OMEGA.. The differential output
of the second band translation device 1460b is connected to the
second signal output 1470b. The second band translation device
1460b is driven with a second LO 1462b. The second LO 1462b
frequency can be tunable to allow the frequency translation of the
second band translation device 1462b to be tuned. Alternatively the
output frequency of the second LO 1462b can be fixed. The second
band translation device 1462b can be configured to frequency
translate the signal or to pass the signal without frequency
translation.
[0195] The second signal input 1412b is connected to the second LNA
1410b and through third and fourth transconductance devices to the
first and second differential buffer amplifiers in a configuration
that is similar to the path from the first signal input 1412a to
the differential buffer amplifiers.
[0196] The second signal input 1412b is connected to the input of
the second LNA 1410b. The differential output of the second LNA is
connected to a pair of emitter followers, one emitter follower for
each of the signal outputs of the second LNA 1410b.
[0197] The in-phase LNA output 1414b is connected to a first
emitter follower 1422b that includes a first current source 1424b
connected to its emitter to provide a bias. The inverted LNA output
1416b is connected to a second emitter follower 1426b that includes
a second current source 1428b connected to its emitter to provide a
bias.
[0198] The output of the first emitter follower 1422b is connected
to the in-phase inputs of third and fourth transconductance
devices. The output of the second emitter follower 1426b is
connected to the inverted inputs of the third and fourth
transconductance devices.
[0199] The third transconductance device includes first and second
transistors 1432b and 1434b arranged in a differential
configuration. The base of the first transistor 1432b is the
in-phase input of the transconductance device and the base of the
second transistor 1434b is the inverted input of the third
transconductance device. The emitters of the first and second
transistors, 1432b and 1434b, are connected via first and second
resistors, 1433b and 1435b, to a controllable current source 1438b.
The controllable current source selectively enables or disables the
third transconductance device. The collectors of the first and
second transistors, 1432b and 1434b, are connected to the
differential inputs of the first differential buffer amplifier.
[0200] Similarly, the fourth transconductance device includes first
and second transistors 1442b and 1444b arranged in a differential
configuration. The base of the first transistor 1442b is the
in-phase input of the transconductance device and the base of the
second transistor 1444b is the inverted input of the fourth
transconductance device. The emitters of the first and second
transistors, 1442b and 1444b, are connected via first and second
resistors, 1443b and 1445b, to a controllable current source 1448b.
The controllable current source 1448b selectively enables or
disables the fourth transconductance device. The collectors of the
first and second transistors, 1442b and 1444b, are connected to the
differential inputs of the second differential buffer amplifier. Of
course, the transconductance devices shown in FIG. 6 only represent
embodiments of typical transconductance devices. Other embodiments
of transconductance devices may be used in other embodiments.
[0201] Thus, a designer can choose between many buffer amplifier
embodiments, crosspoint switch embodiments, and band translation
device embodiments in designing a band translation switch. FIG. 15
is a functional block diagram of a specific embodiment of signal
distribution system 1500 including an integrated crosspoint switch
with band translation (band translation switch) 1510 and external
components. The band translation switch 1510 includes four inputs
for LNB's, a cascadable output corresponding to each of the inputs,
and two outputs configured to interface with set top boxes. The
band translation switch 1510 is configured to interface with LNB's
signals having a dual band-stacked frequency plan. The dual
band-stacked frequency plan includes an upper band block and a
lower band block. The band translation switch outputs maintain the
dual band-stacked frequency plan, but allow the upper or lower band
block from any of the LNB signals to be configured as an output
upper band block. Similarly, the upper or low band block from any
of the LNB signals can be configured as an output lower band block.
A more detailed description of the band translation switch 1510 is
provided below.
[0202] The band translation switch 1510 includes four inputs
configured to interface with up to four LNB's. Each LNB provides a
signal conforming to a dual band-stacking frequency plan having an
upper band block and a lower band block. For example, the LNB
signals can be satellite downlink signals from selected transponder
groups. The lower band block can be 950-1450 MHz and the upper band
block can be 1650-2150 MHz.
[0203] Each of the signal inputs is connected to the input of an
amplifier 1520a-1520d. The amplifiers 1520a-1520d are configured as
Low Noise Amplifiers (LNA's) that both buffer and amplify the input
signals from the LNB's. The output from each of the amplifiers
1520a-1520d is connected to a corresponding input on a crosspoint
switch 1530. Additionally, the output from each of the amplifiers
1520a-1520d is connected to a corresponding cascade output of the
band translation switch 1510.
[0204] The crosspoint switch 1530 is configured as a 4.times.4
switch. Any of the four amplified LNB input signals can be
selectively routed to any of the four outputs of the crosspoint
switch 1530 independently and simultaneously. For example, the
crosspoint switch 1530 can include a two-bit control for each
output. The value of the two-bit control can be programmed to
selectively route the signal from one of the four inputs. The band
translation switch 1510 can, for example, receive the two bit
control words from a set top box. Alternatively, the set top box
may send one or more control messages to a microprocessor
implemented local to the crosspoint switch and the microprocessor
can generate the one or more two bit control words. In the
embodiment shown in FIG. 15, each of the four outputs from the
crosspoint switch 1530 is connected to a band translation device
1540a-1540d. One or more outputs from the crosspoint switch 1530
can be coupled to the same band translation device, for example
1540a.
[0205] The band translation devices 1540a-1540d are configured to
selectively frequency translate the signals or to pass the signals
without frequency translation. Each of the band translation devices
1540a-1540d can select frequency translation or pass through
independently of the other devices. Because a dual band-stacked
frequency plan is used in this embodiment, the band translation
devices 1540a-1540d are configured to swap the positions of the
upper and lower band blocks when frequency translation is
selected.
[0206] Each of the band translation devices 1540a-1540d includes a
mixer. The band translation switch 1510 also includes one or more
local oscillators (LO). In one embodiment with a dual band-stacked
frequency plan, a single LO can be routed to all of the band
translation devices 1540a-1540d. In another embodiment, the local
oscillator frequency can be fixed when a dual band-stacked
frequency plan is implemented. An LO frequency of 3.1 GHz, or
2.times. (the band center mean), can be used to perform the
frequency translation.
[0207] In another embodiment, a plurality of variable frequency LOs
can be used with the band translation devices 1540a-1540d. For
example, each of the band translation devices 1540a-1540d can have
a separate independently controlled LO output frequencies. Thus,
each of the band translation devices 1540a-1540d can frequency
translate its input signal independently of the frequency
translation performed by any other band translation device.
[0208] LO buffer amplifiers (not shown) distribute the signal from
the LO output to each of the band translation devices 1540a-1540d.
The output of the band translation devices 1540a-1540d are
connected to the outputs of the band translation switch 1510.
[0209] Each of the outputs of the band translation switch 1510 is a
dual band-stacked signal. Each of the outputs of the band
translation switch 1510 is connected to a filter 1550a-1550d. The
filters 1550a-1550d are configured to pass signals in one of the
predetermined frequency bands in the dual band-stacked frequency
plan. The filters 1550a-1550d reject signals outside of the
passband, including the signals at the undesired frequency band.
The filters 1550a-1550d can be configured with tunable passbands or
can be configured to have fixed passbands.
[0210] In the present embodiment, the filters 1550a-1550d are
configured as bandpass filters with fixed passbands. The first
filter 1550a is configured as a bandpass filter that passes the
upper band block of the frequency plan. A second filter 1550b is
configured as a bandpass filter that passes the lower band block.
Similarly, a third filter 1550c is configured to pass the upper
band block and a fourth filter 1550d is configured to pass the
lower band block. The outputs of the first and second filters
1550a-1550b are connected to respective first and second inputs of
a first signal combiner 1560a. Similarly, the outputs of the third
and fourth filters 1550c-1550d are connected to first and second
inputs of a second signal combiner 1560b. The filters 1550a-1550d
are not limited to bandpass filters, but can be, for example,
bandpass filters (BPF), lowpass filters (LPF) or highpass filters
(HPF). In other embodiments, other frequency selective devices can
be used to limit the frequency response of the outputs. The filters
1550a-1550d can have passbands that are narrower than the frequency
bandwidth of the input signals. For example, an input to a filter,
for example 1550a, can include multiple carriers. However, the
filter 1550a can be configured to pass a subset of all of the
carriers.
[0211] The signal combiners 1560a-1560b are configured to sum the
signals provided at their inputs and to provide the summed signal
at an output. The outputs from the signal combiners 1560a-1560b are
the band translated outputs of the signal distribution system 1500.
Each of the outputs is connected to a set top box for further
processing and for distribution to an end user device.
[0212] As discussed above, one or more frequency selective devices
can be used as the filters 1550a-1550d. For example, a diplexer can
be used to filter and to band-stack signals. The diplexer can be
used as the filters, for example 1550a and 1550b, and signal
combiner 1560a.
[0213] Of course, the band translation switch 1510 is not limited
to operating with band stacked input signals. For example, each of
the LNB's can provide signals in the same frequency band. The band
translation switch 1510 can be configured to frequency translate
and combine portions of the single band input signals. The
crosspoint switch 1530 can, for example, route the output of the
first amplifier 1520a to a first band translation device 1540a. An
LO in the first band translation device 1540a can be configured to
frequency translate the signal such that one or more channels from
the input signal are translated to desired output frequencies. The
first filter 1550a can be configured to pass only those desired
channels and reject all undesired frequencies and channels.
[0214] Similarly, the crosspoint switch 1530 can be configured to
route the output of the second amplifier 1520b to a second band
translation device 1540b. The second band translation device 1540b
can be configured to frequency translate a portion of the input
signal to desired output frequencies. The second filter 1550b can
be configured to pass only those desired channels and reject all
undesired frequencies and channels.
[0215] The first and second band translation devices 1540a-1540b in
conjunction with the first and second filters 1550a-1550b can be
configured to produce selected channels in mutually exclusive
frequency bands. The combiner 1560a can then sum the filtered
outputs to produce a composite output signal from independent
single band input signals, where each filter includes one or more
channels. In one embodiment of the single band input signal
configuration, each band translation device and filter pair, for
example 1540a and 1550a, is configured to frequency translate one
or more channels from each of one or more input signal bands. The
frequency translated signals can be combined into a single band
signal or a multiple band signal.
[0216] Similarly, some embodiments can have multiple band
translation devices and multiple filters. Each of the multiple band
translation devices can frequency translate one or more channels
from one or more input bands. The outputs of the multiple filters
can be summed to provide a single composite signal having a desired
channel line up.
[0217] FIG. 16 is a functional block diagram of a signal
distribution system 1600 that is typical of a satellite television
system that can be implemented at a residence or other building.
The signal distribution system 1600 includes an antenna 1620 having
antenna feeds 1622, 1624 coupled to two inputs of a low noise block
1626. The outputs of the low noise block 1626 are coupled to two
inputs of a distribution switch 1630. The distribution switch
outputs are connected to first, second, and third set top boxes
1652, 1654, 1656, using first and second transmission lines 1642,
1644. The output of the first set top box 1652 is connected to a
first output device 1662. A signal splitter 1670 splits into two
signals the signal coupled from the distribution switch 1630 by the
second transmission line 1644. A first signal splitter 1670 output
is coupled to the second set top box 1654 and the second signal
splitter 1670 output is coupled to the third set top box 1656. The
output of the second set top box 1654 is connected to a second
output device 1664 and the output of the third set top box 1656 is
connected to a third output device 1666.
[0218] The antenna 1620 includes two antenna feeds 1622, 1624.
However, multiple antennae can be used. Additionally, each antenna
1620 can have one or more antenna feeds 1622, 1624, and each
antenna 1620 is not limited to having only two feeds 1622, 1624.
Alternatively, the antenna 1620 can be a configuration that does
not utilize an antenna feed, such as a whip or horn.
[0219] The antenna 1620 receives one or more signals from a
satellite 1610. Additionally, the satellite 1610 can provide a
signal of a particular polarization and modulation type. Again,
there may be more than one satellite 1610 providing signals to the
antenna 1620. The signals from a particular satellite 1610 can be
in the same frequency band as signals from another satellite (not
shown) or can be in distinct frequency bands. The signals from
multiple satellites can each have the same polarity and modulation
type or can be different from each other.
[0220] In the signal distribution system 1600 of FIG. 16, each of
the antenna feeds 1622, 1624 is connected to an independent input
of a low noise block 1626 that outputs signals to the distribution
switch 1630. Of course, the distribution switch 1630 is not limited
to a 2.times.2 switch but can have any number of input ports and
output ports, for example, the distribution switch 1630 can be, for
example, a 2.times.4 switch, a 4.times.4 switch, or some other
switch arrangement.
[0221] The distribution switch 1630 is configured to process the
received satellite signals. The distribution switch 1630 can, for
example, amplify, filter, and frequency downconvert the received
satellite signals. The distribution switch 1630 can be configured
as a pair of low noise block converters (LNB's) that each block
convert the signals from one of the distribution switch 1630 inputs
to an intermediate frequency. The distribution switch 1630 can also
be configured to allow each of the input signals provided to the
inputs to be connected to any one of multiple switch outputs. Thus,
the signal provided from the first antenna feed 1622 can be block
converted in the distribution switch 1630 and routed to any of the
switch outputs. Similarly, the signal provided from the second
antenna feed 1624 can be block converted in the distribution switch
1630 and routed to any of the switch outputs. Typically, the
distribution switch 1630 is configured such that the signals from
only one signal source are routed to a particular switch output.
Alternatively, one or more of the block converted signals can be
routed to the same distribution switch 1630 output.
[0222] The outputs of the distribution switch 1630 can be connected
to remote locations using cabling when the antenna 1620 and
distribution switch 1630 are installed in a geographically remote
location from the desired signal destinations. The outputs of the
distribution switch 1630 are typically routed to remote
destinations with transmission lines, which can be coaxial cables.
The distribution switch 1630 can be positioned local to the low
noise block 1626 and antenna feeds 1622 and 1624, or may be
positioned remote from the low noise block 1626 and antenna feeds
1622 and 1624.
[0223] In one embodiment, the distribution switch 1630 is
co-located with the antenna 1620, low noise block 1626, and antenna
feeds 1622 and 1624. In another embodiment, the distribution switch
1630 can be located remote from the antenna 1620. For example,
cables or transmission lines can couple the signals from the low
noise block 1626 to a distribution switch 1630 positioned inside a
structure near one or more set top boxes 1652 and 1654. Similarly,
in other embodiments, the distribution switch 1630 can be
positioned in an intermediate location between the antenna 1620 and
the set top boxes 1652 and 1654. In some embodiments, the low noise
block 1626 is omitted and signals from the antenna feeds 1622 and
1624 can be coupled to the distribution switch 1630 using cables.
Similarly, output signals from the distribution switch 1630 can be
coupled to set top boxes or other destination devices using cables
or some other distribution system.
[0224] In a first embodiment, the distribution switch is positioned
local to the low noise block 1626 and antenna 1620. A first
transmission line 1642 distributes the signal from the first output
port of the distribution switch 1630 to a remote location within
the signal distribution system 1600. The end of the first
transmission line 1642 is connected to a first set top box 1652
located remote from the distribution switch 1630.
[0225] A second transmission line 1644 distributes the signal from
the second output port of the distribution switch 1630 to a signal
splitter 1670. A first output of the signal splitter 1670 is
coupled to the second set top box 1654. The second set top box 1654
can be located at a location remote from the distribution switch
1630 and signal splitter 1670 and can also be at a location remote
from the first set top box 1652. A second output of the signal
splitter 1670 is coupled to a third set top box 1656. The output of
the third set top box 1656 is coupled to a third output device
1666.
[0226] The first and second transmission lines, 1642 and 1644, can
be parallel lines, twisted pairs, coaxial line, waveguide, and the
like, or any other means for distributing the signal. Additionally,
although transmission lines are typically used to minimize signal
loss and signal reflections, the system can use other means for
distributing the signal that are not transmission lines. For
example, wires, wire bundles, and the like, can be used for
distributing the signals from the distribution switch 1630 to the
set top boxes 1652, 1654. However, for signals that can be
considered Radio Frequency (RF) signals, the signals are typically
distributed using transmission lines. The RF information signals
can, for example, be in the range of KHz up to several GHz. Of
course, the signal distribution system 1600 is not limited to
distributing RF signals, but can distribute other signals, such as
baseband signals or optical signals.
[0227] The transmission lines 1642, 1644, are typically non-ideal
passive devices. Thus, the transmission lines attenuate the signal
power. However, the attenuation contributed by the transmission
lines 1642, 1644 typically do not attenuate the noise power to the
same degree as the signal power. For example, a passive attenuator,
such as a length of transmission line may not significantly degrade
the thermal noise. Additionally, the transmission lines 1642, 1644
can contribute other types of cable related signal degradation. For
example, the transmission lines can affect flatness, tilt, phase
distortion, group delay distortion, reflection, interference, noise
pick-up and microphonic noise of the distributed signals. Thus, the
losses contributed by the transmission lines 1642, 1644 typically
degrade the SNR of the signal distributed to the set top boxes
1652, 1654.
[0228] The first and second transmission lines 1642 and 1644 are
coupled to corresponding inputs of set top boxes 1652, 1654, and
1656. The second transmission line 1644 couples to the second and
third set top boxes, 1654 and 1656, via the signal splitter 1670.
In one embodiment, the frequency bands for the signals output from
the distribution switch 1630 do not correspond to frequency bands
used by the output devices 1662 and 1664. Thus, the set top boxes
1652, 1654 can further frequency translate the signals to operating
bands compatible with the output devices 1662, 1664, and 1666.
Additionally, the output signals from the distribution switch 1630
can be in a format that is not compatible with the format used by
the output devices 1662, 1664, and 1666. The set top boxes 1652,
1654, and 1656 can then function as signal processing stages. For
example, the satellite downlink signals can be digitally modulated
in a format that is not compatible with the output devices 1662,
1664, and 1666 which can be typical television receivers. The set
top boxes 1652, 1654, and 1656 can be configured to demodulate the
digitally modulated signals, process the demodulated signals, and
then modulate television channel carrier frequencies with the
signals for delivery to the television output devices 1662, 1664,
and 1666.
[0229] Alternatively, if the signals output from the distribution
switch 1630 are in a format and are at a frequency band that is
compatible with the output devices 1662, 1664, and 1666 the set top
boxes 1652, 1654, and 1656 may not be required. In still another
alternative, one or more of the functions performed by the set top
boxes 1652, 1654, and 1656 can be integrated into the output
devices 1662, 1664, 1666. In still another embodiment, the signal
splitter 1670 can be configured to perform signal processing, such
as frequency conversion or demodulation.
[0230] In the embodiment described in FIG. 16, each of the set top
boxes 1652, 1654, and 1656 is connected to a single output device
1662, 1664, and 1666. However, more than one output device e.g.
1662, 1664 can be connected to the output from a single set top
box, for example 1652. Alternatively, outputs from more than one
set top box 1652, 1654, and 1656 can be combined or otherwise
connected to a single output device, for example 1662, although
such a configuration is not typical.
[0231] An output device, for example 1662, can be configured to
tune to a particular channel within the one or more frequency bands
provided by the set top box, such as 1652. The output device 1662
can process the signal from the selected channel to present some
media content, such as video or audio, to a user.
[0232] For example, the output devices, 1662, 1664, and 1666 can be
television receivers and can display a television signal
corresponding to a signal transmitted by the satellite 1610. The
output devices 1662, 1664, and 1666 can be other types of devices
in other signal distribution systems. For example the output
devices 1662, 1664, and 1666 can be telephones, radio receivers,
computers, networked devices, and the like, or other means for
outputting a signal.
[0233] The output devices 1662, 1664, and 1666 can have a range of
signal quality over which the output is considered acceptable. For
example, the output devices 1662, 1664, and 1666 can provide
acceptable outputs for input SNR above a predetermined level, which
may represent a desired minimum SNR. However, the SNR at the input
to the output devices, 1662, 1664, and 1666 is typically determined
by the signal processing performed in the set top boxes 1652, 1654
and 1656. Thus, the signal quality is typically related to the
signal quality at the input of the set top boxes 1652, 1654, and
1656. Thus, the signal distribution system 1600 is typically
configured to provide a signal at the input to the set top boxes
1652, 1654, and 1656 having a SNR greater than the desired
minimum.
[0234] Although FIG. 16 is a functional block diagram of a
satellite signal distribution system, other signal distribution
systems have similar structures. For example, cable distribution
systems, which may distribute television, radio, data, and/or
telephony signals, typically provide a single access point to a
geographic area, such as a residence. The signal from the one
access point is then typically split, amplified, distributed, and
can be combined with other signals, such as, for example the
satellite television signals. Communication systems having wireless
communication links can also have similar structures. For example,
a terrestrial television or radio system can include a single
antenna and distribute the signals received at the single antenna
to multiple output devices using a signal distribution system 100
that can amplify, split, distribute, and/or combine the received
signals.
[0235] The signal distribution system is not limited to a
residence, but can span many residences, businesses, or locations
not associated with dwellings or buildings. The signal distribution
system is characterized by its features and is not limited to any
particular application.
[0236] Additionally, although FIG. 16 shows only the signal
splitter 1670 interposed between the distribution switch 1630 and
set top boxes 1654 and 1656, elements other than the transmission
lines 1642, 1644, and signal splitter 1670 can be interposed
between the distribution switch 1630 and the set top boxes 1652,
1654, and 1656. The additional distribution devices can include
active or passive power dividers, active or passive power
combiners, amplifiers, attenuators, filters, switches, crosspoint
switches, multiplexers, de-multiplexers, frequency translation
devices, encoders, decoders, and the like or any other means for
distributing a signal. Each of these additional signal distribution
devices can contribute to the noise experienced by the distributed
signal.
[0237] For example, a two-way passive power divider allows a signal
at one input to be split equally into two output signals, each
having half the original signal power, while maintaining an
impedance match at all ports. An ideal two-way passive divider
reduces the SNR by 3 dB. However, in practice, the degradation is
often higher.
[0238] Active signal distribution devices can contribute to signal
degradation, for example by generating distortion products that
degrade SNR. The distortion contributed by an active device
typically increases as the input signal power to the device
increases. Additionally, the location of an active device within
the signal distribution system 1600 can affect the impact that the
device has on SNR. An active device located at an input to the
signal distribution system can experience a larger signal power,
and thus degrade the SNR more than an identical device located at a
the end of a transmission line, e.g. 1642, where the signal power
can be significantly attenuated.
[0239] Because the distortion typically increases at a rate greater
than the rate of increase in signal power, the SNR degrades for
input signals that are large in relation to the device
capabilities. A large input signal can be defined as a signal that
generates a predetermined level of distortion in an active device.
For example, a signal can be large when measured in relation to the
input signal level required to generate a 1 dB amplifier output
compression. Alternatively, a signal can be large when measured in
relation to an input signal level required to generate a particular
third order product. That is, a signal can be defined to be large
if a two-tone intermodulation test produces a third order
intermodulation distortion product that is a predetermined level
below the output signal, for example 40 dB. The definition of a
large signal is relative to the signal distribution system in which
a device is used and the previous definitions are not
exhaustive.
[0240] Conversely, when the signal is small, the uncorrelated noise
level may dominate the determination of SNR. Because an attenuator
typically degrades signal power and may not degrade the
uncorrelated noise power by an equivalent amount, the SNR following
the attenuator can degrade. The placement of a passive device can
also affect the amount of SNR degradation contributed by the
device. Attenuators placed where the signal is large may not affect
the SNR while identical attenuators placed where the signal is
small may significantly degrade the SNR.
[0241] Thus, there exists an optimum signal range that maximizes
SNR in the system. The optimum depends on the precise signal
distribution system and the nature of the information signal
distributed. The Automatic Gain Control (AGC) amplifier that is
detailed below can help the system maintain the optimal operating
range and thus help to maintain an optimum SNR in the system. The
AGC amplifier can diminish the effects that subsequent distribution
devices have on the SNR at the set top boxes 1652, 1654, and 1656.
Additionally, the AGC amplifier can minimize adverse effects of
adding or removing distribution paths in the signal distribution
system 1600. The AGC amplifier can, for example, be integrated into
the distribution switch 1630 or signal splitter 1670.
[0242] FIGS. 17A through 17D are functional block diagrams of AGC
amplifiers that can be, for example, integrated into the
distribution switch 1630 and/or signal splitter 1670 of FIG. 16.
The AGC amplifier can also be implemented in an intermediate signal
processing device, such as the signal splitter 1670 or some other
signal distribution device, alternatively referred to as a
distribution device or signal processing device. Typically, the AGC
amplifiers are not added as stand alone devices, but are
implemented in conjunction with other distribution devices.
[0243] In some embodiments, intermediate signal distribution
devices may not include AGC amplifiers. Such intermediate signal
processing devices lacking an AGC amplifier may be configured for
use in particular locations within the signal distribution system.
In other embodiments, the intermediate signal distribution devices
can, for example, include an AGC amplifier as the initial signal
processing element.
[0244] Implementing an AGC amplifier with a signal distribution
device allows the performance of the signal distribution system
1600 to be substantially unaffected by the physical location of the
signal distribution device. That is, the performance of the signal
distribution system 1600 is substantially indifferent to the
placement of a signal distribution device at the front end of a
cable run or at the back end of the cable run.
[0245] Implementing the AGC amplifier in the distribution device
1630 immediately following the low noise block 1626 can compensate
for gain variations in the low noise block 1626. Thus, embodiments
implementing the distribution device 1630 and low noise block 1626
locally or in a single housing may advantageously eliminate a
production adjustment of the low noise block 1626 gain. Thus, the
AGC function implemented in the distribution block 1630 can provide
a lowered production cost by eliminating a production tuning
step.
[0246] Each of the AGC amplifier embodiments shown in FIGS. 17A
through 17D can be implemented with a signal distribution device as
an integrated circuit, as discrete devices, or as a combination of
integrated circuits and discrete devices. An integrated circuit
can, for example, incorporate multiple independent AGC amplifiers
in parallel, with each AGC amplifier controlling the power of a
signal received from a satellite downlink. The integrated circuit
can be manufactured on a variety of substrate materials such as
silicon, germanium, gallium arsenide, indium phosphide, sapphire,
diamond, and the like, or any other suitable substrate material.
Additionally, the AGC amplifier embodiments can be manufactured
using a variety of manufacturing techniques including bipolar, FET,
BiCMOS, CMOS, SiGe, and the like.
[0247] FIG. 17A is a functional block diagram of a first AGC
amplifier embodiment. The AGC amplifier includes a variable gain
amplifier (VGA) 1710 and a detector 1720 connected to the output of
the VGA 1710. An output of the detector 1720 is connected to a gain
control input of the VGA 1710 to control the gain of the
amplifier.
[0248] The AGC amplifier implements an output referred AGC function
to attempt to maintain the output power of the power amplifier at a
predetermined optimal level, also referred to as the AGC set point.
The AGC function is a process that attempts to maintain a signal
power at the AGC setpoint. The AGC function increases the gain of
the amplifier 1710 when the output signal is below the AGC set
point. The AGC function can continue to increase the gain of the
VGA 210 as required, up to a maximum gain value. The VGA 210
continues to provide the maximum gain value as long as the output
signal power remains below the AGC set point.
[0249] Conversely, the AGC function decreases the gain of the VGA
1710 when the output signal power is above the AGC set point. The
AGC function can continue to decrease the gain of the VGA 1710 as
required, down to a minimum value. The AGC function continues to
provide the minimum gain value as long as the output signal power
remains greater than the AGC set point.
[0250] Within a system such as the signal distribution system 1600
of FIG. 16, there is typically a limit of input signal range. That
is, the input to the signal distribution system 1600 typically
falls within a predetermined range. In such a system, it is
possible to configure the AGC range such that one or more of the
AGC limits is not ever reached. For example, the input signal from
the satellite 1610 may vary over a predetermined range. If the AGC
amplifier in the distribution switch 1630 or signal splitter 1670
has an AGC range that is greater than the input signal range, the
AGC function may never reach its limits.
[0251] Initially, an input signal having an input signal power,
Pin, is provided to the input 1715 of the VGA 1710. The control
signal provided to the VGA 1710 can initially be set to control the
VGA 1710 to provide the maximum available gain, Gmax. The VGA 1710
then provides an output signal having an output power, Pout,
substantially equal to Pin +Gmax, for example, measured in terms of
decibels relative to a milliwatt (dBm).
[0252] The output from the VGA 1710 is connected to an input of a
power detector 1720. The power detector 1720 measures the output
signal power and generates a control signal that can correlate with
the output signal power. For example, the power detector 1720 can
be configured to provide an output voltage that correlates with a
given power level. Alternatively, the power detector 1720 can be
configured to provide an output current that correlates with a
given power level.
[0253] The power detector 1720 can be configured to measure the
power of the composite amplifier output signal, including desired
signals, noise, and distortion. Such a power detector 1720 can be a
broadband detector and can detect a power level over a broad
frequency band. Alternatively, the power detector 1720 can measure
the power of only a portion of the output power from the VGA 1710.
For example, the power detector 1720 can measure the power in a
predetermined bandwidth, where the predetermined bandwidth
represents only a portion of the bandwidth of the signal output
from the VGA 1710. The predetermined bandwidth can, for example, be
entirely within a desired signal bandwidth of the output from the
VGA 1710. Alternatively, the predetermined bandwidth can partially
overlap or be exclusive of a desired signal bandwidth of the VGA
1710 output.
[0254] The output of the power detector 1720 is connected to a
control input of the VGA 1710. The AGC amplifier can be configured
to provide an output referred AGC function. For example, the power
detector 1720 can detect an output power of the VGA 1710. The power
detector 1710 can also include a comparator having an AGC setpoint
coupled to one comparator input. The detected output power can be
provided to the second input of the comparator and compared to a
AGC setpoint. The output of the comparator can be filtered, for
example using an integrator. The output of the integrator can be
the detector output control signal that controls the gain of the
amplifier.
[0255] For example, a high power signal, one that is greater than
the AGC set point, at the input to the power detector 1720 produces
a control voltage. The control voltage value corresponds with an
amplifier gain value that is smaller than the original gain value.
The high power detector 1720 output reduces the gain of the VGA
1710 such that the power detected at the output of the VGA 1710 is
substantially equal to the AGC set point.
[0256] Although the VGA 1710 is shown as an amplifier, the AGC
function can be implemented with gain only, a combination of gain
and attenuation, or attenuation only. Additionally, the VGA 1710
can be implemented with multiple stages and multiple devices. For
example, the VGA 1710 can be configured as multiple cascaded
variable gain amplifiers, or as amplifiers cascaded with variable
attenuators, or as multiple variable gain amplifiers in parallel,
and the like.
[0257] Additionally, the power detector 1720 can be a diode
detector, a crystal detector, and the like. The power detector 1720
can be configured to sample mean power, peak power, RMS voltage,
mean voltage, peak voltage, mean current, RMS current, peak
current, or some other value correlated to signal level. The power
detector can be a single device or can be constructed of multiple
devices. As discussed above, the power detector 1720 can include,
for example, a detector, a comparator, and integrator, or some
other signal conditioning block.
[0258] Although the power detector 1720 is shown to provide an
output referred AGC function, the power detector 1720 can be
configured detect the signal power at other locations, such as at
the input of the VGA 1710. The power detector 1720 can be
configured to detect the signal power at some other location that
is remote from the VGA 1710, such as at the input to a set top box
of FIG. 16.
[0259] The actual AGC function can be implemented using a variety
of techniques, including feedback and feed-forward. Regardless of
whether the AGC function is configured as output referred using
feedback, or output referred using feed-forward techniques, the AGC
function can operate to provide a substantially stable output level
over an predetermined AGC range.
[0260] FIG. 17B is a functional block diagram of an embodiment of
an AGC amplifier. The AGC amplifier includes a constant gain
amplifier 1732 at the input to the AGC amplifier. The output of the
constant gain amplifier 1732 is connected to the input of a VGA
1734. The output of the VGA 1734 is connected to a power detector
1740. The output signal from the power detector 1740 is connected
to the control input of the VGA 1734 to control the gain of the VGA
1734.
[0261] The AGC amplifier embodiment in FIG. 17B is similar to the
embodiment of FIG. 17A except that a constant gain amplifier 1732
is implemented before the VGA 1734. The AGC amplifier of FIG. 17B
operates effectively the same as the AGC amplifier of FIG. 17A. The
gain of the constant gain amplifier 1732 can set a lower limit on
the gain of the AGC amplifier. However, the gain of the constant
gain amplifier 1732 can be negated by attenuation in the VGA 1734
if the VGA is configured to provide attenuation. The constant gain
amplifier 1732 can be included in an AGC amplifier, for example, in
order to provide a front end amplifier in the AGC amplifier having
a low noise figure.
[0262] FIG. 17C is a functional block diagram of another AGC
amplifier embodiment. The AGC amplifier includes a VGA 1752 at the
input of the AGC amplifier. The output of the VGA 1752 is connected
to a constant gain amplifier 1754. The output of the constant gain
amplifier 1754 is the output of the AGC amplifier. The output of
the VGA 1752 is also connected to the input of the power detector
1760. The detected output is provided to the control input of the
VGA 1752. Thus, in the embodiment of FIG. 17C, the power detector
1760 detects the power of an intermediate stage, rather than the
input or output of the AGC amplifier. Of course, the embodiment of
FIG. 17A can be modified to correspond to the embodiment of FIG.
17C by cascading the AGC amplifier with a constant gain amplifier.
Although a constant gain amplifier 1754 is implemented after the
VGA 1752, the composite AGC amplifier can be interpreted as being
output referred.
[0263] FIG. 17D is another embodiment of an AGC amplifier. The AGC
amplifier is an embodiment of a VGA coupled with a signal
distribution device. The AGC amplifier includes a VGA 1770 at the
input of the AGC amplifier. The output of the VGA 1770 is connected
to the input of a mixer 1780. A LO 1784 drives an LO port of the
mixer 1780. The output of the mixer 1780 is the output of the AGC
amplifier. The output of the mixer 1780 is also connected to the
input of the power detector 1790. The detected output is provided
to the control input of the VGA 1770.
[0264] In this AGC amplifier configuration, the AGC function is
combined with band translation. The AGC amplifier power controls
the output to track the AGC set point and can also frequency
convert the signal from an input frequency band to an output
frequency band. As noted earlier, a VGA such as 1770 can be
combined with a variety of signal distribution devices. The signal
splitter 1670 of FIG. 16 can represent another embodiment of a VGA
coupled with a signal distribution device.
[0265] The VGA 1770 operates in a manner as described above in
relation to the other AGC amplifier embodiments. The output of the
VGA 1770 is connected to an input port of the mixer 1780. The mixer
1780 operates to frequency convert the signal from a first
frequency band to a second frequency band. An LO 1784, which can be
a fixed frequency LO or a variable frequency LO, drives the LO port
of the mixer 1780. The mixer 1780 provides an output signal that
includes a frequency component that is at the sum of the input
signal frequencies and the LO frequency and a frequency component
that is at the difference of the input signal frequencies and the
LO frequency.
[0266] The power detector 1790 can be configured to detect signals
within a predetermined frequency band. Thus, the power detector
1790 can detect the signals in the desired frequency band while
ignoring signals outside the frequency band of interest. The AGC
amplifier can thus be configured to provide a controlled signal
amplitude combined with a frequency conversion.
[0267] The benefits of including an AGC stage in the signal
distribution system, such as within the distribution switch 1630 or
signal splitter 1670 in the system of FIG. 16, can be illustrated
with a comparison of an AGC signal distribution implementation with
a fixed gain signal distribution implementation. FIGS. 18A and 18B
show embodiments of cascaded amplifier configurations. The
configuration in FIG. 18A includes fixed gain amplifiers while the
configuration of FIG. 18B includes the AGC amplifiers. Such
cascaded amplifier configurations can be included in the signal
distribution switch of FIG. 16, for example, to provide three
independent copies of a single input signal destined for three
different geographic locations within the signal distribution
system.
[0268] FIG. 18A is an embodiment of a fixed gain signal
distribution section 1800, such as a distribution section that can
be implemented in the distribution switch of FIG. 16. For example,
the devices in the distribution section 1800 can be distributed at
front end, intermediate location, or near a termination of a signal
distribution system. The fixed gain distribution section 1800
includes three gain devices 1810, 1820, and 1830 connected in
series. Each of the gain devices, for example 1810, can be
configured as an active power divider having a fixed gain of 0 dB,
a noise figure (NF) of 3 dB, and an input third order intercept
point (IIP3) of +30 dBm. Alternatively, each of the gain devices
can include an amplifier in conjunction with some other type of
signal distribution device.
[0269] A first fixed gain device 1810 includes a fixed gain
amplifier 1812 followed by a passive power divider 1814 having a
first output 1818a and a second output 1818b. The composite gain
through the fixed gain amplifier 1812 and passive power divider
1814 to one of the outputs, for example 1818b, can be configured to
be 0 dB. The second output 1818a of the first fixed gain device
1810 is connected to the input of a second fixed gain device 1820.
The second fixed gain device 1820 also contains a fixed gain
amplifier 1822 and a passive power divider 1824 having a first
output 1828a and a second output 1828b. The second output 1828b of
the second fixed gain device 1820 is connected to the input of a
third fixed gain device 1830. The third fixed gain device 1830 is
similarly configured with a fixed gain amplifier 1832 followed by a
passive power divider 1834 having two outputs 1838a, 1838b.
[0270] An alternative signal distribution section 1850 including
AGC amplifiers is shown in FIG. 18B. The embodiment of the signal
distribution section 1850, including the AGC amplifiers, can be
implemented in the distribution switch of FIG. 16.
[0271] Three gain devices, 1860, 1870, and 1880 are cascaded in the
signal distribution section. Each of the gain devices 1860, 1870,
and 1880 includes an AGC amplifier followed by a passive power
divider. Each of the gain devices, 1860, 1870, and 1880 may also
include an AGC amplifier in conjunction with one or more other
signal distribution devices.
[0272] Each of the gain devices, for example 1860, can have an
output referred AGC function with an AGC set point of 0 dBm, an
IIP3 of +30 dBm, and a NF of 3 dB at 0 dB of gain. The gain device,
for example 1860, can have a gain range of from -20 dB to +20 dB.
Each of the AGC amplifiers can be, for example, one of the AGC
amplifier configurations shown in FIGS. 17A-17C.
[0273] An input signal is provided to an input of the first gain
device 1860. The input signal is coupled to the input of an AGC
amplifier 1862. The output of the AGC amplifier 1862 is connected
to the input of a power detector 1864. The output of the power
detector 1864 is connected to a control input of the AGC amplifier.
The output of the AGC amplifier 1862 is also connected to the input
of a power divider 1866 that has first and second outputs, 1868a
and 1868b respectively.
[0274] The second output 1868b of the first gain device 1860 is
connected to the input of a second gain device 1870. The output
from the first gain device 1860 is coupled to the input of an AGC
amplifier 1872. The output of the AGC amplifier 1872 is connected
to the input of a power detector 1874. The output of the power
detector 1874 is connected to a control input of the AGC amplifier.
The output of the AGC amplifier 1872 is also connected to the input
of a power divider 1876 that has first and second outputs, 1878a
and 1878b respectively.
[0275] The second output 1878b of the second gain device 1870 is
connected to the input of a third gain device 1880. The output from
the second gain device 1870 is coupled to the input of an AGC
amplifier 1882 in the third gain device 1880. The output of the AGC
amplifier 1882 is connected to the input of a power detector 1884.
The output of the power detector 1884 is connected to a control
input of the AGC amplifier. The output of the AGC amplifier 1882 is
also connected to the input of a power divider 1886 that has first
and second outputs, 1888a and 1888b respectively.
[0276] The performance of the fixed gain distribution section 1800
can be compared against the performance of the variable gain
distribution section 1850 for two operating conditions. In the
first operating condition, the input signal is relatively small and
uncorrelated noise is a significant factor limiting the SNR. In the
second operating condition, the input signal is relatively large,
and distortion products are significant factors limiting the
SNR.
[0277] In the first operating condition, the input signal is
relatively small. The configuration of the fixed gain distribution
section 1800 does not change. However, the variable gain
distribution section 1850 automatically configures itself to
provide gain, up to a maximum gain level.
[0278] An active device, such as an amplifier, typically has
multiple noise sources associated with it. The noise contribution
of cascaded amplifiers can be reduced if the front end device has
significant gain. The noise contribution of subsequent stages can
become insignificant, and thus, the degradation to SNR can be
minimized. Additionally, other noise contributors after the first
gain stage, or front end device, degrade the SNR less than without
the front end gain device. Thus, including the front end gain stage
reduces the overall system SNR degradation. The performance of the
fixed gain distribution section 1800 can be compared to the
variable gain distribution section 1850 by examining the noise
figures. The noise figure in a cascaded system is given by the
following formula: nf cascade = nf .times. .times. 1 + i = 2 N
.times. nfi - 1 j = 1 i - 1 .times. .times. A j , .times. where
.times. .times. N = number .times. .times. of .times. .times.
stages , A j = gain .times. .times. of .times. .times. jth .times.
.times. stage ##EQU1##
[0279] The noise figure values in the formula are given as ratios,
while noise figure specified for the devices are given in dB. Thus,
the NF for the gain devices, for example 1810 or 1870, needs to be
converted from decibels to ratios before application of the
formula. Table 1 provides a summary of the cascaded noise figures
for the two gain distribution sections, 1800, 1850. Psig represents
the signal power, in dBm at either the input or output of the gain
devices. The gain of the elements is provided in dB. The noise
figure, in dB, is provided for each gain device and the
corresponding cascaded noise figure, in dB, is provided at the
output of each gain device. TABLE-US-00001 TABLE 1 Fixed Gain
Distribution Section Psig (dBm) -20 -- -20 -- -20 -- -20 Gain (dB)
0 0 0 NF (dB) 3 3 3 NFtot (dB) 3 4.8 6 Variable Gain Distribution
Section Psig (dBm) -20 -- 0 -- 0 -- 0 Gain (dB) 20 0 0 NF (dB) 3 3
3 NFtot (dB) 3 3.02 3.04
[0280] Thus, it can be seen that the ability of the variable gain
distribution section 1850 to include gain in an initial amplifier
section results in greatly reduced signal degradation due to noise
contributed by subsequent stages when compared to the fixed gain
distribution section 1800. Noise contributors after the initial
gain section degrade the SNR less than without the gain section.
Therefore, overall system degradation of SNR can be reduced with
the inclusion of an initial gain section.
[0281] In the second operating condition, the input signal is
relatively large. The configuration of the fixed gain distribution
section 1800 does not change. However, the variable gain
distribution section 1850 automatically configures itself to
provide attenuation, up to a maximum attenuation level. When input
signal levels are relatively large, distortion components, such as
third order intermodulation distortion products, can be the
dominant factor in degrading SNR. A cascaded IIP3 for the signal
distribution sections, 1800, 1850 can be calculated and compared to
illustrate the advantages of variable gain distribution over fixed
gain distribution. The cascaded IIP3 of a gain section is given by
the formula: 1 IP tot = 1 IP .times. .times. 3 1 + i = 2 N .times.
j = 1 i - 1 .times. .times. A j IP .times. .times. 3 i ##EQU2##
[0282] The IP3 values in the formula are the linear terms and are
not the values in dBm. Similarly, the gain values are provided as
ratios and are not in dB. Table 2 provides a summary of the
cascaded IIP3 for the two gain distribution sections 1800, 1850.
Psig represents the signal power, in dBm at either the input or
output of the gain devices. The gain of the elements is provided in
dB. The IIP3, in dBm, is provided for each gain device and the
corresponding cascaded IIP3, in dBm, is provided at the output of
each gain device. TABLE-US-00002 TABLE 2 Fixed Gain Distribution
Section Psig (dBm) +20 +20 +20 +20 Gain (dB) 0 0 0 IIP3 (dBm) +30
+30 +30 IIP3tot (dBm) +30 +27 +25.2 Variable Gain Distribution
Section Psig (dBm) +20 0 0 0 Gain (dB) -20 0 0 IIP3 (dBm) +30 +30
+30 IIP3tot (dBm) +30 +29.96 +29.91
[0283] Thus, it can be seen that the ability of the variable gain
distribution section 1850 to include attenuation in an initial
amplifier section results in greatly reduced signal degradation due
to noise contributed by subsequent stages when compared to the
fixed gain distribution section 1800. Distortion contributors after
the initial attenuation stage degrade the SNR less than without the
attenuation stage. The overall system degradation of SNR can be
reduced with the inclusion of an initial attenuation section.
[0284] The inclusion of an AGC function in a signal distribution
section can thus improve the quality of the signal compared to a
fixed gain configuration. The advantages of the variable gain
section over the fixed gain section under the extreme conditions of
low input signal power and high input signal power show that the
variable gain distribution section has flexibility as to its
position within a signal distribution system. The variable gain
distribution section need not be placed at the front end or as a
final stage in a signal distribution system.
[0285] One embodiment of the band translation switch 1510 can be
used in a signal distribution system designed to provide
distribution of satellite television signals in a residence. The
AGC amplifiers 1520a-1520d provide variable gain and attenuation
based on the power of the input signal. The measurement point for
the AGC function is at the output of the AGC amplifiers 1520a-1520d
and the gain of the crosspoint switch 1530 and the band translation
devices 1540a-1540d are fixed.
[0286] Each AGC amplifier 1520a-1520d followed by a crosspoint
switch 1530, band translation device 1540a-1540d, filter
1550a-1550d, and signal combiner 1560a-1560b can be configured to
provide a total gain that ranges from a minimum of -7 dB to a
maximum of +7 dB. The corresponding NF of a path through the band
translation switch 1510 from the AGC amplifier, for example 1520a,
through to the output of a band translation device, for example
1540a, can vary from, for example, a high of 24 dB to a low of 10
dB. The signal path experiences a higher NF when providing
attenuation and has a lower NF when the gain is unity or greater.
Similarly, the IIP3 associated with the signal path can range from
a minimum of -7 dBm to a maximum of +7 dBm. For example, the IIP3
of the signal path can be -15, -10, -7, -6, -5, -4, -3, -2, -1, 0,
+1, +2, +3, +4, +5, +6, +7, +10, +15, +20, +25, or +30 dBm.
[0287] The IIP3 of the AGC amplifier 1520a is typically higher when
the amplifier is configured to provide attenuation, which
contributes to the composite IIP3 of the signal path. The IIP3 of
the AGC amplifier 1520a can vary in proportion to the gain of the
amplifier.
[0288] Because the AGC amplifier 1520a also provides the signal to
a cascade output, the characteristics of the AGC function at the
cascade output are substantially the same as the characteristics of
the AGC amplifier 1520a. Also because the AGC function is provided
before the cascade output, the benefits of the AGC function are
experienced in the main signal path as well as the signal path
through the cascade output.
[0289] This band translation switch 1510 configuration can be used
in a signal distribution system where the input to the band
translation switch 1510 can be expected to vary over the range of
-50 dBm through -10 dBm. The AGC amplifiers 1520a-1520d can be
configured to have an output referred AGC setpoint of -17 dBm,
where the output refers to the output signal of the switch 1500.
The band translation switch 1510 need not actually measure the
power at the output of the switch 1500. Because the devices
following the band translation switch 1510 have fixed gains, the
AGC output can be interpreted as being output referred to any point
past an AGC amplifier where the gain or attenuation is fixed.
[0290] Using this AGC setpoint, the AGC amplifier, for example
1520a, provides a gain of 7 dB when the input signal is -24 dBm or
below. Additionally, the AGC amplifier 1520a provides -7 dB of
gain, or 7 dB of attenuation, when the input signal is -10 dBm or
greater. Thus, within the input power range of -24 dBm through -10
dBm the AGC amplifier 1520a provides a constant output power of -17
dBm.
[0291] FIG. 19 is a functional block diagram of multiple band
translation switches 1910, 1920, 1930, 1940, and 1950, connected in
a signal distribution system. The band translation switches 1910,
1920, 1930, 1940, and 1950, can be configured with LNBs to provide
the distribution switch of FIG. 16. However, as noted earlier, one
or more of the band translation switches 1910, 1920, 1930, 1940,
and 1950 can be positioned at other locations within the signal
distribution system. For example, one or more of the band
translation devices can be positioned near the signal input, at an
intermediate position within the signal distribution system, or
near a termination or destination device of the signal distribution
system.
[0292] A first band translation switch 1910 includes an LNA input
that can be connected to an LNB that block converts a satellite
downlink transmission. The output of the first band translation
switch 1910 is connected to an input of a second band translation
switch 1920 that, in turn, has an output connected to a third band
translation switch 1930. A cascade output of the first band
translation switch 1910 is connected to the input of a fourth band
translation switch 1940. The output of the fourth band translation
switch 1940 is connected to the input of a fifth band translation
switch 1950.
[0293] Each of the band translation switches, 1910, 1920, 1930,
1940, and 1950, can be the band translation switch of FIG. 15 and
can include one of the AGC amplifiers of FIGS. 17A-17C. Each of the
band translation switches, 1910, 1920, 1930, 1940, and 1950, can be
configured similarly to the first band translation switch 1910. In
the first band translation switch 1910, an input VGA 1912 receives
the input signal from the LNB's. The VGA 1912 typically has a low
noise figure, such that the noise figure of the band translation
switch 1910 from the input to a band translation output is below 3
dB, 4 dB, 5 dB, 6 dB, 8 dB, 10 dB, 12 dB, 14 dB, 15 dB, 20 dB, 25
dB, 30 dB, 35 dB or 40 dB. The noise figure of the band translation
switch 1910 from an input to the cascade output is typically closer
to the value of noise figure of the VGA 1912 and can be, for
example less than 3 dB, 4 dB, 5 dB, 6 dB, 8 dB, 9 dB, 10 dB, 12 dB,
14 dB, 15 dB, 20 dB, 24 dB, 25 dB, 30 dB, 35 dB or 40 dB.
[0294] Additionally, the VGA 1912 contributes to the IIP3 of the
band translation switch 510. The band translation switch 1910
typically has an IIP3, measured from an input to an output of a
band translation device, of greater than -40, -30, -20, -10, -8,
-7, -6, -5, -4, -3, -2, -1, 0, +1, +2, +3, +4, +5, +6, +7, or +8,
+15, +20, +22, +25, +26, +27, +28, +29, or +30 dBm. Similarly, the
band translation switch 1910 typically has an IIP3, measured from
an input to the cascade output of greater than -10, -5, +1, +2, +3,
+4, +5, +6, +7, +8, +9, +10, +15, +20, +25, or +30 dBm.
[0295] The output of the VGA 1912 is connected to a detector 1914
and N.times.M crosspoint switch 1916. The detector 1914 detects the
power output by the VGA 1912 and provides a detected output that is
connected to the control input of the VGA 1912. Additionally, the
output of the VGA 1912 drives the cascade output of the first band
translation switch 1910. The output of the N.times.M crosspoint
switch 1916 is connected to a band translation device 1918.
[0296] Although only one VGA 1912 and detector 1914 are shown in
the first band translation switch 1910, more than one VGA 1912 and
cascade output can be included in a band translation switch, as
shown in FIG. 19. Thus, the benefits of having an AGC function in
line with a signal distribution path can be provided to two signal
paths originating from a single VGA, for example 1912, in a single
band translation switch, 1910.
[0297] Each of the subsequent band translation switches 1920, 1930,
1940 and 1950, can also be connected to signal paths at their
cascade outputs and can likewise control the signal level and
minimize the subsequent noise contributions by utilizing an input
AGC stage. The fourth band translation switch 1940 connected to a
cascade output of the first band translation switch 1910 does not
contribute noise to the originating signal path and further
controls noise contributions from subsequent stages.
[0298] FIG. 20 is a flowchart of a signal distribution method 2000
for use in a signal communication system, such as the satellite
communication system shown in FIG. 16. The method 2000 begins at
block 2002 where the distribution signals are received. The signals
can be received from a satellite, as in FIG. 16, or can be received
from an antenna configured to receive terrestrial signals, a cable,
or an optical link. Additionally, the signals can be received from
a combination of sources.
[0299] After receiving the signals to be distributed, the signals
are amplified, typically by a low noise amplifier, as shown in
block 2010. Because the gain can be varied from a positive gain
value to a negative value, the amplifier may not be a low noise
amplifier under all operating conditions and can be an attenuator
under some operating conditions. In this context, a negative gain
value refers to attenuation.
[0300] After amplification, the output power is measured, block
2012. Because output power is measured after the gain stage, the
subsequent AGC function based on the measured output power can be
referred to as output referred AGC. The measured output power is
then used as a factor for varying the gain, block 2014. As
previously discussed, the gain can typically be varied over a range
spanning positive gain to attenuation.
[0301] A cascade output is also provided, block 2020, and can be
provided after the AGC function. The gain controlled signal can be
provided as a cascade output, as is shown in FIGS. 15 and 19.
[0302] Additionally, the signal is routed to a destination path,
block 2030, such as by the N.times.M crosspoint switch shown in
FIG. 15. The signal that is routed to the destination by the
N.times.M crosspoint switch is typically independent of the signal
provided to the cascade output. Thus, as is shown in the band
translation switch of FIG. 15, the output of the AGC section is
provided as a cascade output and is also provided to the input of
the N.times.M crosspoint switch to be routed to one of M possible
distribution paths.
[0303] The signal that is routed to a distribution path can then be
band translated, block 2040. A band translation block can include a
mixer to selectively translate the signal from a first frequency
block to a second frequency block. Additionally, the band
translation block can be configured to have a pass through path
where the signal is not frequency translated.
[0304] Following band translation, the signal output from the band
translation block can be filtered, block 2050, to remove noise and
unwanted frequency components that are outside of a band of
interest. Two or more of the filtered signals can be combined to
produce a composite signal, block 2060. The two or more filtered
signals can originate from one or more independent signal
distribution paths. Each of the filtered signals can be in a
distinct frequency band. Alternatively, one or more of the filtered
signals can be in a frequency band that overlaps the frequency band
of another of the filtered signals.
[0305] Although the method 2000 is shown with flow from one block
to the next, the order of the method blocks is not limited to the
order shown in FIG. 20.
[0306] A mixer with a fixed frequency LO can be implemented in the
band translation device to provide frequency translation. FIG. 21
is a functional block diagram of a band extraction system 2100 for
a dual band-stacked frequency plan. The band extraction system 2100
is configured to position input signals from either an upper or
lower band block into the upper band block. The band extraction
system 2100 is also configured to bandpass the upper band block and
reject signals from the lower band block.
[0307] The band extraction system 2100 implements a band
translation device 2118 and a bandpass filter 2150. The band
translation device 618 includes a pass through signal path 2122 and
a frequency translation signal path. The frequency translation
signal path includes a mixer 2120 and a LO 2130.
[0308] As discussed earlier, one embodiment of a dual band-stacked
frequency plan has a lower band block at 950-1450 MHz and an upper
band block at 1650-2150 MHz. Thus, the filter 2150 is configured as
a bandpass filter that passes 1650-2150 MHz and rejects signals in
at least the 950-1450 MHz band.
[0309] The input signal is a band stacked signal having an input
lower band signal 2110 and an input upper band signal 2112. Band
extraction for the pass through configuration is trivial. The pass
through signal path 2122 of the band translation device 2118 is
selected. The input signal is pass through the band translation
device 2118 without frequency translation. The upper band block is
then extracted using the filter 2150 to produce an output
signal.
[0310] The frequency translation signal path is used when the input
lower band signal 2110 is to be frequency translated to the upper
band block. The input signals are directed to an input of the mixer
2120 for frequency translation. A LO 2130 is connected to a LO port
of the mixer 2130. A LO 2130 tuned to a frequency of 2.times. the
band block center mean will result in a frequency translation of
signals in the upper band block to the lower band block and
frequency translation of signals in the lower band block to the
upper band block. The LO frequency for the frequency plan shown in
FIG. 21 is 3.1 GHz. Thus, this LO frequency can be used when
extracting either the upper band block or the lower band block.
Only the bandpass filter needs to be changed to extract the desired
band block. The mixer 2130 provides output signals that are at the
sum of the input signal frequency and the LO frequency and output
signals that are at the difference of the input signal frequency
and the LO frequency.
[0311] An input lower band signal is frequency translated by the
mixer 2130 to an upper difference band 2142 and a lower sum band
2146. An input upper band signal 2112 is frequency translated by
the mixer 2130 to a lower difference band 2140 and an upper sum
band 2148. The sum and difference signals, 2140, 2142, 2146 and
2148, are provided to the filter 2150 which passes only the desired
output upper band signal 2160 and rejects at least the other mixer
products as well as the LO frequency. Thus, the input lower band
signal, in the 950-1450 MHZ band, is frequency translated to an
output upper band signal in the 1650-2150 MHz band.
[0312] The sum and difference bands are symmetric about the LO
frequency 2144. Thus, this LO frequency swaps the positions of the
input upper and lower signal. However, the difference bands are
frequency inverted. A signal that is in the upper sideband of the
input lower band 2110 is translated to the lower sideband of the
upper difference signal. Frequency inversion does not pose any
problems for double sideband signals. Similarly, spectral inversion
does not pose problems for digitally modulated signals that are
processed by a demodulator with built in spectral inversion. The
frequency inverted signals can be further processed if the system
designer requires, or desires, a particular frequency relationship.
The two band-stacked configuration allows for a single LO to
simultaneously rearrange the band stack. In other embodiments, each
frequency component in the final composite output signal may be
translated through a different band translation device. For
example, a first band translation device may frequency translate a
first signal to a first output signal band. Similarly, a second
band translation device can frequency translate a second signal to
a second output frequency band, and so on up to M frequency bands.
Multiple output frequency bands can then be combined in one or more
combiners. Each output frequency band can be one or more channels
corresponding to signals from one or more transponders.
[0313] Although the previous embodiments show the frequency
translation device as a single mixer, other frequency translation
means can be used. A single frequency translation device can be a
mixer, sampled switch, switching mixer such as diode ring, Gilbert
cell, FET ring mixers, or nonlinear mixers such as diodes, linear
multipliers such as translinear bipolar devices, variable
resistors, or the like. Alternatively, the frequency translation
device can perform multiple frequency translations. The input band
blocks can be upconverted to a high Intermediate Frequency (IF)
that is at a frequency greater than the highest frequency in the
input band blocks. The IF signal can then be filtered to extract
the band block of interest, where the desired band block can
correspond to one or more channels from one or more transponders.
The desired band block can then be downconverted to the desired
output band block. The high IF can be a common IF or can be
different for different band translation devices. Each high IF
signal can be downconverted to a portion of an output frequency
band. Multiple downconverted signals can be combined into a
composite output signal.
[0314] Alternatively, the input band blocks can be downconverted to
a low IF that is lower than the lowest frequency component of the
input band blocks. The low IF signal can then be filtered to select
the desired band block. The filtered band block is then upconverted
to the desired output band block. For example, independent input
signals can each be downconverted to a common IF where one or more
channels are passed in a filter. The low IF can be a common IF or
can be different for different band translation devices. Each of
the filtered signals at the IF can then be upconverted to a desired
frequency band. Multiple upconverted signals can then be combined
to form a composite signal. The composite signal can then be output
to a set top box or boxes.
[0315] In another alternative, the desired input band block can be
downconverted to a baseband signal. The baseband signal can then be
filtered, for example with a lowpass filter, before being
upconverted to the desired output band block. If the input signals
are quadrature modulated or if the two sidebands have different
information, the input signal can be downconverted into two
baseband channels in quadrature. Typically, a quadrature
downconverter can frequency translate the signals to baseband
In-phase (I) and Quadrature (Q) channels. The baseband I and Q
channels can then be filtered, such as by a LPF, and the filtered
signals I and Q upconverted to the desired band block.
[0316] For example, multiple signals can be frequency translated to
baseband signals in a corresponding number of band translation
devices. Each of the baseband signals can be upconverted to a
portion of an output frequency band. The multiple upconverted
signals can be combined to form a composite signal.
[0317] In other embodiments, more than two frequency translations
can be used with various combinations of simple mixers, image
reject mixers, IQ down converters, Single Side-Band upconverters,
and filters to provide the frequency translation and filtering
functions.
[0318] FIG. 22 is a functional block diagram of an embodiment a
portion of a band translation switch configured to switch and
frequency translate signals from two sources to a single output.
The functional elements are shown along with a depiction of the
signal spectrum output from the element.
[0319] A crosspoint switch 2210 can be configured to have N inputs
and M outputs, where N and M can be the same or different integers.
FIG. 7 shows two of the outputs from the crosspoint switch
2210.
[0320] A first crosspoint switch 2210 output provides a first
composite signal. The first composite signal includes a first lower
input band signal 2212a and a first upper input band signal 2212b.
Similarly, a second crosspoint switch 2210 output provides a second
composite input signal. The second composite signal includes a
second lower input band signal 2214 and a second upper input band
2214b signal.
[0321] The first composite signal is band translated in a band
translation device 2220, such as a mixer having an LO tuned to
2.times. (block center mean). The output of the band translation
device 2220 includes signals at a lower input band 2232a, an upper
band 2232b, a lower sum band 2234a, and an upper sum band 2234b.
The output of the band translation device 2220 is connected to a
first filter 2240 that is configured to pass the upper band and
substantially reject all other signals. The output of the first
filter 2240 includes the upper band 2252 with the remaining signal
components substantially rejected. The output of the first filter
2252 is connected to a first input of a signal combiner 2260.
[0322] The second output of the crosspoint switch 2210 is connected
to a band translation device 2222 configured for pass through.
Thus, the output of the band translation device 2222 appears
substantially the same as the output from the crosspoint switch
2210. The signal includes a lower band 2236a and an upper band
2236b. The output of the band translation device 2222 is connected
to a second filter 2242. The second filter 2242 is configured to
pass the lower band and substantially reject all other signals. The
output of the second filter 2242 is connected to a second input of
the signal combiner 2260.
[0323] The signal combiner 2260 is configured to combine the
signals provided to its first and second inputs. The output of the
signal combiner is thus a band-stacked signal having a lower band
signal 2272 and an upper band signal 2274. The lower band signal
2272 is provided by the second crosspoint switch output and the
upper band signal 2274 is provided by the first crosspoint switch
output. Thus, it can be seen how a band-stacked output having
signal components from any number of predetermined input bands can
be constructed.
[0324] FIG. 23 shows a flowchart of an embodiment of a method 2300
of frequency translating input frequency bands using an integrated
band translation switch. An initial function performed is receiving
satellite transponder signals 2310. The satellite transponder
signals can be received from one or more satellite transponder
groups located on one or more satellites. The satellite transponder
signals are typically received in an antenna. However, satellite
transponder signals can be received from signal distribution
devices, such as a cable television headend, that use an antenna to
directly receive the satellite transponder signals.
[0325] After receiving the satellite transponder signals, the
received signals are block converted to first predetermined bands
2320. The first predetermined bands can also be referred to as
predetermined input bands. One or more block converters can be used
to convert the transponder signals to the first predetermined
bands. The block converters can be low noise block converters
configured to block convert signals from multiple transponder
groups and band-stack the converted transponder group signals into
one or more of the first predetermined bands. Each of the
band-stacked signals can then be provided to an input to a routing
device or assembly.
[0326] The block converted signals are then routed to destinations
2330. The routing can be a selective routing that is controlled
using one or more control lines communicating one or more control
signals. Additionally, the routing can be performed by a switch,
such as a crosspoint switch, that is configured to route any of the
LNB signals to one or more outputs. There can be any number N of
LNB's. The crosspoint switch can be configured to have N inputs
corresponding to the N LNB's. The crosspoint switch can also be
configured to have M outputs, where the number of outputs M can be
the same, or different, from the number of inputs N. The crosspoint
switch can be configured to route a signal from any one of the
crosspoint switch inputs to any one or more of the crosspoint
switch outputs.
[0327] After routing the band-stacked transponder signals, the
signals are band translated from the first predetermined frequency
band to a second predetermined frequency band. Each of the
band-stacked signals are translated independently of the others.
The first predetermined band can be the same, or different, from
the second predetermined band.
[0328] The band translated signals in the second predetermined
frequency bands are then filtered 2350. Alternatively, filtering
can occur prior to band translation. The filter is configured to
pass the desired signal in the desired band and reject
substantially the remaining frequency components.
[0329] The filtered signals are then combined 2360. One or more of
the filtered signals can be combined to produce a band-stacked
output signal. A combiner can be configured to combine any number
of filtered signals. Additionally, more than one combiner can be
used to provide more than one band-stacked output signal. In an
alternative embodiment, a diplexer can be used to filter and
combine signals.
[0330] One or more of the method steps can be performed by a single
integrated circuit. For example, the block conversion can be
performed by an integrated circuit. Similarly, routing the signals
to destinations can be performed by an integrated circuit that can
be the same, or different from the integrated circuit that performs
the block conversion. In another embodiment, band translating the
signals is performed on an integrated circuit that can be the same,
or different from, the integrated circuit used for routing the
signals or block converting the signals. Similarly filtering and
combining the signals can be performed on one or more integrated
circuits that are the same as, or different from, any of the other
integrated circuits. One or more of the integrated circuits can
include a semiconductor substrate such as a silicon substrate.
Alternatively, the integrated circuit substrate can include
germanium, gallium arsenide, indium phosphide, sapphire, diamond,
and the like, or any other suitable substrate material.
Additionally, the integrated circuit can be manufactured using any
suitable technology, such as Silicon Germanium (SiGe), bipolar,
FET, and the like.
[0331] Thus, an integrated crosspoint switch with band translation
has been disclosed. Input band-stacked signals from one or more
satellite transponder groups can be reconfigured to provide
band-stacked output signals having a signal band from any
transponder group positioned in any of the predetermined output
bands. A LNB can receive one or more transponder group signals to
create an band-stacked input signal. A low noise amplifier in an
integrated circuit can amplify the LNB output and also provide a
cascade output signal. The output from the low noise amplifier can
be connected to a crosspoint switch on the same integrated circuit.
The crosspoint switch can be configured to route signals from any
one of its signal inputs to any one of its signal outputs. Each of
the crosspoint signal outputs is connected to a band translation
device. The band translation device can also be configured on the
same integrated circuit. Each of the band translation devices is
configured to frequency translate or pass through an input signal.
The band translation device can frequency translate a predetermined
input band to a predetermined output band. The output from one or
more band translation devices can be combined to produce a
band-stacked output signal.
[0332] Electrical connections, couplings, and connections have been
described with respect to various devices or elements. The
connections and couplings can be direct or indirect. A connection
between a first and second device can be a direct connection or can
be an indirect connection. An indirect connection can include
interposed elements that can process the signals from the first
device to the second device.
[0333] Those of skill in the art will understand that information
and signals can be represented using any of a variety of different
technologies and techniques. For example, data, instructions,
commands, information, signals, bits, symbols, and chips that can
be referenced throughout the above description can be represented
by voltages, currents, electromagnetic waves, magnetic fields or
particles, optical fields or particles, or any combination
thereof.
[0334] Those of skill will further appreciate that the various
illustrative logical blocks, modules, circuits, and algorithm steps
described in connection with the embodiments disclosed herein can
be implemented as electronic hardware, computer software, or
combinations of both. To clearly illustrate this interchangeability
of hardware and software, various illustrative components, blocks,
modules, circuits, and steps have been described above generally in
terms of their functionality. Whether such functionality is
implemented as hardware or software depends upon the particular
application and design constraints imposed on the overall system.
Skilled persons can implement the described functionality in
varying ways for each particular application, but such
implementation decisions should not be interpreted as causing a
departure from the scope of the invention.
[0335] The various illustrative logical blocks, modules, and
circuits described in connection with the embodiments disclosed
herein can be implemented or performed with a general purpose
processor, a digital signal processor (DSP), an application
specific integrated circuit (ASIC), a field programmable gate array
(FPGA) or other programmable logic device, discrete gate or
transistor logic, discrete hardware components, or any combination
thereof designed to perform the functions described herein. A
general-purpose processor can be a microprocessor, but in the
alternative, the processor can be any processor, controller,
microcontroller, or state machine. A processor can also be
implemented as a combination of computing devices, for example, a
combination of a DSP and a microprocessor, a plurality of
microprocessors, one or more microprocessors in conjunction with a
DSP core, or any other such configuration.
[0336] The steps of a method or algorithm described in connection
with the embodiments disclosed herein can be embodied directly in
hardware, in a software module executed by a processor, or in a
combination of the two. A software module can reside in RAM memory,
flash memory, ROM memory, EPROM memory, EEPROM memory, registers,
hard disk, a removable disk, a CD-ROM, or any other form of storage
medium. An exemplary storage medium can be coupled to the processor
such the processor can read information from, and write information
to, the storage medium. In the alternative, the storage medium can
be integral to the processor. The processor and the storage medium
can reside in an ASIC.
[0337] The above description of the disclosed embodiments is
provided to enable any person skilled in the art to make or use the
invention. Various modifications to these embodiments will be
readily apparent to those skilled in the art, and the generic
principles defined herein can be applied to other embodiments
without departing from the spirit or scope of the invention. Thus,
the invention is not intended to be limited to the embodiments
shown herein but is to be accorded the widest scope consistent with
the principles and novel features disclosed herein.
* * * * *