U.S. patent application number 11/506369 was filed with the patent office on 2007-02-22 for method and apparatus for digital demodulation and further processing of signals obtained in the measurement of electrical bioimpedance or bioadmittance in an object.
Invention is credited to Eberhard Gersing, Markus J. Osypka.
Application Number | 20070043303 11/506369 |
Document ID | / |
Family ID | 35058271 |
Filed Date | 2007-02-22 |
United States Patent
Application |
20070043303 |
Kind Code |
A1 |
Osypka; Markus J. ; et
al. |
February 22, 2007 |
Method and apparatus for digital demodulation and further
processing of signals obtained in the measurement of electrical
bioimpedance or bioadmittance in an object
Abstract
Methods and apparatus for digital demodulation of signals
obtained in the measurement of electrical bioimpedance or
bioadmittance of an object. One example comprises: generating an
excitation signal of known frequency content; applying the
excitation signal to the object; sensing a response signal of the
object; sampling and digitizing the response signal to acquire a
digitized response signal representing the response signal with
respect to frequency content, amplitude and phase; correlating, for
each frequency f.sub.AC of the excitation signal applied, digitized
samples of the response signal, with discrete values representing
the excitation signal; calculating, using the correlated signals
for each frequency f.sub.AC of the excitation signal applied,
complex values for the bioimpedance Z(f.sub.AC); providing, over
time, a set of digital bioimpedance waveforms Z(f.sub.AC,t));
separating the base bioimpedance Z.sub.0(f.sub.AC), from the
waveforms; and separating the changes of bioimpedance
.DELTA.Z(f.sub.AC,t), from the waveforms.
Inventors: |
Osypka; Markus J.; (La
Jolla, CA) ; Gersing; Eberhard; (Gottingen,
DE) |
Correspondence
Address: |
TIMOTHY N. ELLIS, PATENT ATTORNEY
8680 VIA MALLORCA, SUITE D
LA JOLLA
CA
92037
US
|
Family ID: |
35058271 |
Appl. No.: |
11/506369 |
Filed: |
August 17, 2006 |
Current U.S.
Class: |
600/547 |
Current CPC
Class: |
A61B 5/7239 20130101;
A61B 5/0535 20130101; A61B 5/0809 20130101; A61B 5/7228
20130101 |
Class at
Publication: |
600/547 |
International
Class: |
A61B 5/05 20060101
A61B005/05 |
Foreign Application Data
Date |
Code |
Application Number |
Aug 17, 2005 |
EP |
EP05017871 |
Claims
1. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of a biological object,
wherein the method comprises the following operations: placing a
first current electrode and a second current electrode in contact
with the biological object; placing a first voltage sensing
electrode and a second voltage sensing electrode in contact with
the biological object; applying an output of an AC current source
to the first current electrode and the second current electrode, to
cause an AC current to flow through the biological object between
the first current electrode and the second current electrode;
measuring a voltage across the first voltage sensing electrode and
the second voltage sensing electrode, wherein the measured voltage
is produced due to application of the output of the AC current
source to the first current electrode and the second current
electrode; digitizing the measured voltage to produce Object
Voltage Samples; obtaining a real part of the bioimpedance of the
biological object, by correlating the Object Voltage Samples with
corresponding reference current samples; and obtaining an imaginary
part of the bioimpedance of the biological object, by correlating
the Object Voltage Samples with corresponding reference current
samples that are shifted in phase by -90 degrees.
2. The method of claim 1, further comprising: calculating a
magnitude of the bioimpedance of the biological object, by
calculating the square root of the sum of the square of the real
part of the bioimpedance of the biological object and the square of
the imaginary part of the bioimpedance of the biological object;
and calculating a phase of the bioimpedance of the biological
object plus a measurement system phase shift, by calculating the
arctan of the ratio of the imaginary part of the bioimpedance of
the biological object to the real part of the bioimpedance of the
biological object.
3. The method of claim 1: wherein the biological object is an
animal; and wherein the operation of digitizing the measured
voltage produces unfitted Object Voltage Samples, and wherein the
operations further comprise fitting the unfitted Object Voltage
Samples to discrete values of an ideal sinusoid to produce the
Object Voltage Samples.
4. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of a biological object,
wherein the method comprises the following operations: placing a
first current electrode and a second current electrode in contact
with the biological object; placing a first voltage sensing
electrode and a second voltage sensing electrode in contact with
the biological object; applying an output of an AC current source
to the first current electrode and the second current electrode, to
cause an AC current to flow through the biological object between
the first current electrode and the second current electrode;
measuring a voltage between the first voltage sensing electrode and
the second voltage sensing electrode, wherein the measured voltage
is produced due to application of the output of the AC current
source to the first current electrode and the second current
electrode; digitizing the measured voltage to produce Object
Voltage Samples; producing a voltage that is directly proportional
to, and in phase with, the output of the AC current source;
digitizing the voltage that is directly proportional to, and in
phase with, the output of the AC current source, to produce Object
Current Samples; obtaining a real part of the bioimpedance of the
biological object, by correlating the Object Voltage Samples with
the Object Current Samples; and obtaining an imaginary part of the
bioimpedance of the biological object, by correlating the Object
Voltage Samples with corresponding Object Current Samples that are
shifted in phase by -90 degrees.
5. The method of claim 4, further comprising: calculating a
magnitude of the bioimpedance of the biological object, by
calculating the square root of the sum of the square of the real
part of the bioimpedance of the biological object and the square of
the imaginary part of the bioimpedance of the biological object;
and calculating a phase of the bioimpedance of the biological
object plus a measurement system phase shift, by calculating the
arctan of the ratio of the imaginary part of the bioimpedance of
the biological object to the real part of the bioimpedance of the
biological object.
6. The method of claim 4, wherein the operation of digitizing the
voltage that is directly proportional to, and in phase with, the
output of the AC current source produces unfitted Object Current
Samples, and wherein the operations further comprise fitting the
unfitted Object Current Samples to discrete values of an ideal
sinusoid to produce the Object Current Samples.
7. The method of claim 4, wherein the operations further comprise:
providing, over time, a set of digital bioimpedance waveforms
Z(f.sub.AC,t)); separating a base bioimpedance Z.sub.0(f.sub.AC)
from the waveforms; separating changes of bioimpedance
.DELTA.Z(f.sub.AC,t) from the waveforms; determining a rate of
change of the bioimpedance dZ(f.sub.AC,t)/dt; and recording a
temporal course of the base bioimpedance Z.sub.0(f.sub.AC) and of
the rate of change of the bioimpedance dZ(f.sub.AC,t)/dt.
8. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of an object, wherein the
method comprises the following operations: placing a first current
electrode and a second current electrode in contact with the
object; placing a first voltage sensing electrode and a second
voltage sensing electrode in contact with the object; applying the
output of an AC current source to the first current electrode and
the second current electrode, to cause an AC current to flow
through the object between the first current electrode and the
second current electrode; measuring a voltage between the first
voltage sensing electrode and the second voltage sensing electrode,
wherein the measured voltage is produced due to application of the
output of the AC current source to the first current electrode and
the second current electrode; digitizing the measured voltage to
produce Object Voltage Samples; calculating an in-phase portion of
the AC current through the object, by correlating the Object
Current Samples with corresponding discrete values of a unity sine
waveform; and calculating a quadrature portion of the AC current
through the object, by correlating the Object Current Samples with
corresponding discrete values of a unity cosine waveform.
9. The method of claim 8, further comprising: calculating a AC
current magnitude through the object, by calculating the square
root of the sum the squared in-phase portion of the AC current
through the object and the squared quadrature portion of the AC
current through the object; calculating a phase of the AC current
through the object plus measurement system current phase shift, by
calculating the arctan of the ratio of the quadrature portion of
the AC current through the object and the in-phase portion of the
AC current through the object; calculating an in-phase portion of
the voltage, by correlating the Object Voltage Samples with
corresponding discrete values of a unity sine waveform; and
calculating a quadrature portion of the voltage, by correlating the
Object Voltage Samples with corresponding discrete values of a
unity cosine waveform.
10. The method of claim 9, further comprising: calculating a
voltage magnitude across the object, by calculating the square root
of the sum the squared in-phase portion of the voltage and the
squared quadrature portion of the voltage; calculating a phase of
the voltage across the object plus measurement system voltage phase
shift, by calculating the arctan of the ratio of the quadrature
portion of the voltage and the in-phase portion of the voltage;
calculating a magnitude of the bioimpedance of the object, by
calculating the ratio of the voltage magnitude across the object to
the AC current magnitude through the object; and calculating a
phase of the bioimpedance of the object plus a measurement system
phase shift, by subtracting the phase of the AC current through the
object plus measurement system current phase shift, from the phase
of the voltage across the object plus measurement system voltage
phase shift.
11. The method of claim 10: wherein the object is a human being;
and wherein the AC current has a plurality of frequencies, and
wherein the applying, measuring, digitizing, and calculating
operations of claims 8, 9 and 10 are repeated for each frequency of
the AC current.
12. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of a biological object,
wherein the method comprises the following operations: applying the
output of an AC current source to a calibration impedance, to cause
an AC current to flow through the calibration impedance; measuring
a voltage across the calibration impedance, wherein the measured
voltage is produced due to application of the AC current source to
the calibration impedance; digitizing the voltage measured across
the calibration impedance to produce Calibration Voltage Samples;
calculating a value proportional to an in-phase portion of the
calibration impedance uncorrected for measurement system phase
shift, by correlating the Calibration Voltage Samples with
corresponding discrete values of a unity sine waveform; and
calculating a value proportional to a quadrature portion of the
calibration impedance uncorrected for measurement system phase
shift, by correlating the Calibration Voltage Samples with
corresponding discrete values of a unity cosine waveform.
13. The method of claim 12, further comprising: calculating a
magnitude of an equivalent to the calibration impedance, by
calculating the square root of the sum of the squared in-phase
portion of the calibration impedance and the squared quadrature
portion of the calibration impedance; calculating a phase of the
calibration impedance including measurement system phase shift, by
calculating the arctan of the ratio of the quadrature portion of
the calibration impedance and the in-phase portion of the
calibration impedance; placing a first current electrode and a
second current electrode in contact with the biological object;
placing a first voltage sensing electrode and a second voltage
sensing electrode in contact with the biological object; applying
the output of the AC current source to the first current electrode
and the second current electrode, to cause an AC current to flow
through the biological object between the first current electrode
and the second current electrode; measuring a voltage between the
first voltage sensing electrode and the second voltage sensing
electrode, wherein the measured voltage is produced due to
application of the output of the AC current source to the first
current electrode and the second current electrode; digitizing the
voltage measured between the first voltage sensing electrode and
the second voltage sensing electrode to produce Object Voltage
Samples; calculating a value proportional to an in-phase portion of
the biological object bioimpedance, by correlating the Object
Voltage Samples with corresponding discrete values of a unity sine
waveform; and calculating a value proportional to a quadrature
portion of the biological object bioimpedance, by correlating the
Object Voltage Samples with corresponding discrete values of a
unity cosine waveform.
14. The method of claim 13, further comprising: calculating a
magnitude of an equivalent to the biological object bioimpedance,
by calculating the square root of the sum of the squared in-phase
portion of the biological object bioimpedance and the squared
quadrature portion of the biological object bioimpedance;
calculating a phase of the biological object bioimpedance plus
measurement system phase shift, by calculating the arctan of the
ratio of the quadrature portion of the biological object
bioimpedance and the in-phase portion of the biological object
bioimpedance; and calculating the magnitude of the biological
object bioimpedance, by determining the ratio of a known
calibration impedance magnitude to the calibration impedance
magnitude equivalent, multiplied by the magnitude equivalent of the
biological object bioimpedance.
15. The method of claim 14: wherein the operation of digitizing the
voltage measured across the calibration impedance produces unfitted
Calibration Voltage Samples, and wherein the operations further
comprise fitting the unfitted Calibration Voltage Samples to
discrete values of an ideal sinusoid to produce the Calibration
Voltage Samples; and wherein the operation of digitizing the
voltage measured between the first voltage sensing electrode and
the second voltage sensing electrode produces unfitted Object
Voltage Samples, and wherein the operations further comprise
fitting the unfitted Object Voltage Samples to discrete values of
an ideal sinusoid to produce the Object Voltage Samples.
16. The method of claim 14, wherein the operations further comprise
calculating the phase of the biological object bioimpedance by
determining the difference between the phase of the calibration
impedance, and the phase of the biological object bioimpedance plus
measurement system phase shift.
17. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of a biological object,
wherein the method comprises the following operations: applying the
output of an AC current source to a calibration impedance, to cause
an AC current to flow through the calibration impedance; producing
a voltage that is directly proportional to, and in phase with, the
output of the AC current source; digitizing the voltage that is
directly proportional to, and in phase with, the output of the AC
current source, to produce Calibration Current Samples; measuring a
voltage across the calibration impedance, wherein the measured
voltage is produced due to application of the AC current source to
the calibration impedance; digitizing the voltage measured across
the calibration impedance to produce Calibration Voltage Samples;
obtaining a value proportional to the real part of the calibration
impedance, by correlating the Calibration Voltage Samples with the
Calibration Current Samples; obtaining a value proportional to the
imaginary part of the calibration impedance, by correlating the
Calibration Voltage Samples with corresponding Calibration Current
Samples that are shifted in phase by -90 degrees; calculating a
calibration impedance magnitude equivalent, by calculating the
square root of the sum of the square of the real part of the
calibration impedance and the square of the imaginary part of the
calibration impedance; and calculating a calibration impedance
phase plus a measurement system phase shift, by calculating the
arctan of the ratio of the imaginary part of the calibration
impedance to the real part of the calibration impedance.
18. The method of claim 17, further comprising: placing a first
current electrode and a second current electrode in contact with
the biological object; placing a first voltage sensing electrode
and a second voltage sensing electrode in contact with the
biological object; applying the output of the AC current source to
the first current electrode and the second current electrode, to
cause an AC current to flow through the biological object between
the first current electrode and the second current electrode;
producing a voltage that is directly proportional to, and in phase
with, the output of the AC current source that is applied to the
first current electrode and the second current electrode;
digitizing the voltage that is directly proportional, and in phase
with, the output of the AC current source, to produce Object
Current Samples; measuring a voltage between the first voltage
sensing electrode and the second voltage sensing electrode, wherein
the measured voltage is produced due to application of the output
of the AC current source to the first current electrode and the
second current electrode; and digitizing the voltage measured
between the first voltage sensing electrode and the second voltage
sensing electrode to produce Object Voltage Samples.
19. The method of claim 18, further comprising: obtaining a real
part of the bioimpedance of the biological object, uncorrected for
measurement system phase shift, by correlating the Object Voltage
Samples with the Object Current Samples; obtaining an imaginary
part of the bioimpedance of the biological object, uncorrected for
measurement system phase shift, by correlating the Object Voltage
Samples with corresponding Object Current Samples that are shifted
in phase by -90 degrees; calculating a biological object
bioimpedance magnitude equivalent, by calculating the square root
of the sum of the square of the real part of the bioimpedance of
the biological object and the square of the imaginary part of the
bioimpedance of the biological object; calculating a phase of the
bioimpedance of the biological object uncorrected for measurement
system phase shift, by calculating the arctan of the ratio of the
imaginary part of the bioimpedance of the biological object to the
real part of the bioimpedance of the biological object; and
calculating a magnitude of the biological object bioimpedance, by
determining the ratio of a known calibration impedance magnitude
and the calibration impedance magnitude equivalent, multiplied by
the magnitude equivalent of the biological object bioimpedance.
20. The method of claim 19, wherein the phase of the biological
object bioimpedance is the difference between the uncorrected phase
of the biological object bioimpedance, and the phase of the
calibration impedance.
21. The method of claim 19: wherein, when the output of the AC
current source is applied to the calibration impedance, the
operation of digitizing the voltage that is directly proportional
and in phase with the output of the AC current source produces
unfitted Calibration Current Samples, and wherein the operations
further comprise fitting the unfitted Calibration Current Samples
to discrete values of an ideal sinusoid to produce the Calibration
Current Samples; wherein the operation of digitizing the voltage
measured across the calibration impedance produces unfitted
Calibration Voltage Samples, and wherein the operations further
comprise filling the unfitted Calibration Voltage Samples to
discrete values of an ideal sinusoid to produce the Calibration
Voltage Samples; wherein, when the output of the AC current source
is applied to the first current electrode and the second current
electrode, the operation of digitizing the voltage that is directly
proportional and in phase with the output of the AC current source
produces unfitted Object Current Samples, and wherein the
operations further comprise fitting the unfitted Object Current
Samples to discrete values of an ideal sinusoid to produce the
Object Current Samples; and wherein the operation of digitizing the
voltage measured between the first voltage sensing electrode and
the second voltage sensing electrode, produces unfitted Object
Voltage Samples, and wherein the operations further comprise
fitting the unfitted Object Voltage Samples to discrete values of
an ideal sinusoid to produce the Object Voltage Samples.
22. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of an object, wherein the
method comprises the following operations: applying the output of
an AC current source to a calibration impedance, to cause an AC
current to flow through the calibration impedance; producing a
voltage that is directly proportional to, and in phase with, the
output of the AC current source; digitizing the voltage that is
directly proportional to, and in phase with, the output of the AC
current source, to produce Calibration Current Samples; measuring a
voltage across the calibration impedance, wherein the measured
voltage is produced due to application of the output of the AC
current source to the calibration impedance; digitizing the voltage
measured across the calibration impedance to produce Calibration
Voltage Samples; calculating a value proportional to an in-phase
portion of the calibration current, by correlating the Calibration
Current Samples with corresponding discrete values of a unity sine
waveform; and calculating a value proportional to a quadrature
portion of the calibration current, by correlating the Calibration
Current Samples with corresponding discrete values of a unity
cosine waveform.
23. The method of claim 22, further comprising: calculating a
magnitude of an equivalent to the current through the calibration
impedance, by calculating the square root of the sum of the squared
in-phase portion of the calibration current and the squared
quadrature portion of the calibration current; and calculating a
phase of the calibration current including measurement system phase
shift, by calculating the arctan of the ratio of the quadrature
portion of the calibration current and the in-phase portion of the
calibration current.
24. The method of claim 23, further comprising: calculating a value
proportional to an in-phase portion of the calibration voltage
uncorrected for measurement system phase shift, by correlating the
Calibration Voltage Samples with corresponding discrete values of a
unity sine waveform; calculating a value proportional to a
quadrature portion of the calibration voltage uncorrected for
measurement system phase shift, by correlating the Calibration
Voltage Samples with corresponding discrete values of a unity
cosine waveform; calculating a magnitude of an equivalent to the
voltage across the calibration impedance, by calculating the square
root of the sum of the square of the value proportional to the
in-phase portion of the calibration voltage and the square of the
value proportional to the quadrature portion of the calibration
voltage; calculating a phase of the calibration voltage including
measurement system phase shift, by calculating the arctan of the
ratio of the value proportional to the quadrature portion of the
calibration voltage and the value proportional to the in-phase
portion of the calibration voltage; and calculating an equivalent
to the calibration impedance magnitude by calculating the ratio of
the voltage magnitude equivalent across the calibration impedance
and the magnitude of the current magnitude equivalent through the
calibration impedance.
25. The method of claim 24, further comprising: placing a first
current electrode and a second current electrode in contact with
the object; placing a first voltage sensing electrode and a second
voltage sensing electrode in contact with the object; applying the
output of the AC current source to the first current electrode and
the second current electrode, to cause an AC current to flow
through the object between the first current electrode and the
second current electrode; producing a voltage that is directly
proportional to, and in phase with, the output of the AC current
source that is applied to the first current electrode and the
second current electrode; digitizing the voltage that is directly
proportional to, and in phase with, the output of the AC current
source, to produce Object Current Samples; calculating an in-phase
portion of the AC current through the object uncorrected for
measurement system phase shift, by correlating the Object Current
Samples with corresponding discrete values of a unity sine
waveform; calculating a quadrature portion of the AC current
through the object uncorrected for measurement system phase shift,
by correlating the Object Current Samples with corresponding
discrete values of a unity cosine waveform; calculating an
equivalent of the object current magnitude through the object, by
calculating the square root of the sum the squared in-phase portion
of the current and the squared quadrature portion of the current
through the object; and calculating a phase of the object current
including measurement system phase shift, by calculating the arctan
of the ratio of the quadrature portion of the current and the
in-phase portion of the current through the object.
26. The method of claim 25, further comprising: measuring a voltage
between the first voltage sensing electrode and the second voltage
sensing electrode, wherein the measured voltage is produced due to
application of the output of the AC current source to the first
current electrode and the second current electrode; digitizing the
voltage measured between the first voltage sensing electrode and
the second voltage sensing electrode to produce Object Voltage
Samples; calculating an in-phase portion of the object voltage
uncorrected for measurement system phase shift, by correlating the
Object Voltage Samples with corresponding discrete values of a
unity sine waveform; calculating a quadrature portion of the object
voltage uncorrected for measurement system phase shift, by
correlating the Object Voltage Samples with corresponding discrete
values of a unity cosine waveform; calculating an equivalent
voltage magnitude across the object, by calculating the square root
of the sum of the squared in-phase portion of the uncorrected
object voltage and the squared quadrature portion of the
uncorrected object voltage; calculating a phase of the voltage
across the object plus measurement system voltage phase shift, by
calculating the arctan of the ratio of the quadrature portion of
the object voltage and the in-phase portion of the object voltage;
calculating the magnitude equivalent of the bioimpedance of the
object, by calculating the ratio of the voltage equivalent
magnitude across the object to the current magnitude equivalent
through the object, multiplied by the cosine of the phase shift
between the voltage across the object bioimpedance and the current
through the object bioimpedance; and calculating the magnitude of
the object bioimpedance by calculating the ratio of a previously
known calibration impedance magnitude, to the magnitude equivalent
of the calibration impedance, multiplied by the magnitude
equivalent of the object bioimpedance.
27. The method of claim 26, wherein the operations further
comprise: providing, over time, a set of digital bioimpedance
waveforms Z(f.sub.AC,t)); separating a base bioimpedance
Z.sub.0(f.sub.AC) from the waveforms; separating changes of
bioimpedance .DELTA.Z(f.sub.AC,t) from the waveforms; determining a
rate of change of the bioimpedance dZ(f.sub.AC,t)/dt; and recording
a temporal course of the base bioimpedance Z.sub.0(f.sub.AC) and of
the rate of change of the bioimpedance dZ(f.sub.AC,t)/dt.
28. The method of claim 26: wherein the AC current has a plurality
of frequencies; and wherein the measurement system phase shift is
the difference between the phase of the calibration voltage and the
phase of the applied AC voltage.
29. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of an object, wherein the
method comprises the following operations: generating an excitation
signal of known frequency content; applying the excitation signal
to the object; sensing a response signal of the object; sampling
and digitizing the response signal to acquire a digitized response
signal representing the response signal with respect to frequency
content, amplitude and phase; correlating, for each frequency
f.sub.AC of the excitation signal applied, digitized samples of the
digitized response signal, with discrete values representing the
excitation signal; calculating, using the correlated signals for
each frequency f.sub.AC of the excitation signal applied, complex
values for the bioimpedance Z(f.sub.AC); providing, over time, a
set of digital bioimpedance waveforms Z(f.sub.AC,t)); separating
the base bioimpedance Z.sub.0(f.sub.AC) from the waveforms; and
separating the changes of bioimpedance .DELTA.Z(f.sub.AC,t) from
the waveforms.
30. The method of claim 29, wherein the operations further
comprise: determining a rate of change of the bioimpedance
dZ(f.sub.AC,t)/dt; and recording a temporal course of the base
bioimpedance Z.sub.0(f.sub.AC) and of the rate of change of the
bioimpedance dZ(f.sub.AC,t)/dt.
31. The method of claim 30, wherein, in the correlating operation,
the discrete values representing the excitation signal are discrete
values of a sinusoidal reference signal for the excitation
signal.
32. The method of claim 29, wherein the operations further
comprise: determining a rate of change of the changes in
bioimpedance d(.DELTA.Z(f.sub.AC,t))/dt; and recording a temporal
course of the base bioimpedance Z.sub.0(f.sub.AC) and of the
changes of the bioimpedance .DELTA.Z(f.sub.AC,t).
33. The method of claim 32, wherein, in the correlating operation,
the discrete values representing the excitation signal are discrete
values of a sinusoidal reference signal for the excitation
signal.
34. The method of claim 29, wherein the excitation signal is a
sinusoidal signal of a known single frequency f.sub.AC.
35. The method of claim 29, wherein the excitation signal has an
amplitude and a phase that are substantially constant over
time.
36. The method of claim 35, wherein the excitation signal has an
alternating current (AC) amplitude in a range of about 0.01 mA to
about 5 mA.
37. The method of claim 29, wherein the operations further comprise
applying the excitation signal to a calibration impedance.
38. The method of claim 37, wherein the calibration impedance is an
ohmic resistor.
39. The method of claim 29, wherein the operation of generating the
excitation signal comprises: reading discrete values of at least
one sinusoidal waveform, stored in an addressable look-up table;
converting the discrete values of the at least one sinusoidal
waveform into analog signals having a desired frequency content,
amplitude, and phase.
40. The method of claim 29, wherein the operation of generating the
excitation signal comprises: employing time-controlled direct
digital synthesizing (DDS); and driving an excitation source.
41. The method of claim 29, wherein the excitation signal contains
frequencies in a range of about 1 kHz to about 1 MHz.
42. The method of claim 29, wherein the excitation signal contains
frequencies in a range of about 10 kHz to about 200 kHz.
43. The method of claim 29, wherein the response signal is sampled
at a rate that is in a range from about 4 to about 20 times the
highest frequency of the excitation signal.
44. The method of claim 29, wherein the response signal is sampled
at a rate that is about 10 times the highest frequency of the
excitation signal.
45. The method of claim 29, wherein the discrete values
representing the excitation signal are produced at a sampling rate
that is in a range from about 4 to about 20 times the highest
frequency of the excitation signal.
46. The method of claim 29, wherein the discrete values
representing the excitation signal are produced at a sampling rate
that is about 10 times the highest frequency of the excitation
signal.
47. The method of claim 29, wherein the operation of separating the
base bioimpedance Z.sub.0(f.sub.AC) from the waveforms comprises
inputting the set of digital bioimpedance waveforms Z(f.sub.AC,t)
to a low pass filter to obtain the base bioimpedance
Z.sub.0(f.sub.AC)
48. The method of claim 29, wherein the operation of separating the
changes of bioimpedance .DELTA.Z(f.sub.AC,t) from the waveforms
comprises inputting the set of digital bioimpedance waveforms
Z(f.sub.AC,t) to a high pass filter to obtain the changes of
bioimpedance .DELTA.Z(f.sub.AC,t).
49. The method of claim 29, wherein the operations further comprise
inputting the set of digital bioimpedance waveforms Z(f.sub.AC,t)
to a differentiator to obtain a rate of change of the changes in
bioimpedance d(.DELTA.Z(f.sub.AC,t))/dt.
50. The method of claim 29, wherein the operations further comprise
inputting the set of digital bioimpedance waveforms Z(f.sub.AC,t)
to a differentiator to obtain a rate of change in the bioimpedance
waveforms dZ(f.sub.AC,t)/dt.
51. The method of claim 29, wherein separate correlation processes
are used for determining, respectively, an in-phase portion
Re(Z(f.sub.AC,t)) and a quadrature portion Im(Z(f.sub.AC,t)) of the
bioimpedance of the object.
52. A method for digital demodulation of signals obtained in the
measurement of electrical bioadmittance of an object, wherein the
method comprises the following operations: generating an excitation
signal of known frequency content; applying the excitation signal
to the object; sensing a response signal of the object; sampling
and digitizing the response signal to acquire a digitized response
signal representing the response signal with respect to frequency
content, amplitude and phase; correlating, for each frequency
f.sub.AC of the excitation signal applied, digitized samples of the
digitized response signal, with discrete values representing the
excitation signal; calculating, using the correlated signals for
each frequency f.sub.AC of the excitation signal applied, complex
values for the bioadmittance Y(f.sub.AC); providing, over time, a
set of digital bioadmittance waveforms Y(f.sub.AC,t); separating
the base bioadmittance Y.sub.0(f.sub.AC) from the waveforms; and
separating the changes of bioadmittance .DELTA.Y(f.sub.AC,t) from
the waveforms.
53. The method of claim 52, wherein the operations further
comprise: determining a rate of change of the bioadmittance
dY(f.sub.AC,t)/dt; and recording a temporal course of the base
bioadmittance Y.sub.0(f.sub.AC) and of the rate of change of the
bioadmittance dY(f.sub.AC,t)/dt.
54. The method of claim 52, wherein the operations further
comprise: determining a rate of change of the changes in
bioadmittance d(.DELTA.Y(f.sub.AC,t))/dt; and recording a temporal
course of the base bioadmittance Y.sub.0(f.sub.AC) and of the
changes of bioadmittance .DELTA.Y(f.sub.AC,t).
55. The method of claim 52, wherein the operation of separating the
base bioadmittance Y.sub.0(f.sub.AC) from the waveforms comprises
inputting the set of digital bioadmittance waveforms Y(f.sub.AC,t)
to a low pass filter to obtain the base bioadmittance
Y.sub.0(f.sub.AC)
56. The method of claim 52, wherein the operation of separating the
changes in bioadmittance .DELTA.Y(f.sub.AC,t) from the waveforms
comprises inputting the set of digital bioadmittance waveforms
Y(f.sub.AC,t) to a high pass filter to obtain the changes of
bioadmittance .DELTA.Y(f.sub.AC,t).
57. The method of claim 52, wherein the operations further comprise
inputting the set of digital bioadmittance waveforms Y(f.sub.AC,t)
to a differentiator to obtain a rate of change of the changes in
bioadmittance d(.DELTA.Y(f.sub.AC,t))/dt.
58. The method of claim 52, wherein the operations further comprise
inputting the set of digital bioadmittance waveforms Y(f.sub.AC,t)
to a differentiator to obtain a rate of the change in the
bioadmittance waveforms dY(f.sub.AC,t)/dt.
59. A method for digital demodulation of signals obtained in the
measurement of electrical bioimpedance of an object, wherein the
method comprises the following operations: applying a calibration
excitation signal to a calibration impedance; measuring, sampling
and digitizing a signal representing the calibration excitation
signal to acquire calibration Excitation Signal Samples; measuring,
sampling and digitizing a calibration response signal across the
calibration impedance to acquire calibration Response Signal
Samples; for each frequency f.sub.AC of the calibration excitation
signal applied to the calibration impedance, correlating the
calibration Excitation Signal Samples with discrete values of an
ideal sine waveform in order to obtain a value proportional to an
in-phase portion of the calibration excitation signal related to
the ideal sine waveform as a reference sine; and for each frequency
f.sub.AC of the calibration excitation signal applied to the
calibration impedance, correlating the calibration Excitation
Signal Samples with discrete values of an ideal cosine waveform in
order to obtain a value proportional to a quadrature portion of the
calibration excitation signal wherein the AC current has a
plurality of frequencies.
60. The method of claim 59, further comprising: correlating the
calibration Response Signal Samples with discrete values of an
ideal sine waveform in order to obtain a value proportional to an
in-phase portion of the calibration response signal; and
correlating the calibration Response Signal Samples with discrete
values of an ideal cosine waveform in order to obtain a value
proportional to a quadrature portion of the calibration response
signal.
61. The method of claim 60, further comprising: calculating an
equivalent for a magnitude and a phase of the calibration
excitation signal; calculating an equivalent for a magnitude and a
phase of the calibration response signal; calculating an equivalent
for a magnitude of the calibration impedance; calculating a system
phase; applying an object excitation signal to the object after the
operation of calculating the system phase; measuring, sampling and
digitizing a signal representing the object excitation signal to
acquire object Excitation Signal Samples; measuring, sampling and
digitizing the object response signal across the bioimpedance of
the object, wherein the samples obtained from sampling the object
response signal are called object Response Signal Samples; for each
frequency f.sub.AC of the object excitation signal applied,
correlating the object Excitation Signal Samples with discrete
values of an ideal sine waveform in order to obtain a value
proportional to an in-phase portion of the object excitation signal
related to the ideal sine waveform; for each frequency f.sub.AC of
the excitation signal applied, correlating the object Excitation
Signal Samples with discrete values of an ideal cosine waveform in
order to obtain a value proportional to a quadrature portion of the
object excitation signal; correlating the object Response Signal
Samples with discrete values of another ideal sine waveform in
order to obtain a value proportional to an in-phase portion of the
object response signal; correlating the object Response Signal
Samples with discrete values of another ideal cosine waveform in
order to obtain a value proportional to a quadrature portion of the
object response signal; calculating an equivalent for the magnitude
and a phase of the object excitation signal; calculating an
equivalent for the magnitude and a phase of the object response
signal; calculating an equivalent for the magnitude and a phase of
the bioimpedance of the object; calculating a magnitude of the
bioimpedance Z(f.sub.AC,t) of the object; and calculating an
in-phase portion Re(Z(f.sub.AC,t)) and a quadrature portion
Im(Z(f.sub.AC,t)) of the bioimpedance of the object.
62. The method of claim 61, further comprising calculating a
cross-correlation signal, wherein the cross-correlation signal is a
function of a time delay .tau. between the object excitation signal
and the object response signal, wherein the cross-correlation
signal is calculated by correlating the object excitation signal
with the object response signal after delay of the object response
signal by the time delay .tau. with respect to the object
excitation signal.
63. The method of claim 61, further comprising calculating a
complex Fourier transform of the cross-correlation signal, to
obtain complex values proportional to a complex bioimpedance.
64. An apparatus for digital demodulation and further processing of
signals obtained to measure electrical bioimpedance in an object,
the apparatus comprising: a voltage controlled current source for
generating an excitation signal of known frequency content; a
current monitor coupled to the voltage controlled current source; a
first electrode coupled to the current monitor; a second electrode
coupled to the current monitor, wherein the first and second
electrodes are configured for applying the excitation signal to the
object; a differential amplifier; a third electrode coupled to the
differential amplifier; a fourth electrode coupled to the
differential amplifier, wherein the third and fourth electrodes are
configured for sensing a response signal across the object due to
application of the excitation signal; a first analog to digital
converter coupled to the differential amplifier, wherein the
differential amplifier and the first analog to digital converter
are configured for acquiring, sampling and digitizing the response
signal, to obtain a digitized response signal representing the
response signal with respect to frequency content, amplitude and
phase; a first buffer coupled to the first analog to digital
converter for temporarily storing the digitized response signal; a
multiplier/accumulator coupled to the first buffer, for
correlating, for each frequency f.sub.AC of the excitation signal
applied, digitized samples of the digitized response signal, with
corresponding discrete values representing the excitation signal;
and a processing unit coupled to the multiplier/accumulator,
wherein the processing unit is configured to calculate, for each
frequency f.sub.AC of the excitation signal applied, complex values
for the bioimpedance, from output values received from the
multiplier/accumulator, and is further configured to provide, over
time, a set of digital bioimpedance waveforms.
65. The apparatus of claim 64, further comprising: a filter coupled
to the processing unit; a monitor coupled to the filter; a second
analog to digital converter coupled to the current monitor, wherein
the current monitor and the second analog to digital converter are
configured for acquiring, sampling and digitizing the excitation
signal to obtain a digitized excitation signal representing the
excitation signal with respect to frequency content, amplitude and
phase; and a second buffer coupled to the second analog to digital
converter, for temporarily storing the digitized excitation
signal.
66. The apparatus of claim 65, wherein the second analog to digital
converter is adapted to sample at a rate that is in a range from
about 4 to about 20 times the highest frequency of the excitation
signal.
67. The apparatus of claim 65, further comprising: a timing
control; a plurality of multiplier/accumulators coupled to the
timing control and the first buffer, for correlating, for each
frequency f.sub.AC of the excitation signal applied, digitized
samples of the digitized response signal, with corresponding
discrete values representing the excitation signal.
68. The apparatus of claim 67, wherein each multiplier/accumulator
outputs a respective digital waveform, and wherein each
multiplier/accumulator is coupled to a first filter set adapted to
separate the base bioimpedance Z.sub.0(f.sub.AC) from each digital
waveform.
69. The apparatus of claim 67, wherein each multiplier/accumulator
outputs a respective digital waveform, and wherein each
multiplier/accumulator is coupled to a second filter set adapted to
separate the changes in the bioimpedance .DELTA.Z(f.sub.AC,t) from
each digital waveform, and wherein each multiplier/accumulator is
coupled to a differentiator for obtaining a rate of change of the
changes in bioimpedance.
70. The apparatus of claim 67, wherein each multiplier/accumulator
outputs a respective digital waveform, and wherein each
multiplier/accumulator is coupled to a differentiator for obtaining
a rate of change in the bioimpedance waveforms dZ(f.sub.AC,t)/dt of
the object.
71. The apparatus of claim 67, wherein: the plurality of
multiplier/accumulators comprises first, second, third, and fourth
multiplier/accumulators; the first multiplier/accumulator in the
plurality of multiplier/accumulators is configured to determine,
for a first frequency f.sub.AC of the excitation signal applied, an
in-phase portion Re(Z(f.sub.AC,t)) of the bioimpedance
Z(f.sub.AC,t); the second multiplier/accumulator in the plurality
of multiplier/accumulators is configured to determine, for the
first frequency f.sub.AC of the excitation signal applied, a
quadrature portion Im(Z(f.sub.AC,t)) of the bioimpedance
z(f.sub.AC,t); the third multiplier/accumulator in the plurality of
multiplier/accumulators is configured to determine, for a second
frequency f.sub.AC of the excitation signal applied, an in-phase
portion Re(Z(f.sub.AC,t)) of the bioimpedance z(f.sub.AC,t); and
the fourth multiplier/accumulator in the plurality of
multiplier/accumulators is configured to determine, for the second
frequency f.sub.AC of the excitation signal applied, a quadrature
portion Im(Z(f.sub.AC,t)) of the bioimpedance Z(f.sub.AC,t).
72. The apparatus of claim 67, wherein the discrete values
representing the excitation signal are discrete values of a
sinusoidal reference signal for the excitation signal.
73. The apparatus of claim 67, further comprising a differentiator
coupled to the processing unit.
74. The apparatus of claim 64, wherein the voltage controlled
current source and the current monitor are configured to generate a
sinusoidal excitation signal of a known single frequency
f.sub.AC.
75. The apparatus of claim 64, further comprising: a first switch
coupled to the current monitor and the first electrode; a second
switch coupled to the current monitor and the second electrode; a
third switch coupled to the differential amplifier and the third
electrode; a fourth switch coupled to the differential amplifier
and the fourth electrode; and a calibration impedance coupled to
the first switch, the second switch, the third switch, and the
fourth switch.
76. The apparatus of claim 64, further comprising: a sine table;
and a digital to analog converter coupled to the sine table and the
voltage controlled current source, for generating the excitation
signal.
77. The apparatus of claim 64, wherein the voltage controlled
current source and the current monitor are configured to generate a
sinusoidal excitation signal in a range of frequencies from about 1
kHz to about 1 MHz.
78. The apparatus of claim 64, wherein the voltage controlled
current source and the current monitor are configured to generate
an excitation alternating current (AC) having an amplitude in a
range from about 0.01 mA to about 5 mA.
79. An apparatus for digital demodulation and further processing of
signals obtained to measure electrical bioadmittance in an object,
the apparatus comprising: a voltage controlled current source for
generating an excitation signal of known frequency content; a
current monitor coupled to the voltage controlled current source; a
first electrode coupled to the current monitor; a second electrode
coupled to the current monitor, the first and second electrodes for
applying the excitation signal to the object; a differential
amplifier; a third electrode coupled to the differential amplifier;
a fourth electrode coupled to the differential amplifier, the third
and fourth electrodes for sensing a response signal across the
object due to the application of the excitation signal; a first
analog to digital converter coupled to the differential amplifier,
the differential amplifier and the first analog to digital
converter for acquiring, sampling and digitizing the response
signal to obtain a digitized response signal representing the
response signal with respect to frequency content, amplitude and
phase; a first buffer coupled to the first analog to digital
converter for temporarily storing the digitized response signal; a
multiplier/accumulator coupled to the first buffer, for
correlating, for each frequency f.sub.AC of the excitation signal
applied, digitized samples of the digitized response signal, with
corresponding discrete values representing the excitation signal; a
processing unit coupled to the multiplier/accumulator, for
calculating, for each frequency f.sub.AC of the excitation signal
applied, complex values for the bioadmittance, from output values
received from the multiplier/accumulator, and for providing, over
time, a set of digital bioadmittance waveforms; a filter coupled to
the processing unit; and a monitor coupled to the filter.
80. An apparatus for digital demodulation and further processing of
signals obtained to measure electrical bioimpedance in an object,
the apparatus comprising: signal generating means for generating an
excitation signal of known frequency content; a first pair of
electrodes for applying the excitation signal to the object; a
second pair of electrodes for sensing the response signal across
the object due to the application of the excitation signal; first
measuring means for acquiring, sampling and digitizing the response
signal to obtain a digitized response signal representing the
response signal with respect to frequency content, amplitude and
phase; memory means for temporarily storing the digitized response
signal; and digital demodulation means for correlating, for each
frequency f.sub.AC of the excitation signal applied, digitized
samples of the response signal with corresponding discrete values
of a sinusoidal reference signal to the excitation signal.
81. The apparatus of claim 80, further comprising: processing means
for calculating for each frequency f.sub.AC of the excitation
signal applied, complex values for the bioimpedance Z(f.sub.AC)
from output values of the digital demodulation means, and for
providing, over time, a set of digital bioimpedance waveforms
Z(f.sub.AC,t); and recording means for recording a temporal course
of the base bioimpedance.
82. The apparatus of claim 81, further comprising: means for, for
each frequency f.sub.AC of the excitation signal applied,
separating the base bioimpedance Z.sub.0(f.sub.AC) from the
corresponding bioimpedance waveform Z(f.sub.AC,t); and means for,
for each frequency f.sub.AC of the excitation signal applied,
separating the changes of bioimpedance .DELTA.Z(f.sub.AC,t) from
the corresponding bioimpedance waveform Z(f.sub.AC,t).
83. The apparatus of claim 82, further comprising: means for
determining a rate of change of the bioimpedance dZ(f.sub.AC,t)/dt,
for each frequency f.sub.AC of the excitation signal applied; and
means for recording a temporal course of the base bioimpedance
Z.sub.0(f.sub.AC) and of the rate of change of the bioimpedance
dZ(f.sub.AC,t)/dt, for each frequency f.sub.AC of the excitation
signal applied.
84. The apparatus of claim 83, further comprising: means for
determining a rate of change of the changes in bioimpedance
d(.DELTA.Z(f.sub.AC,t))/dt, for each frequency f.sub.AC of the
excitation signal applied; and means for recording a temporal
course of the base bioimpedance Z.sub.0(f.sub.AC) and of the
changes of the bioimpedance .DELTA.Z(f.sub.AC,t), for each
frequency f.sub.AC of the excitation signal applied.
85. The apparatus of claim 80, further comprising an addressable
sine look-up table coupled to the signal generating means, and
wherein the signal generating means is adapted to generate the
excitation signal using discrete values of a sinusoidal
waveform.
86. The apparatus of claim 80, further comprising an addressable
sine look-up table coupled to the signal generating means, and
wherein the signal generating means is adapted to generate the
excitation signal by superposition of a number of sinusoidal
waveforms.
87. An apparatus for digital demodulation and further processing of
signals obtained to measure electrical bioimpedance in an object,
the apparatus comprising: signal generating means for generating an
excitation signal of known frequency content; a first pair of
electrodes for applying the excitation signal to the object; a
second pair of electrodes for sensing a response signal across the
object due to the application of the excitation signal; first
measuring means for acquiring, sampling and digitizing the response
signal to obtain a digitized response signal representing the
response signal with respect to frequency content, amplitude and
phase; second measuring means for acquiring, sampling and
digitizing the excitation signal to obtain a digitized excitation
signal representing the excitation signal with respect to frequency
content, amplitude and phase; memory means for temporarily storing
the digitized response signal; processing means for calculating for
each frequency f.sub.AC of the excitation signal applied, complex
values for the bioimpedance Z(f.sub.AC) from the output values of
the digital demodulation means, and for providing, over time, a set
of digital bioimpedance waveforms Z(f.sub.AC,t)); differentiating
means for obtaining the rate of change in the bioimpedance
waveforms dZ(f.sub.AC,t)/dt, and recording means for recording the
rate of change in bioimpedance waveforms.
88. The apparatus of claim 87, further comprising: a calibration
impedance; a plurality of means for switching, for selectively
coupling the signal generating means and the first measuring means
to the calibration impedance, to acquire Calibration Current
Samples and Calibration Voltage Samples; wherein the signal
generating means generates an alternating current (AC); first
correlating means for correlating, for each frequency f.sub.AC of
the alternating current (AC) applied to the calibration impedance,
the Calibration Current Samples with discrete values of an ideal
sine waveform in order to obtain a value proportional to an
in-phase portion of the alternating current applied; second
correlating means for correlating, for each frequency f.sub.AC of
the alternating current (AC) applied to the calibration impedance,
the Calibration Current Samples with the discrete values of an
ideal cosine waveform in order to obtain a value proportional to a
quadrature portion of the alternating current applied; third
correlating means for correlating, for each frequency f.sub.AC of
the alternating current (AC) applied to the calibration impedance,
the Calibration Voltage Samples with the discrete values of an
ideal sine waveform in order to obtain a value proportional to an
in-phase portion of the voltage measured; and fourth correlating
means for correlating, for each frequency f.sub.AC of the
alternating current (AC) applied to the calibration impedance, the
Calibration Voltage Samples with the discrete values of an ideal
cosine waveform in order to obtain a value proportional to a
quadrature portion of the voltage measured.
89. The apparatus of claim 88, further comprising: first
calculating means for determining an equivalent to a current
magnitude from the values proportional to the in-phase portion and
the quadrature portion of the current applied, second calculating
means for determining an equivalent to a voltage magnitude from the
values proportional to the in-phase portion and the quadrature
portion of the voltage measured; third calculating means for
determining a current phase from the values proportional to the
in-phase portion and the quadrature portion of the current applied;
fourth calculating means for determining a voltage phase of the
values proportional to the in-phase portion and the quadrature
portion of the voltage measured; fifth calculating means for
determining a system phase as the difference between the voltage
phase and the current phase; and sixth calculating means for
determining an equivalent for the magnitude of the calibration
impedance from the ratio of the equivalent for the voltage
magnitude and the equivalent of the current magnitude.
90. The apparatus of claim 89: wherein the plurality of means for
switching are further adapted for selectively coupling the signal
generating means and the first measuring means to the object, to
acquire Object Current Samples and Object Voltage Samples; and
wherein the apparatus further comprises: fifth correlating means
for correlating, for each frequency f.sub.AC of an alternating
current (AC) applied to the object, the Object Current Samples with
the discrete values of an ideal sine waveform in order to obtain a
value proportional to an in-phase portion of the alternating
current applied; sixth correlating means for correlating, for each
frequency f.sub.AC of the alternating current (AC) applied to the
object, the Object Current Samples with the discrete values of an
ideal cosine waveform in order to obtain a value proportional to a
quadrature portion of the alternating current applied; seventh
correlating means for correlating, for each frequency f.sub.AC of
the alternating current (AC) applied to the object, the Object
Voltage Samples with the discrete values of an ideal sine waveform
in order to obtain a value proportional to an in-phase portion of
the voltage measured; and eighth correlating means for correlating,
for each frequency f.sub.AC of the alternating current (AC) applied
to the object, the Object Voltage Samples with the discrete values
of an ideal cosine waveform in order to obtain a value proportional
to a quadrature portion of the voltage measured.
91. The apparatus of claim 90, further comprising: seventh
calculating means for determining an equivalent to a current
magnitude from values proportional to the in-phase portion and the
quadrature portion of the current applied; eighth calculating means
for determining an equivalent to a voltage magnitude from the
values proportional to the in-phase portion and the quadrature
portion of the voltage measured; ninth calculating means for
determining a current phase from the values proportional to the
in-phase portion and the quadrature portion of the current applied;
tenth calculating means for determining a voltage phase from the
values proportional to in-phase portion and quadrature portion of
the voltage measured; eleventh calculating means for determining an
object phase as the difference between the voltage phase and the
current phase, corrected for the system phase; twelfth calculating
means for determining an equivalent for a magnitude of the object
bioimpedance from the ratio of the equivalent for the voltage
magnitude and the equivalent of the current magnitude; thirteenth
calculating means for determining a magnitude of the object
bioimpedance from the ratio of the a priori known magnitude of the
calibration impedance and the equivalent for the calibration
impedance magnitude, multiplied by the equivalent for the object
bioimpedance magnitude; fourteenth calculating means for
determining an in-phase portion of the object bioimpedance; and
fifteenth calculating means for calculating a quadrature portion of
the object bioimpedance, from the magnitude and phase of the object
bioimpedance.
92. An apparatus for digital demodulation and further processing of
signals obtained to measure electrical bioimpedance in an object,
the apparatus comprising: signal generating means for generating an
excitation signal of known frequency content; a first pair of
electrodes for applying the excitation signal to the object; a
second pair of electrodes for sensing a response signal across the
object due to the application of the excitation signal; first
measuring means for acquiring, sampling and digitizing the response
signal to obtain a digitized response signal representing the
response signal with respect to frequency content, amplitude and
phase; second measuring means for digitizing a voltage that is
directly proportional to, and in phase with, the excitation signal;
memory means for temporarily storing the digitized response signal;
digital demodulation means for correlating, for each frequency
f.sub.AC of the excitation signal applied, digitized samples of the
digitized response signal with corresponding discrete values of a
sinusoidal reference signal to the excitation signal; processing
means for calculating for each frequency f.sub.AC of the excitation
signal applied, complex values for the bioimpedance Z(f.sub.AC)
from the output values of the digital demodulation means, and for
providing, over time, a set of digital bioimpedance waveforms
Z(f.sub.AC,t)); separating means adapted to separate changes in the
bioimpedance .DELTA.Z(f.sub.AC,t) from the waveforms;
differentiating means for obtaining a rate of change of the changes
in bioimpedance d(.DELTA.Z(f.sub.AC,t))/dt; and recording means for
recording a temporal course of the base bioimpedance and of the
changes in bioimpedance.
93. The apparatus of claim 92, further comprising: a calibration
impedance; a plurality of means for switching, for selectively
coupling the signal generating means and the first measuring means
to the calibration impedance, to acquire Calibration Current
Samples and Calibration Voltage Samples; wherein the signal
generating means generates an alternating current (AC); first
correlating means for correlating, for a frequency f.sub.AC of the
alternating current (AC) applied to the object, the Calibration
Current Samples with the Calibration Voltage Samples in order to
obtain a value proportional to an in-phase portion of the
calibration impedance; second correlating means for correlating,
for the frequency f.sub.AC of the alternating current (AC) applied
to the object, the Calibration Current Samples with the Calibration
Voltage Samples samples, which are shifted in time by -90 degrees,
in order to obtain a value proportional to a quadrature portion of
the calibration impedance; first calculating means for calculating
an equivalent to a magnitude of the calibration impedance from the
in-phase portion and quadrature portion; and second calculating
means for calculating a phase of the calibration impedance from the
in-phase portion and quadrature portion.
94. The apparatus of claim 93: wherein the plurality of means for
switching are further adapted for selectively coupling the signal
generating means and the first measuring means to the object, to
acquire Object Current Samples and Object Voltage Samples; and
wherein the apparatus further comprises: third correlating means
for correlating, for a frequency f.sub.AC of the alternating
current (AC) applied to the object, the Object Current Samples with
the Object Voltage Samples in order to obtain a value proportional
to an in-phase portion of the calibration impedance; fourth
correlating means for correlating, for the frequency f.sub.AC of
the alternating current (AC) applied to the object, the Object
Current Samples with the Object Voltage Samples samples, which are
shifted in time by -90 degrees, in order to obtain a value
proportional to a quadrature portion of the calibration impedance;
third calculating means for calculating an equivalent to a
magnitude of the object bioimpedance from the in-phase portion and
quadrature portion; fourth calculating means for calculating an
uncorrected phase of the object bioimpedance from the in-phase
portion and quadrature portion; fifth calculating means for
calculating a correct phase of the object bioimpedance from the
uncorrected object bioimpedance and from the phase of the
calibration impedance; and sixth calculating means for calculating
a magnitude of the object bioimpedance from the ratio of an a
priori known magnitude of the calibration impedance and the
determined equivalent for the calibration impedance magnitude,
multiplied by the determined equivalent for the object bioimpedance
magnitude.
95. The apparatus of claim 94, further comprising: seventh
calculating means for calculating an in-phase portion; and eighth
calculating means for calculating a quadrature portion from the
magnitude and phase of the object bioimpedance.
96. An apparatus for digital demodulation and further processing of
signals obtained to measure electrical bioimpedance in an object,
the apparatus comprising: signal generating means for generating an
excitation signal of known frequency content; a first pair of
electrodes for applying the excitation signal to the object; a
second pair of electrodes for sensing a response signal across the
object due to the application of the excitation signal; first
measuring means for acquiring, sampling and digitizing the response
signal to obtain a digitized response signal representing the
response signal with respect to frequency content, amplitude and
phase; memory means for temporarily storing the digitized response
signal; digital demodulation means for correlating, for each
frequency f.sub.AC of the excitation signal applied, digitized
samples of the digitized response signal with corresponding
discrete values of a sinusoidal reference signal to the excitation
signal; processing means for calculating for each frequency
f.sub.AC of the excitation signal applied, complex values for the
bioimpedance Z(f.sub.AC) from output values of the digital
demodulation means, and for providing, over time, a set of digital
bioimpedance waveforms Z(f.sub.AC,t)); first separating means
adapted to separate a base bioimpedance Z.sub.0(f.sub.AC) from the
waveforms; and recording means for recording a temporal course of
the base bioimpedance.
97. The apparatus of claim 96, further comprising: a calibration
impedance; a plurality of means for switching, for selectively
coupling the signal generating means and the first measuring means
to the calibration impedance, to acquire Calibration Voltage
Samples; wherein the signal generating means generates an
alternating current (AC); first correlating means for correlating,
for each frequency f.sub.AC of the alternating current (AC) applied
to the calibration impedance, the Calibration Voltage Samples with
the discrete values of an ideal sine waveform in order to obtain a
value proportional to an in-phase portion of the calibration
impedance; second correlating means for correlating, for each
frequency f.sub.AC of the alternating current (AC) applied to the
calibration impedance, the Calibration Voltage Samples with the
discrete values of an ideal cosine waveform in order to obtain a
value proportional to a quadrature portion of the calibration
impedance; first calculating means for determining an equivalent to
a magnitude of the calibration impedance; and second calculating
means for determining a phase of the calibration impedance.
98. The apparatus of claim 97: wherein the plurality of means for
switching are further adapted for selectively coupling the signal
generating means and the first measuring means to the object, to
acquire Object Voltage Samples; and wherein the apparatus further
comprises: third correlating means for correlating, for each
alternating current frequency f.sub.AC applied to the object, the
Object Voltage Samples with samples of an ideal sine waveform in
order to obtain a value proportional to an in-phase portion of the
object bioimpedance; fourth correlating means for correlating, for
each alternating current frequency f.sub.AC applied to the object,
the Object Voltage Samples with the samples of an ideal cosine
waveform in order to obtain a value proportional to a quadrature
portion of the object bioimpedance; third calculation means for
determining an equivalent to a magnitude of the object
bioimpedance; fourth calculation means for determining the
uncorrected phase of the object bioimpedance; fifth calculating
means for determining a correct phase of the object bioimpedance by
subtracting the phase of the calibration impedance from the phase
of the uncorrected object bioimpedance; sixth calculating means for
determining a magnitude of the object bioimpedance from the ratio
of an a priori known magnitude of the calibration impedance and the
determined equivalent for the calibration impedance magnitude,
multiplied by the determined equivalent for the object bioimpedance
magnitude; seventh calculating means for determining an in-phase
portion of the object bioimpedance; and eighth calculating means
for determining a quadrature portion of the object bioimpedance
from the magnitude and phase of the object bioimpedance.
99. The apparatus of claim 98, further comprising: fitting means
for fitting samples of the digitized current signals of the
calibration impedance towards discrete values of an ideal
sinusoidal waveform, and for providing, over time, Calibration
Current Samples; fitting means for fitting samples of the digitized
voltage signals of the calibration impedance towards values of an
ideal sinusoidal waveform, and for providing, over time, the
Calibration Voltage Samples.
100. The apparatus of claim 96, further comprising second measuring
means for digitizing a voltage that is directly proportional to,
and in phase with, the excitation signal.
101. The apparatus of claim 96, further comprising: a plurality of
means for switching, for selectively coupling the signal generating
means and the first measuring means to the object, to acquire
Object Current Samples and Object Voltage Samples; and wherein the
signal generating means generates an alternating current (AC); and
further comprising: first correlating means for correlating, for
each frequency f.sub.AC of the alternating current (AC) applied to
the object, the Object Current Samples with discrete values of an
ideal sine waveform in order to obtain a value proportional to an
in-phase portion of the alternating current; second correlating
means for correlating, for each frequency f.sub.AC of the
alternating current (AC) applied to the object, the Object Current
Samples with discrete values of an ideal cosine waveform in order
to obtain a value proportional to a quadrature portion of the
alternating current; third correlating means for correlating, for
each frequency f.sub.AC of the alternating current (AC) applied to
the object, the Object Voltage Samples with discrete values of
another ideal sine waveform to obtain a value proportional to an
in-phase portion of the voltage; fourth correlating means for
correlating, for each frequency f.sub.AC of the alternating current
(AC) applied to the object, the Object Voltage Samples with
discrete values of another ideal cosine waveform in order to obtain
a value proportional to a quadrature portion of the voltage; first
calculation means for determining a current magnitude from the
values proportional to in-phase portion and quadrature portion of
the current applied; second calculation means for determining a
voltage magnitude from the values proportional to the in-phase
portion and the quadrature portion of the voltage measured; third
calculation means for determining a current phase of the values
proportional to the in-phase portion and the quadrature portion of
the current applied; fourth calculation means for determining a
voltage phase from the values proportional to the in-phase portion
and the quadrature portion of the voltage measured; fifth
calculation means for determining a phase of the object
bioimpedance as the difference between the voltage phase and the
current phase; sixth calculation means for determining a magnitude
of the object bioimpedance from the ratio of voltage magnitude and
current magnitude; seventh calculating means for determining an
in-phase portion of the object bioimpedance; and eighth calculating
means for determining a quadrature portion of the object
bioimpedance from the magnitude and phase of the object
bioimpedance.
102. The apparatus of claim 96, further comprising: a plurality of
means for switching, for selectively coupling the signal generating
means and the first measuring means to the object, to acquire
Object Current Samples and Object Voltage Samples; and wherein the
signal generating means generates an alternating current (AC); and
further comprising: first correlating means for correlating the
Object Current Samples with the Object Voltage Samples in order to
obtain a value proportional to an in-phase portion of the object
bioimpedance; second correlating means for correlating the Object
Current Samples with the Object Voltage Samples, which are shifted
in time by -90 degrees, in order to obtain a value proportional to
a quadrature portion of the object bioimpedance; first calculating
means for determining a magnitude of the object bioimpedance; and
second calculating means for determining a phase of the object
bioimpedance, both from the in-phase portion and quadrature portion
of the object bioimpedance.
103. The apparatus of claim 96, further comprising: a plurality of
means for switching, for selectively coupling the signal generating
means and the first measuring means to the object, to acquire
Object Voltage Samples; and wherein the signal generating means
generates an alternating current (AC) having constant magnitude;
and further comprising: sampling means for providing, for each
frequency f.sub.AC of the alternating current (AC) applied,
discrete values of an ideal sine waveform which represent the
current in magnitude and phase, wherein the discrete values of the
ideal sine waveform are called Reference Current Samples (REF);
first correlating means for correlating, for each frequency
f.sub.AC of the alternating current (AC) applied, the Object
Voltage Samples with the Reference Current Samples (REF), to obtain
a value proportional to an in-phase portion of the object
bioimpedance; second correlating means for correlating, for each
frequency f.sub.AC of the alternating current (AC) applied, the
Object Voltage Samples with the Reference Current Samples (REF), to
obtain a value proportional to a quadrature portion of the object
bioimpedance; first calculating means for determining a magnitude
of the object bioimpedance; and second calculating means for
determining a phase of the object bioimpedance both from the
in-phase portion and quadrature portion of the object
bioimpedance.
104. The apparatus of claim 103, further comprising: fitting means
for fitting the samples of the digitized voltage signals of the
object towards values of an ideal sinusoidal waveform, and for
providing, over time, the Object Voltage Samples.
105. An apparatus for digital demodulation and further processing
of signals obtained to measure electrical bioadmittance in an
object, the apparatus comprising: signal generating means for
generating an excitation signal of known frequency content; a first
pair of electrodes for applying the excitation signal to the
object; a second pair of electrodes for sensing a response signal
across the object due to application of the excitation signal;
first measuring means for acquiring, sampling and digitizing the
response signal to obtain a digitized response signal representing
the response signal with respect to frequency content, amplitude
and phase; memory means for temporarily storing the digitized
response signal; digital demodulation means for correlating, for
each frequency f.sub.AC of the excitation signal applied, digitized
samples of the digitized response signal with corresponding
discrete values of a sinusoidal reference signal to the excitation
signal; processing means for calculating for each frequency
f.sub.AC of the excitation signal applied, complex values for the
bioadmittance Y(f.sub.AC) from output values of the digital
demodulation means, and for providing, over time, a set of digital
bioadmittance waveforms Y(f.sub.AC,t); means for, for each
frequency f.sub.AC of the excitation signal applied, separating the
base bioadmittance Y.sub.0(f.sub.AC) from the bioadmittance
waveform Y(f.sub.AC,t); means for, for each frequency f.sub.AC of
the excitation signal applied, separating the changes of
bioadmittance .DELTA.Y(f.sub.AC,t) from the bioadmittance waveform
Y(f.sub.AC,t); and recording means for recording a temporal course
of the base bioadmittance Y.sub.0(f.sub.AC).
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] Priority for this patent application is being claimed to
European Patent Application No. EP05017871, filed Aug. 17, 2005,
titled "Method and apparatus for digital demodulation and further
processing of signals obtained in the measurement of electrical
bioimpedance or bioadmittance in an object", which is incorporated
herein by this reference.
BACKGROUND
[0002] 1. Technical Field
[0003] This invention is related to the field of digital
demodulation and further processing of signals obtained from the
measurement of electrical bioimpedance or bioadmittance in a
biological object, for instance an animal or a human due to cardiac
and/or respiratory activity, for instance in cardiometry, in
particular to the monitoring through measurement of the change in
thoracic electrical bioimpedance (TEB) or bioadmittance, and
pertains to the processing of the excitation, response and/or
reference signals obtained through sensing and measuring
excitation, response and/or reference signals, e.g., but not
limited to, a voltage resulting from an alternating current (AC)
application.
[0004] 2. Description of Related Art
BACKGROUND OF THE INVENTION
[0005] Noninvasive hemodynamic monitoring utilizes the measurement
of thoracic electrical bioimpedance (TEB) for the determination of
stroke volume, cardiac output and other cardiopulmonary parameters
in humans or animals.
[0006] For this purpose, an alternating current of known (e.g.,
constant amplitude) is applied to current electrodes, e.g. surface
electrodes, located e.g. at one or both sides of the neck and the
lower thorax, approximately at the level of the xiphoid process
(Sramek B.: U.S. Pat. No. 4,450,527; Osypka M. J., Bernstein D. P.:
Electrophysiological Principles and Theory of Stroke Volume
Determination by Thoracic Electrical Bioimpedance. MCN Clinical
Issues 1999; 10, 3: 385-399). The resulting voltage is sensed
through sensing electrodes, e.g. other surface spot electrodes, and
measured.
[0007] Alternatively, current electrodes (surface electrodes) are
applied to the forehead, or substituted by band electrodes around
the circumferences of the neck and the lower thorax (Kubicek W. G.:
U.S. Pat. No. 3,340,867) or by current electrodes located on an
esophageal catheter (Sramek B.: U.S. Pat. No. 4,836,214) or an
implantable pacemaker or defibrillator lead, with the latter
applications focusing on the heart rather than the overall
thorax.
[0008] The voltage resulting from constant alternating current
application, which is proportional to the thoracic electrical
impedance, is modulated onto an alternating voltage signal of the
frequency of the alternating current (AC) applied.
[0009] A common approach in the acquisition of the thoracic
impedance applies an active high-pass or band-pass filter to the
voltage signal obtained from the sensing electrodes. Each
individual signal is further demodulated through a diode rectifier
circuit, and fed into the input of a differential amplifier. The
differential signal which is proportional to Z(t) in the event of a
constant alternating current (AC) application, is then separated,
in the analog domain, into a DC voltage proportional to the base
impedance Z.sub.0 and the change in impedance .DELTA.Z(t). Both
analog signals Z.sub.0 and .DELTA.Z(t) are then digitized for
further processing.
[0010] However, conventional demodulation by diodes and subsequent
low-pass filtering exhibit certain drawbacks. Diode characteristics
change with temperature. Moreover, even a full-wave rectified
sinusoidal signal is difficult to smooth; the time constant of the
smoothing low-pass filter cannot be chosen appropriately high
because the bandwidth of the desired demodulated signal will be
limited, and critical waveform detail can be lost. Digitization of
a signal with ripple can produce unstable data, due to the fact
that the values of the samples of the demodulated signal depend on
the position of the sampling within the period of the carrier
signal.
[0011] In order to improve accuracy and stability and to ease
adaptation to changing conditions, Osypka et al. developed a
technique using a phase sensitive detector, a subsequent integrator
and a high resolution analog-to-digital converter (ADC), followed
by digital signal processing (DSP) (Osypka M. J. and Schafer E. E.,
Impedance Cardiography: Advancements in System Design. In: Riu P J,
Rosell J, Bragos R, Casas O (eds.): Proceedings of the X.
International Conference on Electrical Bio-Impedance (ICEBI),
Barcelona, Spain, Apr. 5-9, 1998). The demodulation is, however,
achieved with analog circuitry. The voltage obtained from the
thorax is fed into a phase-sensitive detector (PSD) circuit. The
reference trigger signal for the PSD is derived from the
alternating sinusoidal current generator, followed by a phase
shifter to adjust the reference phase for detecting the real (or
imaginary) part of impedance, and a comparator switching at the
zero-crossings of the applied sinusoidal current. The output of the
PSD consists of the full wave rectified carrier signal containing
the information on the real or imaginary part of the thoracic
impedance, depending on the reference phase. The following stage
provides the smoothing of the demodulated signal by integration
over an integer number of cycles of the carrier frequency, which
corresponds, for instance, to an integration time of 1 millisecond.
Integration begins after the integration capacitor has been
discharged by a reset signal, and ends prior to the start signal
for the high-resolution ADC. The timing control is initialized by
the reference trigger signal for the PSD, ensuring that the
integration is performed over a number of complete periods of the
carrier signal, and provides the appropriate start pulses for the
ADC. By this process, the demodulated impedance signal Z(t) is
"updated" every millisecond. A high resolution (.gtoreq.20 bit) ADC
measures the charge accumulated during the integration, which is
proportional to the thoracic impedance Z(t). The integration
period, i.e. the time constant of averaging, can be easily
changed.
[0012] For the purpose of determining the change in thoracic
electrical bioimpedance, the theoretical sound approach has
significant practical limitations. First, circuit design will
compromise on the theoretically available resolution of 20 bits.
More realistic, a resolution of 14-16 bits is achievable. The
second limitation is due to the thoracic bioimpedance signal
itself: Z(t) consists of a portion, which is quasi-constant over
time and further referred to as Z.sub.0, and another portion
.DELTA.Z(t), which changes during cardiac and respiration cycles:
Z(t)=Z.sub.0+.DELTA.Z(t).
[0013] In particular the amplitudes of changes in bioimpedance due
to the pump function of the heart are very small compared to
Z.sub.0. With a large offset (Z.sub.0) taking up approximately 8
bits of resolution, the remaining 6-8 bits are available for
quantization of the dynamic portion of Z(t), namely
.DELTA.Z(t).
PROBLEM UNDERLYING THE INVENTION
[0014] It is an object of the present invention to propose a method
and an apparatus for digital demodulation and further processing of
signals obtained in the single and multi-frequency (f.sub.AC)
measurement of electrical bioimpedance
Z(f.sub.AC,t)=Z.sub.0(f.sub.AC)+.DELTA.Z(f.sub.AC,t)
[0015] or electrical bioadmittance
Y(f.sub.AC,t)=Y.sub.0(f.sub.AC)+.DELTA.Y(f.sub.AC,t)
[0016] in a biological object in which the amplitude of changes or
rate of changes in electrical bioimpedance, .DELTA.Z(f.sub.AC,t),
or bioadmittance, .DELTA.Y(f.sub.AC,t), due to biological functions
of the biological object, such as plants or as animals or humans,
with the latter for instance due to respiratory or cardiac
functions including the pump function of the heart, can be
determined with a higher amplitude resolution than before.
SUMMARY
[0017] The method and apparatus according to the present invention
as defined in the appended claims employs digital demodulation by
means of correlation, also called correlation or matched filter
technique, and further digital signal processing followed by a
calculation of a complex value for the bioimpedance or bio
admittance, respectively, for each frequency f.sub.AC of an
excitation signal of known frequency content, preferably an
alternating current (AC), applied, providing, over time, a set
(spectrum) of bioimpedance waveforms, Z(f.sub.AC,t), or
bioadmittance waveforms, Y(f.sub.AC,t), with
Z(f.sub.AC,t)=Z.sub.0(f.sub.AC)+.DELTA.Z(f.sub.AC,t)
Y(f.sub.AC,t)=Y.sub.0(f.sub.AC)+.DELTA.Y(f.sub.AC,t)
[0018] to which a first filter, preferably a low pass filter, is
applied to separate the base impedance, Z.sub.0(f.sub.AC), or the
base admittance, Y.sub.0(f.sub.AC), therefrom, and a second filter,
preferably a high pass filter, is applied to separate the change in
the electrical bioimpedance overtime, .DELTA.Z(f.sub.AC,t), or the
change in the electrical bioadmittance over time,
.DELTA.Y(f.sub.AC,t).
[0019] .DELTA.Z(f.sub.AC,t) or directly Z(f.sub.AC,t), or
.DELTA.Y(f.sub.AC,t) or directly Y(f.sub.AC,t), can be input to a
differentiator in order to obtain the rate of change of the changes
in bioimpedance, d(.DELTA.Z(f.sub.AC,t))/dt, or the rate of change
of the bioimpedance (waveforms), dZ(f.sub.AC,t)/dt, respectively,
or the rate of change of the changes in bioadmittance
d(.DELTA.Y(f.sub.AC,t))/dt or the rate of change of the
bioadmittance (waveforms), dY(f.sub.AC,t)/dt, respectively.
[0020] In the claims the term excitation signal as used is intended
to encompass a voltage signal, a current signal and an
electro-magnetic field signal for application to the object.
[0021] The term signal of known frequency content means that the
signal is defined as regards to a single frequency or a composite
frequency composed of a number of superimposed frequencies. In the
embodiment in which the excitation signal is measured the amplitude
and phase of the excitation signal must not be known a priori.
[0022] The term correlating includes several meanings: a)
correlation of said digitized excitation signal with said digitized
response signal; b) correlation of said digitized excitation signal
delayed by 90.degree. with said digitized response signal; c)
correlation of said digitized excitation signal with the digital
values of an ideal sinusoidal signal (sin, cos) (reference signal
to the excitation signal), and d) correlation of said digitized
response signal with the digital values of an ideal sinusoidal
signal (sin, cos) (reference signal to the excitation signal).
[0023] Like conventional approaches for bioimpedance or
bioadmittance measurements in cardiometry, the excitation signal is
preferably an alternating current of known frequency or frequencies
f.sub.AC with related amplitude(s) and phase(s), preferably of
constant magnitude, and is applied to the object, e.g. a human
thorax, or a portion of it, or arm, or limb, or heart, or trachea,
or the esophagus via electrodes located on the skin surface, or
tracheal or esophageal catheters or probes, or implantable
pacemaker or defibrillator leads,
[0024] Unlike conventional approaches, the response signal, i.e.,
the voltage resulting from the current application, which, in the
event of an AC application with constant magnitude, is proportional
to the bioimpedance or reciprocal to the bioadmittance, is sampled
and digitized as early as possible, prior to any demodulation. The
demodulation is accomplished by digital signal processing (DSP)
directly or indirectly correlating the measured with digitized
signal waveforms representing the response signal, particularly the
voltage signal measured across the object, and the excitation
signal, particularly the alternating current (AC) applied across
the object, for example, the human or animal thorax, or a portion
of it (direct correlation) or a reference signal to the excitation
signal (indirect correlation).
[0025] With the sampling rate being significantly higher than the
highest frequency component of the excitation signal, particularly
an alternating current (AC), applied, the method and apparatus
according to the invention provides measurement results at not only
a sufficient resolution but a very high amplitude resolution.
Unlike common approaches proposed for the display of
multi-frequency bioimpedance (Withers P.O.: U.S. Pat. No.
5,280,429) and in a real-time electrical impedance tomography
system (Brown B. H. and Barber D. C.: U.S. Pat. No. 5,311,878),
bioimpedance or bioadmittance cardiometry requires a high
resolution and accuracy of correlation results because the changes
related to the cardiac cycle are significantly smaller in amplitude
than the quasi-constant portion. If the influence of respiration or
ventilation is suppressed, or the corresponding effect on the
impedance respectively admittance signal is filtered out, only the
cardiac-induced pulsatile impedance or admittance component
remains. By magnitude, .DELTA.Z(f.sub.AC,t) for instance is
approximately 0.3% to 0.5% of Z.sub.0 (Osypka M. J. and Bernstein
D. P.: Electrophysiologic Principles and Theory of Stroke Volume
Determination by Thoracic Electrical Bioimpedance; AACN Clinical
Issues 1999: 10, 3: 385-399).
[0026] Furthermore, the method and apparatus according to the
invention separate for one or more frequencies f.sub.AC of the
excitation signal, e.g. the alternating current (AC) applied, the
change in electrical bioimpedance, .DELTA.Z(f.sub.AC,t), from the
offset, or base impedance Z.sub.0, or the change in electrical
bioadmittance, .DELTA.Y(f.sub.AC,t), from the offset, or base
admittance Y.sub.0, and determine .DELTA.Z(f.sub.AC,t), or
.DELTA.Y(f.sub.AC,t), respectively, .DELTA.Z(f.sub.AC,t) or
directly Z(f.sub.AC,t), or .DELTA.Y(f.sub.AC,t) or directly
Y(f.sub.AC,t), as they are differentiated, e.g. by inputting to a
differentiator in order to obtain the rate of change in
bioimpedance, dZ(f.sub.AC)/dt, or the rate of change in
bioadmittance, dY(f.sub.AC)/dt, respectively, with high
resolution.
[0027] Unlike common approaches proposed for the display of
multi-frequency bioimpedance (Withers P.O.: U.S. Pat. No.
5,280,429), where the complex Fourier transform of the complex
cross-correlation signal (as a function of time delay between the
excitation and response signals) is determined, the method and
apparatus according to one embodiment of the invention perform the
correlation separately for each frequency f.sub.AC, of the
excitation signal applied (in the description further referred to
as indirect correlation), correlating digitized samples of the
measured, sampled and digitized response signals with digital
samples of ideal sinusoids being reference signals to the
excitation signal.
[0028] The method and apparatus according to the invention propose
embodiments with and without incorporation of a calibration
impedance in connection with the suppression of the influence of
electrical circuit properties and its influence on the
measurement.
[0029] Furthermore, signal curve fitting is envisaged as an option
for signals in noisy environments.
[0030] The method and apparatus determine, for one frequency, or
several frequency components, of the excitation signal,
particularly the alternating current applied, the complex
bioimpedance, or complex bioadmittance, i.e. the real part
(in-phase portion) and the imaginary part (quadrature portion) of
the impedance, or admittance, at a high amplitude resolution, which
is required to separate the change in electrical bioimpedance,
.DELTA.Z(f.sub.AC,t), from the offset, or base impedance,
Z.sub.0(f.sub.AC), or the change in electrical bioadmittance,
.DELTA.Y(f.sub.AC,t), from the offset, or base admittance,
Y.sub.0(f.sub.AC), and determine .DELTA.Z(f.sub.AC,t), or
.DELTA.Y(f.sub.AC,t), respectively, and dZ(f.sub.AC,t), or
dY(f.sub.AC,t) respectively, with high resolution.
[0031] Theoretically the correlation process is derived from the
Fourier transform for periodical signals s(t): s .function. ( t ) =
a 0 2 + v = 1 V .times. .times. ( a v .times. cos .function. ( v
.times. .times. .omega. 0 .times. t ) + b v .times. sin .function.
( v .times. .times. .omega. 0 .times. t ) ) ##EQU1## where a 0 2
##EQU2## [0032] designates the offset, [0033] .omega..sub.0 is the
basic frequency, [0034] .nu. is the number of the harmonic of the
base frequency .omega..sub.0, and a v = 2 N .times. n = 1 N .times.
.times. s n .times. cos .function. ( v .times. .times. .omega. 0
.times. t n ) , .times. v = 0 , 1 , .times. , V ##EQU3## represents
the quadrature portion, or imaginary part, of each frequency
component of s(t), and b v = 2 N .times. n = 1 N .times. .times. s
n .times. sin .function. ( v .times. .times. .omega. 0 .times. t n
) , .times. v = 0 , 1 , .times. , V ##EQU4##
[0035] represents the in-phase portion, or real part, of each
frequency component of s(t).
[0036] N represents the number of samples obtained at equidistant
time intervals at points in time t.sub.n(N>2V+1). The samples of
s(t) are referred to as s.sub.n.
[0037] Because the frequency of the alternating current (AC)
applied is known a priori (.nu.=1), the Fourier transform is
reduced to s(t)=a.sub.1 cos(.omega..sub.0t)+b.sub.1
sin(.omega..sup.0t).
[0038] A band pass filter is applied to s(t) for suppression of
noise.
[0039] The application of digital demodulation has already been
proposed for the multi-frequency measurement of impedances (Osypka
M., Schmerbeck A., Gersing E., and Meyer-Waarden K.: Determination
of electric impedances of tissue at a frequency range of 5 Hz to 20
kHz by digital correlation technique. In: Nikiforidis G.,
Pallikaridis N., Proimos B. (eds.): Proceedings V. Mediterranean
Conference on Medical and Biological Engineering (MEDICON 89), Aug.
29-Sep. 1, 1989, University of Patras, Greece; Withers P.O.: U.S.
Pat. No. 5,280,429) and for an electrical impedance tomography
system (Osypka M., Gersing E., and Meyer-Waarden K.: Komplexe
elektrische Impedanztomografie im Frequenzbereich von 10 Hz bis 50
kHz. Z. Med. Phys. 3 (1993), 124-132; Osypka M. and Gersing E.:
Parallel signal processing and multi-electrode current feeding in a
multi-frequency EIT system. Innovation et Technologie en Biologie
et Medicine, Vol. 15, Special Issue 1, 1994, pp. 56-61; Brown B. H.
and Barber D. C.: U.S. Pat. No. 5,311,878). Application for
determination of the base impedance, Z.sub.0(f.sub.AC), the change
in electrical bioimpedance .DELTA.Z(f.sub.AC,t), and the rate of
change in electrical bioimpedance, dZ(f.sub.AC)/dt or the base
admittance, Y.sub.0(f.sub.AC), the change in electrical
bioadmittance .DELTA.Y(f.sub.AC,t), or the rate of change in
bioelectrical admittance dY(f.sub.AC)/dt, with high resolution was
not considered.
BRIEF DESCRIPTION OF THE DRAWINGS
[0040] Preferred embodiments of the apparatus of the invention are
described in the following with respect to the drawings. However,
this description is of exemplary nature and does not limit the
spirit and scope of the invention as defined in the claims and
equivalents thereof.
[0041] FIG. 1a is a basic diagram of an embodiment of the apparatus
for application of a single or multi-frequency alternating current
(AC) as the excitation signal, employing a single
multiplier/accumulator (MACC) for digital demodulation of the
resulting response signal, a voltage signal;
[0042] FIG. 1b is a basic diagram of an embodiment of the apparatus
for application of an alternating current containing 3 frequency
components as the excitation signal, employing a
multiplier/accumulator (MACC) for parallel digital demodulation of
each frequency component of the resulting response signal, a
voltage signal;
[0043] FIG. 1c-1g illustrate the details of embodiments employing a
multi-frequency alternating current (AC) application;
[0044] FIG. 2a is a flowchart of one embodiment of a correlation
method;
[0045] FIG. 2b provides charts for a graphic description of the
correlation method of FIG. 2a;
[0046] FIG. 3 is a systematic overview of the modes of measurement
of the embodiments of FIG. 4-9;
[0047] FIGS. 4a and 4b form jointly a flowchart of a first
embodiment of the invention for the determination of the complex
object impedance (Z.sub.OBJ), i.e., the impedance of interest, by
measurements of the alternating current (AC) as an excitation
signal and the alternating voltage as the response signal, and the
use of a calibration impedance, and by application of indirect
correlation;
[0048] FIG. 5 is a flowchart of a second embodiment for the
determination of the complex object impedance (Z.sub.OBJ), i.e.,
the impedance of interest, by measurements of the alternating
current (AC) as the excitation signal the alternating voltage as
the response signal, and the use of a calibration impedance, and by
application of direct correlation;
[0049] FIG. 6 is a flowchart of a third embodiment for the
determination of the complex object impedance (Z.sub.OBJ), i.e.,
the impedance of interest, with the measurement of the alternating
voltage but without the measurement of the alternating current (AC)
applied and with the use of a calibration impedance;
[0050] FIG. 7 is a flowchart of a fourth embodiment for the
determination of the complex object impedance (Z), i.e., the
impedance of interest, by measurements of the alternating current
(AC) and the alternating voltage, without the use of any
calibration impedance, and by application of indirect
correlation;
[0051] FIG. 8 is a flowchart of a fifth embodiment for the
determination of the complex object impedance (Z), i.e., the
impedance of interest, by measurements of the alternating current
(AC) and the alternating voltage, without the use of any
calibration impedance, and by application of direct
correlation;
[0052] FIG. 9 is a flowchart of a sixth embodiment for the
determination of the complex object impedance (Z), i.e., the
impedance of interest, with the measurement of the alternating
voltage but without the measurement of the alternating current (AC)
applied and without the use of any calibration impedance.
DETAILED DESCRIPTION
[0053] FIG. 1a illustrates a preferred embodiment of the apparatus
according to the invention.
[0054] The embodiment is suitable for determining with high
resolution the thoracic electrical impedance
Z(f.sub.AC,t)=Z.sub.0(f.sub.AC)+.DELTA.Z(f.sub.AC,t)
[0055] where:
[0056] Z(f.sub.AC,t) is the thoracic electrical impedance (TEB),
over time, for a particular frequency f.sub.AC of the alternating
current (AC) applied,
[0057] Z.sub.0(f.sub.AC) is the base impedance, i.e., the
quasi-constant portion, or offset, of Z(f.sub.AC,t), and
[0058] .DELTA.Z(f.sub.AC,t) is the change of thoracic electrical
impedance, i.e., the portion of Z(f.sub.AC,t) which is related to
impedance changes during the cardiac cycle and respiration or
ventilation cycle,
[0059] A single or multi-frequency alternating current (AC) of
known frequency and phase and having as an excitation source a
constant amplitude is applied to an object 10 via a first pair of
electrodes comprising a current electrode 12 located at the
object's left side of the neck and a current electrode 14 located
at the object's left side of the thorax, approximately at the level
of the xiphoid process. Furthermore, a second pair of electrodes
comprises a voltage sensing electrode 16, which is located below
current electrode 12, and a voltage sensing electrode 18, which is
located above current electrode 14. The second pair of electrodes
16 and 18 serves measuring the response signal across the object
due to excitation with the alternating current (excitation signal)
by means of the first pair of electrodes 12 and 14.
[0060] The alternating current (AC) can be switched to the object
10 or to a calibration impedance 20 via an electronic switch 30 and
an electronic switch 32. Accordingly, a differential amplifier (A)
50 can be switched to the object 10 or the calibration impedance 20
via an electronic switch 34 and an electronic switch 36.
[0061] In the preferred embodiment, the alternating current (AC) is
generated by the use of discrete samples of full sinusoidal
waveforms or of portions thereof, the reference signal to the
excitation signal, stored in an addressable sine table 70, which is
connected to a digital-to-analog converter (DAC) 40. The samples in
the sine table 70 are addressed in such a way that the
digital-to-analog converter (DAC) 40 outputs a voltage signal of a
desired frequency content and a desired voltage amplitude. The
application of appropriate low pass filtering following the
digital-to-analog converter (DAC) 40 in order to smooth possible
ripples at its output is known to the art and not further
described. A timing control 62 provides the addresses and clock
signals required therefore. A processing unit 60 initializes the
timing control 62 and the sine table 70.
[0062] The output of the digital-to-analog converter (DAC) 40
drives an excitation means embodied by a voltage-controlled current
source (VCCS) 42, which generates an alternating current (AC) of
the desired frequency content and of a constant AC amplitude.
Typical, but not limited to, are AC frequencies in the range of 10
kHz to 200 kHz, and AC amplitudes in the range of 0.01 mA,
preferably 2 mA, to 5 mA, the AC amplitudes of which are limited
depending on the frequency of the alternating current (AC) applied.
A current monitor (CM) 44 monitors the alternating current (AC)
signal for the purpose of detection of saturation of the
(excitation) current source due to overload or open circuitry and
provides an analog signal reflecting the alternating current (AC)
which is connected to a second fast analog-to-digital converter
(ADC 2) 46. The alternating current (AC) applied from the
excitation source is sampled and digitized by the second
analog-to-digital converter (ADC 2) 46 at a sampling rate which is
controlled by the timing control 62. The samples, further referred
to as the Current Samples, are stored into a second buffer (Buffer
2) 48. The current monitor (CM) 44 and the second analog-to digital
converter (ADC 2) 46 together form a 2.sup.nd measuring means, i.e.
the measuring means for the excitation signal from the excitation
source.
[0063] The application of the alternating current (AC) to the
thorax causes a voltage between the response signal (voltage)
sensing second pair of electrodes 16, 18. The differential
amplifier 50 senses this voltage superimposed by the
electrocardiogram (ECG), and amplifies it.
[0064] The current electrode 12 and the voltage sensing electrode
16, and the current electrode 14 and the voltage sensing electrode
18 may be each combined in a single double purpose electrode which
serves to feed a current (excitation) signal and to retrieve a
response signal.
[0065] The differential amplifier 50 is connected to a first fast
analog-to-digital converter (ADC 1) 52, which digitizes the output
of the differential amplifier 50 at a sampling rate preferably
equal to the sampling rate of the second analog-to-digital
converter (ADC 2) 46, both being controlled by the timing control
62. The digital samples obtained by the first analog-to-digital
converter 52, further referred to as the Voltage Samples, are
stored into a first buffer (Buffer 1) 58. Correlation, i.e., the
process of multiplication and accumulation, is performed by a
multiplier/accumulator (MACC) 80. The differential amplifier 50 and
the first analog-to-digital converter (ADC1) 52 form a 1.sup.st
measuring means.
[0066] For obtaining a value proportional to the in-phase portion
of a frequency f.sub.AC of the alternating current (AC) applied,
the multiplier/accumulator (MACC) 80 correlates the Current Samples
with samples of an ideal sinusoid of the frequency f.sub.AC, which
is obtained from the sine table 70 and represents the corresponding
component of the alternating current (AC) applied. This process,
also referred to as indirect correlation, is reiterated for each
frequency f.sub.AC of the alternating current applied.
[0067] For obtaining a value proportional to the quadrature portion
of a frequency f.sub.AC of the alternating current (AC) applied,
the multiplier/accumulator (MACC) 80 correlates the Current Samples
with samples of an ideal sinusoid of the frequency f.sub.AC shifted
by -90 degrees in phase, which is obtained from the sine table 70.
This process is reiterated for each frequency f.sub.AC of the
alternating current applied.
[0068] For obtaining a value proportional to the in-phase portion
of the sensed voltage at a frequency f.sub.AC of the alternating
current (AC) applied, the multiplier/accumulator (MACC) 80
correlates the Voltage Samples with samples of an ideal sinusoid of
the frequency f.sub.AC, which is obtained from the sine table 70.
This process is reiterated for each frequency f.sub.AC of the
alternating current applied.
[0069] For obtaining a value proportional to the quadrature portion
of the sensed voltage at a frequency f.sub.AC of the alternating
current (AC) applied, the multiplier/accumulator (MACC) 80
correlates the Voltage Samples with samples of an ideal sinusoid of
the frequency f.sub.AC shifted by -90 degrees in phase, which is
obtained from the sine table 70. This process is reiterated for
each frequency f.sub.AC of the alternating current applied.
[0070] Alternatively, in a single-frequency alternating current
(AC) application, the multiplier/accumulator 80 correlates Current
Samples with Voltage Samples directly, a process further referred
to as direct correlation.
[0071] Alternatively, if the alternating current is kept at known
constant amplitude, the measurement of the alternating current and
the second analog-to-digital-converter (ADC 2) 46 and the second
buffer (Buffer 2) 48 can be avoided. Then the samples of an ideal
sinusoid obtained for each frequency from the sine table 70 (as the
reference signal to the excitation signal) and used for correlation
represent, for each frequency, the alternating current applied.
[0072] The differential amplifier 50 is connected to a filter 54
with band-pass characteristics and its output to a third
analog-to-digital converter (ADC 3) 56, which samples the
electrocardiogram (ECG). This separate ECG channel is advantageous
not only for the detection of the intrinsic QRS complexes but also
for the detection of cardiac pacemaker pulses if desired. The
samples are acquired by the processing unit 60. The processing unit
60 applies one or more digital filters to the digitized
electrocardiogram and provides this signal to a R-Wave detector 68,
whose output is received by the processing unit 60.
[0073] In the preferred embodiment, the addresses of the sine table
70, the digital-to-analog converter (DAC) 40, the analog-to-digital
converters (ADC) 46, 52, 56, and the buffers are synchronized with
clock signals provided by the timing control 62. Because the
current applied and the voltage measured are known exactly in
frequency, amplitude and phase, errors due to system properties,
such as propagations delays or phase shifts, can be effectively
eliminated. Utilization of a calibration impedance, preferably a
precision ohmic resistor, to which the system periodically
switches, allows calibration before and in between
measurements.
[0074] For each frequency f.sub.AC of the alternating current (AC)
applied, the output of the correlation process is the digital
demodulated waveform of the in-phase or quadrature portion of the
thoracic electrical bioimpedance Z(f.sub.AC,t)
[0075] In the preferred embodiment, the second analog-to-digital
converter (ADC 1) 46 and the first analog-to-digital converter (ADC
2) 52 are clocked at a rate significantly higher than the highest
frequency of the alternating current (AC). As an example for a
single-frequency (SF) alternating current (AC) application, but not
limited to, the AC frequency is set to 50 kHz and the ADC sampling
rates to 500 kHz. The correlation vector contains 2500 pairs. Thus,
a correlation result is obtained every 5 ms, or 200 results per
second. Doubling the ADC sampling rate would result in 400
correlation results per second.
[0076] The application of appropriate anti-aliasing filtering is
known to the art and not further described.
[0077] The processing unit 60 initializes the timing control 62,
the sine table 70 and the multiplier/accumulator (MACC) 80. It
receives the output of the multiplier/accumulator (MACC) 80, and
calculates the samples for thoracic electrical bioimpedance. The
periodically occurring results of the correlation process form, for
each frequency f.sub.AC of the alternating current (AC) applied, a
digital waveform, which mirrors, within the scope of the embodiment
according to FIG. 1a, the thoracic electrical bioimpedance
Z(f.sub.AC,t). The processing unit 60 aligns the digital waveforms
of thoracic electrical bioimpedance and electrocardiogram (ECG) in
time.
[0078] The application of an appropriate filter (LPF) 64 with low
pass characteristics to Z(f.sub.AC,t) produces a quasi-constant
value, of which the amplitude is known as the base impedance, or
Z.sub.0.
[0079] The further application of an appropriate filter 66 with
high pass characteristics (HPF) produces, for each frequency
f.sub.AC of the alternating current (AC) applied, a waveform
referred to as the change in impedance, in FIG. 1a referred to as
dZ(f.sub.AC). Depending on the filter applied, impedance changes
due to respiration or ventilation may be isolated from impedance
changes related to the cardiac cycle. Alternatively, the high pass
filter is adapted, for example, to the heart rate in order to
perform a separation of cardiac-related impedance changes from
those related to respiration or ventilation at different
physiological states, such as rest versus exercise, or adult versus
pediatric or neonatal objects.
[0080] Impedance cardiometry, for instance, requires determination
of the first time-derivative of the impedance signal. Either
.DELTA.Z(f.sub.AC,t) or directly Z(f.sub.AC,t) (arrow with dotted
line) is input to a differentiator 67. The output of the
differentiator 67 is the rate of change of impedance
d.DELTA.Z(f.sub.AC)/dt or dZ(f.sub.AC)/dt, respectively, with
d.DELTA.Z(f.sub.AC)/dt and dZ(f.sub.AC)/dt being equivalent.
[0081] Alternatively, the timing control 62, the sine table 70, the
multiplier/accumulator (MACC) 80, the buffers 48, 58, the
processing unit 60 and the filters 62, 64, the R-Wave detector 68,
or a part thereof, are integrated into the program of a digital
signal processor (DSP) 100 or a hardwired processor (not shown) or
a fixed programmable gate array (FPGA) (not shown). The digital
waveforms Z.sub.0(f.sub.AC), .DELTA.Z(f.sub.AC,t) and the
electrocardiogram (ECG), or a subset thereof, are input to a
Cardiometry Monitor 110 in addition to data lines synchronizing the
communication and information between processing unit 60, or
digital signal processor 100, and a Cardiometry Monitor 110.
[0082] Alternatively, the cardiometry monitor (110) receives the
bioimpedance waveform Z(f.sub.AC,t) and processes the signals
Z.sub.0(f.sub.AC), .DELTA.Z(f.sub.AC,t) and dZ(f.sub.AC)/dt.
[0083] In the event the object 10 is switched to the
voltage-controlled current source (VCCS) 42 and to the differential
amplifier 50, the demodulated signal consists of a voltage portion
related to the electrical impedance obtained from the object 10 and
the electrocardiogram (ECG). Otherwise, the demodulated signal
represents the value of the calibration impedance 20. The use of a
calibration impedance is an option to eliminate the influence of
electrical circuit properties on measurements. Preferably, the
calibration impedance consists of an ohmic resistor. In the event
of a complex calibration impedance, the frequency-dependent phase
must be considered appropriately in the calculation of the object
impedance.
[0084] Alternatively, the apparatus of the invention is applicable
to other areas of the object 10, such as for example the limbs in
order to determine peripheral blood volume changes.
[0085] Other embodiments according to the apparatus of the
invention include the replacement of the voltage-controlled current
source by a voltage source including a current measuring
circuit.
[0086] Another embodiment relies on the discrete values of a sine
waveform stored in the sine table 70 and the accuracy of the
constant amplitude of the alternating current (AC) applied instead
of measuring the alternating current (AC) signal.
[0087] Another embodiment determines the electrical admittance
Y(f.sub.AC,t)=Y.sub.0(f.sub.AC)+.DELTA.Y(f.sub.AC, t),
[0088] where
[0089] Y(f.sub.AC,t) is the thoracic electrical bioadmittance,
[0090] Y.sub.0(f.sub.AC) is the base admittance, i.e., the
quasi-constant part, or offset, of Y(f.sub.AC,t)
[0091] .DELTA.Y(f.sub.AC,t) is the change in thoracic electrical
bioadmittance, i.e., the part of Y(f.sub.AC,t) which is related to
conductance changes during the cardiac cycle and the respiration or
ventilation cycle, instead of the electrical bioimpedance
Z(f.sub.AC,t), thus, obtaining the base admittance
Y.sub.0(f.sub.AC), the change in bioadmittance .DELTA.Y(f.sub.AC,t)
and the rate of change in bioadmittance dY(f.sub.AC)/dt.
[0092] Another embodiment determines the complex impedance by
separating the in-phase and the quadrature portion of the impedance
(or admittance) with separate correlation processes.
[0093] Another embodiment applies a set of alternating current (AC)
frequencies (further referred to as multi-frequency, or MF
application), instead of a single frequency (further referred to as
SF application) to the object 10 and utilizes a set of
frequency-sensitive demodulation processes, see FIG. 1b.
[0094] The placement of pairs of current application electrodes and
pairs of voltage sensing electrodes is not limited to the placement
shown in FIG. 1.
[0095] The current electrode 12 may be moved from the left side of
the object's neck to the forehead and the current electrode 14 may
be moved from the left side of the thorax, approximately at the
level of the xiphoid process, to the left leg. A current
application electrode placement where the current application
electrode 12 is moved from the left side of the object's neck to
the forehead and the current application electrode 14 is moved from
the left side of the thorax to the left leg is also feasible. This
kind of electrode placement may be advantageous in applications to
pediatric or neonatal objects 10, where space to place electrodes
is sometimes limited.
[0096] Alternatively, the neck electrodes 12, 16 are located at the
right side of the object 10 instead of the left side (as
illustrated in FIG. 1). Alternatively, the neck current electrode
may be applied to the forehead. The current electrode 14 may be
applied either to the thorax, approximately at the level of the
xiphoid process, or to the left leg. This kind of electrode
placement may be advantageous in applications to pediatric or
neonatal objects 10, where space to place electrodes is sometimes
limited.
[0097] Alternatively, the pair of neck electrodes is integrated
into one electrode applying the current and sensing the voltage.
Alternatively, the pair of thorax electrodes is integrated into one
electrode.
[0098] Furthermore, instead of using a first electrode array with
the four electrodes 12, 14, 16, 18 on the left side of the object
10 only, a second electrode array is placed on the right side of
the object 10 (not shown), and the application of the alternating
current (AC) and the voltage measurement is performed using the
first and the second electrode array. Likewise the previous
electrode setups, a pair of current application and voltage sensing
electrodes may be integrated into one electrode.
[0099] For esophageal applications, the electrodes can be located
as electrodes onto an esophageal catheter or probe (not shown) or
tracheal tube (not shown).
[0100] For invasive applications, the electrodes can be integrated
into aortic grafts or, in combination with cardiac pacemakers or
defibrillators, into pacing or defibrillation leads, or other
implants which remain in the body of the object.
[0101] FIG. 1b illustrates an embodiment similar to the embodiment
of FIG. 1a except for an implementation of a parallel in-phase and
quadrature demodulation process aimed at the three different
frequency contents of the measured voltage signal measured by the
differential amplifier (A) 50 and digitized by a first
analog-to-digital converter (ADC) 52. The embodiment of FIG. 1b
applies an alternating current (AC), which consists of three
different frequencies f.sub.1, f.sub.2, and f.sub.3, each of which
is applied at a constant amplitude. The alternating current (AC)
applied is not measured but each frequency component represented,
in frequency and phase, by a first sine table 71, a second sine
table 72 and a third sine table 73. In the multi-frequency (MF)
alternating current (AC) embodiment, demodulation must be performed
for each applied frequency f.sub.AC separately.
[0102] A first MACC 81 accumulates the products of the sampled
values measured, amplified and digitized by the ADC 52 and the
reference sine values of the first frequency f.sub.1 stored in the
first sine table 71, calculating as a result a value proportional
to the in-phase portion, or real part of the impedance at the first
frequency f.sub.1.
[0103] A second MACC 82 accumulates the products of the sampled
values measured, amplified and digitized by the ADC 52 and the
reference cosine values of the first frequency f.sub.1 stored in
the first sine table 71, producing as a result a value proportional
to the quadrature portion, or imaginary part of the impedance at
the first frequency f.sub.1.
[0104] A third MACC 83 accumulates the products of the sampled
values measured, amplified and digitized by the ADC 52 and the
reference sine values of the second frequency f.sub.2 stored in the
second sine table 72, producing as a result a value proportional to
the in-phase portion, or real part of the impedance at the second
frequency f.sub.2.
[0105] A fourth MACC 84 accumulates the products of the sampled
values measured, amplified and digitized by the ADC 52 and the
reference cosine values of the second frequency f.sub.2 stored in
the second sine table 72, producing as a result a value
proportional to the quadrature portion, or imaginary part of the
impedance at the second frequency f.sub.2.
[0106] A fifth MACC 85 accumulates the products of the sampled
values measured, amplified and digitized by the ADC 52 and the
reference sine values of the third frequency f.sub.3 stored in the
third sine table 73, producing as a result a value proportional to
the in-phase portion, or real part of the impedance at the third
frequency f.sub.3.
[0107] A sixth MACC 86 accumulates the products of the sampled
values measured, amplified and digitized by the ADC 52 and the
reference cosine values of the third frequency f.sub.3 stored in
the third sine table 73, producing as a result a value proportional
to the quadrature portion, or imaginary part of the impedance at
the third frequency f.sub.3.
[0108] While the embodiment shows a parallel process of
multiplication and accumulation, the process may be serialized by
implementing memory in order to store intermediate results. More
advantageously, the timing control 62, the processing unit 60, the
sine tables 71, 72, 73 and multiplier/accumulators (MACC) 81, 82,
83, 84, 85, 86 are incorporated into a digital signal processor
(DSP) 100 or into a hard-wired processor or into a fixed
programmable gate array (FPGA).
[0109] FIG. 1c-1g illustrate the application of a multi-frequency
alternating current (AC) application.
[0110] For an example, the alternating current (AC) signal
outputted by the voltage-controlled current source (VCCS) 42 shall
consist of three different sinusoidal signals
S.sub.i(t)=A.sub.isin(2.pi.f.sub.i)t
[0111] of a first frequency f.sub.1, a second frequency f.sub.2 and
a third frequency f.sub.3. In this example, the second frequency is
three times the frequency of the first frequency, i.e,
f.sub.2=3f.sub.1
[0112] and the third frequency is five times the frequency of the
first frequency, i.e., f.sub.3=5f.sub.1
[0113] FIG. 1c illustrates the signal of the first frequency over a
time. FIG. 1d illustrates the signal of the second frequency over
the same time. FIG. 1e illustrates the signal of the third
frequency over the same time. For simplification, all signals are
shown with the same amplitude A.sub.i normalized at 1.
[0114] FIG. 1f superimposes the three signals S.sub.i(t) in the
same diagram. The frequencies and their phases are chosen in such a
way that within the given window the signal begin and end at zero
crossings.
[0115] The sum of the three signals, i.e. S .function. ( t ) = i =
1 3 .times. .times. A i sin .function. ( 2 .times. .pi. .times.
.times. f i ) .times. t ##EQU5##
[0116] is shown in FIG. 1g.
[0117] FIG. 2a provides a flow chart of the correlation process for
a preferred embodiment, without being limited to it.
[0118] In this embodiment, as an example, but not limited to, the
in-phase portion I(Z(t)) of the complex impedance at a single
frequency (SF) of f.sub.AC=50 KHz is of interest. Furthermore, the
alternating current (AC) applied is not measured but held constant.
Accordingly, the voltage due to the alternating current (AC)
application is proportional to the impedance Z(t).
[0119] A sine table 70 (FIG. 1a) is set up in such a way that its
contents represents, in digital values, the alternating current
(AC) amplitudes at all times. For measuring the voltage due to the
alternating current (AC) application, the embodiment utilizes the
first analog-to-digital converter (ADC) 52 operating at a sampling
rate of 500 kHz. In order to achieve, for example, an update rate
of the impedance signal of 200 Hz, i.e., a new value for Z(t) every
5 ms, the correlation vector must include N CORR = 500 .times.
.times. KHz 200 .times. .times. Hz = 2 .times. , .times. 500
##EQU6##
[0120] correlation pairs. Each pair consists of a digitized voltage
sample and a corresponding digital value in the sine table, of
which the digital value represents the amplitude of the alternating
current at the time of the voltage sampling.
[0121] At the beginning of the correlation process 200, the
correlation counter N and the accumulator ACC are reset by circuit
202.
[0122] The multiplier obtains the first digitized voltage sample
V.sub.0 (acquired by the first analog-to-digital converter (ADC) 52
and the corresponding first digital value provided by the sine
table (representing the alternating current (AC) applied), and
calculates the product 204 thereof. The product is added to the
accumulator.
[0123] Then the multiplier obtains the second digitized voltage
sample (acquired by the ADC) and the corresponding second digital
value provided by the sine table (representing the alternating
current (AC) applied), calculates the product thereof and adds the
product to the accumulator 204. This process is reiterated by
circuit 206 until the total number of desired correlation pairs is
reached (in this embodiment, the correlation vector contains 2,500
correlation pairs).
[0124] After multiplying and accumulating the 2,500 correlation
pairs, the content of the accumulator is normalized by normalizer
208 by dividing the result in the accumulator by the number of
pairs N which contributed to the result.
[0125] This normalized result is the result of the correlation
process and represents one discrete sample of the Z(t) waveform.
Consecutive correlation processes are executed until decision
circuit 210 blocks reiteration and monitoring is terminated at
212.
[0126] In the event the quadrature portion Q(Z(t)) of the impedance
is of interest, correlation is performed of the samples obtained by
the first analog-to-digital converter (ADC) 52 with a digitized
sinusoidal waveform shifted in phase by 90 degrees, i.e., a
digitized cosine waveform (not shown).
[0127] In the event the magnitude of the impedance is of interest,
the in-phase portion I(Z(t)) and quadrature portion Q(Z(t)) of the
impedance are determined and added (as vectors). Alternatively, the
magnitude of the impedance can be obtained by performing multiple
correlation processes, which differ in the phase shifts applied to
the sinusoid digitized waveform, the ADC samples are correlated
with. For example, the phase shift can be varied between -90
degrees and +90 degrees in steps of 1 degree. A correlation process
is performed for each phase shift (a feasible task for a digital
signal processor). The correlation with the phase shift providing
the maximum result equates (or is proportional at least) to the
impedance magnitude.
[0128] The ADC sampling rate of 500 kHz and the Z(t) update rate of
200 Hz is an example for an embodiment, and not limited to it. In
another example an ADC sampling rate of 1 MHz in the event an Z(t)
update rate of 400 Hz is specified (not shown).
[0129] FIG. 2b provides a chart which graphically explains the
correlation process for the preferred embodiment of FIG. 2a.
[0130] The horizontal axis is defined as the time axis. An analog
waveform 220 represents the alternating current (AC), which is
generated by the voltage-controlled current source (VCCS) 42. An
analog waveform 222 represents the analog voltage signal, which is
sensed by the differential amplifier 50. The analog waveform 222 is
considered as an example and may vary in amplitude and phase
depending on the actual impedance measured.
[0131] A digitized waveform 224 represents the discrete values of
the alternating current (AC), which are stored in the sine-table 70
or obtained through (current) measurement.
[0132] A digitized waveform 228 is the output of the first
analog-to-digital converter (ADC) 52 digitizing the sensed
voltage.
[0133] A digitized waveform 230 represents the results of the
correlation process at each point in time. FIG. 2b illustrates 4
cycles of the AC frequency only. The number of cycles over which
correlation is performed is determined by design requirements. The
aforementioned preferred embodiment, which specifies a Z(t) update
rate of 200 Hz, generates an AC frequency of 50 kHz and utilizes an
ADC sampling rate of 500 kHz, performs correlation over 2500 cycles
to generate one point of the function Z(t), i.e. one sample of the
electrical impedance over time.
[0134] FIG. 3 provides an overview over possible embodiments of the
invention. These embodiments can be differentiated into whether or
not a calibration impedance is used, whether or not the alternating
current (AC) applied is measured, whether direct or indirect
correlation is performed, and/or whether or not the measured
signals are fitted with ideal waveforms.
[0135] According to a first embodiment, FIG. 4a and 4b, the voltage
controlled current source (VCCS) 42, which generates a single
frequency (SF) or multi-frequency (MF) alternating current (AC),
and the differential amplifier (A) 50 are switched to the
calibration impedance 20, and the alternating current (AC) applied
and the resulting voltage are measured, amplified and
digitized.
[0136] In the event of a single frequency (SF) alternating current
(AC) application, because the frequency of the measured alternating
current (AC) applied and, thus, of the voltage measured is known,
the samples measured, amplified and digitized can be fitted towards
discrete values of an ideal sinusoidal waveform using commonly
known fitting processes.
[0137] Then, in a process further referred to as indirect
correlation, for each frequency f.sub.AC of the alternating current
(AC) applied, the amplified, digitized and optionally fitted
samples obtained from the measurement of the alternating current
(AC) applied are correlated with the discrete values of an ideal
sine waveform in order to obtain a value proportional to the
in-phase portion I.sub.AC(f.sub.AC) of the alternating current (AC)
and are correlated with the discrete values of an ideal cosine
waveform in order to obtain a value proportional to the quadrature
portion Q.sub.AC(f.sub.AC) of the alternating current (AC).
[0138] Furthermore, for each frequency f.sub.AC of the alternating
current (AC) applied, the amplified, digitized and optionally
fitted samples obtained from the measurement of the voltage are
correlated with the discrete values of an ideal sine waveform in
order to obtain a value proportional to the in-phase portion
I.sub.V(f.sub.AC) of the voltage and correlated with the discrete
values of an ideal cosine waveform in order to obtain a value
proportional to the quadrature portion Q.sub.V(f.sub.AC) of the
voltage.
[0139] Thereafter, the aforementioned processes are performed with
the current source and the differential amplifier (A) 50 switched
to the object 10.
[0140] A more detailed description of this embodiment is given
below in connection with the description of FIGS. 4a and 4b.
[0141] According to a second embodiment, FIG. 5, the voltage
controlled current source (VCCS) 42, which generates a
single-frequency (SF) alternating current (AC), and the
differential amplifier (A) 50 are switched to the calibration
impedance 20, and the alternating current (AC) applied and the
resulting voltage are measured, amplified and digitized.
[0142] Because the frequency of the alternating current (AC)
applied and, thus, of the voltage measured is known, the samples
obtained, amplified and digitized can be fitted towards discrete
values of an ideal sinusoidal waveform using commonly known fitting
processes.
[0143] Then, in a process further referred to as direct
correlation, the amplified, digitized and optionally fitted samples
obtained from the measurement of the voltage and obtained from the
measurement of the alternating current (AC) applied are
correlated.
[0144] Thereafter, the aforementioned processes are performed with
the current source (VCCS) 42 and to the differential amplifier 50
switched to the object 10.
[0145] A more detailed description of this embodiment is given
below with FIG. 5.
[0146] According to a third embodiment, FIG. 6, the voltage
controlled current source (VCCS) 42, which generates a single
frequency (SF) or multi-frequency (MF) alternating current (AC),
and the differential amplifier (A) 50 are switched to the
calibration impedance 20 but only the resulting voltage is
measured, amplified and digitized.
[0147] In the event of a single frequency (SF) alternating current
(AC) application, because the frequency of the alternating current
(AC) applied and, thus, the frequency of the measured voltage is
known, the samples obtained, amplified and digitized can be fitted
towards discrete values of an ideal sinusoidal waveform using
commonly known fitting processes.
[0148] Then, for each frequency f.sub.AC of the alternating current
(AC) applied, the amplified, digitized and optionally fitted
samples obtained from the voltage measurement are correlated with
the discrete values of an ideal sine waveform in order to obtain a
value proportional to the in-phase portion IV (f.sub.AC) of the
voltage and correlated with the discrete values of an ideal cosine
waveform in order to obtain a value proportional to the quadrature
portion Q.sub.V(f.sub.AC) of the voltage.
[0149] Thereafter, the aforementioned processes are performed with
the alternating current (AC) source 42 and the differential
amplifier (A) 50 switched to the object 10.
[0150] A more detailed description of this embodiment is given
below with FIG. 6.
[0151] According to a forth embodiment, FIG. 7, the voltage
controlled current source (VCCS) 42, which generates a single
frequency (SF) or multi-frequency (MF) alternating current (AC),
and the differential amplifier (A) 50 are switched to the object
10, and the alternating current (AC), the excitation signal,
applied and the resulting voltage, the response signal, are
measured/acquired, amplified and digitized.
[0152] In the event of a single frequency (SF) alternating current
(AC) application, because the frequency of the alternating current
(AC) applied and, thus, of the voltage measured is known, the
samples obtained, amplified and digitized can be fitted towards
discrete values of an ideal sinusoidal waveform using commonly
known fitting processes.
[0153] Then, in a process further referred to as indirect
correlation, for each frequency f.sub.AC of the alternating current
(AC) applied, the amplified, digitized and optionally fitted
samples obtained from the measurement of the alternating current
(AC) applied are correlated with the discrete values of an ideal
sine waveform in order to obtain a value proportional to the
in-phase portion I.sub.AC(f.sub.AC) of the alternating current (AC)
and correlated with the discrete values of an ideal cosine waveform
in order to obtain a value proportional to the quadrature portion
Q.sub.AC(f.sub.AC) of the alternating current (AC).
[0154] Furthermore, for each frequency f.sub.AC of the alternating
current (AC) applied, the amplified, digitized and optionally
fitted samples obtained from the voltage measurement are correlated
with the discrete values of an ideal sine waveform in order to
obtain a value proportional to the in-phase portion IV(f.sub.AC) of
the voltage and correlated with the discrete values of an ideal
cosine waveform in order to obtain a value proportional to the
quadrature portion Q.sub.V(f.sub.AC) of the voltage.
[0155] A more detailed description of this embodiment is given
below with FIG. 7.
[0156] According to a fifth embodiment, FIG. 8, the voltage
controlled current source (VCCS) 42, which generates a
single-frequency (SF) alternating current (AC), and the
differential amplifier (A) 50 are switched to the object 10, and
the alternating current (AC), the excitation signal, applied and
the resulting voltage, the response signal, are measured/acquired,
amplified and digitized.
[0157] Because the frequency of the excitation signal and of the
response signal is known, the samples acquired, amplified and
digitized can be fitted towards discrete values of an ideal
sinusoidal waveform using commonly known fitting processes.
[0158] Then, in a process further referred to as direct
correlation, the digitized and optionally fitted samples of the
response signal and the excitation signal are correlated.
[0159] A more detailed description of the embodiment is given below
with FIG. 8.
[0160] According to a sixth embodiment, FIG. 9, the voltage
controlled current source (VCCS) 42, which generates a single
frequency (SF) or multi-frequency (MF) alternating current (AC),
the excitation signal, and the differential amplifier (A) 50 are
switched to the object 10 but only the resulting voltage, the
response signal, is measured, amplified and digitized.
[0161] Because the frequency of the excitation signal and the
response signal is known, the samples obtained, amplified and
digitized can be fitted towards discrete values of an ideal
sinusoidal waveform using commonly known fitting processes.
[0162] Then, in a process referred to as indirect correlation, for
each frequency f.sub.AC of the alternating current (AC) applied,
the amplified, digitized and optionally fitted samples obtained
from the voltage measurement are correlated with the discrete
values of an ideal sine waveform (SF), or waveforms (MF), which
represent the alternating current (AC) applied in frequency
f.sub.AC, amplitude and phase.
[0163] A more detailed description is given below with FIG. 9.
[0164] Now the six possible embodiments of the invention outlined
by the above overview of FIG. 3 are described in more detail.
[0165] The flowchart of FIGS. 4a and 4b describes the determination
of the complex object bioimpedance (Z.sub.OBJ), i.e., the impedance
of interest, by measurements of the excitation signal, for example,
an alternating current (AC) applied, and the response signal, in
this example, a resulting alternating voltage, indirect correlation
thereof, and the use of a calibration impedance. The description
encompasses an embodiment of an alternating current (AC) of a
single frequency (SF) and an ohmic resistor as the calibration
impedance, but is not limited to it.
[0166] Measurement of the Calibration Impedance
[0167] FIG. 4a illustrates that, for example, by means 400 an
alternating current (AC) source, including a current monitor 44,
and a differential amplifier 50 are switched to the calibration
impedance 20 (Z.sub.CAL). The current monitor provides a voltage
directly proportional and in phase with the alternating current
(AC) applied, which is sensed, amplified and digitized by a second
analog-to-digital converter (ADC 2) 46. Because the frequency of
the alternating current (AC) is known a priori, the digitized
samples can be fitted towards discrete values of an ideal sinusoid
using commonly known algorithms and are further referred to as the
Calibration Current Samples 402. The differential amplifier senses
the voltage across the calibration impedance, which is amplified
and, simultaneously with the second analog-to-digital converter
(ADC 2) 46, digitized by a first analog-to-digital converter (ADC
1) 52. Because the frequency of the alternating current (AC) and,
thus, of the voltage measured is known a priori, the digitized
samples can be fitted towards discrete values of an ideal sinusoid
using commonly known algorithms and are further referred to as
Calibration Voltage Samples 404.
[0168] Correlation 406, i.e. pair-wise multiplication of
Calibration Current Samples with the corresponding discrete values
of a unity sine waveform (SIN), and accumulation, results in a
value proportional as the in-phase portion of the current (I.sub.AC
CAL') 414, which, at this point, is uncorrected for any phase shift
caused by the measurement system. Correlation, i.e. pair-wise
multiplication of Calibration Current Samples with the
corresponding discrete values of a unity cosine waveform (COS), and
accumulation 408, results in a value proportional to the quadrature
portion of the current (Q.sub.AC CAL') 416, which, at this point,
is uncorrected for any phase shift caused by the measurement
system. Preferably, the unity sine waveform SIN is in phase with
the sinusoidal voltage signal controlling the current source (VCCS
42, FIG. 1a).
[0169] The equivalent of the magnitude of current through the
calibration impedance, |AC.sub.CAL'|, 422, is calculated as the
square root of the sum of squared in-phase portion (IA.sub.C CAL')
and squared quadrature portion of current (Q.sub.AC CAL').
[0170] The phase shift of the current .phi..sub.AC CAL'. 430,
including any phase shift caused by the measurement system, is
calculated as the arctan of the ratio of the quadrature portion
(Q.sub.AC CAL') over the in-phase portion of the current (I.sub.AC
CAL').
[0171] Correlation 410, i.e. pair-wise multiplication of
Calibration Voltage Samples with the corresponding discrete values
of a unity sine waveform (SIN), and accumulation, results in a
value proportional to the in-phase portion of the voltage (I.sub.V
CAL') 418, which, at this point, is uncorrected for any phase shift
caused by the measurement system. Correlation, i.e. pair-wise
multiplication of Calibration Voltage Samples with the
corresponding discrete values of a unity cosine waveform (COS), and
accumulation 412, results in a value proportional as the quadrature
portion of the voltage (Q.sub.V CAL') 420, which, at this point, is
uncorrected for any system phase.
[0172] The equivalent to the voltage magnitude across the
calibration impedance, |V.sub.CAL'|, 424, is calculated as the
square root of the sum of squared in-phase portion (I.sub.V CAL')
and squared quadrature portion of voltage (Q.sub.V CAL').
[0173] The phase shift of the voltage, .phi..sub.V CAL', 432,
including any phase shift caused by the measurement system, is
calculated as the arctan of the ratio of the quadrature portion
(Q.sub.V CAL') and the in-phase portion of the voltage (I.sub.V
CAL').
[0174] In the preferred embodiment, however, in which an ohmic
resistor (with theoretically no phase shift between current and
voltage) is utilized as the calibration impedance, the
aforementioned calculation reveals directly the phase shift of the
system, .phi..sub.SYS, 434, which is determined as the difference
between the phase of the voltage measured and the phase of the
alternating current (AC) applied.
[0175] The equivalent to the calibration impedance magnitude, 436,
is calculated as the ratio of the equivalent of the voltage
magnitude equivalent across the calibration impedance, |V.sub.CAL'|
and the magnitude of the current magnitude equivalent through the
calibration impedance, |AC.sub.CAL'|.
Measurement of the Object Impedance
[0176] Then, by means 450 the alternating current (AC) 42 source
including the current monitor 44 and the differential amplifier 50
are switched to the object 10 or impedance (Z.sub.OBJ),
respectively, (FIG. 4b). The current monitor 44 provides a voltage
directly proportional and in phase with the alternating current
(AC) applied, which is sensed, amplified and digitized by the
second analog-to-digital-converter (ADC 2) 46. Because the
frequency of the alternating current (AC), the excitation signal,
is known a priori, the digitized samples can be fitted towards the
values of an ideal sinusoid using commonly known algorithms and are
further referred to as the Object Current Samples 452. The
differential amplifier senses or acquires, respectively, the
voltage across the object 10, the object impedance Z.sub.OBJ, which
is amplified and, simultaneously with the second analog-to-digital
converter (ADC 2) 46, digitized by the first analog-to-digital
converter (ADC 1) 52. Because the frequency of the alternating
current (AC), the excitation signal, and, thus, of the voltage
measured, the response signal, is known, a priori, the digitized
samples can be fitted towards discrete values of an ideal sinusoid
using commonly known algorithms and are further referred to as
Object Voltage Samples 454.
[0177] Correlation 456, i.e. pair-wise multiplication of Object
Current Samples with the corresponding discrete values of a unity
sine waveform (SIN), and accumulation, results in a value
proportional as the in-phase portion of the current (I.sub.AC OBJ')
464, which, at this point, is uncorrected for any phase shift
caused by the measurement system. Correlation 458, i.e. pair-wise
multiplication of Object Current Samples with the corresponding
discrete values of a unity cosine waveform (COS), and accumulation,
results in a value proportional to the quadrature portion of the
current (Q.sub.AC OBJ') 466, which, at this point, is uncorrected
for any phase shift caused by the measurement system.
[0178] The equivalent of the object current magnitude, through the
object impedance, |AC.sub.OBJ'|, 472, is calculated as the square
root of the sum of squared in-phase portion of current amplitude
(I.sub.AC OBJ') and squared quadrature portion of current (Q.sub.AC
OBJ').
[0179] The phase of the object current, .phi..sub.AC OBJ', 480,
including any phase shift caused by the measurement system, is
calculated as the arctan of the ratio of the quadrature portion and
in-phase portion of the current.
[0180] Correlation 460, i.e. pair-wise multiplication of Object
Voltage Samples with the corresponding discrete values of a unity
sine waveform (SIN), and accumulation, results in a value
proportional to the in-phase portion of the object voltage (I.sub.V
OBJ') 468, which, at this point, is uncorrected for any phase shift
caused by the measurement system. Correlation 462, i.e. pair-wise
multiplication of Object Voltage Samples with the corresponding
discrete values of a unity cosine waveform (COS), and accumulation,
results in a value proportional to the quadrature portion of the
object voltage (Q.sub.V OBJ') 470, which, at this point, is
uncorrected for any phase shift caused by the measurement
system.
[0181] The equivalent to the voltage magnitude across the object
impedance, |V.sub.OBJ'|, 474, is calculated as the square root of
the sum of the squared in-phase portion (I.sub.V OBJ') and the
squared quadrature portion of voltage (Q.sub.V OBJ').
[0182] The phase of the voltage, .phi..sub.V OBJ', 482, including
any phase shift caused by the measurement system, is calculated as
the arctan of the ratio of the quadrature portion (Q.sub.V OBJ')
and the in-phase portion of the voltage (I.sub.V OBJ').
[0183] In the preferred embodiment, in which an ohmic resistor
(with theoretically no phase shift between current and voltage) is
utilized as the calibration impedance 20, the phase shift between
the response signal, the voltage, measured across and the
excitation signal, the alternating current (AC), applied to the
object impedance, cos .phi..sub.OBJ, is calculated as the
difference between the phase of the voltage and the phase of the
current, of which the phase shift of the system, .phi..sub.SYS,
434, is subtracted by circuit 484.
[0184] The magnitude equivalent of the object impedance,
|Z.sub.OBJ'|, is calculated by circuit 486 as the ratio of the
object voltage magnitude equivalent |V.sub.OBJ'| and the current
magnitude equivalent |AC.sub.OBJ'|, I which is multiplied by the
cosine of the phase shift between the voltage across and the
current through the object impedance, cos .phi..sub.OBJ.
[0185] The magnitude of the object impedance, |Z.sub.OBJ|, is
calculated by circuit 488 as the ratio of the (a priori known)
calibration impedance magnitude, |Z.sub.CAL|, to the magnitude
equivalent of the calibration impedance, |Z.sub.CAL'|, times the
magnitude equivalent of the object impedance, |Z.sub.OBJ'|.
[0186] The real part (in-phase portion) of the object impedance
(Re(Z.sub.OBJ)), 490, is calculated from the magnitude of the
object impedance, |Z.sub.OBJ| and the phase of object impedance cos
.phi..sub.OBJ. The imaginary part (quadrature portion) of the
object impedance (Im(Z.sub.OBJ)), 492, is calculated from the
magnitude of object impedance, |Z.sub.OBJ| and the phase of object
impedance
[0187] Alternatively, the second analog-to-digital converter (ADC
2) 46 and the first analog-to-digital converter (ADC 1) 52 can be
replaced by a single analog-to-digital converter (ADC) with
multiplexed inputs (not shown).
[0188] The embodiment of FIG. 4a, 4b can be adapted for a
multi-frequency (MF) alternating current (AC) application by
executing the correlation processes 406, 408, 410, 412 (calibration
impedance) and 456, 458, 460, 462 (object impedance) for each
frequency f.sub.AC of the alternating current (AC) applied, then
obtaining results for complex impedances depending on
frequency.
[0189] The flowchart of FIG. 5 describes the determination of the
complex object bioimpedance (Z.sub.OBJ), i.e., the impedance of
interest, by measurements of the excitation signal, the alternating
current (AC) applied, and the response signal, the alternating
voltage, direct correlation thereof, the calculation of the real
and imaginary part of the object impedance, and the use of a
calibration impedance. The description encompasses an embodiment of
an alternating current (AC) of a single frequency (SF) and an ohmic
resistor as the calibration impedance 20, but is not limited
thereto.
Measurement of the Calibration Impedance
[0190] FIG. 5 illustrates that, for example, by means 500 an
alternating current (AC) source, including a current monitor 44,
and a differential amplifier 50 are switched to the calibration
impedance 20 (Z.sub.CAL). The current monitor 44 provides a voltage
directly proportional and in phase with the alternating current
(AC) applied, which is sensed, amplified and digitized by a second
analog-to-digital converter (ADC 2) 46. Because the frequency of
the alternating current (AC), the excitation signal, is known a
priori, the digitized samples can be fitted towards discrete values
of an ideal sinusoid using commonly known algorithms and are
further referred to as the Calibration Current Samples 502. A
differential amplifier senses the voltage across the calibration
impedance, which is amplified and, synchronously with the second
analog-to-digital converter (ADC 2) 46, digitized by a first
analog-to-digital converter (ADC 1) 52. Because the frequency of
the excitation signal and the response signal is known a priori,
the digitized samples can be fitted towards discrete values of an
ideal sinusoid using commonly known algorithms and are further
referred to as Calibration Voltage Samples 504.
[0191] Correlation accumulation 506, i.e. pair-wise multiplication
of the Calibration Voltage Samples with the corresponding
Calibration Current Samples, and results in a value proportional to
the real part (in-phase portion) of the calibration impedance
(Re(Z.sub.CAL')) 510, which, at this point, is uncorrected for any
phase shift caused by the measurement system. Correlation 508, i.e.
pair-wise multiplication of the Calibration Voltage Samples with
the corresponding Calibration Current Samples, which are shifted in
phase by -90 degrees, and accumulation, results in a value
proportional to the imaginary part (quadrature portion) of the
calibration impedance (Im(Z.sub.CAL')) 512, which, at this point,
is uncorrected for any phase shift due to the measurement
system.
[0192] The calibration impedance magnitude equivalent,
|Z.sub.CAL'|, 514, is calculated as the square root of the sum of
the squared real part of the calibration impedance (Re(Z.sub.CAL'))
and the squared imaginary part of the calibration impedance
(Im(Z.sub.CAL')).
[0193] The calibration impedance phase, .phi..sub.CAL, 516,
including any phase shift due to the measurement system, is
calculated as the arctan of the ratio of the imaginary part and the
real part. In the preferred embodiment, however, in which an ohmic
resistor (with theoretically no phase shift between the voltage
across and the current through it) is utilized as the calibration
impedance, the aforementioned calculation provides directly the
phase shift of the system.
Measurement of the Object Impedance
[0194] Thereafter, by means 520 the alternating current (AC)
source, including a current monitor 44, and the differential
amplifier 50 are switched to the object 10 or object impedance
(Z.sub.OBJ). The current monitor 44 provides a voltage directly
proportional and in phase with the alternating current (AC), the
excitation signal, applied, which is acquired (sensed), amplified
and digitized by the second analog-to-digital converter (ADC 2) 46.
Because the frequency of the alternating current (ADC) is known a
priori, the digitized samples can be fitted towards discrete values
of an ideal sinusoid using commonly known algorithms and are
further referred to as the Object Current Samples 522. The
differential amplifier senses the voltage across the object
impedance, the response signal, which is amplified and,
simultaneously with the second analog-to-digital converter (ADC 2)
46, digitized by a first analog-to-digital converter (ADC 1) 52.
Because the frequency of the alternating current (AC), the
excitation signal, and, thus, of the voltage measured, the response
signal, is known a priori, the digitized samples can be fitted
towards discrete values of an ideal sinusoid and are further
referred to as Object Voltage Samples 524.
[0195] Correlation 526, i.e. pair-wise multiplication of the Object
Voltage Samples with the corresponding Object Current Samples, and
accumulation, results in a value proportional to the real part
(in-phase portion) of the object impedance (Re(Z.sub.OBJ')) 530,
which, at this point, is uncorrected for any phase shift caused by
the measurement system. Correlation 528, i.e. pair-wise
multiplication of the Object Voltage Samples with the corresponding
Object Current Samples, which are shifted in time by -90 degrees,
and accumulation, results in a value proportional to the imaginary
part (quadrature portion) of the object impedance (Im(Z.sub.OBJ'))
532, which, at this point, is uncorrected for any phase shift
caused by the measurement system.
[0196] The object impedance magnitude equivalent, |Z.sub.OBJ'|,
534, is calculated as the square root of the sum of the squared
real part of object impedance (Re(Z.sub.OBJ')) and the squared
imaginary part of object impedance (Im(Z.sub.OBJ')).
[0197] The object impedance phase, .phi..sub.OBJ', 536, including
any phase shift caused by the measurement system, further referred
to as the uncorrected phase, is calculated as the arctan of the
ratio of the imaginary part and the real part.
[0198] In the preferred embodiment, in which an ohmic resistor
(with theoretically no phase shift) is utilized as the calibration
impedance 20, the phase of the object impedance, .phi..sub.OBJ,
540, is calculated as the difference between the previously
determined uncorrected phase, .phi..sub.OBJ', and the phase of the
calibration impedance, .phi..sub.CAL, i.e., the phase shift cause
by the measurement system.
[0199] The magnitude of the object impedance 542, |Z.sub.OBJ|,
calculated as the ratio of the (a priori known) calibration
impedance magnitude, |Z.sub.CAL'|, to the magnitude equivalent of
the calibration impedance, |Z.sub.CAL'|, times the magnitude
equivalent of the object impedance, |Z.sub.OBJ'|.
[0200] The real part of object impedance (Re(Z.sub.OBJ)), 544, is
calculated from the magnitude and phase of object impedance. The
imaginary part of object impedance (Im(Z.sub.OBJ)), 546, is
calculated from the magnitude and phase of object impedance.
[0201] Alternatively, the first analog-to-digital converter (ADC 1)
52 and the second analog-to-digital converter (ADC 2) 46 can be
replaced by a single analog-to-digital converter (ADC) with
multiplexed inputs (not shown).
[0202] The flowchart of FIG. 6 describes the determination of the
complex object bioimpedance (Z.sub.OBJ), i.e., the impedance of
interest, by the application of an alternating current (AC) of
which the amplitude is not measured but held constant, measurement
of the voltage due to the alternating current (AC) applied,
correlation thereof and use of a calibration impedance. The
description encompasses the embodiment of an alternating current
(AC) of a single frequency (SF) and an ohmic resistor as the
calibration impedance 20, but is not limited to.
[0203] By means 600 the alternating current (AC) source, including
a current monitor, and a differential amplifier 50 are switched to
the calibration impedance 20, (Z.sub.CAL). The differential
amplifier acquires/senses 602 the voltage, the response signal,
across the calibration impedance 20, which is amplified and
digitized by a first analog-to-digital converter (ADC 1) 52.
Because the frequency of the exciting signal, the alternating
current (AC), and, thus, of the response signal, the voltage
measured, is known a priori, the digitized samples can be fitted
towards discrete values of an ideal sinusoid using commonly known
algorithms and are further referred to as Calibration Voltage
Samples.
[0204] Correlation 604, i.e. pair-wise multiplication of
Calibration Voltage Samples with the corresponding discrete values
of a unity sine waveform (SIN), and accumulation, results in a
value proportional to the real part (in-phase portion) of the
calibration impedance 608, which, at this point, is uncorrected for
any phase shift caused by the measurement system. Correlation 606,
i.e. pair-wise multiplication of Calibration Voltage Samples with
the corresponding discrete values of a unity cosine waveform (COS),
and accumulation, results in a value proportional to the imaginary
part (quadrature portion) of the calibration impedance 610, which,
at this point, is uncorrected for any phase shift caused by the
measurement system.
[0205] The magnitude of an equivalent to the calibration impedance
612 is calculated as the square root of the sum of the squared
uncorrected real part (in-phase portion) of the calibration
impedance (Re(Z.sub.OBJ')) 608 and the squared uncorrected
imaginary part (quadrature portion) of the calibration impedance
(Im(Z.sub.OBJ')) 610.
[0206] The phase of the calibration impedance, .phi..sub.CAL, 614,
including any phase shift caused by the measurement system, is
calculated as the arctan of the ratio of the imaginary part and the
real part of calibration impedance. In the preferred embodiment,
however, in which an ohmic resistor (with theoretically no phase)
is utilized as the calibration impedance, the aforementioned
calculation provides directly the phase shift of the measurement
system.
[0207] Then, by means 620 the alternating current (AC) source,
including a current monitor 44, and the differential amplifier 50
are switched to the object 10, the impedance (Z.sub.OBJ). The
differential amplifier acquires/senses 622 the voltage, the
response signal, across the object impedance 20, which is amplified
and digitized by a first analog-to-digital converter (ADC 1) 52.
Because the frequency of the excitation signal, the alternating
current (AC), and, thus, of the response signal, the voltage
measured, is known a priori, the digitized samples can be fitted
towards discrete values of an ideal sinusoid using commonly known
algorithms and are further referred to as Object Voltage Samples.
Correlation 624, i.e. pair-wise multiplication of Object Voltage
Samples with the corresponding discrete values of a unity sine
waveform (SIN), and accumulation, results in a value proportional
to the real part (in-phase portion) of the object impedance
(Re(Z.sub.OBJ')) 628, which, at this point, is uncorrected for any
phase shift caused by the measurement system. Correlation 626, i.e.
pair-wise multiplication of Object Voltage Samples with the
corresponding discrete values of a unity cosine waveform (COS), and
accumulation, results in a value proportional to the imaginary part
(quadrature portion) of the object impedance (Im(Z.sub.OBJ')) 630,
which, at this point, is uncorrected for any phase shift caused by
the measurement system.
[0208] The equivalent to the magnitude of object impedance
|Z.sub.OBJ'|, 632, is calculated as the square root of the sum of
the squared uncorrected real part (in-phase portion) of the object
impedance (Re(Z.sub.OBJ')) 628 and the squared imaginary part
(quadrature portion) of the object impedance (Im(Z.sub.OBJ'))
630.
[0209] The phase of the object impedance, .phi..sub.OBJ'. 634,
including any phase shift due to the measurement system, is
calculated as the arctan of the ratio of the imaginary part over
real part of the (uncorrected) object impedance.
[0210] In the preferred embodiment, in which an ohmic resistor
(with theoretically no phase shift between the voltage across and
the current trough it) is utilized as the calibration impedance 20,
the phase of the object impedance, .phi..sub.OJS, 640, is
calculated to the difference of previously determined uncorrected
object phase, .phi..sub.OBJ', and calibration impedance phase,
i.e., the phase shift caused by the measurement system.
[0211] The magnitude of the object impedance, |Z.sub.OBJ|, 642, is
calculated as the ratio of the (a priori known) calibration
impedance magnitude, |Z.sub.CAL|, and the calibration impedance
magnitude equivalent, |Z.sub.CAL'|, times the magnitude equivalent
of the object impedance, |Z.sub.OBJ'|.
[0212] The real part (in-phase portion) of the object impedance
(Re(Z.sub.OBJ)) is calculated from the magnitude and phase of
object impedance by means 644. The imaginary part (quadrature
portion) of the object impedance (Im(Z.sub.OBJ)) is calculated from
the magnitude and phase shifted by -90 degrees of object impedance
by means 646.
[0213] Alternatively, the embodiment of FIG. 6 can be adapted for a
multi-frequency (MF) alternating current (AC) application by
executing the correlation processes 604, 606 (calibration
impedance) and 624, 626 (object impedance) for each frequency
f.sub.AC of the alternating current (AC) applied, then obtaining
results for complex impedances depending on frequency.
[0214] The flowchart of FIG. 7 describes the determination of the
complex object bioimpedance (Z), i.e., the impedance of interest,
by measurements of the alternating current (AC) of a single
frequency (SF) and the alternating voltage, indirect correlation
thereof, and without the use of any calibration impedance.
[0215] By means 700 an alternating current (AC) source, including a
current monitor 44, and a differential amplifier 50 are connected
to the object 10, the object impedance (Z.sub.OBJ). The current
monitor 44 provides a voltage directly proportional and in phase
with the alternating current (AC), the excitation signal, applied,
which is sensed, amplified and digitized by a second
analog-to-digital converter (ADC 2) 46. Because the frequency of
the excitation signal, the alternating current (AC), is known a
priori, the digitized samples can be fitted towards discrete values
of an ideal sinusoid using commonly known algorithms and are
further referred to as the Object Current Samples 702. Each Object
Current Sample equates to an instantaneous value of the current
signal. The differential amplifier 50 senses the response signal,
the voltage across the object impedance, which is amplified and,
simultaneously with the second analog-to-digital converter (ADC 2)
46, digitized by a first analog-to-digital converter (ADC 1) 52.
Because the frequency of the excitation signal, the alternating
current (AC), is known a priori, the digitized samples can be
fitted towards discrete values of an ideal sinusoid using commonly
known algorithms and are further referred to as Object Voltage
Samples. Each Object Voltage Sample 704 equates to an instantaneous
value of the voltage signal. Correlation 706, i.e. pair-wise
multiplication of Object Current Samples with the corresponding
discrete values of a unity sine waveform (SIN), and accumulation,
results in a value equal to the in-phase portion of the current
(I.sub.AC) 714, which, at this point, is uncorrected for any phase
shift caused by the measurement system. Correlation 708, i.e.
pair-wise multiplication of Current Samples (OBJ) with the
corresponding discrete values of a unity cosine waveform (COS), and
accumulation, results in a value equal to the quadrature portion of
the current (Q.sub.AC) 716, which, at this point, is uncorrected
for any phase shift due to the measurement system.
[0216] The current magnitude through the object impedance, |AC|,
722, is calculated as the square root of the sum of squared
in-phase portion of current amplitude (I.sub.AC) and squared
quadrature portion of current (Q.sub.AC).
[0217] The phase of the current, .phi..sub.AC, 730, including any
phase shift due to the measurement system, is calculated as the
arctan of the ratio of the quadrature portion and the in-phase
portion of the current.
[0218] Correlation 710, i.e. pair-wise multiplication of Object
Voltage Samples with the corresponding discrete values of a unity
sine waveform (SIN), and accumulation, results in a value equal to
the in-phase portion of the voltage (I.sub.V) 718, which, at this
point, is uncorrected for any phase shift caused by the measurement
system. Correlation 712, i.e. pair-wise multiplication of Object
Voltage Samples with the corresponding discrete digital samples of
a unity cosine waveform (COS), and accumulation, results in a value
equal to the quadrature portion of the voltage (Q.sub.V) 720,
which, at this point, is uncorrected for any phase shift caused by
the measurement system.
[0219] The voltage magnitude across the calibration impedance, |V|,
724, is calculated as the square root of the sum of the squared
in-phase portion (I.sub.V) and the squared quadrature portion of
voltage (Q.sub.V).
[0220] The phase of the voltage, .phi..sub.V, 732, including any
phase shift caused by the measurement system, is calculated as the
arctan of the ratio of the quadrature portion (Q.sub.V) and the
in-phase portion of the voltage (I.sub.V).
[0221] In the preferred embodiment, in which an ohmic resistor
(with theoretically no phase shift between current and voltage) is
utilized as the calibration impedance 20, the phase shift .phi.,
724, of the impedance is calculated as the difference between the
phase of the voltage and the phase of the current.
[0222] The magnitude of the object impedance, |Z|, 736, is
calculated as the ratio of the object voltage magnitude |V| and the
current magnitude.
[0223] The real part (in-phase portion) of the object impedance
(Re(Z)), 738, is calculated from the magnitude and phase of object
impedance. The imaginary part (quadrature portion) of the object
impedance (Im(Z)), 740, is calculated from the magnitude and the
phase shifted by -90 degrees of object impedance.
[0224] Alternatively, the first analog-to-digital converter (ADC 1)
52 and the second analog-to-digital converter (ADC 2) 46 can be
replaced by a single analog-to-digital converter (ADC) with
multiplexed inputs (not shown).
[0225] The embodiment of FIG. 7 can be adapted for a
multi-frequency (MF) alternating current (AC) application by
executing the correlation processes 706, 708, 710, 712 for each
frequency f.sub.AC of the alternating current (AC) applied, then
obtaining results for complex impedances depending on
frequency.
[0226] The flowchart of FIG. 8 describes the determination of the
complex object bioimpedance (Z), i.e., the impedance of interest,
by measurements of the alternating current (AC) and the alternating
voltage, direct correlation thereof, and without the use of any
calibration impedance.
[0227] By means 800 an alternating current (AC) source, including a
current monitor 44, and a differential amplifier 50 are connected
to the object 10, the object impedance (Z.sub.OBJ). The current
monitor 44 provides a voltage directly proportional and in phase
with the excitation signal, the alternating current (AC) applied,
which is sensed, amplified and digitized by a second
analog-to-digital converter (ADC 2) 46. Because the frequency of
the excitation signal, the alternating current (AC), is known a
priori, the digitized samples can be fitted towards discrete values
of an ideal sinusoid using commonly known algorithms and are
further referred to as the Object Current Samples. Each Object
Current Sample 802 equates to an instantaneous value of the current
signal. The differential amplifier 50 senses the response signal,
the voltage across the object impedance, which is amplified and,
simultaneously with the second analog-to-digital converter (ADC 2)
46, digitized by a first analog-to-digital converter (ADC 1) 52.
Because the frequency of the excitation signal, the alternating
current (AC), and, thus, of the response signal, the voltage
measured, is known a priori, the digitized samples can be fitted
towards discrete values of an ideal sinusoid using commonly known
algorithms and are further referred to as Object Voltage Samples.
Each Object Voltage Sample 804 equates to an instantaneous value of
the voltage signal.
[0228] Correlation 806, i.e. pair-wise multiplication of Object
Voltage Samples with the corresponding discrete values of the
Object Current Samples, and accumulation, results in a value equal
to the real part (in-phase portion) of the impedance (Re(Z))
810.
[0229] Correlation 808, i.e. pair-wise multiplication of Object
Voltage Samples with the corresponding discrete digital samples of
the Object Current Samples, which are shifted in phase by -90
degrees, and accumulation, results in a value equal to the
imaginary part (quadrature portion) of the impedance (Im(Z))
812.
[0230] The magnitude of the object impedance, |Z|, 814, is
calculated as the square root of the sum of the squared real part
(Re(Z)) and the squared imaginary part of the impedance
(Im(Z)).
[0231] The phase of the object impedance, .phi..sub.AC, 816,
including any phase shift caused by the measurement system, is
calculated as the arctan of the ratio of the imaginary part and the
real part of the impedance. In addition, the system-related phase
shift may be compensated for in a phase shift applied to either the
Current or Voltage Samples (not shown).
[0232] Alternatively, the first analog-to-digital converter (ADC)
52 and the second analog-to-digital converter (ADC) 46 can be
replaced by a single analog-to-digital converter (ADC) with
multiplexed inputs (not shown).
[0233] The flowchart of FIG. 9 describes the determination of the
complex object bioimpedance (Z), i.e., the impedance of interest,
by the application of an alternating current (AC) of which the
amplitude is not measured but known and held constant, measurement
of the alternating voltage due to the alternating current (AC)
application, and without the use of any calibration impedance.
[0234] By means 900 the alternating current (AC) source, including
a current monitor 44, and a differential amplifier 50 are connected
to the object 10, the object impedance (Z). The differential
amplifier senses the response signal, the voltage across the object
impedance, which is amplified, sampled and digitized by a first
analog-to-digital converter (ADC) 52. Because the frequency of the
excitation signal, the alternating current (AC), and, thus, of the
response signal, the voltage measured, is known a priori, the
digitized samples can be fitted towards the discrete values of an
ideal sinusoid using commonly known algorithms and are further
referred to as Object Voltage Samples. Each Object Voltage Sample
902 equates to an instantaneous value of the voltage signal.
[0235] Correlation 904, i.e. pair-wise multiplication of the Object
Voltage Samples with the corresponding discrete, a priori
calibrated, Reference Current Samples (REF), and accumulation,
results in a value equal to the real part (in-phase portion) of the
impedance (Re(Z)) 908.
[0236] Correlation 906, i.e. pair-wise multiplication of Object
Voltage Samples with the corresponding discrete, a priori
calibrated, Reference Current Samples (REF), which are shifted in
phase by -90 degrees, and accumulation, results in a value equal to
the imaginary part (quadrature portion) of the impedance (Im(Z))
910.
[0237] The magnitude of the object impedance, |Z|, 912, is
calculated as the square root of the sum of the squared real part
(Re(Z)) and the squared imaginary part of the impedance
(Im(Z)).
[0238] The phase of the object impedance, .phi., 914, including any
phase shift due to the measurement system, is calculated as the
arctan of the ratio of the imaginary part and the real part of the
impedance. In addition, the system-related phase shift may be
compensated for a phase shift applied to the Reference Current
Samples (REF) (not shown).
[0239] The embodiment of FIG. 9 can be adapted for a
multi-frequency (MF) alternating current (AC) application by
executing the correlation processes 904, 906 for each frequency
f.sub.AC of the alternating current (AC) applied, then obtaining
results for complex impedances depending on frequency.
[0240] The following entries illustrate some examples of the
invention:
[0241] Entry 1. Method for digital demodulation and further
processing of signals obtained in the measurement of complex
electrical bioimpedance or bioadmittance in a biological object due
to biological activity, in particular in the measurement of the
change and/or rate of change in electrical bioimpedance or
bioadmittance, by
generating an excitation signal of known frequency content,
applying said excitation signal to the object by a first pair of
electrodes,
sensing the response signal of the object by a second pair of
electrodes,
sampling and digitizing said response signal to acquire a digitized
response signal representing the response signal with respect to
frequency content, amplitude and phase,
[0242] correlating for each frequency f.sub.AC of the excitation
signal applied digitized samples of said digitized response signal
with the discrete values of a sinusoidal reference signal to the
excitation signal (indirect correlation) or of said excitation
signal (direct correlation), respectively, and
[0243] calculating, using said correlated signals for each
frequency f.sub.AC of the excitation signal applied, complex values
for the bioimpedance Z(f.sub.AC), or the bioadmittance Y(f.sub.AC),
respectively, and providing, over time, a set of digital
bioimpedance waveforms Z(f.sub.AC,t)), or digital bioadmittance
waveforms Y(f.sub.AC,t), either
separating the base bioimpedance Z.sub.0(f.sub.AC), or base
bioadmittance Y.sub.0(f.sub.AC), from said waveforms,
separating the changes of bioimpedance .DELTA.Z(f.sub.AC,t), or the
changes of bioadmittance .DELTA.Y(f.sub.AC,t) from said waveforms,
and
determining the rate of change of the changes in bioimpedance
d(.DELTA.Z(f.sub.AC,t))/dt, or the rate of change of the changes in
bioadmittance d(.DELTA.Y(f.sub.AC,t))/dt, or
determining the rate of change in the bioimpedance waveforms
dZ(f.sub.AC,t)/dt, or
the rate of the change in the bioadmittance waveforms
dY(f.sub.AC,t)/dt, and finally
recording the temporal course of said base bioimpedance or
bioadmittance and of said changes or said rate of change in
bioimpedance or bioadmittance.
[0244] Entry 2. Method according to entry 1, wherein
the excitation signal is a sinusoidal signal of a known single
frequency f.sub.AC.
[0245] Entry 3. Method according to entry 1 or 2, wherein
the excitation signal has an amplitude and phase which are
substantially constant over time.
[0246] Entry 4. Method according to any of entries 1 to 3,
wherein
the excitation signal is switched either to the object or to a
calibration impedance, preferably an ohmic resistor.
[0247] Entry 5. Method according to any of entries 1 to 4,
wherein
[0248] the excitation signal is generated by the use of discrete
values of a sinusoidal waveform, or of a number of sinusoidal
waveforms, stored in an addressable sine look-up table which are
converted into analog excitation signals of the desired frequency
content, amplitude and phase.
[0249] Entry 6. Method according to any of entries 1 to 5,
wherein
the excitation signal is generated by time-controlled direct
digital synthesizing (DDS) and in turn driving an excitation source
generating the excitation signals of the desired frequency content,
amplitude and phase.
[0250] Entry 7. Method according to any of entries 1 to 6
wherein
the excitation signal contains frequencies in the range of 1 kHz to
1 MHz, preferably about 10 kHz to 200 kHz.
[0251] Entry 8. Method according to entry 3, wherein
the excitation signal has amplitudes of the alternating current
(AC) in the range of 0.01 mA to 5 mA.
[0252] Entry 9. Method according to any of entries 1 to 8,
wherein
the response signal is sampled by a first fast analog-to-digital
converter (ADC) at a rate significantly higher than the highest
frequency of the excitation signal, preferably by a factor in the
range of 4 to 20, in particular about 10.
[0253] Entry 10. Method according to any of entries 1 to 9
wherein
[0254] the excitation signal or the signal representing the
excitation signal is sampled by a second fast analog-to-digital
converter (ADC) at a rate significantly higher than the highest
frequency of the excitation signal, preferably by a factor in the
range of 4 to 20, in particular about 10.
[0255] Entry 11. Method according to any of entries 1 to 10,
wherein,
for each frequency f.sub.AC of the excitation signal applied, the
results of the correlation processes form digital waveforms
Z(f.sub.AC,t), which are
either input to a low pass filter for obtaining the base impedance
Z.sub.0(f.sub.AC), or base admittance Y.sub.0(f.sub.AC) of the
object, input to a high pass filter for obtaining a waveform
representing the changes in bioimpedance .DELTA.Z(f.sub.AC,t),
or
bioadmittance .DELTA.Y(f.sub.AC,t) of the object, respectively, and
optionally input to a differentiator for obtaining the rate of
change of the changes in bioimpedance d(.DELTA.Z(f.sub.AC,t))/dt,
or the rate of change of the changes in bioadmittance
d(.DELTA.Y(f.sub.AC,t))/dt,
or input to a differentiator for obtaining the rate of change in
the bioimpedance waveforms dZ(f.sub.AC,t)/dt, or the rate of the
change in the bioadmittance waveforms dY(f.sub.AC,t)/dt.
[0256] Entry 12. Method according to any of entries 1 to 11,
wherein separate correlation processes are used to determine the
in-phase portion Re(Z(f.sub.AC,t)) and the quadrature portion
Im(Z(f.sub.AC,t)) of the bioimpedance of the object, or the
in-phase portion Re(Y(f.sub.AC,t)) and the quadrature portion
Im(Y(f.sub.AC,t)) of the bioadmittance of the object,
respectively.
[0257] Entry 13. Method according to entry 4, comprising:
applying the excitation signal to the calibration impedance,
measuring, sampling and digitizing the excitation signal or a
signal representing the excitation signal to acquire Excitation
Signal Samples,
measuring, sampling and digitizing the response signal across the
calibration impedance to acquire Response Signal Samples,
for each frequency f.sub.AC of the excitation signal applied,
correlating the Excitation Signal Samples with discrete values of
an ideal sine waveform in order to obtain a value proportional to
the in-phase portion of the excitation signal related to the ideal
sine waveform as reference sine,
correlating the Excitation Signal Samples with discrete values of
an ideal cosine waveform in order to obtain a value proportional to
the quadrature portion of the excitation signal,
correlating the Response Signal Samples with discrete values of an
ideal sine waveform in order to obtain a value proportional to the
in-phase portion of the response signal,
correlating the Response Signal Samples with discrete values of an
ideal cosine waveform in order to obtain a value proportional to
the quadrature portion of the response signal,
calculating an equivalent for the magnitude and a phase of the
excitation signal,
calculating an equivalent for the magnitude and a phase of the
response signal,
calculating an equivalent for the magnitude of the calibration
impedance,
calculating a system phase,
thereafter applying the excitation signal to the object,
measuring, sampling and digitizing the excitation signal or a
signal representing the excitation signal to acquire the Excitation
Signal Samples,
measuring, sampling and digitizing the response signal across the
bioimpedance of the object, with the samples obtained further
referred to as the Response Signal Samples,
for each frequency f.sub.AC of the excitation signal applied,
correlating the Excitation Signal Samples with discrete values of
an ideal sine waveform in order to obtain a value proportional to
the in-phase portion of the excitation signal related to the
reference sine,
correlating the Excitation Signal Samples with discrete values of
an ideal cosine waveform in order to obtain a value proportional to
the quadrature portion of the excitation signal,
correlating the Response Signal Samples with discrete values of an
ideal sine waveform in order to obtain a value proportional to the
in-phase portion of the response signal,
correlating the Response Signal Samples with discrete values of an
ideal cosine waveform in order to obtain a value proportional to
the quadrature portion of the response signal,
calculating an equivalent for the magnitude and a phase of the
excitation signal,
calculating an equivalent for the magnitude and a phase of the
response signal,
calculating an equivalent for the magnitude and a phase of the
bioimpedance of the object,
calculating the magnitude of the bioimpedance Z(f.sub.AC,t) of the
object,
calculating the in-phase portion Re(Z(f.sub.AC,t)) and the
quadrature portion Im(Z(f.sub.AC,t)) of the bioimpedance of the
object, or the in-phase portion Re(Y(f.sub.AC,t)) and the
quadrature portion Im(Y(f.sub.AC,t)) of the admittance of the
object.
(FIG. 4)
[0258] Entry 14. Method according to any of entries 1 to 13,
wherein a cross-correlation signal is calculated as a function of a
time delay .tau. between the excitation signal and the response
signal by correlating the excitation signal with the response
signal after delay of the response signal by the time delay .tau.
with respect to the excitation signal.
[0259] Entry 15. Method according to entry 14, wherein the complex
Fourier transform of the cross-correlation signal is calculated to
obtain complex values proportional to the complex bioimpedance.
[0260] Entry 16. Apparatus for digital demodulation and further
processing of signals obtained by testing means in the measurement
of electrical bioimpedance or bioadmittance in a biological object,
in particular in the measurement of the change and/or rate of
change in electrical bioimpedance or bioadmittance, the testing
means comprising:
signal generating means (42, 44) generating an excitation signal of
known frequency content,
a first pair of electrodes (12, 14) for applying said excitation
signal to the object,
a second pair of electrodes (16, 18) for sensing the response
signal across the object due to the application of said excitation
signal,
first measuring means (50, 52) for acquiring, sampling and
digitizing said response signal to obtain a digitized response
signal representing the response signal with respect to frequency
content, amplitude and phase,
optional second measuring means (44, 46) for acquiring, sampling
and digitizing said excitation signal to obtain a digitized
excitation signal representing said excitation signal with respect
to frequency content, amplitude and phase,
memory means (48, 58) for temporarily storing said digitized
response signal and optionally said digitized excitation
signal,
[0261] digital demodulation means (80; 81-86) for correlating for
each frequency f.sub.AC of the excitation signal applied digitized
samples of said digitized response signal with corresponding
discrete values of a sinusoidal reference signal to the excitation
signal (indirect correlation) or said excitation signal (direct
correlation), respectively, and
[0262] processing means (60) for calculating for each frequency
f.sub.AC of the excitation signal applied complex values for the
bioimpedance Z(f.sub.AC), or the bioadmittance Y(f.sub.AC),
respectively, from the output values of the digital demodulation
means, providing, over time, a set of digital bioimpedance
waveforms Z(f.sub.AC,t)), or a set of digital bioadmittance
waveforms Y(f.sub.AC,t), either a first separating means (64)
adapted to separate the base impedance Z.sub.0(f.sub.AC), or base
admittance Y.sub.0(f.sub.AC), from said waveforms,
a second separating means (66) adapted to separate the changes in
the bioimpedance .DELTA.Z(f.sub.AC,t), or the changes in the
bioadmittance .DELTA.Y(f.sub.AC,t) from said waveforms, and
a differentiating means (67) for obtaining the rate of change of
the changes in bioimpedance d(.DELTA.Z(f.sub.AC, t))/dt or rate of
change of the changes in bioadmittance d(.DELTA.Y(f.sub.AC,t))/dt,
respectively,
or a differentiating means (67) means for obtaining the rate of
change in the bioimpedance waveforms dZ(f.sub.AC,t)/dt, or the rate
of the change in the bioadmittance waveforms dY(f.sub.AC,t)/dt,
and
recording means (110) for either recording the temporal course of
said base bioimpedance or bioadmittance and of said changes in
bioimpedance or bioadmittance or recording the rate of change in
bioimpedance or bioadmittance waveforms.
[0263] Entry 17. Apparatus according to entry 16, wherein
the signal generating means (42, 44) is adapted to generate a
sinusoidal excitation signal of a known single frequency
f.sub.AC.
[0264] Entry 18. Apparatus of entry 16 or 17 comprising a
calibration impedance (20), especially an ohmic resistor, and
switching means (30, 32, 34, 36) for switching the signal
generating means (42, 44) and first measuring means (50, 52) either
to the object (10) or to the calibration impedance (20.
[0265] Entry 19. Apparatus of any of entries 16 to 18 wherein
[0266] the signal generating means (42, 44) is adapted to generate
the excitation signal by use of discrete values of a sinusoidal
waveform, or by superposition of a number of sinusoidal waveforms,
stored in an addressable sine look-up table (70) and to transform
said waveforms onto a digital-analog-converter DAC (40) connected
to a voltage controlled current source (42) of the signal
generating means.
[0267] Entry 20. Apparatus of any of entries 16 to 19, wherein a
second fast analog-to-digital converter (46) is adapted to sample
the excitation signal or the signal representing the excitation
signal at a rate significantly higher than the highest frequency of
the excitation signal, preferably by a factor in the range of 4 to
20, in particular about 10.
[0268] Entry 21. Apparatus of any of entries 16 to 20 comprising a
direct digital synthesizer (DDS) for the generation of a sinusoidal
waveform, or for superposition of a number of sinusoidal
waveforms.
[0269] Entry 22. Apparatus according to entry 16 or 18 wherein the
signal generating means (42, 44) is adapted to generate a
sinusoidal excitation signal of frequencies in the range of 1 kHz
to 1 MHz.
[0270] Entry 23. Apparatus according to entry 22 wherein the signal
generating means (42, 44) is adapted to generate an excitation
alternating current (AC) of amplitudes in the range of 0.01 mA to 5
mA.
[0271] Entry 24. Apparatus according to any of entries 16 to 23
comprising demodulator means (80; 81-86) for digitally
demodulating, for each frequency f.sub.AC of the alternating
current (AC) applied, the response signal, which is sampled and
digitized by the first analog-to-digital converter (52), by
correlation over a number of cycles, the cycle length being defined
by the frequency f.sub.AC of the alternating current (AC) applied,
of the digitized voltage signal with a digitized signal
representing the frequency-related portion of the alternating
current (AC) applied, the multiplication and accumulation of this
demodulation is performable by a multiplier/accumulator (MACC)
controlled by the timing control (62) multiplying pairs of
digitized voltage samples and digitized values representing the
alternating current, the latter ones taken from a sine table (70;
71-73), and accumulating the products.
[0272] Entry 25. Apparatus of entry 16 and 24 comprising separate
correlation means (81-86) for determining, for each frequency
f.sub.AC of the alternating current (AC) applied, the in-phase
portion Re(Z(f.sub.AC,t)) of the bioimpedance Z(f.sub.AC,t) or
Re(Y(f.sub.AC,t)) of the bioadmittance Y(f.sub.AC,t), respectively,
and the quadrature portion Im(Z(f.sub.AC,t)) or Im(Y(f.sub.AC,t)),
respectively.
[0273] Entry 26. Apparatus according to any of entries 23 to 25
wherein the output of the digital demodulator means (80; 81-86)
forms a digital waveform which is input either to a first filter
set (64) adapted to separate the base impedance Z.sub.0(f.sub.AC),
or base admittance Y.sub.0(f.sub.AC), from said waveforms,
a second filter set (66) adapted to separate the changes in the
bioimpedance .DELTA.Z(f.sub.AC,t), or the changes in the
bioadmittance .DELTA.Y(f.sub.AC,t) from said waveforms, and
a differentiator (67) for obtaining the rate of change of the
changes in bioimpedance d(.DELTA.Z(f.sub.AC,t))/dt or rate of
change of the changes in bioadmittance d(.DELTA.Y(f.sub.AC,t))/dt,
respectively,
or a differentiator (67) for obtaining the rate of change in the
bioimpedance waveforms dZ(f.sub.AC,t)/dt, or the rate of the change
in the bioadmittance waveforms dY(f.sub.AC,t)/dt of the object.
[0274] Entry 27. Apparatus according to any of entries 23 to 26,
wherein for the calibration of the apparatus when the switching
means (30, 32, 34, 36; 400) connect the alternating current (AC)
source (40, 42) and the first measuring means (50, 52) to the
calibration impedance (20) in order to acquire Calibration Current
Samples (402) and Calibration Voltage Samples (404), for each
frequency f.sub.AC of the alternating current (AC) applied,
[0275] a correlating means (406) correlates the Calibration Current
Samples (402) with the discrete values of an ideal sine waveform in
order to obtain a value (414) proportional to the in-phase portion
of the alternating current applied (indirect correlation),
a correlation means (408) correlates the Calibration Current
Samples (402) with the discrete values of an ideal cosine waveform
in order to obtain a value (416) proportional to the quadrature
portion of the alternating current applied,
a correlation means (410) correlates the Calibration Voltage
Samples (404) with the discrete values of an ideal sine waveform in
order to obtain a value (418) proportional to the in-phase portion
of the voltage measured,
a correlation means (412) correlates the Calibration Voltage
Samples (404) with the discrete values of an ideal cosine waveform
in order to obtain a value (420) proportional to the quadrature
portion of the voltage measured,
a calculating means (422) determines an equivalent to the current
magnitude from the values proportional to in-phase portion (414)
and quadrature portion (416) of the current applied,
a calculating means (424) determines an equivalent to the voltage
magnitude from the values proportional to the in-phase portion
(418) and the quadrature portion (420) of the voltage measured,
a calculating means (430) determines a current phase from the
values proportional to the in-phase portion (414) and the
quadrature portion (416) of the current applied,
a calculating means (432) to determines a voltage phase of the
values proportional to in-phase portion (418) and quadrature
portion (420) of the voltage measured,
a calculating means (434) determines a system phase (440) as the
difference between the voltage phase (432) and the current phase
(430) and
a calculating means (436) determines an equivalent for the
magnitude of the calibration impedance (442) from the ratio of the
equivalent for the voltage magnitude (424) and the equivalent of
the current magnitude (422),
wherein further:
[0276] for the digital demodulation when the switching means (30,
32, 34, 36; 450) connect the alternating current (AC) source (70,
40, 42) and the second/first measuring means (50, 52) to the object
(10) in order to acquire Object Current Samples (452) and Object
Voltage Samples (454),
wherein for each frequency f.sub.AC of the alternating current (AC)
applied,
a correlating means (456) correlates the Object Current Samples
(452) with the discrete values of an ideal sine waveform in order
to obtain a value (464) proportional to the in-phase portion of the
alternating current applied (indirect correlation),
a correlation means (458) correlates the Object Current Samples
(452) with the discrete values of an ideal cosine waveform in order
to obtain a value (466) proportional to the quadrature portion of
the alternating current applied,
a correlation means (460) correlates the Object Voltage Samples
(454) with the discrete values of an ideal sine waveform in order
to obtain a value (468) proportional to the in-phase portion of the
voltage measured,
a correlation means (462) correlates the Object Voltage Samples
(454) with the discrete values of an ideal cosine waveform in order
to obtain a value (470) proportional to the quadrature portion of
the voltage measured,
a calculating means (472) determines an equivalent to the current
magnitude from the values proportional to in-phase portion (464)
and quadrature portion (466) of the current applied,
a calculating means (474) determines an equivalent to the voltage
magnitude from the values proportional to the in-phase portion
(468) and the quadrature portion (470) of the voltage measured,
a calculating means (480) determines a current phase from the
values proportional to the in-phase portion (464) and the
quadrature portion (466) of the current applied,
a calculating means (482) determines a voltage phase from the
values proportional to in-phase portion (468) and quadrature
portion (470) of the voltage measured,
a calculating means (484) determines an object phase as the
difference between the voltage phase (480) and the current phase
(482), corrected for the system phase (440) and
a calculating means (486) determines an equivalent for the
magnitude of the object impedance from the ratio of the equivalent
for the voltage magnitude (474) and the equivalent of the current
magnitude (472)
[0277] a calculating means (488) determines the magnitude of the
object impedance from the ratio of the a priori known magnitude of
the calibration impedance (20) and the equivalent for the
calibration impedance magnitude (442), multiplied by the equivalent
for the object impedance magnitude (486),
and further comprising in the event the real or imaginary portion
of the object impedance is further processed:
a calculating means (490) to determines the in-phase portion (real
part) and/or a calculating means (492) to calculates the quadrature
portion (imaginary part) from the magnitude (488) and phase (484)
of the object impedance. (FIG. 4a,b)
[0278] Entry 28. Apparatus according to any of entries 23 to 26,
wherein:
[0279] for the calibration of the apparatus when the switching
means (30, 32, 34, 36; 500) connect the alternating current (AC)
source (40, 42) and the first measuring means (50, 52) to the
calibration impedance (20) in order to acquire Calibration Current
Samples (502) and Calibration Voltage Samples (504),
for the frequency f.sub.AC of the alternating current (AC)
applied
a correlating means (506) correlates the Calibration Current
Samples (502) with the Calibration Voltage Samples (504) in order
to obtain a value (510) proportional to the in-phase portion of the
calibration impedance (direct correlation), and
[0280] a correlation means (508) correlates the Calibration Current
Samples (502) with the Calibration Voltage Samples (504) samples,
which are shifted in time by -90 degrees, in order to obtain a
value (512) proportional to the quadrature portion of the
calibration impedance,
a calculating means (514) calculates an equivalent to the magnitude
of the calibration impedance from the in-phase portion (510) and
quadrature portion (512),
a calculating means (516) calculates the phase of the calibration
impedance from the in-phase portion (510) and quadrature portion
(512),
wherein further:
[0281] for the digital demodulation when the switching means (30,
32, 34, 36; 520) connect the alternating current (AC) source (40,
42) and the first measuring means (50, 52) to the object (10) in
order to acquire Object Current Samples (522) and Object Voltage
Samples (524),
for the frequency f.sub.AC of the alternating current (AC)
applied
a correlating means (526) correlates the Object Current Samples
(522) with the Object Voltage Samples (524) in order to obtain a
value (530) proportional to the in-phase portion of the calibration
impedance (direct correlation), and
[0282] a correlation means (528) correlates the Object Current
Samples (522) with the Object Voltage Samples (524) samples, which
are shifted in time by -90 degrees, in order to obtain a value
(532) proportional to the quadrature portion of the calibration
impedance,
a calculating means (534) calculates an equivalent to the magnitude
of the object impedance from the in-phase portion (530) and
quadrature portion (532),
a calculating means (536) calculates the uncorrected phase of the
object impedance from the in-phase portion (530) and quadrature
portion (532),
calculating means (540) for calculating the correct phase of the
object impedance from the uncorrected object impedance (536) and
from the phase of the calibration impedance (516),
[0283] calculating means (542) for calculating the magnitude of the
object impedance from the ratio of the a priori known magnitude of
the calibration impedance (20) and the determined equivalent for
the calibration impedance magnitude (514), multiplied by the
determined equivalent for the object impedance magnitude (534),
and wherein further in the event the real or imaginary portion of
the object impedance is further processed,
calculating means (544) to calculate the in-phase portion (real
part) and/or calculating means (546) to calculate the quadrature
portion (imaginary part) from the magnitude (542) and phase (540)
of the object impedance.
(FIG. 5)
[0284] Entry 29. Apparatus according to any of entries 23 to 26,
wherein:
[0285] for the calibration of the apparatus when the switching
means (30, 32, 34, 36; 600) connect the alternating current (AC)
source (70, 40, 42) and the first measuring means (50, 52) to the
calibration impedance (20) in order to acquire Calibration Voltage
Samples (602),
for each alternating current frequency f.sub.AC applied,
a correlating means (604) correlates the Calibration Voltage
Samples (602) with the discrete values of an ideal sine waveform in
order to obtain a value (608) proportional to the in-phase portion
of the calibration impedance,
a correlation means (606) correlates the Calibration Voltage
Samples (602) with the discrete values of an ideal cosine waveform
in order to obtain a value (610) proportional to the quadrature
portion of the calibration impedance,
a calculation means (612) determines an equivalent to the magnitude
of the calibration impedance,
a calculation means (614) determines the phase of the calibration
impedance,
wherein further:
for the digital demodulation when the switching means (30, 32, 34,
36; 620) connect the alternating current (AC) source 70, 40, 42)
and the first measuring means (50, 52) to the object (10) in order
to acquire Object Voltage Samples (622),
for each alternating current frequency f.sub.AC applied,
a correlating means (624) correlates the Object Voltage Samples
(622) with the samples of an ideal sine waveform in order to obtain
a value (628) proportional to the in-phase portion of the object
impedance,
a correlation means (626) correlates the Object Voltage Samples
(622) with the samples of an ideal cosine waveform in order to
obtain a value (630) proportional to the quadrature portion of the
object impedance,
a calculation means (632) determines an equivalent to the magnitude
of the object impedance,
a calculation means (634) determines the phase of the uncorrected
object impedance,
a calculating means (640) determines the correct phase of the
object impedance by subtracting the phase of the calibration
impedance (614) from the phase of the uncorrected object impedance
(634),
[0286] a calculating means (642) determines the magnitude of the
object impedance from the ratio of the a priori known magnitude of
the calibration impedance and the determined equivalent for the
calibration impedance magnitude (612), multiplied by the determined
equivalent for the object impedance magnitude (632),
and wherein further in the event the real or imaginary portion of
the object impedance is further processed,
calculating means (644) for determining the in-phase portion (real
part) and/or calculating means (646) for determining the quadrature
portion (imaginary part) from the magnitude (642) and phase (640)
of the object impedance.
(FIG. 6)
[0287] Entry 30. Apparatus according to any of entries 23 to 26,
wherein:
[0288] for the demodulation when the switching means (30, 32, 34,
36; 700) connect the alternating current (AC) source (70, 40, 42)
and the first measuring means (50, 52) to the object (10) in order
to acquire Object Current Samples (702) and Object Voltage Samples
(704),
for each frequency f.sub.AC of the alternating current (AC)
applied,
a correlating means (706) correlates the Object Current Samples
(702) with the discrete values of an ideal sine waveform in order
to obtain a value (714) proportional to the in-phase portion of the
alternating current (indirect correlation),
a correlation means (708) correlates the Object Current Samples
(702) with the discrete values of an ideal cosine waveform in order
to obtain a value (716) proportional to the quadrature portion of
the alternating current,
a correlation means (710) correlates the Object Voltage Samples
(704) with the discrete values of an ideal sine waveform to obtain
a value (718) proportional to the in-phase portion of the
voltage,
a correlation means (712) correlates the Object Voltage Samples
(704) with the discrete values of an ideal cosine waveform in order
to obtain a value (720) proportional to the quadrature portion of
the voltage,
a calculation means (722) determines a current magnitude from the
values proportional to in-phase portion (714) and quadrature
portion (716) of the current applied,
a calculation means (724) determines a voltage magnitude from the
values proportional to in-phase portion (718) and quadrature
portion (720) of the voltage measured,
[0289] a calculation means (730) determines a current phase of the
values proportional to in-phase portion (714) and quadrature
portion (716) of the current applied, calculation means (732)
determines a voltage phase from the values proportional to in-phase
portion (718) and quadrature portion (720) of the voltage
measured,
a calculation means (734) determines the phase of the object
impedance as the difference between the voltage phase (730) and the
current phase (732),
a calculation means (736) determines the magnitude of the object
impedance from the ratio of voltage magnitude (724) and current
magnitude (722),
and wherein further in the event the real or imaginary portion of
the object impedance is further processed,
calculating means (738) for determining the in-phase portion (real
part) of the object impedance and/or
calculating means (740) for determining the quadrature portion of
the object impedance from the magnitude (736) and phase (734) of
the object impedance.
(FIG. 7)
[0290] Entry 31. Apparatus according to any of entries 23 to 26,
comprising:
[0291] for the demodulation when the switching means (30, 32, 34,
36; 800) connect the alternating current (AC) source (40, 42) and
the first measuring means (50, 52) to the object (10) in order to
acquire Object Current Samples (802) and object Voltage Samples
(804),
a correlating means (806) for correlating the Object Current
Samples (802) with the Object Voltage Samples (804) in order to
obtain a value (810) proportional to the in-phase portion (real
part) of the object impedance (direct correlation),
[0292] a correlating means (808) for correlating the Object Current
Samples (802) with the Object Voltage Samples (804), which are
shifted in time by -90 degrees, in order to obtain a value (812)
proportional to the quadrature portion (imaginary part) of the
object impedance,
and further comprising
a calculating means (814) for determining the magnitude of the
object impedance and/or
a calculating means (816) for determining the phase of the object
impedance, both from the in-phase portion (810) and quadrature
portion (812) of the object impedance.
(FIG. 8)
[0293] Entry 32. Apparatus according to any of entries 23 to 26,
wherein:
[0294] for the demodulation when the switching means (30, 32, 34,
36; 900) connect the alternating current (AC) source (70, 40, 42)
of constant magnitude and the first measuring means (50, 52) to the
object (10) in order to acquire Object Voltage Samples (902),
for each frequency f.sub.AC of the alternating current (AC)
applied,
a sampling means for providing discrete values of an ideal sine
waveform which represent the current in magnitude and phase,
further referred to as the Reference Current Samples (REF),
a correlation means (904) for correlating the Object Voltage
Samples (902) with the Reference Current Samples (REF), to obtain a
value (908) proportional to the in-phase portion (real part) of the
object impedance,
a correlation means (906) for correlating the Object Voltage
Samples (902) with the Reference Current Samples (REF), to obtain a
value (910) proportional to the quadrature portion (imaginary part)
of the object impedance,
and further comprising in the event the real or imaginary portion
of the object impedance is further processed,
a calculating means (912) for determining the magnitude of the
object impedance and/or
a calculating means (914) for determining the phase of the object
impedance both from the in-phase portion (908) and quadrature
portion (910) of the object impedance.
(FIG. 9)
[0295] Entry 33. Apparatus of any of entries 27 to 32, wherein
[0296] fitting means are provided to fit the samples of the
digitized current signals of the calibration impedance (20) and/or
the object (10) towards discrete values of an ideal sinusoidal
waveform, providing, over time, the Calibration and/or Object
Current Samples (402, 452; 502, 522; 702, 802),
[0297] and/or fitting means to fit the samples of the digitized
voltage signals of the calibration impedance (20) and/or the object
(10) towards values of an ideal sinusoidal waveform, providing,
over time, the Calibration and/or Object Voltage Samples (404, 454;
504, 524; 602, 622; 704; 804, 902).
[0298] Entry 34. Apparatus of any of entries 27 to 32, wherein the
complex bioadmittance is determined instead of the complex
bioimpedance.
[0299] Some examples of the invention may be summarized as follows.
A method and apparatus for digital demodulation by means of
correlation and further processing of signals obtained in the
single and multi-frequency measurement of electrical bioimpedance
or bioadmittance in which the amplitude of changes or rate of
changes thereof can be determined with a higher amplitude
resolution than before. It comprises: signal generation means which
apply an excitation signal and a first measuring means (44, 46; 50,
52) for acquiring, sampling and digitizing a response signal to
said excitation signal with respect to frequency content, amplitude
and phase, whereas said excitation signal is either held at a
constant, known amplitude and defined by a digital excitation
waveform or measured by a second measuring means for acquiring,
sampling and digitizing said excitation signal; memory means (48,
58) for temporarily storing said digitized response signal and,
optionally, said digitized excitation signal; digital demodulation
means (80; 81-86) for correlating for each frequency of the
excitation signal applied digitized samples of said digitized
response signal with corresponding discrete values of a sinusoidal
reference signal to the excitation signal or said excitation
signal, respectively; processing means (60) for calculating for
each frequency of the excitation signal applied complex values for
the bioimpedance or bioadmittance from the output values of the
digital demodulation means, a first separating means (64) for
separation the base bioimpedance or bioadmittance from said
waveforms; and either a second separating means (66) for separation
the changes in the bioimpedance or bioadmittance from said
waveforms, and a differentiating means (67) for obtaining the rate
of change of said changes, or a differentiating means (67) means
for obtaining the rate of change in the bioimpedance or
bioadmittance waveforms; as well as recording means (110) for
either recording the temporal course or recording the rate of
change in said waveforms.
[0300] This invention is related to the field of digital
demodulation and further processing of signals obtained from the
measurement of electrical bioimpedance or bioadmittance in a
biological object, for instance in a plant or a fruit thereof due
to biological activity, or in an animal or a human due to cardiac
and/or respiratory activity, for instance in cardiometry, in
particular to the monitoring through measurement of the change in
thoracic electrical bioimpedance (TEB) or bioadmittance, and
pertains to the processing of the excitation, response and/or
reference signals obtained through sensing and measuring
excitation, response and/or reference signals, e.g., but not
limited to, a voltage resulting from an alternating current (AC)
application.
[0301] A number of illustrative embodiments of the invention have
been described herein. It will be apparent to persons skilled in
the art that various changes and modifications can be made to the
described embodiments without departing from the scope of the
invention as defined by the following claims.
* * * * *