U.S. patent application number 10/536742 was filed with the patent office on 2007-02-01 for switching power supply circuit.
This patent application is currently assigned to Sony Corporation. Invention is credited to Masayuki Yasumura.
Application Number | 20070024255 10/536742 |
Document ID | / |
Family ID | 32708473 |
Filed Date | 2007-02-01 |
United States Patent
Application |
20070024255 |
Kind Code |
A1 |
Yasumura; Masayuki |
February 1, 2007 |
Switching power supply circuit
Abstract
Enhancement of the power conversion efficiency and reduction of
switching noise of a power supply circuit are achieved. The power
supply circuit includes, on the primary side, a composite resonance
type converter formed from a current resonance type converter and a
partial voltage resonance circuit in combination, and is configured
so as to produce a plurality of secondary side DC output voltages.
A particular one of the plural secondary side DC output voltages is
controlled to a constant voltage by variably controlling the
switching frequency of a primary side switching converter. Each of
the remaining secondary side DC output voltages is controlled to a
constant voltage by adjusting the level of control current to be
supplied to a controlling winding of a control transformer in
response to the level of the secondary side DC output voltage to
adjust the inductance of a controlling winding of the control
transformer inserted in a rectification current path.
Inventors: |
Yasumura; Masayuki;
(Kanagawa, JP) |
Correspondence
Address: |
LERNER, DAVID, LITTENBERG,;KRUMHOLZ & MENTLIK
600 SOUTH AVENUE WEST
WESTFIELD
NJ
07090
US
|
Assignee: |
Sony Corporation
Tokyo
JP
141-0001
|
Family ID: |
32708473 |
Appl. No.: |
10/536742 |
Filed: |
November 13, 2003 |
PCT Filed: |
November 13, 2003 |
PCT NO: |
PCT/JP03/14457 |
371 Date: |
March 22, 2006 |
Current U.S.
Class: |
323/267 ;
363/16 |
Current CPC
Class: |
H02M 3/3385 20130101;
H02M 1/009 20210501 |
Class at
Publication: |
323/267 ;
363/016 |
International
Class: |
H02M 3/335 20060101
H02M003/335; G05F 1/577 20060101 G05F001/577 |
Foreign Application Data
Date |
Code |
Application Number |
Dec 27, 2002 |
JP |
2002-381227 |
Claims
1. A switching power supply circuit, comprising: switching means
including a switching element for receiving a DC input voltage as
an input thereto to perform switching operation; switching driving
means for switching driving said switching element; a converter
transformer formed by winding thereon at least a primary winding to
which a switching output obtained by the switching operation of
said switching means is supplied and a plurality of secondary
windings in each of which an alternating voltage as the switching
output obtained in said primary winding is excited; first DC output
voltage production means for receiving the alternating voltage
obtained in one of said plural secondary windings of said converter
transformer as an input thereto to perform rectification operation
to produce a first DC output voltage; second DC output voltage
production means for receiving the alternating voltage obtained in
a different one of said plural secondary windings as an input
thereto to perform rectification operation to produce a second DC
output voltage, said second DC output voltage production means
supplying a load thereto with power lower than that supplied by
said first DC output voltage production means; frequency controlled
type constant voltage controlling means for controlling said
switching driving means in response to the level of a first
secondary side DC output voltage produced by said first DC output
voltage production means to adjust the switching frequency of said
switching means to perform constant voltage control for the first
DC output voltage; and inductance controlled type constant voltage
controlling means including a control transformer as a saturable
reactor having a controlling winding and a controlled winding wound
thereon, said controlled winding being inserted between the
different secondary winding and said secondary DC output voltage
production means, said inductance controlled type constant voltage
controlling means adjusting the level of control current to be
supplied to said controlling winding in response to the level of
the second DC output voltage from said second DC output voltage
production means to adjust the inductance of said controlled
winding to perform constant voltage control for the second DC
output voltage.
2. The switching power supply circuit according to claim 1, wherein
said converter transformer is a transformer which includes a core
having a central magnetic leg and has a gap provided on said
central magnetic leg and wherein said primary winding and said
plural secondary windings are wound on said central magnetic leg
such that said transformer has a coupling coefficient which
represents rough coupling.
3. The switching power supply circuit according to claim 2, wherein
said switching means is a switching means including two switching
elements connected between the DC input voltage and a reference
potential and connected in cascade connection to each other through
a nodal point, and said switching power supply circuit further
comprises a series resonance circuit formed from a leakage
inductance component of said primary winding of said converter
transformer and a capacitance of a primary side series resonance
capacitor connected in series to said primary winding and connected
between the nodal point of the two switching elements and the
reference potential, said series resonance circuit causing the
switching output obtained by said switching means to resonate.
4. The switching power supply circuit according to claim 3, further
comprising a partial resonance circuit including a parallel
resonance capacitor connected in parallel to one of said two
switching elements which is connected to the reference potential
for performing partial resonance operation together with the
leakage inductance component of said primary winding of said
converter transformer.
5. The switching power supply circuit according to claim 3, wherein
each of said plural secondary windings is divided into two
divisional winding portions by a center tap connected to the
reference potential such that alternating voltages whose periods of
time within which the alternating voltages have the positive
polarity or the negative polarity are different from each other are
induced in the divisional winding portions, and each of said first
and second DC output voltage production means is a full wave
rectification and smoothing circuit which includes two rectifiers
individually connected at one end portions thereof to the two
divisional winding portions divided by the center tap and connected
at the other end thereof commonly for rectifying the induced
alternating voltages and a smoothing capacitor connected between
the other ends of the two rectifiers and the reference
potential.
6. The switching power supply circuit according to claim 5, further
comprising a resistor connected between the two divided winding
portions of each of said plural secondary windings.
7. The switching power supply circuit according to claim 1, wherein
said inductance controlled type constant voltage controlling means
includes: a shunt regulator for producing an output of a level
corresponding to an error of the level of the second DC output
voltage; and an amplification circuit connected, while the first DC
output voltage is supplied to one end of said controlling winding
as power supply, to the other end of said controlling winding for
amplifying and outputting the output of said shunt regulator as
control current to said controlling winding.
8. The switching power supply circuit according to claim 7, wherein
said inductance controlled type constant voltage controlling means
stops the supply of power to said amplification circuit in response
to occurrence of load short-circuiting with regard to the second DC
output voltage, and said switching power supply circuit further
comprises a switch circuit including a required number of
transistor elements.
9. The switching power supply circuit according to claim 5, wherein
said control transformer is an orthogonal controlled type
transformer which includes a pair of controlled windings whose
inductance is adjusted in response to adjustment of the level of
the control current to be supplied to said controlling winding and
on which said controlling winding is disposed so as to be
orthogonal to the pair of controlled windings.
10. The switching power supply circuit according to claim 5,
further comprising: a full wave rectification and smoothing circuit
provided in parallel to said first DC output voltage production
means and including two rectifiers individually connected at one
end portions thereof to the two divisional winding portions for
rectifying the induced alternating voltages and a smoothing
capacitor connected between the other ends of the two rectifiers
connected commonly and the reference potential for supplying a
third DC output voltage; a control transformer including a pair of
controlled windings and a pair of controlling windings individually
connected in series between the two divisional winding portions and
the two rectifiers; and second inductance controlled type constant
voltage control means for adjusting the level of the control
current to be supplied to said controlling winding in response to
the level of the third DC output voltage from said full wave
rectification and smoothing circuit to adjust the inductance of
said controlled winding to perform constant voltage control for the
third DC output voltage.
Description
TECHNICAL FIELD
[0001] This invention relates to a switching power supply circuit
provided as a power supply in various electronic apparatus.
BACKGROUND ART
[0002] Various switching power supply circuits are widely known
including, for example, a switching power supply circuit of the
flyback converter type or the forward converter type. The switching
converters of the types mentioned are restricted in suppression of
switching noise because the switching operation waveform is a
rectangular waveform. Further, it is known that the switching
converters are limited in enhancement in the power conversion
efficiency from their operation characteristics.
[0003] Therefore, various switching power supply circuits which
rely upon various resonance type converters have been proposed by
the applicant of the present application. According to the
resonance type converters, a high power conversion efficiency can
be obtained readily, and low noise is achieved because the
switching operation waveform is a sine waveform. Further, the
resonance type converters have a merit also that they can be formed
from a comparatively small number of parts.
[0004] One of the power supply circuits having been applied for
patent formerly by the applicant of the present invention is
configured such that it includes a voltage resonance type converter
as a primary side switching converter and produces and outputs a
plurality of secondary side DC output voltages on the secondary
side.
[0005] Further, for the stabilization of the secondary side DC
output voltages, a switching frequency control method which depends
upon control of the switching frequency of the primary side
switching converter is adopted, for example, with regard to a main
one of the primary side DC output voltages. Meanwhile, with regard
to another required one of the secondary side DC output voltages, a
controlled winding of an orthogonal type control transformer
(saturable reactor) is inserted in series to a rectification
current path for producing the secondary side DC output voltage.
Further, the level of control current to be supplied to a
controlling winding of the orthogonal type control transformer is
varied in response to the level of the secondary side DC output
voltage to vary the inductance of the controlled winding thereby to
control the level of current to be supplied to the rectification
current path to make the secondary side DC output voltage constant
(refer to the official gazette of Japanese Patent Laid-Open No.
2000-064981).
[0006] A power supply circuit as a prior art which is formed based
on the power supply circuit described above is shown in FIG. 14.
Also the power supply circuit shown in FIG. 14 is configured such
that it includes a resonance type converter as a primary side
switching converter and a plurality of secondary side DC output
voltages are produced on the secondary side. However, while the
primary side switching converter in the power supply circuit
described above is a voltage resonance type converter, that of the
power supply circuit shown in FIG. 14 is a current resonance type
converter. For example, while the rectification circuit system for
producing secondary side DC output voltages adopts a configuration
as a half-rectification circuit where the primary side switching
converter is a voltage resonance type converter, where the primary
side switching converter is a current resonance type converter, it
is possible to form a full-wave rectification circuit as the
rectification circuit system. This can increase the current
capacity as the switching power supply circuit.
[0007] In the power supply circuit shown in FIG. 14, a common mode
noise filter formed from one common mode choke coil CMC and two
across capacitors CL is connected to a commercial AC power supply
AC. The common mode noise filter suppresses noise to be
transmitted, for example, from the switching converter side to the
commercial AC power supply AC.
[0008] Further, a full-wave rectification circuit formed from a
bridge rectification circuit Di and a smoothing capacitor Ci is
provided in the line of the commercial AC power supply AC at the
next stage to the common mode noise filter. A rectification
smoothed voltage Ei having a level equal to that of an AC input
voltage VAC is obtained as a voltage across the smoothing capacitor
Ci by rectification smoothing operation by the full wave
rectification circuit.
[0009] A switching converter which receives and operates with the
rectification smoothed voltage Ei as a DC input voltage in this
instance has a configuration as a composite resonance type
converter which includes at least a partial voltage resonance
circuit on the primary side in a basic configuration as a current
resonance type converter.
[0010] Here, two switching devices Q1 (high side) and Q2 (low side)
each formed from a MOS-FET are connected to each other in a half
bridge coupling scheme as seen in FIG. 14. Damper diodes DD1 and
DD2 are individually connected in parallel to each other and in
such directions as seen in FIG. 14 between the drain-source of the
switching devices Q1 and Q2, respectively.
[0011] Further, a partial resonance capacitor Cp is connected in
parallel between the drain-source of the switching device Q2. A
parallel resonance circuit (partial voltage resonance circuit) is
formed from the capacitance of the partial resonance capacitor Cp
and the leakage inductance L1 of a primary winding N1. The partial
voltage resonance circuit performs partial voltage resonance
operation wherein it voltage resonates only upon turning off of the
switching devices Q1 and Q2.
[0012] A control IC 2 includes an oscillation circuit for driving
the current resonance type converter in a separately excited
fashion, a control circuit, a protection circuit and so forth and
is formed as an analog IC (Integrated Circuit) for universal use
including a bipolar transistor in the inside thereof.
[0013] The control IC 2 operates with a DC voltage inputted to a
power supply input terminal Vcc. In this instance, the power supply
input terminal Vcc is connected to the line of the rectification
smoothed voltage Ei through a starting resistor Rs. The control IC2
is started by and operates with the rectification smoothed voltage
Ei inputted thereto through the starting resistor Rs. Further, the
control IC 2 is grounded to the primary side ground through a
ground terminal E.
[0014] Further, the control IC 2 includes two drive signal output
terminals VGH and VGL as terminals for outputting a drive signal
(gate signal) to the switching elements.
[0015] A drive signal for switching driving the high side switching
element is outputted from the drive signal output terminal VGH, and
another drive signal for switching driving the low side switching
element is outputted from the drive signal output terminal VGL.
[0016] The high side drive signal outputted from the drive signal
output terminal VGH is applied to the gate of the switching element
Q1. Meanwhile, the low side drive signal outputted from the drive
signal output terminal VGL is applied to the gate of the switching
element Q2.
[0017] The switching elements Q1 and Q2 perform switching operation
such that they are alternately switched on/off with a required
switching frequency by the drive signals outputted from the drive
signal output terminals VGH and VGL, respectively.
[0018] An insulating converter transformer PIT transmits switching
outputs of the switching elements Q1 and Q2 to the secondary side
and in this instance has a primary winding N1 and two secondary
windings N2 and N2A wound thereon.
[0019] In this instance, the primary winding N1 of the insulating
converter transformer PIT is connected at one end thereof to a node
(switching output point) between the source of the switching
element Q1 and the drain of the switching element Q2 through a
series connection of a series resonance capacitor C1. The primary
winding N1 is connected at the other end thereof to the primary
side ground.
[0020] According to the connection scheme described above, a series
circuit of the series resonance capacitor C1 and the primary
winding N1 is connected to the switching output point of the
switching elements Q1 and Q2. Consequently, a primary side series
resonance circuit is formed from the capacitance of the series
resonance capacitor C1 and a leakage inductance L1 of the
insulating converter transformer PIT including the primary winding
N1. Since the primary side series resonance circuit is connected to
the switching output point in such a manner as described above, the
switching output of the switching elements Q1 and Q2 is transmitted
to the primary side series resonance circuit. The primary side
series resonance circuit performs resonance operation in response
to the switching output transmitted thereto. Consequently,
operation of the primary side switching converter becomes that of
the current resonance type.
[0021] Accordingly, operation of the current resonance type by the
primary side series resonance circuit (C1-L1) and partial resonance
operation by the partial voltage resonance circuit (Cp//L1)
described hereinabove are obtained by the primary side switching
converter shown in FIG. 14.
[0022] In other words, the power supply circuit shown in FIG. 14
adopts a form which includes a combination of a resonance circuit
for making the primary side switching converter that of the
resonance type with another resonance circuit. In short, the power
supply circuit adopts a configuration as a composite resonance type
converter
[0023] An alternating voltage is induced in each of the secondary
windings N2 and N2A wound on the secondary side of the insulating
converter transformer PIT in response to the switching output
transmitted to the primary winding N1.
[0024] The secondary winding N2 has a center tap provided thereon
as shown in FIG. 14 and connected to the secondary side ground, and
a full-wave rectification circuit formed from rectification diodes
Do1 and Do2 and a smoothing capacitor Co is connected to the
secondary winding N2. Consequently, a secondary side DC output
voltage Eo is obtained as a voltage across the smoothing capacitor
Co. The secondary side DC output voltage Eo is supplied to the load
side not shown and is branched and inputted also as a detection
voltage for a control circuit 1 described below.
[0025] The control circuit 1 produces a voltage or current whose
level is adjusted in response to the level of the secondary side DC
output voltage Eo described hereinabove as a control output
thereto. The control output is outputted to a control terminal Vc
of the control IC 2.
[0026] The control IC 2 operates to adjust the frequency of the
drive signals in response to the control output level inputted to
the control terminal Vc to adjust the frequency of the drive signal
for the high side and the drive signal for the low side to be
outputted from the drive signal output terminals VGH and VGL while
the drive signals maintain the timings at which they are turned
on/off alternately.
[0027] Consequently, the switching frequency of the switching
elements Q1 and Q2 is variably controlled in response to the
control output level (that is, the secondary side DC output voltage
level) inputted to the control terminal Vc.
[0028] When the switching frequency varies, the resonance impedance
of the primary side series resonance circuit varies. When the
resonance impedance varies, the amount of current to be supplied to
the primary winding N1 of the primary side series resonance circuit
varies and also the power to be transmitted to the secondary side
varies. Consequently, the level of the secondary side DC output
voltage Eo varies, and constant voltage control for the secondary
side DC output voltage Eo is implemented.
[0029] In this instance, a step-down type converter formed from a
switching element Q3 formed from a MOS-FE, a rectification diode
D3, a choke coil L10, and a smoothing capacitor Co3 in such a
manner as seen in FIG. 14 is connected to the secondary side output
voltage Eo.
[0030] The step-down type converter produces a secondary side DC
output voltage Eo2 stepped down from the secondary side DC output
voltage Eo by receiving the secondary side DC output voltage Eo as
an input thereto and half-wave rectifying an alternating voltage
obtained by switching performed by the switching element Q3 by
means of the rectification diode D3 and the choke coil L10 to
charge the smoothing capacitor Co3.
[0031] The constant voltage control of the secondary side DC output
voltage Eo2 is performed by a control circuit 3.
[0032] The control circuit 3 receives the secondary side DC output
voltage Eo2 as an input thereto and, for example, varies the pulse
width within one cycle of a drive signal to be outputted to the
gate of the switching element Q3 in response to the level of the
secondary side DC output voltage Eo2 inputted thereto while
controlling the switching frequency constant. In other words, the
control circuit 3 preforms PWM control. Consequently, the on-angle
of the switching element Q3 within one switching cycle is varied,
and as a result, also the level of the secondary side DC output
voltage Eo2 varies. By variably controlling the secondary side DC
output voltage Eo2 in this manner, stabilization of the secondary
side DC output voltage Eo2 is achieved.
[0033] Further, a center tap is provided also for the secondary
winding N2A and connected to the secondary side ground, and
besides, a full-wave rectification circuit is formed from
rectification diodes Do3 and Do4 and a smoothing capacitor Co1 in
such a manner as seen in FIG. 15. A DC voltage is obtained across
the smoothing capacitor Co1.
[0034] In this instance, a three-terminal regulator 4 is connected
to the DC voltage of an output of the smoothing capacitor Co1 so
that a stabilized secondary side DC output voltage Eo1 is obtained
as a voltage across the smoothing capacitor Co2.
[0035] Here, the load conditions with regard to the secondary side
DC output voltages Eo, Eo1 and Eo2 obtained on the secondary side
in such a manner as described above are such as given below:
[0036] Eo: 5.0 V/6 A to 2 A
[0037] Eo1: 12.0 V/1 A to 0.2 A
[0038] Eo2: 3.3 V/6A to 2 A
[0039] According to the load conditions given above, the highest
load power is applied to the secondary side DC output voltage Eo.
Therefore, the constant voltage control of the secondary side DC
output voltage Eo is performed using switching frequency control
which has the highest controlling capability and provides
comparatively low power loss.
[0040] The second highest load power next to the secondary side DC
output voltage Eo is applied to the secondary side DC output
voltage Eo2. Since the load current amount is considerably great
also with regard to the secondary side DC output voltage Eo2, in
this instance, a step-down type converter is provided as means
other than the switching frequency control to achieve a constant
voltage.
[0041] The remaining secondary side DC output voltage Eo1 is
stabilized by simple and easy means by the three-terminal regulator
4 since the load current amount is small.
[0042] However, the power supply circuit described hereinabove with
reference to FIG. 14 has the following problems.
[0043] While the DC/DC power conversion efficiency (.eta. DC/DC) of
the power supply circuit shown in FIG. 14 is 94% with regard to the
secondary side DC output voltage Eo, it is 80% with regard to the
secondary side DC output voltage Eo1 and 92% with regard to the
secondary side DC output voltage Eo2, and is totally approximately
88%.
[0044] In particular, while the circuit shown in FIG. 14 adopts a
configuration wherein a series regulator such as the three-terminal
regulator 4 and a step-down type converter are added in order to
individually stabilize a plurality of secondary side DC output
voltages, the series regulator and the step-down type converter
inevitably exhibit high power loss. Therefore, where the load power
variation is great as a condition of the load side, the power loss
further increases, and consequently, also it becomes necessary to
provide a heat radiating plate for the series regulator and/or the
step-down type converter, which gives rise to, for example,
expansion of the circuit scale and/or increase of the cost.
[0045] Further, in the power supply circuit shown in FIG. 14, while
the switching frequency of the primary side composite resonance
type converter is 75 KHz to 100 KHz, the switching frequency of the
switching element Q3 in the step-down type converter on the
secondary side is fixed, for example, at 100 KHz. Where a plurality
of switching frequencies are involved in one power supply circuit
in this manner, the switching frequencies interfere with each other
and also the level of noise generation becomes higher. Therefore,
such a countermeasure against noise such as various types of noise
filters or shield plates is required, and also in this regard,
expansion of the circuit scale and increase of the cost are
invited.
[0046] Therefore, it is known to adopt a magnetic amplifier as
means for stabilizing the secondary side outputs in place of such a
series regulator and a step-down type converter as described
above.
[0047] FIG. 15 shows an example of a configuration where the
secondary side of the power supply circuit shown in FIG. 14 adopts
such a magnetic amplifier as described above. It is to be noted
that like elements those of FIG. 14 are denoted by like reference
characters and description of them is omitted herein.
[0048] Referring to FIG. 15, a circuit system for producing the
stabilized secondary side DC output voltage Eo1 is configured in
the following manner.
[0049] First, the secondary winding N2A has a center tap provided
thereon and connected to the secondary side ground, and the
rectification diodes Do3 and Do4 and the smoothing capacitor Co1
are connected to the secondary winding N2A in such a manner as seen
in FIG. 15 to form a full-wave rectification circuit. Thus, the
secondary side DC output voltage Eo1 is produced as a voltage
across the smoothing capacitor Co1.
[0050] In addition, the full-wave rectification circuit of the
secondary winding N2A includes a constant voltage circuit (magnetic
amplifier constant voltage circuit) which includes a magnetic
amplifier, and the secondary side DC output voltage Eo1 is
stabilized by the magnetic amplifier constant voltage circuit.
[0051] The magnetic amplifier constant voltage circuit includes a
saturable inductor (choke coil) SR1 interposed between an end of
the secondary winding N2A and the anode of the rectification diode
Do3 and another saturable inductor SR2 interposed between the other
end of the secondary winding N2A and the anode of the rectification
diode Do4. Further, the cathode of a reset voltage adjusting diode
DV1 is connected to the anode of the rectification diode Do3, and
the cathode of another reset voltage adjusting diode DV2 is
connected to the anode of the rectification diode Do4. The anodes
of the diodes DV1 and DV2 are connected to the collector of the
transistor Q4. The emitter of the transistor Q4 is connected to the
positive line for the secondary side DC output voltage Eo1 through
a resistor Rc.
[0052] The control circuit 3 in this instance controls the magnetic
fluxes of the saturable inductors SR1 and SR2 in order to stabilize
the secondary side DC output voltage Eo1.
[0053] The control circuit 3 is formed as an error amplifier
including shunt regulator and so forth and variably controls the
base current level of the transistor Q4 in response to the level of
the secondary side DC output voltage Eo1 inputted thereto. The
collector current level of the transistor Q4 is adjusted by the
variable control of the base current level of the transistor Q4.
Since the collector of the transistor Q4 is connected to a node
between the anodes of the reset voltage adjusting diodes DV1 and
DV2, when the collector current level is adjusted, the control
voltage for adjusting the reset voltage for the magnetic fluxes of
the saturable inductors SR1 and SR2 is adjusted.
[0054] Here, the saturable inductor SR (SR1, SR2) is formed by
winding a winding Ln of a solid wire by a required number of turns
on a circular toroidal core CR, for example, in such a manner as
seen in FIG. 16.
[0055] FIG. 17 illustrates a B-H diagram where a cobalt type
amorphous material is selected as a material of the core of the
saturable inductor SR formed in such a manner as described above.
The B-H characteristic of the saturable inductor SR exhibits a
hysteresis characteristic having a high rectangular ratio as can be
seen from the figure.
[0056] The magnetic amplifier including such a saturable inductor
SR as described above operates in such a manner as seen in FIG. 18.
Referring to FIG. 18, the voltage V3 represents a potential between
the node between the saturable inductor SR1 and the secondary
winding N2A and the center tap of the secondary winding N2A.
Meanwhile, the voltage VL1 indicates a voltage across the saturable
inductor SR1. The current ID1 represents rectification current
flowing to the rectification diode Do3.
[0057] Within a period from t0 to t1, the voltage V3 exhibits a
state of the positive polarity, and at this time, the saturable
inductor SR1 is in an unsaturated state (B0>B>B1). At this
time, since the relationship between the voltages V3 and VL1 is
V3.apprxeq.VL1, the current ID1 does not flow to the rectification
diode Do3.
[0058] Within another period from t1 to t2, since the saturable
inductor SR1 exhibits a saturated state (B=B1), the voltage VL1 has
the substantially 0 level. Consequently, since the relationship
between the voltages V3 and VL1 is V3>VL1, the current ID1
begins to flow to the rectification diode Do3.
[0059] Then within a further period from t2 to t3, an output
voltage adjustment circuit 11 shown equivalently in FIG. 19
operates. This output voltage adjustment circuit 11 is, in FIG. 15,
a control circuit 3 to which the secondary side DC output voltage
Eo2 is inputted. As can be seen also from FIG. 19, the control
circuit 3 adopts a configuration as an error amplifier. In short,
the control circuit 3 compares the level of the secondary side DC
output voltage Eo2 divided by the voltage dividing resistors Ro1
and Ro2 with a reference voltage Vref and amplifies an error
between them by means of an amplifier formed from an operational
amplifier OP and a feedback circuit (Ca, Ra), and then outputs the
amplified error through a resistor Rb.
[0060] Then, a reset circuit 10 supplies reset current to the
saturable inductor SR1 in response to the output from the output
voltage adjustment circuit 11 obtained in such a manner as
described above. The reset circuit 10 equivalently indicates a
function as the reset circuit formed from the resistor Rc,
transistor Q4, diodes DV1 and DV2 and saturable inductors SR1 and
SR2 shown in FIG. 15.
[0061] Supplying operation of reset current by the reset circuit 10
at this time is performed by supplying current of a level
corresponding to the output level from the output voltage
adjustment circuit 11 to the saturable inductor SR1 through the
resistor Rc.fwdarw.transistor Q4.fwdarw.diode DV1. By the reset
current, resetting of the saturable inductor SR1 is performed so
that the magnetic flux density may be returned to B0.
[0062] The time length of the period from t0 to t1 in which the
saturable inductor SR1 has an unsaturated state is determined by
the reset amount (reset current level) within the period from t2 to
t3.
[0063] Therefore, the reset amount is increased in response to a
rise of the level of the secondary side DC output voltage Eo1 as
the tendency to a lighter load increases. Consequently, since the
remaining magnetic flux density B0 becomes BOA as seen in FIG. 17,
also the period from t0 to t1 which is a period of the unsaturated
state can be increased so as to become a period from t0A to t1A as
seen in FIG. 18. As the period of the unsaturated state increases
in this manner, also the period within which the current ID1 does
not flow increases, and consequently, also the power supply period
to the load per unit time decreases and also the level of the
secondary side DC output voltage Eo1 drops as much.
[0064] Then, such operation as described above is performed also on
the saturable inductor SR2 side but at a timing at which the
waveform illustrated in FIG. 18 exhibits a phase difference of
180.degree..
[0065] In this manner, the circuit shown in FIG. 15 achieves
stabilization of the secondary side DC output voltage Eo1 obtained
by full-wave rectification.
[0066] Further, in FIG. 15, also for the secondary side DC output
voltage Eo2 produced on the secondary winding N2 side, a
configuration wherein constant voltage control is performed by a
magnetic amplifier constant voltage circuit is adopted similarly
for the secondary side DC output voltage Eo1 described above.
[0067] In short, as a basic configuration for obtaining the
secondary side DC output voltage Eo2, a full-wave rectification
circuit formed from rectification diodes Do5 and Do6 and a
smoothing capacitor Co1 is connected to the secondary winding
N2.
[0068] In addition, saturable inductors (choke coils) SR3 and SR4,
the diodes DV1 and DV2 for reset voltage adjustment, the transistor
Q3 for reset current outputting, the resistor Rc and a control
circuit 3 are connected to the full-wave rectification circuit in
such a manner as seen in FIG. 15 to form a magnetic amplifier
constant voltage circuit.
[0069] Where such a magnetic amplifier constant voltage circuit as
described hereinabove with reference to FIG. 15 is adopted, the
constant voltage control by the magnetic amplifier constant voltage
circuit is of the type wherein the periods within which the
saturable inductor SR exhibits saturated/unsaturated states. This
operation is performed in accordance with cyclic timings of the
voltage V3 obtained in the secondary winding as can be recognized
from the foregoing description. In other words, operations of the
saturable inductor SR, reset voltage adjusting diodes DV1 and DV2,
and reset current outputting transistors Q3 and Q4, which form the
magnetic amplifier constant voltage circuit are held in synchronism
with the switching frequency of the primary side switching
converter. From this, the problem of increase of the generation
amount of noise by interference between different switching
frequencies as in the case of, for example, the power supply
circuit shown in FIG. 14 is eliminated.
[0070] However, also in the circuit shown in FIG. 15, the power
loss by the toroidal core CR which forms the saturable inductor SR
and the power loss by semiconductor elements such as the reset
voltage adjusting diodes DV1 and DV2 and the reset current
outputting transistors Q3 and Q4 which form the magnetic amplifier
constant voltage circuit are high. Consequently, the problem that
the total power conversion efficiency of the power supply circuit
drops remains. For example, the total power conversion efficiency
(.eta. DC/DC) where, for example, the circuit shown in FIG. 15 is
used is approximately 86% and is lower than that by the circuit
configuration shown in FIG. 14.
[0071] Further, in order to form the magnetic amplifier constant
voltage circuit, a toroidal core as the saturable inductor SR and
semiconductor elements such as diode elements for reset voltage
adjustment and transistors for reset current outputting are
required. For example, in an actual case, a Schottky diode is
selectively used for the diode elements for reset voltage
adjustment. Meanwhile, a transistor for 50 V/2 A is selectively
used for the transistors for the reset current outputting. Since
the semiconductor elements mentioned are comparatively expensive,
the circuit shown in FIG. 15 is still disadvantageous in terms of
the cost.
[0072] This problem is significant particularly where the
configuration of the power supply circuit shown in FIG. 14 is used
as a basic configuration. In particular, as described hereinabove,
where a current resonance type configuration is used as the basic
configuration of the primary side switching converter, the
rectification circuit system for producing a secondary side DC
output voltage can be formed as a full-wave rectification circuit
thereby to achieve a configuration by which a greater current
capacity can be obtained. However, if it is tried to add a magnetic
amplifier constant voltage circuit to a full-wave rectification
circuit, then two sets each including a saturable inductor SR and a
diode element for reset voltage adjustment are required
corresponding to positive/negative rectification current paths.
[0073] In this manner, where constant voltage control is performed
individually for a plurality of secondary side DC output voltages
produced on the secondary side by a switching power circuit which
includes, for example, a current resonance type converter on the
primary side, the problems of drop of the power conversion
efficiency, increase of the cost and so forth caused by addition of
the circuit elements for the constant voltage control are
involved.
DISCLOSURE OF INVENTION
[0074] Therefore, according to the present invention, a switching
power supply circuit is configured in the following manner taking
the subject described above into consideration.
[0075] In particular, the switching power supply circuit includes
switching means having a switching element for receiving a DC input
voltage as an input thereto to perform switching operation, and
switching driving means for switching driving the switching
element.
[0076] The switching power supply circuit further includes an
insulating converter transformer formed by winding thereon at least
a primary winding to which a switching output obtained by the
switching operation of the switching means is supplied and a
plurality of secondary windings in each of which an alternating
voltage as the switching output obtained in the primary winding is
excited.
[0077] The switching power supply circuit further includes
plurality of DC output voltage production means for receiving the
alternating voltages obtained in the plural secondary windings of
the insulating converter transformer as inputs thereto to perform
rectification operation to produce secondary side DC output
voltages, and frequency controlled type constant voltage
controlling means for controlling the switching driving means in
response to the level of the secondary side DC output voltage which
supplies comparatively high power to a load from among the plural
secondary side DC output voltages produced by the plural DC output
voltage production means to adjust the switching frequency of the
switching means to perform constant voltage control for the
secondary side DC output voltage.
[0078] The switching power supply circuit further includes
inductance controlled type constant voltage controlling means
provided corresponding to each of those of the plural secondary
side DC output voltages other than the secondary side DC output
voltage controlled to a constant voltage by the frequency
controlled type constant voltage control means and including a
control transformer as a saturable reactor having a controlling
winding and a controlled winding wound thereon, the controlled
winding being inserted in a secondary side rectification current
path for producing the secondary side DC output voltage of a
control object, the inductance controlled type constant voltage
controlling means adjusting the level of control current to be
supplied to the controlling winding in response to the level of the
secondary side DC output voltage of the control object to adjust
the inductance of the controlled winding to perform constant
voltage control for the secondary side DC output voltage of the
control object.
[0079] According to the configuration described above, the
secondary side of the switching power supply circuit of the present
invention produces a plurality of secondary side DC output
voltages.
[0080] Further, that one of the plural secondary side DC output
voltages which supplies comparatively high power to a load is
controlled to a constant voltage by variably controlling the
switching frequency of the primary side switching converter in
response to the level of the secondary side DC output voltage.
[0081] Meanwhile, each of the remaining secondary side DC output
voltages is controlled to a constant voltage by means of the
inductance controlled type constant voltage control means. In
particular, the inductance controlled type constant voltage control
means includes a control transformer as a saturable reactor which
includes a controlling winding and a controlled winding, and the
controlled winding is inserted in a rectification current path for
producing the secondary side DC output voltage of a control object
of the control transformer. Then, the level of control current to
be supplied to the controlling winding is adjusted in response the
level of the secondary side DC output voltage of the control object
to vary the inductance of the controlled winding thereby to
stabilize the secondary side DC output voltage of the control
object.
[0082] With the inductance controlled type constant voltage control
means having such a configuration as described above, for example,
the power loss by the controlling winding is low, and also the
required control power for supplying the control current to the
controlling winding is very low.
BRIEF DESCRIPTION OF DRAWINGS
[0083] FIG. 1 is a circuit diagram showing an example of a
configuration of a power supply circuit as a first embodiment of
the present invention;
[0084] FIG. 2 is a circuit diagram showing an example of a
structure of an orthogonal type control transformer;
[0085] FIG. 3 is a characteristic diagram illustrating an
inductance characteristic of a controlled winding;
[0086] FIG. 4 is a circuit diagram showing an example of a
configuration of a control circuit on the secondary side
corresponding to the first embodiment;
[0087] FIG. 5 is a waveform diagram illustrating constant voltage
control operation by the control circuit on the secondary side in
the power supply circuit of the first embodiment;
[0088] FIG. 6 is a circuit diagram illustrating a relationship
between a modification to the secondary side rectification circuit
and the insertion position of the controlled winding in the present
embodiment;
[0089] FIG. 7 is a circuit diagram illustrating a relationship
between another modification to the secondary side rectification
circuit and the insertion position of the controlled winding in the
present embodiment;
[0090] FIG. 8 is a circuit diagram illustrating a relationship
between a further modification to the secondary side rectification
circuit and the insertion position of the controlled winding in the
present embodiment;
[0091] FIG. 9 is a circuit diagram illustrating a relationship
between a still further modification to the secondary side
rectification circuit and the insertion position of the controlled
winding in the present embodiment;
[0092] FIG. 10 is a circuit diagram showing an example of a
configuration of a power supply circuit as a second embodiment;
[0093] FIG. 11 is a circuit diagram showing an example of a
configuration of a control circuit on the secondary side as the
second embodiment;
[0094] FIG. 12 is a waveform diagram illustrating load
short-circuiting protection operation by the control circuit on the
secondary side in the second embodiment;
[0095] FIG. 13 is a waveform diagram illustrating operation of a
secondary side rectification circuit where a resistor for parasitic
oscillation noise removal is inserted in the second embodiment;
[0096] FIG. 14 is a circuit diagram showing an example of a
configuration of a power supply circuit as a prior art;
[0097] FIG. 15 is a circuit diagram showing another example of the
configuration on the secondary side of the power supply circuit as
the prior art;
[0098] FIG. 16 is a view showing an example of a structure of a
saturable inductor;
[0099] FIG. 17 is a characteristic diagram illustrating a B-H
characteristic of the saturable inductor;
[0100] FIG. 18 is a waveform diagram illustrating constant voltage
control operation by a magnetic amplifier which includes the
saturable inductor; and
[0101] FIG. 19 is a circuit diagram equivalently showing a magnetic
amplifier constant voltage circuit shown in FIG. 15.
BEST MODE FOR CARRYING OUT THE INVENTION
[0102] FIG. 1 shows an example of a configuration of a switching
power supply circuit as a first embodiment of the present
invention.
[0103] In the power supply circuit shown in FIG. 1, a common mode
noise filter formed from one common mode choke coil CMC and two
across capacitors CL is connected to a commercial AC power supply
AC. The common mode noise filter suppresses noise to be
transmitted, for example, from the switching converter side to the
commercial AC power supply AC.
[0104] Further, a full-wave rectification circuit formed from a
bridge rectification circuit Di and a smoothing capacitor Ci is
provided in the line of the commercial AC power supply AC at the
next stage to the common mode noise filter. A rectification
smoothed voltage Ei having a level equal to that of an AC input
voltage VAC is obtained as a voltage across the smoothing capacitor
Ci by rectification smoothing operation by the full wave
rectification circuit.
[0105] A primary side switching converter which receives and
operates with the rectification smoothed voltage Ei as a DC input
voltage in this instance has a configuration as a composite
resonance type converter which includes at least a partial voltage
resonance circuit on the primary side in a basic configuration as a
current resonance type converter.
[0106] Here, two switching devices Q1 (high side) and Q2 (low side)
each formed from a MOS-FET are connected to each other in a half
bridge coupling scheme as seen in FIG. 1. Damper diodes DD1 and DD2
are individually connected in parallel to each other and in such
directions as seen in FIG. 1 between the drain-source of the
switching devices Q1 and Q2, respectively.
[0107] Further, a partial resonance capacitor Cp is connected in
parallel between the drain-source of the switching device Q2. A
parallel resonance circuit (partial voltage resonance circuit) is
formed from the capacitance of the partial resonance capacitor Cp
and the leakage inductance L1 of a primary winding N1. The partial
voltage resonance circuit performs partial voltage resonance
operation wherein it voltage resonates only upon turning off of the
switching devices Q1 and Q2.
[0108] A control IC 2 includes an oscillation circuit for driving
the current resonance type converter in a separately excited
fashion, a control circuit, a protection circuit and so forth and
is formed as an analog IC (Integrated Circuit) for universal use
including a bipolar transistor in the inside thereof.
[0109] The control IC 2 operates with a DC voltage inputted to a
power supply input terminal Vcc. In this instance, the power supply
input terminal Vcc is connected to the line of the rectification
smoothed voltage Ei through a starting resistor Rs. The control IC2
is started by and operates with the rectification smoothed voltage
Ei inputted thereto through the starting resistor Rs. Further, the
control IC 2 is grounded to the primary side ground through a
ground terminal E.
[0110] Further, the control IC 2 includes two drive signal output
terminals VGH and VGL as terminals for outputting a drive signal
(gate signal) to the switching elements.
[0111] A drive signal for switching driving the high side switching
element is outputted from the drive signal output terminal VGH, and
another drive signal for switching driving the low side switching
element is outputted from the drive signal output terminal VGL.
[0112] The high side drive signal outputted from the drive signal
output terminal VGH is applied to the gate of the switching element
Q1. Meanwhile, the low side drive signal outputted from the drive
signal output terminal VGL is applied to the gate of the switching
element Q2.
[0113] It is to be noted that, though not shown in FIG. 1, a boot
strap circuit formed from a peripheral externally provided part is
actually connected to the control IC 2. The boot strap circuit is
for shifting the level of the drive signal to be applied to the
switching element Q1 for the high side so that it has a level with
which it can appropriately drive the switching element Q1.
[0114] Further, although also such part elements as a gate resistor
and a gate-source resistor are actually connected to the switching
elements Q1 and Q2, they are omitted in FIG. 1.
[0115] In the control IC 2, an oscillation signal of a required
frequency is produced by an oscillation circuit therein. It is to
be noted that the oscillation circuit adjusts the frequency of the
oscillation signal in response to the level of a control output
inputted from a control circuit 1 to a terminal Vc as hereinafter
described.
[0116] Thus, the control IC 2 makes use of the oscillation signal
produced by the oscillation circuit to produce a drive signal for
the high side and another drive signal for the low side. The drive
signal for the high side is outputted from a drive signal output
terminal VGH, and the signal for the low side is outputted from
another drive signal output terminal VGL.
[0117] The drive signal for the high side and the drive signal for
the low side have a waveform wherein an on period within which a
pulse of a rectangular wave of the positive polarity is generated
and an off period within which 0 V is generated are obtained within
one switching period. Further, while the drive signals have such a
common waveform as described above, they have output timings having
a phase difference of 180.degree. from each other.
[0118] Since the drive signals having the waveforms are applied to
the switching elements Q1 and Q2, the switching elements Q1 and Q2
perform switching operation such that they are alternately switched
on and off with a switching frequency which depends upon the
oscillation frequency of the oscillation circuit.
[0119] It is to be noted that, in actual switching operation, a
short period of dead time wherein both of the switching elements Q1
and Q2 exhibit an off state is formed within a period of time after
the switching element Q1 is turned off until the switching element
Q2 is turned on and within another period of time after the
switching element Q2 is turned off until the switching element Q1
is turned.
[0120] The dead time is a period of time within which both of the
switching elements Q1 and Q2 are off. The dead time is formed in
order that charging and discharging operation can be obtained with
certainty within a short period of time at timings at which the
switching elements Q1 and Q2 are turned on/off as partial voltage
resonance operation. The time length of such dead time as just
described can be set, for example, by the control IC 2 side. In
particular, the control IC 2 adjusts the duty ratio of the pulse
width of the drive signals to be outputted from the drive signal
output terminals VGH and VGL so that a dead time period of the set
time length may be formed.
[0121] An insulating converter transformer PIT is provided in order
to transmit switching outputs of the switching elements Q1 and Q2
to the secondary side.
[0122] In this instance, the primary winding N1 of the insulating
converter transformer PIT is connected at one end thereof to a node
(switching output point) between the source of the switching
element Q1 and the drain of the switching element Q2 through a
series connection of a series resonance capacitor C1. The primary
winding N1 is connected at the other end thereof to the primary
side ground.
[0123] According to the connection scheme described above, a series
circuit of the series resonance capacitor C1 and the primary
winding N1 is connected to the switching output point of the
switching elements Q1 and Q2. Consequently, a primary side series
resonance circuit is formed from the capacitance of the series
resonance capacitor C1 and a leakage inductance L1 of the
insulating converter transformer PIT including the primary winding
N1. Since the primary side series resonance circuit is connected to
the switching output point in such a manner as described above, the
switching output of the switching elements Q1 and Q2 is transmitted
to the primary side series resonance circuit. The primary side
series resonance circuit performs resonance operation in response
to the switching output transmitted thereto. Consequently,
operation of the primary side switching converter becomes that of
the current resonance type.
[0124] In other words, operation of the current resonance type by
the primary side series resonance circuit (C1-L1) and partial
resonance operation by the partial voltage resonance circuit
(Cp//L1) described hereinabove are obtained by the primary side
switching converter shown in FIG. 1.
[0125] In other words, the power supply circuit shown in FIG. 1
adopts a configuration as a composite resonance type converter
which includes a combination of a resonance circuit for making the
primary side switching converter that of the resonance type with a
different resonance circuit. It is to be noted that, in the
composite resonance type converter described in the present
specification, the different resonance circuit may be provided on
the primary side or otherwise on the secondary side. In the case of
the circuit shown in FIG. 1, a partial voltage resonance circuit is
provided as the different resonance circuit.
[0126] Although description with reference to the drawings is
omitted, the insulating converter transformer PIT is structured
such that it includes, for example, an EE type core which is formed
from a combination of E type cores made of a ferrite material.
Further, a wiring receiving portion of the insulating converter
transformer PIT is divided into winding receiving portions for the
primary side and the secondary side, and the primary winding N1 and
secondary windings N2 and N2A which are described below are wound
on a central magnetic leg of the EE type core.
[0127] Further, a gap of 1.0 mm to 1.5 mm is formed in the central
magnetic leg of the EE type core of the insulating converter
transformer PIT. Consequently, a rough coupling state with a
coupling coefficient of approximately 0.7 to 0.8 is obtained.
[0128] The secondary side of the power supply circuit shown in FIG.
1 produces and outputs a plurality of secondary side DC output
voltages and in this instance outputs three secondary side DC
output voltages including secondary side DC output voltages Eo, Eo1
and Eo2.
[0129] In order to obtain three secondary side DC output voltages
in this manner, in the circuit shown in FIG. 1, two secondary
windings N2 and N2A are wound on the secondary side of the
insulating converter transformer PIT. The secondary side DC output
voltages Eo and Eo2 are produced from the secondary winding N2
side, and the secondary side DC output voltage Eo1 is produced from
the secondary winding N2A side and outputted. Further, the present
embodiment is configured such that constant voltage control is
performed individually for each of the secondary side DC output
voltages Eo, Eo1 and Eo2 obtained in this manner.
[0130] The secondary winding N2 has a center tap provided thereon
as shown in FIG. 1 and connected to the secondary side ground, and
a full-wave rectification circuit formed from rectification diodes
Do1 and Do2 and a smoothing capacitor Co is connected to the
secondary winding N2. Consequently, a secondary side DC output
voltage Eo is obtained as a voltage across the smoothing capacitor
Co. The secondary side DC output voltage Eo is supplied to the load
side not shown and is branched and inputted also as a detection
voltage for a control circuit 1 described below.
[0131] While stabilization of the secondary side DC output voltage
Eo is performed by execution of constant voltage control according
to a switching frequency controlling method by a constant voltage
control circuit system which includes the control circuit 1, this
is hereinafter described.
[0132] To the secondary winding N2, a full-wave rectification
circuit for producing the secondary side DC output voltage Eo is
connected in parallel to the full-wave rectification circuit for
the secondary side DC output voltage Eo described above.
[0133] In particular, to one of the opposite end portions of the
secondary winding N2, the anode of a rectification diode Do5 is
connected through a series connection of a controlled winding NR1
of an orthogonal control transformer PRT-2 hereinafter described.
Meanwhile, to the other end portion of the secondary winding N2,
the anode of another rectification diode Do6 is connected through a
series connection of the controlled winding NR1 of the orthogonal
control transformer PRT-2. The cathodes of the rectification diodes
Do5 and Do6 are connected to the positive terminal of a smoothing
capacitor Co2. The negative terminal of the smoothing capacitor Co2
is connected to the secondary side ground. Here, the controlled
winding NR1 of the orthogonal control transformer PRT-2 is
connected at a winding ending end portion thereof to the secondary
winding N2 side and connected at a winding starting end portion
thereof to the rectification diode Do5 side. In contrast, another
controlled winding NR2 is connected at a winding starting end
portion thereof to the secondary winding N2 side and connected at a
winding ending end portion thereof to the rectification diode Do5
side.
[0134] By the full-wave rectification circuit formed in this
manner, the secondary side DC output voltage Eo2 is obtained as a
voltage across the smoothing capacitor Co2.
[0135] While description is hereinafter given, stabilization of the
secondary side DC output voltage Eo2 is performed by a constant
voltage control circuit system formed from a control circuit 3-2
and the orthogonal control transformer PRT-2.
[0136] A center tap is provided for the secondary winding N2A of
the different secondary winding set and is connected to the
secondary side ground, and a full-wave rectification circuit formed
from rectification diodes Do3 and Do4 and a smoothing capacitor Co1
is connected in a connection scheme similar to that for the
secondary side DC output voltage Eo2 described hereinabove to the
secondary winding N2A. The full-wave rectification circuit produces
the secondary side DC output voltage Eo1 as a voltage across the
smoothing capacitor Co1. Also in this instance, the controlled
windings NR1 and NR2 of the orthogonal control transformer PRT-1
are inserted in series between the anodes of the rectification
diodes Do3 and Do4 and the end portions of the secondary winding
N2A, respectively.
[0137] Stabilization of the secondary side DC output voltage Eo1 is
performed by a constant voltage control circuit system formed from
a control circuit 3-1 and the orthogonal control transformer
PRT-1.
[0138] The load conditions with regard to the secondary side DC
output voltages Eo, Eo1 and Eo2 in the power supply circuit shown
in FIG. 1 are such as given below:
[0139] Eo=5.0 V/6 A to 2 A
[0140] Eo1=12.0 V/1 A to 0.2 A
[0141] Eo2=3.3 V/6A to 2 A
[0142] Now, a configuration for stabilizing the secondary side DC
output voltages Eo, Eo1 and Eo2 described above is described.
[0143] First, stabilization of the secondary side DC output voltage
Eo to which the highest load power is applied is performed in the
following manner.
[0144] In order to make the secondary side DC output voltage Eo a
constant voltage, the secondary side DC output voltage Eo is
inputted as a detection voltage to the control circuit 1. The
control circuit 1 produces current or a voltage, whose level is
adjusted in response to the level of the secondary side DC output
voltage Eo, as a control output. The control output is outputted to
the control terminal Vc of the control IC 2.
[0145] The control IC 2 operates in response to the control output
level inputted to the control terminal Vc so as to vary the drive
signal for the high side and the drive signal for the low side to
be outputted from the drive signal output terminals VGH and VGL,
respectively, with the frequencies of the drive signals kept in
synchronism with each other while the timings at which the drive
signals are turned on/off alternately are maintained. In
particular, the control IC 2 varies the oscillation frequency of
the internal oscillation circuit.
[0146] Consequently, the switching frequency of the switching
elements Q1 and Q2 is adjustably controlled in response to the
control output level (that is, the secondary side DC output voltage
level) inputted to the control terminal Vc.
[0147] When the switching frequency is adjusted, the resonance
impedance of the primary side series resonance circuit varies. When
the resonance impedance varies in this manner, the amount of
current to be supplied to the primary winding N1 varies and also
the power to be transmitted to the secondary side varies.
Consequently, the level of the secondary side DC output voltage Eo
varies, and constant voltage control of the secondary side DC
output voltage Eo is achieved thereby.
[0148] Now, a configuration for stabilizing the secondary side DC
output voltage Eo2 is described.
[0149] As can be recognized also from the foregoing description,
the constant voltage control circuit system for the secondary side
DC output voltage Eo2 includes the orthogonal control transformer
PRT-2 and the control circuit 3-2.
[0150] Here, a structure of the orthogonal control transformer
PRT-2 is described with reference to FIG. 2. It is to be noted that
also the orthogonal control transformer PRT-1 provided for the
secondary side DC output voltage Eo2 side has a similar
structure.
[0151] As seen in FIG. 2, the orthogonal control transformer PRT
(PRT-1, PRT-2) includes two double channel-shaped cores 21 and 22
made of, for example, ferrite. Each of the double channel-shaped
cores 21 and 22 has four magnetic legs as seen in FIG. 2. A solid
core 20 is formed by joining end portions of the magnetic legs of
the two double channel-shaped cores 21 and 22 to each other.
[0152] In this instance, the joining portions of the magnetic legs
of the double channel-shaped cores 21 and 22 do not have a gap
formed therein.
[0153] Further, a controlled winding NR (NR1, NR2) is wound on two
magnetic legs of the double channel-shaped core 21 while a
controlling winding NC is wound on two magnetic legs of the double
channel-shaped core 21 in such a manner as seen in FIG. 2. In this
instance, the winding directions of the controlled winding NR and
the controlling winding NC are such that the controlling winding NC
is orthogonal to the controlled winding NR as seen in FIG. 2.
Consequently, no transformer coupling is provided between the
controlled winding NR and the controlling winding NC. Thus, a
configuration as a saturable rector having a characteristic that
the inductance of the controlled winding NR varies in response to
the level of DC current flowing through the controlling winding NC
is obtained.
[0154] In the orthogonal control transformer PRT in the present
embodiment, the controlled winding NR (NR1, NR2) is formed by
winding a polyurethane-coated copper wire of 60 .mu.m.phi./80
strands by 4 T (turns). Meanwhile, the controlling winding NC is
formed by winding, for example, a polyurethane-coated copper wire
of 60 .mu.m.phi. by 1,000 T (turns).
[0155] Under such winding specifications as described above, the
inductance LR of the controlled winding NR exhibits such a DC
superposition characteristic as illustrated in FIG. 3. In FIG. 3,
the axis of ordinate indicates the inductance LR and the axis of
abscissa indicates the current IR flowing to the controlled winding
NR.
[0156] As can be recognized from FIG. 3, the inductance LR
indicates a variation range of LR=20 pH to 1.5 pH with respect to
the variation range of the control current Ic=10 mA to 40 mA.
[0157] The control circuit 3-2 adjusts the DC current level of the
control current to be supplied to the controlling winding NC in
response to an error of the level of the secondary side DC output
voltage Eo2 inputted as a detection voltage thereto and outputs the
adjusted DC current level.
[0158] As the level of control current flowing to the controlling
winding NC of the orthogonal control transformer PRT-2 is adjusted
in this manner, the inductance LR of the controlled winding NR
varies in such a manner as seen in FIG. 3.
[0159] Here, where the inductance of the secondary winding N2 by
which an alternating voltage on which the secondary side DC output
voltage Eo2 is based is represented by L2 and the potential between
a node between the secondary winding N2 and the controlled winding
NR2 of the orthogonal control transformer PRT-2 and the secondary
side ground is represented by V2, the potential V3 between the
anode of the rectification diode Do6 which forms the rectification
circuit system for the secondary side DC output voltage Eo2 and the
secondary side ground is represented in the following manner.
V3=V2.times.(L2/(L2+LR))=V2.times.(1+(LR/L2)) (Expression 1)
[0160] As can be recognized from Expression 1, the level of the
potential V3 varies by adjustment of the inductance LR. When the
level of the potential V3 varies, also the level of the secondary
side DC output voltage Eo2 varies in response to the variation of
the same. Accordingly, when the inductance LR of the controlled
winding NR is adjusted in response to the level error of the
secondary side DC output voltage Eo2 in such a manner as described
above, the level of the secondary side DC output voltage Eo2 is
controlled so as to be stabilized.
[0161] An example of an internal configuration of the control
circuit 3-2 is shown in FIG. 4.
[0162] In the control circuit 3-2 shown in FIG. 4, the secondary
side DC output voltage Eo2 is divided by voltage dividing resistors
R11-R12, and the divided voltage level is inputted to a control
terminal of a shunt regulator Q3. Through the shunt regulator Q3,
current of a level corresponding to the error of the divided
voltage level (level of the secondary side DC output voltage Eo2)
inputted to the control terminal flows from a power supply line of
12 V through a resistor R13. It is to be noted that the power
supply line of 12 V may be derived, for example, from the secondary
side DC output voltage Eo1.
[0163] The base of a transistor Q4 is connected to a node between
the shunt regulator Q3 and a series circuit of resistors R14-R15
connected in parallel to each other. Meanwhile, the collector of
the transistor Q4 is connected to the 12 V power supply line
through a series connection of the controlling winding NC of the
orthogonal control transformer PRT-2. The emitter of the transistor
Q4 is connected to the secondary side ground, and a resistor R16 is
connected in parallel to the emitter of the transistor Q4.
[0164] Where such a connection scheme as described above is used,
DC current of the level amplified in response to the level of
current flowing to the shunt regulator Q3 flows to the collector of
the transistor Q4. The DC current becomes the control current to
the controlling winding NC.
[0165] In this manner, the control circuit 3-2 detects the level
error of the secondary side DC output voltage Eo2 by means of the
shunt regulator Q3 and supplies collector current of the transistor
Q4, whose level is adjusted in response to the detected error, as
control current to the controlling winding NC. In other words, the
control circuit 3-2 is configured so as to supply DC current, whose
level is adjusted in response to the level error of the secondary
side DC output voltage Eo2, as control current to the controlling
winding NC.
[0166] It is to be noted that the control circuit 3-2 formed based
on the circuit configuration shown in FIG. 4 is actually configured
such that the resistance value of the controlling winding NC of the
orthogonal control transformer PRT-2 is set to 20.OMEGA. so that
the control current Ic may have a control range of Ic=10 mA to 30
mA. By setting such a control range as just described, constant
voltage control ready for the actual load variation of the
secondary side DC output voltage Eo2 is implemented.
[0167] A waveform diagram of FIG. 5 illustrates operation of the
constant voltage control circuit system for the secondary side DC
output voltage Eo2 described above. The operation illustrated in
FIG. 5 indicates operation where the secondary side DC output
voltage Eo2 is Eo2=3.3 V/6 A.
[0168] First, for the potential V2, a waveform is obtained which is
reversed between the positive and negative polarities for each one
cycle in response to the cycle of the alternating voltage obtained
by the secondary winding N2 in such a manner as seen in FIG. 5. The
peak levels of the positive and negative polarities are 6 Vp.
[0169] Also the potential V3 has a waveform which is reversed
between the positive and negative polarities at cyclical timings
same as those of the potential V2. However, the waveform of the
positive polarity of the potential V3 within the on period of the
rectification diode Do6 exhibits a drop to 4 V. This arises from
the fact that the potential V3 is controlled so as to change from 6
V to 4 V by an influence of the variable control of the inductance
LR2 of the controlled winding NR2 as indicated by Expression 1
given hereinabove. In other words, it is indicated from comparison
between the potential V3 and the potential V2 in FIG. 5 that
constant voltage control operation by the control circuit 3-2 and
the orthogonal control transformer PRT-2 is obtained.
[0170] Then, within a period within which the potential V3 has the
positive polarity, rectification current I2 flows through the
rectification diode Do6 to charge the smoothing capacitor Co2.
[0171] Further, though not illustrated in FIG. 5, within a period
within which the potential V3 has the negative polarity in FIG. 5,
similar operation is obtained in the rectification current path
system of the rectification diode Do5 side because the inductance
LR1 of the controlled winding NR1 is variably controlled.
[0172] As a result of such operation as described above obtained in
this manner, the secondary side DC output voltage Eo2 formed as a
constant voltage so as to be, for example, 3.3 V fixed is
obtained.
[0173] It is to be noted that, also in the constant voltage control
circuit system for the secondary side DC output voltage Eo1
connected to the secondary winding N2A, the orthogonal control
transformer PRT-1 and the control circuit 3-1 operate similarly to
the orthogonal control transformer PRT-2 and the control circuit
3-2 described above, respectively. Consequently, also the secondary
side DC output voltage Eo1 is stabilized similarly to the secondary
side DC output voltage Eo2.
[0174] It is to be noted that the control range of the control
current Ic in the control circuit 3-1 and so forth are arbitrarily
set taking actual load variation of the secondary side DC output
voltage Eo1 and so forth into consideration.
[0175] In this manner, the power supply circuit shown in FIG. 1
produces and outputs a plurality of secondary side DC output
voltages. Then, for the secondary side DC output voltage Eo to
which the highest load power is applied, constant voltage control
by the switching frequency control method is performed, and for the
other remaining secondary side DC output voltages Eo1 and Eo2, the
orthogonal control transformer PRT is provided such that
stabilization is achieved by adjusting the inductance of the
controlled winding NR of the orthogonal control transformer PRT
inserted in the rectification current path.
[0176] Here, the power loss by the controlled winding NR of the
orthogonal control transformer PRT is low, and the control power
required by the control circuits 3-1 and 3-2 and so forth for
adjustment of the inductance of the controlled winding NR is
approximately 0.4 W.
[0177] As a result, the power conversion efficiency (.eta.DC/DC) of
the power supply circuit shown in FIG. 1 is 94% with regard to the
secondary side DC output voltage Eo, 95% with regard to the
secondary side DC output voltage Eo1 and approximately 94% with
regard to the secondary side DC output voltage Eo2, and totally is
approximately 90%.
[0178] In contrast, for example, the DC/DC power conversion
efficiency of the circuits shown in FIGS. 14 and 15 is
approximately 88% or 86%, and it can be seen that the circuit shown
in FIG. 1 is enhanced in the power conversion efficiency. It is to
be noted that, if the power losses of the power supply circuit of
FIG. 1 and the power supply circuit of FIG. 14 are compared with
each other, then the power loss of the power supply circuit of FIG.
1 exhibits a reduction by 1.5 W from that of the power supply
circuit of FIG. 14. Further, the power loss of the power supply
circuit of FIG. 1 indicates a reduction by 3.2 W from that of the
power supply circuit of FIG. 15. Consequently, the circuit shown in
FIG. 1 can eliminate the necessity for a heat radiating plate or
the like for the secondary side constant voltage control circuit
system. Where the necessity for a heat radiating plate is
eliminated, reduction in size and weight of the circuit can be
achieved as much.
[0179] Further, in the case of the constant voltage control circuit
system on the secondary side shown in FIG. 1, even where the
rectification circuit is a full-wave rectification circuit, the
constant voltage control circuit system can be formed from the
orthogonal control transformer PRT which is a variable inductance
element and one control circuit 3.
[0180] For example, the orthogonal control transformer PRT requires
a very low cost if compared with a saturable inductor which forms a
magnetic amplifier. Also semiconductor elements which are provided
in the control circuit 3 are only the transistor Q4 of 50 V/0.1 A
and the shunt regulator Q3 which are less expensive as seen in FIG.
4.
[0181] As a result, when compared with a circuit configuration
which includes, for example, the magnetic amplifier constant
voltage circuit shown in FIG. 15, the circuit shown in FIG. 1 can
be produced with a cost suppressed to approximately 1/2.
[0182] In this manner, the power supply circuit of the present
embodiment is enhanced in the power conversion efficiency and can
be produced at a much reduced cost when compared with the power
supply circuit shown in FIG. 14 or 15. Further, the constant
voltage control circuit on the secondary side in the power supply
circuit of the present embodiment performs, when it performs
adjustment of the inductance of the controlled winding NR of the
orthogonal control transformer PRT, DC control of adjusting the
control current (DC current level) to be supplied to the
controlling winding NC, and involves no switching operation.
Accordingly, the power supply circuit of the present embodiment
does not suffer from the problem of interference between different
switching frequencies which is a problem to the circuit of FIG.
14.
[0183] Incidentally, the secondary side shown in FIG. 1 adopts a
configuration wherein the secondary side DC output voltages Eo, Eo1
and Eo2 are produced by respective full-wave rectification
circuits. However, the configuration of the rectification circuits
on the secondary side is not limited to a full-wave rectification
circuit, but a rectification circuit of a different type may be
used in accordance with the level of an actually required secondary
side DC output voltage, an actually required load current amount
and so forth.
[0184] Thus, four different modifications with regard to the
configuration of the rectification circuit system which is ready
for one secondary side DC output voltage are described with
reference to FIGS. 6 to 9. It is to be noted that the secondary
side DC output voltage obtained from the rectification circuits
shown in the figures is, in the case of FIG. 1, the secondary side
DC output voltage Eo1 or Eo2 which is stabilized by the orthogonal
control transformer PRT.
[0185] First, the rectification circuit system shown in FIG. 6
includes a half-wave rectification circuit formed from a single
rectification diode Do and a single smoothing capacitor Co and
connected to the secondary winding N2.
[0186] Here, for example, when constant voltage control by the
orthogonal control transformer PRT is performed, in the case of the
full-wave rectification circuit shown in FIG. 1, it is necessary to
insert the controlled winding NR of the orthogonal control
transformer PRT into each of the rectification current paths formed
within periods within which the alternating voltage of the
secondary winding N2 has the positive and negative polarities.
[0187] In contrast, in the case of the half-wave rectification
circuit shown in FIG. 6, within a period of one half wave within
which rectification operation is performed, rectification current
flows to a node between an end portion of the secondary winding N2
and the anode of the rectification diode Do5 without fail. In other
words, rectification current does not flow along different
rectification current paths from each other within different
periods within which the alternating voltage of the secondary
winding N2 has the positive and negative polarities within one
cycle as in the case of a full-wave rectification circuit.
[0188] Accordingly, in the case wherein a half-wave rectification
circuit is formed in such a manner as seen in FIG. 6, it is only
necessary to insert a single controlled winding NR of the
orthogonal control transformer PRT in series between the end
portion of the secondary winding N2 and the node of the anode of
the rectification diode Do5. In other words, in the case of the
rectification circuit shown in FIG. 6, the number of controlled
windings NR to be wound on the orthogonal control transformer PRT
can be reduced from that in the case of a full-wave rectification
circuit. By reducing the number of windings to be wound on the
orthogonal control transformer PRT in this manner, for example,
reduction of the cost required for the reduced number of windings
can be anticipated, and the production efficiency of the orthogonal
control transformer PRT is enhanced. Also it becomes possible to
further miniaturize the orthogonal control transformer PRT.
[0189] FIG. 7 shows an example wherein a full-wave rectification
circuit formed from a bridge rectification circuit DBR and a
smoothing capacitor Co is connected to the secondary winding
N2.
[0190] In the case of such a full-wave rectification circuit as
just described, rectification current flows without fail along a
line, for example, between the positive input terminal of the
bridge rectification circuit DBR and the end portion of the
secondary winding N2 within both periods within which the
alternating voltage of the secondary winding N2 has the positive
and negative polarities.
[0191] Therefore, also in this instance, it is only necessary to
insert a single controlled winding NR of the orthogonal control
transformer PRT into the line between the positive input terminal
of the bridge rectification circuit DBR and the end portion of the
secondary winding N2. It is to be noted that, even if the
controlled winding NR is inserted into another line, for example,
between the negative input terminal of the bridge rectification
circuit DBR and the other end portion of the secondary winding N2,
equivalent constant voltage control operation can be achieved.
[0192] FIG. 8 shows a case wherein a voltage doubler rectification
circuit is connected to the secondary winding N2.
[0193] The voltage doubler rectification circuit in this instance
is formed by connecting two rectification diodes Do1 and Do2 and
two smoothing capacitors CoA-CoB connected in series in such a
connection scheme as shown in FIG. 8.
[0194] Also in this instance, the orthogonal control transformer
PRT includes a single controlled winding NR which is inserted in
series between the end portion of the secondary winding N2 and a
node between smoothing capacitors CoA-CoB connected in series.
[0195] Here, rectification operation of the voltage doubler
rectification circuit shown in FIG. 8 is described. First, within a
period within which the alternating voltage of the secondary
winding N2 has the positive polarity, rectification current flows
along a path of the secondary winding N2.fwdarw.rectification diode
Do1.fwdarw.smoothing capacitor CoA.fwdarw.controlled winding NR
secondary winding N2 to charge the smoothing capacitor CoA.
Consequently, a rectification smoothed voltage of a level equal to
the level of the alternating voltage obtained by the secondary
winding N2 is obtained across the smoothing capacitor CoA.
[0196] On the other hand, within another period of time within
which the alternating voltage of the secondary winding N2 has the
negative polarity, rectification current flows along another path
of the secondary winding N2.fwdarw.controlled winding
NR.fwdarw.smoothing capacitor CoB.fwdarw.rectification diode
Do2.fwdarw.secondary winding N2 to charge the smoothing capacitor
CoB. Consequently, a rectification smoothed voltage of a level
equal to the level of the alternating voltage obtained by the
secondary winding N2 is obtained also across the smoothing
capacitor CoB.
[0197] Since such rectification operation as described above is
repeated after every one cycle, a rectification smoothed voltage of
a level equal to twice the level of the alternating voltage of the
secondary winding N2 is obtained as a voltage across the series
circuit of the smoothing capacitors CoA-CoB. This rectification
smoothed voltage becomes a secondary side DC output voltage. The
secondary side DC output voltage by the voltage doubler
rectification operation is obtained in this manner.
[0198] According to the rectification operation described above,
rectification current flows through the single controlled winding
NR, which is inserted between the end portion of the secondary
winding N2 and the node between the smoothing capacitors CoA-CoB,
within both periods within which the alternating voltage of the
secondary winding N2 has the positive and negative polarities. In
other words, also in this instance, the controlled winding NR is
inserted in the paths along which rectification current flows
within both periods within which the alternating voltage of the
secondary winding N2 has the positive and negative polarities.
Accordingly, also in this instance, only one controlled winding NR
is required.
[0199] The rectification circuit shown in FIG. 9 forms a voltage
quadrupler rectification circuit. In the voltage quadrupler
rectification circuit, voltage doubler rectification operation is
performed by a rectification circuit section formed from
rectification diodes Do1 and Do2 and smoothing capacitors CoA and
CoC, and a rectification smoothed voltage of a level equal to twice
the level of the alternating voltage of the secondary winding N2 is
obtained as a voltage across the smoothing capacitor CoA.
[0200] Meanwhile, voltage doubler rectification operation is
performed also by another rectification circuit section formed from
rectification diodes Do3 and Do4 and smoothing capacitors CoB and
CoD, and a rectification smoothed voltage of a level equal to twice
the level of the alternating voltage of the secondary winding N2 is
obtained as a voltage across the smoothing capacitor CoB.
[0201] As a result, as a secondary side DC output voltage which is
a voltage across the smoothing capacitors CoA-CoB connected in
series, a rectification smoothed voltage of a level equal to four
times the level of the alternating voltage of the secondary winding
N2 is obtained.
[0202] Also in this instance, only one controlled winding NR is
required for the orthogonal control transformer PRT, and the
orthogonal control transformer PRT is inserted in series between
the end portion of the secondary winding N2 and the node between
the smoothing capacitors CoA-CoB.
[0203] First, the voltage doubler rectification operation of the
rectification circuit section formed from the rectification diodes
Do1 and Do2 and the smoothing capacitors CoA and CoC is such as
follows.
[0204] Within a period within which the alternating voltage of the
secondary winding N2 has the negative polarity, rectification
current flows along a path of the secondary winding
N2.fwdarw.controlled winding NR.fwdarw.rectification diode
Do2.fwdarw.smoothing capacitor CoC.fwdarw.secondary winding N2 to
charge the smoothing capacitor CoC. Consequently, a DC voltage of a
level equal to the level of the alternating voltage obtained by the
secondary winding N2 is obtained as a voltage across the smoothing
capacitor CoC.
[0205] On the other hand, within another period within which the
alternating voltage of the secondary winding N2 has the positive
polarity, rectification current flows along a path of the secondary
winding N2.fwdarw.smoothing capacitor CoC.fwdarw.rectification
diode Do1.fwdarw.smoothing capacitor CoA.fwdarw.controlled winding
NR.fwdarw.secondary winding N2. At this time, since charging into
the smoothing capacitor CoA is performed in such a manner that the
voltage obtained across the smoothing capacitor CoC is superposed,
a DC voltage of a level equal to twice the alternating voltage of
the secondary winding N2 is obtained as a voltage across the
smoothing capacitor CoA. Voltage doubler rectification operation is
performed in this manner.
[0206] On the other hand, in the rectification circuit section
formed from the rectification diodes Do3 and Do4 and the smoothing
capacitors CoB and CoD, first within a period within which the
alternating voltage of the secondary winding N2 has the positive
polarity, rectification current flows along a path of the secondary
winding N2.fwdarw.smoothing capacitor CoD.fwdarw.rectification
diode Do3.fwdarw.controlled winding NR.fwdarw.secondary winding N2.
Consequently, a DC voltage of a level equal to the level of the
alternating voltage of the secondary winding N2 is obtained as a
voltage across the smoothing capacitor CoD.
[0207] On the other hand, within another period within which the
alternating voltage of the secondary winding N2 has the negative
polarity, rectification current flows along another path of the
secondary winding N2.fwdarw.controlled winding NR.fwdarw.smoothing
capacitor CoB.fwdarw.rectification diode Do4.fwdarw.smoothing
capacitor CoD.fwdarw.secondary winding N2. At this time, since
charging into the smoothing capacitor CoB is performed in such a
manner that the voltage obtained across the smoothing capacitor CoD
is superposed, a DC voltage of a level equal to twice the
alternating voltage of the secondary winding N2 is obtained as a
voltage across the smoothing capacitor CoA. In short, voltage
doubler rectification operation is performed.
[0208] According to such rectification operation as described
above, in the rectification operation of any of the two
rectification circuit sections, rectification current flows
commonly through the controlled winding NR within both periods
within which the alternating voltage of the secondary winding N2
has the positive and negative voltages. In short, also in this
instance, only one controlled winding NR is required by insertion
of the controlled winding NR at the position shown in FIG. 9.
[0209] FIG. 10 shows an example of a configuration of a power
supply circuit as a second embodiment. It is to be noted that, in
FIG. 10, like elements to those of FIG. 1 are denoted by like
reference characters and description thereof is omitted herein.
[0210] The orthogonal control transformers PRT (orthogonal control
transformers PRT-1 and PRT-2) shown in FIG. 10 may have a
configuration similar to that of FIG. 2 described hereinabove.
[0211] In the power supply circuit shown in FIG. 10, the control
circuit 3-2 on the secondary side which controls the inductance of
the controlled windings NR (NR1, NR2) of the orthogonal control
transformer PRT (PRT-1, PRT-2) on the secondary side is configured
in such a manner as shown in FIG. 11. It is to be noted that, in
FIG. 11, like elements to those of FIG. 4 are denoted by like
reference characters and description thereof is omitted herein.
[0212] In the control circuit 3-2 shown in FIG. 11, a capacitor C2
is inserted with the polarities illustrated in FIG. 11 between the
secondary side DC output voltage Eo1 which is a power supply line
of 12 V and the controlling winding NC of the orthogonal control
transformer PRT. The capacitor C2 in this instance is an
electrolytic capacitor, and the positive electrode of the capacitor
C2 is connected to the line of the secondary side DC output voltage
Eo1 while the negative electrode of the capacitor C2 is connected
to an end portion of the controlling winding NC.
[0213] Further, the emitter of a transistor Q5 is connected to the
positive electrode of the capacitor C2, and the collector of the
transistor Q5 is connected to the negative electrode of the
capacitor C2. A resistor R17 is a base-emitter resistor of the
transistor Q5.
[0214] The base of the transistor Q5 is connected to the collector
of a transistor Q6 through a resistor R18. The emitter of the
transistor Q6 is connected to the secondary side ground. The base
of the transistor Q6 is connected to the line of the secondary side
DC output voltage Eo2 through a resistor R20. A resistor R19 is a
base-emitter resistor of the transistor Q6.
[0215] According to such a configuration of the control circuit 3-2
as described above, for example, if a commercial AC power supply AC
is made available and the secondary side DC output voltage Eo1
builds up to a prescribed level, then control current Ic first
flows from the line of the secondary side DC output voltage Eo1 to
the controlling winding NC through the capacitor C2.
[0216] Thereupon, also the secondary side DC output voltage Eo2
builds up in response to the starting of the commercial AC power
supply AC. However, if the secondary side DC output voltage Eo2
rises to a level higher than a predetermined level (for example, 2
V), then a base-emitter voltage sufficient to render the transistor
Q6 conducting is obtained and places the transistor Q6 into an on
state. In response to this, also the transistor Q5 is placed into
an on state.
[0217] After the transistor Q5 is placed into an on state, the path
of the control current Ic changes over from the line of the
secondary side DC output voltage Eo1 to the path which includes the
emitter-collector of the transistor Q5, and the control current Ic
flows along the new path. Thereafter, the level of the control
current Ic is controlled by operation of the error amplifier formed
from the shunt regulator Q3 and the transistor Q4 in response to
the level of the secondary side DC output voltage Eo2 as described
hereinabove with reference to FIG. 4. Consequently, the secondary
side DC output voltage Eo2 is stabilized, for example, at 3.3
V.
[0218] It is assumed here that load short-circuiting occurs with
the secondary side DC output voltage Eo2. In this instance, the
secondary side DC output voltage Eo2 drops to the zero level, and
the transistor Q6 is controlled so as to change over from the on
state till then to an off state in response to the drop of the
secondary side DC output voltage Eo2. Thereupon, also the
transistor Q5 is changed over into an off state.
[0219] After the transistor Q5 is placed into an off state in this
manner, since the path along which the control current Ic is
supplied is disconnected from the line of the secondary side DC
output voltage Eo1, the control current Ic decreases to the 0
level. Consequently, the inductance of the controlled wirings NR1
and NR2 increases as described hereinabove with reference to FIG.
3.
[0220] The waveforms of the secondary side DC output voltage Eo2
and the rectification current I2 which flows through the
rectification diodes Do5 and Do6 at this time are illustrated in
FIG. 12.
[0221] As described hereinabove, when load short-circuiting occurs,
the secondary side DC output voltage Eo2 continues to have the 0
level.
[0222] Then, although the rectification current I2 should
originally increase to a very high level as a result of the load
short-circuiting, since the control current Ic is suppressed to the
0 level and the controlled wirings NR1 and NR2 have increased
inductances, the level of the rectification current I2 which flows
in high frequency in accordance with the switching period is
suppressed. For example, while the rectification current I2 flows
at the level of 15 Ap as seen in FIG. 5 also in a steady state, it
can be seen that, according to the waveform diagram of FIG. 12
which shows the waveforms upon load short-circuiting, the
rectification current I2 is suppressed down to 1.2 Ap.
[0223] In other words, the control circuit 3-2 shown in FIG. 11 is
furnished also with a function as a protection circuit ready for
load short-circuiting of the secondary side DC output voltage Eo2.
Such a load short-circuiting protection function as just described
can be implemented by a simple and low-cost circuit which includes
several resistance elements and so forth in addition to the
transistors Q5 and Q6 and the capacitor C2 which are provided as
principal components.
[0224] For example, in the configurations shown in FIGS. 14 and 15,
if it is tried to provide a protection function against load
short-circuiting, then it is necessary to form and connect a more
complicated load short-circuiting protection circuit. Consequently,
the circuit scale increases and increase of the cost is invited
when compared with the present embodiment.
[0225] Further, in the power supply circuit as the second
embodiment shown in FIG. 10, a resistor R1 is connected in such a
manner as seen in FIG. 10, for example, to the constant voltage
control circuit system for the secondary side DC output voltage
Eo1. The resistor R1 is inserted between end portions of the two
controlled wirings NR1 and NR2 on the rectification diode side.
[0226] Also to the constant voltage control circuit system
corresponding to the secondary side DC output voltage Eo2, a
resistor R2 is connected in a similar connection scheme.
[0227] For example, where the resistor R1 or R2 is not inserted in
such a manner as described above, noise caused by parasitic
oscillation is generated on the potential V3 between the node
between the rectification diode and the controlled winding NR and
the secondary side ground at a timing at which the rectification
diode turns off.
[0228] Therefore, where the resistors R1 and R2 are inserted in
such a manner as seen in FIG. 10, such noise as parasitic
oscillation as described above is removed as seen from the
potential V3 of FIG. 13. It is to be noted that the current IQ2
illustrated in FIG. 13 is switching current which flows through the
switching element Q2. The current IQ2 is illustrated in FIG. 13 in
order to indicate that the variation of the potential V3
corresponds to the switching period.
[0229] For example, the peak level of a voltage as parasitic
oscillation generated on the secondary side DC output voltage Eo1
on the 12 V line is 45 Vp, and corresponding to this, it is
necessary to select a rectification diode part having a voltage
resisting property of 60 V for the rectification diodes Do3 and
Do4.
[0230] In contrast, where the resistor R1 is connected to remove
parasitic oscillation as in the case of the present embodiment, a
voltage withstanding property of only 40 V is required for the
rectification diodes Do3 and Do4, and the cost for the
rectification diodes can be reduced as much. Further, since a
better characteristic can be obtained with a part whose voltage
withstanding property is low, the reliability of the circuit is
enhanced.
[0231] It is to be noted that the load short-circuiting protection
circuit (refer to FIG. 11) of the control circuit 3-2 on the
secondary side described as the second embodiment and the resistors
(R1, R2) for parasitic oscillation removal need not necessarily be
adopted in combination. In particular, for example, only the
control circuit 3-2 which includes the load short-circuiting
protection circuit shown in FIG. 11 may be added to the power
supply circuit as the first embodiment shown in FIG. 1 which is
used as a base circuit, or a configuration which includes only the
resistors for parasitic oscillation removal may be adopted.
[0232] Also in the power supply circuit as the second embodiment,
the rectification circuit systems for producing a secondary side DC
output voltage shown in FIGS. 6 to 9 can be applied.
[0233] Further, the present invention is not limited to the
configurations of the power supply circuits described
hereinabove.
[0234] For example, for the switching elements, an element other
than a MOS-FET such as, for example, an IGBT (Insulated Gate
Bipolar Transistor) may be adopted only if it can be used in a
separately excited fashion. Further, also the constants of the part
elements described hereinabove may be changed in accordance with
actual conditions and so forth.
[0235] Further, according to the present invention, the power
supply circuit can be configured including a current resonance type
converter which is of the self-excited type and uses a half bridge
coupling system. In this instance, for example, a bipolar
transistor can be selectively used for the switching elements.
[0236] Furthermore, for example, also for the circuit configuration
for producing a secondary side DC output voltage on the secondary
side of the insulating converter transformer PIT, a configuration
different from those shown in the drawings may be adopted.
[0237] Further, the number of secondary side DC output voltages to
be produced by the power supply circuit according to the present
invention may be changed suitably, for example, in response to the
load power for which the power supply circuit is to be ready, the
number of required DC power supplies and so forth. Also the number
of secondary windings to be wound on the secondary side of the
insulating converter transformer PIT may be changed in response to
the number of secondary side DC output voltages.
INDUSTRIAL APPLICABILITY
[0238] As described above, the switching power supply circuit of
the present invention includes a current resonance type converter
as a basic configuration of a primary side switching converter.
Further, a plurality of secondary side DC output voltages are
produced on the secondary side.
[0239] A particular one of the plural secondary side DC output
voltages is controlled to a constant voltage by variably
controlling the switching frequency of the primary side switching
converter in response to the level of the secondary side DC output
voltage.
[0240] Further, in order to control any one of the remaining
secondary side DC output voltages which requires stabilization to a
fixed voltage, a control transformer as a saturable reactor which
includes a controlling winding and a controlled winding is provided
corresponding to the secondary side DC output voltage. Then, the
controlled winding is inserted into a rectification current path
for producing the secondary side DC output voltage of the control
object. Then, the level of control current to be supplied to the
controlling winding is adjusted in response to the level of the
secondary side DC output voltage of the control object to adjust
the inductance of the controlled winding thereby to achieve
stabilization of the secondary side DC output voltage of the
control object.
[0241] With the configuration described above, since the power loss
decreases when compared with an alternative case wherein a
secondary side DC output voltage is stabilized by means of, for
example, a series regulator, a step-down type converter or a
magnetic amplifier, the power conversion efficiency of the power
supply circuit is enhanced.
[0242] Further, as an actual circuit, it is necessary to provide
only a control transformer as a saturable reactor and a circuit for
adjusting the level of DC current (control current) to be supplied
to the controlling winding of the control transformer in response
to the secondary side DC output voltage level of the control
object. Consequently, when compared similarly with an alternative
case wherein a secondary side DC output voltage is stabilized by
means of a series regulator, a step-down type converter or a
magnetic amplifier, a configuration for stabilization can be
obtained at a very low cost.
[0243] Furthermore, constant voltage control which utilizes a
control transformer is control of variably controlling the level of
DC current (control current) to be supplied to the controlling
winding of the control transformer, and switching operation
independent of the primary side switching converter is not
involved. Accordingly, such interference between different
switching frequency as in the case wherein a step-down type
converter is adopted does not occur, and also the amount of noise
which may be generated in the power supply circuit is decreased as
much.
* * * * *