U.S. patent application number 11/210032 was filed with the patent office on 2007-01-25 for pfc pre-regulator frequency dithering circuit.
This patent application is currently assigned to Texas Instruments Incorporated. Invention is credited to Michael J. O'Loughlin.
Application Number | 20070019446 11/210032 |
Document ID | / |
Family ID | 37678888 |
Filed Date | 2007-01-25 |
United States Patent
Application |
20070019446 |
Kind Code |
A1 |
O'Loughlin; Michael J. |
January 25, 2007 |
PFC PRE-REGULATOR FREQUENCY DITHERING CIRCUIT
Abstract
A circuit and method of dithering the switching frequency of an
off-line power factor corrected (PFC) pre-regulator. The circuitry
used to dither the frequency is advantageously accomplished by
taking advantage of the PWM's internal timing circuitry. This
invention reduces narrow band EMI and eliminates the need to
provide specialty PWM controllers to achieve dithering.
Inventors: |
O'Loughlin; Michael J.;
(Nashua, NH) |
Correspondence
Address: |
TEXAS INSTRUMENTS INCORPORATED
P O BOX 655474, M/S 3999
DALLAS
TX
75265
US
|
Assignee: |
Texas Instruments
Incorporated
|
Family ID: |
37678888 |
Appl. No.: |
11/210032 |
Filed: |
August 23, 2005 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
60702031 |
Jul 22, 2005 |
|
|
|
Current U.S.
Class: |
363/41 |
Current CPC
Class: |
Y02B 70/10 20130101;
H02M 1/12 20130101; Y02B 70/12 20130101; Y02P 80/112 20151101; Y02P
80/10 20151101; H02M 1/42 20130101 |
Class at
Publication: |
363/041 |
International
Class: |
H02M 1/12 20060101
H02M001/12 |
Claims
1. (canceled)
2. A pulse width modulator (PWM) comprising: circuitry configured
to receive an input signal at an input and generate a pulse width
modulated output signal having a switching frequency at an output;
and control circuitry controlling the switching frequency as a
function of a rectified sine wave input signal, wherein the control
circuitry is configured to dither the switching frequency by
varying a charging current for a timing capacitor of an oscillator
with a current path parallel to a timing resistor of the oscillator
where the current path is controlled by a voltage divided from said
rectified sine wave input signal.
3. The pulse width modulator as specified in claim 2, wherein the
control circuitry is configured to dither the switching frequency
without the requirement of a remote controller.
4. The pulse width modulator as specified in claim 3, wherein the
control circuitry is configured to dither the switching frequency
up to about 20%.
5. The pulse width modulator as specified in claim 2, wherein the
control circuitry is configured to generate a voltage feed forward
(VFF) signal that is proportional to the rectified sine wave
input.
6. The pulse width modulator as specified in claim 5, wherein the
control circuit comprises a transistor configured to sink a current
as a function of the VFF signal.
7. The pulse width modulator as specified in claim 6, wherein the
transistor sinks the current as a function of a varying peak line
voltage of the input.
Description
CLAIM OF PRIORITY
[0001] This application claims priority of U.S. Provisional Patent
application Ser. No. 60/702,031 filed Jul. 22, 2005 entitled "PFC
Pre-Regulator Frequency Dithering Circuit", the teaching of which
are incorporated herein by reference.
FIELD OF THE INVENTION
[0002] The present invention relates to pulse width modulators
(PWMs).
BACKGROUND OF THE INVENTION
[0003] Many off-line power converters implement a pulse width
modulator to manage the power conversion. In off-line AC to DC
converters, dithering the PWM frequency has proven to reduce
electromagnetic interference (EMI) by changing it from narrowband
to broadband.
SUMMARY OF INVENTION
[0004] The present invention achieved technical advantages as a
circuit and method of dithering the switching frequency of an
off-line power factor corrected (PFC) pre-regulator. The circuitry
used to dither the frequency is advantageously accomplished by
taking advantage of the PWM's internal timing circuitry. This
invention reduces narrow band EMI and eliminates the need to
provide specialty PWM controllers to achieve dithering.
[0005] The frequency dithering technique reduces the narrowband EMI
by spreading out the noise spectrum of the power supply. There are
limitations on how much the oscillator frequency (fs) can be
dithered. Some of the limiting factors are switching losses and
magnetic design. To keep the boost inductor as small as possible
and to keep the switching losses in check the frequency dithering
should be no more than about 20$ to 30% of the fundamental
frequency.
BRIEF DESCRIPTION OF THE DRAWINGS
[0006] FIG. 1 is a block diagram of a typical PFC boost regulator
with a PWM controller;
[0007] FIG. 2 depicts the desired oscillator frequency fs versus
changes in the input sine wave;
[0008] FIG. 3 depicts a functional block diagram of internal timing
circuitry of the PWM;
[0009] FIG. 4 depicts a control circuit for dithering the
oscillator frequency according to one embodiment of the
invention.
[0010] FIGS. 5-7 depict the oscillator frequency varying with the
rectified line voltage. This shows that the dithering circuit does
what it was designed to do.
DETAILED DESCRIPTION OF THE PRESENT INVENTION
[0011] FIG. 1 is a schematic of a 250 W AC to DC off-line power
converter 10 with PFC that is designed for a universal input
voltage. A good portion of conventional PWMs develop timing through
internal circuitry, and it is this internal circuitry that is taken
advantage according to the present invention to dither the
converter's switching frequency to lower narrowband EMI.
[0012] The duty cycle varies from 80% to 100% as a function of the
input voltage sine wave. At input line crossover, the oscillator
frequency fs is at 80% of its maximum and increases and decreases
proportionally with changes in the line voltage. FIG. 2 shows how
frequency fs varies with input voltage sine wave.
[0013] FIG. 3 shows a functional block diagram of the internal
timing circuitry of converter 10 that generates the oscillator
signal of the PWM controller in FIG. 1. The timing is set up by a
resistor R and capacitor C and a comparator. Resistor RT sets the
charging current in the timing capacitor (CT) through the current
mirror formed by transistors Q1 and Q2. An internal comparator with
hysteresis controls when capacitor CT charges and discharges,
forming the PWM timing. To dither the switching frequency of the
PWM, circuitry is added to the PWM to modulate capacitor CT's
charging current.
[0014] One embodiment of the present invention is a PWM with
additional circuitry such as shown at 40 in FIG. 4. When this
circuitry 40 is added to the power converter 10 in FIG. 1, the PWM
oscillator frequency (fs) dithers by 20% as a function of the input
voltage sine wave. This is advantageously accomplished by varying
the charging current of capacitor CT by 20% as the line voltage
varies. This may be accomplished by using resistors RA through RE
and transistor Q2 to vary the oscillating timing circuitry's
charging current 20%. Resistor R1 is the timing resistor in FIG. 1.
Capacitor CA is used filter out any high frequency switching
noise.
[0015] Conventional PFC pre-regulators are typically designed for a
universal input line voltage of 85V to 265V. These changes in line
amplitude would cause changes in the PWM oscillator frequency (fs)
if not corrected. A majority of PFC PWM ICs conventionally use a
multiplier to shape the input current waveform. The multiplier
typically has a voltage feed forward function for power limiting
and easier control loop compensation. The voltage feed forward
circuitry generates a DC voltage (VFF) that is proportional to the
rectified line voltage amplitude. In the UCC3817 control IC, this
VFF voltage comes out at pin 8 of the IC and is filtered with
resistor and a capacitor.
[0016] Advantageously, according to the present invention, the
dithering circuitry 10 uses the VFF signal to correct for changes
in line amplitude and is accomplished by replacing resistor R6 in
FIG. 1 with R6A and R6B and electrical components resistor RF and
transistor Q1. As the input voltage amplitude varies with peak line
voltage, the VFF signal changes proportionately. The VFF voltage
controls the current sink formed by Q1 and resistor RF and sinks
any changes in current through resistor RA caused by varying peak
line voltage. This function ensures that oscillator frequency fs
varies as a function of a rectified input sine wave, and not
function of changing line amplitude.
Illustrative Embodiment
[0017] In the circuit 40 of FIG. 4, a minimum frequency (fmin) of
80 kHz and a maximum frequency (fmax) of 100 kHz is established,
although limitation to these design frequencies is not to be
inferred. A timing capacitor CT of 560 pF may be chosen based on
the IC's manufacturer data sheet. Equation 1 describes the
oscillator frequency behavior of the control IC used in this
design, circuit 40, which frequency will vary with different
manufacturers. fs = 0.6 RT * CT 1 ) ##EQU1##
[0018] Resistor R1 is selected for fmin frequency and which is the
frequency where the converter operates at line crossover. R .times.
.times. 1 = 0.6 f .times. .times. min * CT .apprxeq. 13 .times. k 2
) ##EQU2##
[0019] The transistors in this circuit 40 will not see excessive
voltage or current, and 2N2222-type transistors may be used for
transistors Q1 and Q2. These transistors are setup in this circuit
40 to have a Vce of 500 mV to keep the transistors out of
saturation.
[0020] Resistor RE is selected based on voltage at RT (V.sub.RT)
and the 500 mV design requirement of Vce and fmax. The voltage at
resistor RT is determined by the ICs internal circuitry and can
typically be found in the data sheet. The voltage at RT is
determined by the ICs internal circuitry and can typically be found
in the data sheet. For circuit 40, V.sub.RT may be 3V. Resistor RT
is the equivalent resistance required to attain fmax, and IRT is
the current through resistor RT. I.sub.R1 and I.sub.RE are the
currents in resistors R1 and RE respectively. RT = 0.6 f .times.
.times. max * CT .apprxeq. 10.7 .times. k 3 ) I RT = V RT RT 4 ) I
R .times. .times. 1 = V RT R .times. .times. 1 5 ) I RE = I RT - I
R .times. .times. 1 6 ) RE = V RT - Vce I RE .apprxeq. 47.5 .times.
K 7 ) ##EQU3##
[0021] The components for the divider are formed by resistors RA
through RD. The peak voltage at the base of Q2 (V1) may be 3.2V,
and is calculated with the use of equation 8. Voltage Vbe for this
design may be roughly 0.6V. V1=V.sub.RT-Vce+Vbe 8)
[0022] The voltage divider resistors can be selected based on
equation 9 trying to minimize loss. Vinmin is the minimum RMS line
voltage of 85V for this illustrative embodiment. The rectified line
voltage is obtained by connecting to the output of diode bridge D3
in the schematic of FIG. 1. V .times. .times. 1 RD = Vin .times.
.times. min .times. 2 - V .times. .times. 1 RA + RB + RC 9 )
##EQU4##
[0023] To prevent frequency fs from varying with line amplitude,
components RF, R6A, R6B and Q1 are properly selected. The
determining factors on selecting these components are changes in
current though resistor RA caused by varying line amplitude
variations in the VFF signal. The changes in current through
resistor RA are denoted by variable I.sub.RA. The design of the
power converter determines the VFF signal. In this circuit 40,
signal VFF varies proportionally with the line voltage from 1.4 V
to 4.2 V. Resistor R6 is also determined by the design requirements
and is 30 K ohms. To implement the frequency dithering, resistor R6
in FIG. 1 is replaced with resistor R6A and R6B of FIG. 2. These
two resistors can be sized with equations 10 and 11. R .times.
.times. 6 .times. B = Vbe * R .times. .times. 6 VFF .times. .times.
min .apprxeq. 12.1 .times. .times. k 10 ) R .times. .times. 6
.times. A = R .times. .times. 6 - R .times. .times. 6 .apprxeq.
18.2 .times. k 11 ) ##EQU5##
[0024] Transistor Q1 and resistor RF form a controllable current
sink that suppresses the excess current caused by changing peak
line voltages. The resistor RF is selected based on the maximum VFF
voltage (VFFmax) and the change in peak current as the line varies
between the minimum (Vinmin) and the maximum (Vinmax) RMS input
voltages. The following equation 12 can be used to select RF for
proper current sinking. RF = VFF .times. .times. max .times. R
.times. .times. 6 .times. B R .times. .times. 6 .times. A + R
.times. .times. 6 .times. B - Vbe Vin .times. .times. max .times. 2
- Vin .times. .times. max .times. 2 RA + RB + RC .apprxeq. 3.72
.times. k 12 ) ##EQU6##
[0025] The circuit 40 in FIG. 4 is shown added to the circuitry 10
in FIG. 1 and the oscillator frequency dithers as a function of the
rectified sine wave input voltage and does not vary with the
changes in line amplitude. FIGS. 5 through 7 show the oscillator
frequency varying with rectified line voltage, where CH1 is the
rectified line voltage and CH2 is the PWM oscillator frequency. The
oscillator frequency varies with the input from 82 kHz to 100
kHz.
[0026] Advantageously, this frequency dithering circuit and method
takes advantage of the internal timing circuitry that is found in
most PWMs. This technique is a cost effective solution for lowering
narrowband EMI.
[0027] Though the invention has been described with respect to a
specific preferred embodiment, many variations and modifications
will become apparent to those skilled in the art upon reading the
present application. It is therefore the intention that the
appended claims be interpreted as broadly as possible in view of
the prior art to include all such variations and modifications.
* * * * *