U.S. patent application number 11/492523 was filed with the patent office on 2006-11-16 for noise cancelling method and apparatus.
Invention is credited to Takashi Kaku, Takahiro Kurakata, Hideo Miyazawa, Hiroyasu Murata.
Application Number | 20060256890 11/492523 |
Document ID | / |
Family ID | 18831626 |
Filed Date | 2006-11-16 |
United States Patent
Application |
20060256890 |
Kind Code |
A1 |
Kaku; Takashi ; et
al. |
November 16, 2006 |
Noise cancelling method and apparatus
Abstract
In a noise canceling method and an apparatus therefor which
notices the colored noise looked macroscopically, positively
cancels the dominant noise component in the low frequency band,
shifts the S/N value to plus, and can extract the reception signal
buried in the low frequency band and having a comparatively high
level, a signal in which a time axis, an amplitude, and a phase are
specified or a zero-point signal is inserted into a transmission
signal by an inserter, a noise component is interpolated by using
the specified signal or the zero-point signal by a noise canceler,
and an originally transmitted signal is regenerated by subtracting
the noise component from the reception signal.
Inventors: |
Kaku; Takashi; (Kawasaki,
JP) ; Miyazawa; Hideo; (Kawasaki, JP) ;
Kurakata; Takahiro; (Kawasaki, JP) ; Murata;
Hiroyasu; (Kawasaki, JP) |
Correspondence
Address: |
KATTEN MUCHIN ROSENMAN LLP
575 MADISON AVENUE
NEW YORK
NY
10022-2585
US
|
Family ID: |
18831626 |
Appl. No.: |
11/492523 |
Filed: |
July 25, 2006 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
09819395 |
Mar 28, 2001 |
7110465 |
|
|
11492523 |
Jul 25, 2006 |
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Current U.S.
Class: |
375/285 |
Current CPC
Class: |
H04L 27/2647 20130101;
H04B 1/1036 20130101; H04B 1/123 20130101 |
Class at
Publication: |
375/285 |
International
Class: |
H04B 15/00 20060101
H04B015/00 |
Foreign Application Data
Date |
Code |
Application Number |
Nov 27, 2000 |
JP |
2000-359949 |
Claims
1. A noise canceling method comprising the steps of: periodically
inserting a zero-point into a transmission signal, establishing
synchronization based on a received signal, extracting the
zero-point based on the established synchronization and
interpolating a noise component of the received signal by using the
zero-point, and subtracting the noise component from the received
signal.
2. The noise canceling method as claimed in claim 1, wherein one or
more zero-points are inserted at intervals of an integer number of
samples.
3. The noise canceling method as claimed in claim 2, wherein an
inserted number of the zero-points is determined by deciding a
signal quality on the reception side to be notified to the
transmission side.
4. The noise canceling method as claimed in claim 1 wherein a
transmission line of the received signal includes a transparent
transmission line.
5. The noise canceling method as claimed in claim 4, wherein the
transparent transmission line includes a Nyquist transmission
line.
6. The noise canceling method as claimed in claim 1 wherein an
automatic equalizing process is further performed so as to remove
an intersymbol interference at a former or latter stage of a noise
cancellation.
7. A noise canceling apparatus comprising: means periodically
inserting a zero-point into a transmission signal, means
establishing synchronization based on a received signal, means
extracting the zero-point based on the established synchronization
and interpolating a noise component of the received signal by using
the zero-point, and means subtracting the noise component from the
received signal.
8. The noise canceling apparatus as claimed in claim 7, wherein one
or more zero-points are inserted at intervals of an integer number
of samples.
9. The noise canceling apparatus as claimed in claim 8, wherein an
inserted number of the zero-points is determined by deciding a
signal quality on the reception side to be notified to the
transmission side.
10. The noise canceling apparatus as claimed in claim 7 wherein a
transmission line of the received signal includes a transparent
transmission line.
11. The noise canceling apparatus as claimed in claim 10, wherein
the transparent transmission line includes a Nyquist transmission
line.
12. The noise canceling apparatus as claimed in claim 7 wherein an
automatic equalizer is further provided for removing an intersymbol
interference at a former or latter stage of a noise
cancellation.
13. A noise canceling method comprising the steps of: receiving a
signal periodically including a zero-point, establishing
synchronization based on a received signal, extracting the
zero-point based on the established synchronization, interpolating
a noise component of the received signal by using the zero-point,
and subtracting the noise component from the received signal.
14. A noise canceling apparatus comprising: means receiving a
signal periodically including a zero-point, means establishing
synchronization based on a received signal, means extracting the
zero-point based on the established synchronization, means
interpolating a noise component of the received signal by using the
zero-point, and means subtracting the noise component from the
received signal.
Description
BACKGROUND OF THE INVENTION
Field of the Invention
[0001] The present invention relates to a noise canceling method
and an apparatus therefor, and in particular to a noise canceling
method and an apparatus therefor for faithfully taking out a signal
buried in a noise.
[0002] Such a noise canceling method and an apparatus therefor has
become remarkably necessary in various industrial fields as
described in the following.
[0003] Power-line carrier modem field which attempts to realize a
data transmission at a high speed under circumstances with many
noises such as in a power-line carrier;
[0004] CATV modem, ADSL modem, VDSL modem, 2.4 G wireless LAN,
wireless transmission field, optical transmission field, and the
like;
[0005] Magnetic disk or optical disk which attempts to realize a
high recording density by taking out such a signal as is buried in
the noise due to the transmission rate accelerated;
[0006] Semiconductor of accelerated multi valued transmission
technology;
[0007] Voice recognition, image compression, demodulation of bar
code scanner, and the like under noisy circumstances.
[0008] Hereinafter, the presence of such a noise will be described
by taking a power-line carrier modem as an example, while the same
applies to the other fields as mentioned above.
[0009] In a power system shown in FIG. 20, power in a distribution
substation 100 is firstly supplied to a pole transformer 103
through a 6.6 KV high voltage distribution line 102, and is further
supplied to a home 105 through a 100 V/200 V low voltage
distribution line 104.
[0010] Upon performing a power-line carrier communication, an
optical fiber (not shown) is set up parallel with the high voltage
distribution line 102 between an access node 101 in the
distribution substation 100 and a modem set up in the pole
transformer. Through the optical fiber, the communication between
the modem in the pole transformer 103 and the modem inserted into
the convenience outlet connected to an interior distribution line
106 in the home 105 is performed through the 100 V/200 V low
voltage distribution line 104.
[0011] In this case, as shown in FIGS. 21A-21C, the low voltage
distribution line 104 looks like an inductor of 1 .mu.H/m, as shown
in FIG. 21B, for a spectrum of a transmission signal TX shown in
FIG. 21A, and looks like an inductor of 150 .mu.H if the line
length is assumed to be 150 m.
[0012] Also, a service wire 107 connected to the low voltage
distribution line 104 looks like a capacitor of 75 pF/m, and looks
like a capacitor of 0.1125 .mu.F if a 50 m service wire is assumed
to be connected to the home 105. Not only the service wire, but
also various household electric appliances in the home 105 look
like a capacitive load (see FIG. 21B), since the capacitors for
canceling the noise are connected to AC 100V
[0013] Consequently, the portion between the utility pole where the
pole transformer 103 is placed and the convenience outlets in the
home looks like a low-pass filter (LPF), as shown in FIG. 21B, and
a reception signal RX greatly attenuates in a high frequency band,
as shown in FIG. 21C. Therefore, when arriving at the terminal
side, the high frequency band signal is buried in a noise N in the
worst case.
[0014] On the other hand, although the loss in the low frequency
band is not so large compared with the high frequency band, random
noises (white noises) from the household electric appliances such
as inverter appliances are extremely large. Therefore, the low
frequency band signals are also buried in the noise N as shown in
FIG. 21C, and the high-speed data communication can not be put into
practice, so that the solution has been demanded for a long
time.
DESCRIPTION OF THE RELATED ART
[0015] Hereinafter, the prior art technologies for which such
solutions have been proposed will be described over three
generations.
<1st Generation>
[0016] FM modulation method, FSK modulation method, PSK modulation
method, and the like said to be resistive to the noise were adopted
as a modulation method of a power-line carrier modem. However,
since the noise level of the actual power line was high, their
practical uses were limited to the applications for a low speed,
equal to or less than 1200 bps.
<2nd Generation>
[0017] The spread spectrum system was introduced. Since the spread
spectrum system was resistive to the noise, it was greatly expected
for practical use of the power-line carrier.
[0018] However, according to Shannon theory limit, the transmission
capacity decreases rapidly when the S/N value is minus or negative
(see FIG. 21C) subject to the white noise, and it is theoretically
impossible to realize the high-speed transmission. Accordingly, the
Shannon limit could not be exceeded, so that the communication of
100 kbps at the maximum could be performed or the communication was
disabled in the worst case.
<3rd Generation>
[0019] OFDM (Orthogonal Frequency Division Multiplexing) method has
appeared. The OFDM method adopts the multi carrier modulation
method, and is a technology which avoids the carrier bandwidth with
a number of noises without using it. Therefore, a large noise can
be avoided, resulting in a prospect of realizing enhanced
speed.
[0020] However, the inverter built-in rate of the household
electric appliances forming the main sources of the noises has been
increasing, and the reduction in the high frequency band
accompanied with the capacitive load has been also increasing. As a
result, although it has been possible to perform a low speed
communication depending on circumstances in the prior art
technology, it has been impossible to realize a higher speed
communication of several Mbps.
[0021] Thus, building the inverters in the household electric
appliances is a great trend, so that the noise due to the inverters
built in the household electric appliances has a tendency to
increase more and more hereafter. Also, with the increase of the
noise, the capacitive load has a tendency to increase for a noise
preventive measure.
[0022] Under such circumstances, the idea of avoiding the noise
like the solution of the 3rd generation is not enough for the
trend. Rather, for the solution of the 4th generation, the attempt
of positively facing the noise to cancel the noise, and realize a
high-speed data communication is important.
[0023] As shown in FIG. 21C, although the noises are little in the
high frequency band, the reception signal greatly attenuates by the
capacitive load, and becomes lower than the noise level. Although
the attenuation of the reception signal is not so large in the low
frequency band, the S/N value is still minus due to the noise from
the household electric appliances.
SUMMARY OF THE INVENTION
[0024] It is accordingly an object of the present invention to
provide a method and an apparatus for realizing a high-speed
transmission by canceling a noise in a low frequency where signal
attenuation is little even if in a state where an S/N value is
minus and by regenerating a buried reception signal.
[0025] Even though a low frequency band (noise component N1, signal
component S1) where the level of a noise N is high, as shown in
FIG. 1A, is cut to transmit data only by using a high frequency
band, the S/N value remains minus.
[0026] On the other hand, if a noise in a power line is carefully
observed in FIG. 21C and FIG. 1A, it is seen that there are a
number of noises emphasized in a low frequency band, in which if
being microscopically observed they are white noises while if being
macroscopically observed (from the entire frequency bandwidth),
they are colored noises. Namely, if being observed at any narrow
bandwidth over the entire frequency band, they are the same white
noises.
[0027] Accordingly, in the present invention, the colored noise
observed macroscopically in that way is noticed, and as shown in
FIG. 1B, the noise component N1 dominant in the low frequency band
is positively canceled to shift the S/N value to plus, so that the
extraction of a reception signal S buried in the low frequency
band, with a comparatively high level, is attempted.
[0028] Therefore, the prevent invention realizes a noise canceling
method and an apparatus therefor for interpolating a noise
component based on a signal (hereinafter, occasionally referred to
as specific signal) in which a time, an amplitude, and a phase are
specified which is included into a received signal, and for
canceling the noise component from the received signal.
[0029] Also, the present invention realizes a noise canceling
method and an apparatus therefor for periodically inserting a
zero-point into a signal on a transmission side, for interpolating
a noise component by using the zero-point on a reception side, and
for subtracting the noise component from a received signal.
[0030] Hereinafter, such a noise canceling method and an apparatus
therefor according to the present invention will be described
referring to the figures.
[0031] FIG. 2A shows a prior art transmission/reception system of a
signal, in which a transmission signal from a transmission signal
generator 32 is sent to a reception signal regenerator 33 through a
Nyquist transmission line 31 as a transparent transmission
line.
[0032] In the present invention, as shown in FIG. 2B, an inserter
(inserting portion) 1 of a specific signal or a zero-point
(hereinafter, generally referred to as zero-point) is provided
between the transmission signal generator 32 and the Nyquist
transmission line 31 in such a transmission/reception system, and a
noise canceler 2 is provided between the Nyquist transmission line
31 and the reception signal regenerator 33. It is to be noted that
the noise canceler 2 is composed of a frequency shift portion 3, a
decimator (DCM) (decimating portion) 4, an interpolater (IPL)
(interpolating portion) 5, a frequency reverse shift portion 6, and
a subtracter (subtracting portion) 7, as described later.
[0033] First of all, the symbol rate of the transmission signal
generated by the transmission signal generator 32 is assumed to be
e.g. 192 kB as shown in FIG. 3A. If such a transmission signal is
provided to the zero-point inserter 1, the zero-point inserter 1
inserts the zero-point, as shown in FIG. 3B, into the transmission
signal of FIG. 3A to be transmitted to the Nyquist transmission
line 31. If the signal S is also transmitted at the same rate, the
transmission rate assumes 384 kB.
[0034] The reception side, as shown in FIG. 3C, receives the
reception signal S and the zero-point on which the noise N of the
transmission line 31 is respectively put.
[0035] The noise canceler 2 cancels the signal S including the
noise N (S+N), and leaves only the noise N at the zero-point. Then,
as shown in FIG. 3D, a noise interpolation signal N' is generated
at each reception signal point from the noises N on both sides.
[0036] The noise canceler 2 further subtracts the noise
interpolation signal N' shown in FIG. 3D from the reception signal
shown in FIG. 3C, so that the noise N assumes N-N' as shown in FIG.
3E. Thus, the signal (corresponding to the transmission signal),
having substantially removed therefrom the noise, only composed of
the signal component S can be regenerated.
[0037] The operation of the noise canceler 2 will now be described
in more detail referring to FIGS. 4-6.
[0038] The above-mentioned transmission signal is firstly
transmitted at the rate of 192 kB as shown in FIG. 4A. The spectrum
in this case is shown by the scalar, in which the abscissa denotes
frequency bandwidth kHz, in the right of FIG. 4A.
[0039] When the zero-points are inserted into such a transmission
signal, the zero-points are to be inserted into signal points as
shown in FIG. 4B, so that the frequency bandwidth after the
insertion assumes 384 kB. In this case, a spectrum is copied around
+192 kHz can be obtained.
[0040] The reception signal at the time when such a transmission
signal into which the zero-points are inserted is transmitted to
the reception side assumes the noise components N being overlapped
with the signals S and the zero-points respectively, as shown in
FIG. 4C. The spectrum in this case is the same as that of the
transmission signal shown in FIG. 4B.
[0041] The operation at the time when the reception signal is sent
to the decimator 4 after being shifted by the frequency shift
portion 3 in the noise canceler 2 is shown in FIGS. 5A-5D.
[0042] Namely, a sample value and a spectrum of a reception signal
S(n) are as shown in FIG. 5A, and the Z transformation A of the
signal S(n) is expressed by the following equation:
A=S(z)=.SIGMA.S(n)z.sup.-n Eq.(1)
[0043] It is to be noted that the spectrum in the right of FIG. 5A
shows that the noises are distributed over 0-f.sub.s/2 (f.sub.s is
sample frequency) since the noises are added by the transmission
line 31. The Z transformation B of the inversion signal of the
reception signal S(n) is expressed by the following equation:
B=Z[(-1).sup.nS(n)]=S(-z) Eq.(2)
[0044] The inverted signal in this case has a coefficient
(-1).sup.n because the inversion is made only to the signal
component at the signal point.
[0045] The Z transformation C of a signal t(n) obtained after
adding the inversion signal (-1).sup.n*S(n) to the reception signal
S(n) shown in FIG. 5A is given by the following equation:
C=Z[t(n)]=T(z)=(1/2)[S(z)+S(-z)] Eq.(3)
[0046] Namely, the amplitude at the signal point becomes zero, so
that not only the signal component S but also the noise component N
overlapped with the signal S is removed. The signal t(n) in which
t(1), t(3),,,=0 is expressed by the following equation:
T(z)=.SIGMA.t(2n)*Z.sup.-2n Eq.(4)
[0047] A signal D after the signal point of the signal t(n)
obtained in this way shown in FIG. 5C is decimated is expressed by
the following equation: D=u(n)=T(z.sup.1/2) Eq.(5)
[0048] Since the transmission rate falls to 192 kB in this case,
the spectrum is aliased or folded as shown in FIG. 5D.
[0049] A final signal E=U(z) is given by the following equation:
E=[S(z.sup.1/2)+S(-z.sup.1/2)]/2 Eq.(6)
[0050] The thus obtained decimation signal u(n) provided to the
interpolater 5 shown in FIG. 2B would exhibit the operations shown
in FIGS. 6A and 6B.
[0051] Namely, the signal u(n) from the decimator 4 is only
composed of the noise component having the sample value and the
spectrum shown in FIG. 6A. The signal t(n) with the zero-point
inserted into the noise component has a sample value and a spectrum
such as shown in FIG. 6B, and the Z transformation A is expressed
by the following equation: A=(z)=.SIGMA.t(n)z.sup.-n Eq. (7)
[0052] Since t(1), t(3),,,=0, A=.SIGMA.t(2n)z.sup.-n=u(n)z.sup.-2n
Eq.(8)
[0053] Then, the following equation is obtained: T(z)=U(z.sup.2)
Eq. (9)
[0054] If the zero-points are interpolated with the noise
components N on their both sides in the signal T(z), the signal has
the same transmission rate as the reception signal S(n) shown in
FIG. 5A and has only the noise component.
[0055] Accordingly, by subtracting the interpolated signal from the
reception signal S(n), the transmission signal into which the
zero-points are inserted shown in FIG. 4B can be obtained.
[0056] It is to be noted that in order to obtain the transmission
signal shown in FIG. 4A the zero-points only have to be
decimated.
[0057] While in the above description, how the transmission signal
is regenerated on the reception side has been mentioned, FIG. 7
shows how the noise component is canceled by paying attention only
to the noise component.
[0058] Namely, when the transmission signal has the transmission
bandwidth of 192 kB (.+-.96 kB), and the zero-points are inserted
thereto, the bandwidth is doubled, so that the copied component is
generated to be sent to the Nyquist transmission line 31.
[0059] At the noise canceler 2, as shown in a noise distribution
characteristic {circle around (1)}, the noise distribution firstly
extends over .+-.192 kHz. The noise level is high especially in the
left half of the frequency bandwidth of -192-0 kHz as shown in
FIGS. 1A and 1B, and is low in the frequency bandwidth of 0-+192
kHz.
[0060] When the frequency shift portion 3 shifts the frequency by
+96 kHz in this state, a noise component A+B will be shifted by +96
kHz for the noise characteristic {circle around (1)}, as shown in a
noise characteristic {circle around (2)}. With this shifting, a
noise component D in the noise characteristic {circle around (1)}
will be aliased to -192 kHz-96 kHz. Thus, the noise bandwidth for
which the interpolation (interpolated prediction) is desired to be
performed is shifted to the interpolation bandwidth, thereby more
effectively canceling the noise.
[0061] It is to be noted that the shift amount of +96 kHz is only
one example for convenience' sake description.
[0062] If the decimation operation is performed by the decimator 5
in this state, the frequency becomes half. Therefore, the noise
component A is aliased in +96-+192 kHz, the noise component B is
aliased to -192--96 kHz, the noise component C is aliased to -96-0
kHz, and the noise component D is aliased to 0-+96 kHz. The
bandwidth where the aliased component becomes the least is selected
here.
[0063] If the interpolater 5 interpolates the zero-points and
performs a filter canceling of the noise components A+C and B+D on
both sides, the noise components A+C and B+D only between -96-+96
kHz remain as shown in a noise characteristic {circle around
(4)}.
[0064] If the interpolated noise components are shifted in the
reverse direction to the above-mentioned frequency shift, that is,
by -96 kHz, the noise components A+C and B+D only between -192-0
kHz remain as shown in a noise characteristic {circle around
(5)}.
[0065] Accordingly, the subtracter 7 subtracts such noise
components from the entire noise components shown in the
characteristic {circle around (1)} thereby completely canceling the
noise components A and B between -192-0 kHz as shown in a
characteristic {circle around (6)}. It is to be noted that although
the noise components C and D remain, their noise level is low, as
shown in FIG. 1B, so that the S/N value is not greatly
influenced.
[0066] The reception signal from which the noise is canceled in
that way is regenerated substantially corresponding to the
transmission signal.
[0067] It is to be noted that the reason for performing the
frequency shift as mentioned above is because the interpolation
bandwidth is set e.g. to the bandwidth where the most noises exist
(in low frequency band in this example) to select the high
frequency band with less noise for the aliased frequency
bandwidth.
[0068] While in the above-mentioned FIGS. 3 and 4, a case where one
zero-point is inserted between the signal points has been
mentioned, FIGS. 8A-8E show various patterns of the zero-point
insertion.
[0069] Namely, FIG. 8A shows the case where the zero-points are
inserted into every 4th signal S, whereby the interpolated noise
bandwidth assumes 96 kHz.
[0070] Also, FIG. 8B shows a case where the zero-points are
inserted into every 3rd signal S, whereby the interpolated noise
bandwidth assumes 128 kHz.
[0071] FIG. 8C shows a case where the zero-points are inserted into
every other signal in the same way as the above-mentioned example,
whereby the interpolated noise bandwidth assumes 192 kHz.
[0072] FIG. 8D shows an example in which two zero-points are
inserted between the signals S, whereby the interpolated noise
bandwidth assumes 256 kHz.
[0073] Furthermore, FIG. 8E shows an example in which three
zero-points are inserted between signals S, whereby the
interpolated noise bandwidth assumes 288 kHz.
[0074] By increasing the number of the zero-point as shown in FIGS.
8D and 8E, the noise canceling over a wider bandwidth is made
possible. Although the data transmission rate may decrease in some
cases in exchange for the increase of the noise proof, it becomes
possible to withstand worse circumstances.
[0075] Since an intersymbol interference increases upon passing the
narrow bandwidth for example, the noise is canceled while the
signal itself is partially canceled. In such a case, the system
parameter may be optimized in order to effectively cancel the noise
without decreasing the entire rate and attenuating the signal.
Alternatively, an equalizer may be inserted at the former stage of
the noise canceler.
[0076] Also, if the signal quality is judged on the reception side,
the inserted number of the zero-point is determined according to
the decision result, and the number is notified to the transmission
side, it becomes possible to adaptively change the zero-point
inserted number.
[0077] Moreover, the zero-point insertion may be performed on the
transmission side by using e.g. the PN (pseudo random) system.
Thus, the reception side can interpolate the random noise by the PN
system.
[0078] As an example of the PN system, the followings can be given:
[0079] 15 chips:1111010110010000 [0080] 31
chips:1111100110100100001010111011000
[0081] In this case, it is also possible to sequentially insert the
zero-points with the time axis being shifted like the MUSE system
as performed by the image compression method.
[0082] There are various other methods of the zero-point insertion.
The optimization may be performed according to the system
characteristic.
[0083] The interpolation 5 shown in FIG. 2B can perform the
interpolating operation by using various filter characteristics as
shown in FIGS. 9A-9C.
[0084] Namely, in the low-pass filter shown in FIG. 9A, the
interpolation bandwidth is made a transmission bandwidth. There is
no aliasing waveform outside the interpolation bandwidth in this
case. However, being composed of a transversal filter or the like,
the filter has a characteristic that the number of taps is large
and the cancelation range is small.
[0085] Moreover, in case of a cos-squired filter shown in FIG. 9B,
the interpolation bandwidth is made the Nyquist bandwidth, and the
filter has a characteristic that the number of taps is small, the
cancelation range is large, but the aliasing waveform occurs
outside the interpolated bandwidth.
[0086] Furthermore, in case of a cos filter shown in FIG. 9C, the
interpolation bandwidth is also made the Nyquist bandwidth, and the
filter has a characteristic that the number of taps is large, the
calculation amount is much, and the aliasing waveform occurs
outside the interpolation bandwidth.
[0087] Moreover, in the present invention, the above-mentioned
frequency shift amount may be automatically determined for the
frequency bandwidth by detecting the frequency bandwidth in which
the noise frequency component of the received signal is large.
[0088] Furthermore, an automatic equalizing process for removing
the intersymbol interference at the former or latter stage of the
noise cancelation may be performed.
BRIEF DESCRIPTION OF THE DRAWINGS
[0089] FIGS. 1A and 1B are graphs for illustrating a basic
principle of a noise canceling method and an apparatus therefor
according to the present invention;
[0090] FIGS. 2A and 2B are block diagrams comparing a basic
arrangement of the present invention with a prior art example;
[0091] FIGS. 3A-3E are diagrams showing a schematic operation of
the present invention;
[0092] FIGS. 4A-4C are diagrams showing in detail an operation on a
transmission side of the present invention;
[0093] FIGS. 5A-5D are diagrams illustrating a decimating operation
of the present invention;
[0094] FIGS. 6A and 6B are diagrams illustrating an interpolating
operation of the present invention;
[0095] FIG. 7 is a diagram showing a canceling process of a noise
component of the present invention;
[0096] FIGS. 8A-8E are diagrams showing various states of
zero-point insertion by the present invention;
[0097] FIGS. 9A-9C are diagrams showing interpolation filter
examples used in the present invention;
[0098] FIGS. 10A and 10B are block diagrams showing an embodiment
in which the present invention is applied to a modem;
[0099] FIG. 11 is a block diagram showing an embodiment of a noise
canceler used in the present invention;
[0100] FIG. 12 is a block diagram showing an embodiment of an
interpolater used in the present invention;
[0101] FIG. 13 is a block diagram showing an embodiment of a timing
extractor and a VCXO type PLL circuit used in the present
invention;
[0102] FIG. 14 is a waveform diagram of a timing extractor by the
present invention;
[0103] FIG. 15 is a block diagram showing an example of a
zero-point control system by the present invention;
[0104] FIG. 16 is a block diagram showing an arrangement of an
automatic frequency shift by the present invention;
[0105] FIG. 17 is a diagram showing an example of a noise decrease
by a frequency shift amount by the present invention;
[0106] FIGS. 18A-18C are diagrams of a frequency bandwidth showing
an example for obtaining a frequency shift amount to a noise
bandwidth desired to be canceled by the present invention;
[0107] FIG. 19 is a block diagram showing a modification, added
with an equalizer, of the present invention;
[0108] FIG. 20 is a schematic diagram for illustrating an
applicable field of the present invention; and
[0109] FIGS. 21A-21C are diagrams for illustrating prior art
problems. Throughout the figures, like reference numerals indicate
like or corresponding components.
DESCRIPTION OF THE EMBODIMENTS
[0110] FIGS. 10A and 10B show an embodiment of a modem using a
noise canceling method and an apparatus therefor according to the
present invention.
[0111] Namely, a scramble process is performed to a transmission
signal SD by a scrambler (SCR) 11, and a serial signal is converted
into a parallel signal. The parallel signal is converted from a
Gray code (G) of which the transmission signal is originally formed
into a Natural code (N) by a vector sum circuit 12. After a vector
sum calculation is performed corresponding to a vector difference
circuit 28 for detecting the phase at the reception side, a signal
generator 13 transmits the transmission signal as shown in FIGS. 3A
and 4A.
[0112] The zero-points are inserted into the transmission signal by
the zero-point inserter 1 according to the present invention, and
the waveform is shaped by a roll-off filter (ROF) 14. The output
signal of the roll-off filter 14 is modulated by a modulation
circuit (MOD) 15 and is further converted from the digital signal
into the analog signal by a D/A conversion circuit 16. Then, a
low-pass filter (LPF) 17 extracts a signal only in a low frequency
bandwidth including a frequency bandwidth (10 kHz-450 kHz) of a
power carrier wave to be transmitted to the transmission line.
[0113] When the transmission signal from the transmission line is
received through the reception line, only a predetermined frequency
bandwidth component (10-450 kHz for a power carrier modem) is
firstly extracted by a band-pass filter (BPF) 19, and then the
analog signal is restored to the digital signal by an A/D
conversion circuit 20.
[0114] The analog signal expressed in the digital form is
demodulated into the baseband signal by a demodulation circuit
(DEM) 21, so that the waveform is shaped by a roll-off filter
22.
[0115] The output of the roll-off filter 22 is sent to a timing
extractor 23 and a VCXO type PLL circuit 24, thereby extracting the
phase of the zero-point signal and providing a sampling timing
signal to the A/D converter 20.
[0116] The noise component of the transmission line is canceled by
the noise canceler 2 according to the present invention from the
output signal of the roll-off filter 22, an intersymbol
interference is removed by an equalizer (EQL) 25, and a phase
adjustment is performed by a carrier automatic phase controller
(CAPC) 26, so that a decision circuit (DEC) 27 further outputs a
signal component from which the noise is removed.
[0117] A vector difference (error) calculation by the Natural code,
opposite to the vector sum circuit 12, is performed by the vector
difference circuit 28, and the Natural code is restored to the Gray
code, so that the parallel Gray code is converted into a serial
signal by a descrambler (DSCR) 29 for the descramble process to be
outputted as a reception signal RD.
[0118] In addition, a transmission clock generation circuit
(TX-CLK) 18 provides a transmission clock to the zero-point
inserter 1 and the D/A converter 16, and distributes the same to
other portions. Also, on the reception side, a reception clock
generation circuit (RX-CLK) 30 extracts the reception clock to be
provided to the noise canceler 2 and the portions of the
receiver.
[0119] It is to be noted that the reception clock generation
circuit 30 only passes the zero-point signal extracted from the PLL
circuit 24. Also, the zero-point signal is a mere symbol timing
signal in the prior art example.
[0120] Also, the hatched portion of FIG. 10A corresponds to a
Nyquist transmission line 31 as a transparent transmission line.
The Nyquist transmission line, as shown in FIG. 1B, transmits
signals with the interval of the transmission signal points being
the Nyquist interval (384 kB).
[0121] FIG. 11 shows an embodiment of a noise canceler 2 shown in
FIG. 1A, which corresponds to the noise canceler 2 shown in FIG.
2B.
[0122] Namely, a received signal A (384 kB) is outputted as a
signal C whose frequency is shifted by a desired rotation vector
signal B by the frequency shift portion 3.
[0123] The signal C is sent to the decimator 4, where the signal is
converted into a signal D (192 kB) having only the noise component
shown in FIG. 5D, based on the zero-point signal (192 kB) extracted
from the PLL circuit 24 shown in FIG. 10A.
[0124] The signal D is sent to the interpolater 5 to be outputted
as a signal E (384 kB) interpolated by the filter process. Since
the signal E is sent to the frequency reverse shift portion 6 and
shifted toward the reverse direction to the rotation vector signal
B used by the frequency shift portion 3, the signal is rotated in
the reverse direction by a signal F composing a conjugated complex
number with the signal B to be outputted as a signal G. It is to be
noted that a delay circuit 8 is provided on the course in order
that the signal F is adjusted to the timing of the output signal of
the interpolater 5.
[0125] The output signal G of the frequency reverse shift portion 6
is subtracted from the received signal A by the subtracter 7 to
assume an output signal K. It is to be noted that a delay circuit 9
is also provided to the received signal A in order to adjust the
timing to the output signal of the interpolater 5 in this case.
[0126] Thus, the signal K that is the received signal A from which
the noise component is canceled is outputted.
[0127] FIG. 12 shows an embodiment of the interpolater 5 shown in
FIG. 11, which is composed of a zero-point inserter 51 and an
interpolation filter 52.
[0128] Namely, the zero-point inserter 51 inserts the zero-points
between the noises, as shown in FIG. 6B, with respect to the signal
D (192 kB) composed of only the noise component outputted from the
decimator 4 to be provided to the interpolation filter 52 as a
signal of 384 kB transmission bandwidth.
[0129] The interpolation filter 52 can be composed of a transversal
filter, which can compose various filters as shown in FIGS. 9A-9C
with a delay circuit 521 and filter coefficients C1-Cn of a
multiplication circuit 522. The interpolation signal E outputted
therefrom is outputted as a signal having a certain amplitude where
the noise component N' at each zero-point is interpolated by the
noise components N on both sides of the zero-point at the signal
shown in FIG. 3D.
[0130] FIG. 13 shows an embodiment of the timing extractor 23 and
the VCXO type PLL circuit 24 shown in FIG. 10A. The timing
extractor 23 is composed of a power calculation circuit (PVR) 231,
a band-pass filter 232, and a vectorizing circuit 233. The PLL
circuit 24 is composed of a comparator 241, a low-pass filter 242,
a secondary PLL circuit 243, a D/A conversion circuit 244, a VCXO
(Voltage Controlled Crystal Oscillator) circuit 245, and a
frequency divider 246.
[0131] Namely, the vector signal outputted from the roll-off filter
22 is squared by the power calculation circuit 231 to calculate the
power. The spectrum at this time is shown in FIG. 14, where the
line spectrum in the center of the photograph indicates the
zero-point signal of 192 kHz. Namely, since the zero-point is
periodically transmitted on the transmission side, the energy for
this section is zero, whereas it becomes possible to extract the
power spectrum according to the insertion degree of the
zero-point.
[0132] The power value thus obtained is passed through the
band-pass filter 232. Since the band-pass filter having the center
frequency of 192 kHz is used in this example, desired zero-point
signal information is outputted to the vectorizing circuit 233.
[0133] The vectorizing circuit 233 vectorizes the input signal by
synthesizing the input signal with a signal whose phase is
different by 90 degrees, and provides the same to the PLL circuit
24 as timing phase information.
[0134] In the PLL circuit 24, the timing phase information from the
vectorizing circuit 233 is firstly compared with the phase of a
reference point preliminarily known at the comparator 241. The
phase difference is filtered to include only a low component by the
low-pass filter 242, so that the controlled voltage of the VCXO 245
is controlled by the secondary PLL circuit 243 composed of two
integrators and the D/A conversion circuit 244.
[0135] After performing the frequency division at the frequency
divider 246, the phase information is fed back to the comparator
241 to be compared with the phase at the reference point. Thus, the
phase difference between the timing phase information from the
vectorizing circuit 233 and the reference point is pulled in or
nullified thereby enabling the extraction of the zero-point signal
whose synchronization is established. Also, the sample timing
signal to the A/D converter 16 is outputted from the VCXO circuit
245, and is finally fed back to the comparator 241 to compose a
phase locked loop.
[0136] As for the zero-point inserted into the transmission signal,
various embodiments as shown in FIGS. 8A-8E can be taken into
account in the above embodiment. Namely, it is not necessary to fix
this zero-point interval, but it is possible to control the
zero-point interval as shown in FIG. 15.
[0137] Namely, in the transmission/reception system where two
transmitters 34 and 38 are respectively connected to receivers 35
and 39 across full duplex transmission lines 31a and 31b, a
deciding portion 36 for deciding the signal quality by the output
signal from the receiver 35 is provided. If the decision result is
provided to a zero-point inserted number setting/notifying portion
37, which determines the zero-point inserted number, and notifies
the information to a zero-point inserted number setting/notifying
portion 41 through the transmission line 31b to set the zero-point
number of the transmitter 34.
[0138] Similarly, the signal quality is decided by a signal quality
deciding portion 40 provided in the same way as the signal quality
deciding portion 36, based on the signal received by the receiver
39 from the transmitter 38 through the transmission line 31b. If
the zero-point inserted number setting/notifying portion 41
determines the zero-point inserted number based on the decision
result and notifies the same to the transmitter 38, this
transmitter 38 also performs the insertion control of the
zero-point in the same way as the transmitter 34.
[0139] Accordingly, the zero-point interval as shown in FIGS. 8A-8E
can be adaptively changed based on the signal quality of the
transmission line.
[0140] The above-mentioned frequency shift portion 3, as shown in
FIG. 11, provides the fixed rotation vector signal B. However, it
is also possible to automatically change this vector signal.
[0141] FIG. 16 shows an arrangement of such an automatic frequency
shift, in which multipliers 42 and 43, decimators 44 and 45, FFT
calculators (operating unit) 46 and 47, and a shift amount
determining portion 48 are additionally provided in the noise
canceler 2 in FIG. 11.
[0142] In operation, carrier signals .DELTA.f1 and .DELTA.f2, where
the frequencies of the outputs from the roll-off filter 22 are
respectively shifted by 90 degrees mutually by the multipliers 42
and 43, are multiplied, and are respectively decimated to obtain
the signals of the rate 192 kB at the decimators 44 and 45. Then,
the signals are converted into the frequency signals by the FFT
calculators 46 and 47, and determines which frequency bandwidth has
the largest noise bandwidth at the frequency shift amount
determining portion 48, so that the determined frequency shift
amount is provided to the frequency shift portion 3.
[0143] It is to be noted that the reason for using two FFT
calculators is because the frequency bandwidth of the input signal
is 192 kB, a half of the whole bandwidth.
[0144] FIG. 17 shows a noise reduction state in case where the
frequency shift amount is variously shifted. In this case, it is
found that the reduction effect is the largest over the range of
128 kHz-224 kHz.
[0145] What kind of frequency shift is determined for a desired
noise canceling range will now be described referring to FIGS.
18A-18C.
[0146] Firstly in an example of the power-line carrier
communication, 165.2 kHz (165 kHz+0.24 kHz=165.24 kHz) is
prescribed for a special carrier AM modulation method, and 162 kHz
(132 kHz+30 kHz=162 kHz) is prescribed for a special carrier PM
modulation method. Therefore, in case the interpolation filter 52
(see FIG. 12) of the noise canceler 2 uses the cos-squared filter
with a roll-off rate 14.5% as shown in FIG. 9B, a noise canceling
range equal to or less than 174 kHz and equal to or more than 10
kHz (10 kHz-174 kHz) can be applied by considering this roll-off
rate, so that both methods can favorably coexist.
[0147] This is shown by the transmission bandwidth in FIG. 18A. In
the permitted transmission bandwidth 10 kHz-450 kHz, 230 kHz
corresponds to the center frequency, and the noise canceling range
10 kHz-174 kHz corresponds to the hatched portion. The baseband
signal bandwidth at the time when the signal of such a transmission
bandwidth is outputted from the roll-off filter 22 of the modem
shown in FIG. 10A is shown in FIG. 18B. In this case, the bandwidth
of .+-.192 kHz around 0 kHz can be obtained. Accordingly, 10
kHz-174 kHz in FIG. 18A corresponds to -220 kHz--56 kHz in FIG.
18B.
[0148] On the other hand, the frequency bandwidth processed by the
interpolation filter 52 is decimated to the half, that is 192 kHz
by the decimator 4. Therefore, considering the roll-off rate 14.5%
of the cos-squared filter, 164 kHz can be obtained, so that the
baseband assumes 164/2=.+-.82 kHz.
[0149] Accordingly, in order that the upper bound frequency +82 kHz
of the interpolation filter 52 accords with the upper bound
frequency -56 kHz of the noise bandwidth shown in FIG. 18B, the
shift of 56+82=138 kHz has only to be performed. In the example of
FIG. 7, +96 kHz is only used for convenience' sake description.
[0150] It is to be noted that in case the automatic frequency shift
as shown in FIG. 16 is performed, the noise bandwidth 10 kHz-174
kHz of FIG. 18A itself shifts.
[0151] Also, in the embodiment shown in Fig. OA, the noise canceler
2 of the present invention inputs the output signal of the roll-off
filter 22 as it is. However, if an equalizer (EQL) 33 is provided
on the output side (former stage of noise canceler 2) of a
demodulator/roll-off filter (DEM/ROF) 21 (corresponding to filter
22 in FIG. 10A) as in a modification shown in FIG. 19, and the
intersymbol interference is preliminarily removed, it becomes
possible to perform a more effective noise cancelation.
[0152] In this case, the equalizing process is divided so that the
equalizer 33 may perform a time equalization, for example, and the
equalizer 25 may perform a frequency equalization, for example.
[0153] Although the time equalization and the frequency
equalization are known for an equalizer in the ADSL field, a quite
large effect can be achieved by the time axis equalization
alone.
[0154] As described above, a noise canceling method and an
apparatus therefor according to the present invention is arranged
so that a signal in which a time axis, an amplitude, and a phase
are specified or a zero-point signal is inserted into a
transmission signal, a noise component is interpolated by using the
specified signal or the zero-point signal, and an originally
transmitted signal is regenerated by subtracting the noise
component from a received signal. Therefore, it becomes possible to
realize a high-speed data transmission by an effective noise
cancelation in a low frequency band with less signal attenuation,
even in circumstances with a number of noises, especially in a
state where the S/N value is minus such as in a power-line
carrier.
[0155] Also, a noise canceling method and an apparatus therefor
according to the present invention can be similarly applied not
only to the power-line carrier modem, but also to the CATV modem,
the ADSL modem, the VDSL modem, the 2.4 G wireless LAN, the
wireless transmission field, and the optical transmission
field.
[0156] Furthermore, the high recording density can be realized for
a signal buried in the noise due to the enhanced speed such as a
magnetic disk by accurately taking out the signal.
[0157] Moreover, a noise canceling method and an apparatus therefor
according to the present invention can be applied to a process of
taking out the signal from the noise in the multi-valued
transmission technology of semiconductors, and can contribute to
the enhanced speed of the semiconductor performance. Furthermore,
the application to the fields troubled by various noises such as in
the voice recognition, the image compression, the demodulation of
the bar code scanner, and the like is made possible.
* * * * *