U.S. patent application number 11/454171 was filed with the patent office on 2006-10-19 for analytical scanning evanescent microwave microscope and control stage.
This patent application is currently assigned to Intematix Corporation. Invention is credited to Fred Duewer, Chen Gao, Xiao-Dong Xiang, Hai Tao Yang.
Application Number | 20060231756 11/454171 |
Document ID | / |
Family ID | 40765916 |
Filed Date | 2006-10-19 |
United States Patent
Application |
20060231756 |
Kind Code |
A1 |
Xiang; Xiao-Dong ; et
al. |
October 19, 2006 |
Analytical scanning evanescent microwave microscope and control
stage
Abstract
A scanning evanescent microwave microscope (SEMM) that uses
near-field evanescent electromagnetic waves to probe sample
properties is disclosed. The SEMM is capable of high resolution
imaging and quantitative measurements of the electrical properties
of the sample. The SEMM has the ability to map dielectric constant,
loss tangent, conductivity, electrical impedance, and other
electrical parameters of materials. Such properties are then used
to provide distance control over a wide range, from to microns to
nanometers, over dielectric and conductive samples for a scanned
evanescent microwave probe, which enable quantitative non-contact
and submicron spatial resolution topographic and electrical
impedance profiling of dielectric, nonlinear dielectric and
conductive materials. The invention also allows quantitative
estimation of microwave impedance using signals obtained by the
scanned evanescent microwave probe and quasistatic approximation
modeling. The SEMM can be used to measure electrical properties of
both dielectric and electrically conducting materials.
Inventors: |
Xiang; Xiao-Dong; (Danville,
CA) ; Gao; Chen; (Hefei, CN) ; Duewer;
Fred; (Albany, CA) ; Yang; Hai Tao; (San Jose,
CA) |
Correspondence
Address: |
BUCHANAN, INGERSOLL & ROONEY PC
POST OFFICE BOX 1404
ALEXANDRIA
VA
22313-1404
US
|
Assignee: |
Intematix Corporation
Fremont
CA
|
Family ID: |
40765916 |
Appl. No.: |
11/454171 |
Filed: |
June 14, 2006 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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09608311 |
Jun 30, 2000 |
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11454171 |
Jun 14, 2006 |
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09158037 |
Sep 22, 1998 |
6173604 |
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11454171 |
Jun 14, 2006 |
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08717321 |
Sep 20, 1996 |
5821410 |
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09158037 |
Sep 22, 1998 |
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60141698 |
Jun 30, 1999 |
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60059471 |
Sep 22, 1997 |
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Current U.S.
Class: |
250/309 |
Current CPC
Class: |
G01Q 60/22 20130101 |
Class at
Publication: |
250/309 |
International
Class: |
G01N 23/00 20060101
G01N023/00 |
Goverment Interests
[0002] This invention was made with U.S. Government support under
Contract No. DE-AC03-76SF00098 between the U.S. Department of
Energy and the University of California for the operation of
Lawrence Berkeley Laboratory. The U.S. Government may have certain
rights in this invention.
Claims
1-53. (canceled)
54. A method for measuring an electromagnetic property of a sample
using an evanescent wave probe, the method comprising: measuring
probe parameters selected from the group consisting of resonant
frequency shift and quality factor shift, wherein the resonant
frequency shift and the quality factor shift results from an
interactin between the sample and an evanscent electromagnetic
field emitted from said probe.
55. The method for measuring an electromagnetic property according
to claim 54, wherein the measurement is made using quasistatic
approximation modeling.
Description
[0001] This application claims benefit of provisional application
Ser. No. 60/141,698 filed Jun. 30, 1999.
FIELD OF THE INVENTION
[0003] This invention relates generally to scanning probe
microscopy and more specifically to scanning evanescent
electromagnetic wave microscopy and/or spectroscopy.
BACKGROUND OF THE INVENTION
[0004] Quantitative dielectric measurements are currently performed
by using deposited electrodes on large length scales (mm) or with a
resonant cavity to measure the average dielectric constant of the
specimen being tested. Quantitative conductivity measurements of
the test specimen can only be accurately performed with a
four-point probe. A drawback associated with performing the
aforementioned measurements is that the probe tip used to measure
the dielectric and conductivity properties of the test specimen
often comes into contact with the test specimen. Repeated contact
between the probe tip and the test specimen causes damage to both
the probe tip and the specimen, thereby making the resulting test
measurements unreliable.
[0005] Another drawback associated with the aforementioned
measurements is that gap distance between the probe tip and the
test specimen can only be accurately controlled over a milli-meter
(mm) distance range.
SUMMARY OF THE INVENTION
[0006] The present invention allows quantitative non-contact and
high-resolution measurements of the complex dielectric constant and
conductivity at RF or microwave frequencies. The present invention
comprises methods of tip-sample distance control over dielectric
and conductive samples for the scanned evanescent microwave probe,
which enable quantitative non-contact and high-resolution
topographic and electrical impedance profiling of dielectric,
nonlinear dielectric and conductive materials. Procedures for the
regulation of the tip-sample separation in the scanned evanescent
microwave probe for dielectric and conducting materials are also
provided.
[0007] The present invention also provides methods for quantitative
estimation of microwave impedance using signals obtained by scanned
evanescent microwave probe and quasistatic approximation modeling.
The application of various quasistatic calculations to the
quantitative measurement of the dielectric constant, nonlinear
dielectric constant, and conductivity using the signal from a
scanned evanescent microwave probe are provided. Calibration of the
electronic system to allow quantitative measurements, and the
determination of physical parameters from the microwave signal is
also provided.
[0008] The present invention also provides methods of fast data
acquisition of resonant frequency and quality factor of a
resonator; more specifically, the microwave resonator in a scanned
evanescent microwave probe.
[0009] A piezoelectric stepper for providing coarse control of the
tip-sample separation in a scanned evanescent microwave probe with
nanometer step size and centimeter travel distances is
disclosed.
BRIEF DESCRIPTION OF THE DRAWINGS
[0010] FIGS. 1(A)-1(B) are schematic views of equivalent circuits
used for modeling the tip-sample interaction;
[0011] FIG. 2 is a schematic view of the tip-sample geometry;
[0012] FIG. 3 illustrates the infinite series of image charges used
to determine the tip-sample impedance;
[0013] FIG. 4 is a graph showing the agreement between the
calculated and measure frequency shifts for variation of the
tip-sample separation.
[0014] FIG. 5 illustrates the calculation of C.sub.tip-sample;
[0015] FIG. 6 is a schematic view of the setup used to measure the
nonlinear dielectric constant;
[0016] FIG. 7 shows images of topography and .di-elect
cons..sub.333 for a periodically poled single-crnstal LiNbO.sub.3
wafer;
[0017] FIG. 8 is a graph showing f.sub.r and .DELTA.(1/Q) as a
function of conductivity.
[0018] FIG. 9 illustrates the tip-sample geometry modeled;
[0019] FIG. 10 is a graph showing C.sub.r using the model
approximation;
[0020] FIG. 11 is a schematic view of the operation of the
microscope for simultaneous measurement of the topography and
nonlinear dielectric constant;
[0021] FIG. 12 shows images of topography and .di-elect
cons..sub.333 for a periodically poled single-crystal LiNbO.sub.3
wafer;
[0022] FIG. 13 is a graph for the C.sub.tip-sample calculation;
[0023] FIG. 14(a)-14(b) are two graphs showing the variation of the
derivative signal versus tip-sample variation;
[0024] FIG. 15 is a graph showing calibration for regulation of
tip-sample separation;
[0025] FIG. 16 illustrates the measurement of topographic and
resistivity variations;
[0026] FIG. 17 is a schematic view of the architecture of the data
acquisition and control electronics;
[0027] FIG. 18 is a flow chart for of the architecture of the
inventive data acquisition and control electronics;
[0028] FIG. 19 illustrates the design and operation of the
piezoelectric stepper;
[0029] FIG. 20 illustrates the sequence of motion of the
piezoelectric stepper;
[0030] FIG. 21 illustrates the integration of an AFM tip with
SEMM.
DETAILED DESCRIPTION OF THE INVENTION
Embodiment A
[0031] To determine quantitatively the physical properties, such as
the complex dielectric constant, nonlinear dielectric constant and
conductivity, through measurements of changes in resonant frequency
(f.sub.r)and quality factor (Q) as function of different materials,
bias electric and magnetic fields, tip-sample distance and
temperature, etc. by a scanned evanescent microwave probe (SEMP), a
quantitative model of the electric and magnetic fields in the
tip-sample interaction region is necessary. A number of quasistatic
models can be applied to the calculation of the probe response to
dielectric, nonlinear dielectric and conductive materials. For the
present invention, these models are applied to the calculation of
the complex dielectric constant, nonlinear dielectric constant, and
conductivity.
[0032] To determine the electrical properties of a sample, the
variation in resonant frequency (f.sub.r) and quality factor (Q) of
a resonant cavity is measured. (FIG. 1(A)). The tip-sample
interaction is modeled using the equivalent RLC circuit shown in
FIG. 1(B). FIG. 1(A) shows the novel evanescent probe structure
comprising a microwave resonator such as illustrated microwave
cavity 10 with coupling loops for signal input and output. The
sharpened metal tip 20, which, in accordance with the invention
acts as a point-like evanescent field emitter as well as a
detector, extends through a cylindrical opening or aperture 22 in
endwall 16 of cavity 10. Mounted immediately adjacent sharpened tip
20 is a sample 80. Cavity 10 comprises a standard quarter or half
wave cylindrical microwave cavity resonator having a central metal
conductor 18 with a tapered end 19 to which is attached sharpened
metal tip or probe 20. The tip-sample interaction appears as an
equivalent complex tip-sample capacitance (C.sub.tip-sample). Given
f.sub.r and Q, the complex tip-sample capacitance can be extracted.
.DELTA. .times. .times. f f 0 = - C r 2 .times. C ( 1 ) .DELTA.
.function. ( 1 Q ) = - ( 1 Q 0 + 2 .times. C i C r ) .times.
.DELTA. .times. .times. f f 0 ( 2 ) ##EQU1## where C tip - sample =
C r + iC l , .DELTA. .times. .times. f = f r - f 0 , .DELTA.
.function. ( 1 Q ) = 1 Q - 1 Q 0 , ##EQU2## and f.sub.0 and Q.sub.0
are the unloaded resonant frequency and quality factor. The
calculation of C.sub.tip-sample is described in greater detail
below. a) Modeling of the Cavity Response for Dielectric
Materials
[0033] To allow the quantitative calculation of the cavity response
to a sample with a certain dielectric constant, a detailed
knowledge of the electric and magnetic fields in the probe region
is necessary. FIG. 2 describes the tip-sample geometry. The most
general approach is to apply an exact finite element calculation of
the electric and magnetic fields for a time-varying three
dimensional region. This is difficult and time consuming,
particularly for the tip-sample geometry described in FIG. 2. Since
the tip is sharply curved, a sharply varying mesh size should be
implemented. Since the spatial extent of the region of the
sample-tip interaction is much less than the wavelength of the
microwave radiation used to probe the sample (.lamda..about.28 cm
at 1 GHz, .about.14 cm at 2 GHz), the quasistatic approximation can
be used, i.e. the wave nature of the electric and magnetic fields
can be ignored. This allows the relatively easy solution of the
electric fields inside the dielectric sample. A finite element
calculation of the electric and magnetic fields under quasistatic
approximation for the given tip-sample geometry can be applied. A
number of other approaches can be employed for the determination of
the cavity response with an analytic solution, which are much more
convenient to use. The calculation of the relation between complex
dielectric constant and SEMP signals for bulk and thin film
dielectric materials by means of an image charge approach is now
outlined.
Spherical Tip
[0034] i. Calculation of the Complex Dielectric Constant for Bulk
Materials.
[0035] The complex dielectric constant measured can be determined
by an image charge approach if signals from the SEMP are obtained.
By modeling the redistribution of charge when the sample is brought
into the proximity of the sample, the complex impedance of the
sample for a given tip-sample geometry can be determined with
measured cavity response (described below). A preferred model is
one that can have an analytic expression for the solution and is
easily calibrated and yields quantitatively accurate results. Since
the tip geometry will vary appreciably between different tips, a
model with an adjustable parameter describing the tip is required.
Since the region close to the tip predominately determines the
sample response, the tip as a metal sphere of radius R.sub.0 can be
modeled.
[0036] FIG. 3 illustrates the infinite series of image charges used
to determine the tip-sample impedance. For dielectric samples, the
dielectric constant is largely real (.di-elect
cons..sub.i/.di-elect cons..sub.r<0.1), where .di-elect
cons..sub.r and .di-elect cons..sub.i are the real and imaginary
part of the dielectric constant of the sample, respectively.
Therefore, the real portion of the tip-sample capacitance, C.sub.r,
can be calculated directly and the imaginary portion of the
tip-sample capacitance, C.sub.i, can be calculated by simple
perturbation theory.
[0037] Using the method of images, the tip-sample capacitance is
calculated by the following equation: C r = 4 .times. .times. .pi.
.times. .times. 0 .times. R 0 .times. n = 1 .infin. .times. bt n a
1 + a n , ( 3 ) ##EQU3## where t.sub.n and a.sub.n have the
following iterative relationships: a n = 1 + a ' - 1 1 + a ' + a n
- 1 .times. .times. .times. and ( 4 ) t n = bt n - 1 1 + a ' + a n
- 1 ( 5 ) ##EQU4## with a 1 = 1 + a ' , t 1 = 1 , b = - 0 + 0 , and
.times. .times. a ' = d R , ##EQU5## where .di-elect cons. is the
dielectric constant of the sample, .di-elect cons..sub.0 is the
permittivity of free space, d is the tip-sample separation, and R
is the tip radius.
[0038] This simplifies to: C r = 4 .times. .times. .pi. .times.
.times. 0 .times. R 0 .function. [ ln .function. ( 1 - b ) b + 1 ]
( 6 ) ##EQU6## as the tip-sample gap approaches zero.
[0039] Since the dielectric constant of dielectric materials is
primarily real, the loss tangent (tan .delta.) of dielectric
materials can be determined by perturbation theory. The imaginary
portion of the tip-sample capacitance will be given by:
C.sub.i=C.sub.r tan .delta.. (7)
[0040] Given the instrument response, C.sub.tip-sample, the complex
tip-sample capacitance. and therefore the complex dielectric
constant of the sample can be estimated.
[0041] FIG. 4 illustrates agreement between the calculated and
measure frequency shifts for variation of the tip-sample
separation. The dielectric constant and loss tangent can be
determined from C.sub.tip-sample by a number of methods. One simple
approach is to construct a look-up table which yields the
dielectric constant corresponding to a given C.sub.tip-sample.
Alternatively, it can be directly calculated from the signals using
the formula. Table 1 shows a comparison between measured and
reported values for dielectric constant and loss tangent.
[0042] The perturbed electric field inside the sample is: E
.fwdarw. 1 .function. ( , d ) = q 2 .times. .pi. .function. ( + 0 )
.times. n = 1 .infin. .times. t n .times. r .times. e _ r + ( z + a
n ) .times. e _ z [ r 2 + ( z + a n ) 2 ] 3 / 2 ( 8 ) ##EQU7##
where q=4.pi..di-elect cons..sub.0RV.sub.0, V.sub.0 is the voltage,
and {right arrow over (e)}.sub.r and {right arrow over (e)}.sub.z
are the unit vectors along the directions of the cylindrical
coordinates r and z, respectively.
[0043] ii. Calculation of the Complex Dielectric Constant for Thin
Films.
[0044] The image charge approach can be adapted to allow the
quantitative measurement of the dielectric constant and loss
tangent of thin films. FIG. 5 illustrates the calculation of
C.sub.tip-sample. In the strict sense, the image charge approach
will not be applicable to thin films due to the divergence of the
image charges shown in FIG. 5. However, if the contribution of the
substrate to the reaction on the tip can be modeled properly, the
image charge approach is still a good approximation. According to
the present invention, it is expected that all films can be
considered as bulk samples if the tip is sharp enough since the
penetration depth of the field is only about R. The contribution
from the substrate will decrease with increases in film thickness
and dielectric constant. This contribution can be modeled by
replacing the effect of the reaction from the complicated image
charges with an effective charge with the following format: b eff =
b 20 + ( b 10 - b 20 ) .times. exp .function. ( - 0.18 .times. a 1
- b 20 ) , ( 9 ) ##EQU8## where b 20 = 2 - 1 2 + 1 , b 10 = 1 - 0 1
+ 0 , ##EQU9## .di-elect cons..sub.2 and .di-elect cons..sub.1 are
the dielectric constants of the film and substrate, respectively, a
= d R , ##EQU10## and d is the thickness of the film. This format
reproduces the thin and thick film limits for the signal. The
constant 0.18 was obtained by calibrating against interdigital
electride measurements at the same frequency on SrTiO.sub.3 thin
film. Following a similar process to the previous derivation,
yields: C r = 4 .times. .pi. .times. .times. 0 .times. R .times. n
= 1 .infin. .times. m = 0 .infin. .times. b eff n - 1 .times. b 21
m .times. b 10 m .function. ( b 20 n + 1 + 2 .times. mna - b 21 n +
1 + 2 .times. ( m - 1 ) .times. na ) , ( 10 ) C i = 4 .times. .pi.
.times. .times. 0 .times. R .times. n = 1 .infin. .times. m = 0
.infin. .times. b eff n - 1 .times. b 21 m .times. b 10 m
.function. ( tan .times. .times. .delta. 2 .function. ( b 20 n + 1
+ 2 .times. .times. mna - b 21 ( n + 1 + 2 .times. ( m + 1 )
.times. na ) ) + 2 .times. 1 .times. 2 .times. tan .times. .times.
.delta. 1 ( 2 + 1 ) .times. ( 2 + 0 ) .times. b 21 ( n + 1 + 2
.times. ( m + 1 ) .times. na ) ) , ( 11 ) ##EQU11## where b 21 = 2
- 1 2 + 1 , ##EQU12## tan .delta..sub.2 and tan .delta..sub.1 are
the tangent losses of the film and substrate. Table 2 lists the
results of thin film measurements using the SEMP and interdigital
electrodes at the same frequency (1 GHz). b) Calculation of the
Nonlinear Dielectric Constant
[0045] The detailed knowledge of the field distribution in Eqn. 8
allows quantitative calculation of the nonlinear dielectric
constant. The component of the electric displacement D
perpendicular to the sample surface is given by:
D.sub.3=P.sub.3+.di-elect
cons..sub.33(E.sub.l+E.sub.m)+1/2.di-elect
cons..sub.333(E.sub.l+E.sub.m).sup.2+1/6.di-elect
cons..sub.333(E.sub.l+E.sub.m).sup.3+ . . . (12) where D.sub.3 is
the electric displacement perpendicular to the sample surface,
P.sub.3 is the spontaneous polarization, .di-elect cons..sub.ij,
.di-elect cons..sub.ijk, .di-elect cons..sub.ijkl, . . . are the
second-order (linear) and higher order (nonlinear) dielectric
constants, respectively.
[0046] Since the field distribution is known for a fixed tip-sample
separation, an estimate of the nonlinear dielectric constant from
the change in resonance frequency with applied voltage can be made.
For tip-sample separations much less than R, the signal mainly
comes from a small region under the tip where the electric fields
(both microwave electric field E.sub.m and low frequency bias
electric field E.sub.l) are largely perpendicular to the sample
surface. Therefore, only the electric field perpendicular to the
surface needs to be considered.
[0047] From Eqn. 12, the effective dielectric constant with respect
to E.sub.m can be expressed as a function of E.sub.l: 33 .function.
( E 1 ) = .differential. D 3 .differential. E m = 33 + 333
.function. ( E l + E m ) + 1 2 .times. 3333 .function. ( E l + E m
) 2 + .times. , ( 13 ) ##EQU13## and the corresponding dielectric
constant change caused by E.sub.l is: .DELTA..di-elect
cons.=.di-elect cons..sub.333E.sub.l+1/2.di-elect
cons..sub.3333E.sub.l.sup.2+ . . . (14)
[0048] The change f.sub.r for a given applied electric field,
E.sub.l is related to the change in the energy stored in the
cavity. Since the electric field for a given dielectric constant is
known and the change in the dielectric constant is small, this can
be calculated by integrating over the sample: df r f r = - .intg. V
s .times. .DELTA. .times. .times. .times. .times. E m 2 .times. d V
.intg. V t .times. ( .times. .times. E 0 2 + .mu. .times. .times. H
0 2 ) .times. d V .times. .times. = - .times. .intg. V s .times. (
333 .times. E l + 1 .times. 2 .times. .times. 3333 .times. .times.
E 1 2 + ) .times. .times. E m 2 .times. d V .times. .intg. V t
.times. ( .times. .times. E 0 2 + .mu. .times. .times. H 0 2 )
.times. .times. d V ( 15 ) ##EQU14## where V.sub.s is the volume of
the sample containing electric field, H.sub.0 is the microwave
magnetic field. and V.sub.l is the total volume containing electric
and magnetic fields, possibly with dielectric filling of dielectric
constant .di-elect cons.. E.sub.m is given by Eqn. 8. The
application of a bias field requires a second electrode located at
the bottom of the substrate. If the bottom electrode to tip
distance is much larger than tip-sample distance and tip radius,
Eqn. 8 should also hold for E.sub.l. The upper portion can be
calculated by integrating the resulting expression. The lower
portion of the integral can be calibrated by measuring the
dependence of f.sub.r versus the tip-sample separation for a bulk
sample of known dielectric constant. If the tip-sample separation
is zero, the formula can be approximated as: C r .function. ( V ) =
C r .function. ( V = 0 ) + 4 .times. .pi. .times. .times. 0 .times.
R .times. 1 32 .times. A 33 .times. V R .times. 33 + 0 2 .times.
.times. 0 .times. 333 ( 16 ) ##EQU15## where V is the low frequency
voltage applied to the tip. This calculation can be generalized in
a straightforward fashion to consider the effects of other
nonlinear coefficients and thin films.
[0049] FIG. 6 illustrates the setup used to measure the nonlinear
dielectric constant. To measure .di-elect cons..sub.333, an
oscillating voltage V.sub..OMEGA., of frequency f.sub..OMEGA., is
applied to the silver backing of the sample and the output of the
mixer is monitored with a lock-in amplifier (SR 830). This bias
voltage will modulate the dielectric constant of a nonlinear
dielectric material at f.sub..OMEGA.. By measuring f.sub.r and the
first harmonic variation in the phase output simultaneously, sample
topography and .di-elect cons..sub.333 can be measured
simultaneously.
[0050] FIG. 7 shows images of topography and .di-elect
cons..sub.333 for a periodically poled single-crystal LiNbO.sub.3
wafer. The crystal is a 1 cm.times.1 cm single crystal substrate,
poled by periodic variation of dopant concentration. The poling
direction is perpendicular to the plane of the substrate. The
dielectric constant image is essentially featureless, with the
exception of small variations in dielectric constant correlated
with the variation in dopant concentration. The nonlinear image is
constructed by measuring the first harmonic of the variation in
output of the phase detector using a lock-in amplifier. Since
.di-elect cons..sub.ijk reverses when the polarization switches,
the output of the lock-in switches sign when the domain direction
switches. The value (-2.4.times.10.sup.-19 F/V) is within 20% of
bulk measurements. The nonlinear image clearly shows the
alternating domains.
c) Calculation of the Conductivity
[0051] i. Low Conductivity
[0052] The dielectric constant of a conductive material at a given
frequency f may be written as: = r + 2 .times. .times. i .times.
.times. .sigma. f , ##EQU16## where .di-elect cons..sub.r is the
real part of the permittivity and .sigma. is the conductivity. The
quasistatic approximation should be applicable when the wavelength
inside the material is much greater (>>) than the tip-sample
geometry. For R.sub.0.about.1 um and .lamda..apprxeq.14 cm, max
.apprxeq. ( .lamda. R 0 ) 2 .apprxeq. 2 10 10 .times. .times.
.times. or .times. .times. .times. .sigma. max .apprxeq. f .times.
.times. max 2 .apprxeq. 2 10 7 .times. 1 .OMEGA. - cm . ( 17 )
##EQU17##
[0053] For .sigma.<<.sigma..sub.max, the quasistatic
approximation remains valid. C.sub.tip-sample can be calculated by
the method of images. Each image charge will be out of phase with
the driving voltage. By calculating the charge (and phase shift)
accumulated on the tip when it is driven by a voltage V, frequency
f, one can calculate a complex capacitance. For moderate tip-sample
separations, it is primarily a capacitance with a smaller real
component.
[0054] Using the method of images, we find that the tip-sample
capacitance is given by: C tip - sample = 4 .times. .pi. .times.
.times. 0 .times. R 0 .times. n = 1 .infin. .times. bt n a 1 + a n
, ( 18 ) ##EQU18## where t.sub.n and a.sub.n have the following
iterative relationships: a n = 1 + a ' - 1 1 + a ' + a n - 1
.times. .times. and ( 19 ) t n = bt n - 1 1 + a ' + a n - 1 ( 20 )
##EQU19## with a.sub.l a 1 = 1 + a ' , .times. t 1 = 1 , .times. b
= - 0 + 0 , ##EQU20## and a ' = d R , ##EQU21## where .di-elect
cons.=.di-elect cons..sub.r-i.di-elect cons..sub.i is the complex
dielectric constant of the sample, .di-elect cons..sub.0 is the
permittivity of free space, d is the tip-sample separation, and R
is the tip radius.
[0055] By expressing b=b.sub.r+ib.sub.i=|b|.di-elect
cons..sub.i.phi., the real and complex capacitances can be
separated. C r = 4 .times. .pi. .times. .times. 0 .times. R 0
.times. n = 1 .infin. .times. b n .times. cos .function. ( n
.times. .times. .phi. ) .times. g n a 1 + a n .times. .times. and (
21 ) C i = 4 .times. .times. .pi. .times. .times. 0 .times. R 0
.times. n = 1 .infin. .times. b n .times. sin .function. ( n
.times. .times. .phi. ) .times. g n a 1 + a n , .times. where
.times. .times. g 1 = 1 .times. .times. and .times. .times. .times.
g n .times. .times. is .times. .times. given .times. .times. by
.times. : ( 22 ) g n = g n - 1 1 + a ' + a n - 1 . ( 23 ) ##EQU22##
[0056] This calculation can be generalized in a straightforward
fashion to thin films. FIG. 8 illustrates f.sub.r and .DELTA.(1/Q)
as a function of conductivity. The curve peaks approximately where
the imaginary and real components of .di-elect cons. become equal.
For those plots, it is assumed that the complex dielectric constant
was given by = 10 + I .times. 2 .times. .sigma. f . ##EQU23##
[0057] ii. High Conductivity
[0058] For .sigma.>.sigma..sub.max, the magnetic field should
also be considered. The real portion of C.sub.tip-sample can be
derived using an image charge approach. This is identical to
letting b=1. C tip - sample = 4 .times. .pi. 0 .times. R 0 .times.
n = 1 .infin. .times. t n a 1 + a n , ( 24 ) ##EQU24## where
t.sub.n and a.sub.n have the following iterative relationships: a n
= 1 + a ' - 1 1 + a ' + a n - 1 .times. .times. and ( 25 ) t n = t
n - 1 1 + a ' + a n - 1 ( 26 ) ##EQU25## with a 1 = 1 + a ' ,
.times. t 1 = 1 , .times. b = - 0 + 0 , .times. and ##EQU26## a ' =
d R , ##EQU26.2## where .di-elect cons. is the dielectric constant
of the sample, .di-elect cons..sub.0 is the permittivity of free
space, d is the tip-sample separation, and R is the tip radius.
[0059] In this limit, the formula can also be reduced to a sum of
hyperbolic sines; see E. Durand, Electrostatique, 3 vol. (1964-66).
C r = 4 .times. .pi. 0 .times. R 0 .times. sinh .times. .times.
.alpha. .times. n = 2 .infin. .times. 1 sinh .times. .times. n
.times. .times. .alpha. ( 27 ) ##EQU27## where .alpha.=cos
h.sup.-1(1+.alpha.').
[0060] The magnetic and electric fields at the surface of the
conductor are given by: E _ s .function. ( r ) = R 0 2 .times. .pi.
0 .times. n = 1 .infin. .times. a n ' .times. q n [ r 2 + ( a n '
.times. R 0 ) 2 ] 3 / 2 .times. e _ z ( 28 ) H _ s .function. ( r )
= - I .times. .omega. 2 .times. .pi. .times. .times. r .times. n =
1 .infin. .times. q n .times. [ r 2 + ( a n ' .times. R 0 ) 2 ] 1 /
2 - a n ' .times. R 0 [ r 2 + ( a n ' .times. R 0 ) 2 ] 1 / 2
.times. e _ .PHI. ( 29 ) ##EQU28## The knowledge of this field
distribution allows the calculation of the loss in the conducting
sample. Hyperbolic Tip
[0061] For tip-sample separations <R.sub.0, a spherical tip
turns out to be an excellent approximation, but the approximation
does not work well for tip-sample separations >R.sub.0. To
increase the useful range of the quasistatic model, additional
modeling is performed to accurately model the electric field over a
larger portion of the tip. An exact solution exists for a given
hyperbolic tip at a fixed distance from a conducting plane. The
potential is: .PHI. = .PHI. 0 .times. log .function. [ 1 + v 1 - v
] log .function. [ 1 + v 0 1 - v 0 ] , ( 30 ) ##EQU29## The surface
charge/unit area is: .sigma. = .times. 1 4 .times. .pi. .times. E
.times. | v 0 = .times. - 1 4 .times. .pi. .times. 2 .times. .PHI.
0 a .function. ( 1 - v 0 2 ) .times. 1 log .function. [ 1 + v 0 1 -
v 0 ] .function. [ 1 - v 0 2 u 2 + 1 - v 0 2 ] 1 / 2 = .times. - 1
4 .times. .pi. .times. 2 .times. .PHI. 0 a .function. ( 1 - v 0 2 )
.times. 1 log .function. [ 1 + v 0 1 - v 0 ] [ 1 y 2 a 2 .function.
( 1 - v 0 2 ) 2 + 1 ] 1 / 2 ( 31 ) ##EQU30## The area/dy is: y = au
.function. ( 1 - v 0 2 ) 1 / 2 .times. .times. x = av 0 .function.
( 1 + u 2 ) 1 / 2 .times. .times. d x d y = av 0 .times. y a 2
.function. ( 1 - v 0 2 ) [ 1 + y 2 a 2 .function. ( 1 - v 0 2 ) ] -
1 / 2 , ( 32 ) d A d y = 2 .times. .pi. .times. .times. y
.function. ( 1 + ( d x d y ) 2 ) 1 / 2 = 2 .times. .pi. .times.
.times. y [ 1 + y 2 a 2 .function. ( 1 - v 0 2 ) 2 1 + y 2 a 2
.function. ( 1 - v 0 2 ) ] 1 / 2 , ( 33 ) ##EQU31## So, for a given
v.sub.0 and .alpha., the charge on a hyperbolic tip between
y.sub.min and y.sub.max is given by: Q = .intg. y .times. .times.
min y .times. .times. max .times. .sigma. .times. d A d y .times.
.times. d y = .PHI. 0 a .function. ( 1 - v 0 2 ) .times. 1 log
.function. [ 1 + v 0 1 - v 0 ] .times. .intg. y .times. min y
.times. max .times. .times. d yy [ 1 + y 2 a 2 .function. ( 1 - v 0
2 ) ] - 1 / 2 = .PHI. 0 .times. a .times. .times. 1 log .function.
[ 1 + v 0 1 - v 0 ] .function. [ 1 + y 2 a 2 .function. ( 1 - v 0 2
) ] 1 / 2 .times. | y min y .times. max ( 34 ) ##EQU32## Given
proper choice of the limits of integration and tip parameters, Eq.
34 may be used to more accurately model the tip-sample capacitance
as detailed below.
[0062] iii. Large Tip-Sample Separations
[0063] The long-range dependence is modeled by calculating
C.sub.tip-sample. For large tip-sample separations, there is no
problem. (separations roughly>tip radius). The capacitance is
calculated as the sum of the contribution from a cone and a
spherical tip. For the cone, the charge only outside the tip radius
is considered. This solution can be approximately adapted to a
variable distance. FIG. 9 describes the tip-sample geometry
modeled; where:
[0064] Tip radius: R.sub.0
[0065] Opening angle: .theta.
[0066] Wire radius: R.sub.wire
[0067] Tip-sample separation: x.sub.0
[0068] To approximate the contribution of the conical portion of
the tip to the tip-sample capacitance, the conical portion of the
tip is divided into N separate portions, each portion n extending
from y.sub.n-1 to y.sub.n. For each portion, the hyperbolic
parameters (a, v.sub.0) are found for which the hyperbola
intersects and is tangent to the center of the portion.
[0069] Given points x, y on a hyperbola, and the slope s, find the
hyperbola intersecting and tangent to point (x, y). x = av
.function. ( u 2 + 1 ) 1 / 2 ( 35 ) y = au .function. ( 1 - v 2 ) 1
/ 2 ( 36 ) ##EQU33##
[0070] Eliminate u. x = av ( 1 + y 2 ( 1 - v 2 ) .times. a 2 ) 1 /
2 ( 37 ) d x d y = av .times. y .times. ( 1 - v 2 ) .times. a 2
.times. ( 1 + y 2 ( 1 - v 2 ) .times. a 2 ) - 1 / 2 ( 38 )
##EQU34##
[0071] Eliminate a.
[0072] From derivative equation: a 2 = y 2 s 2 .times. ( v 2
.function. ( 1 + s 2 ) - s 2 ( 1 - v 2 ) 2 ) ( 39 ) ##EQU35##
Substitute into equation for hyperbola: x 2 v 2 - y 2 1 - v 2 =
.times. a 2 = .times. y 2 s 2 .times. ( v 2 .function. ( 1 + s 2 )
- s 2 ( 1 - v 2 ) 2 ) .times. ( x 2 .times. ( 1 - v 2 ) ) 2 - y 2
.times. v 2 .function. ( 1 - v 2 ) = .times. y 2 .function. ( v 4 -
v 2 ) + y 2 .times. v 4 s 2 ( 40 ) v = xs xs + y , ( 41 ) ##EQU36##
get a from above. Finally, the charges accumulated on each portion
of the tip are summed and a capacitance is obtained. The
contribution from each conical portion of the tip is given
approximately by: C n .times. .times. cone .times. - .times. sample
= a .times. .times. 1 log .function. [ 1 + v 0 1 - v 0 ] [ 1 + y 2
a 2 .function. ( 1 - v 0 2 ) ] 1 / 2 .times. | y n - 1 y n . , ( 42
) ##EQU37## where C.sub.n cone-sample is the contribution to the
tip-sample capacitance from the n th portion of the cone. Assuming
N equally spaced cone portions, for the sphere+cone tip modeled,
here: y n = x 0 + R 0 + cot .times. .times. .theta. .function. ( n
.times. .times. R wire - R 0 N ) ##EQU38## The total tip-sample
capacitance is then given by the sum of the portions of the cone
and the spherical portion of the tip, C.sub.sphere-sample, as given
by Eq. 24. (C.sub.sphere-sample is substituted for C.sub.tip-sample
to reduce confusion.) C tip .times. - .times. sample = C sphere
.times. - .times. sample + n = 1 N .times. C c .times. .times. cone
.times. - .times. sample ##EQU39##
[0073] For small tip-sample separations, this model does not work
well. So the capacitance is calculated using the spherical model
(which dominates) and the line tangent to the contribution from the
cone is made. FIG. 10 shows C.sub.r using this approximation.
Embodiment B
[0074] The above described models are applied to the regulation of
the tip-sample separation for dielectric and conductive materials.
In principle, with above models, the relationship between tip
sample distance, electrical impedance and measured signals (f.sub.r
and Q as function of sample difference, bias fields and other
variables) is known precisely, at least when the tip is very close
to the sample. If measured f.sub.r and Q signal points (and their
derivatives with respect to electric or magnetic fields, distance
and other variables) are more than unknown parameters, the unknowns
can be uniquely solved. If both tip-sample distance and electrical
impedance can be determined simultaneously, then the tip-sample
distance can be easily controlled, so that the tip is always kept
above the sample surface with a desired gap (from zero to microns).
Both topographic and electrical impedance profiles can be obtained.
The calculation can be easily performed by digital signal processor
or any computer in real time or after the data acquisition.
[0075] Since the physical properties are all calculated from the fr
and Q and their derivatives, the temperature stability of the
resonator is crucial to ensure the measurement reproducibility. The
sensitivity very much depends on the temperature stability of the
resonator. Effort to decrease the temperature variation of
resonator using low thermal-coefficient-ceramic materials to
construct the resonator should be useful to increase the
sensitivity of the instrument.
d) Tip-Sample Distance Control for Dielectric Materials
[0076] i.) For Samples of Constant Dielectric Constant, the
Tip-Sample Separation can be Regulated by Measurement of
f.sub.r.
[0077] Other physical properties, i.e. nonlinear dielectric or loss
tangent can be measured simultaneously.
[0078] FIG. 11 illustrates the operation of the microscope for
simultaneous measurement of the topography and nonlinear dielectric
constant. From the calibration curve of resonant frequency versus
tip-sample separation, a reference frequency f.sub.ref is chosen to
correspond to some tip-sample separation. To regulate the
tip-sample distance, a phase-locked loop (FIG. 11) is used. A
microwave signal of frequency f.sub.ref is input into the cavity
10, with the cavity output being mixed with a signal coming from a
reference path. The length of the reference path is adjusted so
that the mixed output is zero when the resonance frequency of the
cavity matches f.sub.ref. The output of the phase detector 41 is
fed to an integrator 44, which regulates the tip-sample distance by
changing the extension of a piezoelectric actuator 50 (Burleigh
PZS-050) to maintain the integrator output near zero. For samples
with uniform dielectric constant, this corresponds to a constant
tip-sample separation. To measure .di-elect cons..sub.333, an
oscillating voltage V.sub..OMEGA., of frequency f.sub..OMEGA., is
applied to the silver backing of the sample and the output of the
phase detector is monitored with a lock-in amplifier (SR 830). This
bias voltage will modulate the dielectric constant of a nonlinear
dielectric material at f.sub..OMEGA.. Since f.sub..OMEGA. exceeds
the cut-off frequency of the feedback loop, the high frequency
shift in .di-elect cons. from V.sub..OMEGA. does not affect the
tip-sample separation directly. By measuring the applied voltage to
the piezoelectric actuator and the first harmonic variation in the
phase output simultaneously, sample topography and .di-elect
cons..sub.333 can be measured simultaneously.
[0079] FIG. 12 shows images of topography and .di-elect
cons..sub.333 for a periodically poled single-crystal LiNbO.sub.3
wafer. The crystal is a 1 cm.times.1 cm single crystal substrate,
poled by application of a spatially periodic electric field. The
poling direction is perpendicular to the plane of the substrate.
The topographic image is constructed by measuring the voltage
applied to the piezoelectric actuator. It is essentially
featureless, with the exception of a constant tilt and small
variations in height correlated with the alternating domains. The
small changes are only observable if the constant tilt is
subtracted from the figure. Since .di-elect cons..sub.ijk is a
third-rank tensor, it reverses sign when the polarization switches,
providing an image of the domain structure. The nonlinear image is
constructed by measuring the first harmonic of the variation in
output of the phase detector using a lock-in amplifier. Since
.di-elect cons..sub.ijk reverses when the polarization switches,
the output of the lock-in switches sign when the domain direction
switches. The value (-2.4.times.10.sup.-19 F/V) is within 20% of
bulk measurements. The nonlinear image clearly shows the
alternating domains.
[0080] Ferroelectric thin films, with their switchable nonvolatile
polarization, are also of great interest for the next generation of
dynamic random access memories. One potential application of this
imaging method would be in a ferroelectric storage media. A number
of instruments based on the atomic force microscope have been
developed to image ferroelectric domains either by detection of
surface charge or by measurement of the piezoelectric effect. The
piezoelectric effect, which is dependent on polarization direction,
can be measured by application of an alternating voltage and
subsequent measurement of the periodic variation in sample
topography. These instruments are restricted to tip-sample
separations less than 10 nanometers because they rely on
interatomic forces for distance regulation, reducing the possible
data rate. Since the inventive microscope measures variations in
the distribution of an electric field, the tip-sample separation
can be regulated over a wide range (from nanometers to
microns).
[0081] ii.) For Samples with Varying Dielectric Constant, f.sub.r
Changes with .di-elect cons..
[0082] To extract the dielectric constant and topography
simultaneously, an additional independent signal is required. This
can be accomplished in several ways:
[0083] 1. Measuring more than one set of data for f.sub.r and Q at
different tip-sample distances. This method is especially effective
when the tip-sample distance is very small. The models described in
Embodiment A can then be used to determine the tip-sample distance
and electrical impedance through DSP or computer calculation. In
this approach, as the tip is approaching the sample surface the DSP
will fit the tip-sample distance, dielectric constant and loss
tangent simultaneously. These values should converge as the
tip-sample distance decreases. Therefore, this general approach
will provide a true non-contact measurement mode, as the tip can
kept at any distance away from the sample surface as long as the
sensitivity (increase as tip-sample distance decreases) is enough
for the measurement requirement. This mode is referred to as
non-contact tapping mode. During the scanning, at each pixel the
tip is first pull back to avoid crash before lateral movement. Then
the tip will approach the surface as DSP calculate the dielectric
properties and tip-sample distance. As the measurement value
converge to have less error than specified or calculated tip-sample
distance is less than a specified value, the tip stop approaching
and DSP record the final values for that pixel. In this approach, a
consistent tip-sample moving element is critical, i.e. the element
should have a reproducible distance vs, e.g. control voltage.
Otherwise, it increases the fitting difficulty and measurement
uncertainty. This requirement to the z-axis moving element may be
hard to satisfy. An alternative method is to independently encode
the z-axis displacement of the element. Capacitance sensor and
optical interferometer sensor or any other distance sensor can be
implemented to achieve this goal.
[0084] 2. In particular, when the tip is in soft contact (only
elastic deformation is involved) to the surface of the sample,
there will be a sharp change in the derivatives of signals (fr and
Q) as function of approaching distance. This method is so sensitive
that it can be used to determine the absolute zero of tip-sample
distances without damaging the tip. Knowing the absolute zero is
very useful and convenient for further fitting of the curve to
determine the tip-sample distance and electrical impedance. A soft
contact "tapping mode" (as described in above) can be implemented
to perform the scan or single point measurements. The approaching
of tip in here can be controlled at any rate by computer or DSP. It
can be controlled interactively, i.e. changing rate according to
the last measurement point and calculation.
[0085] 3. The method described in 1) can be alternatively achieved
by a fixed frequency modulation in tip-sample distance and detected
by a lock-in amplifier to reduce the noise. The lock-in detected
signal will be proportional to the derivative of f.sub.r and Q as
function of tip-sample distance. A sharp decrease in this signal
can be used as a determination of absolute zero (tip in soft
contact with sample surface without damaging the tip). Using
relationships described in Embodiment A, any distance of tip-sample
can be maintained within a range that these relationship is
accurate enough.
Details
[0086] At a single tip-sample separation, the microwave signal is
determined by the dielectric constant of the sample. However, the
microwave signal is a function of both the tip-sample separation
and the dielectric constant. Thus, the dielectric constant and
tip-sample separation can be determined simultaneously by the
measurement of multiple tip-sample separations over a single point.
Several methods can be employed to achieve simultaneous measurement
of tip-sample separation and dielectric constant. First, the
derivative of the tip-sample separation can be measured by varying
the tip-sample separation. Given a model of the tip-sample
capacitance, (Eqn. 3), the tip-sample separation and dielectric
constant can then be extracted. The dependence on tip-sample
separation and dielectric constant can be modeled using a modified
fermi function. C r = 4 .times. .pi. 0 .times. R .times. ln .times.
( 1 - b ) / b + 1 exp .times. { G .function. ( ) .function. [ ln
.times. .times. a ' - x 0 .function. ( ) ] } + 1 ( 43 ) ##EQU40##
where b=(.di-elect cons.-.di-elect cons..sub.0)/(.di-elect
cons.+.di-elect cons..sub.0). Furthermore, G(.di-elect cons.) and
x.sub.0(.di-elect cons.) can be fitted well with rational functions
as: { G .function. ( ) = 9.57 .times. 10 - 1 + 2.84 .times. 10 - 2
.times. + 3.85 .times. 10 - 5 .times. 2 1 + 4.99 .times. 10 - 2
.times. + 1.09 .times. 10 - 4 .times. 2 x 0 .function. ( ) = 5.77
.times. 10 - 1 + 1.31 .times. 10 - 1 .times. + 3.55 .times. 10 - 4
.times. 2 1 + 3.68 .times. 10 - 2 .times. + 5.16 .times. 10 - 5
.times. 2 ( 44 ) ##EQU41## FIG. 13 illustrates the agreement
between Eqn. 3 and Eqn. 43.
[0087] Equation 43 is suitable for rapid calculation of the
tip-sample separation and dielectric constant. The tip-sample
separation and dielectric constant can also be extracted by
construction of a look-up table. The architecture described in f)
below is then used to regulate the tip-sample separation. FIG. 14a
and FIG. 14b show the variation of the derivative signal versus
tip-sample variation. FIG. 14a is modeled assuming a 10 .mu.m
spherical tip. FIG. 14b is a measured curve. Maintaining the
tip-sample separation at the maximum point of the derivative signal
can also regulate the tip-sample separation. This has been
demonstrated by scanning the tip-sample separation and selecting
the zero slope of the derivative signal. It can also be
accomplished by selecting the maximum of the second harmonic of the
microwave signal.
e) Tip-Sample Distance Control for Conductive Materials
[0088] For conductive materials, the tip-sample separation and
microwave resistivity can be measured simultaneously in a similar
fashion. Since Eqn. 24 is independent of conductivity for good
metals, C.sub.tip-sample can be used as a distance measure and
control. This solution should be generally applicable to a wide
class of scanned probe microscopes that include a local electric
field between a tip and a conducting sample. It should prove widely
applicable for calibration and control of microscopes such as
scanning electrostatic force and capacitance microscopes.
[0089] From the calibration curves, a frequency f.sub.ref is chosen
to correspond to some tip sample separation (FIG. 15). The
tip-sample separation is then regulated to maintain the cavity
resonance frequency at f.sub.ref. This can be accomplished
digitally through the use of the digital signal processor described
in f) below. An analog mechanism can and has also been used. A
phase-locked loop described in FIG. 11 has also been used to
regulate the tip-sample separation. FIG. 16(a)(c) illustrate the
measurement of topography with constant microwave conductivity.
FIG. 16(b)(d) illustrate the measurement of conductivity
variations.
[0090] This method allows submicron imaging of the conductivity
over large length scales. This method has the advantage of allowing
distance regulation over a wide length scale (ranging from microns
to nanometers) giving rise to a capability analogous to the optical
microscope's ability to vary magnification over a large scale.
Force Sensor Distance Feedback Control
[0091] In many cases, an absolute and independent determination of
zero-distance is desirable as other methods can rely on model
calculations and require initial calibration and fitting. A method
that relies on measuring the vibration resonant frequency of the
tip as a force sensor has been implemented. When the tip approaches
the sample surface, the mechanical resonant frequency of the tip
changes. This change can be used to control the tip-sample
distance. This approach has been shown to be feasible and that this
effect does exist. This effect has been found to exist over a very
long range (.about.1 micron). The long-range effect is believed to
come from the electrostatic force and the short-range effect is
from a shear force or an atomic force. A low frequency DDS-based
digital frequency feedback control electronics system similar to
the microwave one discussed above is implemented to track the
resonant frequency and Q of this mechanical resonator. The measure
signals will then be used to control the distance. This is similar
to shear force measurement in near field scanning microscope
(NSOM). The signal is derived from the microwave signal, not from
separate optical or tuning fork measurements as in other methods.
This feature is important since no extra microscope components are
required. The electronics needed is similar to the high frequency
case. This features is critical for high-resolution imaging and
accurate calibration of the other two methods.
Integration of AFM
[0092] It is very useful to integrate an atomic force microscope
(AFM) tip with SEMM in some applications where a high-resolution
topography image is desirable. The tip resonant frequency (10 kHz)
and quality factor limit the bandwidth of shear force feedback in
closed loop applications. Previously, in a scanning capacitance
microscope (SCM), an AFM tip has been connected to a microwave
resonator sensor to detect the change in capacitance between the
tip and the bottom electrode. In principle, SCM is similar to our
SEMM. However, insufficient coupling between the tip and resonator,
very large parasitic capacitance in the connection, and lack of
shielding for radiation prevented the SCM from having any
sensitivity in direct (dc mode) measurements of the tip-sample
capacitance without a bottom electrode and ac modulation. (SCM only
can measure dC/dV).
[0093] A new inventive design will allow easy integration of an AFM
tip to the SEMM without losing the high sensitivity experienced
with the SEMM. This new desion is based on a modification stripline
resonator with the inventors' proprietary tip-shielding structure
(see FIG. 21). Since the AFM tip is connected within less than 1 mm
of the central strip line, the sensitivity will not be reduced
seriously and parasitic capacitance with be very small (shielding
also reduces the parasitic capacitance). The AFM tip is custom
designed to optimize the performance. With this unique inventive
design, nanometer resolution can be achieved in both topography and
electrical impedance imaging, which is critical in gate oxide
doping profiling and many other applications.
Embodiment C
[0094] By contrast to most types of microscope, SEMM measures a
complex quantity, i.e., the real and imaginary parts of the
electrical impedance. This is realized by measuring the changes in
the resonant frequency (f.sub.r) and quality factor (Q) of the
resonator simultaneously. A conventional method of measuring these
two quantities is to sweep the frequency of the microwave generator
and measure the entire resonant curve. For each measurement, this
can take seconds to minutes depending on the capabilities of the
microwave generator. These measurements are limited by the
switching speed of a typical microwave generator to roughly 20 Hz.
With the use of a fast direct digital synthesizer based microwave
source, the throughput can be improved to roughly 10 kHz, but is
still limited by the need to switch over a range of frequencies.
Another method is to implement an analog phase-locked loop for
frequency feedback control. This method can track the changing
resonant frequency in real time and measure f.sub.r and Q quickly.
However, one has to use a voltage-controlled-oscillator (VCO) as a
microwave generator which usually only has a frequency stability of
10.sup.-4. This low frequency stability will seriously degrade the
sensitivity of the instrument. Since .DELTA..di-elect
cons./.di-elect cons..about.500 .DELTA.f.sub.r/f.sub.r, frequency
instability in the VCO will limit measurement accuracy. Another
problem is that interaction between this frequency feedback loop
and the tip-sample distance feedback loop can cause instability and
oscillation, which will seriously limit the data rate.
[0095] A direct digital synthesizer (DDS) based microwave generator
is used to implement the method according to the present invention.
In a preferred embodiment, the DSS has a frequency stability of
better than 10.sup.-9. The inventive method fixes the frequency of
the microwave signal at the previous resonant frequency and
measures I/Q signals simultaneously. Since the microwave frequency
is fixed, the DDS switching speed does not limit the data rate. By
measuring the in-phase and quadrature microwave signals, the
inventors can derive f.sub.r and Q. Near resonance, the in-phase
and quadrature signals are given by: i=A sin .theta.
[0096] Q=A cos .theta., where A is the amplitude of the microwave
signal on resonance and .theta. is the phase shift of the
transmitted wave. Given i, q, the current input microwave
frequency, and the input coupling constants, the current f.sub.r
and Q can be calculated.
[0097] For a resonator with initial quality factor Q.sub.0,
transmitted power A.sub.0, and resonant frequency f.sub.0, driven
at frequency f, .DELTA. .times. .times. f = 1 .times. 2 .times. tan
.times. .times. .theta. .times. .times. .times. f .times. r .times.
Q = 1 .times. 2 .times. c .times. .times. sin .times. .times.
.theta. .times. A , f .times. r = f - .DELTA. .times. .times. f Q =
1 .times. c .times. f .times. r .times. A .times. cos .times.
.times. .theta. , where c = .times. A .times. 0 .times. .times. f
.times. 0 .times. Q .times. 0 ##EQU42## Then, f.sub.r and Q can be
obtained by f r = f - 1 2 .times. c .times. i i 2 + q 2 ##EQU43## Q
= 1 c .times. f r .times. i .times. 2 + q 2 q ##EQU43.2##
[0098] Initially, since the I/Q mixer does not maintain perfect
phase or amplitude balance, these quantities are calibrated. To
calibrate the relative amplitudes at a given frequency of the i and
q outputs of the mixer, the relative values of the outputs are
measured when the reference signal is shifted by 90 degrees. This
can easily be extended by means of a calibration table.
[0099] To calibrate the relative phases of the i and q outputs at a
given frequency, the i output of the mixer is measured on
resonance. At resonance,
[0100] i/q=.delta., where .delta. is the phase error of the I/Q
mixer. Near resonance, i = A .times. .times. sin .function. (
.theta. + .delta. ) = A .times. .times. sin .times. .times.
.theta.cos .times. .times. .delta. + A .times. .times. sin .times.
.times. .delta. .times. .times. cos .times. .times. .theta. = A
.times. .times. sin .times. .times. .theta. + A .times. .times.
.delta.cos .times. .times. .theta. = A .times. .times. sin .times.
.times. .theta. + q .times. .times. .delta. ##EQU44## This allows
the correction of the phase error of the mixer.
[0101] This method of measurement only requires one measurement
cycle. Therefore, it is very fast and limited only by the DSP
calculation speed. To increase the working frequency range, the DSP
is used to control the DDS frequency to shift when the resonant
frequency change is beyond the linear range. This method allows
data rates around 100 kHz-1 MHz (limited by the bandwidth of the
resonator) and frequency sensitivity below 1 kHz ( .DELTA. .times.
.times. f r f r .apprxeq. 10 - 6 .times. .times. .times. to .times.
.times. 10 - 2 ) . ##EQU45## f) Data Acquisition and Control
Electronics
[0102] FIGS. 17 and 18 illustrate the architecture of the inventive
data acquisition and control electronics. FIG. 17 contains a
schematic for the EMP. FIG. 18 is a flow chart describing the
operation of the SEMP.
[0103] To eliminate the communication bottleneck between data
acquisition, control electronics and the computer, the high
performance PCI bus is adopted for every electronic board. A main
board with four high-speed digital signal processors (DSPs) is used
to handle data acquisition, feedback control loops and other
control functions separately. Four input data signals (A/Ds) and
six control signals (D/As) were implemented.
[0104] The input signals include:
[0105] 1) in-phase (I=A sin .theta., where A is the amplitude and
.theta. the phase) signal,
[0106] 2) quadrature (q=A cos .theta.) signal,
[0107] 3) in-phase signal of tip vibration resonant frequency or
tapping mode signal or electric/magnetic/optical field
modulations,
[0108] 4) quadrature signal of tip vibration resonance,
[0109] The output signals include:
[0110] 1-3) fine x-y-z piezo-tube control signals,
[0111] 4) z-axis coarse piezo-step-motor signal,
[0112] 5-6) coarse x-y stage signals.
[0113] The DSPs are dedicated to specific functions as follows:
[0114] DSP1: microwave frequency f.sub.r and Q data
acquisition,
[0115] DSP2: tip mechanical resonant frequency f.sub.m and Q.sub.m
or microwave derivative signal data acquisition,
[0116] DSP3: tip-sample distance feedback control,
[0117] DSP4: x-y fine/coarse scan and image data acquisition,
The division of labor between a number of fast processors
simplifies design and allows rapid processing of multiple tasks. A
fast system bus is necessary to allow rapid transfer of data to the
display and between neighboring boards.
Stepping Motor--Coarse Positioning
Embodiment D
[0118] For all scanned probe microscopes, increasing resolution
decreases the measurable scan range. Microscopes must be able to
alter their scan position by millimeters to centimeters while
scanning with high resolution over hundreds of microns. Given
samples of macroscopic scale, means must exit to adjust the
tip-sample separation over macroscopic (mm) distances with high
stability over microscopic distances (nm). Conventional
piezoelectric positioners are capable only of movements in the
range of hundreds of microns and are sensitive to electronic noise
even when stationary. In addition, they are subject to large
percentage drifts. (>1% of scan range) As such, separate means
of coarse and fine adjustment of position are necessary. A coarse
approach inventive mechanism by means of a novel piezoelectric
stepper motor has been designed to accomplish this.
[0119] FIG. 19 illustrates the design and operation of the stepper
motor. The cross-sectional view illustrates that the motor consists
of a sapphire prism in the form of an equilateral triangle clamped
into an outer casing. There are 3 Piezoelectric stacks topped by a
thin sapphire plate contact each side of the prism. Each
piezoelectric stack consists of a lower expansion plate, which is
used to grip and release the prism, an upper shear plate, which is
used to move the prism, and a thin sapphire plate, which is used to
provide a uniform surface. FIG. 20 shows the sequence of motion. At
step (a), the motor is stationary. At step (b), a smooth rising
voltage is applied to each shear place and the center prism moves.
At step (c), the voltage to the expansion plates labeled (2) is
reduced. This reduces the pressure applied by those plates and thus
the frictional force. The center points the motor fixed. At step
(d), the voltage to the shear plates labeled (1) is reduced. Since
the plates have been retracted, the friction between these plates
and the prism is reduced, letting the center plates hold the prism.
At step (e), the voltage to the expansion plates labeled (2) is
increased, increasing the pressure applies by the other plates. At
step (F), the voltage to the expansion plates labeled (4) is
reduced. The outer points hold the motor fixed. At step (g), the
voltage to the shear plates labeled (3) is reduced. At step (h),
the voltage to the expansion plates labeled (4) is increased,
restoring the pressure applied by the center points to its original
value. This motor has a number of advantages by comparison to
earlier designs. Prior designs had only 2 piezoelectric elements
per side and relied on a stick-slip motion, similar to pulling a
tablecloth from under a wineglass. The use of a third piezoelectric
element on each side and the addition of an expansion piezoelectric
have several benefits. First, since the method of motion does not
involve slip-stick, it is less prone to vibration induced by the
necessary sharp motions. Second, the requirements for extreme
cross-sectional uniformity of the central element are reduced by
the use of the expansion piezoelectric plates.
[0120] While the present invention has been particularly described
with respect to the illustrated embodiment, it will be appreciated
that various alterations, modifications and adaptations may be made
based on the present disclosure, and are intended to be within the
scope of the present invention. While the invention has been
described in connection with what is presently considered to be the
most practical and preferred embodiments, it is to be understood
that the present invention is not limited to the disclosed
embodiments but, on the contrary, is intended to cover various
modifications and equivalent arrangements included within the scope
of the appended claims.
* * * * *