U.S. patent application number 11/177851 was filed with the patent office on 2005-12-01 for method and apparatus for code phase tracking.
Invention is credited to Goodings, Christopher J..
Application Number | 20050265494 11/177851 |
Document ID | / |
Family ID | 9891481 |
Filed Date | 2005-12-01 |
United States Patent
Application |
20050265494 |
Kind Code |
A1 |
Goodings, Christopher J. |
December 1, 2005 |
Method and apparatus for code phase tracking
Abstract
A method of code phase tracking for CDMA type communication is
disclosed together with apparatus including a GPS receiver (10 to
15) for the same. The method uses a modified early-minus-late
correlation function to determine the code phase error between a
target pseudorandom noise code of an incoming signal and locally
generated replica codes. The early-minus-late correlation function
is modified compared to the true function whereby the gradient of
the modified function at zero code phase error is increased. This
may be achieved my modifying the early-minus-late correlation
function after its derivation. Alternatively, it may be modified by
modifying variables from which it is derived and in particular, by
modifying the either or both of the power spectrums of the subject
signal or the early and late replica code signals whereby at least
one odd harmonic is reduced in size or removed; at least one even
harmonic is increased in size; or the bandwidth is truncating
between harmonics so as to excise an adjacent even harmonic.
Inventors: |
Goodings, Christopher J.;
(Fleet, GB) |
Correspondence
Address: |
PHILIPS INTELLECTUAL PROPERTY & STANDARDS
P.O. BOX 3001
BRIARCLIFF MANOR
NY
10510
US
|
Family ID: |
9891481 |
Appl. No.: |
11/177851 |
Filed: |
July 8, 2005 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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11177851 |
Jul 8, 2005 |
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09854402 |
May 11, 2001 |
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6931056 |
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Current U.S.
Class: |
375/343 ;
342/357.69; 375/150; 375/E1.016 |
Current CPC
Class: |
H04J 13/0022 20130101;
G01S 19/30 20130101; H04B 1/7085 20130101 |
Class at
Publication: |
375/343 ;
375/150 |
International
Class: |
H04B 001/707 |
Foreign Application Data
Date |
Code |
Application Number |
May 13, 2000 |
GB |
0011493.4 |
Claims
1-10. (canceled)
11. A receiver comprising a signal processor for modifying the
power spectrum of a received subject signal containing a target
pseudorandom noise code so that the power spectrum of the subject
signal has either at least one odd harmonic which is reduced in
size or removed; at least one even harmonic which is increased in
size; or a reduced bandwidth which is truncated between harmonics
so as to excise an adjacent even harmonic.
12-20. (canceled)
Description
[0001] This invention relates to a method of code phase tracking
for code division multiple access (CDMA) type communication and to
apparatus for the same.
[0002] The invention is of particular benefit to the field of
global positioning systems (GPS) and is described with reference to
GPS hereafter. However, such reference should not be interpreted as
limiting the scope of the invention to merely GPS. For example, the
invention is equally applicable to CDMA communication between
mobile cellular telephones and associated networks.
[0003] At present, GPS is most notably associated with the
Navigation System with Time and Ranging (NAVSTAR) GPS, an all
weather, spaced based navigation system developed and operated by
the US Department of Defense, however, the general principles
underlying GPS are universal and not merely limited to NAVSTAR.
Accordingly, GPS hereafter refers to any global positioning system
comprising a plurality of CDMA radio transmitters at different
locations and a receiver which determines its location based on the
time of arrival of the transmissions of the radio transmitters.
[0004] As is well known, a GPS receiver may implement a
pseudorandom noise (PRN) code correlation loop in which early (E),
prompt (P) and late (L) replica codes of satellite PRN codes are
continuously generated, and compared to the incoming satellite PRN
codes as received by the receiver. Using a code phase discriminator
calculated as a function of the correlation, it is then possible to
determine whether a target incoming code has been acquired; if the
code phase discriminator exceeds a predetermined threshold level,
the target incoming code and the locally generated replica codes
can be assumed to be in phase, i.e. the code is acquired. If not,
the code generator produces the next series of replicas with a
phase shift, typically of one chip, and the code phase
discriminator is recalculated. Assuming carrier phase lock, a
linear code sweep should eventually result in the target incoming
code being in phase with that of the locally generated replica
codes and therefore, if detected, code acquisition.
[0005] Once the code is acquired, a code phase lock loop may be
used to track the code, typically employing an early-minus-late
code correlation function in order to determine the code phase
error between the target incoming code and the locally generated
replica codes. As code phase shift is the basis for pseudorange
measurements, the accuracy to which the pseudoranges can be
measured and ultimately to which the position of the GPS receiver
can be estimated depends on the accuracy to which the code phase
lock loop can determine the code phase error.
[0006] It is therefore an object of the invention to provide a
method and apparatus for code phase tracking able to more
accurately determine the code phase error between a target
pseudorandom noise code of an incoming signal and locally generated
replica codes.
[0007] According to the present invention, a method of code phase
tracking is provided together with a receiver comprising an antenna
and a signal processor for the same. The method comprises the steps
of:
[0008] (a) receiving a subject signal containing a target
pseudorandom noise code;
[0009] (b) generating a series of signals containing early and late
replica codes corresponding to the target code;
[0010] (c) correlating the subject signal with the early and late
replica code signals and returning respective early and late
correlation values; and
[0011] (d) determining the code phase error between the target code
and the replica codes from a modified early-minus-late correlation
function derived from the early and late correlation values, the
modified early-minus-late correlation function being such that its
gradient at zero code phase error is increased compared to the true
early-minus-late correlation function.
[0012] Increasing the gradient of the early-minus-late correlation
function at the zero code phase error enables the occurrence of
zero code phase error to be determined more accurately.
[0013] The early-minus-late correlation function may be modified
after its derivation or, alternatively, prior to its derivation by
modifying either the subject signal, the early and late replica
code signals or the early and late correlation values.
[0014] Modifying any one of the parameters from which the
early-minus-late correlation function is derived may enable the
necessary signal processing to be simplified. For example,
modification of the subject signal may be achieved using relatively
simple front-end analogue circuitry and modification of the early
and late replica code signals may be achieved either by processing
unmodified replica signals, say by filtering, or by direct
generation of the appropriate sequence, especially if done in the
digital domain.
[0015] Modification of the subject signal and/or the early and late
replica code signals preferably includes modification of their
respective power spectrums by at least one of the following:
reducing in size or removing at least one odd harmonic; increasing
in size at least one even harmonic; and truncating the bandwidth
between harmonics so as to excise an adjacent even harmonic.
[0016] There is believed there is a direct relationship between the
power spectrums of the subject signal/replica signals and the
early-minus-late correlation function such that the aforementioned
harmonic manipulation increases the gradient of the
early-minus-late correlation function at the point of zero code
phase error. This relationship together with its effect on the
accuracy to which the occurrence of zero code phase error can be
determined is explained below.
[0017] The invention further provides a receiver comprising a
signal processor for modifying the power spectrum of a received
subject signal containing a target pseudorandom noise code whereby
the power spectrum of the subject signal has either at least one
odd harmonic which is reduced in size or removed; at least one even
harmonic which is increased in size; or a reduced bandwidth which
is truncated between harmonics so as to excise an adjacent even
harmonic.
[0018] The invention yet further provides a receiver comprising an
antenna for receiving a subject signal containing a target
pseudorandom noise code; and a signal processor comprising a
generator for generating a series of signals containing early and
late replica codes corresponding to the target code, a correlator
for correlating the subject signal with the early and late replica
code signals and returning respective early and late correlation
values, and means for determining the code phase error between the
target code and the replica codes from a modified early-minus-late
correlation function derived from the early and late correlation
values, such that the gradient of the modified early-minus-late
correlation function at zero code phase error is increased compared
to the true early-minus-late correlation function.
[0019] The above receiver may modify the early-minus-late
correlation function by modifying either the subject signal, the
early and late replica code signals or the early and late
correlation values prior to deriving the early-minus-late
correlation function. In particular, but not exclusively, the
receiver may modify the early-minus-late correlation function by
modifying the subject signal or the early and late replica code
signals whereby either or both of their respective power spectrums
have either at least one odd harmonic which is reduced in size or
removed; at least one even harmonic which is increased in size; or
a reduced bandwidth which is truncated between harmonics so as to
excise an adjacent even harmonic.
[0020] The above and other features and advantages of the present
invention will be apparent from the following description, by way
of example, of an embodiment of a method of code phase tracking and
a GPS receiver according to the present invention with reference to
the accompanying drawings in which:
[0021] FIG. 1 shows, schematically, an SPS GPS receiver according
to the present invention;
[0022] FIG. 2 shows, schematically, the receiver channels and
receiver processor of the GPS receiver of FIG. 1 in greater
detail;
[0023] FIG. 3 is a graph showing an idealised early-minus-late
correlation function for 1.0 MHz and 4.0 MHz bandwidths;
[0024] FIG. 4 is a graph showing the gradient of the
early-minus-late correlation function plotted against the bandwidth
of the incoming signal for both unmodified and modified incoming
signals; and
[0025] FIGS. 5A and 5B are graphs showing normalised power
spectrums for both an unmodified and a modified incoming signal
respectively.
[0026] The general principles underlying GPS and methods and
apparatus for its implementation are known. For example, see GPS
Principles and Applications (Editor, Kaplan) ISBN 0-89006-793-7
Artech House, hereinafter "Kaplan".
[0027] As is well known, each NAVSTAR GPS satellite transmits two
carrier frequencies; L1, the primary frequency at 1575.42 MHz and
L2, the secondary frequency at 1227.60 MHz. The carrier frequencies
are modulated by spread spectrum codes with a PRN sequence unique
to each satellite and also by the navigation data message. The L1
signal is modulated the coarse/acquisition (C/A) code and the
precision (P[Y]) code whereas the L2 signal is modulated by the
P[Y] code only. The P[Y] codes relate to the precise positioning
service (PPS) primarily for military and select government agency
users whereas the C/A relates to the standard positioning service
(SPS) for which there is currently unrestricted access.
[0028] FIG. 1 shows, schematically, the architecture of an SPS GPS
receiver according to the present invention. GPS radio frequency
(RF) signals are received by an antenna 10 and pre-processed in a
pre-processor 11 by preamplification, passive bandpass filtering in
order to minimise out-of-band RF interference, further filtering to
filter out first harmonics of the power spectrum of the incoming
signal; down converting to an intermediate frequency (IF) and
analog to digital conversion. Digitised IF signals are then
provided to each of n digital receiver channels 12 in which they
are acquired and tracked in co-operation with the receiver
processor 13 for the purpose of acquiring navigation information.
Such methods for acquisition and tracking are well known, for
example, see chapter 4 (GPS satellite signal characteristics) &
chapter 5 (GPS satellite signal acquisition and tracking), Kaplan
ibid. Using acquired navigation information and the time of arrival
of the transmissions, the navigation processor 14 calculates the
position of the receiver using conventional algorithms and that
position is displayed on a display 15 to the user.
[0029] The pre-processor 11 will be typically implemented in the
form of front end analogue circuitry with the digital receiver
channels 12, the receiver processor 13 and the navigation processor
14 implemented in the form of a general purpose microprocessor or a
microprocessor embedded in a GPS application specific integrated
circuit (ASIC).
[0030] FIG. 2 shows, schematically, the receiver channel
co-operating with the receiver processor in greater detail. In
order to retrieve the information on the incoming signal, a carrier
wave (CW) must be removed and this is done by the receiver
generating in-phase (I) and quadrature phase (Q) replica carrier
wave signals using a carrier wave generator 21. The replica carrier
waves ideally have the same frequency as the received signal,
however, due to Doppler shift caused by the relative movement
between the receiver and orbiting satellites, the frequency of the
GPS signals as received in the receiver normally differs from the
precise satellite transmission frequency. In order to accurately
replicate the frequency of the received carrier wave, a
conventional carrier wave phase lock loop is employed. Although
perhaps undesirable, it is possible to omit the carrier phase lock
stage altogether as the Doppler shift of the carrier and its
associated effect on the code phase discriminator is reasonably
small.
[0031] In order to acquire code phase lock, early (E), prompt (P)
and late (L) replica codes of the PRN sequences are continuously
generated by a code generator 22 at a frequency related to the
received carrier (i.e. nominal plus Doppler). The replica codes are
then correlated with the I and Q signals to produce three in-phase
correlation components (IE, IL, IP) and three quadrature phase
correlation components (QE, QL, QP), typically by integration in an
integrator 23 over substantially the whole of the PRN code. In the
receiver processor 13, a code phase discriminator is calculated as
a function of the correlation components on which a threshold test
is applied and a phase match declared if the code phase
discriminator is high. If not, the code generator produces the next
series of replicas with single chip phase advance and the code
phase discriminator is recalculated. Any declared phase match is
validated by recalculating the discriminator. A linear phase sweep
will eventually result in the incoming PRN code being in phase with
that of the locally generated replica and thus code
acquisition.
[0032] Once the code is acquired, a code phase lock loop employing
an early-minus-late correlation function is used to track the code.
The early-minus-late correlation function is modified compared to
the true early-minus-late correlation function because of the
aforementioned filtering out of the first harmonics of the power
spectrum of the incoming signal. As a consequence, the code phase
lock loop provides an enhanced code phase error signal relating to
the time difference between the incoming signal and the two local
generated replicas.
[0033] Without wishing to be bound by any theory, the inventor
believes the following analysis explains the relationship between
the accuracy to which the code phase lock loop can determine the
code phase error, the gradient of the early-minus-late correlation
function at zero code phase error and the aforementioned power
spectrum manipulation.
[0034] As previously stated, the accuracy to which the pseudoranges
can be measured and ultimately to which the position of the GPS
receiver can be estimated The relationship between the code phase
error measurement (.DELTA.t) and an associated pseudorange error
measurement (.DELTA.d) can be expressed as:
.DELTA.d.apprxeq.c.DELTA.t [Equation 1]
[0035] where c is the speed of light. Thus, if the accuracy of the
pseudorange measurement is to be determined to 1 metre, the code
loop error (after averaging for noise) must be to within 3.3 ns, or
about {fraction (1/300)} T.sub.c where T.sub.c is the GPS L1 C/A
signal chip period.
[0036] Curves 31 and 32 of FIG. 3 shows the true, normalised,
early-minus-late (E-L) correlation function for signal bandwidths
of 1.0 MHz and 4.0 MHz respectively where the E-L correlation is
normalised by the prompt (P) correlation value. In a GPS receiver
using early-minus-late code phase tracking, the code phase tracking
error (.DELTA.t) can therefore be approximated whereby: 1 t = 1 g {
E - L n e n 1 P n p } [ Equation 2 ]
[0037] where g is the gradient of the (E-L)/P graph at t=0; E, L
and P are correlation measurements; and n.sub.e, n.sub.i and
n.sub.p are early, late and prompt correlation measurement noise
respectively.
[0038] Assuming the noise to be small and uncorrelated, at E-L=0,
the code phase error .DELTA.t can be expressed as follows: 2 t 1 g
2 n P [ Equation 3 ]
[0039] where n is the magnitude of the effective noise.
[0040] It is therefore possible to define a figure of merit (FOM)
which is approximately directly proportional to the receiver
accuracy: 3 FOM = ( E - L ) t t = 0 .times. n [ Equation 4 ]
[0041] A finite length pseudorandom noise code may be approximated
to an ideal random binary code. Accordingly, there is no
periodicity of the correlation and therefore no line quantisation
of the correlation power spectrum.
[0042] The power spectrum of the correlation is given by the
Fourier transform of the correlation power. Since there is no
Fourier transform of a truly random binary code, in order to obtain
the power spectrum, the Fourier transform of the autocorrelation
function is calculated. For a binary code having an amplitude of
.+-.A, the autocorrelation function is the classic triangular shape
with a base length of .+-.1 code chip and a maximum correlation
power of A.sup.2. This can be expressed as follows: 4 R ( ) = A 2 (
1 - T c ) for T c R ( ) = 0 for > T c [ Equation 5 ]
[0043] where T.sub.c is the chip period for the C/A code). The
Fourier transform of this is given by: 5 S ( ) = A 2 T c sin c 2 Tc
2 [ Equation 6 ]
[0044] As the correlation power can be derived from the inverse
Fourier transform of the correlation power spectrum, it is
therefore possible to use the power spectrum as a starting point
for the derivation of the correlation function shape and size.
Also, the power spectrum can be modified to reflect the
consequences of any front-end filter characteristics and the
Fourier transform taken of the resulting spectrum to derive the
correlation characteristics. Although this does not take account of
several significant effects, such as the low resolution ADC in an
actual system, the qualitative results appear to be consistent with
theorised models.
[0045] To do this, an idealised power spectrum is first generated
over a range of frequencies (say .+-.8 MHz or .+-.12 MHz in 0.02
MHz steps) and held in a vector. The vector can then be "filtered"
by multiplying individual elements by a transmission factor. The
simplest form of filtering only uses filter transmissions of zero
or one. The power spectrum is normalised to an amplitude, A, of
unity.
[0046] The precise correlation can then be generated from the power
spectrum by taking the inverse Fourier transform. This is a vector
containing the correlation as a function of time, expressed in
terms of the chip period Tc. The total length of the vector (i.e.
the range of time) is inversely proportional to the frequency step
size of the power spectrum and the correlation step size is
inversely proportional to the total range of the power
spectrum.
[0047] The early-minus-late correlation can be generated by a
sparse matrix operation on the precise correlation vector.
EL(t)=M.sub.(E-L)-P(t)
[0048] where EL(t) is a column vector containing the (E-L)
correlation result, P(t) is a column vector containing the precise
correlation result and M.sub.(E-L) is a square matrix whose
dimension is the same as the vector lengths. The matrix M.sub.(E-L)
is mainly zero but contains elements of +1 on diagonals which
perform the (E-L) arithmetic operation. An example is shown below:
6 M ( E - L ) = ( 0 0 1 0 0 0 0 0 0 0 1 0 0 0 - 1 0 0 0 1 0 0 0 - 1
0 0 0 1 0 0 0 - 1 0 0 0 1 0 0 0 - 1 0 0 0 0 0 0 0 - 1 0 0 ) [
Equation 7 ]
[0049] The distance of the leading element of each diagonal from
the top-left corner of the matrix gives the early-minus-late
spacing and will depend on the time resolution of the correlation
vector.
[0050] In FIG. 4, curve 41 shows the FOM, i.e. the gradient of the
early-minus-late correlation function, varying with the
half-bandwidth of incoming signal. This is due to the shape of the
correlation peak being modified by the varying sum of its Fourier
components. The reason for the variation in FOM with bandwidth is
further explained by the power spectrum of the correlation
described by curve 51 in FIG. 5A. The power spectrum has nulls at
1,2,3 . . . n MHz. Between 0 and 1 MHz, even Fourier components
contribute to the power, between 1 and 2 MHz odd Fourier components
contribute to the power (etc.). These even and odd components
eventually mix to form the ideal triangular correlation shape.
However, even components serve to increase the steepness of the
correlation shape around t=Tc/2 while odd components serve to
decrease the steepness. For half-chip (E-L) correlators, an
increase in the steepness of the correlation shape at Tc/2 results
in an increased (E-L) gradient at t-0 and therefore an improved
FOM.
[0051] The idea behind this invention is to selectively allow the
even components of the signals to contribute to the correlation
while blocking the odd components. In practice, to limit the
required bandwidth, only the first few components would be used.
The resulting modified power spectrum is described by curve 52 in
FIG. 5B, suggesting that the FOM can be increased by about 14% in
this way.
[0052] Even greater increases in the FOM can be achieved by
artificially adapting the shape of the correlation. Curve 53 of
FIG. 5B described the situation where the odd components have been
removed and the first harmonic even components increased by a
factor of 2. The FOM is now increased by a factor of about 28%.
Thus, this invention provides a method of increasing the accuracy
of a GPS system which uses code tracking only.
[0053] As previously stated, in a GPS receiver of the type shown
schematically in FIGS. 1 and 2, the pre-processing, receiver
channel and receiver processor will typically be implemented in the
form of front end analogue circuitry combined with either a general
purpose microprocessor or a microprocessor embedded in a GPS
application specific integrated circuit. Implementation of a method
of code phase tracking according to the present invention,
including the example as described below, may be accomplished by
appropriate analogue circuitry design and/or microprocessor
programming. Of course, such design and programming is well known
and would be accomplished by one of ordinary skill in the art of
GPS and CDMA communication without undue burden.
* * * * *