U.S. patent application number 10/761014 was filed with the patent office on 2005-07-21 for self-calibrating wideband phase continuous synthesizer and associated methods.
This patent application is currently assigned to Harris Corporation. Invention is credited to Coleman, John Roger, Mashburn, Travis Sean.
Application Number | 20050156781 10/761014 |
Document ID | / |
Family ID | 34750130 |
Filed Date | 2005-07-21 |
United States Patent
Application |
20050156781 |
Kind Code |
A1 |
Coleman, John Roger ; et
al. |
July 21, 2005 |
Self-calibrating wideband phase continuous synthesizer and
associated methods
Abstract
The synthesizer and method provide a relatively wideband swept
frequency signal and include generating a first swept frequency
signal with a first generator, and successively switching between
different frequency signals with a second generator. Such switching
creates undesired phase discontinuities in the output swept
frequency signal. The first swept frequency signal is combined with
the successively switched different frequency signals to produce
the relatively wideband swept frequency signal, and the second
generator is calibrated to reduce the undesired phase
discontinuities during switching based upon the output swept
frequency signal.
Inventors: |
Coleman, John Roger; (Palm
Bay, FL) ; Mashburn, Travis Sean; (Melbourne,
FL) |
Correspondence
Address: |
ALLEN, DYER, DOPPELT, MILBRATH & GILCHRIST P.A.
1401 CITRUS CENTER 255 SOUTH ORANGE AVENUE
P.O. BOX 3791
ORLANDO
FL
32802-3791
US
|
Assignee: |
Harris Corporation
Melbourne
FL
32919
|
Family ID: |
34750130 |
Appl. No.: |
10/761014 |
Filed: |
January 20, 2004 |
Current U.S.
Class: |
342/174 ;
342/201; 342/202 |
Current CPC
Class: |
G01S 7/4008 20130101;
G01S 7/282 20130101; G01S 13/90 20130101 |
Class at
Publication: |
342/174 ;
342/201; 342/202 |
International
Class: |
G01S 007/40 |
Claims
That which is claimed is:
1. An apparatus for generating a relatively wideband swept
frequency signal comprising: a first generator for generating a
first swept frequency signal; a second generator successively
switching between different frequency signals and creating
undesired phase discontinuities during switching; a mixer connected
to said first and second generators for mixing the first swept
frequency signal and the successively switched different frequency
signals to produce the relatively wideband swept frequency signal;
and a calibrator for calibrating said second generator to reduce
the undesired phase discontinuities during switching based upon the
relatively wideband swept frequency signal.
2. The apparatus according to claim 1 wherein said calibrator
comprises a self-calibration feedback loop including: a phase
locked loop (PLL) receiving a reference frequency signal; a mixer
receiving the relatively wideband swept frequency signal and a
phase reference signal from the PLL; an analog-to-digital (a/d)
converter receiving an output signal of the mixer; and a controller
connected to the a/d converter and providing a calibration signal
to the second generator.
3. The apparatus method according to claim 1, wherein said second
generator generates an offset frequency signal and successively
combines the offset frequency signal with a reference frequency
signal to produce the successively switched different frequency
signals.
4. The apparatus according to claim 3, wherein said first generator
comprises a first digital synthesizer and a translator to translate
an output of said first digital synthesizer to generate the first
swept frequency signal; and wherein said second generator
comprises: a second digital synthesizer to generate the offset
frequency signal; a plurality of frequency converters to
successively combine the offset frequency signal with a reference
frequency signal to produce the successively switched different
frequency signals; and a controller for controlling the operation
of said second digital synthesizer to maintain phase continuity
between the successively switched different frequency signals.
5. The apparatus according to claim 1, wherein said second
generator comprises: a plurality of frequency converters receiving
a reference frequency signal; and a second digital synthesizer
providing an offset frequency signal to the plurality of frequency
converters to successively combine the offset frequency signal with
the reference frequency signal to produce the different frequency
signals.
6. An apparatus for generating a relatively wideband swept
frequency signal comprising: a first digital synthesizer for
generating a first swept frequency signal; a second generator
successively switching between different frequency signals and
creating undesired phase discontinuities during switching; a mixer
connected to said first and second digital synthesizers for mixing
the first swept frequency signal and the successively switched
different frequency signals to produce the relatively wideband
swept frequency signal; and a self-calibration feedback loop to
reduce the undesired phase discontinuities during switching based
upon the relatively wideband swept frequency signal, the
self-calibration feedback loop comprising a phase locked loop (PLL)
receiving a reference frequency signal, a mixer receiving the
relatively wideband swept frequency signal and a phase reference
signal from the PLL, an analog-to-digital (a/d) converter receiving
an output signal of the mixer, and a controller connected to the
a/d converter and providing a calibration signal to the second
generator.
7. The apparatus according to claim 6, wherein said second
generator generates an offset frequency signal and successively
combines the offset frequency signal with a reference frequency
signal to produce the successively switched different frequency
signals.
8. The apparatus according to claim 7, wherein said second
generator comprises: a second digital synthesizer to generate the
offset frequency signal; and a plurality of frequency converters to
successively combine the offset frequency signal with a reference.
frequency signal to produce the successively switched different
frequency signals.
9. A method for generating a relatively wideband swept frequency
signal comprising: generating a first swept frequency signal with a
first generator; successively switching between different frequency
signals with a second generator while creating undesired phase
discontinuities during switching; combining the first swept
frequency signal and the successively switched different frequency
signals to produce the relatively wideband swept frequency signal;
and calibrating the second generator to reduce the undesired phase
discontinuities during switching based upon the relatively wideband
swept frequency signal.
10. The method according to claim 9, wherein successively switching
between different frequency signals comprises generating an offset
frequency signal and successively combining the offset frequency
signal with a reference frequency signal to produce the respective
different frequency signals.
11. The method according to claim 9, wherein the first generator
comprises a first digital synthesizer; and wherein the second
generator comprises a second digital synthesizer generating an
offset frequency signal and successively combining the offset
frequency signal with a reference frequency signal to produce the
respective different frequency signals.
12. The method according to claim 11, wherein calibrating the
second generator comprises comparing the phase of the relatively
wideband swept frequency signal before and after successively
switching between different frequency signals to determine the
undesired phase discontinuities created during switching, and
adjusting the phase of the offset frequency signal generated by the
second digital synthesizer to reduce the undesired phase
discontinuities created during switching.
13. The method according to claim 12, wherein determining the
undesired phase discontinuities created during switching and
adjusting the phase of the offset frequency signal generated by the
second digital synthesizer comprises providing a self-calibration
feedback loop including: a phase locked loop (PLL) receiving the
reference frequency signal; a mixer receiving the relatively
wideband swept frequency signal and a phase reference signal from
the PLL; an analog-to-digital (a/d) converter receiving an output
signal of the mixer; and a controller connected to the a/d
converter and providing a calibration signal to the second digital
synthesizer.
14. The method according to claim 9, wherein successively switching
between different frequency signals comprises: connecting a
plurality of frequency converters to an output of a reference
frequency signal generator; and coupling an offset frequency signal
to the plurality of frequency converters to successively combine
the offset frequency signal with the reference frequency signal to
produce the different frequency signals.
15. A method for generating a relatively wideband swept frequency
signal comprising: generating a first swept frequency signal with a
first generator; successively switching between different frequency
signals with a second generator and creating undesired phase
discontinuities during switching; combining the first swept
frequency signal and the successively switched different frequency
signals to produce the relatively wideband swept frequency signal;
and calibrating the second generator by comparing the phase of the
relatively wideband swept frequency signal before and after
successively switching between different frequency signals to
determine the undesired phase discontinuities created during
switching, and adjusting the second generator to reduce the
undesired phase discontinuities.
16. The method according to claim 15, wherein successively
switching between different frequency signals comprises generating
an offset frequency signal and successively combining the offset
frequency signal with a reference frequency signal to produce the
respective different frequency signals.
17. The method according to claim 15, wherein the first generator
comprises a first digital synthesizer; and wherein the second
generator comprises a second digital synthesizer generating an
offset frequency signal and successively combining the offset
frequency signal with a reference frequency signal to produce the
respective different frequency signals.
18. The method according to claim 16, wherein determining the
undesired phase discontinuities created during switching and
adjusting the second generator comprises providing a
self-calibration feedback loop including: a phase locked loop (PLL)
receiving a reference frequency signal; a mixer receiving the
relatively wideband swept frequency signal and a phase reference
signal from the PLL; an analog-to-digital (a/d) converter receiving
an output signal of the mixer; and a controller connected to the
a/d converter and providing a calibration signal to the second
generator.
19. The method according to claim 15, wherein successively
switching between different frequency signals comprises: connecting
a plurality of frequency converters to an output of a reference
frequency signal generator; and coupling an offset frequency signal
to the plurality of frequency converters to successively combine
the offset frequency signal with the reference frequency signal and
produce the different frequency signals.
Description
FIELD OF THE INVENTION
[0001] The present invention relates to the field of communication
systems, and is particularly directed to a digitally controlled
phase continuous synthesizer for relatively high frequency chirp
applications, such as synthetic aperture radar (SAR) and the like,
in which the synthesizer includes continuous phase calibration.
BACKGROUND OF THE INVENTION
[0002] Synthetic aperture radar (SAR) systems, typically located on
board an aircraft or satellite platform, provide SAR imagery of the
radar return signals in both the range dimension and the
cross-range or azimuth dimension. Range resolution is achieved in a
well known manner by using either a high bandwidth fixed frequency
transmit pulse or a frequency modulated (FM) transmit pulse.
Resolution in the cross-range dimension is achieved by synthesizing
a large antenna aperture using the motion of the radar platform.
The key to SAR is the data processing of reflected return data. For
an overview of SAR, reference is made to "An Introduction To
Synthetic Aperture Radar" by W. M. Brown and L. J. Porcello, IEEE
Spectrum (September, 1969), pages 52-62.
[0003] For optimal performance, the frequency content of relatively
high frequency communication signal processing systems, such as
those used for generating wideband chirps for SAR, should be as
pure as possible, in particular, they should exhibit phase
continuity or coherency through the entire output frequency range.
Analog synthesizer-based systems, which offer a relatively wide
tuning range, suffer from arbitrary phase steps when switching
between local oscillators. A direct digital synthesizer (DDS), on
the other hand, provides phase continuity with low noise when
switching, but is capable of operation within a relatively narrow
tuning range (e.g., 100 MHz).
[0004] One technique currently used to generate a wideband chirp
involves multiplying up the output chirp of a DDS to realize the
desired output frequency range of the system. Unfortunately,
successive multiplications also multiply noise by the same factor.
This problem is compounded because radiation requirements typically
limit the choice of DDS to those having relatively low frequency
rates, which means that even higher multiplication factors are
required. Another conventional approach is to limit the frequency
range (width) of the chirp and use receiver processing to resolve
phase errors associated with the discontinuities.
[0005] One conventional approach is disclosed in U.S. Pat. No.
5,878,335 to Kushner which is directed to a low-power digital
frequency synthesizer that combines direct digital frequency
synthesis techniques with serrodyne frequency translation
principles to produce a wideband frequency response with high
spectral purity. A DDS is used to generate a high-resolution analog
carrier signal from a low-speed digital clock signal. The carrier
signal is phase modulated by a low-resolution signal generated from
a high-speed digital clock signal. The modulation signal is a
higher frequency signal than the carrier signal, and the phase
modulation is accomplished by exact decoded gain elements.
SUMMARY OF THE INVENTION
[0006] In view of the foregoing background, it is therefore an
object of the present invention to provide a phase continuous
synthesizer and method in which the synthesizer includes continuous
phase calibration.
[0007] This and other objects, features, and advantages in
accordance with the present invention are provided by an apparatus
for generating a relatively wideband swept frequency signal
including a first generator for generating a first swept frequency
signal, and a second generator successively switching between
different frequency signals while creating undesired phase
discontinuities during switching. A mixer is connected to the first
and second generators for mixing the first swept frequency signal
and the successively switched different frequency signals to
produce the relatively wideband swept frequency signal, and a
calibrator is provided for calibrating the second generator to
reduce the undesired phase discontinuities during switching based
upon the relatively wideband swept frequency signal.
[0008] The calibrator preferably includes a self-calibration
feedback loop including a phase locked oscillator receiving a
reference frequency signal, a mixer receiving the relatively
wideband swept frequency signal and a phase reference signal from
the phase locked loop (PLL), an analog-to-digital (a/d) converter
receiving an output signal of the mixer, and a controller connected
to the a/d converter and providing a calibration signal to the
second generator. The second generator generates an offset
frequency signal and successively combines the offset frequency
signal with a reference frequency signal to produce the
successively switched different frequency signals. The first
generator comprises a first digital synthesizer to generate the
first swept frequency signal.
[0009] The second generator may include a second digital
synthesizer to generate the offset frequency signal, a plurality of
cascaded frequency converters to successively combine the offset
frequency signal with a reference frequency signal to produce the
successively switched different frequency signals, and a controller
for controlling the operation of the second digital synthesizer to
maintain phase continuity between the successively switched
different frequency signals.
[0010] Objects, features, and advantages in accordance with the
present invention are also provided by a method for generating a
relatively wideband swept frequency signal including generating a
first swept frequency signal with a first generator, and
successively switching between different frequency signals with a
second generator while creating undesired phase discontinuities
during switching. The first swept frequency signal is combined with
the successively switched different frequency signals to produce
the relatively wideband swept frequency signal, and the second
generator is calibrated to reduce the undesired phase
discontinuities during switching based upon the relatively wideband
swept frequency signal.
[0011] Successively switching between different frequency signals
preferably includes generating an offset frequency signal and
successively combining the offset frequency signal with a reference
frequency signal to produce the respective different frequency
signals. Successively switching between different frequency signals
may include connecting a plurality of frequency converters to an
output of a reference frequency signal generator and coupling an
offset frequency signal to the plurality of frequency converters to
successively combine the offset frequency signal with the reference
frequency signal to produce the different frequency signals. The
first generator may be a first digital synthesizer, and the second
generator may be a second digital synthesizer generating the offset
frequency signal.
[0012] Calibrating the second generator preferably includes
comparing the phase of the relatively wideband swept frequency
signal before and after successively switching between different
frequency signals to determine the undesired phase discontinuities
created during switching, and adjusting the phase of the offset
frequency signal generated by the second digital synthesizer to
reduce the undesired phase discontinuities created during
switching. The calibration may include providing a self-calibration
feedback loop including a phase locked loop (PLL) receiving the
reference frequency signal, a mixer receiving the relatively
wideband swept frequency signal and a phase reference signal from
the PLL, an analog-to-digital (a/d) converter receiving an output
signal of the mixer, and a controller connected to the a/d
converter and providing a calibration signal to the second digital
synthesizer.
BRIEF DESCRIPTION OF THE DRAWINGS
[0013] FIG. 1 is schematic block diagram of a phase continuous
self-calibrating synthesizer according to the present
invention.
[0014] FIG. 2 is a timing diagram of the phase continuous swept
frequency signal output from the synthesizer of FIG. 1.
[0015] FIGS. 3A and 3B are enlarged views of a portion of the
timing diagram of FIG. 2 illustrating detail of the phase
continuous swept frequency signal at a switching point.
[0016] FIG. 4 is schematic block diagram of a phase continuous
synthesizer including phase coasting according to another
embodiment of the present invention.
[0017] FIG. 5 is a more detailed schematic block diagram of the
phase coasting unit of the phase continuous synthesizer of FIG.
4.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0018] The present invention will now be described more fully
hereinafter with reference to the accompanying drawings, in which
preferred embodiments of the invention are shown. This invention
may, however, be embodied in many different forms and should not be
construed as limited to the embodiments set forth herein. Rather,
these embodiments are provided so that this disclosure will be
thorough and complete, and will fully convey the scope of the
invention to those skilled in the art. Like numbers refer to like
elements throughout, and prime notation is used to indicate similar
elements in alternative embodiments.
[0019] Before describing in detail the phase-continuous frequency
synthesizer of the present invention, it should be noted that the
invention resides primarily in a modular arrangement of
communication circuits and components and an associated controller
therefor, that controls the operations of such circuits and
components. In a practical implementation that facilitates their
being packaged in a hardware-efficient equipment configuration,
this modular arrangement may be implemented via an application
specific integrated circuit (ASIC) chip set, for example.
[0020] Consequently, the architecture of the arrangement of
circuits and components has been illustrated in the drawings by a
readily understandable block diagram, which shows only those
specific details that are pertinent to the present invention, so as
not to obscure the disclosure with details which will be readily
apparent to those skilled in the art having the benefit of the
description herein. Thus, the block diagram illustration is
primarily intended to show the major components of the invention in
a convenient functional grouping, so that the present invention may
be more readily understood.
[0021] Referring initially to FIG. 1, an embodiment of the
phase-continuous frequency synthesizer of the present invention
will now be described. The phase-continuous frequency synthesizer
is diagrammatically illustrated as comprising a controlled `fine`
tune direct digital synthesizer (DDS) 10 that is operative, under
the control of a controller 100, to produce a linearly swept or
ramp frequency output. By `fine` tune is meant that DDS 10 has the
finest spectral granularity of various frequency tuning components
of the system. As a non-limiting example, the frequency ramp
produced by DDS 10 may be swept over a range from 100 to 200 MHz.
Thus, in this example, the `inest` tuning range within the system
is 100 MHz.
[0022] The DDS 10 is coupled to a prescribed reference frequency
(e.g., 100 MHz) produced by a phase locked oscillator (PLO) 20,
which is coupled to receive a frequency reference from an external
source (not shown). This reference frequency is used to synchronize
the various components of the synthesizer, and may be provided to
the PLO 20 via a power divider 46.
[0023] The fine tune DDS 10 is coupled to a mixer 30. Frequency
translation oscillator 50 is operative to produce a relatively high
radio frequency (RF) output, e.g., an RF frequency on the order of
1.0 GHz. The output of mixer 30 is coupled to a first band pass
filter 40 which is then coupled to a frequency mixer 70, which is
also coupled to the output of a switch (S1) 80. Switch 80 is
operative under processor control, via controller 100, to switch
among a plurality of coarse frequency inputs (four in the
illustrated example at 81, 82, 83 and 84), that are used to define
a coarse range of operation of the synthesizer (the fine tuning
range of which is established by DDS 10, as described above).
[0024] For this purpose, the respective inputs 81, 82, 83 and 84 of
switch 80 are coupled to PLO 20 and to a set of cascaded frequency
offset converters 110, 120 and 130. Each frequency offset converter
produces an output frequency that is equal to the sum of its input
frequencies and under the phase control of the offset frequency DDS
140. PLO 20 generates a base coarse frequency F0, while the
frequency offset converters 110, 120 and 130 produce respective
coarse frequencies F1, F2 and F3, that are combinations of the base
frequency F0 and an a coarse offset frequency Foff generated by an
offset DDS 140. Offset DDS 140 is operative under the control of
the controller 100 to produce the coarse offset frequency Foff
equal to the sweep range of fine tune DDS 10, which, in the present
example, may be 50 MHz, as described above.
[0025] The output frequency F1 produced by frequency offset
converter 110 is equal to the sum of the offset frequency Foff
supplied by DDS 140 and the base frequency F0 supplied by PLO 20;
the output frequency F2 produced by offset converter 120 is equal
to the sum of the offset frequency Foff and the frequency F1
supplied by offset converter 110; and the output frequency F3
produced by offset converter 130 is equal to the sum of the offset
frequency Foff and the frequency F2 supplied by offset converter
120. Under the control of controller 100, the phase of the offset
frequency Foff produced by offset DDS 140 is controllably
adjustable, so as to provide for phase-continuity at the instances
of switching among the respective input frequencies to switch 80.
In particular, controller 100 sets the phase of the offset
frequency Foff produced by offset DDS 140 to be equal to the
negative of the measured phase error (discussed below), so that at
the instant of switching between any of its inputs the new
frequency to which switch 80 switches will be at zero degrees and
phase continuous with the frequency from which switch 80 has
switched. Of course, more than one switch 80 may be provided in
combination with an associated multiplier and mixer to increase the
sweep range of the chirp, as would be appreciated by the skilled
artisan.
[0026] Mixer 70 provides the output chirp or swept frequency signal
to another band pass filter 42 and a power divider 44. The chirp is
then typically provided to an up converter, transmitter and antenna
as would be appreciated by the skilled artisan. However, undesired
phase discontinuities in the chirp would normally occur during
switching as will be described below.
[0027] Operation of the frequency synthesizer of FIG. 1 will now be
described. For purposes of the present example, the offset
frequency Foff is 50 MHz, as referenced above. Initially, at time
t0 the phase of the offset frequency Foff produced by offset DDS
140 is controllably pre-set at zero phase. Also, switch 80 is
coupled to receive the frequency F0 from PLO 20. As pointed out
above, controller 100 sets the phase of the offset frequency Foff
produced by offset DDS 140 to be equal to the negative value of
overall phase error, so that at the instant of switching between
any of their inputs the new frequencies to which switches 80
transition will be at zero degrees and phase continuous with the
previous frequency. (It is to be understood that by "phase" is
meant the relative difference between the pre-switched frequency
and the post-switched frequency at the instant of switching, i.e.,
zero degrees difference and phase continuous.)
[0028] Whenever a transition is made to a new coarse frequency, the
fine tune DDS 10 is reset to the beginning of its sweep and
thereupon proceeds to ramp over its sweep range (100 MHz in the
present example). Upon DDS 10 reaching the upper end of its sweep
range, switch 80 switches to the next offset frequency F1 following
F0 and the sweep of DDS 10 is restarted. The switch 80 sequentially
transitions through its coarse frequency inputs 81-82-83-84.
Therefore, referring to the timing diagram of FIG. 2, at time t0,
the output of the synthesizer is equal to the product of the
frequency output Fx of the mixer 30 plus the lowest coarse
frequency F0. Between time t0 and time t1, as the frequency output
of the fine tune DDS 10 ramps over its 100 Mhz range, the output of
the synthesizer is linearly swept from Fx+F0 to Fx+F0+Foff which
equals Fx+F1. Upon reaching the frequency Fx+F0+Foff at time t1,
fine tune DDS 10 returns to the base translation frequency Fx.
However, since switch 80 is switched from input 81 to input 82, the
output of the synthesizer begins sweeping from Fx+F1 to Fx+F1+Foff,
and so on as the switch 80 is stepped through its additional inputs
83 and 84.
[0029] Referring to FIG. 3A, the enlarged portion 3 of the timing
diagram of FIG. 2 will be described. As discussed above, during the
switching transients, e.g. at time t1, undesired phase
discontinuities A would normally be created. Because phase
continuity at the switching transients is needed to reduce SAR
image degradation, such a synthesizer would typically require
pre-mission calibration including precise phase measurements made
on ground. Such measurements are intended to match the phase of the
chirp across PLO 20 switching interval. This calibration and
measurements would increase pre-mission set up time and may become
a drift term over time and temperature after calibration.
[0030] In the present invention, the phase continuous synthesizer
includes a calibrator 90 to reduce the undesired phase
discontinuities created during switching. Calibrating includes
comparing the phase of the relatively wideband swept frequency
signal output at the power divider 44 before and after successively
switching between different frequency signals to determine the
undesired phase discontinuities created during switching, and
adjusting the phase of the offset frequency signal Foff generated
by the offset DDS 140 to reduce the undesired phase discontinuities
created during switching. The calibration includes a
self-calibration feedback loop defined by a phase locked loop (PLL)
92 receiving the reference frequency signal, a mixer 94 receiving
the chirp or swept frequency signal output at the power divider 44
and a phase reference signal from the PLL 92, an analog-to-digital
(a/d) converter 96 receiving an output signal of the mixer, and the
controller 100 which is connected to the a/d converter and provides
a calibration signal to the offset DDS 140. Referring to FIG. 3B,
the phase discontinuity B created during switching is reduced or
eliminated at time t1 in the next chirp.
[0031] The PLL 92 is an auxiliary synthesizer added as a phase
reference and is tuned to the frequency of the chirp at each switch
point of the switch 80. The phase of the chirp is compared with the
PLL 92 before and after switching to determine the change in phase
due to switching. In other words, the instantaneous phase
difference between chirp and PLL 92 is measured before and after
switching. Settling time is not a challenge because the PLL 92 is
set to frequency well in advance of the calibration need. The
result is correlated in the controller 100 against a mathematical
chirp waveform and the error is returned as feedback to adjust the
phase of the offset DDS 140. Thus, the next chirp will reduce the
phase difference during switching. The controller 100 determines
the phase change needed, provides for averaging and resolves any
0.degree./180.degree. ambiguity. Multiple settings may be provided
in an adaptable lookup table to provide calibration at each switch
point.
[0032] A plurality of approaches would be appropriate for
determining the phase error. For example, a correlation method to
maximize the cross-correlation between the desired and measured
phase. Such a cross-correlation would be calculated between ideal
and measured or between mirror-image of pre-switched and
post-switched waveforms. Another approach may be to calculate the
standard deviation on the phase difference between before and after
switching. Also, in yet another approach, the arcsine could be
taken of mirrored pre-switch and non-mirrored post switch waveforms
while linear and quadratic time is removed from the phase function.
In these various approaches, linearization may be needed.
[0033] In sum, the fine tune DDS 10 produces a chirp but does not
fully cover the required output sweep range. Switch 80 selects
offset to put the chirp within the output sweep range. At switch
points, DDS 10 is at highest frequency and is switched to lowest
frequency as the coarse step is made with the switch 80. The phase
transient normally produced is adjusted out by changing the phase
of the offset DDS 140 during the switching interval.
[0034] Previous embodiments utilized short duration RF switches to
minimize the switching interval and reduce corresponding phase
disturbances. However, such switches may not be readily available,
and typically do not have good isolation from the switching signal
to output. Phase continuity on either side of a switching transient
is provided by phase calibration of the offset DDS 140, but
switching is not instantaneous. Break-before-make switches have
reduced output during switching and transients caused by varying
return loss. Make-before-break switches have increased output and
reflection induced phase disturbances during switching. So the
output of the synthesizer has continuous phase except during the
switching interval where the transient distorts the phase.
[0035] Referring to FIGS. 4 and 5, another embodiment of the phase
continuous synthesizer will be described. In the simplified
diagram, the calibrator 90 of the previous embodiment is not
illustrated but may certainly be included in the present
embodiment. Here, a phase coasting unit 150 is connected downstream
of the mixer 70 and band pass filter 42 to reduce the undesired
phase discontinuities created during switching in the output
frequency signal. In other words, the phase coasting unit will
track the phase of the output frequency signal and coast over the
region of phase/amplitude disturbance. The phase coasting unit 150
is preferably a high gain second or third-order phase locked loop
(PLL) so there is little or no static phase error during frequency
ramp. As an example of a third-order loop, the phase coasting unit
150 includes a phase detector 152, a switch 154 connected to the
phase detector and controlled to open during a switching interval,
a plurality of integrators 156, 158 downstream from the switch, and
a voltage controlled oscillator 160 downstream from the plurality
of integrators.
[0036] The analog switch 154 opens during the switching interval
and holds the integrators 156, 158 constant. The VCO 160 continues
to ramp since the conditions are the same as if ramp were present
during switching. Also, the controller 100 may pretune the phase
coasting loop 150 to set initial conditions at the beginning of the
chirp for fast acquisition. The output of the phase coasting unit
150 is a ramp of frequency due to the integrators 156, 158 having
fixed or zero voltage at their inputs. The blocks 151 and 153
schematically illustrate the swept frequency signal before and
after phase coasting respectively.
[0037] The phase continuous self-calibrating synthesizer and method
of the present invention can increase the performance of SAR. The
invention improves the ability to widen the chirp range and allows
higher radiation tolerant technologies to be used without sweep
range degradation. The present invention also allows the use of
frequency division to reduce spurious frequencies rather than the
conventional approach of multiplication. The coasting loop provides
a near-ideal chirp that is continuous without transients. Also, the
typical stairstep frequency ramp from the DDS is smoothed out and
appears as infinite granularity.
[0038] Other features of the phase continuous synthesizer may be
described in greater detail in copending applications to the
Assignee entitled "PHASE-CONTINUOUS FREQUENCY SYNTHESIZER"
(attorney docket No. 51327) and "PHASE CONTINUOUS SYNTHESIZER WITH
PHASE COASTING AND ASSOCIATED METHODS" (attorney docket No. 51358),
and filed concurrently herewith, the entire disclosures of each of
which are incorporated by reference herein in their entirety.
[0039] Many modifications and other embodiments of the invention
will come to the mind of one skilled in the art having the benefit
of the teachings presented in the foregoing descriptions and the
associated drawings. Therefore, it is understood that the invention
is not to be limited to the specific embodiments disclosed, and
that modifications and embodiments are intended to be included
within the scope of the appended claims.
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