U.S. patent application number 11/043086 was filed with the patent office on 2005-06-16 for power-supply device.
Invention is credited to Hayashi, Katsunori, Kanouda, Akihiko, Sase, Takashi, Tateno, Koji, Yoshida, Shinichi.
Application Number | 20050127886 11/043086 |
Document ID | / |
Family ID | 29717440 |
Filed Date | 2005-06-16 |
United States Patent
Application |
20050127886 |
Kind Code |
A1 |
Sase, Takashi ; et
al. |
June 16, 2005 |
Power-supply device
Abstract
A step-down type DC-DC power supply device implements both the
stabilization of the control loop and the responsibility at the
same time. In the power-supply device, an output power signal is
fed back to an error amplifier after having passed through a CR
smoothing filter provided independently of a power LC smoothing
filter. Also, independently of the duty controls over Power
MOSFETs, i.e., upper-side/lower-side semiconductor switching
components in the steady state, an output from the power LC
smoothing filter is added to an upper and lower limit-mode-equipped
control circuit, thereby, at the transient state, forcefully
setting the duty .alpha. at either 0% or 100%.
Inventors: |
Sase, Takashi; (Hitachi,
JP) ; Tateno, Koji; (Hitachi, JP) ; Kanouda,
Akihiko; (Hitachinaka, JP) ; Hayashi, Katsunori;
(Odawara, JP) ; Yoshida, Shinichi; (Takasaki,
JP) |
Correspondence
Address: |
MATTINGLY, STANGER, MALUR & BRUNDIDGE, P.C.
1800 DIAGONAL ROAD
SUITE 370
ALEXANDRIA
VA
22314
US
|
Family ID: |
29717440 |
Appl. No.: |
11/043086 |
Filed: |
January 27, 2005 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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11043086 |
Jan 27, 2005 |
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10937397 |
Sep 10, 2004 |
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6879137 |
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10937397 |
Sep 10, 2004 |
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10462680 |
Jun 17, 2003 |
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6798180 |
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Current U.S.
Class: |
323/282 |
Current CPC
Class: |
Y02B 70/10 20130101;
H02M 3/33507 20130101; Y02B 70/1466 20130101; H02M 3/1588 20130101;
H02M 3/1584 20130101 |
Class at
Publication: |
323/282 |
International
Class: |
G05F 001/40 |
Foreign Application Data
Date |
Code |
Application Number |
Jun 17, 2002 |
JP |
2002-175172 |
Claims
1. A power-supply device including a step-down DC-DC converter,
comprising: power semiconductor switching components, driving means
for driving said power semiconductor switching components, a
pulse-width modulation oscillator to supply said driving means with
a driving signal, an error amplifier to supply said pulse-width
modulation oscillator with an error signal indicating a comparison
result between a reference value and an output power, wherein a
control loop of said power-supply device includes a filter provided
independently of a power output filter through which said output
power passes, an output signal corresponding to said output power
being fed back to said error amplifier after having passed through
said independently provided filter, and wherein said power output
filter is an LC filter comprised of an inductor and a capacitor,
said independently provided filter being a CR filter comprised of a
capacitor and a resistor, said CR filter being connected to both
ends of said inductor of said power output filter, and said output
signal being fed back to said error amplifier after having passed
through said CR filter, and a transient variation detecting
circuit, said transient variation detecting circuit: detecting said
output power from an output terminal of said power output filter,
and, if said output power has been found to exceed a predetermined
upper-limit value, outputting a signal for setting the duty .alpha.
of said pulse-width modulation oscillator at 0%, and, if said
output power has been found to be lower than a predetermined
lower-limit value, outputting a signal for setting said duty
.alpha. of said pulse-width modulation oscillator at 100%.
Description
[0001] The present application is a continuation of application
Ser. No. 10/937,397, filed Sep. 10, 2004, which is a continuation
of application Ser. No. 10/462,680, filed Jun. 17, 2003, now U.S.
Pat. No. 6,798,180, the contents of which are incorporated herein
by reference.
BACKGROUND OF THE INVENTION
[0002] The present invention relates to a power-supply device
wherein, independently of a power LC smoothing filter, a signal is
caused to pass through a CR smoothing filter and is then fed back
so that the control loop will be stabilized.
DESCRIPTION OF THE RELATED ART
[0003] A prior art on the loop stabilizing method for a
power-supply device has been described in "Low-Voltage On-Board
DC/DC Modules for Next Generations of Data Processing Circuits",
Zhang et al., IEEE Tran. on Power Elect. Vol. 11, No. 2, March
1996. In the power-supply device according to the prior art, a
signal is fed back to an error amplifier from a power LC smoothing
filter. Then, the error amplifier compensates the phase, thereby
implementing the stabilization of the control loop. In this prior
art, an aluminum electrolytic capacitor is used as the power LC
smoothing filter.
[0004] U.S. Pat. No. 5,877,611 discloses a power supply system in
which an output of a CR smoothing filter connected across an
inductor of an output LC smoothing filter is fed back to an error
amplifier having a low input impedance. In the U.S. patent prior
art, voltage and current signals of a power supply output are
extracted using the CR smoothing filter, so that the resistance
value of the CR smoothing filter must be set to be small. The
component constants of the CR smoothing filter are a capacitance of
0.47 .mu.F and a resistance of 100.OMEGA.. Accordingly, the CR
smoothing filter having such constants cannot be formed on chip in
a power supply IC and must be formed externally of the IC chip,
resulting in a problem that the power supply device cannot be made
in small size totally.
SUMMARY OF THE INVENTION
[0005] In order to downsize the power-supply device, instead of
using the aluminum electrolytic capacitor as the power LC smoothing
filter, there has occurred a necessity for using a ceramic
capacitor of a chip-part as the power LC smoothing filter. However,
the equivalent series resistance (ESR) of the chip ceramic
capacitor is equal to several m.OMEGA., which is considerably
small. What is more, the ceramic capacitors are connected in
parallel under an actual use condition. Accordingly, the total of
the ESRs in this case becomes less than 1 m.OMEGA., which is even
smaller. This makes it impossible to expect the damping of the ESR
as is expected in the case of using the aluminum electrolytic
capacitor. Consequently, it becomes difficult to stabilize the
control loop.
[0006] In the above-described prior art, when using the ceramic
capacitor with the small ESR as the power LC smoothing filter, it
becomes impossible to expect the damping effect of the ESR. This
causes a signal to oscillate, thereby making the phase compensation
difficult. Also, if, in the prior art, it were to become possible
to implement the phase compensation by narrowing the operation
bandwidth of the error amplifier, a response from the power-supply
is delayed exceedingly. Moreover, in modifying the LC smoothing
filter's constants, there exists a troublesome task of adjusting
the phase compensation condition of the error amplifier on each
that occasion.
[0007] It is an object of the present invention to provide a
power-supply device that employs a novel control method where,
independently of a power LC smoothing filter, a signal is caused to
pass through a CR smoothing filter and is then fed back so that the
control loop will be stabilized.
[0008] A power-supply device according to one aspect of the present
invention is as follows: In the control loop of the power-supply
device of a step-down type DC-DC converter, a CR smoothing filter
is provided independently of a power LC smoothing filter. Moreover,
a signal corresponding to the output power is fed back to an error
amplifier after having passed through the CR smoothing filter.
[0009] Also, a power-supply device according to another aspect of
the present invention includes the following unit: Independently of
the duty controls over Power MOSFETs, i.e., upper-side/lower-side
semiconductor switching components in the steady state, the unit
adds the output from a power LC smoothing filter to an upper and
lower limit value detecting circuit, thereby, at the transient
state, forcefully setting the duty at either 0% or 100%.
[0010] Moreover, a power-supply device according to still another
aspect of the present invention is as follows: The power-supply
device includes power-supply device units prepared in plural
number. In order to perform a parallel operation of these
power-supply device units, the power-supply device further includes
an oscillator and a phase shift circuit that the plural
power-supply device units have in common. Furthermore, in the
steady state, phases of driving pulses of upper-side/lower-side
Power MOSFETs in the respective power-supply device units are
respectively shifted to phases that result from dividing
360.degree. by the number of the parallelism. At the transient
state, all of the parallel power-supply device units are operated
by driving pulses of one and the same phase.
[0011] Other objects, features and advantages of the invention will
become apparent from the following description of the embodiments
of the invention taken in conjunction with the accompanying
drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
[0012] FIG. 1 is a circuit block diagram for illustrating a
power-source device of a first embodiment in the present
invention;
[0013] FIG. 2 is an explanatory diagram for explaining an IC where
a CR filter is built in a semiconductor chip in the power-supply
device in FIG. 1;
[0014] FIG. 3 is a circuit block diagram for illustrating a
power-supply device of a second embodiment in the present
invention;
[0015] FIG. 4 is an explanatory diagram for explaining an IC where
a CR filter is built in a semiconductor chip in the power-supply
device in FIG. 3;
[0016] FIG. 5 is a circuit block diagram for illustrating a
power-supply device of a third embodiment in the present
invention;
[0017] FIG. 6 is a circuit diagram for illustrating the details in
FIG. 5;
[0018] FIG. 7 is a diagram for illustrating the operation state
mode in FIG. 6;
[0019] FIG. 8 is a circuit block diagram for illustrating a
power-supply device of a fourth embodiment in the present
invention;
[0020] FIG. 9 is a circuit block diagram for illustrating another
power-supply device of the fourth embodiment;
[0021] FIG. 10 is a circuit block diagram for illustrating still
another power-supply device of the fourth embodiment;
[0022] FIG. 11 is a circuit block diagram for illustrating a
multi-phase power-source device of a fifth embodiment in the
present invention;
[0023] FIG. 12 is a circuit diagram for illustrating the details in
FIG. 11;
[0024] FIG. 13 is a diagram for illustrating the operation state
mode in FIG. 12;
[0025] FIG. 14 is a circuit block diagram for illustrating an
example of the chip configuration of a power-source device of a
sixth embodiment in the present invention;
[0026] FIG. 15 is an explanatory diagram for explaining a VID code
input D/A converter applied to FIG. 14;
[0027] FIG. 16 is a circuit block diagram for illustrating a
multi-phase compatible chip of a seventh embodiment in the present
invention;
[0028] FIG. 17 is an explanatory diagram for explaining the printed
wiring board implementation of a power-source control IC of an
eighth embodiment;
[0029] FIG. 18 is an explanatory diagram for explaining a HDD
device of a ninth embodiment;
[0030] FIG. 19 is an explanatory diagram for explaining a tenth
embodiment in the present invention;
[0031] FIG. 20 is an explanatory diagram for illustrating another
embodiment of a pulse-width modulation oscillator PWM;
[0032] FIG. 21 is an explanatory diagram for explaining an eleventh
embodiment in the present invention applied to a
commercially-available power-source IC; and
[0033] FIG. 22 is a diagram for illustrating the operation state
mode in FIG. 21.
DETAILED DESCRIPTION OF THE EMBODIMENTS
[0034] Referring to the accompanying drawings, the explanation will
be given below concerning the details of the present invention.
[0035] Embodiment 1
[0036] FIG. 1 illustrates a power-supply device of the present
embodiment. In FIG. 1, reference notations Vi and Vo denote an
input terminal and an output terminal, respectively. An upper-side
Power MOSFET Q1 is connected to the input terminal Vi, and a
lower-side Power MOSFET Q2 is connected to a ground potential side.
An LC smoothing filter, i.e., a power output filter consisting of
an inductor L and a capacitor Co, and a CR smoothing filter
consisting of a resistor R and a capacitor C are connected in
parallel to a midpoint of the Power MOSFETs Q1 and Q2. Moreover,
the output terminal Vo is connected to a midpoint of the LC
smoothing filter, and one input (-) of an error amplifier EA is
connected to a midpoint of the CR smoothing filter. Here, the
capacitor Co of the LC smoothing filter is a chip ceramic
capacitor.
[0037] Also, a reference voltage Vref is connected to the other
input (+) of the error amplifier EA. A pulse-width modulation
(abbreviated as PWM) oscillator PWM, and gates of the Power MOSFETs
Q1 and Q2 via a driver DRV are connected to an output of the error
amplifier EA. The Power MOSFETs Q1 and Q2 are driven in opposite
phases to each other, and thus are electrically conducted
alternately. In the present embodiment, an output voltage Vout is
smaller than an input voltage Vin.
[0038] Next, the explanation will be given below regarding the
circuit operation in FIG. 1. The input voltage Vin applied to the
input terminal Vi is converted into a voltage by on/off controls
over the upper-side Power MOSFET Q1 and the lower-side Power MOSFET
Q2 via the CR smoothing filter. This converted voltage VFB is
compared with the reference voltage Vref by the error amplifier EA.
As a consequence, an error voltage is generated in a state of being
amplified at the output of the error amplifier EA. This error
voltage is converted into a PWM pulse by the pulse-width modulation
oscillator PWM. This PWM pulse is further converted by the driver
DRV into an on/off-time ratio (i.e., duty: .alpha.) at which the
driver DRV drives the upper-side Power MOSFET Q1 and the lower-side
Power MOSFET Q2. Moreover, a negative-feedback control is performed
over the PWM pulse so that the error voltage becomes equal to 0. As
a result of this, the converted voltage VFB becomes equal to the
reference voltage Vref. In this case, the converted voltage VFB
acquired through the CR smoothing filter in the steady state is
proportional to the duty .alpha. of the input voltage Vin.
Consequently, the following relational expression holds:
VFB=Vref=.alpha..multidot.Vin
[0039] where the duty .alpha. assumes a value in the range of 0 to
1, since .alpha. is defined as the on-time/(.alpha. total of the
on-time and the off-time).
[0040] In the case of the ordinary step-down type converter, it has
been found out that the voltage-converted ratio in the steady state
is equal to the ratio, i.e., the duty, between the output voltage
and the input voltage. Accordingly, assuming that the input voltage
is Vin and the duty is .alpha., the output from the LC smoothing
filter, i.e., the output voltage Vout acquired at the output
terminal Vo, can be determined by a relational expression:
Vout=.alpha..multidot.Vin.
[0041] From the above-described 2 expressions, the following
relational expression holds:
Vout=VFB=.alpha..multidot.Vin.
[0042] Consequently, even if no direct negative-feedback control is
performed over the output from the LC smoothing filter, if an
indirect control over the duty a using some other method proves
successful, this successful indirect control becomes equivalent to
a direct control over the output voltage Vout at the output
terminal Vo. As a result, it becomes possible to acquire, at the
output terminal Vo, the voltage that is proportional to the duty
.alpha. of the input voltage Vin. In other words, the Power MOSFETs
Q1 and Q2 are driven, thereby performing the negative-feedback
control over the output from the CR smoothing filter. This
operation allows the desired voltage, which is proportional to the
duty .alpha. of the input voltage Vin, to be also acquired at the
output from the LC smoothing filter as the output voltage Vout.
[0043] As the voltage converting method based on the duty control
over the upper-side Power MOSFET Q1 and the lower-side Power MOSFET
Q2, the present embodiment is a primary-delay control method where
the CR smoothing filter is used for the control loop. Accordingly,
since there exists none of the secondary delay by the LC smoothing
filter as was found in the prior art, the control loop does not
become the oscillating system. This prevents the oscillating
waveform from occurring in the output, thereby making the loop
stable. Consequently, according to the present embodiment, even if
the chip ceramic capacitor with a small ESR is used as the
capacitor of the LC smoothing filter, it is possible to stabilize
the control loop.
[0044] Next, the explanation will be given below concerning the
large-or-small relationship among the corner frequencies and the
switching frequency of the above-described 2 smoothing filters.
Let's assume that the corner frequency of the CR smoothing filter
and that of the LC smoothing filter are equal to fCR and fLC
respectively, and that the switching frequency is equal to fSW.
Then, setting the relationship among these frequencies as
fLC<fCR<fSW makes it possible to ensure the stability of the
loop. Moreover, from this relationship, the feedback from the CR
smoothing filter results in a higher operation frequency as
compared with the feedback from the LC smoothing filter, which
allows the implementation of the high-speed response. Also, fLC and
fCR are set as frequencies that are different to some extent. This
setting, even if the LC smoothing filter's constants are modified,
makes it unnecessary to change the CR smoothing filter's constants,
thereby allowing an increase in the degree-of-freedom of the
design. With respect to the high-speed operation of a 1-to-6-MHz
switching frequency, values usable as the LC smoothing filter's
constants and the CR smoothing filter's constants are, e.g., 0. 2
.mu.H, 220 .mu.F, and 20 pF, 200 k.OMEGA., respectively. If the
values of these capacitors and this resistor are of these orders,
it becomes possible to mount (i.e. on-chip) the CR smoothing filter
on a semiconductor integrated circuit chip, thereby making
externally-attached components unnecessary. This means the
following: By merely replacing the power-supply device illustrated
in FIG. 1 by an IC whose terminal location is the same (i.e.,
pin-compatible) as that of the power-supply control IC in the prior
art, the printed wiring board in the prior art can be utilized with
no modification added thereto.
[0045] FIG. 2 is an explanatory diagram for explaining the chip
layout in the case where, in the power-supply device in FIG. 1, the
CR smoothing filter is built in a semiconductor chip. In FIG. 2,
reference notations C and R denote a built-in capacitor and a
built-in resistor, respectively. These components are mounted on a
semiconductor board that is the same as the one that mounts thereon
the error amplifier EA, the pulse-width modulation oscillator PWM,
the driver DRV, and the Power MOSFETs Q1 and Q2.
[0046] So far, the explanation has been given selecting, as the
example, the CR smoothing filter whose output is fed back to the
error amplifier in the control loop. Instead of the CR smoothing
filter, however, the use of another high-response filter circuit
allows the acquisition of basically the same effects. Also,
although the explanation has been given selecting the Power MOSFETs
as the example of the semiconductor switching components, the IGBTs
may be used instead.
[0047] Embodiment 2
[0048] FIG. 3 illustrates the present embodiment. In FIG. 3, the
same reference notations are attached to the same configuration
components in FIG. 1. The point in which FIG. 3 differs from FIG. 1
is that the CR smoothing filter is set up at both ends of the
inductor L of the LC smoothing filter. In the present embodiment,
since the electrostatic capacitance of the capacitor Co of the
output LC smoothing filter is large, the inductor-connected edge
side of the capacitor Co can also be regarded as the ground
potential. The present embodiment also allows the acquisition of
basically the same effects in FIG. 1. Furthermore, the present
embodiment makes it possible to perform the negative feedback of an
infinitesimal capacitance change caused by a temperature change in
the capacitor Co of the LC smoothing filter. Consequently, even if
the chip ceramic capacitor with a small ESR is used, the present
embodiment permits an enhancement in the stability of the control
loop. In this case as well, the constants of the embodiment in FIG.
1 are usable as the CR smoothing filter's constants. FIG. 4
illustrates an explanatory diagram for explaining the chip layout
in the case where, in the power-supply device in FIG. 3, the CR
smoothing filter is built in a semiconductor chip.
[0049] Embodiment 3
[0050] FIG. 5 illustrates a power-supply device obtained by further
providing a transient variation detecting circuit TVD into the 1st
embodiment. This transient variation detecting circuit TVD controls
the duty of the pulse-width modulation oscillator PWM by detecting
a transient load variation between the output voltage Vout at the
output terminal Vo and a voltage that results from adding a upper
and lower limit-voltage width .+-..DELTA. to the reference voltage
Vref. FIG. 6 illustrates a concrete example of the pulse-width
modulation oscillator PWM and that of the transient variation
detecting circuit TVD.
[0051] In FIG. 6, the pulse-width modulation oscillator PWM is a
variable oscillator including a voltage-to-current converting
circuit V/I, current-source MOSs 110, 120, inverters INV11, INV12,
a capacitor 105, and a flip-flop FF. Also, the transient variation
detecting circuit TVD includes comparators CMP1, CMP2, switching
MOSs SW1 to SW4, constant current-sources I1 to I4, and inverters
INV1 to INV8.
[0052] The transient variation detecting circuit TVD includes a
wind comparator consisting of the 2 comparators CMP1, CMP2. The
circuit TVD compares the output voltage Vout with the voltage that
results from adding the upper and lower limit-voltage width
.+-..DELTA. to the reference voltage Vref, thereby detecting the
operation state of the output voltage Vout and determining the
pulse duty .alpha. of the oscillator PWM indicated in FIG. 7. This
means that, in the transient variation detecting circuit TVD, the
control method in the steady state and the one at the transient
state are switched into control modes that match the operation
state.
[0053] From the outputs from the 2 comparators CMP1, CMP2, the
following 3-way information is acquired: (a) a case where the load
current is decreased, (b) the steady state, (c) a case where the
load current is increased. Using FIG. 7, these cases will be
explained below:
[0054] The case (a) is under a condition
Vout.gtoreq.(Vref+.DELTA.). At this time, the output duty .alpha.
of the pulse-width modulation oscillator PWM is forcefully set at
0%. For this purpose, the switching MOSs SW1 and SW4 are turned on,
and the switching MOSs SW3 and SW2 are turned off. As a result, a
current from the constant current-source I1 is added to a current
from the current-source MOS 110, then flowing together to the
inverter INV11. A current from the constant current-source I4 is
subtracted by a current to the current-source MOS 120, so that the
current value flowing to the inverter INV12 becomes equal to 0.
Consequently, the upper-side Power MOSFET Q1 is switched off, and
the lower-side Power MOSFET Q2 is switched on, which, eventually,
makes the duty .alpha. equal to 0%. In this case, in order to set
the duty 60 at 0% completely, it is preferable that current values
from the constant current-sources I1 to I4 be each set at the total
current of differential pair operation currents of the
voltage-to-current converting circuit V/I.
[0055] The case (b) is under a condition (Vref+.DELTA.)
>Vout>(Vref-.DELTA.). In this case, all of the switching MOSs
SW1 to SW4 are turned off, and are operated in accordance with a
current ratio determined by a control instruction from the error
amplifier EA. Since this current ratio is equal to the rate of the
duty, the voltage that is proportional to the duty a of the input
voltage Vin can be acquired as the output voltage Vout.
[0056] The case (c) is under a condition
Vout.ltoreq.(Vref-.DELTA.), where the duty .alpha. is forcefully
set at 100%. In this case, the switching MOSs SW3 and SW2 are
turned on, and the switching MOSs SW1 and SW4 are turned off. As a
result, a current from the constant current-source I3 is added to
the current from the current-source MOS 120, then flowing together
to the inverter INV12. A current from the constant current-source
I2 is subtracted by the current to the current-source MOS 110, so
that the current value flowing to the inverter INV11 becomes equal
to 0. Consequently, the upper-side Power MOSFET Q1 is switched on,
and the lower-side Power MOSFET Q2 is switched off, which,
eventually, makes the duty .alpha. equal to 100%. In this case, in
order to set the duty .alpha. at 100% completely, it is preferable
that the current values from the constant current-sources I1 to I4
be each set at the total current of the differential pair operation
currents of the voltage-to-current converting circuit V/I.
[0057] The present embodiment forcefully switches the duty .alpha.
of the pulse-width modulation oscillator PWM to either 0% or 100%
so that the voltage generated at the output terminal Vo at the
transient state will fall within the upper and lower limit-voltage
width .+-..DELTA. added to the reference voltage Vref. This rapidly
suppresses the output voltage Vout within (Vref .+-..DELTA.).
Moreover, when the operation state enters the steady state, the
present embodiment causes the output voltage to be stabilized as
the voltage that is proportional to the duty .alpha. of the input
voltage.
[0058] In this way, in the present embodiment, the control mode is
automatically switched depending on whether the operation state is
the transient state or the steady state. As a consequence, with
respect to even, e.g., an about 10A transient load variation having
the high current slew rate (i.e., di/dt) of 500 A/.mu.s, it becomes
possible to simultaneously implement both the high-speed response
and the stabilization of the output voltage in the steady
state.
[0059] Next, using FIG. 20, the description will be given below
concerning another embodiment of the pulse-width modulation
oscillator PWM. A circuit illustrated in FIG. 20 can be implemented
by a combination of an oscillator OSC, a one-shot multivibrator
OSM, and a V/I converter V/I. A constant time-period pulse can be
generated by the oscillator OSC as follows: A MOS 130 and a
constant current-source I5 set a constant current which is needed
for determining the desired time-period. Next, this constant
current is made to flow to the current-source MOSs 110, 120 of the
pulse-width modulation oscillator PWM in FIG. 6. Also, when this
constant time-period pulse is applied to a clock terminal CLK of
the one-shot multivibrator OSM, the terminal voltage of a capacitor
CT becomes equal to 0 on a temporary basis. At the next moment,
however, the capacitor CT is electrically charged by a current that
results from converting the error voltage of the error amplifier EA
by the V/I converter V/I. Moreover, a time that has elapsed until
this charge voltage attains to a predetermined threshold value is
acquired as the PWM pulse. In this way, the series of pulse-width
modulation oscillating operations can be repeated. Namely, it
becomes possible to acquire the PWM pulse that is proportional to
the error voltage of the error amplifier EA.
[0060] This pulse-width modulation oscillator PWM is used as an
effective unit in a multi-phase control in FIG. 11 and FIG. 12
which will be described later. In this case, in order to implement
the multi-phase operation, a phase shift circuit needs to be
inserted after the oscillator OSC.
[0061] Embodiment 4
[0062] FIGS. 8 to 10 illustrate the present embodiment. The
embodiment in FIG. 8, which is obtained by providing the transient
variation detecting circuit TVD into the embodiment in FIG. 3,
allows the acquisition of basically the same effects in FIG. 5. The
configurations in FIG. 9 and FIG. 10 are as follows: In the circuit
diagrams in FIG. 1 and FIG. 3, the input into the transient
variation detecting circuit TVD is drawn from the midpoint of a
series circuit that consists of a capacitor C3 and a resistor R3
which are set up at both ends of the inductor L of the LC smoothing
filter. As a result of this, the phase of an inductor L current,
which can be detected by the series circuit of the capacitor C3 and
the resistor R3, and the charge/discharge phase of the output
capacitor Co can be made to coincide with each other. Consequently,
it becomes possible to eliminate as much as possible
excessive/redundant electric charges produced by the
charge/discharge of the output capacitor Co from the inductor L
current. This makes it possible not only to implement the
high-speed response and the high stability, but also to reduce a
variation (i.e., ripple) in the output voltage at the transient
state.
[0063] Embodiment 5
[0064] The present embodiment is a multi-phase embodiment where the
plural power-supply device units in the 1st to the 4th embodiments
are operated in parallel. The present embodiment combines the 2 or
more same-type power-supply devices indicated in the 1st to the 4th
embodiments. Hereinafter, the explanation will be given below
selecting the 2-phasing as the example.
[0065] FIG. 11 illustrates the embodiment that results from
multi-phasing the power-supply device unit in FIG. 8. In order to
implement the multi-phasing, the embodiment in FIG. 11 newly
includes the oscillator OSC and a phase shift circuit PSFT, which
generate 2-phase pulses whose phases are shifted to each other by
180.degree.. This embodiment inputs each of the 2-phase pulses into
each of pulse-width modulation oscillators PWM1 and PWM2, thereby
implementing the multi-phase control.
[0066] FIG. 12 illustrates, in more detail, the embodiment of the
power-supply device in FIG. 11. In FIG. 12, the pulse-width
modulation oscillator PWM1 includes a voltage-to-current converting
circuit V/I1 and a one-shot multivibrator OSM1. In the steady
state, the oscillator PWM1 operates by receiving a pulse signal
from the phase shift circuit PSFT.
[0067] Using an operation state mode in FIG. 13, the explanation
will be given below regarding the operation of the embodiment in
FIG. 12. This operation state mode will be explained in much the
same way as the case of the 3rd embodiment. Hereinafter, the
explanation will be given concerning the Phase 1 power-supply
illustrated on the upper-half side in FIG. 12. (a) In the case of
Vout .gtoreq.(Vref+.DELTA.), the output duty of the pulse-width
modulation oscillator PWM1 is forcefully set at 0%. For this
purpose, the reset RST of the one-shot multivibrator OSM1 is turned
on, which makes the duty equal to 0%.
[0068] (b) In the case of (Vref+.DELTA.)>Vout>(Vref-.DELTA.),
as an ordinary operation of the one-shot multivibrator, the OSM1
receives the pulse from the phase shift circuit PSFT as a clock
CLK, thereby generating an on-pulse width. The on-pulse width is
determined by the current value from the current-source MOS 210 and
the capacitance value of a capacitor CT1, i.e., a timing capacitor.
This on-pulse width is of a control mode that operates in
accordance with the current ratio determined by the control from
the error amplifier EA. Namely, since this current ratio is equal
to the duty, the output voltage Vout becomes equal to the voltage
that is proportional to the duty .alpha. of the input voltage
Vin.
[0069] (c) In the case of Vout.ltoreq.(Vref-.DELTA.), the duty is
forcefully set at 100%. For this purpose, both ends of the
capacitor CT1, i.e., the timing capacitor, are short-circuited by a
MOS switch M21 so as to maintain the on-state, which makes the duty
equal to 100%. Incidentally, a detection result by an overcurrent
detecting circuit OC1 is also added to the reset RST, thereby
preventing a component breakdown caused by an overcurrent from the
upper-side Power MOSFET Q1. Concerning the Phase 2 power-supply on
the lower-half side in FIG. 12, the explanation will be omitted
because the operation is the same as the Phase 1 power-supply.
[0070] In the operations described so far, in the steady state, the
inductor currents from the 2 power-sources operate in opposite
phases, i.e., in phases shifted to each other by 180.degree..
Meanwhile, at the transient time, the inductor currents from the 2
power-supplies become the same in their phases, thereby dealing
with a rapid load variation. The present embodiment not only
increases the output current by using the plural power-supplies,
but also reduces a ripple in the output voltage.
[0071] In the case of providing the 2 or more power-supply device
units, there are provided an oscillator and a phase shift circuit
that the plural power-supply device units have in common. Moreover,
in the steady state, phases of driving pulses of the
upper-side/lower-side Power MOSFETs in the respective power-supply
device units are respectively shifted to phases that result from
dividing 360.degree. by the number of the power-supply device units
located in parallel. At the transient state, as are the cases with
the above-described (a) and (c), all of the parallel power-supply
device units are operated by driving pulses of one and the same
phase. In the case of, e.g., the 4 power-supply device units, it is
advisable to shift the phases to the respective phases of 0.degree.
(i.e., criterion), 90.degree., 180.degree., and 270.degree..
[0072] Embodiment 6
[0073] Next, the explanation will be given below concerning an
embodiment of the IC chip configuration of the power-supply control
device in the present invention.
[0074] FIG. 14 illustrates the embodiment of the one-chip
configuration of the circuit configuration illustrated in FIG. 8.
In FIG. 14, circuits and functions are all implemented on-chip on
one semiconductor board except for the following externally-mounted
components: The LC smoothing filter, the CR circuit consisting of
the capacitor C3 and the resistor R3 for detecting the current
phase of the transient variation detecting circuit TVD, and a boost
circuit consisting of a diode DBT and a capacitor CBT.
[0075] The on-chip implemented circuits and functions are as
follows: The CR smoothing filter consisting of the capacitor C and
the resistor R, the error amplifier EA, the reference voltage Vref,
the pulse-width modulation oscillator PWM, a band circuit DBU, a
dead band circuit DBL, a level shift circuit LS, a driver DRVU, a
driver DRVL, the upper-side/lower-side Power MOSFETs Q1, Q2, an
overcurrent detecting circuit OC, the transient variation detecting
circuit TVD, an upper and lower limit-voltage generating circuit
V.DELTA., a soft-start circuit SS, an under-voltage lockout circuit
UVLO, and a power-good circuit PWRGD. Incidentally, instead of
acquiring the reference voltage Vref from a band-gap reference
circuit, the reference voltage Vref may be acquired by receiving a
digital signal corresponding to a VID (: Voltage Identification)
code, using an on-chip D/A converter illustrated in FIG. 15.
Although there exist not-illustrated circuits and functions, the
1-chip power-supply control IC in the present embodiment is
equipped with the functions implemented in compliance with the VRM
9. 1 expounded by the Intel Corporation.
[0076] Although, in FIG. 14, the explanation has been given
selecting the case where the upper-side Power MOSFET Q1 is the
NMOS, the MOSFET Q1 may also be a PMOS. In this case, the
externally-mounted boost circuit becomes unnecessary. However,
since it is necessary to drive the gate of the PMOS at the electric
potential from the input terminal Vi, a voltage-generating supply
for this necessity is implemented on-chip.
[0077] The voltage fed to the input terminal Vi and the one fed to
a power-supply terminal Vcc may be made equal to each other, e.g.,
5V or 12V. Otherwise, the voltages may be made different, e.g., 12V
is fed to the input terminal Vi, and 5V is fed to the power-supply
terminal Vcc. When the voltage fed to the input terminal Vi and the
one fed to the power-supply terminal Vcc are different, 5V to the
power-supply terminal Vcc may be fed from the outside. Otherwise,
5V may be generated by the on-chip circuit from 12V fed to the
input terminal Vi, then being supplied thereto. Incidentally, when
feeding 12V to the input terminal Vi, an about 7V Zener diode is
connected to the boost circuit in FIG. 14 in series with the diode
DBT, thereby preventing the gate voltage of the upper-side Power
MOSFET from becoming too large.
[0078] Also, in the operation of the soft-start circuit, at the
time of injecting the power-supply, it is preferable to mask the
output signal from the transient variation detecting circuit for
the high-speed response.
[0079] Embodiment 7
[0080] FIG. 16 illustrates a multi-phase-compatible IC chip
configuration in the present embodiment. The configuration in FIG.
16 results from multi-phasing the circuit configuration of the IC
chip illustrated in FIG. 14. The point that differs from the 6th
embodiment in FIG. 14 is that the oscillator OSC and the phase
shift circuit PSFT are added to the IC chip. As IC pins that become
necessary for implementing the multi-phasing, there exist terminals
for providing its-own/the other IC chips with phase pulses .phi.1
to .phi.4 that correspond to the number of the multi phases, and
terminals for supplying the reference voltage Vref, and outputs
from the upper and lower limit-voltage generating circuit V.DELTA.
to the transient variation detecting circuit TVD.
[0081] In the case of configuring the multi phases, at first, IC
chips are prepared by the number of the desired multi phases, and,
from among the IC chips, one IC chip is selected as a master.
Concretely, a selection signal SEL0 for selecting the master IC
chip activates the oscillator OSC and a switch SWr, and 2 bits of
selection signals SEL1 and SEL2 specify the desired multi-phase
number. Next, the master IC chip supplies the phase pulses .phi.2
to .phi.4, the reference voltage Vref, and the outputs Vref+.DELTA.
and Vref-.DELTA. from the upper and lower limit-voltage generating
circuit V.DELTA.. As a result, it turns out that .phi.2 to .phi.4,
Vref, Vref+.DELTA., and Vref-.DELTA. are added to the other IC
chips, respectively. This allows the implementation of the
multi-phasing.
[0082] Although, in the present embodiment, the multi-phase number
has been illustrated as 4, no limitation is imposed on the
multi-phase number. The selection-signal number for setting the
multi-phase number is modified, and the circuit configuration of
the phase shift circuit PSFT is modified to a circuit configuration
that matches the multi-phase number, and these pieces of
information are installed into the IC chips. This allows the
multi-phase number to be increased or decreased depending on the
requirements.
[0083] Embodiment 8
[0084] FIG. 17 illustrates an embodiment where the power-supply
control IC chip in the present invention is implemented on a
printed wiring board. In FIG. 17, the power-supply control ICs, and
the inductor L and the capacitor Co are mounted on a printed wiring
board PB with the use of a BGA (: Ball Grid Array) and chip
components, respectively, thereby allowing the downsized
high-density implementation. Here, the capacitor Co is the chip
ceramic capacitor. Incidentally, although not illustrated, in
addition to these components, the CR circuit of the capacitor C3
and the resistor R3, the boost circuit, and the input capacitor are
mounted on the printed wiring board PB with the use of chip
components in this embodiment. Also, other than the on-chip
mounting by the BGA, the CSP (: Chip Size Package) mounting may
also be employed.
[0085] Furthermore, in the case of the multi-phase compatibility,
other than the on-chip mounting of the plural power-supply control
ICs, the MCM (: Multi Chip Module) mounting may also be employed.
In addition to these mountings, components divided onto 2 IC chips,
such as a control unit including the error amplifier, the
oscillator PWM, and the like, and a driver unit where the Power
MOSFETs are built-in, may also be mounted on the printed wiring
board in much the same way.
[0086] As described above, according to the present embodiment, it
becomes possible to implement the elimination of a pin neck, an
enhancement in the heat-dissipating capability, and the downsizing
of the printed wiring board of the power-supply device.
[0087] Embodiment 9
[0088] FIG. 18 illustrates the present embodiment. FIG. 18
illustrates the embodiment that results from applying the present
invention to HDDs (: Hard Disk Drives). Each of the HDDs includes a
magnetic storage disk, a magnetic head, a magnetic-disk rotating
drive, a magnetic-head drive, a magnetic-head position controller,
and an input/output signal controller. DC-DC converters DC-DC1 to
DC-DCn, i.e., the power-supply devices described in the first to
the eigth embodiments, supply electric power to these HDDs HDD1 to
HDDn. As the DC-DC converters DC-DC1 to DC-DCn, i.e., the
power-supply devices illustrated in FIG. 18, the single-phase
power-supply devices or the multi-phase power-supply devices are
used, depending on the current capacities of the HDDs, i.e. the
targets of the power supply.
[0089] Embodiment 10
[0090] Next, the explanation will be given below concerning an
embodiment where the control scheme in the present invention is
applied to isolation type DC-DC converters. FIG. 19 illustrates the
embodiment applied to a forward type converter. In FIG. 19, as is
the case with FIG. 3, the CR smoothing filter of C and R is set up
at both ends of an inductor L of the forward type converter. Next,
the error amplifier EA generates an amplified error voltage, using
the relationship between a voltage VFB at the midpoint of the CR
smoothing filter and a reference voltage Vref. Moreover, the use of
the pulse-width modulation oscillator PWM converts this amplified
error voltage into a PWM pulse. This PWM pulse is passed through a
transformer T2, and is applied to the gate of a Power MOSFET QD for
driving a transformer T1, then being subjected to the
negative-feedback control. This allows a desired output voltage to
be acquired in a stationary manner at the output terminal Vo. The
present method performs no negative-feedback control over the
output from the power LC filter, thereby making it possible to
configure a high loop-stability power-supply system. Consequently,
the present embodiment is especially effective when the ceramic
capacitor is used as C of the LC filter.
[0091] Although, so far, the explanation has been given using the
CR filter in FIG. 3, the explanation is also possible using the
method in FIG. 1. Also, instead of the transformer T2, the
implementation is also possible using a photo coupler. In FIG. 19,
the explanation has been given selecting the 1-stone forward type
converter. The above-described control scheme, however, is also
applicable to the other isolation type DC-DC converters such as
2-stone forward type, push-pull type, half-bridge type, and
full-bridge type.
[0092] Embodiment 11
[0093] Next, the illustration will be given below regarding an
embodiment where the control scheme in the present invention is
applied to a commercially-available power-supply IC. FIG. 21
illustrates the case where a PWM control IC HIP6311A and a
driver-built-in Power MOSFET IC ILS6571 of the Intersil Corporation
are used as the commercially-available power-supply IC. The
midpoint of C and R of one-side CR smoothing filter set up at both
ends of an inductor L is connected to a feedback terminal FB of the
PWM control IC HIP6311A. The midpoint of C3 and R3 of the
other-side CR smoothing filter is connected to a transient
variation detecting circuit TVD comprised of a reference
power-source LT1790A and a converter LT1715 of the Linear
Technology Corporation through a high-input-impedance buffer
amplifier BA and a resistor RN. Moreover, from the relationship
between logical levels "H" and "L" of two signals a and b acquired
by the transient variation detecting circuit TVD, 3 operation state
modes, i.e., a PWM pulse signal PWM1 (desired duty .alpha.)
outputted from the PWM control IC, a duty 0% .alpha.0, and a duty
100% .alpha.100, are switched selectively as indicated in FIG. 22
by a selector HD74HC153, HD74HC157. Furthermore, its selected
signal Y is outputted to a PWM terminal of the driver-built-in
Power MOSFET IC. This shows that the control scheme in the present
invention is also applicable easily to the power-supply device
configured using the commercially-available power-supply IC. The
application of the present invention is not limited to the products
described in the above-described embodiment. Incidentally, when the
transient variation detecting circuit TVD is not used, the PWM
pulse signal PWM1 outputted from the PWM control IC is directly
connected to the PWM terminal of the driver-built-in Power MOSFET
IC, thereby making it possible to implement the present
invention.
[0094] It is needless to say that, although not illustrated, the
power-supply devices in the first to the eighth embodiments can be
applied and expanded to the other apparatuses, e.g., a VRM, a DC-DC
converter for portable-appliances, and a general-purpose DC-DC
converter.
[0095] In the power-supply device of the present invention, none of
the secondary delay by the power LC smoothing filter enters the
control loop, which enhances the stability of the control loop.
This further makes it possible to use the small-ESR chip ceramic
capacitor in the LC smoothing filter, thereby implementing the
downsizing of the power-supply device.
[0096] In the power-supply device of the present invention, the
upper and lower limit value detecting circuit controls the
high-speed response at the transient state. This allows the
power-supply device to make response to even the high current slew
rate (i.e., di/dt).
[0097] The power-supply device of the present invention can be
easily multi-phased. This makes it possible to simultaneously
implement both the large output current and the ripple-voltage
reduction.
[0098] It should be further understood by those skilled in the art
that although the foregoing description has been made on
embodiments of the invention, the invention is not limited thereto
and various changes and modifications may be made without departing
from the spirit of the invention and the scope of the appended
claims.
* * * * *