U.S. patent application number 10/821244 was filed with the patent office on 2005-01-06 for motor control device.
This patent application is currently assigned to Hitachi, Ltd.. Invention is credited to Fujino, Shinichi, Goto, Kosei, Hashimoto, Keita, Inaba, Masamitsu, Innami, Toshiyuki, Iwamura, Masahiro, Sakano, Junichi, Sakurai, Yoshimi, Shirakawa, Shinji, Tsuchiya, Masanori.
Application Number | 20050001582 10/821244 |
Document ID | / |
Family ID | 32871255 |
Filed Date | 2005-01-06 |
United States Patent
Application |
20050001582 |
Kind Code |
A1 |
Goto, Kosei ; et
al. |
January 6, 2005 |
Motor control device
Abstract
The invention is intended to provide a control device for a
vehicular AC motor, which has higher efficiency of voltage
utilization in a power running mode and has higher efficiency of
electricity generation in an electricity generation mode. The motor
control device comprises rectifying devices and switching devices
for three phases, which are connected between a DC power source and
armature coils of an AC motor operatively coupled to an internal
combustion engine. The motor control device has the inverter
function of converting a DC power from the DC power source into an
AC power and supplying the AC power to the armature coils, and the
converter function of converting an AC power generated by the AC
motor into a DC power and supplying the DC power to the DC power
source. Rectangular-wave driving control of applying rectangular
wave voltages to the armature coils of the AC motor is performed
when the AC motor is operated for power running, and synchronous
rectification control for making synchronous rectification of the
AC power generated by the AC motor is performed when the AC motor
is operated for electricity generation.
Inventors: |
Goto, Kosei; (Takahagi,
JP) ; Innami, Toshiyuki; (Mito, JP) ; Fujino,
Shinichi; (Mito, JP) ; Sakurai, Yoshimi;
(Hitachiohta, JP) ; Inaba, Masamitsu; (Hitachi,
JP) ; Iwamura, Masahiro; (Hitachi, JP) ;
Shirakawa, Shinji; (Hitachi, JP) ; Sakano,
Junichi; (Hitachi, JP) ; Hashimoto, Keita;
(Hitachinaka, JP) ; Tsuchiya, Masanori;
(Hitachinaka, JP) |
Correspondence
Address: |
CROWELL & MORING LLP
INTELLECTUAL PROPERTY GROUP
P.O. BOX 14300
WASHINGTON
DC
20044-4300
US
|
Assignee: |
Hitachi, Ltd.
|
Family ID: |
32871255 |
Appl. No.: |
10/821244 |
Filed: |
April 9, 2004 |
Current U.S.
Class: |
318/802 |
Current CPC
Class: |
F02N 11/0859 20130101;
H02M 7/797 20130101; F02N 11/04 20130101; H02P 9/08 20130101; F02N
2011/0896 20130101 |
Class at
Publication: |
318/802 |
International
Class: |
H02P 005/28 |
Foreign Application Data
Date |
Code |
Application Number |
Apr 10, 2003 |
JP |
2003-106907 |
Jan 26, 2004 |
JP |
2004-17043 |
Claims
What is claimed is:
1. A motor control device comprising rectifying devices and
switching devices for three phases, which are connected between a
DC power source and armature coils of an AC motor operatively
coupled to an internal combustion engine, said motor control device
having the inverter function of converting a DC power from said DC
power source into an AC power and supplying the AC power to said
armature coils, and the converter function of converting an AC
power generated by said AC motor into a DC power and supplying the
DC power to said DC power source, wherein rectangular-wave driving
control of applying rectangular wave voltages to said armature
coils of said AC motor is performed when said AC motor is operated
for power running, and synchronous rectification control for making
synchronous rectification of the AC power generated by said AC
motor is performed when said AC motor is operated for electricity
generation.
2. A motor control device according to claim 1, wherein when the
rectangular wave voltages are applied to said armature coils in the
rectangular-wave driving control, currents flowing through said
switching devices are held below maximum allowable current values
of said switching devices.
3. A motor control device according to claim 2, wherein pulse
widths of the rectangular wave voltages applied to said armature
coils are set such that the currents flowing through said switching
devices are held below the maximum allowable current values of said
switching devices.
4. A motor control device according to claim 2, wherein resistance
values of said armature coils are set such that the currents
flowing through said switching devices are held below the maximum
allowable current values of said switching devices.
5. A motor control device according to claim 1, wherein, in the
rectangular-wave driving control, pulse widths of the rectangular
wave voltages applied to said armature coils are set such that each
pulse width is equal to a half cycle (180.degree.) of an electrical
angle of said AC motor at maximum and gradually decreases as a
rotation speed of said AC motor lowers.
6. A motor control device according to claim 1, wherein, in the
rectangular-wave driving control, pulse widths of the rectangular
wave voltages applied to said armature coils are set such that each
pulse width is equal to a half cycle (180.degree.) of an electrical
angle of said AC motor at maximum and gradually decreases as a
voltage of said AC motor lowers.
7. A motor control device according to claim 6, wherein, in the
rectangular-wave driving control, the pulse widths of the
rectangular wave voltages applied to said armature coils are set
such that, when a voltage of said DC power source is relatively
high, currents flowing through said switching devices are held
below the maximum allowable current values of said switching
devices.
8. A motor control device according to claim 1, wherein, in the
rectangular-wave driving control, pulse widths of the rectangular
wave voltages applied to said armature coils are set such that each
pulse width is equal to a half cycle (180.degree.) of an electrical
angle of said AC motor at maximum and gradually decrease as
temperatures of said armature coils or temperatures of said
switching devices rise.
9. A motor control device according to claim 8, wherein, in the
rectangular-wave driving control, the pulse widths of the
rectangular wave voltages applied to said armature coils are set
such that, when temperatures of said armature coils or temperatures
of said switching devices are relatively high, currents flowing
through said switching devices are held below the allowable
temperatures of said armature coils or said switching devices.
10. A motor control device according to claim 1, wherein the power
running under the rectangular-wave driving control is performed
when the rotation speed of said AC motor lowers is lower than a
predetermined speed, and the electricity generation under the
synchronous rectification control is performed when the rotation
speed of said AC motor lowers is higher than the predetermined
speed.
11. A motor control device according to claim 1, wherein the power
running under the rectangular-wave driving control is performed
when a predetermined time has not yet lapsed from start of
operation of said internal combustion engine, and electricity
generation under the synchronous rectification control is performed
when the predetermined time has lapsed from start of operation of
said internal combustion engine.
12. A motor control device for performing power running control and
electricity generation control on an AC motor connected to an
internal combustion engine, said motor control device comprising: a
power module including rectifying devices and switching devices,
and having the inverter function of converting a direct current
into an alternating current and the converter function of
converting an alternating current into a direct current; a
power-running/electricity-generation changing-over unit for
changing over the power running control and the electricity
generation control to be performed on said AC motor; and a power
running control unit for performing the power running control on
said AC motor when the power running control is selected by said
power-running/electricity-generation changing-over unit, said power
running control unit comprising: a magnetic pole position detecting
unit for detecting a magnetic pole position or an electrical angle
of said AC motor; a speed computing unit for computing a rotation
speed .omega. of said AC motor; an interlinkage magnetic-flux
amount computing unit for computing an amount of magnetic flux
.phi. interlinking with armature coils of a stator of said AC
motor; a voltage vector phase computing unit for computing a phase
.theta.v of a voltage vector V applied to the armature coil of the
stator of said AC motor; and a pulse generating unit for generating
a switching signal applied to the switching device of said power
module, thereby performing rectangular-wave driving control on said
AC motor in a power running mode.
13. A motor control device for performing power running control and
electricity generation control on an AC motor connected to an
internal combustion engine, said motor control device comprising: a
power module including rectifying devices and switching devices,
and having the inverter function of converting a direct current
into an alternating current and the converter function of
converting an alternating current into a direct current; a
power-running/electricity-generation changing-over unit for
changing over the power running control and the electricity
generation control to be performed on said AC motor; and an
electricity generation control unit for performing the electricity
generation control on said AC motor when the electricity generation
control is selected by said power-running/electricity-generation
changing-over unit, said electricity generation control unit
comprising: a magnetic pole position detecting unit for detecting a
magnetic pole position or an electrical angle of said AC motor; a
speed computing unit for computing a rotation speed .omega. of said
AC motor; an interlinkage magnetic-flux amount computing unit for
computing an amount of magnetic flux .phi. interlinking with
armature coils of a stator of said AC motor; an induced voltage
computing unit for computing induced voltages Vue, Vve and Vwe of
respective phases in the armature coils of said rotor of said AC
motor; a DC voltage detecting unit for detecting a voltage VB of a
DC power source connected to said power module; a voltage comparing
unit for comparing inter-line values of the induced voltages Vue,
Vve and Vwe with the voltage VB of said DC power source; a pulse
cycle computing unit for computing a pulse cycle of a switching
signal applied to the switching device of said power module based
on a result of the comparison made by said voltage comparing unit;
and a pulse generating unit for generating the switching signal
applied to the switching device of said power module based on the
pulse cycle computed by said pulse cycle computing unit, thereby
performing synchronous rectification control on said AC motor in an
electricity generation mode.
14. A motor control device according to claim 13, wherein the
induced voltages generated in said AC motor when said internal
combustion engine is running at idle is higher than the voltage of
said DC power source.
15. A motor control device comprising: upper arm driving means for
receiving a power-running or rectification mode command, selecting
an upper-arm power running drive signal or an upper-arm
rectification drive signal in response to the power-running or
rectification mode command, and outputting the upper-arm power
running drive signal or the upper-arm rectification drive signal to
a control terminal of an upper-arm switching device; lower arm
driving means for receiving a power-running or rectification mode
command, selecting a lower-arm power running drive signal or a
lower-arm rectification drive signal in response to the
power-running or rectification mode command, and outputting the
lower-arm power running drive signal or the lower-arm rectification
drive signal to a control terminal of a lower-arm switching device;
phase-correction drive signal distributing means for receiving a
magnetic pole position detected signal from an AC motor, advancing
a phase of the magnetic pole position detected signal depending on
a rotation speed of said AC motor, and distributing the magnetic
pole position detected signal having the advanced phase as the
upper-arm power running drive signal and the lower-arm power
running drive signal, the upper-arm power running drive signal
being outputted to said upper arm driving means and the lower-arm
power running drive signal being outputted to said lower arm
driving means; upper-arm rectification detecting means for
comparing the magnitude of a potential at a higher potential
terminal for a main power source with the magnitude of a potential
at an output terminal, and outputting the upper-arm rectification
drive signal to said upper arm driving means when the magnitude of
the potential at the output terminal is larger; and lower-arm
rectification detecting means for comparing the magnitude of the
potential at the output terminal with the magnitude of a potential
at a lower potential terminal for said main power source, and
outputting the lower-arm rectification drive signal to said lower
arm driving means when the magnitude of the potential at the output
terminal is smaller.
16. A motor control device according to claim 15, wherein said
phase-correction drive signal distributing means comprises: a
frequency-voltage conversion circuit for receiving the magnetic
pole position detected signal from said AC motor and converting
rotation frequency of said AC motor into a DC voltage; a constant
current source having a current value, changed depending on an
output voltage of said frequency-voltage conversion circuit;
triangular wave generating means for generating a triangular wave
with a constant current supplied from said constant current source;
a capacity charging switch for delivering the current from said
constant current source to said triangular wave generating means; a
capacity discharging switch for drawing the current from said
triangular wave generating means into said constant current source;
a voltage comparator for comparing the triangular wave outputted
from said triangular wave generating means with a reference
voltage, and generating a voltage pulse; a drive signal
distribution circuit for distributing the voltage pulse outputted
from said voltage comparator as the upper-arm power running drive
signal with the potential at the lower potential terminal for said
main power source being a reference, and as lower-arm power running
drive signal with the potential at the lower potential terminal for
said main power source being a reference, the lower-arm power
running drive signal being outputted to said lower arm driving
means; and a level shift-up circuit for converting the reference
potential of the upper-arm power running drive signal from the
potential at the lower potential terminal for said main power
source into a potential at an output terminal, and outputting the
converted potential to said upper arm driving means.
17. A motor control device according to claim 15, wherein said
upper-arm rectification detecting means comprises: negative voltage
detecting means for receiving the potential at the higher potential
terminal for said main power source and the potential at the output
terminal, outputting a negative voltage to the output terminal when
the magnitude of the potential at the output terminal is larger,
and outputting a positive voltage to the output terminal when the
magnitude of the potential at the output terminal is smaller; and
amplifying means for amplifying a voltage level when said negative
voltage detecting means outputs a negative voltage, and wherein
said lower-arm rectification detecting means comprises: negative
voltage detecting means for receiving the potential at the output
terminal and the potential at the lower potential terminal for said
main power source, outputting a negative voltage to the lower
potential terminal for said main power source when the magnitude of
the potential at the output terminal is smaller, and outputting a
positive voltage to the output terminal when the magnitude of the
potential at the output terminal is smaller; and amplifying means
for amplifying a voltage level when said negative voltage detecting
means outputs a negative voltage.
18. A motor control device according to claim 15, further
comprising: an upper arm switching device having a drain connected
to the higher potential terminal for said main power source and
having a source connected to the output terminal; and a lower arm
switching device having a drain connected to the output terminal
and having a source connected to the lower potential terminal for
said main power source.
Description
BACKGROUND OF THE INVENTION
[0001] 1. Field of the Invention
[0002] The present invention relates to a control device for a
vehicular AC motor, and more particularly to a motor control device
for controlling outputting of power (i.e., power running) and
generation of electric power (i.e., electricity generation).
[0003] 2. Description of the Related Art
[0004] As a control method for a vehicular AC motor, there is known
PWM. (Phase Width Modulation) control for switching a device with a
high-frequency PWM signal and changing the phase of a current
flowing through an armature coil of the AC motor relative to an
induced voltage of the AC motor, thereby performing power running
or electricity generation.
[0005] JP,A 2000-197204 discloses rectangular-wave driving control
for applying, to an armature coil of an AC motor, a rectangular
wave voltage each phase of which is changed over per half cycle
(180.degree.), and changing the phase of a current flowing through
the armature coil relative to an induced voltage of the AC motor,
thereby controlling torque of the AC motor. Further, in relation to
electricity generation control, JP,A 2002-218797 discloses
synchronous rectifying control for switching a device, which
rectifies a current generated by an AC motor, in a region where an
induced voltage of the AC motor is higher than a source voltage,
thereby reducing a loss during the rectification.
[0006] In addition, as a known power running control method for an
AC motor, JP,A 2001-45789, for example, discloses a method of
changing over a rectangular-wave driving signal depending on a
rotation speed. The turning-on width of the rectangular-wave
driving signal is changed over in accordance with information,
stored in advance, regarding the rotation speed and the motor
efficiency.
SUMMARY OF THE INVENTION
[0007] In the case of driving an AC motor based on the PWM control,
high switching frequency increases the number of times of switching
to be made for a certain time and hence increases a switching loss
as a whole. Also, because of a high switching speed, a ripple of DC
voltage is increased and a smoothing capacitor having a large
capacity is required to suppress the ripple, thus resulting in a
larger size of a motor control device. Since an induced voltage of
the AC motor must be held lower than the voltage capable of being
controlled by the motor control device, the field intensity is
weakened in the range of high rotation speeds. This requires an
additional current and therefore reduces the efficiency. Further,
noise caused with the switching operation may invite a problem in
some cases.
[0008] In the case of driving an AC motor based on the
rectangular-wave driving control, the switching frequency is lower
than that in the PWM control and therefore a smoothing capacitor is
not required. However, the current generated by the AC motor in the
electricity generation mode flows only through a diode connected in
anti-parallel relation to a switching device having a large loss.
Accordingly, a loss during the electricity generation is increased
as compared with the synchronous rectifying control in which a
switching device and a diode are used in a combined manner for the
rectification.
[0009] In order to obtain maximum torque in the power running mode,
the phase of an applied voltage command for a switching device per
phase must be controlled so as to lead relative to the phase of an
induced voltage per phase of an armature coil. The angle of such a
lead changes depending on the rotation speed and the interlinkage
magnetic flux generated in an excitation coil. With the above in
mind, in the power running control method disclosed in the
above-cited JP,A 2001-45789, the turning-on width of the
rectangular-wave driving signal is changed over depending on the
rotation speed in accordance with the information, stored in
advance, regarding the rotation speed and the motor efficiency.
That changing-over control requires a microcomputer and therefore
pushes up the cost.
[0010] Recently, it has been proposed to restart an engine without
standing at idle after the engine operation is automatically
stopped upon stop of a vehicle, by using alternator (generator) as
a motor. In that case, because the engine is in the warmed-up
state, it can be restarted with no need of more complicated control
than that required in start from the cold state. Thus, complicated
control using a microcomputer, for example, is not required. The
use of a microcomputer also pushes up the cost.
[0011] Another problem experienced in the past is that a current
generated by an alternator is rectified with diodes and hence
efficiency is poor.
[0012] A first object of the present invention is to provide a
control device for a vehicular AC motor, which has higher
efficiency of voltage utilization in a power running mode and has
higher efficiency of electricity generation in an electricity
generation mode.
[0013] A second object of the present invention is to provide a
motor control device, which can perform the power running control
without using a microcomputer and has higher rectification
efficiency.
[0014] To achieve the first object, according to the present
invention, rectangular-wave driving control is performed when the
AC motor is operated for power running, and synchronous rectifying
control is performed when the AC motor is operated for electricity
generation.
[0015] Since the rectangular-wave driving control is performed in
the power running mode, the voltage applied to an armature coil of
the AC motor is provided as a rectangular wave (1 pulse). In other
words, a maximum voltage of the DC power source is applied to the
armature coil of the AC motor and the utilization factor of voltage
can be increased. It is therefore possible to increase the
efficiency of control for weakening the field intensity, which is
performed in the range of high rotation speeds. Further, since the
switching frequency is low, a switching loss can be reduced. In
addition, since the switching speed can be held slow, a smoothing
capacitor having a large capacity is no longer required and the
size of the motor control device can be reduced.
[0016] In the electricity generation mode, the synchronous
rectification control is performed and the generated voltage is
rectified by combined use of a switching device and a diode
connected to the switching device in anti-parallel connection.
Accordingly, a loss during the rectification can be reduced and the
efficiency of the electricity generation can be increased. Further,
since the switching speed is relatively slow as in the
rectangular-wave driving control, a smoothing capacitor is no
longer required and the size of the motor control device can be
reduced.
[0017] In any of the power running mode and the electricity
generation mode, noises caused with the switching operation can be
eliminated.
[0018] Also, to achieve the second object, a motor control device
according to the present invention comprises an upper arm driving
unit for receiving a power-running or rectification mode command,
selecting an upper-arm power running drive signal or an upper-arm
rectification drive signal in response to the power-running or
rectification mode command, and outputting the upper-arm power
running drive signal or the upper-arm rectification drive signal to
a control terminal of an upper-arm switching device; a lower arm
driving unit for receiving a power-running or rectification mode
command, selecting a lower-arm power running drive signal or a
lower-arm rectification drive signal in response to the
power-running or rectification mode command, and outputting the
lower-arm power running drive signal or the lower-arm rectification
drive signal to a control terminal of a lower-arm switching device;
a phase-correction drive signal distributing unit for receiving a
magnetic pole position detected signal from an AC motor, advancing
a phase of the magnetic pole position detected signal depending on
a rotation speed of the AC motor, and distributing the magnetic
pole position detected signal having the advanced phase as the
upper-arm power running drive signal and the lower-arm power
running drive signal, the upper-arm power running drive signal
being outputted to the upper arm driving means and the lower-arm
power running drive signal being outputted to the lower arm driving
means; an upper-arm rectification detecting unit for comparing the
magnitude of a potential at a higher potential terminal for a main
power source with the magnitude of a potential at an output
terminal, and outputting the upper-arm rectification drive signal
to the upper arm driving unit when the magnitude of the potential
at the output terminal is larger; and a lower-arm rectification
detecting unit for comparing the magnitude of the potential at the
output terminal with the magnitude of a potential at a lower
potential terminal for the main power source, and outputting the
lower-arm rectification drive signal to the lower arm driving unit
when the magnitude of the potential at the output terminal is
smaller.
[0019] With the arrangement set forth above, the power running
control can be performed without using a microcomputer, and the
efficiency of rectification can be increased.
BRIEF DESCRIPTION OF THE DRAWINGS
[0020] FIG. 1 is a block diagram of a driving system of a vehicle
mounted with a motor control device for a vehicle according to one
embodiment of the present invention;
[0021] FIG. 2 is an electrical circuit diagram including the motor
control device for the vehicle according to one embodiment of the
present invention;
[0022] FIG. 3 is a chart showing a voltage vector, decomposed into
d- and q-axis components, applied to an armature coil in
rectangular-wave driving control in the power running mode
according to one embodiment of the present invention;
[0023] FIG. 4 is a time chart showing an induced voltage per phase,
an applied voltage command for an armature coil, a magnetic pole
position signal, and a pulse applied to each switching device,
which are used in the rectangular-wave driving control in the power
running mode according to one embodiment of the present
invention;
[0024] FIG. 5 is a block diagram of a power running control unit
for executing the rectangular-wave driving control according to one
embodiment of the present invention;
[0025] FIG. 6 is a flowchart of processing executed in the
rectangular-wave driving control in the power running mode
according to one embodiment of the present invention;
[0026] FIG. 7 is a time chart showing the induced voltage per
phase, the applied voltage command for the armature coil, the
magnetic pole position signal, and the pulse applied to each
switching device, which are used in the rectangular-wave driving
control in the power running mode according to one embodiment of
the present invention when the pulse width of the applied voltage
command for the armature coil is narrowed;
[0027] FIG. 8 is a graph showing the relationship between the
rotation speed of an AC motor and the pulse width of the voltage
applied to the armature coil resulting in the rectangular-wave
driving control in the power running mode according to one
embodiment of the present invention;
[0028] FIG. 9 is a graph showing the relationship between the
battery voltage and the pulse width of the voltage applied to the
armature coil resulting in the rectangular-wave driving control in
the power running mode according to one embodiment of the present
invention;
[0029] FIG. 10 is a graph showing the relationship between the
temperature of the switching device and the pulse width of the
applied voltage resulting in the rectangular-wave driving control
in the power running mode according to one embodiment of the
present invention;
[0030] FIG. 11 is a graph showing the relationship between the
temperature of the armature coil and the pulse width of the applied
voltage resulting in the rectangular-wave driving control in the
power running mode according to one embodiment of the present
invention;
[0031] FIG. 12 is a block diagram of an electricity generation
control unit for executing synchronous rectification control
according to one embodiment of the present invention;
[0032] FIG. 13 is a time chart showing the induced voltage per
phase, the magnetic pole position signal, and the pulse applied to
each switching device, which are used in the synchronous
rectification control in the electricity generation mode according
to one embodiment of the present invention when the battery voltage
is lower than an induced voltage inter-line value Ve;
[0033] FIG. 14 is a time chart showing the induced voltage per
phase, the magnetic pole position signal, and the pulse applied to
each switching device, which are used in the synchronous
rectification control in the electricity generation mode according
to one embodiment of the present invention when the battery voltage
is higher than the induced voltage inter-line value Ve;
[0034] FIG. 15 is a flowchart of processing executed in the
synchronous rectification control in the electricity generation
mode according to one embodiment of the present invention;
[0035] FIG. 16 is a graph showing one example of changeover timing
between the power running control and the electricity generation
control according to one embodiment of the present invention;
[0036] FIG. 17 is a graph showing another example of changeover
timing between the power running control and the electricity
generation control according to one embodiment of the present
invention;
[0037] FIG. 18 is a block diagram of a motor control system using a
motor control device according to another embodiment of the present
invention;
[0038] FIG. 19 is a block diagram of a lead-angle drive signal
distribution circuit used in the motor control device according to
another embodiment of the present invention;
[0039] FIG. 20 is a graph for explaining the relationship between
the rotation frequency and the lead angle, which is provided by the
lead-angle drive signal distribution circuit used in the motor
control device according to another embodiment of the present
invention;
[0040] FIG. 21 is a graph for explaining the relationship between
the rotation frequency and a current value of a constant current
source, which is provided by the lead-angle drive signal
distribution circuit used in the motor control device according to
another embodiment of the present invention;
[0041] FIG. 22 is a circuit diagram of the constant current source
used in the lead-angle drive signal distribution circuit of the
motor control device according to another embodiment of the present
invention;
[0042] FIG. 23 is a circuit diagram of a capacity charging switch
and a capacity discharging switch both used in the lead-angle drive
signal distribution circuit of the motor control device according
to another embodiment of the present invention;
[0043] FIG. 24 is a waveform chart showing the operation of the
lead-angle drive signal distribution circuit of the motor control
device according to another embodiment of the present
invention;
[0044] FIG. 25 is a waveform chart showing the operation of the
lead-angle drive signal distribution circuit of the motor control
device according to another embodiment of the present
invention;
[0045] FIG. 26 is a waveform chart of MOSFET gate drive signals of
the U, V and W phases in the power running control when the motor
control device according to another embodiment of the present
invention is employed;
[0046] FIG. 27 is a block diagram of a rectification
detecting/driving circuit used in the motor control device
according to another embodiment of the present invention;
[0047] FIG. 28 is a waveform chart of the rectification
detecting/driving circuit used in the motor control device
according to another embodiment of the present invention; and
[0048] FIG. 29 is a waveform chart of MOSFET gate drive signals of
the U, V and W phases in the synchronous rectification control
executed by the motor control device according to another
embodiment of the present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0049] Embodiments of the present invention will be described below
with reference to the accompanying drawings. FIG. 1 shows the
construction of a driving system of a vehicle mounted with a motor
control device for a vehicle according to one embodiment of the
present invention. As shown in FIG. 1, the driving system of the
vehicle comprises an internal combustion engine 1, an AC motor
(motor/generator) 9, a motor control device 3, and a DC power
source (battery) 5. A crankshaft of the internal combustion engine
1 and an output shaft of the AC motor 9 are coupled to each other
through a power transmitting means 2 such as a belt. Also, the AC
motor 9 and the motor control device 3 are connected to each other
by 3-phase power cables (output lines) 4 and an excitation cable 7.
The motor control device 3 is connected to the DC power source
(battery) 5 via DC power cables (power source lines) 6.
[0050] The motor control device 3 operates as an inverter circuit
for converting a DC power from the DC power source 5 into an AC
power when it performs power running control while employing the AC
motor 9 as a motor. The motor control device 3 also operates as a
converter (rectifier) circuit for converting an AC power from the
AC motor 9 into a DC power when it performs electricity generation
control while employing the AC motor 9 as a generator.
[0051] When the internal combustion engine 1 is started up, the
motor control device 3 performs the power running control for the
AC motor 9. In other words, the AC power is supplied to the AC
motor 9 from the DC power source 5 through the motor control device
3. The output shaft of the AC motor 9 produces a powering torque
for rotating the crankshaft of the internal combustion engine 1
through the power transmitting means 2. When the internal
combustion engine 1 reaches a predetermined revolution speed,
firing is started. In this case, the AC motor 9 serves as a starter
motor.
[0052] When the internal combustion engine 1 is in a stable
self-sustained operating state, the motor control device 3 stops
the power running control for the AC motor 9 and starts the
electricity generation control. In other words, the AC motor 9 is
driven by the power of the internal combustion engine 1 to generate
electricity. An AC power obtained with the electricity generation
is converted into a DC power by the motor control device 3 and then
charged in the DC power source 5.
[0053] Thus, the AC motor 9 functions not only as a motor supplied
with electricity from the DC power source 5 and generating power,
but also as a generator supplied with power from the internal
combustion engine 1 and generating electricity.
[0054] The electrical configuration of the driving system shown in
FIG. 1 will be described below with reference to FIG. 2. The motor
control device 3 of this embodiment comprises a power module 10, a
power module driving circuit 11, a controller 12, and an excitation
driving circuit 15.
[0055] The power module 10 is constituted as a 3-phase bridge
circuit including switching devices (UP to WN) and rectifying
devices connected respectively to the switching devices in
anti-parallel relation. In this embodiment, a field effect
transistor (FET) is used as the switching device, and a diode is
used as the rectifying device.
[0056] The AC motor 9 comprises a stator and a rotor, and it is
constituted as a 3-phase AC motor of winding field type. An
excitation coil 14 is wound over the rotor coupled to the power
transmitting means 2 for driving, and armature coils 16 for the U,
V and W phases are wound over the stator.
[0057] The AC motor 9 is provided with a magnetic pole position
detecting means 13 for detecting the rotational position of the
rotor. The rotor's excitation coil 14 of the AC motor 9 is supplied
with electricity from the excitation driving circuit 15. Also, the
voltage applied to the excitation coil 14 is adjusted by the
excitation driving circuit 15. The output lines 4 for the
respective phases extended from the stator's armature coils 16 of
the AC motor 9 are connected to the power source lines 6, which are
connected to a high-potential side terminal and a low-potential
side terminal of the DC power source 5, through the 3-phase
switching devices (UP to WN) and associated rectification devices
of the power module 10.
[0058] When the power running control is performed while employing
the AC motor 9 as a motor, the power module 10 operates as an
inverter circuit for converting a DC power accumulated in the DC
power source 5 into an AC power and supplying the AC power to the
armature coils 16. Also, when the electricity generation control is
performed while employing the AC motor 9 as a generator, the power
module 10 operates as a converter (rectifier) circuit for
converting an AC power outputted from the armature coils 16 with
the electricity generation into a DC power and supplying the DC
power to the power source lines 6. Turning-on/off of the switching
devices (UP to WN) to realize the above-described operation of the
power module 10 is controlled by the power module driving circuit
11.
[0059] On the other hand, the controller 12 includes, as described
later, a changeover means for changing over the power running mode
and the electricity generation mode so that one of the power
running control and the electricity generation control is
selectively performed.
[0060] Rectangular-wave driving control executed as the power
running control will be described below with reference to FIG. 3.
FIG. 3 shows the relationship, represented in the form conversed
into two axes (d- and q-axis), between a current and a voltage
supplied in the power running control to the stator's armature coil
16 of the AC motor 9. Upon supply of electricity from the
excitation driving circuit 15, interlinkage magnetic flux .phi. is
generated in the rotor's excitation coil 14 in the positive
direction of the d-axis. The power running operation is performed
by causing a current to flow in the positive direction of the
q-axis that is perpendicular to the vector of the interlinkage
magnetic flux .phi..
[0061] More specifically, assuming that Iq is a q-axis component of
the current flowing through the armature coil 16 and Id is a d-axis
component of the current flowing through the armature coil 16, a
phase .theta.v of a voltage vector V applied to the armature coil
16 is controlled so that the relationship expressed by the
following formula (1) is satisfied:
Iq>0, Id=0 (1)
[0062] The phase .theta.v of the voltage vector V applied to the
armature coil 16 is derived from the following formula (2):
.theta.v=.theta.+tan.sup.-1(Vq/Vd) (2)
[0063] where .theta. is the magnetic pole position or the
electrical angle detected by the magnetic pole position detecting
means 13, Vq is a q-axis component of the voltage vector V applied
to the armature coil 16, and Vd is a d-axis component of the
applied voltage vector V. Further, the following formulae are held
for the voltage vector V applied to the armature coil 16:
V=Vq+Vd (3)
Vq=Iq.multidot.R+.omega..multidot..phi.-.omega..multidot.Ld.multidot.Id
(4)
Vd=Id.multidot.R-.omega..multidot.Lq.multidot.Iq (5)
[0064] where R is the resistance of the armature coil 16, .omega.
is the rotation speed of the AC motor 9, Lq is a q-axis inductance
component of the armature coil 16, and Ld is a d-axis inductance
component of the armature coil 16.
[0065] As will be seen from the above relationship, the phase
.theta.v of the voltage vector V applied to the armature coil 16
changes successively depending on changes of both the rotation
speed .omega. and the interlinkage magnetic flux .phi.. In this
embodiment of the present invention, therefore, respective values
of the d-axis component Vd of the voltage vector V and the q-axis
component Vq of the applied voltage vector V applied to the
armature coil 16 are computed from a map prepared with the rotation
speed .omega. and the interlinkage magnetic flux .phi. being
parameters.
[0066] A further description is made with reference to FIG. 4. (A)
in FIG. 4 represents voltages induced in the armature coils 16 per
phase, and (B) represents voltage commands Vu, Vv and Vw applied to
the armature coils 16 per phase. Also, (C) represents the magnetic
pole position, i.e., the electrical angle .theta., and (D)
represents switching command signals applied to the switching
devices (UP to WN). In this embodiment, the electrical angle
.theta. shown in FIG. 4(C) represents the angle of the induced
voltage of the U phase.
[0067] In this embodiment, as shown in FIG. 4(B), the applied
voltage commands Vu, Vv and Vw are each supplied at the electrical
angle covering the range of 180.degree..
[0068] As will be seen from comparison between the U-phase applied
voltage command Vu shown in FIG. 4(B) and the magnetic pole
position .theta. shown in FIG. 4(C), the phase of the U-phase
applied voltage command Vu is advanced relative to the magnetic
pole position .theta. by the phase .theta.v of the applied voltage
vector. In other words, there holds the relationship expressed by
the above formula (2). Assuming the present magnetic pole position
to be 0.degree., for example, the phase of the U-phase applied
voltage command Vu is equal to the phase .theta.v of the applied
voltage vector.
[0069] Further, as will be seen from comparison between FIG. 4(B)
and FIG. 4(D), once the voltage commands Vu, Vv and Vw to be
applied to the armature coils 16 per phase are determined,
switching command signals having the same phases as the applied
voltage command phases are produced.
[0070] With reference to FIG. 5, a description is now made of a
portion of the construction of the controller 12 which is related
to the power running control. The controller 12 comprises a
power-running/electricity-- generation changing-over section 19 for
determining which one of the power running control and the
electricity generation control is to be performed, and changing
over the power running control and the electricity generation
control from one to the other, and a power running control section
20D for executing the rectangular-wave driving control. The power
running control section 20D comprises a magnetic pole position
detecting unit 21 for detecting the magnetic pole position or the
electrical angle of the AC motor 9, a speed computing unit 22 for
computing the rotation speed .omega. of the AC motor 9, an
interlinkage magnetic-flux amount computing unit 23 for computing
the amount (number) of magnetic flux .phi. interlinking with the
armature coils 16 of the stator, a voltage vector phase computing
unit 24 for computing the phase .theta.v of the voltage vector V
applied to the armature coil 16, and a pulse generating unit 25 for
producing the switching signals applied to the switching devices
(UP to WN) of the power module 10. The operations of those
components will be described below with reference to FIG. 6.
[0071] The operation of the rectangular-wave driving control in the
power running control according to this embodiment of the present
invention will be described with reference to FIG. 6. First, in
step S1, it is determined, in accordance with a changeover command
signal 18 from an external controller, which one of the power
running control and the electricity generation control is to be
performed. If the power running control is to be performed, the
control flow proceeds to step S2. Then, in step S2, the magnetic
pole position .theta. is computed based on a pulse signal outputted
from the magnetic pole position detecting means 13. In step S3, the
actual rotation speed .omega. of the AC motor 9 is computed based
on a time-dependent change of the pulse signal outputted from the
magnetic pole position detecting means 13. In step S4, the amount
of magnetic flux .phi. interlinking with the armature coils 16 of
the stator is computed from both the rotation speed .omega.
computed in step S3 and the excitation current If in the excitation
coil 14 of the rotor detected by the excitation driving circuit 15.
In this embodiment, a map for the amount of interlinkage magnetic
flux .phi. is prepared in advance from calculations based on
experimental results, for example, with the excitation current If
and the rotation speed .omega. being parameters, and the
corresponding amount of interlinkage magnetic flux .phi. is
determined from the map.
[0072] In step S5, the d-axis component Vd of the voltage vector V
applied to the armature coil 16 and the q-axis component Vq of the
applied voltage vector V are computed from the amount of
interlinkage magnetic flux .phi., the rotation speed .omega. and
the magnetic pole position .theta., which were computed
respectively in steps 4, 3 and 2, by using the above formulae (4)
and (5). Then, the phase .theta.v of the voltage vector V is
computed from Vd, Vq and .theta. by using the above formula
(2).
[0073] In step S6, phases or timings of the applied voltages Vu, Vv
and Vw are obtained based on the voltage vector phase .theta.v that
was computed in step S5. Switching command signals are then
produced based on the phases or timings of the applied voltages Vu,
Vv and Vw.
[0074] In this embodiment, since the rectangular-wave driving
control is performed in the power running mode, the voltage applied
to the armature coil of the AC motor is provided as a rectangular
wave (1 pulse), whereby a maximum voltage of the DC circuit is
given and the utilization factor of voltage can be increased. It is
therefore possible to increase the efficiency of control for
weakening the field intensity, which is performed in the range of
high rotation speeds. Further, since the switching frequency is
low, a switching loss can be reduced. In addition, since the
switching speed can be held slow, a smoothing capacitor having a
large capacity is no longer required and the size of the motor
control device can be reduced.
[0075] In the rectangular-wave driving control according to this
embodiment of the present invention, unlike the PWM control in
which the current flowing through the armature coil 16 is subjected
to feedback control, the voltage of the DC power source 5 is
directly applied to the armature coil 16 at the duty of a half
cycle (180.degree.) of the electrical angle .theta. as shown at (B)
and (D) in FIG. 4. Accordingly, when the vehicle is running at low
speeds or is stopped and the impedance of the armature coil 16 is
reduced, there is a risk that a large current flows through the
power module 10 and the AC motor 9 to such an extent that the
switching devices (UP to WN) in the power module 10 may break or
the AC motor 9 may come into an overheating state. For that reason,
the impedance of the armature coil 16 must be set so as to prevent
a current from flowing through the power module 10 and the AC motor
9 at a value beyond the allowable current value when the voltage of
the DC power source 5 is applied. In this embodiment, the value of
resistance R of the armature coil 16, shown in the above formulae
(4) and (5), is set so that a current beyond the allowable current
value will now flow through the power module 10 and the AC motor
9.
[0076] In order to prevent an overcurrent state, i.e., a state of a
current flowing beyond the allowable current value, when the
vehicle is running at low speeds or is stopped and the impedance of
the armature coil 16 is reduced, the pulse width of the applied
voltage may be narrowed from the original setting, i.e.,
180.degree., to a value smaller than 180.degree. to reduce the
current flowing through the armature coil 16. In other words, the
voltage may be applied to the armature coil 16 at the duty smaller
than the half cycle (180.degree.) of the electrical angle
.theta..
[0077] Such a case will be described below with reference to FIG.
7. (A) in FIG. 7 represents voltages induced in the armature coils
16 per phase, and (B) represents voltage commands Vu, Vv and Vw
applied to the armature coils 16 per phase. Also, (C) represents
the magnetic pole position, i.e., the electrical angle .theta., and
(D) represents switching command signals applied to the switching
devices (UP to WN).
[0078] In this embodiment, as shown in FIG. 7(B), the applied
voltage commands Vu, Vv and Vw are each supplied at the electrical
angle covering the range of 180.degree..
[0079] FIG. 8 shows the relationship between the pulse width (duty)
of the voltage applied to the armature coil 16 per phase and the
rotation speed .omega. of the AC motor 9. In the region where the
rotation speed .omega. of the AC motor 9 is high, the pulse width
of the applied voltage is set to 180.degree.. In the region where
the rotation speed .omega. of the AC motor 9 is lower than a
predetermined speed .omega.1, the pulse width of the applied
voltage is gradually narrowed toward a lower limit value x.sup.o as
the rotation speed reduces. As a result, the overcurrent state can
be prevented from occurring when the vehicle is running at low
speeds or is stopped.
[0080] When the voltage of the DC power source 5 increases, there
is also a risk that an overcurrent may flow in the armature coils
16. Accordingly, when the voltage of the DC power source 5
increases in excess of a predetermined value, it is also possible
to narrow the pulse width of the applied voltage as shown in FIG.
7.
[0081] FIG. 9 shows the relationship between the pulse width of the
voltage applied to the armature coil 16 per phase and the voltage
of the DC power source (battery) 5. In the region where the voltage
of the DC power source 5 is not higher than a predetermined voltage
V1, the pulse width of the applied voltage is set to 180.degree..
When the voltage of the DC power source 5 becomes higher than the
predetermined voltage V1, the pulse width of the applied voltage is
gradually narrowed toward a lower limit value x.sup.o. As a result,
the overcurrent state can be prevented from occurring when the
voltage of the DC power source 5 increases excessively.
[0082] Further, when the temperature of the switching devices (UP
to WN) or the temperature of the AC motor 9 rises, the pulse width
of the applied voltage may be narrowed to reduce the currents
flowing through the armature coils 16 and the switching devices (UP
to WN) so that those temperatures will not exceed respective
allowable values.
[0083] FIG. 19 shows the relationship between the pulse width of
the voltage applied to the armature coil 16 per phase and the
temperature of the switching devices (UP to WN). In the region
where the temperature of the switching devices (UP to WN) is not
higher than a predetermined temperature TI1, the pulse width of the
applied voltage is set to 180.degree.. When the temperature of the
switching devices (UP to WN) becomes higher than the predetermined
temperature TI1, the pulse width of the applied voltage is
gradually narrowed toward a lower limit value x.sup.o.
[0084] FIG. 11 shows the relationship between the pulse width of
the voltage applied to the armature coil 16 per phase and the
temperature of the armature coil 16. In the region where the
temperature of the armature coil 16 is not higher than a
predetermined temperature TM1, the pulse width of the applied
voltage is set to 180.degree.. When the temperature of the armature
coil 16 becomes higher than the predetermined temperature TM1, the
pulse width of the applied voltage is gradually narrowed toward a
lower limit value y.sup.o.
[0085] Thus, in this embodiment, the armature coils 16 and the
switching devices (UP to WN) are prevented from coming into the
excessively high temperature state (overheated state) by narrowing
the pulse width of the voltage applied to the armature coil 16 per
phase. In this respect, the pulse width used in practice is
determined by comparing the narrowed pulse width of the applied
voltage necessary to prevent an excessive temperature rise of the
switching devices (UP to WN) with the narrowed pulse width of the
applied voltage necessary to prevent an excessive temperature rise
of the armature coil 16, and then selecting a smaller one.
[0086] Of the construction of the controller 12, a portion
regarding the electricity generation control will be described
below with reference to FIG. 12. The controller 12 comprises the
power-running/electricity-genera- tion changing-over section 19 for
determining which one the power running control and the electricity
generation control is to be performed, and changing over the power
running control and the electricity generation control, and an
electricity generation control section 20G for executing the
synchronous rectification control. The electricity generation
control section 20G comprises a magnetic pole position detecting
unit 21 for detecting the magnetic pole position or the electrical
angle of the AC motor 9, a speed computing unit 22 for computing
the rotation speed .omega. of the AC motor 9, an interlinkage
magnetic-flux amount computing unit 23 for computing the amount
(number) of magnetic flux .phi. interlinking with the armature
coils 16 of the stator, an induced voltage computing unit 26 for
computing the voltages Vue, Vve and Vwe of the respective phases
induced in the armature coils 16 of the AC motor 9, a DC voltage
detecting unit 27 for detecting a voltage VB of the DC power source
5, a voltage comparing unit 28 for comparing inter-line values of
the induced voltages Vue, Vve and Vwe with the voltage VB of the DC
power source 5, a pulse cycle computing unit 29 for computing,
based on the comparison results of the voltage comparing unit 28,
pulse cycles of the switching signals applied to the switching
devices (UP to WN) of the power module 10, and a pulse generating
unit 25 for producing the switching signals.
[0087] The operation of the pulse cycle computing unit 29 will be
described below with reference to FIGS. 13 and 14. FIG. 13 is a
time chart for explaining the operation when the battery voltage VB
is lower than a lower limit Ve of the induced voltage inter-line
value, and FIG. 14 is a time chart for explaining the operation
when the battery voltage VB is higher than the lower limit Ve of
the induced voltage inter-line value. In FIGS. 13 and 14, (A)
represents waveforms of the induced voltages Vue, Vve and Vwe of
the respective phases in the AC motor 9, (B) represents the
inter-line values of the induced voltages, (C) represents the
magnetic pole position signal .theta., and (D) represents waveforms
of the pulse signals applied to the switching devices (UP to WN)
per phase, which are controlled based on the induced voltages and
the battery voltage VB.
[0088] Assuming that maximum values Eu0, Ev0 and Ew0 of the induced
voltages of the respective phases are each em, the relationships
between the magnetic pole position .theta. and the induced voltages
of the respective phases are expressed by the following
formulae:
Vue=-Eu0.multidot.sin .theta.=-em.multidot.sin .theta. (6)
Vve-Ev0.multidot.sin (.theta.+2.PI./3)=-em.multidot.sin
(.theta.+2n/3) (7)
Vwe=-Ew0.multidot.sin (.theta.-2.PI./3)=-em.multidot.sin
(.theta.-2.PI./3) (8)
[0089] Further, as shown at (A) and (B) in FIGS. 13 and 14, the
time at which the induced voltage inter-line value reaches the
lower limit value Ve is the same as the time at which the phase
voltage value of the induced voltage becomes em/2.
[0090] A description is first made of the operation in the
electricity generation mode when the battery voltage VB is lower
than the lower limit Ve of the induced voltage inter-line value as
shown in FIG. 13. In this case, the rectifying operation is
performed such that the switching devices (UP to WN) are
successively made conducted in order depending on the magnitudes of
the induced voltages Vue, Vve and Vwe for converting the induced
voltages into a DC voltage and charging the DC voltage in the DC
power source 5. More specifically, the switching devices UP, VP and
WP on the upper arm side are successively turned on into the
conducted state in the phases where the induced voltages of the
respective phases take maximum values. On the other hand, the
switching devices UN, VN and WN on the lower arm side are
successively turned on into the conducted state in the phases where
the induced voltages of the respective phases take minimum values.
By thus successively bringing the switching devices into the
conducted state depending on the magnitudes of the induced
voltages, the rectifying operation is performed and the resulting
DC voltage is charged in the DC power source 5.
[0091] A description is next made of the operation when the battery
voltage VB is higher than the lower limit Ve of the induced voltage
inter-line value as shown in FIG. 14. In the case of the battery
voltage VB being relatively high as shown in FIG. 14, the switching
devices (UP to WN) on the upper arm side are turned on into the
conducted state, as seen from FIG. 14, in the phases where the
values of the phase voltage values of the induced voltages are
maximized and in the range where the induced voltage inter-line
values are higher than the battery voltage VB. On the lower arm
side, the switching devices (UP to WN) are turned on into the
conducted state in the phases where the values of the induced phase
voltages are minimized and in the range shown in FIG. 14(D).
[0092] The operation of the synchronous rectification control in
the electricity generation mode according to this embodiment of the
present invention will be described below with reference to FIG.
15. First, in step S1, it is determined, in accordance with a
changeover command signal 18 from the external controller, which
one of the power running control and the electricity generation
control is to be performed. If the synchronous rectification
control is to be performed, the control flow proceeds to step S10.
Then, in step S10, the magnetic pole position .theta. is computed
based on a pulse signal outputted from the magnetic pole position
detecting means 13. In step S11, the actual rotation speed .omega.
f the AC motor 9 is computed based on a time-dependent change of
the pulse signal outputted from the magnetic pole position
detecting means 13. In step S12, the amount of magnetic flux .phi.
interlinking with the armature coils 16 of the stator is computed
from both the rotation speed .omega. computed in step S11 and the
excitation current If in the excitation coil 14 of the rotor
detected by the excitation driving circuit 15. In this embodiment,
a map for the amount of interlinkage magnetic flux .phi. is
prepared in advance from calculations based on experimental
results, for example, with the excitation current If and the
rotation speed .omega. being parameters, and the corresponding
amount of interlinkage magnetic flux .phi. is determined from the
map.
[0093] In step S13, the induced voltages (Vue, Vve, Vwe) of the
respective phases are computed from the amount of interlinkage
magnetic flux .phi., the rotation speed .omega. and the magnetic
pole position .theta., which were computed respectively in steps
12, 11 and 10, by using the above formulae (6), (7) and (8).
Further, inter-line values (Vuv, Vvw, Vwu) of the induced voltages
are computed in accordance with the following formulae:
Vuv=Vue-Vve (9)
Vvw=Vve-Vwe (10)
Vwu=Vwe-Vue (11)
[0094] Then, in step S14, the voltage of the DC power source 5 is
detected. In step S15, the battery voltage VB detected in step S14
is compared with the inter-line values (Vuv, Vvw, Vwu) of the
induced voltages computed in step S13. Based on the comparison
results in step S15, cycles of respective pulses applied to the
switching devices (UP to WN) are computed in step S16 as described
above in connection with FIGS. 13 and 14. Finally, pulse output
processing is executed in step S6, whereby the synchronous
rectification control is completed.
[0095] With this embodiment, in the electricity generation mode,
the synchronous rectification control is performed and the
generated voltage is rectified by combined use of the switching
device and the diode connected to the switching device in
anti-parallel relation. Accordingly, a loss during the
rectification can be reduced and the efficiency of the electricity
generation can be increased. Further, since the switching speed is
relatively slow as in the rectangular-wave driving control, a
smoothing capacitor is no longer required and the size of the motor
control device can be reduced.
[0096] In the synchronous rectification control described above,
the timings of switching over the switching devices (UP to WN) are
computed through steps of computing the induced voltages (Vue, Vve,
Vwe) and comparing the inter-line values (Vuv, Vvw, Vwu) of the
induced voltages with the battery voltage VB. As an alternative,
however, it is also possible to provide means for detecting the
potential between two terminals (i.e., the source-drain potential)
of each of the switching devices (UP to WN), and to switch over the
switching device when the detected potential has become lower than
a predetermined value. This modification utilizes the fact that
when the current generated by the AC motor 9 is rectified by each
switching device, the diode connected to the switching device in
anti-parallel relation is made conducted and the potential between
two terminals (i.e., the source-drain potential) of the switching
device connected tp the conducted diode is reduced to a value near
the forward voltage of the conducted diode.
[0097] The AC motor 9 is coupled to the internal combustion engine
1 through the power transmitting means 2. Even with the internal
combustion engine 1 running at idle, therefore, it is required that
the AC motor 9 be capable of generating electricity. As described
above, the AC motor 9 is capable of generating electricity when the
inter-line values (Vuv, Vvw, Vwu) of the induced voltages are
higher than the battery voltage VB. Hence, the inter-line values
(Vuv, Vvw, Vwu) of the induced voltages in the AC motor 9 must be
set to have regions of levels higher than the battery voltage VB
even when the internal combustion engine 1 is running at idle.
[0098] One example of a method for changing over the power running
control and the electricity generation control from one to the
other will be described below with reference to FIG. 16. In the
above description, the power running control and the electricity
generation control are changed over from one to the other in
accordance with the changeover command signal 18 from the external
controller. However, the power running control and the electricity
generation control may be changed over in accordance with the
rotation speed .omega. of the AC motor 9. In this case, as shown in
FIG. 16, when the rotation speed .omega. of the AC motor 9 is not
higher than a predetermined speed .omega.0, the power running
control is performed, and when it is higher than the predetermined
speed .omega.0, the electricity generation control is performed.
Stated another way, when the rotation speed .omega. of the AC motor
9 is in the range of 0 to the predetermined speed .omega.0, the
power running control is performed to start up the internal
combustion engine 1 until the revolution speed of the internal
combustion engine 1 increases to an idle speed. Then, when the
internal combustion engine 1 comes into a complete firing state and
the rotation speed .omega. of the AC motor 9 exceeds the
predetermined speed .omega.0, the mode is changed over to the
electricity generation control.
[0099] Another example of the method for changing over the power
running control and the electricity generation control will be
described below with reference to FIG. 17. In this example, the
power running control is performed during a period from inputting
of a start command signal to the internal combustion engine 1 to a
predetermined time t0. After that, the mode is changed over to the
electricity generation control. Stated another way, during the
period from inputting of the start command signal to the internal
combustion engine 1 to the predetermined time t0, the power running
control is performed to start up the internal combustion engine 1.
When the predetermined time t0 has lapsed, this is judged as
indicating that the internal combustion engine 1 is in the complete
firing state, and the mode is changed over to the electricity
generation control.
[0100] The construction of a motor control system according to
another embodiment of the present invention will be described below
with reference to FIGS. 18 to 29.
[0101] A description is first made of the construction of the motor
control system using the motor control device according to this
embodiment with reference to FIG. 18.
[0102] FIG. 18 is a block diagram of the motor control system using
the motor control device according to another embodiment of the
present invention.
[0103] A motor control device 3 according to this embodiment
comprises a power module 10 and a power module control circuit 12A.
A DC voltage of a battery 5 is converted into an AC voltage by the
power module 10, and the AC voltage is supplied to armature coils
16 of a stator of an AC motor 9. Magnetic pole positions of U, V
and W phases are detected respectively by a U-phase magnetic pole
position detecting means. 13U, a V-phase magnetic pole position
detecting means 13V, and a W-phase magnetic pole position detecting
means 13W. An excitation current flowing through an excitation coil
14 of a rotor of the AC motor 9 is controlled by an excitation
current control circuit 15. The excitation current control circuit
15 is controlled by an external controller (host controller). The
external controller (host controller) supplies a mode changeover
signal 18 to the power module control circuit 12A for changing over
the power running mode and the electricity generation mode from one
to the other.
[0104] In the power module 10, UP denotes a U-phase upper arm
MOSFET, DUP denotes a parasitic diode of UP, VP denotes a V-phase
upper arm MOSFET, DVP denotes a parasitic diode of VP, WP denotes a
W-phase upper arm MOSFET, and DWP denotes a parasitic diode of WP.
Respective drains of the MOSFET's are connected in common to a
positive pole of the battery 5. Also, UN denotes a U-phase lower
arm MOSFET, DUN denotes a parasitic diode of UN, VN denotes a
V-phase lower arm MOSFET, DVN denotes a parasitic diode of VN, WN
denotes a W-phase lower arm MOSFET, and DWN denotes a parasitic
diode of WN. Respective sources of the MOSFET's are connected in
common to a negative pole of the battery 5 and are grounded. The
source of the U-phase upper arm MOSFET (UP) and the drain of the
U-phase lower arm MOSFET (UN) are connected in common and also
connected to a U-terminal of the AC motor 9. The source of the
V-phase upper arm MOSFET (VP) and the drain of the V-phase lower
arm MOSFET (VN) are connected in common and also connected to a
V-terminal of the AC motor 9. The source of the W-phase upper arm
MOSFET (WP) and the drain of the W-phase lower arm MOSFET (WN) are
connected in common and also connected to a W-terminal of the AC
motor 9.
[0105] In the power module control circuit 12A, a U-phase upper arm
drive signal changing-over circuit 30a selectively changes over a
power running drive signal 34a and a rectification drive signal 33a
in accordance with the mode changeover signal 18, and supplies the
selected drive signal to a gate of the U-phase upper arm MOSFET
(UP). A U-phase lower arm drive signal changing-over circuit 30b
selectively changes over a power running drive signal 34b and a
rectification drive signal 33b in accordance with the mode
changeover signal 18, and supplies the selected drive signal to a
gate of the U-phase lower arm MOSFET (UN).
[0106] A U-phase upper arm rectification detecting/driving circuit
31a compares the magnitude of a positive pole voltage VB of the
battery with the magnitude of a voltage VU at the U-terminal and
generates a positive voltage pulse in the condition of VU>VB. A
U-phase lower arm rectification detecting/driving circuit 31b
compares the magnitude of the voltage VU at the U-terminal with the
magnitude of a negative pole voltage VG (0 V) of the battery and
generates a positive voltage pulse in the condition of
VU<VG.
[0107] A U-phase lead-angle drive signal distribution circuit 32
generates, in the power running mode, a voltage pulse with a phase
leading relative to the phase of an output voltage pulse signal hu
from the U-phase magnetic pole position detecting means 13U.
Although a phase angle advancing method executed by the U-phase
lead-angle drive signal distribution circuit 32 is described later
with reference to FIG. 19, a voltage pulse with a phase leading in
appearance relative to the phase of the output voltage pulse signal
hu is generated by causing the phase of an output voltage pulse
signal hw from the W-phase magnetic pole position detecting means
13W to lag depending on the rotation speed of the armature coils
16.
[0108] A V-phase upper arm drive signal changing-over circuit 40a
selectively changes over a power running drive signal 44a and a
rectification drive signal 43a in accordance with the mode
changeover signal 18, and supplies the selected drive signal to a
gate of the V-phase upper arm MOSFET (VP). A V-phase lower arm
drive signal changing-over circuit 40b selectively changes over a
power running drive signal 44b and a rectification drive signal 43b
in accordance with the mode changeover signal 18, and supplies the
selected drive signal to a gate of the V-phase lower arm MOSFET
(VN).
[0109] A V-phase upper arm rectification detecting/driving circuit
41a compares the magnitude of the positive pole voltage VB of the
battery with the magnitude of a voltage W at the V-terminal and
generates a positive voltage pulse in the condition of VV>VB. A
V-phase lower arm rectification detecting/driving circuit 41b
compares the magnitude of the voltage VV at the V-terminal with the
magnitude of the negative pole voltage VG (0 V) of the battery and
generates a positive voltage pulse in the condition of
VV<VG.
[0110] A V-phase lead-angle drive signal distribution circuit 42
generates, in the power running mode, a voltage pulse with a phase
leading relative to the phase of an output voltage pulse signal hv
from the V-phase magnetic pole position detecting means 13V.
[0111] A W-phase upper arm drive signal changing-over circuit 50a
selectively changes over a power running drive signal 54a and a
rectification drive signal 53a in accordance with the mode
changeover signal 18, and supplies the selected drive signal to a
gate of the W-phase upper arm MOSFET (WP). A W-phase lower arm
drive signal changing-over circuit 50b selectively changes over a
power running drive signal 54b and a rectification drive signal 53b
in accordance with the mode changeover signal 18, and supplies the
selected drive signal to a gate of the W-phase lower arm MOSFET
(WN).
[0112] A W-phase upper arm rectification detecting/driving circuit
51a compares the magnitude of the positive pole voltage VB of the
battery with the magnitude of a voltage VW at the W-terminal and
generates a positive voltage pulse in the condition of VW>VB. A
W-phase lower arm rectification detecting/driving circuit 51b
compares the magnitude of the voltage VW at the W-terminal with the
magnitude of the negative pole voltage VG (0 V) of the battery and
generates a positive voltage pulse in the condition of
VW<VG.
[0113] A W-phase lead-angle drive signal distribution circuit 52
generates, in the power running mode, a voltage pulse with a phase
leading relative to the phase of an output voltage pulse signal hw
from the W-phase magnetic pole position detecting means 13W.
[0114] The circuit operation in this embodiment will be described
below. Note that since the operation is the same with respect to
the U, V and W phases in this embodiment, a description is made of
only the operation in U-phase. The motor control device of this
embodiment has two functions of executing power running control and
synchronous driving control for the AC motor/AC generator 9.
[0115] The power running control will first be described. When a
signal for starting the operation for the power running control
(i.e., the mode changeover signal 18) from the external controller
is inputted to the U-phase upper arm drive signal changing-over
circuit 30a, the U-phase upper arm drive signal changing-over
circuit 30a selects the power running drive signal 34a and outputs
a signal identical to the power running drive signal 34a to the
gate of the U-phase upper arm MOSFET (UP). The power running drive
signal 34a has a low level at the same potential as that at the
U-terminal and has a high level at a potential higher than the
threshold of the U-phase upper arm MOSFET (UP). Similarly, when the
signal for starting the operation for the power running control
(i.e., the mode changeover signal 18) is inputted to the U-phase
lower arm drive signal changing-over circuit 30b, the U-phase lower
arm drive signal changing-over circuit 30b selects the power
running drive signal 34b and outputs a signal identical to the
power running drive signal 34b to the gate of the U-phase lower arm
MOSFET (UN). The power running drive signal 34b has a low level at
the same potential as the negative pole potential VG (0 V) of the
battery and has a high level at a potential higher than the
threshold of the U-phase lower arm MOSFET (UN). In order to prevent
the U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET
(UN) from turning on at the same time, the power running drive
signal 34b takes a low level when the power running drive signal
34a takes a high level, and conversely the power running drive
signal 34b takes a high level when the power running drive signal
34a takes a low level. Additionally, depending on an input capacity
Ciss and gate resistance Rg of the U-phase upper arm MOSFET (UP)
and the U-phase lower arm MOSFET (UN), the power running drive
signal 34a and the power running drive signal 34b are set to have a
dead time (period during which both the power running drive signals
34a, 34b take a low level concurrently) not shorter than the time
constant (product of Ciss and Rg).
[0116] The U-phase upper arm MOSFET (UP) and the U-phase lower arm
MOSFET (UN) starts switching with the power running drive signals
34a, 34b, respectively, whereupon a current flows into the armature
coil 16 and the rotor stars rotation. At the same time, the
magnetic pole position detecting means 13U, 13V and 13W for the
respective phases detect the magnetic pole positions and then
output the voltage pulse signals hu, hv and hw, respectively.
Although the phase relationships among the voltage pulse signals
differ depending on the number of magnetic poles of the rotor and
mechanical angles of the magnetic pole position detecting means,
the high level and the low level of each of the voltage pulse
signals hu, hv and hw are reversed for each 180.degree. of
electrical angle in this embodiment. The voltage pulse signal hw is
applied to the U-phase lead-angle drive signal distribution circuit
32 to delay the phase of the voltage pulse signal hw depending on
the rotation speed of the armature coil 16. A voltage pulse with a
phase leading in appearance relative to the phase of the voltage
pulse signal hu is thereby generated. The generated voltage pulse
is distributed as the upper-arm power running drive signal 34a and
the lower-arm power running drive signal 34b. The detailed
construction of the U-phase lead-angle drive signal distribution
circuit 32 will be described later with reference to FIG. 19. The
motor control device operates in such a way for the power running
control in the U phase, and it also similarly operates in the other
V and W phases.
[0117] The operation for the rectification driving control will be
described below. When the mode changeover signal 18 is inputted to
the U-phase upper arm drive signal changing-over circuit 30a, the
U-phase upper arm drive signal changing-over circuit 30a selects
the rectification drive signal 33a and outputs a signal identical
to the rectification drive signal 33a to the gate of the U-phase
upper arm MOSFET (UP). The rectification drive signal 33a has a low
level at the same potential as that at the U-terminal and has a
high level at a potential higher than the threshold of the U-phase
upper arm MOSFET (UP). Similarly, when the mode changeover signal
18 is inputted to the U-phase lower arm drive signal changing-over
circuit 30b, the U-phase lower arm drive signal changing-over
circuit 30b selects the rectification drive signal 33b and outputs
a signal identical to the rectification drive signal 33b to the
gate of the U-phase lower arm MOSFET (UN). The rectification drive
signal 33b has a low level at the same potential as the negative
pole potential VG (0 V) of the battery and has a high level at a
potential higher than the threshold of the U-phase lower arm MOSFET
(UN).
[0118] With the rotation of the rotor, the U-, V- and W-terminals
generate positive induced voltages VU, VV and VW, respectively.
When the positive pole voltage VB of the battery and the voltage VU
at the U-terminal are inputted to the U-phase upper arm
rectification detecting/driving circuit 31a, the circuit 31a
compares the magnitude of the voltage VB with the magnitude of the
voltage VU and generates a positive voltage with respect to the
potential VU at the U-terminal as a reference in the condition of
VU >VB. The potential difference of this positive voltage
provides a voltage Vra higher than the threshold Vth of the U-phase
upper arm MOSFET (UP). Also, in the condition of VU.ltoreq.VB, the
U-phase upper arm rectification detecting/driving circuit 31a
outputs a voltage substantially at the same potential as that at
the U-terminal. Thus, the rectification drive signal 33a outputted
from the U-phase upper arm rectification detecting/driving circuit
31a is given as a voltage pulse having the voltage Vra (high level)
in the condition of VU>VB with the potential at the U-terminal
being a reference, and having substantially 0 V (low level) in the
condition of VU.ltoreq.VB. The U-phase lower arm rectification
detecting/driving circuit 31b operates substantially in a similar
manner.
[0119] When the voltage VU at the U-terminal and the negative pole
voltage VG of the battery are inputted to the U-phase lower arm
rectification detecting/driving circuit 31b, the circuit 31b
compares the magnitude of the voltage VU with the magnitude of the
voltage VG and generates a positive voltage with respect to VG (0
V) as a reference in the condition of VG>VU. The potential
difference of this positive voltage provides a voltage Vrb higher
than the threshold Vth of the U-phase lower arm MOSFET (UN). Also,
in the condition of VG.ltoreq.VU, the U-phase lower arm
rectification detecting/driving circuit 31b outputs a voltage
substantially at the same potential as VG (0 V). Thus, the
rectification drive signal 33b outputted from the U-phase lower arm
rectification detecting/driving circuit 31b is given as a voltage
pulse having the voltage Vrb (high level) in the condition of
VG>VU with VG (0 V) being a reference, and having substantially
0 V (low level) in the condition of VG.ltoreq.VU. Those
rectification drive signals 33a, 33b are applied to the gates of
the U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET
(UN) through the U-phase upper arm rectification detecting/driving
circuit 31a and the U-phase lower arm rectification
detecting/driving circuit 31b, respectively, thereby switching over
the U-phase upper arm MOSFET (UP) and the U-phase lower arm MOSFET
(UN). The motor control device operates in such a way for the MOS
rectification driving control in the U phase, and it also similarly
operates in the other V and W phases.
[0120] With the operation described above, in the power running
mode, the phases of the magnetic pole position voltage pulses are
advanced and the switching devices for the respective phases are
switched over by the advanced voltage pulses. Therefore, the
currents of the respective phases are advanced in phase relative to
the induced voltages of the respective phases. As a result, high
torque can be obtained even at high rotation speeds. Further, the
rectification efficiency can be increased by utilizing the MOS
rectification.
[0121] The construction and operation of the U-phase lead-angle
drive signal distribution circuit 32 used in the motor control
device of this embodiment will be described below with reference to
FIGS. 19 to 25. Although the following description is made of, by
way of example, the operation of the U-phase lead-angle drive
signal distribution circuit 32 for the convenience of explanation,
a V-phase lead-angle drive signal distribution circuit 42 and a
W-phase lead-angle drive signal distribution circuit 52 also have
similar constructions and operations.
[0122] With reference to FIG. 19, a description is first made of
the construction of the U-phase lead-angle drive signal
distribution circuit 32 used in the motor control device according
of this embodiment.
[0123] FIG. 19 is a block diagram of the lead-angle drive signal
distribution circuit used in the motor control device according to
another embodiment of the present invention.
[0124] A frequency-voltage (f-V) converter 209 comprises a one-shot
multivibrator 201 and an integrator 202. The one-shot multivibrator
201 detects the rising of the voltage pulse signal hw outputted
from the magnetic pole position detecting means 13W, and then
outputs a voltage pulse having a predetermined pulse width TW1.
Assuming the width of the pulse outputted from the magnetic pole
position detecting means 13W to be TW2, there is a relationship of
TW1<TW2. The integrator 202 receives the voltage pulse outputted
from the one-shot multivibrator 201 and integrates it over time,
thereby outputting a DC voltage VDC. Thus, the frequency-voltage
(f-V) converter 209 produces a voltage signal V.sub.DC in
proportion to the frequency of the voltage pulse signal hw
outputted from the magnetic pole position detecting means 13W,
i.e., the rotation speed of the AC motor 9.
[0125] A capacity charging/discharging current source 210 comprises
a capacity charging current source 203a, a capacity discharging
current source 203b, a capacity charging switch 204a, and a
capacity charging switch 204b, and an inverter 205. The capacity
charging current source 203a delivers a DC current from a power
source V.sub.CC, which is supplied to operate the U-phase
lead-angle drive signal distribution circuit 32, depending on the
voltage value of the DC voltage V.sub.DC. The capacity discharging
current source 203b draws a DC current depending on the voltage
value of the DC voltage V.sub.DC. The capacity charging switch 204a
receives the voltage pulse signal hw and makes switching of the
output current from the capacity charging current source 203a. The
inverter 205 receives the voltage pulse signal hw and outputs an
inverted one 205a of the inputted voltage pulse signal. The
capacity discharging switch 204b receives the inverted voltage
pulse signal outputted from the inverter 205 and makes switching of
the output current from the capacity discharging current source
203b.
[0126] A capacity C integrates the DC current and generates a
triangular wave. The slope of the triangular wave changes depending
on the voltage value of the DC voltage V.sub.DC, i.e., the rotation
speed of the AC motor. The higher the rotation speed of the AC
motor, the larger is the slope of the triangular wave. A comparator
206 receives the triangular wave at its non-inverting input
terminal and a reference voltage (V.sub.CC/2) at its inverting
input terminal, and compares those two voltage values, thereby
outputting an output voltage pulse hu'.
[0127] An upper/lower arm drive signal distribution circuit 207
distributes the voltage pulse hu' as an upper arm drive signal and
a lower arm drive signal. Namely, it outputs an upper arm drive
signal 207a and the lower-arm power running drive signal 34b. A
level shift-up circuit 208 converts a reference voltage level of
the upper arm drive signal 207a from VG (0 V) into VU. An output
voltage of the level shift-up circuit 208 serves as the upper-arm
power running drive signal 34a. Incidentally, the upper-arm power
running drive signal 34a is the voltage pulse signal described
above with reference to FIG. 18.
[0128] The one-shot multivibrator 201 is a generally used circuit
and can be constituted by, e.g., a universal IC or a bipolar
transistor. The integrator 202 may be formed using an operational
amplifier. The upper/lower arm drive signal distribution circuit
207 can be constituted by a general logic gate. Further, the level
shift-up circuit 208 can be easily formed using a switching element
and a resistor.
[0129] The construction and operation of a phase angle advancing
circuit in the U-phase lead-angle drive signal distribution circuit
32 will be described below.
[0130] When the voltage pulse hw is inputted to the one-shot
multivibrator 201, the one-shot multivibrator 201 detects the
rising of the voltage pulse hw and generates a voltage pulse that
sustains a high level for a period of the pulse width TW1 (sec)
(TW1<TW2) from the time of the rising. Assuming that the voltage
pulse hw has the frequency f (Hz) (>0), the pulse width TW2
(sec), and the duty of 50%, TW2=1/(2f) is resulted and the
frequency of the voltage pulse outputted from the one-shot
multivibrator 201 remains f, i.e., the same as that of the voltage
pulse hw. Thus, the one-shot multivibrator 201 has the function of
converting the voltage pulse hw into a voltage pulse having the
frequency f and the pulse width TW1. The one-shot multivibrator
itself is known as a general universal analog IC. In general, the
pulse width TW1 can be optionally set with the time constant
defined by an external capacity and resistance. When the voltage
pulse is inputted to the integrator 202, it is integrated over time
and converted into the DC voltage V.sub.DC. The frequency f and the
DC voltage VDC are proportional to each other and expressed by: 1 V
DC = kf ( k > 0 ) = V H TW1 / T = V H TW1 f ( 13 )
[0131] In the above formula, V.sub.H is the high level voltage of
the voltage pulse hw. The frequency-voltage converter 209
comprising the one-shot multivibrator 201 and the integrator 202
may be constituted by a general universal analog IC.
[0132] The output V.sub.DC of the integrator 202 is applied to the
capacity charging current source 203a and the capacity discharging
current source 203b for conversion into a constant current
I.sub.DC. The relationship between V.sub.DC and I.sub.DC is
described later with reference to FIG. 22. When the capacity
charging switch 204a is turned on by the voltage pulse signal hw,
the constant current I.sub.DC is charged in the capacity C, and
when the capacity charging switch 204b is turned on by the inverted
voltage pulse signal 205a of the voltage pulse signal hw, the
constant current I.sub.DC is discharged from the capacity C.
Assuming that the time until the charged voltage reaches V.sub.CC/2
from the start of charging into the capacity C is Td (sec), the
following relationship holds:
I.sub.DC=C.multidot.V.sub.CC/(2Td) (14)
[0133] As a result of repeating the charging and discharging into
and from the capacity C, a triangular wave is generated and applied
to the non-inverting input terminal of the comparator 206. In the
comparator 206, the voltage of the triangular wave is compared with
the reference voltage V.sub.CC/2 on the non-inverting input
terminal. Then, the comparator 206 outputs a voltage V.sub.CC (high
level) when the voltage at the non-inverting input terminal is
larger than V.sub.CC/2, and outputs a voltage 0 V (low level) when
the voltage at the non-inverting input terminal is smaller than or
equal to V.sub.CC/2. Accordingly, the voltage pulse hu' outputted
from the comparator 206 is delayed Td in phase relative to the
voltage pulse hw. The delay time Td is converted into an electrical
angle .theta.d (degree) as follows:
.theta.d=360.multidot.Td.multidot.f (15)
[0134] While the phase of the voltage pulse hu' lags .theta.d
(degree) relative to the phase of the voltage pulse hw, it is
supposed that the phase of the voltage pulse hu' leads .theta.c
(degree) relative to the phase of the voltage pulse hu. Because the
phase of the voltage pulse hw leads 120 (degrees) relative to the
phase of the voltage pulse hu, the following formula is
obtained:
.theta.c=.theta.d-120 (16)
[0135] Using the formulae (13), (14) and (15), the formula (16) is
modified into:
.theta.c={360.multidot.C.multidot.V.sub.CC.multidot.f/(2I.sub.DC)}-120
(17)
[0136] With reference to FIG. 20, a description is now made of the
relationship between the rotation frequency and the lead angle,
which is provided by the lead-angle drive signal distribution
circuit 32 used in the motor control device of this embodiment.
[0137] FIG. 20 is a graph for explaining the relationship between
the rotation frequency and the lead angle, which is provided by the
lead-angle drive signal distribution circuit used in the motor
control device according to another embodiment of the present
invention.
[0138] Assuming, as shown in FIG. 20, that the following
relationship holds between .theta.c and f,
.theta.c=-Kf(K>0, f1>0) (18)
[0139] the relationship between I.sub.DC and f is expressed as
follows from the formula (17);
I.sub.DC=180.multidot.C.multidot.V.sub.CC.multidot.f/(120-Kf)
(19)
[0140] where Kf<120 must be satisfied.
[0141] With reference to FIG. 21, a description is now made of the
relationship between the rotation frequency and a current value of
a constant current source, which is provided by the lead-angle
drive signal distribution circuit 32 used in the motor control
device of this embodiment.
[0142] FIG. 21 is a graph for explaining the relationship between
the rotation frequency and a current value of a constant current
source, which is provided by the lead-angle drive signal
distribution circuit used in the motor control device according to
another embodiment of the present invention.
[0143] FIG. 21 plots a curve representing the formula (19). The
curve is defined in the range of 0<f<120/K and is downwardly
convexed. By designing the constant current sources 203a and 203b
so as to satisfy the relationship of the formula (19), therefore,
the relationship of the formula (18) always holds and the phase
.theta.c leads a larger angle as the frequency f increases.
[0144] With the lead-angle drive signal distribution circuit
constructed as described above, in the power running mode, the
phases of the magnetic pole position voltage pulses are advanced
and the switching devices for the respective phases are switched
over by the advanced voltage pulses. Therefore, the currents of the
respective phases are advanced in phase relative to the induced
voltages of the respective phases. As a result, high torque can be
obtained even at high rotation speeds.
[0145] With reference to FIG. 22, a description is now made of the
construction of the constant current source 203 (203a or 203b) used
in the lead-angle drive signal distribution circuit 32 of the motor
control device of this embodiment.
[0146] FIG. 22 is a circuit diagram of the constant current source
used in the lead-angle drive signal distribution circuit of the
motor control device according to another embodiment of the present
invention.
[0147] Because the current characteristic shown in FIG. 21 is
similar to a voltage-current characteristic of a diode, the
constant current sources 203a, 203b used in this embodiment are
each formed by utilizing an exponential function characteristic of
a diode. Note that the construction shown in FIG. 22 is merely one
example of a constant current source circuit corresponding to the
equation (19), and any other suitable circuit can also be used
instead.
[0148] The DC voltage V.sub.DC is inputted to a voltage input
terminal 501. The constant current source 203 comprises an
operational amplifier 502, a first pnp-transistor 503, an
npn-transistor 504, a first resistor 505, a diode 506, a second
resistor 507, a second pnp-transistor 508, and an output terminal
509 from which a DC current is outputted.
[0149] The operational amplifier 502 operates with the power source
V.sub.CC. It has a non-inverting input terminal connected to the
voltage input terminal 501, an inverting input terminal connected
to an emitter of the npn-transistor 504, an output terminal
connected to a base of the npn-transistor 504. One terminal of the
first resistor 505 is connected to the emitter of the
npn-transistor 504, and the other terminal thereof is connected to
an anode of the diode 506 and one terminal of the second resistor
507. Further, a cathode of the diode 506 is connected to the other
terminal of the second resistor 507 and is also grounded. The first
pnp-transistor 503 and the second pnp-transistor 508 constitute a
current mirror circuit.
[0150] The operation of the constant current source 203 will be
described below. When the DC voltage V.sub.DC is inputted to the
voltage input terminal 501, the operational amplifier 502 operates
so as to hold the voltage at the non-inverting input terminal
thereof to be V.sub.DC, and hence the emitter voltage of the
npn-transistor 504 also becomes V.sub.DC. Accordingly, a current I
flows through the first resistor 505 and the second resistor
507.
[0151] Assuming that the resistance values of the first resistor
505 and the second resistor 507 are R1 and R2, respectively, and
the turning-on voltage of the diode 506 is Vd, a current flowing
through the first resistor 505 is expressed by I1=V.sub.DC/(R1+R2)
in the condition of I.times.R2<Vd. Also, the operational
amplifier 502 delivers a current to the base of the npn-transistor
504, thus causing the npn-transistor 504 to operate. If the
npn-transistor 504 has a sufficiently large current amplification
factor h.sub.FE (e.g., 300 or more), a collector current is almost
equal to the emitter current. Therefore, the collector current also
becomes I1. This current I1 is delivered from the output terminal
509 through the current mirror circuit.
[0152] In the condition of I1.times.R2.gtoreq.Vd, since the diode
506 is turned on, the current flowing through the first resistor
505 is expressed by I2=(V.sub.DC-Vd)/R2. This current I2 is
likewise delivered from the output terminal 509 through the current
mirror circuit. Thus, when V.sub.DC<Vd.times.(R1+R2)/R2 is
satisfied, the current I1=V.sub.DC/(R1+R2) is delivered from the
output terminal 509, and when V.sub.DC.gtoreq.Vd.times.(R1+R2)/R2
is satisfied, the current I2=(V.sub.DC-Vd)/R2 is delivered from the
output terminal 509. By eliminating V.sub.DC in those formula based
on the formula (13), I.sub.DC can be expressed as follows:
In the case of f<Vd.times.(R1+R2)/(kR2) I.sub.DC=kf/(R1+R2)
(20)
In the case of f.gtoreq.Vd.times.(R1+R2)/(kR2) I.sub.DC=(kf-Vd)/R2)
(21)
[0153] By adjusting R1 and R2 in the above formulae, a curve close
to that represented by the formula (19) can be realized.
Incidentally, from the formula (13), k=V.sub.H.multidot.TW1 is
given in which V.sub.H is the high level voltage of the voltage
pulse hw.
[0154] With reference to FIG. 23, a description is now made of the
construction of the capacity charging switch 204a and the capacity
discharging switch 204b both used in the lead-angle drive signal
distribution circuit 32 of the motor control device of this
embodiment.
[0155] FIG. 23 is a circuit diagram of the capacity charging switch
and the capacity discharging switch both used in the lead-angle
drive signal distribution circuit of the motor control device
according to another embodiment of the present invention. Note that
the construction shown in FIG. 23 is merely one example of the
capacity charging switch and the capacity discharging switch, and
any other suitable circuit can also be used instead.
[0156] A switch circuit 204 comprises a constant current input
terminal 601, a first npn-transistor 602, a second npn-transistor
603, a first pnp-transistor 604, a second pnp-transistor 605, a
third npn-transistor 606, a third pnp-transistor 607, a fourth
pnp-transistor 608, a fourth npn-transistor 609, a fifth
npn-transistor 610, a first pMOS-transistor 611, and a first
nMOS-transistor 612. Numeral 611a denotes a gate terminal of the
first pMOS transistor 611, 612a denotes a gate terminal of the
first nMOS transistor 612, and 613 denotes a constant current
output terminal.
[0157] The constant current input terminal 601 is connected to the
output terminal 509 shown in FIG. 22. The gate terminals 611a, 612a
are both connected to the output hw of the magnetic pole position
detecting means 13W shown in FIG. 19. The first npn-transistor 602,
the second npn-transistor 603, and the third npn-transistor 606
constitute a current mirror circuit drawing the same current as the
DC current I.sub.DC, which is inputted from the constant current
input terminal 601, through a collector terminal of the second
npn-transistor 603. Also, this current mirror circuit similarly
draws the same current as the DC current I.sub.DC, which is
inputted from the constant current input terminal 601, through a
collector terminal of the third npn-transistor 606.
[0158] The first pnp-transistor 604 and the second pnp-transistor
605 constitute a current mirror circuit. An emitter of the first
pnp-transistor 604 is connected to a source of the first
pMOS-transistor 611, and a collector of the first pnp-transistor
604 is connected to a drain of the first pMOS-transistor 611. The
first pMOS-transistor 611 serves to turn on/off the current
I.sub.DC flowing in this current mirror circuit. This current
mirror circuit and the first pMOS-transistor 611 cooperatively
realize the function of the charging constant current switch.
[0159] When the voltage pulse hw outputted from the magnetic pole
position detecting means 13W has a high level, the potential
difference between the gate and source of the first pMOS-transistor
611 is substantially zero and hence the first pMOS-transistor 611
is turned off. At this time, if the current I.sub.DC flows in the
collector of the first pnp-transistor 604, a voltage Vd (about 0.7
V) generates between the emitter and the base of the first
pnp-transistor 604. As a result, the voltage Vd (about 0.7 V) is
also applied between the emitter and the base of the second
pnp-transistor 605, whereby the second pnp-transistor 605 is turned
on and the current I.sub.DC flows into the collector of the second
pnp-transistor 605.
[0160] On the other hand, when the voltage pulse hw outputted from
the magnetic pole position detecting means 13W has a low level, the
potential difference between the gate and source of the first
pMOS-transistor 611 is increased and the first pMOS-transistor 611
is turned on when the potential difference exceeds the threshold of
the first pMOS-transistor 611. At this time, even if the current
I.sub.DC flows in the collector of the first pnp-transistor 604,
there generates just substantially zero voltage between the emitter
and the base of the first pnp-transistor 604. As a result, the
voltage between the emitter and the base of the second
pnp-transistor 605 also becomes substantially zero, whereby the
second pnp-transistor 605 is turned off and no current flows into
the collector of the second pnp-transistor 605.
[0161] Thus, this current mirror circuit is turned on and off
depending on the voltage pulse hw, thus causing the current
I.sub.DC to be delivered to the constant current output terminal or
stopped.
[0162] The third pnp-transistor 607 and the fourth pnp-transistor
608 constitute a current mirror circuit delivering the same current
as the DC current I.sub.DC, which is drawn from the collector of
the third npn-transistor 607, through a collector of the fourth
pnp-transistor 608. This current mirror circuit is intended to
produce the delivery current I.sub.DC for use in the discharging
constant current switch of the next stage. The fourth
npn-transistor 609 and the fifth npn-transistor 610 constitute a
current mirror circuit. An emitter of the fourth npn-transistor 609
is connected to a source of the first nMOS-transistor 612, and a
collector of the fourth npn-transistor 609 is connected to a drain
of the first nMOS-transistor 612. The first NMOS-transistor 612
serves to turn on/off the current I.sub.DC flowing in this current
mirror circuit. This current mirror circuit and the first
nMOS-transistor 612 cooperatively realize the function of the
discharging constant current switch.
[0163] When the voltage pulse hw outputted from the magnetic pole
position detecting means 13W has a high level, the potential
difference between the gate and source of the first nMOS-transistor
612 is increased and the first nMOS-transistor 612 is turned on
when the potential difference exceeds the threshold of the first
nMOS-transistor 612. At this time, even if the current I.sub.DC
flows in the collector of the fourth npn-transistor 609, there
generates just substantially zero voltage between the emitter and
the base of the fourth npn-transistor 609. As a result, the voltage
between the emitter and the base of the fifth npn-transistor 610
also becomes substantially zero, whereby the fifth npn-transistor
610 is turned off and no current flows into the collector of the
fifth npn-transistor 610.
[0164] On the other hand, when the voltage pulse hw outputted from
the magnetic pole position detecting means 13W has a low level, the
potential difference between the gate and source of the first
nMOS-transistor 612 is substantially zero and hence the first
nMOS-transistor 612 is turned off. At this time, if the current
I.sub.DC flows in the collector of the fourth npn-transistor 609, a
voltage Vd (about 0.7 V) generates between the emitter and the base
of the fourth npn-transistor 609. As a result, the voltage Vd
(about 0.7 V) is also applied between the emitter and the base of
the fifth npn-transistor 610, whereby the fifth npn-transistor 610
is turned on and the current I.sub.DC flows into the collector of
the fifth npn-transistor 610.
[0165] Thus, this current mirror circuit is turned on and off
depending on the voltage pulse hw, thus causing the current
I.sub.DC to be drawn from the constant current output terminal or
stopped.
[0166] The operation of the lead-angle drive signal distribution
circuit 32 of the motor control device of this embodiment will be
described below with reference to FIGS. 24 and 25.
[0167] FIGS. 24 and 25 are each a waveform chart showing the
operation of the lead-angle drive signal distribution circuit of
the motor control device according to another embodiment of the
present invention. FIG. 24 shows the angle advancing operation in
the case of the frequency f being low (f=f1). FIG. 25 shows the
angle advancing operation in the case of the frequency f being high
(f=f2).
[0168] With reference to FIG. 24, a description is first made of
the angle advancing operation in the case of the low frequency f
(f=f1). (A) in FIG. 24 represents the waveform of the voltage pulse
hu, (B) in FIG. 24 represents the waveforms of the voltage pulse hw
and a triangular wave, and (C) in FIG. 24 represents the waveform
of the voltage pulse hu' outputted from the comparator 206.
[0169] The phase of the voltage pulse hw shown in FIG. 24(B) leads
120.degree. relative to the phase of the voltage pulse hu shown in
FIG. 24(A). The triangular wave shown in FIG. 24(B) is a voltage
pulse applied to the non-inverting input terminal of the comparator
206 shown in FIG. 19. The comparator 206 compares the triangular
wave voltage with the reference voltage V.sub.CC/2 applied to the
inverting input terminal thereof, and outputs the voltage pulse hu'
as shown in FIG. 24(C).
[0170] The phase of the voltage pulse hu' shown in FIG. 24(C) leads
.theta.c1 relative to the phase of the voltage pulse hu shown in
FIG. 24(A). The lead angle .theta.c1 at the frequency f1 is given
by .theta.c1=-Kf1. K can be obtained by eliminating I.sub.DC1 based
on I.sub.DC1=180.multidot.C.multidot.V.sub.CC.multidot.f1/(120-Kf1)
derived from the formula (19), I.sub.DC1=kf1/(R1+R2) derived from
the formula (20), and k=V.sub.H.multidot.TW1 derived from the
formula (13). Note that although the lead angle .theta.c1 is here
determined for the constant current circuit shown in FIG. 22, the
formula for determining the lead angle becomes different one when a
circuit other than that described above is employed. Also, the
formula (20) is applied on condition that f1 satisfies
f1<Vd.times.(R1+R2)/(kR2).
[0171] With reference to FIG. 25, a description is next made of the
angle advancing operation in the case of the high frequency f
(f=f2). (A) in FIG. 25 represents the waveform of the voltage pulse
hu, (B) in FIG. 25 represents the waveforms of the voltage pulse hw
and a triangular wave, and (C) in FIG. 25 represents the waveform
of the voltage pulse hu' outputted from the comparator 206. For the
purpose of comparison with the case of FIG. 24, FIG. 25 shows an
example in which the frequency is twice that shown in FIG. 24.
[0172] The phase of the voltage pulse hw shown in FIG. 25(B) leads
120.degree. relative to the phase of the voltage pulse hu shown in
FIG. 25(A). The triangular wave shown in FIG. 25(B) is a voltage
pulse applied to the non-inverting input terminal of the comparator
206 shown in FIG. 19. The comparator 206 compares the triangular
wave voltage with the reference voltage V.sub.CC/2 applied to the
inverting input terminal thereof, and outputs the voltage pulse hu'
as shown in FIG. 25(C).
[0173] When the frequency is high, the current value I.sub.DC of
the constant current source is increased, as shown in FIG. 21, so
that the slope of the triangular wave becomes steeper and the lead
angle .theta.c is increased in comparison with that in the case of
FIG. 24. The phase of the voltage pulse hu' leads .theta.c2
relative to the phase of the voltage pulse hu. The lead angle
.theta.c2 at the frequency f2 is given by .theta.c2=-Kf2. K can be
obtained by eliminating I.sub.DC2 based on
I.sub.DC2=180.multidot.C.multidot.V.sub.CC.multidot.f2/(120-Kf2)
derived from the formula (19), I.sub.DC2=(kf2-Vd)/R2 derived from
the formula (21), and k=V.sub.H.multidot.TW1 derived from the
formula (13). Note that although the lead angle .theta.c2 is here
determined for the constant current circuit shown in FIG. 22, the
formula for determining the lead angle becomes different one when a
circuit other than that described above is employed. Also, the
formula (21) is applied on condition that f2 satisfies
f2.gtoreq.Vd.times.(R1+R2)/(kR2). However, the formula (20) must be
applied in the case of f2<Vd.times.(R1+R2)/(kR2).
[0174] In such a way, the relationship between the frequency and
the lead angle, shown in FIG. 20, can be realized.
[0175] With reference to FIG. 26, a description will be made of
MOSFET gate drive signals of the U, V and W phases in the power
running control when the motor control device of this embodiment is
employed.
[0176] FIG. 26 is a waveform chart of MOSFET gate drive signals of
the U, V and W phases in the power running control when the motor
control device according to another embodiment of the present
invention is employed.
[0177] Corresponding to the induced voltages of the respective
phases shown in FIG. 26(A), magnetic pole position detected
voltages hu, hv and hw of the respective phases are generated as
shown in FIG. 26(B). The phases of those voltage pulses are shifted
from one another by 120.degree.. Voltage pulses hup and hun, shown
in FIG. 26(C), serve as drive signals for the U-phase upper arm
MOSFET and lower arm MOSFET. In response to each of those voltage
pulses, the corresponding MOSFET is turned on when the voltage
pulse has a high level, and is turned off when the voltage pulse
has a low level. Also, the voltage pulse hun takes a low level when
the voltage pulse hup takes a high level, and conversely the
voltage pulse hun takes a high level when the voltage pulse hup
takes a low level. In addition, to prevent the upper arm MOSFET and
the lower arm MOSFET from turning on at the same time, those
voltage pulses are set to have a period (i.e., a dead time) during
which both the voltage pulses are turned off at the same time. The
phase of the voltage pulse hup leads .theta.c relative to the phase
of the voltage pulse hun. Similarly, voltage pulses hvp and hvn
serve as drive signals for the V-phase upper arm MOSFET and lower
arm MOSFET, respectively, and voltage pulses hwp and hwn serve as
drive signals for the W-phase upper arm MOSFET and lower arm
MOSFET, respectively.
[0178] The construction of the rectification detecting/driving
circuit 31 (31a or 31b) used in the motor control device of this
embodiment will be described below with reference to FIG. 27.
[0179] FIG. 27 is a block diagram of the rectification
detecting/driving circuit used in the motor control device
according to another embodiment of the present invention. The
following description is made of the basic constructions and
operations of the U-phase upper arm rectification detecting/driving
circuit 31a and the U-phase lower arm rectification
detecting/driving circuit 31b shown in FIG. 18, the V-and W-phase
rectification detecting/driving circuits also have similar basic
constructions and operations.
[0180] The rectification detecting/driving circuit 31a comprises a
U-phase upper-arm negative voltage detection circuit 1001a and a
U-phase upper-arm voltage amplification circuit 1002a. The
rectification detecting/driving circuit 31b comprises a U-phase
lower-arm negative voltage detection circuit 1001b and a U-phase
lower-arm voltage amplification circuit 1002b. Additionally, UVP
denotes an upper arm power source, and LVP denotes a lower arm
power source. The upper arm power source UVP outputs a
predetermined voltage with respect to the U-terminal as a
reference. The lower arm power source LVP outputs a predetermined
voltage with respect to the G-terminal as a reference. Other
symbols denote the same components as those shown in FIG. 18.
[0181] The U-phase upper-arm negative voltage detection circuit
1001a compares the magnitude of the voltage VB at the B-terminal
with the magnitude of the voltage VU at the U-terminal, and outputs
a compared result, i.e., (VB-VU). In the condition of VU.ltoreq.VB,
the U-phase upper-arm negative voltage detection circuit 1001a
outputs a positive voltage with respect to the U-terminal, and in
the condition of VU>VB, it outputs a negative voltage with
respect to the U-terminal. The U-phase upper-arm negative voltage
detection circuit 1001a can be realized, for example, by a
subtracter using an operational amplifier.
[0182] Also, the U-phase upper-arm voltage amplification circuit
1002a amplifies a negative voltage portion of an output of the
U-phase upper-arm negative voltage detection circuit 1001a to the
threshold voltage of the U-phase upper arm MOSFET. The U-phase
upper-arm voltage amplification circuit 1002a can be realized, for
example, by a half-wave rectification circuit and an amplification
circuit using an operational amplifier.
[0183] On the other hand, the U-phase lower-arm negative voltage
detection circuit 1001b compares the magnitude of the voltage VG at
the G-terminal with the magnitude of the voltage VU at the
U-terminal, and outputs a compared result, i.e., (VU-VG). In the
condition of VG.ltoreq.VU, the U-phase lower-arm negative voltage
detection circuit 1001b outputs a positive voltage with respect to
the G-terminal, and in the condition of VG>VU, it outputs a
negative voltage with respect to the G-terminal. The U-phase
lower-arm negative voltage detection circuit 1001b can be realized,
for example, by a subtracter using an operational amplifier. Also,
the U-phase lower-arm voltage amplification circuit 1002b amplifies
a negative voltage portion of an output of the U-phase lower-arm
negative voltage detection circuit 1001b to the threshold voltage
of the U-phase lower arm MOSFET. The U-phase lower-arm voltage
amplification circuit 1002b can be realized, for example, by a
half-wave rectification circuit and an amplification circuit using
an operational amplifier.
[0184] With the construction described above, it is possible to
perform the MOS rectification by detecting the rectification timing
from the U-terminal voltage VU and turning on/off the U-phase upper
arm MOSFET and the U-phase lower arm MOSFET. As a result, a
rectification loss can be reduced in comparison with that resulting
in the case of using the diode rectification.
[0185] The operation of the rectification detecting/driving circuit
31 used in the motor control device of this embodiment will be
described below with reference to FIG. 28.
[0186] FIG. 28 is a waveform chart of the rectification
detecting/driving circuit used in the motor control device
according to another embodiment of the present invention.
[0187] In FIG. 28, VB indicates the B-terminal voltage, VU
indicates the U-terminal voltage, and VG indicates the G-terminal
voltage. (A) in FIG. 28 represents the induced voltages, (B)
represents the output of the upper-arm negative voltage detection
circuit 1001a, and (C) represents the output of the upper-arm
voltage amplification circuit 1002a. (D) in FIG. 28 represents the
output of the lower-arm negative voltage detection circuit 1001b,
and (E) represents the output of the lower-arm voltage
amplification circuit 1002b.
[0188] As shown in FIG. 28(B), the upper-arm negative voltage
detection circuit 1001a outputs a voltage value of (VB-VU) with
respect to the VU as a reference. In the condition of VU.ltoreq.VB,
the circuit 1001a outputs a positive voltage with respect to the VU
as a reference, and in the condition of VU>VB, it outputs a
negative voltage with respect to the VU as a reference. Also, as
shown in FIG. 28(C), the upper-arm voltage amplification circuit
1002a amplifies a negative voltage portion level of the output of
the upper-arm negative voltage detection circuit 1001a after
half-wave rectification.
[0189] As shown in FIG. 28(D), the lower-arm negative voltage
detection circuit 1001b outputs a voltage value of (VU-VG) with
respect to the VG as a reference. In the condition of VG.ltoreq.VU,
the circuit 1001b outputs a positive voltage with respect to the VG
as a reference, and in the condition of VG>VU, it outputs a
negative voltage with respect to the VG as a reference. Also, as
shown in FIG. 28(E), the lower-arm voltage amplification circuit
1002b amplifies a negative voltage portion level of the output of
the lower-arm negative voltage detection circuit 1001b after
half-wave rectification.
[0190] With reference to FIG. 29, a description will be made of
MOSFET gate drive signals of the U, V and W phases in the
synchronous rectification control executed by the motor control
device of this embodiment.
[0191] FIG. 29 is a waveform chart of MOSFET gate drive signals of
the U, V and W phases in the synchronous rectification control
executed by the motor control device according to another
embodiment of the present invention.
[0192] A voltage pulse RUP shown at (B1) in FIG. 29 serves as a
drive signal for the U-phase upper arm MOSFET, and a voltage pulse
RUN shown at (B2) in FIG. 29 serves as a drive signal for the
U-phase lower arm MOSFET. In response to each of those voltage
pulses, the corresponding MOSFET is turned on when the voltage
pulse has a high level, and is turned off when the voltage pulse
has a low level. Also, the voltage pulse RUN takes a low level when
the voltage pulse RUP takes a high level, and conversely the
voltage pulse RUN takes a high level when the voltage pulse RUP
takes a low level. In addition, to prevent the upper arm MOSFET and
the lower arm MOSFET from turning on at the same time, those
voltage pulses are set to have a period (i.e., a dead time) during
which both the voltage pulses are turned off at the same time.
[0193] When the U-terminal voltage VU exceeds above the B-terminal
voltage VB, the RUP turns to a high level, and when the U-terminal
voltage VU exceeds below the B-terminal voltage VB, the RUP turns
to a low level. When the U-terminal voltage VU exceeds below the
G-terminal voltage VG, the RUN turns to a high level, and when the
U-terminal voltage VU exceeds above the G-terminal voltage VG, the
RUN turns to a low level. The other voltage pulses of the V and W
phases behave in a similar way. Incidentally, a voltage pulse RVP
shown at (B3) in FIG. 29 serves as a drive signal for the V-phase
upper arm MOSFET, and a voltage pulse RVN shown at (B4) serves as a
drive signal for the V-phase lower arm MOSFET. A voltage pulse RWP
shown at (B5) in FIG. 29 serves as a drive signal for the W-phase
upper arm MOSFET, and a voltage pulse RWN shown at (B6) serves as a
drive signal for the W-phase lower arm MOSFET.
[0194] With the construction described above, it is possible to
perform the MOS rectification by detecting the rectification timing
from each of the terminal voltages VU, VV and VW, and turning
on/off the upper arm MOSFET and the lower arm MOSFET for the
respective phases. As a result, a rectification loss can be reduced
in comparison with that resulting in the case of using the diode
rectification.
[0195] The preferred embodiments of the present invention have been
described above, but the present invention is not limited to the
embodiments described above. It is obvious to those skilled in the
art that the present invention can be modified in various forms
without departing the scope of the invention defined in the
attached claims.
[0196] The advantages of the present invention are summarized
below. Since the rectangular-wave driving control is performed in
the power running mode, the utilization factor of voltage can be
increased.
[0197] Since the synchronous rectification control is performed in
the electricity generation mode, the efficiency of electricity
generation can be increased.
[0198] In any of the power running mode and the electricity
generation mode, noises caused with the switching operation can be
eliminated.
[0199] The power running control can be performed without using a
microcomputer, and the efficiency of rectification can be
increased.
* * * * *