U.S. patent application number 10/822425 was filed with the patent office on 2004-11-25 for electromagnetic resonant sensor.
This patent application is currently assigned to Rosemount, Inc.. Invention is credited to Frick, Roger L..
Application Number | 20040233458 10/822425 |
Document ID | / |
Family ID | 35197637 |
Filed Date | 2004-11-25 |
United States Patent
Application |
20040233458 |
Kind Code |
A1 |
Frick, Roger L. |
November 25, 2004 |
Electromagnetic resonant sensor
Abstract
An electromagnetic resonant sensor has a dielectric sensor body
through which electromagnetic wave energy is propagated. The sensor
body has a cavity, with surfaces facing one another to define a gap
that varies as a function of a parameter to be measured. The
resonant frequency of an electromagnetic standing wave in the body
and the variable gap changes as a function of the gap
dimension.
Inventors: |
Frick, Roger L.;
(Minneapolis, MN) |
Correspondence
Address: |
KINNEY & LANGE, P.A.
THE KINNEY & LANGE BUILDING
312 SOUTH THIRD STREET
MINNEAPOLIS
MN
55415-1002
US
|
Assignee: |
Rosemount, Inc.
Eden Prairie
MN
|
Family ID: |
35197637 |
Appl. No.: |
10/822425 |
Filed: |
April 12, 2004 |
Related U.S. Patent Documents
|
|
|
|
|
|
Application
Number |
Filing Date |
Patent Number |
|
|
10822425 |
Apr 12, 2004 |
|
|
|
09996143 |
Nov 28, 2001 |
|
|
|
60253703 |
Nov 28, 2000 |
|
|
|
60253704 |
Nov 28, 2000 |
|
|
|
60253705 |
Nov 28, 2000 |
|
|
|
Current U.S.
Class: |
356/480 ; 331/65;
332/118; 372/92 |
Current CPC
Class: |
G01L 1/25 20130101; G01K
5/72 20130101; G01D 5/35312 20130101; G01L 9/0011 20130101; G01L
9/0008 20130101; G01D 5/268 20130101; G01K 11/32 20130101; G01F
1/383 20130101; G01D 5/35345 20130101; G01L 11/00 20130101 |
Class at
Publication: |
356/480 ;
372/092; 332/118; 331/065 |
International
Class: |
G01N 021/00; H03C
007/00; G05D 027/02; G01D 005/243; G01D 005/00 |
Claims
1. For use with a source of electromagnetic energy, a sensor for
measuring a measurable parameter, the sensor comprising: an
electromagnetic resonator, disposed to receive at least a portion
of the electromagnetic energy, the electromagnetic resonator having
a dielectric body with a sensing surface responsive to changes in
the measurable parameter at the sensing surface and the
electromagnetic resonator defining a cavity forming a variable gap
that varies in response to the sensing surface and that is
positioned such that a resonant frequency associated with an
electromagnetic standing wave in the dielectric body and the
variable gap changes in response to changes in the measurable
parameter.
2. The sensor of claim 1, wherein the resonator comprises a
dielectric resonator.
3. The sensor of claim 1, wherein the resonator comprises a
resonant antenna.
4. The sensor of claim 1, wherein the resonator comprises a
resonant transmission line.
5. The sensor of claim 1, wherein the measurable parameter is
selected from the group consisting of pressure, temperature, flow
rate, material composition, force, and strain.
6. The sensor of claim 1, further comprising a measuring apparatus
for measuring a repetition rate of the energy.
7. The sensor of claim 1, wherein the resonator is external to the
source.
8. The sensor of claim 1, wherein the resonator is internal to the
source, forming a cavity of a mode-locked source.
9. For use with a source of electromagnetic energy, a sensor for
use in measuring a measurable parameter, the sensor comprising: a
resonator having a dielectric body with a variable cavity gap
responsive to changes in the measurable parameter at a sensing
surface, the resonator defining a resonant frequency of a standing
electromagnetic wave in the dielectric body and the variable cavity
gap that is dependent upon the measurable parameter at the sensing
surface, the resonator being disposed such that a signal from the
sensor is a function of the resonant frequency.
10. The sensor apparatus of claim 9 wherein the resonator is
internal to the source and forms a cavity of the source.
11. The sensor apparatus of claim 9, wherein the resonator forms a
resonator that is external to the source.
12. The sensor of claim 9, wherein the resonator comprises a
dielectric resonator.
13. The sensor of claim 9, wherein the resonator comprises a
resonant antenna.
14. The sensor of claim 9, wherein the resonator comprises a
resonant transmission line.
15. The sensor of claim 9, wherein the measurable parameter is
selected from the group consisting of pressure, temperature, flow
rate, material composition, force, and strain.
16. The sensor apparatus of claim 9, further comprising a measuring
apparatus for measuring the frequency of the signal.
17. An apparatus for modulating, based on a measurable parameter,
the output of a source producing electromagnetic energy, the
apparatus comprising: a coupler coupled to receive the energy; and
a high Q resonator having a dielectric body with a variable
configured to produce an effective dielectric constant that varies
in response to changes in the measurable parameter, the high Q
resonator coupled to the coupler for receiving the energy and
creating an electromagnetic standing wave within the dielectric
body and the variable cavity gap at a resonant frequency that is a
function of the measurable parameter.
18. The apparatus of claim 35, wherein the measurable parameter is
selected from the group consisting of pressure, temperature, flow
rate, material composition, force, and strain.
19. The apparatus of claim 17, wherein the source has a resonator
characterized by a first Q value Q1, and the high Q resonator is
characterized by a second Q value Q2, that is substantially higher
then Q1.
20. The apparatus of claim 19, wherein Q2 is at least 100.
21. A variable frequency resonator comprising an electromagnetic
resonator having a dielectric body and a cavity defining a variable
gap, the resonator producing an output at a resonant frequency that
is dependent upon the variable gap which is disposed to alter a
ratio of stored electric field and magnetic field energy of an
electromagnetic standing wave in response to changes in the
measurable parameter.
22. A method of sensing a measurable parameter, the method
comprising: providing a resonator characterized by a resonant
frequency that is a function of a variable gap in an internal
cavity of a dielectric body of the resonator, the variable gap
being responsive to the measurable parameter; supplying
electromagnetic energy to the resonator to produce an
electromagnetic standing wave in the dielectric body and the
variable gap; and sensing a resonant frequency of the
electromagnetic standing wave to determine the measurable
parameter.
23. A method of sensing a measurable parameter, the method
comprising the steps of: providing a pulsed electromagnetic signal
characterized by a repetition rate; providing a resonator having a
dielectric body with a variable gap that varies in response to
changes in the measurable parameter; supplying the pulsed
electromagnetic signal to the resonator to produce a pulsed
electromagnetic wave pattern in the dielectric body and the
variable gap; and sensing variations in the repetition rate of the
pulsed electromagnetic signal in response to variations in the
variable gap.
24. For use with a electromagnetic source, a resonator having a
dielectric body with a variable gap that varies in response to
changes in a measurable parameter, the resonator receiving
electromagnetic energy from the source to produce an
electromagnetic standing wave in the dielectric body and the
variable gap so that a characteristic of the energy changes in
response to variations in the variable gap.
25. The resonator of claim 30, wherein the electromagnetic energy
is a continuous wave and the characteristic is frequency.
26. The resonator of claim 30, wherein the electromagnetic energy
is a pulsed energy and the characteristic is repetition rate.
27. An electromagnetic resonant sensor comprising: a sensor body;
and a cavity within the sensor body having a variable gap between
interior surfaces of the sensor body that varies as a function of a
measurable parameter, the cavity being positioned within the sensor
body so that an electromagnetic standing wave is formed within the
body and the variable gap, and a resonant frequency of the sensor
is a function of the measurable parameter.
28. The electromagnetic resonant sensor of claim 27 wherein the
sensor body is a dielectric material.
29. The electromagnetic resonant sensor of claim 28 and further
comprising: a conductor on one of the interior surfaces.
30. The electromagnetic resonant sensor of claim 29 wherein the
conductor is configured to cause the sensor to resonate as a ring
resonator.
31. The electromagnetic resonant sensor of claim 29 wherein the
conductor is configured to cause the sensor to resonate as a
transmission line.
32. The electromagnetic resonant sensor of claim 29 wherein the
conductor is configured to cause the sensor to resonate as a slot
antenna.
33. The electromagnetic resonant sensor of claim 29 wherein the
conductor is configured to cause the sensor to resonate as a dipole
antenna.
34. The electromagnetic resonant sensor of claim 29 wherein the
conductor is configured to cause the sensor to resonate as a port
antenna.
35. The electromagnetic resonant sensor of claim 28 wherein the
sensor body and cavity are configured to resonate at suboptical
frequencies.
36. An electromagnetic resonant sensor for receiving
electromagnetic energy and producing an output based upon an
electromagnetic standing wave having a resonant frequency that is a
function of a parameter to be measured, the sensor characterized by
a dielectric body with a variable gap that changes dimension as a
function of the parameter, the dielectric body and the variable gap
being configured so that the electromagnetic standing wave extends
within the dielectric body and the variable gap and a change in gap
dimension causes a change in the resonant frequency.
Description
CROSS-REFERENCE TO RELATED APPLICATION(S)
[0001] This application is a continuation-in-part of application
Ser. No.09/996,143, filed Nov. 28, 2001 that claims priority from
U.S. Provisional Applications 60/253,703, 60,253,704 and
60,253,705, all filed Nov. 28, 2000.
BACKGROUND OF THE INVENTION
[0002] The present invention relates generally to sensors. More
specifically, the invention relates to electromagnetic resonant
sensors used to measure parameters commonly measured in industrial
process or flow systems--parameters such as pressure, temperature,
flow rate, strain, and material composition.
BRIEF SUMMARY OF THE INVENTION
[0003] The invention is a electromagnetic resonant sensor having a
body with a cavity gap that changes in dimension as a function of a
parameter to be measured. Electromagnetic energy is supplied to the
sensor, to produce an electromagnetic standing wave within the
sensor body and the cavity gap. The resonant frequency of the
electromagnetic standing wave is a function of the cavity gap. An
output of the sensor is produced based upon the resonant frequency,
so that the output varies as a function of the parameter to be
measured.
BRIEF DESCRIPTION OF THE DRAWINGS
[0004] FIG. 1 is a block illustration of an external high Q
resonator coupled to a laser in accordance with an embodiment.
[0005] FIG. 2 is a block illustration of a laser with an internal
resonator according to an embodiment.
[0006] FIG. 3A is a block illustration of a mode-locked laser
having an optical gain medium in accordance with an embodiment.
[0007] FIG. 3B is a block illustration of a mode-locked laser
having an optical gain medium in accordance with an embodiment.
[0008] FIG. 4 is a cross-sectional view of an optical fiber in
accordance with an embodiment.
[0009] FIG. 5 is a cross-sectional view of the optical fiber of
FIG. 4 after a measurable parameter has changed at a sensing
surface of the fiber.
[0010] FIG. 6 is an illustration of the optical fiber of FIG. 4
used in an optical sensor in accordance with an embodiment.
[0011] FIG. 7 is an illustration of a ring resonator in accordance
with an embodiment.
[0012] FIG. 8 is an illustration of a ring resonator in accordance
with another embodiment.
[0013] FIG. 9 is a cross-sectional profile of an optical fiber in
accordance with an embodiment alternative to that of FIGS. 4 and
5.
[0014] FIG. 10 is an illustration of a microsphere resonator in
accordance with an alternative embodiment.
[0015] FIG. 11 is an illustration of the microsphere resonator of
FIG. 10 in an exemplary optical sensor.
[0016] FIG. 12 is an illustration of the microsphere resonator of
FIG. 10 in another exemplary optical sensor.
[0017] FIG. 13 is a cross-sectional view of an alternative optical
sensor having a microsphere.
[0018] FIG. 14 is a perspective view of an unassembled integrated
optical sensor with first module and second module in accordance
with an embodiment.
[0019] FIG. 15 is a cross-sectional view of the integrated optical
sensor of FIG. 14 assembled.
[0020] FIG. 16 is a cross-sectional view of the an alternative
integrated optical sensor to that shown in FIG. 15.
[0021] FIG. 17 is a top view of an alternative to the first module
of FIG. 14 using a photonic crystal array to form a resonator.
[0022] FIG. 18 is a cross-sectional view of a microdisc resonator
and VCSEL in accordance with an embodiment.
[0023] FIG. 19 is a block diagram of electromagnetic high Q
dielectric resonant sensor driven by a microwave oscillator.
[0024] FIG. 20 is a block diagram of a sensor system having a
microwave oscillator and a high Q dielectric resonant sensor in a
self-resonant configuration.
[0025] FIG. 21 is a block diagram of a sensor system having a maser
in conjunction with a high Q dielectric resonant sensor.
[0026] FIGS. 22A and 22B show an embodiment of electromagnetic ring
resonator sensor for use with microwave energy in accordance with
the present invention.
[0027] FIGS. 23A and 23B show an embodiment of an electromagnetic
ring resonator sensor with a cavity of two different gap
thicknesses.
[0028] FIGS. 24A and 24B show an embodiment of an electromagnetic
ring resonator sensor having square faces and a circular cavity
operating at a lowest resonant mode.
[0029] FIGS. 25A and 25B show the ring resonator sensor of FIGS.
24A and 24B operated at a higher resonant mode.
[0030] FIGS. 26A and 26B show a ring resonator having a conductor
on one surface of the cavity.
[0031] FIGS. 27A and 27B show sectional views of an embodiment of a
resonant transmission line sensor having conductors on opposing
surfaces of a cavity.
[0032] FIGS. 28A and 28B show sectional views of an embodiment of a
resonant transmission line microwave sensor having a slot antenna
in a conductor on one surface on the cavity.
[0033] FIGS. 29A and 29B are sectional views of an embodiment of a
coaxial resonant transmission line sensor.
[0034] FIGS. 30A and 30B, 31A and 31B, 32A and 32B, 33A and 33B,
34A and 34B, and 35A and 35B are sectional views of additional
embodiments of slot antenna resonant transmission line sensors.
[0035] FIGS. 36A and 36B, and 37A and 37B are sectional views of
embodiments of circular dipole resonant transmission line
sensors.
[0036] FIGS. 38A and 38B are sectional views of a high Q resonator
sensor having a rectangular cavity with a rectangular conductor and
a circular port antenna.
[0037] FIG. 39 is a block diagram of an embodiment using an array
of mode-locked resonant sensors.
DETAILED DESCRIPTION
[0038] FIGS. 1-18 show embodiments of the high Q resonant sensors
of the invention operating with electromagnetic radiation generally
in optical ranges from long-infrared to ultraviolet wavelengths.
FIGS. 19-38B show embodiments of the high Q resonant sensors of the
invention operating with electromagnetic radiation in the microwave
portion of the spectrum (i.e. up to long-infrared wavelengths).
[0039] Generally, FIGS. 1-18 include lasers, preferably operating
in a pulsed output mode, that produce a laser signal of a
repetition rate or frequency modulated by a measurable parameter.
By using a mode locked laser, conventional high-speed electronics
can be used to measure the modulated signal, and the repetition
rate or frequency of the laser signal can be measured with high
resolution. By measuring the frequency of the laser signal, a value
for the measurable parameter can be derived.
[0040] In some embodiments, a high Q optical resonator produces the
laser signal with a frequency dependent upon the measurable
parameter. The high Q resonator can be internal or external to the
laser. The devices have lower power consumption and improved
accuracy over the state of the art. The devices may be used to
measure changes in a measurable parameter or they may be used to
make absolute measurements. Further applications and further
embodiments will be apparent to persons of ordinary skill in the
art. For example, the devices described may be adapted for use as
tunable optical filters, tunable laser sources, and other all
optical applications.
[0041] FIGS. 1 and 2 show exemplary embodiments. FIG. 1 shows a
laser 100 coupled to an external high Q resonator 102 through a
coupler 104, generally shown. The laser 100 may be a fiber-doped
laser, a ruby laser, or a diode laser. Other laser sources are
contemplated. The laser 100 may be an amplification stage, such as
an optical parametric amplifier or fiber amplifier stage pumped by
a laser source. The laser source 100 may also be replaced with a
light emitting diode (LED). By way of example only, the laser
source 100 is shown receiving power from a power source 103, which
as it would be understood by persons of ordinary skill in the art
may take the form of an optical or electrical power source. The
laser energy is preferably at a wavelength in the visible or
infrared region, though the laser energy may be within the
far-infrared and microwave regions as well. Embodiments using
microwave energy will be further discussed in conjunction with
FIGS. 19-38B.
[0042] In the preferred embodiment, the coupler 104 is an optical
fiber or optical waveguide, and coupling is achieved through
low-loss evanescent coupling. Coupling may be achieved through
partially transmissive mirrors, waveguide taps, or other known
means for coupling optical signals.
[0043] The laser source 100 provides a laser energy to the high Q
resonator 102 through the coupler 104. The laser energy coupled
from the laser 100 to the resonator 102 is at a wavelength
corresponding to the resonance of the laser cavity within the laser
100. Such laser cavities, however, have low Q and produce an output
of relatively large bandwidth. The Q of the external resonator 102
is preferably substantially higher than the Q of the laser cavity
within the laser 100. For example, in the preferred embodiment, the
Q of the resonator 102 would be at least 100. Typical resonators
only have Q values between 3 and 100 and are limited by the mirrors
forming the resonant cavity and the desired power output.
[0044] It is generally known that a low Q oscillator system will
lock onto the frequency of a high Q resonator if there is
sufficient coupling between the oscillator and the resonator and if
the frequency ranges of the two regions overlap. The low Q laser
cavity of laser 100 locks onto the resonance frequency of a high Q
resonator 102. That is, the exchange of energy between the high Q
resonator 102 and the lower Q laser 100 will lock the laser signal
of the entire system to a frequency and bandwidth defined by the
resonator 102. The result is that the system produces a laser
signal of a narrow bandwidth and centered at a resonance frequency
of the resonator 102.
[0045] The laser source 100 may be a continuous wave (CW) source or
a preferably a pulse mode locked laser source. If the source 100 is
a CW source, then the laser signal from the system locks onto the
resonant frequency of the resonator 102 and has a narrowed
bandwidth induced by the high Q of the resonator 102. Here, a
spectrometer would be used to measure the frequency of the laser
signal. If the source 100 is a pulse mode locked laser source, the
resonator 102 additionally determines the repetition rate of the
pulse train. Here, conventional electronic detectors can be used to
measure the sub 100 GHz repetition rate.
[0046] Optical resonators have multiple resonant frequencies. It is
desirable, however, that only a single resonant frequency be
located within the bandwidth of the laser energy supplied by the
laser sources. That is, the laser signal exists at a single
consistent resonant frequency. This condition will reduce the mode
hopping that occurs in some state of the art laser systems. A
distributed feedback laser (DFB) laser may be used as the laser
source 100 to achieve a bandwidth that allows a single resonant
frequency. The physical parameters of the resonator 102 could be
altered to achieve single resonance, as well.
[0047] The resonator 102 is formed of an optically transparent
material. The material may be a lasing material or a non-lasing
material. Suitable materials include sapphire, quartz, doped
quartz, polysilicon, and silicon. These materials exhibit low
optical losses. These materials also exhibit good mechanical
properties that respond precisely and consistently to changes in
the measurable parameters and do not permanently deform under
pressure or force, but instead return to their original shapes
after the measurable parameter has returned to a steady state
value. Preferably, materials that allow propagation under total
internal reflection are used. The total internal reflection and low
optical losses allow for very high Q resonators.
[0048] The resonator 102 is characterized by having resonant
frequencies that depend upon measurable parameters near the
resonator 102. Herein, "measurable parameters" means those
parameters associated with an external force or pressure. Pressure
(absolute and differential), temperature, flow rate, material
composition, force, and strain are examples. Laser source 100 and
high Q resonator 102 collectively form an optical sensor 105, which
produces a laser signal or sensed signal, dependent upon a
measurable parameter.
[0049] The laser signal is supplied to a measuring apparatus 106
through structure generally shown as couplers 108A and 108B and
isolator 109. If the laser source 100 is a pulse mode locked laser,
the measuring apparatus 106 could be a conventional high-speed
electronics detector. If the laser source 100 is a continuous wave
source, the measuring apparatus 106 is preferably a spectrometer or
other suitable device for measuring signal frequency.
[0050] The isolator 109 prevents back reflected signals of the
measuring apparatus 106 from entering into the resonator 102. As
the laser signal is dependent upon a measurable parameter, the
measuring apparatus 106 may additionally derive a value for the
measurable parameter by measuring the frequency or repetition rate
of the laser signal and calculating a measurable parameter value
corresponding to that measurement. This derivation is performed in
known ways. The coupling between the resonator 102 and the
measuring apparatus 106 may be achieved through fiber coupling,
mirror coupling, taps, evanescent coupling or other suitable
coupling types.
[0051] The resonator 102 has a high Q value and, therefore, the
energy coupled between the laser 100 and the resonator 102 may be
very low and proper locking onto a resonant frequency of the
resonator 102 will occur. Another advantage of using the high Q
external resonator 102 is that the signal/noise (S/N) ratio of the
system improves. Generally, the S/N ratio and resolution of a
frequency modulated laser system is limited by the frequency jitter
in the lasing mechanism. This jitter has many sources; mode
hopping, power supply noise, thermal noise, quantum fluctuations
and gain noise in the lasing media are a few. Amplitude noise
fluctuations modulate the lasing frequency itself so that the
resulting frequency noise cannot be filtered out once it has been
generated.
[0052] By way of example, the S/N ratio is proportional to the GF,
defined above, under the following proportionality:
S/N=GF-f.sub.rf.sub.n. The value f.sub.n is the noise frequency
dither inherent in f.sub.r. A high GF results in a high S/N ratio.
Coupling a laser output into an external high Q resonator, like
resonator 102, means that laser output frequency will be determined
by the resonator and, therefore f.sub.n will be low and the S/N
ratio characteristics will be determined primarily by the
characteristics in the external resonator. The resonators shown
herein are also characterized by high GF and therefore high S/N
ratio.
[0053] The external resonator structure of FIG. 1 is useful to make
measurements in environments hostile to laser operation, because
the sensing mechanism (i.e., the resonator 102) is remote to the
laser source 100. Also, in this embodiment, the external resonator
102 is not susceptible to the high temperatures produced by the
laser source 100.
[0054] FIG. 2 is block depiction of an alternative embodiment of an
optical sensor 130, in which a laser 132 is formed of an internal
high Q resonator. Here, the high Q resonator forms the laser 132
and, therefore, functions as the laser gain cavity. The high Q
resonator is formed of a material that lases upon being pumped by
an appropriate feed energy. By way of example only, the laser
source 132 is shown receiving power from a power source 133, which
as it would be understood by persons of ordinary skill in the art
may take the form of an optical or electrical power source.
Semiconductor materials, doped sapphire, doped quartz, and other
materials may also be used to form the internal resonator. Doped
quartz is particularly beneficial because when quartz is doped with
the rare earth element erbium the quartz can be made to emit laser
light of 1550 nm, i.e., the preferred low-loss wavelength for
current optical fiber communications. The laser signal is produced
by the laser 132 and provided to the measuring apparatus 106
through couplers 134A and 134B and an isolator 136, similar to that
described above.
[0055] Provided in a preferred embodiment is a frequency modulated
laser source that produces a pulse train as the laser signal. The
repetition rate of the pulses varies as a function of a measurable
parameter acting on a resonator, and, therefore, the entire
structure forms a high resolution and high accuracy optical sensor.
For example, simply counting of a 1 GHz change (induced by a change
in a measurable parameter) in a 100 GHz signal would give a
resolution of 1 ppb over a 1 second measurement. State of the art
lasers have noise bandwidths of a few KHz with a base frequency of
about 200,000 GHz, indicating that a resolution of more than 30
bits is attainable with such an optical sensor.
[0056] There are various ways known in the art to set-up a
mode-locked laser such as using either passive mode-locking or
active mode-locking. Ti:Sapphire mode-locked lasers pumped with an
Argon continuous wave laser source or pumped with a Nd:Yag laser
continuous wave source have been shown. Others have shown passive
mode-locking using semiconductor lasers and micro-mechanical
tuning. Any mode-locked laser would be suitable as the laser source
100 of FIG. 1 and various mode-lock laser set-ups may be used in
the embodiment of FIG. 2. Thus, pulse mode locked operation is
contemplated in both-the external and internal resonator
embodiments.
[0057] An exemplary mode-locked laser is shown in FIG. 3A, showing
a laser 140 coupled to an external High Q resonator 142 for
producing a mode-locked laser signal 144 at an operating frequency
related to a measurable parameter acting on the resonator 142. The
laser 140 includes a mode-locking mechanism, which can take a
variety of forms as known in the art. For instance, a saturable
amplifier can be introduced into the lasing cavity such that only
short pulses are able to pass without attenuation. The laser signal
144 is a pulsed laser signal having a repetition rate dependent
upon the measurable parameter acting on a sensing (e.g., outer)
surface of the high Q resonator 142. In particular, pulsed laser
signals 144 of the laser 140 locked to the external high Q
resonator 142 have a repetition rate determined by F=F.sub.in
(1-h/nL) where h is the round trip length of the external
resonator, L is the round trip length of the mode locked laser, n
is the effective refractive index, and F.sub.in is the round trip
frequency of the mode locked laser. (See "Passively Mode-Locked
Micromechanically-Tunable Semiconductor Lasers", Katagirie et al.,
IEICE Trans. Electron., Vol. E81-C. No. 2, Feb. 1998). A measuring
apparatus 146 measures this repetition rate and uses that
measurement to derive a value for the measurable parameter. The
laser 140 is exemplarily shown being pumped by a power source 148,
which represents any of the known sources of pump energy for a
mode-locked laser.
[0058] Referring to FIG. 3B, a mode locked laser 150 may
alternatively incorporate the high Q resonator internal to the
lasing cavity (similar to FIG. 2) to produce the mode locked laser
signal 152. With this internal resonator structure, the repetition
rate of the laser signal 152 is simply the round trip time of the
resonator. As noted above, the mode locking mechanism of the laser
150 can be accomplished through a variety of techniques known in
the art such as introducing a saturable amplifier section into the
loop or using active mode locking. The measuring apparatus 146 then
measures the repetition rate of the laser signal 152 to derive a
value for the measurable parameter acting on a sensing surface of
the laser/resonator. Though not shown, the embodiments of FIGS. 3A
and 3B may be implemented with isolators and other optical
components (such as varied optical couplers) as desired.
[0059] Some exemplary resonator structures characterized by a
resonant frequency dependent upon a measurable parameter will now
be discussed.
[0060] FIGS. 4 and 5 depict a cross section of an optical fiber 160
that may be used to form the resonator 102 or 132. The optical
fiber 160 may be formed of any of the standard materials used in
optical fibers and is preferably a single mode fiber. The optical
fiber 160 is characterized by a cladding region 162 and a higher
index of refraction core region 164. This configuration confines
propagation primarily to the core 164 and a signal propagating
within the core 164 propagates under total internal reflection.
[0061] The optical fiber 160 contains a cavity 166 defining a
variable gap, which may be evacuated or contain a gas or other
suitable material. The cavity 166 is formed in the fiber 160
through known processing methods, such as etching or drawing down a
blank or preform that includes the cavity. In FIG. 4, the cavity
166 is entirely disposed within the core 164.
[0062] The cavity 166 may also be partially within the core 164 or
entirely external to the core 164, as shown in the embodiment of
FIG. 9. In the preferred embodiment, the cavity 166 is similarly
shaped in cross-section to that of the core 164. Also, in a
preferred embodiment the cavity 166 would be symmetric to the core
164. While both the cavity 166 and the core 164 are shown with a
rectilinear cross-section, it would be understood that other
cross-sectional profiles may be used. For example, other shapes for
the cavity could be used such as a multiplicity of closely spaced
round holes which change size in response to a sensed variable or
measurable parameter. The cavity 166 extends longitudinally with
the core 164 along at least a portion of the fiber 160.
[0063] The gap of the cavity 166 varies in response to changes in
measurable parameters, for example, changes in pressure or force
external to the fiber 160. In operation, an increase in the
pressure on the outside of the fiber 160 applies force to an outer
wall or sensing surface 167 of the fiber 160, which results in
radial forces being applied through the cladding region 162 and on
the cavity 166. Due to the geometry of the cavity 166, some of the
radial forces will not alter the cavity shape. Other forces,
principally forces represented by arrows 168 (see, FIG. 2), will
act to compress the cavity 166. Therefore, an increase in pressure
at the sensing surface 167 of the fiber 160 will result in the
compression, i.e., inward displacement, of the cavity 166. Though,
not shown it would be understood that a decrease in pressure would
result in an expansion of the cavity 166.
[0064] Changes in the other measurable parameters would alter the
cavity 166, as well. For example, the fiber 160 may be placed
within a processing flow system such that changes in flow rate,
temperature, or material composition alter the geometry of the
cavity 166. Changes in any of the measurable parameters would
result in changes in the variable gap of the cavity 166. The cavity
166, therefore, provides an alterable perturbation within the
propagating core 164, a perturbation that alters in response to a
measurable parameter.
[0065] It is preferred that the cavity 166 have a cross-sectional
shape that can be compressed and decompressed in response to
relatively small changes in the measurable parameter. It is also
preferred that the cavity displacements be relatively small, i.e.,
in the micron and sub-micron range so that the propagation
characteristics within the core 164 change a detectable amount, but
not an amount that will detrimentally affect the mode profile of a
wave propagating within the core, 164. In the examples of FIGS. 4
and 5, an elongated rectangular profile is used for the cavity 166.
The cavity 166 has a first side 170 longer than a second side 172.
The cavity 166 has dimensions on the order of the wavelength(s) to
propagate within the fiber 160. The steady state cavity profile
(e.g., at atmospheric pressure) can be changed depending on the
desired sensitivity and the parameter to be measured.
[0066] Changes to the shape of the cavity 166 will alter the
propagation characteristics within the core 164. In particular, a
wave traveling within the core 164 experiences a particular index
of refraction within the core 164. A fiber core is typically
characterized by a material dependent index of refraction. A wave
propagating within the core 164 experiences an effective index of
refraction that is dependent upon the various materials that the
wave propagates through. A propagating wave has an electric field
that passes primarily through the core 164 and cavity 166 but also
extends substantially into the cladding 162. The energy stored in
the electric field is thus dependent on the refractive indices and
geometries of the three regions, and the energy stored in the
electric field determines the velocity of propagation of the
electromagnetic wave along the length of the fiber 160. The
propagating wave thus behaves as though it is traveling through a
uniform material with an effective index of refraction that is a
geometry weighted average of the indices of refraction of the three
regions. The effective index of refraction experienced by a
propagating wave changes with changes to the geometry, i.e.,
compression or expansion of the cavity 166. Described now are some
exemplary applications in which the change to the effective index
of refraction of a wave propagating in a core can be used in an
optical sensor.
[0067] FIG. 6 shows the fiber 160 of FIGS. 4 and 5 in a optical
sensor 176. The structure may also be considered an apparatus of
modulating laser signal frequency. The depicted configuration is
similar to that of the laser system of FIG. 1. The optical sensor
176 has a light source 178 supplying an output to the fiber 160
through an isolator 182 and couplers 180A and 180B. Preferably, a
semiconductor laser or LED source is used as the light source 178.
The source 178 could be a continuous-wave laser or a pulse mode
locked laser, though in the latter case the optical medium forming
the fiber 160 is not a lasing medium. The fiber 160 is doped to
form a lasing material, and the output from the source 178 is a
pump energy of a wavelength sufficient to allow lasing action in
the fiber 160.
[0068] The fiber 160 has a middle portion 184 over which changes to
a measurable parameter are measured. A first end of the fiber 160
has a first reflector in the form of Bragg reflector 188 and a
second end has a second reflector in the form of Bragg reflector
190. The middle portion 184 extends between the Bragg reflectors
188 and 190 and coincides with the sensing surface 167. The Bragg
reflectors 188 and 190 define a resonator 192 within the optical
fiber 160. In the depicted environment, the resonator 192 extends
along the length of the optical fiber coinciding with the middle
portion 184 and extending slightly into the Bragg reflectors 188,
190. In the preferred embodiment, the cavity 166 (not shown) does
not extend into the Bragg reflectors 188, 190. However, the cavity
166 may extend into the Bragg reflectors 188, 190 if desired.
Though shown as Bragg reflectors 188 and 190, the first and second
reflectors could alternatively be mirrors or other highly
reflective structures formed on the fiber 160 or external
thereto.
[0069] In operation, the pump energy produced by light source 178
is supplied to the resonator 192 through the partially transmitting
Bragg reflector 188. A laser signal emits from the Bragg reflector
190 along couplers 194A and 194B and through isolator 186. The
laser signal has a wavelength corresponding to a resonance
frequency of the resonator 192. The frequency of the signal on
coupler 194B is measured by a measuring apparatus 196.
[0070] Upon a change to a measurable parameter at the sensing
surface 167, specifically over the middle portion 184, the cavity
166 will be altered and thereby alter the effective index of
refraction experienced by a signal propagating within the core 164.
The effective index of refraction will determine the velocity of
propagation of the light wave in the resonator 192. This in turn
will determine the resonant frequency of the resonator 192 and,
therefore, the frequency of the laser signal on coupler 194A in CW
operation. In mode-locked operation, the repetition rate is
altered. Changes to a measurable parameter will be detected by the
detector 196 in the form of changes in the frequency of the laser
signal.
[0071] In CW operation of the sensor 176, the measurement apparatus
196 is a detector in which the laser signal frequency is compared
to the frequency of a reference laser to allow for the measurement
of very fine changes to the frequency of the laser signal. In pulse
mode operation, the measurement apparatus 196 is an electronic
detector that measures changes in the repetition rate of the laser
signal pulse train. In either case, cavity displacements of a
micron or below will result in frequency changes that can be
measured by the detector 196. Though not shown, a CPU or other
processor is used to compute a value for the measurable parameter
based on the detected laser signal frequency. Changes in measurable
parameters are detectable as well as absolute measurements. It
would be understood, that an initial normalization may be used to
calibrate the detector 196 and/or processor for accurate
measurements the measurable parameter. For example, a normalization
may be performed before a different measurable parameter is to be
sensed. It would be further appreciated that multiple sensors can
be used with a processor to make varying sorts of other
measurements, like measuring AP between two separate locations
within a flow system. With a typical gage factor of 0.01 to 0.1 and
a Q of 160 or more measurements with 0.01% to 0.001% resolution may
be made with the sensor 176.
[0072] Alternative resonators are contemplated. An example of an
alternative resonator is shown in FIG. 7. Here, a waveguide 200
forms a circular resonator also known as a circulator or ring
resonator and will be termed as such henceforth. The ring resonator
200 may be formed by joining ends of a optical fiber using
commercially available fusing techniques in the preferred
embodiment. The ring resonator 200 has a cladding, core region, and
cavity like those of the optical fiber 160 when viewed in
cross-section. The ring resonator 200 is characterized by high Q
and high gage factor and forms part of an optical sensor 202. If
the cavity extends the entire length of the ring resonator 200,
then the entire outer surface of the ring resonator 200 would act
as a sensing surface.
[0073] Coupling of a signal into the closed loop of the ring
resonator 200 is achieved through evanescent coupling. A primary
waveguide 204 is brought within evanescent coupling contact of the
ring resonator 200 over a region generally shown by reference
numeral 208. The waveguide 204 is an optically transparent
waveguide formed, for example, of a polysilicon material. Sapphire
and quartz would also be useful for creating total internal
reflection propagation and the waveguide could be another optical
fiber. A laser signal 206 from laser source 205 is made to
propagate through the waveguide 204.
[0074] The signal 206 locks onto the resonant frequency within the
resonator 200 and has a narrow bandwidth induced by the coupling
into the resonator 200. The signal 206 therefore is dependent upon
the properties with the resonator 200, i.e., it is dependent upon a
measurable parameter at an outer surface of the resonator 200.
Signal 206 is provided to a measurement apparatus 209, such as
those described above. It is noted that in one embodiment the ring
resonator 200 may be formed from a lasing material such that the
resonator constitutes the laser cavity such as shown in the
internal resonator embodiment of FIG. 2.
[0075] An alternative embodiment is shown in FIG. 8, in which a
signal 210 propagating with the resonator 200 is coupled as signal
212 to a secondary or output waveguide 214 that is within coupling
contact with the ring resonator 200 over a region generally shown
as 216. This embodiment is particularly useful where the resonator
200 is formed of a non-lasing material. The output waveguide 214 is
outside of coupling contact with the waveguide 204. To direct the
signal 212, the waveguide 214 has a snubbed end 220 and extends
distally therefrom, so that the signal 212 propagates in a parallel
direction to that of the signal 206. The signal 212 is coupled to
the measuring device 209. The ring resonator 200 and waveguides 214
and 204 are preferably created during the same overall process to
reduce device cost and fabrication times.
[0076] The embodiment of FIG. 8 may be particularly useful in
filtering out a wavelength from an incoming signal. For example,
with signal 206 as a broad bandwidth LED energy or a white light
energy, the resonator 200 would remove that part of the propagating
energy coinciding with the resonance frequency and bandwidth of the
resonator 200. The signal 212 would be at the removed frequency.
With resonator 200, near complete removal of the resonant frequency
from the signal 206 is achievable.
[0077] FIG. 9 shows a suitable alternative embodiment to FIGS. 4
and 5 of a fiber 300 having a cladding region 302, a core 304 and a
cavity 306. The outer surface 308 is the sensing surface of the
fiber 300. The cavity 306 defines a variable gap similar to that of
the cavity 166 in that the cross sectional profile of the gap
changes in response to changes in measurable parameters. Here,
however, the cavity 306 is disposed entirely within the cladding
region 302. The cavity 306 is nonetheless close enough to the core
304 to change the effective index experienced by a signal
propagating therein. As with the above embodiments, changes to the
cavity 306 induced by changes in measurable parameters would alter
the effective index. Thus, the fiber 300 can be used in a resonator
or laser cavity to produce an output signal that is dependent upon
measurable parameters. The fiber 300 is suitable for use in the
Bragg reflector resonator or the ring resonator configurations
described above, as well as other resonator configurations. As with
the fiber 100, the core 304 would be formed of higher index
optically transparent material, preferably transparent in the
infrared region. The core 304 and the cavity 306 can have different
cross sectional profiles and still achieve the desired dependency
of the resonant frequency upon changes in the measurable
parameters.
[0078] Another type of resonator encompassed within the present
teachings is a microsphere resonator such as resonator 400 shown in
FIGS. 10-12. Optical microspheres are known to have exceedingly
high Q values, exceeding 1,000,000,000. Microspheres, therefore,
provide an ideal resonator for measuring very small changes in a
measurable parameter. Known microspheres, however, are formed of
unitary structures without gaps or spacings that can be made to
vary.
[0079] The microsphere 400 is hollow and operates in a whispering
gallery mode where light travels along the outer surface of the
microsphere 400 like known microspheres. Light is confined by total
internal reflection at the surface of the sphere. The microsphere
400 is separated into a first hemisphere 402 and a second identical
hemisphere 404, and the two hemispheres 402, 404 are separated by a
variable gap 406. The gap 406 is small enough such that a signal
propagating within either of the hemispheres 402, 404 will be able
to couple into the other for propagation therein.
[0080] The microsphere 400 is characterized by a resonant frequency
defined by the hemispheres 402 and 404. The spacing of the gap 406
affects the resonant frequency in a similar manner to that of
cavity 166 on fiber 160. Referring to FIG. 11, a portion of a laser
signal 408 propagating in waveguide 410 is coupled into hemisphere
402. The laser signal 408 will lock onto the resonant frequency
within the high Q resonator 400. In operation, as a measurable
parameter changes at sensing surface 413A and/or 413B, the variable
gap 406 will vary the spacing between the hemisphere 402 and the
hemisphere 404 and thereby alter the resonant frequency of the
microsphere 400. The resulting variation in the variable gap 406
alters the output frequency of the laser signal 408. The signal 408
is coupled to a measuring apparatus, not shown.
[0081] FIG. 12 shows an alternative embodiment, in which the
microsphere 400 is disposed between two waveguides 410 and 412, and
the microsphere 400 functions as a resonant frequency filter or
sensor 414 similar to the structure shown in FIG. 8, producing a
filter laser signal 416 in waveguide 412.
[0082] The microsphere 400 and waveguides 408 and 410 may be formed
over a substrate and mounted using mounting means suitable for
ordinary microspheres. The hemispheres 402 and 404 are preferably
mounted for movement relative to one another. By way of example,
MEMS mounting structures may be used for this purpose. A MEMS
fabrication process could be used to create an actuation mounting
that biases the hemispheres 402 and 404 to a desired variable gap
spacing, but that will allow the variable gap spacing to contract
and expand in response to small changes in pressure, temperature,
etc. The microsphere 400 is preferably formed of a lasing material,
like doped quartz. Though, it may be formed of a nonlasing material
instead. Multiple microspheres may be used to increase the signal
to noise ratio of the output signal measured at the detector. Other
modifications will be apparent.
[0083] FIG. 13 shows an alternative optical sensor 450, in the form
of an optical sensor capsule, formed with a microsphere 452. In the
preferred embodiment, the laser is doped to form a microlaser which
lases when excited by pump light. The sensor 450 is comprised of
two modules 454 and 456. The first module 454 is formed of a
dielectric material and contains a receiving cavity 458. The module
454 has a sensing surface 460 disposed above a flexible membrane or
portion 462, such that changes in a measurable parameter at the
sensing surface 460 will deflect the membrane 462. The module 456
is formed of a dielectric material and is disposed in contact with
the microsphere 452. For example, the microsphere 452 may be
disposed in a small spot indentation in the module 456. The
microsphere 452 could be supported by a pedestal. The microsphere
452 is preferably a unitary structure and not formed of halves like
the embodiments of FIGS. 10-12. The microsphere 452 is positioned
below the membrane 462 and collectively the two define a variable
gap 464, In this configuration, light is coupled into the
microsphere 452 and changes in the variable gap 464, i.e., changes
due to changes in the measurable parameter at the sensing surface
460, will affect the resonance condition in the microsphere 452,
thereby changing the frequency of a laser source in CW operation or
changing repetition rate of a pulsed laser source in mode-locked
operation. By way of example, a waveguide 466 is shown for coupling
light in and out of the microsphere 452. Alternatively light could
be coupled to the microsphere 452 through the transparent module
454 by focusing a light beam unto the microsphere 452.
[0084] The ring resonator embodiments like those of FIGS. 7-8 may
be formed integral to a substrate, thereby providing a unitary
structure protecting the resonator and waveguides from damage. An
exemplary integrated optical sensor 500 is shown (unassembled) in
FIG. 14 having a first module 502 and a second module 504. The
first module 502 includes a ring resonator 506 formed using an
implantation, an etch and growth, or other suitable processes. In a
preferred embodiment, the substrate 508 is formed of sapphire and
the ring resonator 506 is formed of gallium arsenide or polysilicon
which have higher indices of refraction than sapphire and thus
provide total internal reflection. A primary waveguide 510 and a
snubbed secondary waveguide 512, similar to the waveguides
described above with respect to FIG. 8, have also been formed in
the substrate 508. The waveguides 510 and 512 and the ring
resonator 506 have top surfaces flush with the top surface 514 of
the substrate 508. Signals propagating with the waveguides 510 and
512, as well as the ring resonator 506, do so under total internal
reflection.
[0085] The module 504 is formed of a substrate 516 which in the
preferred embodiment would be the same material as that of
substrate 508. Module 504 includes a cavity 518 defining a variable
gap. As with the cavity 166 previously described, the cavity 518
has a geometry such that the gap of the cavity 518 will vary in
response to changes to a measurable parameter, like pressure, force
or temperature. Furthermore, while a rectilinear shape is shown in
FIG. 14, it will be understood that other shapes are suitable; for
example, a non-planar shape may be used. The integrated optical
sensor 500 is formed by mounting module 504 on module 502 forming
the structure shown in FIG. 15.
[0086] As shown in cross-section in FIG. 15, the cavity 518 is
external to the ring resonator 506, but close enough to alter the
effective index of refraction experienced by a wave propagating
within the ring resonator 506. The shape of the cavity 518 is
altered in response to changes to the measurable parameters
described above at sensing surfaces 519, and variations in the
variable gap change the resonant frequency of the resonator 506.
The module 504 may, for example, form a diaphragm above the
resonator 506. The output signal on waveguide 512 is coupled to a
detector and processor. The configuration could be used in a CW or
pulse mode operation in accordance with the above teachings.
[0087] Numerous alternatives to the optical sensor 500 will be
apparent to persons of ordinary skill in the art. For example, a
Bragg grating may be formed on a surface of the resonator 506 to
further narrow the bandwidth of the output signal from waveguide
512 or otherwise affect operation. The ring resonator 506 could be
doped to provide integral lasing action or the ring resonator 506
could be coupled to an external laser to provide a variable
frequency output. Additionally, concentric ring resonators may be
used, for example, to compensate for temperature fluctuations. This
alternative is particularly useful as each of the concentric ring
resonators would have different pressure sensitivities due to
differing geometries (in this case radii).
[0088] Even further, the optical sensor 500 could be formed of
symmetrical and identical opposing ring resonators existing on
opposite sides of the cavity 518. The two waveguides would act as a
single mode waveguide with a variable internal gap. FIG. 16 shows
the cross-section of an embodiment in which a second ring resonator
520 is disposed above the cavity 518 and over the ring resonator
506.
[0089] Alternative to the two module structure shown in FIG. 14,
the optical sensor 500 can be formed in a single substrate
structure, i.e., without modules. Here, multiple step processing
may be employed in which a first portion of a substrate layer is
grown and then implanted or etched processed to form the integral
ring resonator and necessary waveguides and then a subsequent
growth stage would be performed to form the cavity on the top
surface of the sensor, The first module 502 is alternatively formed
of a photonic crystal module 530 in FIG. 17. The module 530 has a
primary waveguide 532, a resonator .534, and a secondary waveguide
536. The module 530 may be used with the module 504, as an
alternative to module 502, or the module 530 may be formed with a
variable gap cavity in a unitary structure. The resonator 534 is
formed by a change in the spacing in the photonic crystal array as
is known in the art.
[0090] The waveguides 532, 536 and the resonator 534 are formed in
a 2D photonic crystal lattice array. A photonic crystal generally
consists of a repetitive array of elements where the dimensions for
each element are similar to or on the order of the wavelengths of
light to propagate within the array. Photonic crystals are
desirable because they have tight mode confinement and low losses
even with sharp corners in the waveguide. They also allow for
evanescent coupling. As a result, the module 530 is a low loss
optical coupler, and the resonator 534 is a high Q resonator.
[0091] The module 530 is formed with a 2D array of holes or posts
538 configured in a triangular lattice orientation. The array may
be formed using known photonic crystal formation techniques. For
example, collimated optical beams may bore holes through an optical
substrate material. Lithographic processes by which electron beams
directly write the patterns to be etched in thin membrane or
heterostructures are also known. The formed 2D photonic crystal
array defines the resonator 534 as well as the waveguides 532, 536
and therefore a single processing step may be used to form these
three structures simultaneously.
[0092] In the module 530, a laser signal propagating within the
primary waveguide 532 will evanescent couple into the resonator
534, a ring resonator. As with the other resonators described
herein, the resonator 534 may be formed of lasing material or
non-lasing material. The signal from the resonator 534 is coupled
to the waveguide 536. The module 530 is preferably used with the
external cavity 518, where changes to the sensing surface 519 will
alter the variable gap of the cavity 518 and the frequency of the
resonator signal from the resonator 534.
[0093] Though the embodiments shown in FIGS. 14-17 show a primary,
or input, waveguide and a secondary, or output, waveguide, it will
be understood that a single waveguide may be used as in FIG. 7.
[0094] FIG. 18 shows another embodiment of an optical sensor 600.
The optical sensor 600 is formed of a vertical cavity surface
emitting laser (VCSEL) 602 having an output coupled to an external
resonator 604. The VCSEL 602 exemplarily includes an active region
606 and two reflectors 612 and-614, each being distributed Bragg
reflector layers in the preferred embodiment. The resonator 604 is
a microdisc resonator operating on the principle of total internal
reflection and thus has low losses and a high Q. The resonator 604
has a cavity 620 defining a variable gap that varies in response to
changes in a measurable parameter at a sensing surface 616. The
resonator 604 is mounted to the top surface of the VCSEL 602 for
receiving the output from the VCSEL 602. By way of example, a
transparent dielectric 622 is shown for this purpose. The entire
sensor 600 could be mounted on a substrate or support layer 624 for
easy packaging and placement in existing applications.
[0095] In this embodiment, the output from the resonator 604
depends upon the resonant frequency thereof. The resonant frequency
is a function of the variable gap of the cavity 620 and that
variable gap is a function of the measurable parameters like
pressure and temperature. The VCSEL output is coupled to the high Q
microdisc 604 to determine the frequency of the VCSEL 602.
[0096] As can be seen from the foregoing, a high Q optical
resonator with a resonant frequency that is dependent upon a
measurable parameter such as pressure, temperature, flow rate,
force, material composition, or strain is shown. The resonant
frequency of the resonator determines the output frequency of a
laser, by having the laser lock onto the resonant frequency, or the
resonant frequency may determine the output of a resonator acting
as a filter. The output of either is dependent upon the measurable
parameter at a sensing surface and can be used to calculate an
absolute or differential value for the measurable parameter. The
resonator may be formed of an optical medium external to a laser or
light source or the optical medium may be internal to the source
making the laser cavity the resonator. Numerous waveguides are
described above including dielectric resonators like the microdisc
and microsphere that rely only upon total internal reflection, as
well as resonators that do have waveguides for confining
propagating signals. In addition to those shown, other resonator
structures will be apparent.
[0097] Numerous applications for these teachings have been
described above and yet others will be apparent. The high accuracy
of the described optical devices is well suited for industrial
process and flow system applications, particularly those with low
signal strengths where conventional electronic based semiconductor
sensors often do not work. In one application, an optical remote
pressure sensor could be used where the optical resonator replaces
an oil filled capillary tubing. Another application includes AP
flow meters where pressure is measured in physically separated
locations, and a meter is used to determine the change in pressure.
In contrast, conventional AP sensors require an oil filled isolator
system to couple the two physically separated pressures to a common
sensor. The optical sensors are also suitable for pressure
measurement in high temperature applications where conventional
sensors and electronics do not operate, for example, measuring
pressure in jet engines, measuring pressure in oil wells and
measuring steam. The structures shown could also be used in AP
transmitters where the AP must be measured at high line pressure
(AP). Here dual AP optical sensors with high sensitivity could be
used. Even further, temperature measurement applications where
conventional wiring is not suitable due to electrical interference
or safety considerations can now be achieved through the use of all
optical sensors. Other sensor applications include using optical
sensors to measure flow rate and material composition.
[0098] FIGS. 19-38B show embodiments of the high Q resonant sensor
operating at frequencies in the microwave portion of the
electromagnetic spectrum (i.e. the suboptical portion of the
spectrum including microwave and terahertz frequencies). FIGS.
19-21 show block diagrams of exemplary sensing systems, while FIGS.
22A-38B show embodiments of high Q dielectric resonant sensors
usable with microwave or exitation.
[0099] FIG. 19 shows sensing system including power source 700,
oscillator 702, high Q dielectric resonant sensor 704, and output
sensor 706. Power source 700 provides electrical power or
electromagnetic wave energy to oscillator 702. The output of
oscillator 702 is high frequency electomagnetic radiation in the
microwave portion of the spectrum. For simplicity in description,
the electromagnetic wave energy used in the embodiments shown in
FIGS. 19-21 will be referred to as "microwave" energy.
[0100] The microwave output of oscillator 702 is supplied through a
microwave coupling or transmission line (such as a coaxial cable or
waveguide) to high Q dielectric resonant sensor 704. As will be
discussed in more detail with reference to FIGS. 22A-38B,
dielectric resonant sensor is a body of dielectric material having
an internal cavity which changes geometry in response to a
measurable parameter, such as pressure, differential pressure,
temperature, force or the like. The cavity defines a gap which
changes size in response to changes in the measurable parameter.
Changes in the size of the gap cause a change in the effective
dielectric constant encountered by the electric field component of
the electromagnetic microwave energy (supplied by oscillator 702)
that propagates through sensor 704. The velocity of propagation is
a function of the effective dielectric constant, and therefore the
resonant frequency of resonant sensor 704 changes as a function of
gap size (which is a function of the measurable parameter).
[0101] The embodiment shown in FIG. 19, output sensor 706 is
coupled to dielectric resonant sensor 704 to receive microwave
energy from sensor 704. As frequency of microwave energy supplied
from oscillator 702 is swept through a range of frequencies, output
sensor 706 detects when the input microwave energy is at the
resonant frequency of dielectric resonant sensor 704. The
embodiment shown in FIG. 19, output sensor 706 is connected in a
feedback loop to the frequency sweep control of oscillator 702, so
that the output frequency of oscillator 702 is adjusted to the
resonant frequency of sensor 704. The output frequency of
oscillator 702 as sensed by output sensor 706, is representative of
the measured parameter.
[0102] FIG. 20 shows an embodiment in which oscillator 702 and high
Q dielectric resonant sensor 704 arranged in a self-resonant
configuration. DC power from source 700 powers oscillator 702 which
supplies microwave energy over two-way connection 707 to excite
resonant sensor 704. Impedance changes in sensor 704 lock
oscillator 702 onto the resonant frequency of sensor 704. The
oscillator frequency of oscillator 702 is a function of the
resonant frequency of sensor 704, and is counted by counter 708 to
provide a measure of the measurable parameter.
[0103] FIG. 21 shows an embodiment including power source (or pump)
700, maser 710, high Q dielectric resonant sensor 704, and
measurement apparatus 712. Power source 700 is a microwave
oscillator providing power at a pumping frequency f.sub.1. In this
embodiment, sensor 704 is either within the cavity of maser 710, or
is positioned outside the cavity, and has a resonant frequency
f.sub.2 that is a function of the parameter to be measured.
Measurement apparatus 712 receives the output beam from maser 710,
as affected by resonant sensor 704. Measurement apparatus 712 may
be a spectrometer that detects the frequency of the maser beam, or
may count maser pulses. Both the frequency of the beam, and the
repetition rate of the pulses of the beam are functions of the
resonant frequency of high Q dielectric resonant sensor 704. Since
the cavity geometry (i.e. gap size) and the effective dielectric
constant of sensor 704 varies as a function of the parameter,
frequency and repetition rate of the maser beam received by
measurement apparatus 712 both vary as a function of the measurable
parameter.
[0104] One example of a maser usable with the present invention is
a sapphire (ruby) resonator that can be made to mase at about 32
GHz when excited with a 66.4 GHz pump source. This allows the input
parameter sensitive 32 GHz signal from maser 710 and resonator 704
to be easily distinguished by measurement apparatus 712 from the
pump frequency produced by source 700. A description of a planar
sapphire coupled-cavity maser is shown in J. S. Shell et al.,
T.M.O. Progressive Report 42-142 (Aug. 15, 2002, at page 17.) The
Shell et al. article describes continuous mode masers. Although
FIG. 21 shows a single resonant sensor 704 used with maser 710,
multiple resonant sensors constructed so that they are tuned a
slightly different resonant frequencies close to 32 GHz can also be
used.
[0105] FIGS. 22A-38B show various embodiments of an electromagnetic
resonant sensor which is responsive to pressure (or another
parameter to be sensed) and which operates at microwave
frequencies. The electromagnetic resonance sensor is a dielectric
structure having a cavity gap that changes in response to the
parameter being sensed. The dielectric material may be a single
crystal material such as sapphire, a dielectric ceramic, a glass,
or quartz. In the following discussions, sapphire will be used as
an example of the dielectric material forming the body of the
electromagnetic resonant sensor. The resonant frequency of the
sensor is a function of the square root of the ratio of stored
electric field energy to stored magnetic field energy in the
dielectric body and cavity gap of the sensor. As the cavity gap
changes as a function of the parameter, the effective dielectric
constant of the sensor changes. This causes a change in stored
electric field energy, and therefore a change in resonant
frequency.
[0106] The resonant sensor is typically suspended in a metal cavity
that is either evacuated, or filled with a fluid such as oil or a
gas. Because of the large dielectric constant of sapphire (or other
dielectric material) compared to free space (or the gas or oil
filling), microwave resonant modes will exist within the dielectric
structure that are almost independent of the surrounding cavity.
The electromagnetic energy in selected resonant modes will be
confined to the internal high dielectric structure as an
electromagnetic standing wave within the dielectric body and the
cavity gap. Microwave energy can be fed to the structure and
impedance changes at resonance can be detected, or the system can
be made self resonant. Because of the low dielectric loss of
sapphire at microwave frequencies, the Q of the resonant sensor can
be very high. Q factors of greater then 100,000 at room temperature
and greater then 1,000,000 at cyrogenic temperatures are typical.
Dimensions of the resonant sensor are typically in the millimeter
to centimeter range to be compatible with microwave wavelengths.
Small changes in dimensions of the resonant sensor can be used to
tune the resonant sensor over small frequency ranges. The resonant
sensor has excellent short-term stability because of the high Q.
Long-term stability is excellent because frequency depends only on
the stability of the dimensions and the dielectric constant of the
dielectric of the resonant sensor body.
[0107] FIGS. 22A and 22B show electromagnetic resonant sensor 800
that operates at microwave frequencies. Sensor 800 is a ring or
toroidal shaped dielectric body 802 with an internal ring-shaped
cavity or chamber 804. Faces 806 and 808 of body 802 are responsive
to a parameter to be measured, such as pressure applied through a
oil fill medium that surrounds sensor 800. The pressure causes a
deflection of the walls of body 802 inward, which causes a change
in gap 810, which is defined by the distance between interior
surfaces 804A and 804B of cavity 804.
[0108] Sensor 800 is operated in a whispering gallery mode, which
provides a high gage factor in conjunction with a high Q. This
structure provides a high Q at room temperature, since modes are
available in which the internal electromagnetic radiation is
totally reflected at the interface between dielectric body 802 and
the surrounding medium. This mode is similar to the acoustic
whispering gallery in which sounds are transmitted with very low
loss around the periphery of a circular chamber.
[0109] By interrupting the electric field component of the
electromagnetic standing wave within body 802 with variable gap 810
(which has lower dielectric constant than dielectric body 802), a
relatively high gauge factor can be achieved. An oil fill medium
can be used to transfer pressure to faces 806 and 808 in order to
change the geometry of cavity 804, and particularly to change the
thickness of the pressure sensitive gap. Since the dielectric
constant of the oil is significantly lower than the dielectric
constant of body 802, a high Q resonance that is relatively
independent of the oil dielectric properties can be maintained.
Microwave power can be coupled to sensor 800 by simply introducing
microwave energy into the surrounding oil medium, or by coupling
the microwave energy into discontinuities in dielectric body 802,
such as a small hole or projection. Also, microwave energy can be
transferred to dielectric body 802 by placing the center conductor
of a coaxial cable close to or in contact with body 802.
[0110] FIG. 22A shows sensor 800 operating in a fourth order mode,
with an electromagnetic standing wave having four magnetic field
loops 812 and four electric field loops 814. The mode shown in FIG.
22A has maximum pressure sensitivity, since electric field loops
814 pass through gap 810 of cavity 804. If electric field loops 814
and magnetic field loops 812 are transposed, frequency will have
minimum pressure sensitivity, and sensor 800 can be used for
temperature sensing purposes. In that case, resonant frequency will
depend on the dielectric constant of the sapphire, which is
temperature dependent.
[0111] The number of loops shown in FIG. 22A is by way of example.
Resonant mode with other numbers of loops can be accommodated. High
order modes generally have higher Q and higher resonant
frequencies.
[0112] Electromagnetic resonant modes are based on distributed
rather than lumped effects, and the structures are complex. Exact
solutions are difficult to obtain, but approximations can be made
using rough lumped constants. The electromagnetic standing wave
pattern will consist of interlocking looped magnetic and electric
fields. In the ring resonator structure shown in FIGS. 22A-22B, the
pattern is a circular loop linked chain with alternate horizontal
and vertical links representing electric and magnetic fields. By
interrupting the electrical field loops with variable gap 810, a
change in the measurable parameter effectively changes capacitance
of an LC resonator. The resonant frequency will thus change since
F=K/SqrtLC. In a conventional capacitance sensor, all of the
capacitance will be confined to the gap since the conducting plates
do not support internal capacitance. In the structure shown in
FIGS. 22A-22B, on the other hand, energy is also stored in the
electric field within the sapphire so that a significant part of
the total lumped capacitance is fixed and the gage factor will be
reduced. This will be compensated for by the increased stability
and signal to noise ratio of the resonator. Roughly: 1 C t C S + C
g C S C g where C s S A S L S and C g 0 A g X g
[0113] Where C.sub.s is capacitance due to sapphire and .di-elect
cons..sub.s is sapphire dielectric constant and A.sub.s is
effective area of electric field in sapphire and L, is effective
length of electric field in sapphire and C.sub.g is capacitance due
to gap 810 and .di-elect cons..sub.o is dielectric constant of
vacuum and A.sub.g is effective area of gap 810 and X.sub.g is the
size of gap 810.
[0114] An alternate approach is to examine the effective wavelength
of the standing wave electromagnetic field. In general the smallest
dimension will determine the effective wavelength. If an air (or
vacuum) gap is inserted in the electric field loop in a high
dielectric material it will effectively increase the wavelength by
a ratio equal to the ratio of the dielectric constants of the
dielectric material and the gap.
[0115] The embodiment illustrated in FIG. 22A-22C, approximate
operating frequencies and gage factors can be calculated, depending
upon the ring thickness T.sub.s, the pressure sensitive gap
X.sub.o, and the maximum deflection or change in the gap .DELTA.X.
When .DELTA.X is 0.5 mil, X.sub.o is 1 ml, .di-elect cons..sub.s is
10 and T.sub.s is 50 mil, the approximate operating frequency is 40
GHz. 2 f c s 40 GHz
[0116] The approximate gage factor GF is 0.1. 3 GF 1 2 X s T s s 0
1 2 X X 0 s 0 X 0 T s 1 2 10 1 1 50 0.1
[0117] It may be desirable to lower the resonant frequency for a
given size sensor in order to reduce power and to increase
compatability of the sensor with semiconductor circuitry. One way
of accomplishing lowered resonant frequency is to use a sensor
design that is intermediate between a lumped LC circuit and a
waveguide. FIGS. 23A and 23B show sensor 820, which has such a
configuration.
[0118] Sensor 820 is a cylindrical dielectric body 822 with
internal cavity 820 divided into two zones: central zone 826 and
surrounding zone 828. Central zone 826 is thinner and presents a
pressure responsive variable gap between interior surfaces 826A and
826B.
[0119] FIGS. 23A and 23B show magnetic field loop 830. Electric
field loop 832 is shown in FIG. 23B. As shown in FIGS. 23A and 23B,
the electric field predominates in zone 826 and represents a
capacitance which varies as the variable gap X changes in response
to pressure. The magnetic field predominates in zone 828 and
represents and inductance. In essence, sensor 820 is a single turn
torroidal inductor coupled to a capacitance.
[0120] FIGS. 24A and 24B show another embodiment of a dielectric
resonator sensor in accordance with the present invention. Sensor
840 is a generally rectangular block of dielectric material 842
having a generally square front and rear faces 844 and 846 and a
circular cylindrical internal cavity 848 with a variable gap X
defined by interior surfaces 848A and 848B. The thickness of body
842 is approximately one-half wavelength of the resonant frequency
sensor 840. FIGS. 24A and 24B illustrate sensor 840 operating at
the lowest resonant mode, with a single magnetic field loop 850 and
a single electric field loop 852.
[0121] FIGS. 25A and 25B illustrate sensor 840 operating in a
higher resonance mode, with higher Q and higher resonant frequency.
In this embodiment, resonance is established in a whispering
gallery mode with a standing wave which consists of a link chain of
magnetic field loops 850 and electric field loops 852, with an
effective diameter D. The wavelength .lambda. is given by the
following relationship: 4 2 D 4
[0122] Other forms of resonant sensor using the present invention
have resonant transmission line or resonant antenna configurations.
FIGS. 26A and 26B show a resonant antenna embodiment of sensor 840
in which conducting film 854 has been added to one side of internal
cavity 848 to reduce the operating frequency of sensor 840. A half
wavelength of the resonant frequency is approximately equal to the
radius of conducting film 854. Magnetic field loop 850 and electric
field loops 852 of the electromagnetic standing wave are shown.
Sensor 840 acts as a resonant circular antenna.
[0123] FIGS. 27A and 27B show a resonant transmission line sensor
900, which includes dielectric body 902 with cavity 904 and
transmission line 906 located within cavity 904. Transmission line
906 is formed by conductors 908 and 910, which are positioned
opposite one another on opposing walls 904A and 904B of cavity
904.
[0124] In the embodiment shown in FIGS. 27A and 27B, transmission
line 906 is a strip dual transmission line. The speed of
propagation down transmission line 906 will depend on the energy
stored in the electromagnetic field, which is a function of the gap
spacing between surfaces 904A and 904B of cavity 904. If
transmission line 906 is terminated either open or shorted, there
will be an associated resonance which will also vary as the gap X
of cavity 904 changes. This resonance can be sensed to indicate
pressure applied to outer faces 912 and 914 of sensor body 902.
[0125] For a given sensor size, the resonant frequency of resonant
transmission line sensor 900 is lower than the operating
frequencies of the resonant sensors having shown in FIGS. 22A-25B a
resonant cavity with no electrodes. A lower operating frequency has
the advantage of greater compatibility with solid state
electronics. Conversely, for a given operating frequency, the
sensor can be smaller and thus lower cost.
[0126] Other configurations are possible. Both conductors 908-910
can be located on the same side of cavity 904. Transmission line
906 can fold back and forth in order to lower the operating
frequency even further. Conductors 908 and 910 can be separated
physically and capacitively coupled before being brought through a
feedthrough so that the resonance is primarily determined within
cavity 904. External coupling electrodes can be used to sense the
resonance. In still another embodiment, a single conductor can be
used with an external ground to form a coaxial type resonator.
[0127] The resonant frequency of sensor 900 is determined by the
velocity of propagation on transmission line 906 and its length 5 f
= V = V 2 L
[0128] where V is propagation velocity and .lambda. is the
wavelength of the standing wave, and the mode is a resonant half
wavelength transmission line with both ends shorted or open. The
propagation velocity in this structure is a function of the average
or effective dielectric constant through the electromagnetic wave
energy is traveling. Sensor 900 is configured so that the standing
wave is confined such that a significant percentage of its total
energy is stored in the gap. The remainder of the energy is stored
in the sapphire which has a dielectric constant of about 10 times
the dielectric constant in the gap. Under these circumstances 6 V =
1 0 _ = c 0 _
[0129] where c is the speed of light, and .di-elect cons..sub.0 is
the dielectric constant of the vacuum, and {overscore (.di-elect
cons.)} is the energy averaged dielectric constant that the
electromagnetic wave is exposed to. Thus the propagation velocity
will vary from approximately c/3 when the gap is totally closed to
a value approaching c when the gap is very large. The resonant
frequency will also change by a 3:1 ratio since it is directly
related to velocity. For a practical design the change will be
smaller than this but a gage factor of 0.1 to 0.5 is probably
reasonable.
[0130] Q (or quality factor) is an important factor in a resonant
circuit since it determines the sharpness of the resonance and thus
determines how accurately the resonant frequency can be determined.
Q should be much larger than 1 for a practical device. Q is an
inverse measure of the energy losses is the system. Most of the
losses will occur in the resistance of the metal films creating the
conductors. Losses in the sapphire and gap dielectric will be low,
and shielding can prevent energy loss due to radiation. The
geometry will determine the relationship between Q and the film
resistivity.
[0131] FIGS. 28A and 28B show another embodiment of the
invention--resonant transmission line sensor 920. Dielectric body
922 has an internal cavity 924 with a pressure responsive variable
gap X between interior surfaces 924A and 924B. Positioned within
cavity 924 is slot antenna 926. The width of the major conductive
lines 928 forming slot antenna 926 is W, and the spacing between
the two major conductive lines is S.
[0132] When the distances between conductive lines 928 is about
equal to the relative dielectric constant of sapphire times cavity
gap X, the sensor 920 will have a reasonable gage factor in
combination with a reasonable Q. The width W of the lines 928 can
be varied to optimize the Q versus the gage factor. When they are
very wide, the structure resembles a resonant slot more than a
transmission line but the principle of operation is very similar.
In both cases charge resonates back and forth between the facing
central portions of the two lines 928. It does this via a current
flow around closed ends 930 that connect lines 928. Q can be
estimated as follows: 7 Q = T 2 R e C e 8 T R t C t
[0133] where T is the period of oscillation and R.sub.e is the
effective resistance that charges the effective capacitance of
C.sub.e transmission line 926. R and C are distributed and the
charging voltages and currents are distributed sinusoidal. Thus the
effective values R.sub.e and C.sub.e are less than the total values
R.sub.t and C.sub.t. Since T is relatively constant for any given
line length L, maximizing Q is a matter of finding a structure with
a small R in combination with a small C. The-edge-to-edge
orientation of this structure gives a small capacitance in
conjunction with a low resistance. It allows relatively low
resistance since the lines can be made wide without appreciably
increasing the capacitance. One example of values using the
structure of sensor 920 is:
[0134] Cavity gap X=6 microns.
[0135] Line spacing S=50 microns.
[0136] Line width W=200 microns.
[0137] Line length L=0.25 cm
[0138] Resistivity R.sub.s=0.5 ohms/square
V=c/sqrt9=c/3=10.sup.10 cm/scc
f=V/2L=10.sup.-10 cm/sec/2*0.25 cm=20 GHz
T=0.5.times.10.sup.-10 sec
Rt=# of squares.times.Rs=28 squares.times.0.5 ohms/square=14
ohms
C.sub.t=2 pf/cm.times.0.25 cm =5 pf 8 Q = 8 3.14 5 .times. 10 - 10
sec 14 ohms .times. 5 .times. 10 - 12 fd = 18
[0139] Higher Q can be obtained by increasing film thickness, or
other dipole and slot antenna structures can be used.
[0140] FIG. 29A and 29B show sensor 940 having a dielectric body
942, internal cavity 944, and coaxial transmission line 946. Sensor
940 offers a simple structure with a transmission line 946 which
can be considered as either a coaxial transmission line with open
terminations, or a dipole antenna. Length and velocity determine
the resonant frequency. Velocity of the electromagnetic wave along
transmission line 946 will depend upon the dimension gap X between
surfaces 944A and 944B of cavity 944. The structure shown in FIGS.
29A and 29B have only a few squares of sheet resistance impeding
current flow, and thus have higher Q. In addition, the structure
shown in FIGS. 29A and 29B tends to have lower capacitance per unit
length, which also helps to raise Q.
[0141] In the embodiment shown in FIGS. 29A and 29B, capacitance
sensitivity of sensor 940 is at a maximum near the ends of body
942. As a result, the antenna 946 should be oriented as shown in
FIG. 29A.
[0142] Slot type antennas can also be used in conjunction with the
present invention. Examples of resonant sensor 950 with different
of slot type antennas are illustrated in FIGS. 30A and 30B, 31A and
31B, and 32A and 32B, 33A and 33B, 34A and 34B, and 35A and 35B. In
these figures, resonant sensor 950 has a dielectric body 952,
cavity 954 with interior surfaces 954A and 954B defining variable
gap X, and conductor 956 on surface 954B. Conductor 956 contains a
slot antenna of a different configuration in each FIG. 30A-35A. In
FIG. 30A the length of slot antenna 958 and the velocity of the
electromagnetic wave energy (which is a function of gap size X)
will determine resonant frequency. For maximum Q, slot type
antennas have to be relatively short, since current must flow
unimpeded around the ends of the slot. This results in
correspondingly higher resonant frequencies.
[0143] In FIG. 31A, a single vertically oriented slot 960 is shown.
The frequency determining length is the vertical length of slot
960.
[0144] In FIG. 32A, slot 962, surrounded by a pair of circular
cutouts 964 and 966, form a slot antenna. Cutouts 964 and 966
enhance Q by reducing capacitance near the sensitive central
area.
[0145] FIG. 33A shows a slot antenna formed by five parallel slots
968. This structure has capability of very high Q if it oscillates
with alternative polarities on adjacent slots. Current flow should
be in a horizontal direction between the slots to minimize the
losses.
[0146] FIGS. 34A and 35A show further embodiments in which slots
are "widened" to form port type antennas. FIG. 34A shows circular
port 970, while FIG. 35A shows an elliptical port 972.
[0147] In variations of the embodiments shown in FIGS. 30A-35B
multiple antennas (slot or circular port antennas) can be formed in
the conductor resulting in multiple resonant frequencies. The use
of different antennas and different lengths in one sensor allows
the potential of providing a reference or a temperature signal,
along with the signal representing the parameter of interest.
[0148] FIGS. 36A and 36B and FIGS. 37A and 37A show two additional
embodiments of sensor 950 featuring dipole antennas. Circular
conductor 980 is shown in FIG. 36A and elliptical conductor 982 is
shown in FIG. 37A. The frequency determining length is in the
horizontal direction as shown in FIGS. 36A and 37A.
[0149] FIGS. 38A and 38B show still another embodiment of a sensor
having a resonant antenna. Sensor 990 has a dielectric sensor body
992 with square front and real surfaces 994 and 996 with a square
internal cavity 998 defining variable gap X between interior
surfaces 998A and 998B. Conductor 1000 is formed on 998B wall of
cavity 994 and has an elliptical opening 1002 which forms a port
antenna.
[0150] As illustrated in FIGS. 19-38B, resonant sensor is operating
with microwave oscillators can be used to sense various parameters,
so long as the parameter causes a change in the cavity gap of the
resonant sensor. In general, the size of the device is operating at
microwave frequencies will be roughly equal to one half the
wavelength of the electromagnetic standing wave. If the resonant
sensor is made small to reduce cost, it will tend to have a
resonant frequency that is too high to measure directly with low
cost semiconductor frequency coning techniques. For example, 1 GHz
operation requires a 7.5 cm sensor size.
[0151] A solution to this problem is to operate the microwave
oscillator in a mode-locked configuration in which the fundamental
output frequency is amplitude modulated (AM) such that a lower
frequency AM output signal is directly proportional to the
fundamental frequency. Various detection techniques are then
available to convert the power fluctuations of the AM signal to an
electronic signal that can be counted. For instance, a high
frequency diode can rectify the signal.
[0152] FIG. 39 illustrates an embodiment of the present invention
which operates high frequency microwave resonant sensors in a
mode-locked configuration. Sensing system 1050 includes power
supply 1052, sensor structure 1054, mode locking coupling system
1056, and AM frequency output device 1058.
[0153] Sensor structure 1054 includes multiple resonators that are
responsive to a common sensed parameter such as pressure. In this
particular embodiment, sensor structure 1054 includes resonator A
and resonator B which operate at fundamental frequencies 10 GHz and
10.1 GHz respectively. Each resonator includes a GHz oscillator,
such as a Gunn diode oscillator coupled to a microwave resonant
sensor, such as the sensors shown in FIGS. 22A-38B. Power supply
1052 supplies power to each oscillator, so that it oscillates at a
different fundamental frequency, in this case 10 GHz and 10.1
GHz.
[0154] The output of resonators A and B is combined by mode locking
coupling system 1056 to produce a unitary mode-locked AM output.
The output of coupling system 1056 is sensed by output detector
1058.
[0155] With the system shown in FIG. 39, the size of each resonant
sensor is reduced by a factor of 100, since the output frequency
(100 MHz) to be detected is a factor of 100 lower than the
fundamental frequencies (.about.10 GHz) at which each of the
resonators is operating. This allows a resonator size of about 7.5
mm, which can be fabricated relatively inexpensively.
[0156] The operation of multiple oscillators in a mode-locked
configuration is described in "Mode-Locked Oscillator Arrays" R. A.
York and R. C. Compton, IEEE Microwave and Guided Wave Letters,
Vol. 1, No. 8, Aug. 1991, pp. 215-218.
[0157] Although FIG. 39 shows an embodiment using two resonators in
a mode-locked configuration, additional resonators can be included
in the sensor structure, with the resonators all having frequencies
that are a function of the sensed parameter and are equally spaced
in the frequency domain at a given value of the sensed parameter.
In addition, although Gunn diode oscillators are one desirable form
of oscillator for use in the embodiment shown in FIG. 39, other
forms of oscillators can be used as well. For example, maser
oscillators could also be used, where the pump operates at a
desired pump frequency which is different than either the
fundamental frequencies of the resonators or the difference
frequencies between the sensor oscillators.
[0158] Many additional changes and modifications could be made to
the disclosed embodiments without departing from the fair scope and
spirit thereof. The scope of some changes is discussed above. The
scope of others will be come apparent from the appended claims.
* * * * *