U.S. patent application number 10/718338 was filed with the patent office on 2004-05-27 for space time transmit diversity for tdd/wcdma systems.
Invention is credited to Dabak, Anand G., Schmidl, Timothy M., Sengupta, Chaitali.
Application Number | 20040101032 10/718338 |
Document ID | / |
Family ID | 32330007 |
Filed Date | 2004-05-27 |
United States Patent
Application |
20040101032 |
Kind Code |
A1 |
Dabak, Anand G. ; et
al. |
May 27, 2004 |
Space time transmit diversity for TDD/WCDMA systems
Abstract
A circuit is designed with a matched filter circuit including a
plurality of fingers (700, 702, 704) coupled to receive a data
symbol. Each finger corresponds to a respective path of the data
symbol. Each finger produces a respective output signal. A
plurality of decoder circuits (706, 708, 710) receives the
respective output signal from a respective finger of the plurality
of fingers. Each decoder circuit produces a respective output
signal. A joint detector circuit (1310) is coupled to receive each
respective output signal from the plurality of decoder circuits.
The joint detector circuit produces an output signal corresponding
to a predetermined code.
Inventors: |
Dabak, Anand G.; (Plano,
TX) ; Schmidl, Timothy M.; (Dallas, TX) ;
Sengupta, Chaitali; (Dallas, TX) |
Correspondence
Address: |
TEXAS INSTRUMENTS INCORPORATED
P O BOX 655474, M/S 3999
DALLAS
TX
75265
|
Family ID: |
32330007 |
Appl. No.: |
10/718338 |
Filed: |
November 18, 2003 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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10718338 |
Nov 18, 2003 |
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09514452 |
Feb 25, 2000 |
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60121541 |
Feb 25, 1999 |
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60121657 |
Feb 25, 1999 |
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60135263 |
May 21, 1999 |
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Current U.S.
Class: |
375/143 ;
375/152; 375/343; 375/347; 375/E1.025; 375/E1.032 |
Current CPC
Class: |
H04B 1/71052 20130101;
H04B 1/7115 20130101; H04L 1/005 20130101; H04B 1/7093 20130101;
H04B 1/71075 20130101; H04B 7/0897 20130101; H04B 7/0669 20130101;
H04L 1/0618 20130101; H04L 25/03063 20130101; H04B 1/7105 20130101;
H04L 25/0224 20130101 |
Class at
Publication: |
375/143 ;
375/152; 375/343; 375/347 |
International
Class: |
H04K 001/00; H04B
001/69; H04B 001/707 |
Claims
What is claimed:
1. A circuit, comprising: a matched filter circuit coupled to
receive a plurality signals within a time slot, the plurality of
signals including a sequence of predetermined signals interposed
within a plurality of data signals, the matched filter circuit
producing an output signal in response to the data signals; and a
decode circuit coupled to receive the output signal, the output
signal including a first data symbol and a transform of a second
data symbol, the decode circuit producing a decoded first data
symbol and a decoded second data symbol.
2. A circuit as in claim 1, wherein the decode circuit produces
each of the decoded first data symbol and the decoded second data
symbol in response to the first data symbol and the transform of
the second data symbol.
3. A circuit as in claim 2, wherein the transform of the second
data symbol is a complex conjugate of the second data symbol.
4. A circuit as in claim 1, wherein the predetermined signals
comprise a midamble.
5. A circuit as in claim 1, wherein the matched filter circuit
further comprises a plurality of fingers coupled to receive the
plurality of signals, and wherein each finger corresponds to a
respective path of the plurality of signals, each finger producing
a respective output signal.
6. A circuit as in claim 5, wherein the respective output signal of
each finger of the plurality of fingers comprises plural output
signals, and wherein each of the plural output signals corresponds
to a respective shift of a code of the predetermined signals.
7. A circuit as in claim 6, wherein said each respective shift of a
code of the predetermined signals is a respective shifted sample of
a code sequence.
8. A circuit as in claim 6, further comprising: a plurality of
decode circuits, each decode circuit coupled to receive the plural
output signals from a respective finger, each decode circuit
arranged to produce a respective first decoded symbol and a
respective second decoded symbol; and a joint detector circuit
coupled to receive the respective first decoded symbol and the
respective second decoded symbol from each respective decode
circuit, the joint detector circuit combining the respective first
decoded symbol and the respective second decoded symbol from each
said finger corresponding to each said respective code.
9. A circuit as in claim 8, wherein the joint detector attenuates
interference by parallel interference cancellation.
10. A circuit as in claim 8, wherein the joint detector attenuates
interference by zero forcing interference cancellation.
11. A circuit as in claim 8, wherein the joint detector attenuates
interference by minimum mean squared error interference
cancellation.
12. A circuit as in claim 1, wherein the predetermined signals
within each time slot correspond a respective user, and wherein the
predetermined signals corresponding to each user are encoded with a
respective shift of a code sequence.
13. A circuit, comprising: a matched filter circuit having a
plurality of fingers and coupled to receive a respective plurality
signals, the plurality of signals including a sequence of
predetermined signals and a plurality of data signals having a
spreading factor of one, the matched filter circuit producing an
output signal in response to the data signals; a decode circuit
coupled to receive the output signal, the output signal including a
first data symbol and a transform of a second data symbol, the
decode circuit producing a decoded first data symbol and a decoded
second data symbol; and an equalizer circuit coupled to receive the
decoded first data symbol and the decoded second data symbol, the
equalizer circuit producing an output signal corresponding to a
predetermined code.
14. A circuit as in claim 13, wherein each finger corresponds to a
respective path of the plurality of signals, and wherein each
finger produces a respective output signal.
15. A circuit as in claim 13, wherein the decode circuit produces
each of the decoded first data symbol and the decoded second data
symbol in response to the first data symbol and the transform of
the second data symbol.
16. A circuit as in claim 15, wherein the transform of the second
data symbol is a complex conjugate of the second data symbol.
17. A circuit as in claim 13, wherein the decode circuit is further
coupled to receive a transform of the first data symbol and the
second data symbol.
18. A circuit as in claim 13, wherein the equalizer circuit
attenuates interference by zero forcing interference
cancellation.
19. A circuit as in claim 13, wherein the equalizer circuit
attenuates interference by minimum mean squared error interference
cancellation.
20. A circuit, comprising: a matched filter circuit coupled to
receive a first data symbol and a transform of a second data
symbol, the matched filter circuit producing an output signal; and
a joint detector circuit coupled to receive each respective output
signal from the plurality of rake receiver circuits, the joint
detector circuit producing an output signal corresponding to a
predetermined code.
21. A circuit as in claim 20, wherein the predetermined code
corresponds to a mobile receiver.
22. A circuit as in claim 20, wherein the predetermined code is a
subset of a code sequence corresponding to a plurality of mobile
receivers.
23. A circuit as in claim 20, wherein the joint detector circuit
further comprises a decoding circuit.
24. A circuit as in claim 20, wherein the decode circuit is further
coupled to receive a transform of the first data symbol and the
second data symbol.
25. A circuit as in claim 20, wherein the equalizer circuit
attenuates interference by zero forcing interference
cancellation.
26. A circuit as in claim 20, wherein the equalizer circuit
attenuates interference by minimum mean squared error interference
cancellation.
27. A circuit as in claim 20, wherein the equalizer circuit
attenuates interference by parallel interference cancellation.
28. A circuit, comprising an encoder circuit coupled to receive a
plurality of first and second symbols, the encoder circuit
producing the plurality of first symbols at a first output terminal
and a transform of the plurality of second symbols at a second
output terminal within a time slot, the encoder circuit producing a
sequence of predetermined signals interposed within the plurality
of first symbols.
29. A circuit as in claim 28, further coupled to receive a control
signal, the encoder circuit producing the plurality of first
symbols at the first output terminal and the transform of the
plurality of second symbols at the second output terminal in
response to a first value of the control signal, the encoder
circuit producing the plurality of first symbols at the first
output terminal and not producing the transform of the plurality of
second symbols at the second output terminal in response to a
second value of the control signal.
30. A circuit as in claim 28, further comprising a diversity
control circuit coupled to receive a first input signal, the
diversity control circuit producing the control signal
corresponding to the first input signal.
31. A circuit as in claim 28, wherein the first input signal
corresponds to a Doppler frequency.
32. A circuit as in claim 28, wherein the diversity control circuit
is further coupled to receive a second input signal corresponding
to a handoff signal.
33. A circuit as in claim 28, wherein the first input signal
corresponds to a handoff signal.
34. A circuit as in claim 28, wherein the encoder circuit produces
a midamble after the first symbol and before the second symbol.
35. A circuit as in claim 28, wherein the sequence of predetermined
signals comprises a code sequence, and wherein a first shift of the
code sequence corresponds to the first output terminal and a second
shift of the code sequence corresponds to the second output
terminal.
36. A circuit, comprising a filter circuit coupled to receive a
sequence of predetermined signals from a first and a second remote
antenna, wherein the sequence of predetermined signals comprises a
code sequence, and wherein a first shift of the code sequence
corresponds to the first remote antenna and a second shift of the
code sequence corresponds to the second remote antenna, the filter
circuit producing an output signal in response to the data
signals.
37. A circuit as in claim 36, wherein the predetermined signals
comprise a midamble.
38. A circuit as in claim 36, wherein said each respective shift of
a code of the predetermined signals is a respective shifted sample
of a code sequence.
39. A circuit, comprising an encoder circuit coupled to receive a
plurality of symbols, the encoder circuit producing the plurality
of symbols and a sequence of predetermined signals at a first and a
second output terminal, wherein the sequence of predetermined
signals comprises a code sequence, and wherein a first shift of the
code sequence corresponds to the first output terminal and a second
shift of the code sequence corresponds to the second output
terminal.
Description
CLAIM TO PRIORITY OF PROVISIONAL APPLICATION
[0001] This application claims priority under 35 U.S.C. .sctn.
119(e)(1) of provisional application serial No. 60/121,541, filed
Feb. 25, 1999, provisional application serial No. 60/121,657, filed
Feb. 25, 1999, and provisional application serial No. 60/135,263,
filed Feb. 21, 1999.
FIELD OF THE INVENTION
[0002] This invention relates to wideband code division multiple
access (WCDMA) for a communication system and more particularly to
space time block coded transmit antenna diversity for WCDMA.
BACKGROUND OF THE INVENTION
[0003] Present code division multiple access (CDMA) systems are
characterized by simultaneous transmission of different data
signals over a common channel by assigning each signal a unique
code. This unique code is matched with a code of a selected
receiver to determine the proper recipient of a data signal. These
different data signals arrive at the receiver via multiple paths
due to ground clutter and unpredictable signal reflection. Additive
effects of these multiple data signals at the receiver may result
in significant fading or variation in received signal strength. In
general, this fading due to multiple data paths may be diminished
by spreading the transmitted energy over a wide bandwidth. This
wide bandwidth results in greatly reduced fading compared to narrow
band transmission modes such as frequency division multiple access
(FDMA) or time division multiple access (TDMA).
[0004] New standards are continually emerging for next generation
wideband code division multiple access (WCDMA) communication
systems as described in U.S. patent application Ser. No.
90/205,029, filed Dec. 3, 1998, and incorporated herein by
reference. Therein, Dabak et al. describe a method of space-time
transmit diversity (STTD) for frequency division duplex (FDD) WCDMA
systems. These FDD systems are coherent communications systems with
pilot symbol assisted channel estimation schemes. These pilot
symbols are transmitted as quadrature phase shift keyed (QPSK)
known data in predetermined time frames to any receivers within
range. The frames may propagate in a discontinuous transmission
(DTX) mode. For voice traffic, transmission of user data occurs
when the user speaks, but no data symbol transmission occurs when
the user is silent. Similarly for packet data, the user data may be
transmitted only when packets are ready to be sent. The frames
include pilot symbols as well as other control symbols such as
transmit power control (TPC) symbols and rate information (RI)
symbols. These control symbols include multiple bits otherwise
known as chips to distinguish them from data bits. The chip
transmission time (T.sub.C), therefore, is equal to the symbol time
rate (T) divided by the number of chips in the symbol (G).
[0005] Time division duplex (TDD) provides an alternative
communication standard for WCDMA, FDD systems. TDD data are
transmitted as QPSK symbols in data packets of a predetermined
duration or time slot. Each data packet includes a predetermined
training sequence or midamble within the time slot. Data packets
are exchanged within a cell formed by a base station in
communication with nearby mobile units. Data in adjacent cells are
modulated by different periodic codes. The midamble is formed by
adding time shifted versions of the same basic sequence, wherein
each time shift corresponds to a mobile unit within the cell. The
spreading factor (SF) or chips per symbol of the modulation is
preferably sixteen or less. The basic periodic code that modulates
midamble symbols within the cell is shifted to uniquely identify
each mobile unit within the cell. Since the periodic code within
the cell is the same and the spreading factor is small, however,
interference from the base station and other mobile units within
the cell is not received as Gaussian noise. Typical matched filter
circuits used in FDD systems, therefore, are unsuitable for
eliminating this intra cell interference. A solution to this
problem was presented by Anja Klein et al., Zero Forcing and
Minimum Mean-Square-Error Equalization for Multiuser Detection in
Code-Division Multiple-Access Channels, IEEE Trans. on Vehicular
Technology, 276-287 (1996), and incorporated by reference herein.
Therein, Klein et al. teach zero forcing (ZF) and minimum
mean-square-error (MMSE) equalization with and without decision
feedback (DF) to reduce both inter-symbol interference (ISI) and
multiple-access interference (MAI). Klein et al. further cites P.
Jung, J. Blanz, and P. W. Baier, Coherent Receiver Antenna
Diversity for CDMA Mobile Radio Systems Using Joint Detection,
Proc. IEEE Int. Symp. Pers. Indoor and Mobile Radio Communications,
488-492 (1993), for the proposition that these techniques may be
used in combination with antenna diversity. A. Naguib, N. Seshadri
and A. R. Calderbank, Applications of Space-Time Block Codes and
Interference Suppression for High Capacity and High Data Rate
Wireless Systems, Proc. of the Asilomar Conference, 1803-1810
(1998) further expand the work of Klein et al. Space time transmit
diversity, however, was unknown at the time of either work. Thus,
neither Klein et al. nor Jung et al. teach or suggest a method to
combine STTD with joint detection of TDD systems. Moreover, neither
Klein et al. nor Jung et al. teach a communication system having
the advantages of STTD and joint detection of TDD systems.
SUMMARY OF THE INVENTION
[0006] These problems are resolved by a circuit designed with a
matched filter circuit including a plurality of fingers coupled to
receive a data symbol. Each finger corresponds to a respective path
of the data symbol. Each finger produces a respective output
signal. A plurality of decoder circuits receives the respective
output signal from a respective finger of the plurality of fingers.
Each decoder circuit produces a respective output signal. A joint
detector circuit is coupled to receive each respective output
signal from the plurality of decoder circuits. The joint detector
circuit produces an output signal corresponding to a predetermined
code.
[0007] The present invention improves reception by providing at
least 2L diversity over time and space for TDD systems. No
additional transmit power or bandwidth is required. Power is
balanced across multiple antennas.
BRIEF DESCRIPTION OF THE DRAWINGS
[0008] A more complete understanding of the invention may be gained
by reading the subsequent detailed description with reference to
the drawings wherein:
[0009] FIG. 1 is a block diagram of a transmitter of the present
invention using diversity control;
[0010] FIG. 2 is a block diagram of a communication system of the
present invention showing communication with mobile units with and
without diversity;
[0011] FIG. 3 is a diagram of a TDD radio frame;
[0012] FIG. 4 is a diagram of a time slot within the radio frame of
FIG. 3;
[0013] FIG. 5A is a diagram showing an embodiment of the symbol
transmit sequence for TDD with STTD encoding;
[0014] FIG. 5B is a diagram showing an embodiment of the midamble
structure that is used for channel estimation;
[0015] FIG. 6A is a block diagram showing signal flow for a single
user for a TDD receiver of the present invention using STTD
encoding;
[0016] FIG. 6B is a schematic diagram of the STTD decoder of FIG.
6A;
[0017] FIG. 7 is a block diagram showing signal flow for multiple
users for a TDD receiver of the present invention using STTD
encoding;
[0018] FIG. 8 is a block diagram showing parallel interference
cancellation of the present invention for TDD with STTD
encoding;
[0019] FIG. 9A is a block diagram of interference cancellation with
an STTD decoder and a zero forcing STTD equalizer;
[0020] FIG. 9B is a detailed block diagram of FIG. 9A showing the
zero forcing STTD equalizer with decision feedback;
[0021] FIG. 10A is a block diagram of interference cancellation
with an STTD decoder and a minimum mean squared error STTD
equalizer:
[0022] FIG. 10B is a detailed block diagram of FIG. 10A showing the
minimum mean squared error STTD equalizer with decision
feedback;
[0023] FIG. 11 is a simulation diagram showing bit error rate (BER)
as a function of bit energy to noise (Eb/N.sub.0) with and without
diversity for vehicular Doppler rates with a spreading factor of
16;
[0024] FIG. 12 is a simulation diagram showing bit error rate (BER)
as a function of bit energy to noise (Eb/N.sub.0) with and without
diversity for pedestrian Doppler rates with a spreading factor of
16 and 8 users;
[0025] FIG. 13A is a block diagram of a receiver of the present
invention including STTD decoders before the rake receivers and
joint detector;
[0026] FIG. 13B is a block diagram of a receiver of the present
invention having a combined joint detector and STTD decoder;
and
[0027] FIG. 13C is a block diagram of a receiver of the present
invention including a joint detector followed by an STTD
decoder.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0028] Referring to FIG. 1, there is a simplified block diagram of
a transmitter of the present invention using Space-Time Transit
Diversity (STTD). The transmitter circuit includes a diversity
control circuit 100 that is coupled to receive a Doppler control
signal on lead 102 and a handoff control signal on lead 104. The
Doppler control signal is determined by comparing sequential
midamble symbols from mobile units in the same cell as the
transmitter. An increasing difference between received midamble
symbols indicates a greater Doppler rate due to velocity of the
mobile unit with respect to the transmitter. The handoff signal is
determined by mobile unit reports indicating received signal
strength from surrounding base stations. For low Doppler rates when
no base station handoff is required, the diversity control circuit
produces a first value of a control signal on lead 108. This first
value applied to STTD encoder circuit 110 directs the encoder
circuit to apply switched transmit diversity (STD) to transmit
antennas 112 and 114. Thus, received symbols on leads 106 are
alternately transmitted at antennas 112 and 114.
[0029] Alternatively, for high Doppler rates or when a base station
handoff is required, the diversity control circuit produces a
second value of a control signal on lead 108. This second value
directs the STTD encoder circuit 110 to apply STTD to transmit
antennas 112 and 114. Thus, the encoder circuit simultaneously
produces a symbol S.sub.1 at antenna 112 and transformed symbol
-S.sub.2.sup.* at antenna 114. These symbols are transmitted to a
remote mobile antenna 120 along multiple paths 116 and 118. This
design is highly advantageous in providing improved communication
via STTD encoding for high Doppler rates as well as during weak
signal periods such as during base station handoff. For broadcast
channels such as the primary common control channel (PCCPCH), for
example, STTD encoding is preferably used for all transmissions.
This is advantageous, since the broadcast channel transmission is
directed to all mobile receivers without regard to their specific
diversity requirements.
[0030] Turning now to FIG. 2, there is a block diagram of a
communication system of the present invention showing communication
with mobile units with and without diversity. The exemplary
configuration provides STTD for users 1 through Z and no diversity
for users Z+1 through K. The communication system, therefore,
provides STTD for data symbols on lead 202 as well as no diversity
for data symbols on lead 218. Data symbols D.sup.1 at lead 202 are
STTD encoded by encoder circuit 200 to produce encoded data symbols
D.sub.1.sup.1 on lead 204 and encoded data symbols D.sub.2.sup.1 on
lead 206. Encoded data symbols D.sub.1.sup.1 on lead 204 are
multiplied by a predetermined user specific code or sequence
C.sup.1 by circuit 208 and applied to summation circuit 212.
Summation circuit 212 sums these encoded data symbols together with
other user specific data symbols and applies them to antenna 1 at
lead 230. Likewise, data symbols D.sub.2.sup.1 on lead 206 are
multiplied by the same user specific code C.sup.1 by circuit 214
and applied to summation circuit 216. Summation circuit 216 sums
these encoded data symbols together with other user specific data
symbols and applies them to antenna 2 at lead 236. These summed
symbols are transmitted over radio channel 261 to a mobile receiver
antenna at lead 250. The transmitted symbols are effectively
multiplied by channel impulse response matrices H.sub.1 232 and
H.sup.2 238 on respective paths 234 and 240 and summed by path 242.
Noise N is added by path 246 to produce the received signal at
antenna 250. A joint STTD decoder circuit 260 receives the
composite signal and produces user specific symbol sequences
{circumflex over (D)}.sup.1 on lead 252, {circumflex over
(D)}.sup.k on lead 254 and {circumflex over (D)}.sup.K on lead 256,
corresponding respectively to K users.
[0031] In the case where no transmit diversity is employed or where
other forms of diversity such as switched transmit diversity (STD)
or transmit adaptive array diversity (TxAA) are employed, the
transmitter produces symbol sequence D.sub.1.sup.K on lead 218.
This sequence is multiplied by user specific code C.sup.K by
circuit 220 and applied to summation circuit 212. The symbol
sequence D.sub.1.sup.K is summed by circuit 212 together with other
user specific signals and transmitted over the radio channel 261 as
previously described. The communication circuit of the present
invention, therefore, is compatible with STTD as well as no
diversity transmission.
[0032] Referring now to FIG. 3, there is a diagram of a TDD radio
frame that may be transmitted by the communication system of FIG.
2. The radio frame, for example radio frame 300, has a duration of
10 ms. The radio frame is divided into 15 equal time slots 302-310.
Each of these time slots is further divided into 2560 chip times
T.sub.C. The diagram of FIG. 4 illustrates the structure of the TDD
time slot. The time slot includes a first group of data symbols 420
having 1104 chips. This first group corresponds to 69 data symbols
for an exemplary spreading factor of 16. The first group is
followed by a midamble 422 having 16 symbols for the exemplary
spreading factor of 16. These midamble symbols are a predetermined
training sequence similar to pilot symbols of FDD systems. The
midamble symbols are cyclically time shifted for different users in
the cell as previously discussed. A second group of data symbols
424 having another 1104 chips follows the midamble. Finally, the
second group of data symbols is followed by a guard period 426 of
96 chips.
[0033] Referring now to FIG. 5A, there is a diagram showing an
embodiment of the symbol transmit sequence for TDD with STTD
encoding. The exemplary symbol sequence S.sub.1-S.sub.8 shows a
partial sequence of data symbols presented to the transmit circuit
on lead 106 (FIG. 1). This symbol sequence corresponds to data
symbols 420 that precede midamble 422 (FIG. 4). The symbols are
rearranged and transformed for transmission from antennas ANT 1 and
ANT 2 according to symbol transmit times 0T, 2T, . . . . (N+3)T.
There are 2NT symbol transmit times corresponding to the first
group of data symbols 420. Symbol transmit time NT, therefore, is
approximately in the middle of the transmit sequence of data
symbols 420. For example, symbols S.sub.1 and S.sub.2 are
transmitted at transmit times T and 2T, respectively, from antenna
ANT 1. Transformed symbols -S.sub.3.sup.* and -S.sub.4.sup.* are
transmitted simultaneously at transmit times T and 2T,
respectively, from antenna ANT 2. These transformed symbols are
complements of complex conjugates of respective symbols S.sub.3 and
S.sub.4. The sequence continues for symbols 420 and 424 (FIG. 4).
This transmit sequence advantageously provides reduces the
complexity of the zero forcing (ZF-STTD) and the minimum mean
squared error (MMSE-STTD) STTD decoders by allowing the receiver to
neglect the intersymbol interference (ISI) of the block of data
symbols 0 to (N-1)T on the NT of 2NT symbols.
[0034] Referring to FIG. 5B, there is a diagram showing an
embodiment of the midamble pattern that is used for channel
estimation. The basic sequence extends for the entire length of the
midamble except for the cyclic prefix. This basic sequence is
circularly shifted as taught by B. Steiner and P. W. Baier, Low
Cost Channel Estimation in the Uplink Receiver of CDMA Mobile Radio
Systems, Frequenz., vol. 47, 292-298 (1993), to obtain channel
estimates for different users. The cyclic prefix 516 is obtained by
copying over the tail end 514 of the circularly shifted basic
sequence. The shaded region 510 is the first 64 bits of the basic
midamble sequence. In the present invention, the first two time
shifts 511-512 are allotted for channel estimates for antenna 1 and
antenna 2, respectively, of the broadcast channel. The broadcast
channel, therefore, transmits midamble shift 511 from antenna 1
midamble shift 512 from antenna 2. Receivers using STTD preferably
use two midamble shifts for channel estimation similar to the
broadcast channel. Alternatively, a non-STTD receiver preferably
uses the same midamble shift from both antennas with a suitable
weighting corresponding to respective transmit beam forming for
that user.
[0035] Turning now to FIG. 6A, there is a block diagram showing
signal flow at the receiver for a single user for a TDD receiver of
the present invention using STTD encoding. The receiver includes
matched filters 600-604. Each of the matched filter circuits is
coupled to a respective STTD decoder circuit 606-610. The STTD
decoder circuits 606-610 are coupled to rake combiner circuit 612.
Each matched filter and respective STTD decoder corresponds to a
finger of the rake combiner circuit 612. These fingers are coupled
to selectively pass different multipath signals such as Path 1
(116) through Path j (118) of FIG. 1. The selected multipath
signals are then combined by the rake combiner 612 and sent to a
channel decoder such as a Turbo decoder or a Viterbi decoder for
further processing.
[0036] An exemplary STTD decoder 606 shown at FIG. 6B may be used
for the STTD decoders 606-610 of FIG. 6A. Rayleigh fading
parameters are determined from channel estimates of midamble
symbols transmitted from respective antennas at leads 112 and 114.
For simplicity of analysis, a Rayleigh fading parameter
.alpha..sub.j.sup.1 is assumed for a signal transmitted from the
first antenna 112 along the j.sup.th path. Likewise, a Rayleigh
fading parameter .alpha..sub.j.sup.2 is assumed for a signal
transmitted from the second antenna 114 along the j.sup.th path.
Each i.sup.th chip or bit signal r.sub.j(i+.tau..sub.j) of a
respective symbol is subsequently received at a remote mobile
antenna 120 after a transmit time .tau..sub.j corresponding to the
j.sup.th path. The chip signals at lead 620 are multiplied by a
channel orthogonal code at lead 622 by circuit 624 to produce a
user specific signal on lead 626. The signals on lead 626 are
applied to a despreader input circuit 628 where they are summed
over each respective symbol time to produce output signals
R.sub.j.sup.1 at lead 632 and R.sub.j.sup.2 at lead 634
corresponding to the j.sup.th of L multiple signal paths. Circuit
630 delays signal R.sub.j.sup.1 by one symbol time so that it is
synchronized with signal R.sub.j.sup.2 at lead 634. A phase
correction circuit receives signals R.sub.j.sup.1 and R.sub.j.sup.2
as input signals on leads 632 and 634 as shown in equations [1-2],
respectively. 1 R j 1 = i = 0 N - 1 r j ( i + j ) = j 1 S 1 - j 2 S
2 * [ 1 ] R j 2 = i = N 2 N - 1 r j ( i + j ) = j 1 S 1 + j 2 S 1 *
[ 2 ]
[0037] The phase correction circuit receives a complex conjugate of
a channel estimate of a Rayleigh fading parameter
.alpha..sub.j.sup.1* corresponding to the first antenna on lead 644
and a channel estimate of another Rayleigh fading parameter
.alpha..sub.j.sup.2 corresponding to the second antenna on lead
646. Complex conjugates of the input signals are produced by
circuits 636 and 638 at leads 648 and 650, respectively. These
input signals and their complex conjugates are multiplied by
Rayleigh fading parameter estimate signals and summed as indicated
to produce path-specific first and second symbol estimates at
respective output leads 668 and 670 as in equations [3-4].
R.sub.j.sup.1.alpha..sub.j.sup.1*+R.sub.j.sup.2*.alpha..sub.j.sup.2=(.vert-
line..alpha..sub.j.sup.1.vertline..sup.2+.vertline..alpha..sub.j.sup.2.ver-
tline..sup.2)S.sub.1 [3]
-R.sub.j.sup.1*.alpha..sub.j.sup.2+R.sub.j.sup.2.alpha..sub.j.sup.1*=(.ver-
tline..alpha..sub.j.sup.1.vertline..sup.2+.vertline..alpha..sub.j.sup.2.ve-
rtline..sup.2)S.sub.2 [4]
[0038] These path-specific symbol estimates are then applied to the
rake combiner circuit 612 to sum individual path-specific symbol
estimates, thereby providing net soft symbols as in equations [5-6]
at lead 616. 2 S ~ 1 = j = 1 L R j 1 j 1 * + R j 2 * j 2 [ 5 ] S ~
2 = j = 1 L - R j 1 * j 2 + R j 2 j 1 * [ 6 ]
[0039] These soft symbols or estimates provide a path diversity L
and a transmit diversity 2. Thus, the total diversity of the STTD
system is 2L. This increased diversity is highly advantageous in
providing a reduced bit error rate.
[0040] Referring now to FIG. 7, there is a block diagram showing
signal flow for multiple users for a TDD receiver of the present
invention using STTD encoding. This diagram is an extension of the
circuits of FIG. 6A and FIG. 6B to perform parallel interference
cancellation for multiple users as will be described in detail.
There are L fingers which despread received signals from K users.
Matched filter circuits 700-704, therefore, selectively pass L
signals corresponding to each respective multipath for each of K
users. These matched filter output signals are applied to
respective STTD decoder circuits 706-710 and, subsequently, to rake
combiner circuit 712. The rake combiner circuit 712 combines L
multipath signals for each of K. The combined signals for the K
users are applied to symbol decision circuit 714. Each of the K
symbols are determined and produced as output signals on bus
716.
[0041] As previously mentioned, the spreading factor (SF) or chips
per symbol of the modulation is preferably sixteen or less for
these TDD data symbols. Furthermore, the basic periodic code that
modulates midamble symbols within a cell is shifted to uniquely
identify each mobile unit within the cell. Since the periodic code
within the cell is the same and the spreading factor is small,
therefore, interference from the base station and other mobile
units within the cell is not received as Gaussian noise. Typical
matched filter circuits used in FDD systems are unsuitable for
eliminating this intra cell interference. The circuit of FIG. 8 is
a block diagram of a first embodiment of the present invention
showing parallel interference cancellation of the present invention
for TDD with STTD encoding. Data symbols from matched filter
circuits 700-704 are stored in memory circuit 800 as shown in
equation [7].
Y=(y.sub.1,1, y.sub.1,2, . . . , y.sub.1,K, . . . , y.sub.2,1,
y.sub.2,2, . . . , y.sub.2,K, . . . , y.sub.L,1, y.sub.L,2, . . . ,
y.sub.L,K).sup.T [7]
[0042] A cross-correlation matrix is calculated and stored in
memory circuit 802 to determine the interference effect of each
path of each finger of each user on all the other paths. The
cross-correlation matrix R is calculated by first computing all the
cross-correlations between each symbol of each finger for each
user. This step is completed for preceding symbols, present time
symbols and for next time symbols, thereby producing three LK
matrices. Then the middle LK matrix diagonal is set to zero to
exclude self-correlation. Thus, cross-correlation matrix R is
LK.times.3LK. Initial channel estimates for each antenna given by
equations [8-9] are stored in memory circuit 804.
a.sup.(0)=(a.sub.1,1.sup.(0), a.sub.1,2.sup.(0), . . . ,
a.sub.1,K.sup.(0), . . . , a.sub.2,1.sup.(0), a.sub.2,2.sup.(0), .
. . , a.sub.2,K.sup.(0), . . . , a.sub.L,1.sup.(0),
a.sub.L,2.sup.(0), . . . , a.sub.L,K.sup.(0)).sup.T [8]
b.sup.(0)=(b.sub.1,1.sup.(0), b.sub.1,2.sup.(0), . . . ,
b.sub.1,K.sup.(0), . . . , b.sub.2,1.sup.(0), b.sub.2,2.sup.(0), .
. . , b.sub.2,K.sup.(0), . . . , b.sub.L,1.sup.(0),
b.sub.L,2.sup.(0), . . . , b.sub.L,K.sup.(0)).sup.T [9]
[0043] Initial data symbol estimates include two symbols for each
user and are given by equation [10] are stored in memory circuit
818.
D.sub.(0)=(d.sub.1,1.sup.(0), d.sub.1,2.sup.(0), . . . ,
d.sub.1,K.sup.(0), . . . , d.sub.2,1.sup.(0), d.sub.2,2.sup.(0), .
. . , d.sub.2,K.sup.(0)).sup.T [10]
[0044] These initial data symbols are STTD encoded and multiplied
by the initial channel estimates stored in memory circuit 804 as
shown in equations [11-12] for path p and stored in circuit
814.
e.sub.1,p.sup.(0)=(a.sub.p,1.sup.(0)d.sub.1,1.sup.(0)-b.sub.p,1.sup.(0)d.s-
ub.2,1.sup.*(0),
a.sub.p,2.sup.(0)d.sub.1,2.sup.(0)-b.sub.p,1.sup.(0)d.sub-
.2,2.sup.*(0), . . .
,a.sub.p,K.sup.(0)d.sub.1,K.sup.(0)-b.sub.p,K.sup.(0)-
d.sub.2,K.sup.*(0)).sup.T [11]
e.sub.2,p.sup.(0)=(a.sub.p,1.sup.(0)d.sub.2,1.sup.(0)+b.sub.p,1.sup.(0)d.s-
ub.1,1.sup.*(0),a.sub.p,2.sup.(0)d.sub.2,2.sup.(0)+b.sub.p,1.sup.(0)d.sub.-
1,2.sup.*(0), . . .
,a.sub.p,K.sup.(0)d.sub.2,K.sup.(0)+b.sub.p,K.sup.(0)d-
.sub.1,K.sup.*(0)).sup.T [12]
[0045] Circuit 814 multiplies these STTD encoded data symbols of
equations [11-12] by cross correlation matrix R from circuit 802 to
produce a signal estimate given by equation [13].
E=(e.sub.2,1.sup.(-1), e.sub.2,2.sup.(-1), . . . ,
e.sub.2,L.sup.(-1), e.sub.1,1.sup.(0), e.sub.1,2.sup.(0), . . . ,
e.sub.1,L.sup.(0), e.sub.2,1.sup.(0), e.sub.2,2.sup.(0), . . . ,
e.sub.2,L.sup.(0)).sup.T [13]
[0046] This signal estimate is then multiplied by the
cross-correlation matrix R to generate the inter-symbol
interference (ISI) estimate at lead 812. Circuit 820 subtracts this
ISI estimate at lead 812 from the stored matched filter symbols Y
at lead 806 to produce a first iteration of corrected data symbols
on lead 822. This first iteration of new data symbols is decoded
and rake combined at circuit 824 to produce new symbol decisions on
lead 826. These new symbols on lead 826 then replace initial
symbols stored in memory circuit 818. The previous procedure is
then repeated to produce second and subsequent iterations of
corrected data symbols on lead 822. New symbol decisions Y.sub.i
are made for a predetermined number of iterations according to
equation [14] until ISI is effectively cancelled. Thus, the
parallel interference cancellation circuit of FIG. 8 produces new
symbol decisions Y.sub.i as a difference between previous symbol
decisions Y.sub.i-1 and a product of correlation matrix R and the
previous signal estimate matrix E.sub.i-1.
Y.sub.i=Y.sub.i-1-RE.sub.i-1 [14]
[0047] Referring back to FIG. 2, a system model for alternative
embodiments of interference cancellation of the present invention
will be explained in detail. The circuit of FIG. 2 includes a base
station to the left of radio channel 261. The base station
transmits STTD encoded data symbols for L of K users from antenna 1
at 230 given by equation [15]. The base station transmits
corresponding data symbols for these same users at antenna 2 (236)
given by equation [16]. 3 D 1 k = D k 2 ; k = 1 , , L [ 15 ] D 2 k
= ( ( - d 2 k 2 ) * , ( d 1 k 2 ) * , ( - d 4 k 2 ) * , ( d 3 k 2 )
* , , ( - d M k 2 ) * , ( d M - 1 k 2 ) * ) T ; k = 1 , 2 , , L [
16 ]
[0048] The term {square root}{square root over (2)} in the
denominator of equations [15-16] is due to the balanced transmit
power at each antenna for STTD encoding. Data symbols for the
remaining K-L users are transmitted without STTD encoding only from
antenna 1 at 230 given by equation [17].
D.sub.1.sup.k=D.sup.k;k=L+1, . . . ,K [17]
[0049] Transmit data rates for all users are the same. Each data
symbol is repeated G times and multiplied by a respective
user-specific orthogonal code as in equation [18] by circuits 208,
220 and 214.
C.sup.k=(c.sub.1.sup.k, c.sub.2.sup.k, . . . ,
c.sub.G.sup.k).sup.T;k=1, . . . , K [18]
[0050] The chip period for each data symbol is T.sub.c=T.sub.S/G.
After the user-specific spreading, the signals data symbols for all
K users are summed by circuit 212 and applied to antenna 1 at 230.
The radio channel further imposes an impulse response of length W
at 232 on data symbols transmitted by antenna 1 sampled at a chip
rate as in equation [19]. A corresponding impulse response on data
symbols transmitted by antenna 2 is given by equation [20].
H.sub.1=(h.sub.1.sup.1, h.sub.1.sup.2, . . . , h.sub.1.sup.W).sup.T
[19]
H.sub.2=(h.sub.2.sup.1, h.sub.2.sup.2, . . . , h.sub.2.sup.W).sup.T
[20]
[0051] A value of W greater than 1 for a given user results in
inter-symbol interference (ISI) of the user's symbols and multiple
access interference (MAI) of other users symbols due to the loss or
orthogonality. Even though an exemplary chip rate sampling is
assumed for the purpose of illustration, the channel may have to be
sampled at twice the chip rate to implement a fractionally spaced
equalizer at the mobile as will be appreciated by one of ordinary
skill in the art. However, analysis of the STTD decoder for a
fractionally spaced equalizer and multi-user detector is the same
as for the exemplary chip rate sampling. A combined channel
response for antennas 1 and 2 is given by equations [21] and [22],
respectively.
U.sup.k=(u.sub.1.sup.k,u.sub.2.sup.k, . . .
,u.sub.G+W-1.sup.k)=C.sup.kH.s- ub.1 [21]
V.sup.k=(v.sub.1.sup.k,v.sub.2.sup.k, . . .
,v.sub.G+W-1.sup.k)=C.sup.kH.s- ub.2 [22]
[0052] A composite data symbol vector for a block of M symbols from
both antennas is produced by 242 at transmit path 244 as in
equation [23].
{overscore (D)}=((D.sup.1).sup.T,(D.sup.2).sup.T, . . .
,(D.sup.k).sup.T)=(d.sub.1.sup.1,d.sub.2.sup.1, . . . ,
d.sub.M.sup.1, d.sub.1.sup.2, . . . , d.sub.M.sup.2, . . . ,
d.sub.1.sup.k, d.sub.2.sup.k, . . . , d.sub.M.sup.k, . . . ,
d.sub.1.sup.k, d.sub.2.sup.k, . . . , d.sub.M.sup.k) [23]
[0053] Additive Gaussian noise {overscore (N)} at the sampled at
the chip rate is added at 246 as in equation [24] to produce a
composite signal at the mobile receiver antenna 250.
{overscore (N)}=(n.sub.1,n.sub.2, . . . ,n.sub.M*G+W-1).sup.T
[24]
[0054] This received sequence {overscore (R)} at 250 sampled at the
chip rate is of length (MG+W-1) and it is the sum of the signals
from the two antennas and the additive Gaussian noise given by
equation [25]. 4 R _ = A D _ + B D _ * + N _ = [ A B ] [ D _ D _ *
] + N _ [ 25 ]
[0055] Elements of the matrices A=(A.sub.ij) and B=(B.sub.ij) are
given by equations [26-30], where i=1,2, . . . M*G+W-1 and j=1,2, .
. . ,K*M. Elements of matrix B are given by equations [28] and [30]
for an even number of elements and by equations [29-30] for an odd
number of elements. 5 A G * ( m - 1 ) + l , m + M * ( k - 1 ) = { u
l k 2 for k = 1 , 2 , J , m = 1 , 2 , M , l = 1 , 2 , G + W - 1 u l
k for k = J + 1 , , K , m = 1 , 2 , M , l = 1 , 2 , G + W - 1 [ 26
] A G * ( m - 1 ) + l , m + M * ( k - 1 ) = 0 otherwise [ 27 ] B G
* ( m - 2 ) + l , m - 1 + M * ( k - 1 ) = v l k 2 for k = 1 , 2 , L
, m = 2 , 4 , 6 , 8 , M , l = 1 , 2 , G + W - 1 [ 28 ] B G * m + l
, m + 1 + M * ( k - 1 ) = - v l k 2 for k = 1 , 2 , L , m = 1 , 3 ,
5 , ( M - 1 ) , l = 1 , 2 , G + W - 1 [ 29 ] B G * ( m - 1 ) + l ,
m + M * ( k - 1 ) = 0 otherwise [ 30 ]
[0056] The structure of the matrix B occurs because of the STTD
encoding. This structure is substantially different from the prior
art. For example, Klein et al. and Naguib et al. teach a structure
corresponding to equation [25] with matrix B equal to zero. These
formulations of the prior art work in the absence of a multi-path
channel. In the presence of a multi-path channel, however, the
structure of equation [25] cannot cancel either inter-symbol
interference (ISI) or multiple access interference (MAI). This is
only accomplished by including the received signal matrix
{overscore (R)} together with the complex conjugate matrix
{overscore (R)}.sup.* to remove both ISI and MAI. This structure is
highly advantageous in the joint detector design of the present
invention. The structure of the matrix B, represented in equation
[25], therefore, is rewritten in conjugate form in equation [31]. 6
[ R _ R _ * ] = [ A B B * A * ] [ D _ D _ * ] + [ N _ N _ * ] [ 31
]
[0057] Even though the equations for {overscore (R)}* are related
to the equations for {overscore (R)} in equation [25], the
equations [31] are linearly independent if the original equations
[25] are linearly independent. Thus, conjugate matrices are
rewritten as in equations [32] and equation [31] is rewritten as
equation [33]. 7 R ~ = [ R _ R _ * ] , A ~ = [ A B B * A * ] , D ~
= [ D _ D _ * ] and N ~ = [ N _ N _ * ] [ 32 ] {tilde over
(R)}={tilde over (D)}+ [33]
[0058] Turning now to FIG. 9A, there is a block diagram of another
embodiment of interference cancellation circuit of the present
invention with an STTD decoder and a zero forcing STTD equalizer.
The received signal {tilde over (R)} of equation [33] is applied to
via lead 900 to a whitening matched filter 902. This whitening
matched filter includes the multiple finger matched filters 700-704
and their respective sampling STTD decoders 706-710 of FIG. 7. A
product of this received signal and the whitening matched filter is
applied to a zero forcing STTD equalizer circuit 904 to produce
data symbol matrix {overscore ({circumflex over (D)})} at lead 906.
The term inside the zero forcing STTD equalizer circuit 904 yields
a zero forcing solution to equation [33] without any intersymbol
interference (ISI) or multiple access interference (MAI) as given
in equation [34], where .zeta..sub., is the covariance of the noise
vector . and (.).sup.H denotes the Hermitian operation on a matrix.
8 [ D _ ^ D _ ^ * ] ZF - STTD = ( A ~ H N ~ , N ~ - 1 A ~ ) - 1 A ~
H N ~ , N ~ - 1 R ~ [ 34 ]
[0059] For the special case of .zeta..sub.,=.sigma..sup.2
I.sub.2*(M*G+W-1).times.2*(M*G+W-1) the ZF-STTD is given by
equation [35]. 9 [ D _ ^ D _ ^ * ] ZF - STTD = ( A ~ H A ~ ) - 1 A
~ H R ~ [ 35 ]
[0060] Since {overscore ({circumflex over (D)})} and {overscore
({circumflex over (D)})}.sup.* yield the same estimate for received
data symbols, it is only necessary to calculate one of them.
However, the intermediate steps that are involved that is,
calculating .sup.H{tilde over (R)} and the (.sup.H).sup.-1 have to
be performed completely. A Cholesky decomposition of the matrix
.sup.H.zeta..sub.,.sup.-1 is given by equation [36].
.sup.H.zeta..sub.,.sup.-1=(.SIGMA.H).sup.-1.SIGMA.H [36]
[0061] The term .SIGMA. is a diagonal matrix and H is an upper
triangular matrix. The Cholesky decomposition in equation [36]
greatly reduces the calculation complexity of equation [35] by
eliminating the term (.sup.H).sup.-1. The Cholesky formulation of
equation [36] provides a means for solving equation [35] using a
forward equation obtained from the upper triangular matrix H. The
detailed block diagram of FIG. 9B illustrates the iterative
solution to equation [34] of the zero forcing STTD equalizer with
decision feedback. Derivation and use of the feedback operator 924
is explained in detail Anja Klein et al. at 280.
[0062] Referring now to FIG. 10A, there is a block diagram of a
third embodiment of interference cancellation of the present
invention with an STTD decoder and a minimum mean squared error
STTD equalizer. For data covariance matrix .zeta..sub.{tilde over
(D)},{tilde over (D)}, the minimum mean squared error solution for
STTD decoding (MMSE-STTD) is given by equation [37]. 10 [ D _ ^ D _
^ * ] MMSE - STTD = ( A ~ H N ~ , N ~ - 1 A ~ + D ~ , D ~ - 1 ) - 1
A ~ H N ~ , N ~ - 1 R ~ [ 37 ]
[0063] For the special case of
.zeta..sub.,=.sigma..sup.2I.sub.2*(M*G+W-1)- .times.2*(M*G+W-1) and
.zeta..sub.{tilde over (D)},{tilde over
(D)}=I.sub.2*(M*G+W-1).times.2*(M*G+W-1) the MMSE-STTD decoder
solution is given by equation [38]. 11 [ D _ ^ D _ ^ * ] MMSE -
STTD = ( A ~ H A ~ + I ) - 1 A ~ H R ~ [ 38 ]
[0064] Again since the {overscore ({circumflex over (D)})} and
{overscore ({circumflex over (D)})}.sup.* yield the same estimate
for data, only one of them needs to be calculated in the end while
the intermediate steps have to calculated completely.
[0065] Cholesky decomposition of the matrix (.sup.H+I) given by
equation [39].
.sup.H+I=(.SIGMA.H).sup.-1.SIGMA.H [39]
[0066] The Cholesky decomposition in equation [39] reduces the
complexity of equation [38]. This is highly advantageous due to the
calculation complexity of the term (.sup.H+I). The Cholesky
formulation of equation [39] provides a means for solving equation
[38] using a forward equation obtained from the upper triangular
matrix H. The block diagram of FIG. 10B shows an iterative minimum
mean squared error STTD equalizer with decision feedback.
Derivation and use of the feedback operator 1024 is explained in
detail by Klein et al., Id at 281.
[0067] Referring now to FIG. 11, there is a simulation diagram
showing bit error rate (BER) as a function of bit energy to noise
(Eb/N.sub.0) with and without diversity for vehicular Doppler rates
with a spreading factor of 16. For an exemplary BER of 10.sup.-2
the zero forcing STTD receiver shows a 2.5 dB improvement over a
comparable receiver without STTD. The a simulation diagram of FIG.
12 shows bit error rate (BER) as a function of bit energy to noise
(Eb/N.sub.0) with and without diversity for pedestrian Doppler
rates with a spreading factor of 16 and 8 users. Both curves show
improved bit energy to noise ratios compared to the simulation of
FIG. 11 for the relatively higher vehicular Doppler rate. Moreover,
the STTD curve for a pedestrian Doppler rate shows a 3 dB
improvement over the solid curve without STTD. Thus, STTD for TDD
of the present invention provides significantly improved reception
over systems of the prior art.
[0068] Referring to FIG. 13A, there is a block diagram of a
receiver of the present invention including STTD decoders before
the rake receivers and joint detector. This circuit design is
similar to that of FIG. 7. The circuit provides STTD decoder
circuits 1302-1304 corresponding to respective multipath signals.
Each STTD decoder produces plural output signals that are coupled
to respective rake receivers to combine multipath signals for each
respective user. The combined signals are then applied to joint
STTD detector circuit 1310. The joint detector circuit utilizes
detected signals for other users to eliminate interference from the
intended user signal as previously described. The circuit of FIG.
13B is an alternative embodiment of the present invention. This
embodiment includes rake receivers 1312-1314 arranged to combine
multipath signals for each respective user. These combined signals
are then applied to the combined joint detector and STTD decoder
circuit 1316. The joint detector 1316 decodes the received signals
for each user and subtracts interference signals for unintended
users to produce the intended output signal D.sub.0 on lead 1320.
The circuit of FIG. 13C is yet another embodiment of the present
invention. This embodiment includes rake receivers 1312-1314 as
previously described. Combined signals from the rake receivers are
applied to the joint detector circuit 1318 for user identification
and interference cancellation. The resulting signal is applied to
STTD decoder 1319. The STTD decoder produces decoded output signal
D.sub.0 on lead 1320 for the intended user.
[0069] Although the invention has been described in detail with
reference to its preferred embodiment, it is to be understood that
this description is by way of example only and is not to be
construed in a limiting sense. For example, several variations in
the order of symbol transmission would provide the same 2L
diversity. Moreover, the exemplary diversity of the present
invention may be increased with a greater number of transmit or
receive antennas. Furthermore, novel concepts of the present
invention are not limited to exemplary circuitry, but may also be
realized by digital signal processing as will be appreciated by
those of ordinary skill in the art with access to the instant
specification. For example, an alternative embodiment of the
present invention having a spreading factor of one is equivalent to
a time division multiple access (TDMA) system. Thus, IS-136,
Enhanced Data GSM Environment (EDGE) and other cellular systems may
use the present invention with STTD encoded multi-path signals for
received channel equalization.
[0070] It is to be further understood that numerous changes in the
details of the embodiments of the invention will be apparent to
persons of ordinary skill in the art having reference to this
description. It is contemplated that such changes and additional
embodiments are within the spirit and true scope of the invention
as claimed below.
* * * * *