U.S. patent application number 10/345487 was filed with the patent office on 2003-07-31 for transmission system for reduction of amateur radio interference.
Invention is credited to Anderson, Carl William, Duxbury, Guy M.A.F..
Application Number | 20030142759 10/345487 |
Document ID | / |
Family ID | 22277670 |
Filed Date | 2003-07-31 |
United States Patent
Application |
20030142759 |
Kind Code |
A1 |
Anderson, Carl William ; et
al. |
July 31, 2003 |
Transmission system for reduction of amateur radio interference
Abstract
The invention is a novel method and transmission system for
mitigating the adverse effects of amateur (HAM) radio interference
on digital signals transmitted across a twisted pair and reducing
the interference caused by the digital signals to amateur radio
communications. At the heart of the invention is strategical
placement of the frequency spectrum of the transmitted signal
relative to the center frequency of the HAM bands. In particular,
the signal spectrum is permitted to straddle one or more HAM bands,
which permits the transport of high data rates with a single
carrier. At the transmitter, the signal is filtered with a
real-valued symmetric baseband filter in order to remove signal
content in the HAM bands. At the receiver, the signal undergoes
adaptive interference cancellation, followed by processing through
a decision feedback equalizer. In some cases, the signal is
filtered at baseband with a real-valued filter to remove
interference in the HAM bands. In the context of a
telecommunications access system, the inventive technique can be
applied to either or both directions of traffic flow, i.e.,
downstream and upstream, and multiple carriers can be added to
increase capacity in any given direction.
Inventors: |
Anderson, Carl William;
(Kanata, CA) ; Duxbury, Guy M.A.F.; (Nepean,
CA) |
Correspondence
Address: |
SMART & BIGGAR
Suite 3400
1000 de la Gauchetiere Street West
Montreal
QC
H3B 4W5
CA
|
Family ID: |
22277670 |
Appl. No.: |
10/345487 |
Filed: |
January 17, 2003 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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10345487 |
Jan 17, 2003 |
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09100010 |
Jun 19, 1998 |
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6522699 |
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Current U.S.
Class: |
375/298 ;
375/261; 375/350 |
Current CPC
Class: |
H04L 27/2647 20130101;
H04L 2025/03414 20130101; H04L 2025/0349 20130101; H04B 3/30
20130101; H04L 25/03057 20130101 |
Class at
Publication: |
375/298 ;
375/350; 375/261 |
International
Class: |
H04L 005/12; H04L
023/02; H04L 027/36 |
Claims
We claim:
1. A transmitter for transmitting a modulated signal across a
transmission medium, the transmitter comprising: an encoder for
encoding a digital data stream into one or more encoded digital
signals; one or more substantially identical baseband notching
filters connected to the encoder, for respectively receiving the
one or more encoded digital signals, each baseband notching filter
having a notch at zero frequency; and a modulator connected to the
one or more baseband notching filters, for producing the modulated
signal centered about a carrier frequency, wherein the carrier
frequency is approximately equal to the center frequency of an
interference band.
2. A transmitter according to claim 1, further comprising one or
more substantially identical Nyquist filters respectively connected
between the one or more notching filters and the modulator.
3. A transmitter according to claim 1, wherein the baseband
notching filters have real-valued coefficients.
4. A transmitter according to claim 1, wherein the encoder is a
quadrature amplitude modulation (QAM) encoder producing two encoded
signals, of which one is an in-phase encoded signal and the other
is an in-quadrature encoded signal, and wherein the modulator is a
quadrature modulator.
5. A transmitter according to claim 1, wherein the interference
band is an amateur radio band having a center frequency
approximately equal to one of 1.9, 3.75, 7.15, 10.125, 14.175,
18.118 and 21.225 Megahertz.
6. A transmitter according to claim 1, wherein the interference
band has a bandwidth of less than or approximately equal to 10% of
the carrier frequency.
7. A transmitter for transmitting a modulated signal across a
transmission medium, the transmitter comprising: an encoder for
encoding a digital data stream into one or more encoded digital
signals having a symbol rate f.sub.S; one or more substantially
identical baseband notching filters connected to the encoder, for
respectively receiving the one or more encoded digital signals,
each baseband notching filter having a pair of notches
symmetrically disposed about zero frequency at frequencies
approximately equal to .+-.f.sub.S/4; and a modulator connected to
the one or more baseband notching filters, for producing the
modulated signal centered about a carrier frequency f.sub.C;
wherein f.sub.S and f.sub.C are related to the center frequencies
f.sub.H1 and f.sub.H2 of two interference bands by f.sub.C being
approximately equal to 1/2(f.sub.H1+f.sub.H2) and f.sub.S being
approximately equal to 2.vertline.f.sub.H2-f.sub.H1.vertline..
8. A transmitter according to claim 7, further comprising one or
more substantially identical Nyquist filters respectively connected
between the one or more notching filters and the modulator.
9. A transmitter according to claim 7, wherein the baseband
notching filters have real-valued coefficients.
10. A transmitter according to claim 7, wherein the encoder is a
quadrature amplitude modulation (QAM) encoder producing two encoded
signals, of which one is an in-phase encoded signal and the other
is an in-quadrature encoded signal, and wherein the modulator is a
quadrature modulator.
11. A transmitter according to claim 7, wherein the interference
bands are adjacent amateur radio bands, having corresponding center
frequencies f.sub.H1 and f.sub.H2 chosen from the set of 1.9, 3.75,
7.15, 10.125, 14.175, 18.118 and 21.225 Megahertz,
approximately.
12. A transmitter according to claim 7, wherein the interference
bands have bandwidths of less than or approximately equal to
0.1.times.f.sub.C.
13. A receiver for extracting a digital data stream from a
modulated signal, said modulated signal being centered about a
carrier frequency, the receiver comprising: adaptive interference
cancellation (AIC) means for controllably reducing narrowband
interference present in the modulated signal around the carrier
frequency, thereby to produce an interference-reduced signal; a
demodulator connected to the AIC means for receiving and
demodulating the interference-reduced signal, thereby to produce
one or more baseband demodulated signals; a power estimator
connected to the demodulator and to the AIC means, for receiving
the one or more baseband demodulated signals, calculating the power
of residual interference present in the one or more baseband
demodulated signals around the carrier frequency and providing the
AIC means with an interference estimate signal; one or more
substantially identical baseband notching filters connected to the
demodulator, for receiving the one or more baseband demodulated
signals and producing respective filtered demodulated signals, each
baseband notching filter having a notch at zero frequency; and a
decision-feedback equalizer (DFE) connected to the one or more
baseband notching filters, for receiving the filtered demodulated
signals, decoding digital data embedded therein and producing the
digital data stream.
14. A receiver according to claim 13, wherein the baseband notching
filters have real-valued coefficients.
15. A receiver according to claim 13, further comprising one or
more Nyquist filters respectively connected between the demodulator
and the baseband notching filters.
16. A receiver according to claim 13, further comprising a
variable-gain amplifier connected between the AIC means and the
demodulator, for maintaining a substantially constant signal level
of the interference-reduced signal.
17. A receiver according to claim 13, wherein the demodulator is a
quadrature demodulator and wherein the number of baseband
demodulated signals is two, one of which is an in-phase baseband
demodulated signal and the other of which is an in-quadrature
baseband demodulated signal.
18. A receiver for extracting a digital data stream from a
modulated signal, said modulated signal being centered about a
carrier frequency f.sub.C and having a bandwidth f.sub.S, the
receiver comprising: adaptive interference cancellation (AIC) means
for controllably reducing narrowband interference present in the
modulated signal around frequencies f.sub.C.+-.f.sub.S/4, thereby
to produce an interference-reduced signal; a demodulator connected
to the AIC means for receiving and demodulating the
interference-reduced signal, thereby to produce one or more
baseband demodulated signals; a power estimator connected to the
demodulator and to the AIC means, for receiving the one or more
baseband demodulated signals and providing the AIC means with an
estimate of residual interference in the one or more baseband
demodulated signals around frequencies f.sub.C.+-.f.sub.S/4; one or
more substantially identical baseband notching filters connected to
the demodulator, for receiving the one or more baseband demodulated
signals and producing respective filtered demodulated signals, each
baseband notching filter having a pair of notches symmetrically
disposed about zero frequency at frequencies approximately equal to
.+-.f.sub.S/4; and a decision-feedback equalizer (DFE) connected to
the one or more baseband notching filters, for receiving the
filtered demodulated signals, decoding digital data embedded
therein and producing the digital data stream.
19. A receiver according to claim 18, wherein the baseband notching
filters have real-valued coefficients.
20. A receiver according to claim 18, further comprising one or
more Nyquist filters respectively connected between the demodulator
and the baseband notching filters.
21. A receiver according to claim 18, further comprising a
variable-gain amplifier connected between the AIC means and the
demodulator, for maintaining a substantially constant signal level
of the interference-reduced signal.
22. A receiver according to claim 18, wherein the demodulator is a
quadrature demodulator.
23. A receiver for extracting a digital data stream from a
modulated signal, said modulated signal being centered about a
carrier frequency, the receiver comprising: adaptive interference
cancellation (AIC) means for controllably reducing narrowband
interference present in the modulated signal around the carrier
frequency in accordance with first and second residual interference
estimates, thereby to produce an interference-reduced signal; a
demodulator connected to the AIC means for receiving and
demodulating the interference-reduced signal, thereby to produce
one or more baseband demodulated signals, and for providing the AIC
means with the first estimate of residual interference in the one
or more baseband demodulated signals around the carrier frequency;
one or more substantially identical baseband notching filters
connected to the demodulator, for receiving the one or more
baseband demodulated signals and producing respective filtered
demodulated signals, each baseband notching filter having a notch
at zero frequency; and a decision-feedback equalizer (DFE)
connected to the one or more baseband notching filters, for
receiving the filtered demodulated signals, decoding digital data
embedded therein and producing the digital data stream, wherein the
DFE is further connected to the AIC means for providing the AIC
means with the second estimate of residual interference in the
filtered demodulated signals around the carrier frequency;
24. A receiver according to claim 23, wherein the baseband notching
filters have real-valued coefficients.
25. A receiver according to claim 23, further comprising one or
more Nyquist filters respectively connected between the demodulator
and the baseband notching filters.
26. A receiver according to claim 23, further comprising a
variable-gain amplifier connected between the AIC means and the
demodulator, for maintaining a substantially constant signal level
of the interference-reduced signal.
27. A receiver according to claim 23, wherein the demodulator is a
quadrature demodulator.
28. A method of transmitting digital data, comprising the steps of:
encoding the data into one or more baseband digital signals;
filtering the one or more baseband digital signals with respective
baseband filters having notches at zero frequency; modulating the
filtered baseband digital signals about a carrier frequency,
thereby to produce a modulated signal; and transmitting the
modulated signal across a transmission medium; wherein the carrier
frequency is approximately equal to the center frequency of an
interference band.
29. A method according to claim 28, wherein the center frequency of
the interference band is approximately equal to one of 1.9, 3.75,
7.15, 10.125, 14.175, 18.118 and 21.225 Megahertz.
30. A method according to claim 28, wherein the encoding step
further comprises further encoding the data in accordance with a
partial response filter having a null at zero frequency.
31. A method of transmitting digital data, comprising the steps of:
encoding the data into one or more baseband digital signals;
filtering the one or more baseband digital signals with respective
baseband filters having notches at frequencies approximately equal
to .+-..DELTA.f; modulating the filtered baseband digital signals
about a carrier frequency f.sub.C, thereby to produce a modulated
signal; and transmitting the modulated signal across a transmission
medium; wherein f.sub.C+.DELTA.f is approximately equal to the
center frequency of a first interference band and f.sub.C-.DELTA.f
is approximately equal to the center frequency of a second
interference band.
32. A method according to claim 31, the baseband digital signals
having a common symbol rate, wherein .DELTA.f is approximately
equal to one-quarter the symbol rate.
33. A method according to claim 31, wherein the first and second
interference bands are adjacent, and wherein the center frequencies
of the first and second interference bands are chosen from the set
of 1.9, 3.75, 7.15, 10.125, 14.175, 18.118 and 21.225 Megahertz,
approximately.
34. A method of recovering a digital data stream from a received
signal modulated about a carrier frequency, comprising:
demodulating the modulated signal, thereby to produce one or more
baseband demodulated signals; filtering the one or more baseband
demodulated signals with respective baseband filters having notches
at zero frequency; and decoding the filtered baseband demodulated
signals, thereby to recover the digital data; wherein the carrier
frequency is approximately equal to the center frequency of an
interference band.
35. A method according to claim 34, wherein the center frequency of
the interference band is approximately equal to one of 1.9, 3.75,
7.15, 10.125, 14.175, 18.118 and 21.225 Megahertz.
36. A method of recovering a digital data stream from a received
signal modulated about a carrier frequency f.sub.C, comprising:
demodulating the modulated signal, thereby to produce one or more
baseband demodulated signals; filtering the one or more baseband
demodulated signals with respective baseband filters having notches
at frequencies .+-..DELTA.f; and decoding the filtered baseband
demodulated signals, thereby to recover the digital data; wherein
f.sub.C.+-..DELTA.f is approximately equal to the center frequency
of a first interference band and f.sub.C-.DELTA.f is approximately
equal to the center frequency of a second interference band.
37. A method according to claim 36, wherein the first and second
interference bands are adjacent, and wherein the center frequencies
of the first and second interference bands are chosen from the set
of 1.9, 3.75, 7.15, 10.125, 14.175, 18.118 and 21.225 Megahertz,
approximately.
38. A modem for transmitting a first modulated signal onto a
twisted pair and for receiving a second modulated signal from the
twisted pair, comprising: a hybrid for interfacing with the twisted
pair; a transmitter connected to the hybrid, for producing the
first modulated signal from a first digital data stream, the
transmitter comprising an encoder for encoding the first digital
data stream into one or more encoded digital signals; one or more
substantially identical baseband notching filters connected to the
encoder, for respectively receiving the one or more encoded digital
signals, each baseband notching filter having a notch at zero
frequency; and a modulator connected to the one or more baseband
notching filters, for producing the first modulated signal centered
about a carrier frequency, wherein the carrier frequency is
approximately equal to the center frequency of an interference
band; and a receiver connected to the hybrid, for extracting a
digital data stream from the second modulated signal, the receiver
comprising adaptive interference cancellation (AIC) means for
controllably reducing narrow band interference present in the
second modulated signal around the carrier frequency, thereby to
produce an interference-reduced signal; a demodulator connected to
the AIC means for receiving and demodulating the
interference-reduced signal, thereby to produce one or more
baseband demodulated signals; a power estimator connected to the
demodulator and to the AIC means, for receiving the one or more
baseband demodulated signals and providing the AIC means with an
estimate of residual interference in the one or more baseband
demodulated signals around the carrier frequency; one or more
substantially identical baseband notching filters connected to the
demodulator, for receiving the one or more baseband demodulated
signals and producing respective filtered demodulated signals, each
baseband notching filter having a notch at zero frequency; and a
decision-feedback equalizer (DFE) connected to the one or more
baseband notching filters, for receiving the filtered demodulated
signals, decoding digital data embedded therein and producing the
second digital data stream.
39. A modem according to claim 38 wherein the twisted pair carries
a differential signal.
40. A modem according to claim 38, wherein the center frequency of
the interference band is approximately equal to one of 1.9, 3.75,
7.15, 10.125, 14.175, 18.118 and 21.225 Megahertz.
41. A method of allocating a frequency spectrum, comprising:
selecting a first portion of the frequency spectrum for
transmission of a first signal, said first portion having a first
center frequency f.sub.C1; wherein f.sub.C1 is related to the
center frequency f.sub.H1 of an interference band by f.sub.C1 being
approximately equal to f.sub.H1.
42. A method according to claim 41, further comprising: selecting a
second portion of the frequency spectrum for transmission of a
second signal, said second portion having a second center frequency
f.sub.C2; wherein f.sub.C2 is related to the center frequency
f.sub.H2 of an interference band by f.sub.C being approximately
equal to f.sub.H2.
43. A method according to claim 41, further comprising: selecting a
second portion of the frequency spectrum for transmission of a
second signal, said second portion having a second center frequency
f.sub.C2 and a bandwidth f.sub.S; wherein f.sub.S and f.sub.C2 are
related to the center frequencies f.sub.H2 and f.sub.H3 of two
interference bands by f.sub.C2 being approximately equal to
1/2(f.sub.H2+f.sub.H3) and f.sub.S being approximately equal to
2.vertline.f.sub.H3-f.sub.H2.vertline..
44. A method of allocating a frequency spectrum, comprising:
selecting a first portion of the frequency spectrum for
transmission of a first signal, said first portion having a center
frequency f.sub.C1 and a bandwidth f.sub.S1; wherein f.sub.S1 and
f.sub.C1 are related to the center frequencies f.sub.H1 and
f.sub.H2 of two interference bands by f.sub.C1 being approximately
equal to 1/2(f.sub.H1+f.sub.H2) and f.sub.S1 being approximately
equal to 2.vertline.f.sub.H2-f.sub.H1.vertline..
45. A method according to claim 44, further comprising: selecting a
second portion of the frequency spectrum for transmission of a
second signal, said second portion having a second center frequency
f.sub.C2 and a bandwidth f.sub.S2; wherein f.sub.S2 and f.sub.C2
are related to the center frequencies f.sub.H3 and f.sub.H4 of two
interference bands by f.sub.C2 being approximately equal to
1/2(f.sub.H3+f.sub.H4) and f.sub.S2 being approximately equal to 2
.vertline.f.sub.H4-f.sub.H3.vertline..
Description
FIELD OF THE INVENTION
[0001] This invention relates to the transmission and reception of
digital signals along subscriber telephone loops and is partially
concerned with mitigating the adverse effects of the digital
signals interfering with amateur radio transmissions and vice
versa.
BACKGROUND OF THE INVENTION
[0002] In an access system for use in a fiber-to-the-neighbourhood
(FTTN) network, digital data is exchanged between a host digital
terminal (HDT) and a plurality of optical network units (ONUs) via
optical fibers. Each ONU is responsible for exchanging downstream
(ONU-to-subscriber) and upstream (subscriber-to-ONU) data with a
respective plurality of subscribers via copper twisted pairs.
Typically, the downstream and upstream data are modulated about
separate carrier frequencies and occupy spectral bands which have a
width proportional to the transmitted data rate. For example, a
data stream at 20 Megabits per second, modulated using 16-QAM
(quadrature amplitude modulation with 16 constellation points,
i.e., 4 bits per symbol), can be transported by a spectral band
having a width of 5 MHz and being centered about a given carrier
frequency.
[0003] Two major factors affecting the performance of a system
which transmits data along a twisted pair are loop attenuation and
crosstalk. In order to combat these two elements, it is commonplace
to use a lower-order modulation format, such as 16-QAM, and to
restrict the frequency range of the upstream and downstream data to
approximately between 1 MHz and 20 MHz.
[0004] Unfortunately, this frequency range is also subject to
interference from amateur radio communications carried by HAM bands
commonly known as the 160-, 80-, 40-, 30-, 20-, 17- and 15-meter
bands, whose spectral characteristics are illustrated in the
following table:
1 HAM band Frequency Center frequency (meters) range (MHz) (MHz)
160 1.800-2.000 1.900 80 3.500-4.000 3.750 40 7.000-7.300 7.150 30
10.100-10.150 10.125 20 14.000-14.350 14.175 17 18.068-18.168
18.118 15 21.000-21.450 21.225
[0005] Clearly, a downstream or upstream spectrum of any
considerable width (i.e., data rate) will straddle at least one HAM
band. By way of example, FIG. 1 shows a signal spectrum 11 having a
center frequency f.sub.C of approximately 9.0 MHz and a bandwidth
f.sub.S of approximately 5 MHz. The signal spectrum 11 is seen to
span two HAM bands, specifically bands 14 and 13 (centered at 7.150
MHz and 10.125 MHz, respectively).
[0006] Due to various regulations governing electromagnetic
interference, it is imperative that a signal transmitted along the
twisted pair contain reduced signal energy in any HAM band so as
not to interfere with HAM radio communications occurring in those
bands. This requirement is somewhat in conflict with the goal of
transmitting high data rates, and has forced telecommunications
companies to provide ways of reducing or eliminating the signal
energy in these bands, while still delivering the high data rates
demanded by today's customers.
[0007] In one prior art approach, the total downstream (or
upstream) data rate is handled by transporting the data using
multiple separate carriers, which allows placement of the
individual spectra between HAM bands, thereby avoiding any
intersections of the signal spectra with HAM bands. However, this
technique increases the complexity of the transmitter and receiver,
as both must now be equipped to deal with multiple parallel
modulations or demodulations.
[0008] Another prior art approach acknowledges that the transmitted
signal spectrum will straddle one or more HAM bands, and the signal
to be transmitted is passed through a series of digital notch
filters in the transmitter, each filter notch being centered about
a HAM band. These filters may operate at passband or at baseband.
If a passband filter (or series of filters) is used, then the
sharpness required of the filter (or filters) is very high,
necessitating the use of a large number of taps. On the other hand,
if baseband filters are employed, the notches are generally
asymmetrically disposed about zero frequency, requiring the use of
complex coefficients. Either solution leads to a relatively high
complexity for the transmitter.
[0009] Even if notching at the transmitter were able to prevent the
signal spectrum from interfering with the HAM bands, it cannot
control the onset of interference due to transmissions occurring in
the HAM bands themselves. That is to say, the receiver will
necessarily accept a signal whose spectrum has undergone
radio-frequency interference due to HAM transmissions. In order to
combat this effect, a conventional approach applies band-pass and
notch filtering at the receiver in order to pass only those
components of the signal spectrum lying outside the HAM bands.
Again, regardless of whether filtering is performed at passband or
at baseband, the complexity of the required filters is relatively
high.
[0010] Finally, the art has seen the development of a technique
known as DMT (discrete multi-tone), which relies on the
transmission of a large number of carriers generated by inverse
Fourier transform techniques, each carrier being associated with a
small amount of the overall required bandwidth. Radio-frequency
interference is sidestepped in the DMT system simply by employing
only those carriers which do not overlap with the HAM bands.
However, the processing requirements of an actual implementation of
DMT are often too demanding to permit cost-effective use of this
technique.
SUMMARY OF THE INVENTION
[0011] It is an object of the present invention to mitigate or
obviate one or more disadvantages of the prior art.
[0012] Therefore, the invention may be summarized according to a
first broad aspect as a transmitter for transmitting a modulated
signal across a transmission medium, the transmitter comprising: an
encoder for encoding a digital data stream into one or more encoded
digital signals; one or more substantially identical baseband
notching filters connected to the encoder, for respectively
receiving the one or more encoded digital signals, each baseband
notching filter having a notch at zero frequency; and a modulator
connected to the one or more baseband notching filters, for
producing the modulated signal centered about a carrier frequency,
wherein the carrier frequency is approximately equal to the center
frequency of an interference band.
[0013] According to a second broad aspect, the present invention
may be summarized as a receiver for extracting a digital data
stream from a modulated signal, said modulated signal being
centered about a carrier frequency, the receiver comprising:
adaptive interference cancellation (AIC) means for controllably
reducing narrowband interference present in the modulated signal
around the carrier frequency, thereby to produce an
interference-reduced signal; a demodulator connected to the AIC
means for receiving and demodulating the interference-reduced
signal, thereby to produce one or more baseband demodulated
signals; a power estimator connected to the demodulator and to the
AIC means, for receiving the one or more baseband demodulated
signals, calculating the power of residual interference present in
the one or more baseband demodulated signals around the carrier
frequency and providing the AIC means with an interference estimate
signal; one or more substantially identical baseband notching
filters connected to the demodulator, for receiving the one or more
baseband demodulated signals and producing respective filtered
demodulated signals, each baseband notching filter having a notch
at zero frequency; and a decision-feedback equalizer (DFE)
connected to the one or more baseband notching filters, for
receiving the filtered demodulated signals, decoding digital data
embedded therein and producing the digital data stream.
[0014] The invention can be summarized according to another broad
aspect as a method of transmitting digital data, comprising the
steps of: encoding the data into one or more baseband digital
signals; filtering the one or more baseband digital signals with
respective baseband filters having notches at zero frequency;
modulating the filtered baseband digital signals about a carrier
frequency, thereby to produce a modulated signal; and transmitting
the modulated signal across a transmission medium; wherein the
carrier frequency is approximately equal to the center frequency of
an interference band.
[0015] According to yet another broad aspect, the present invention
may be summarized as a method of recovering a digital data stream
from a received signal modulated about a carrier frequency,
comprising: demodulating the modulated signal, thereby to produce
one or more baseband demodulated signals; filtering the one or more
baseband demodulated signals with respective baseband filters
having notches at zero frequency; and decoding the filtered
baseband demodulated signals, thereby to recover the digital data;
wherein the carrier frequency is approximately equal to the center
frequency of an interference band.
[0016] According to still another broad aspect, the present
invention may be summarized as a modem for transmitting a first
modulated signal onto a twisted pair and for receiving a second
modulated signal from the twisted pair, comprising: a hybrid for
interfacing with the twisted pair; a transmitter connected to the
hybrid, for producing the first modulated signal from a first
digital data stream, the transmitter comprising an encoder for
encoding the first digital data stream into one or more encoded
digital signals; one or more substantially identical baseband
notching filters connected to the encoder, for respectively
receiving the one or more encoded digital signals, each baseband
notching filter having a notch at zero frequency; and a modulator
connected to the one or more baseband notching filters, for
producing the first modulated signal centered about a carrier
frequency, wherein the carrier frequency is approximately equal to
the center frequency of an interference band; and a receiver
connected to the hybrid, for extracting a digital data stream from
the second modulated signal, the receiver comprising adaptive
interference cancellation (AIC) means for controllably reducing
narrowband interference present in the second modulated signal
around the carrier frequency, thereby to produce an
interference-reduced signal; a demodulator connected to the AIC
means for receiving and demodulating the interference-reduced
signal, thereby to produce one or more baseband demodulated
signals; a power estimator connected to the demodulator and to the
AIC means, for receiving the one or more baseband demodulated
signals and providing the AIC means with an estimate of residual
interference in the one or more baseband demodulated signals around
the carrier frequency; one or more substantially identical baseband
notching filters connected to the demodulator, for receiving the
one or more baseband demodulated signals and producing respective
filtered demodulated signals, each baseband notching filter having
a notch at zero frequency; and a decision-feedback equalizer (DFE)
connected to the one or more baseband notching filters, for
receiving the filtered demodulated signals, decoding digital data
embedded therein and producing the second digital data stream.
[0017] The present invention may be summarized according to still
another broad aspect as a method of allocating a frequency
spectrum, comprising: selecting a first portion of the frequency
spectrum for transmission of a first signal, said first portion
having a first center frequency f.sub.C1; wherein f.sub.C1 is
related to the center frequency f.sub.H1 of an interference band by
f.sub.C1 being approximately equal to f.sub.H1.
[0018] According to a further broad aspect, the present invention
may be summarized as a method of allocating a frequency spectrum,
comprising: selecting a first portion of the frequency spectrum for
transmission of a first signal, said first portion having a center
frequency f.sub.C1 and a bandwidth f.sub.S1; wherein f.sub.S1 and
f.sub.C1 are related to the center frequencies f.sub.H1 and
f.sub.H2 of two interference bands by f.sub.C1 being approximately
equal to 1/2(f.sub.H1+f.sub.H2) and f.sub.S1 being approximately
equal to 2.vertline.f.sub.H2-f.sub.H1.vertline..
BRIEF DESCRIPTION OF THE DRAWINGS
[0019] The preferred embodiment of the present invention will now
be described with reference to the accompanying drawings, in
which:
[0020] FIG. 1 shows the placement of downstream and upstream signal
spectra relative to HAM bands in accordance with the prior art;
[0021] FIG. 2 is a block diagram of a telco modem connected to a
customer premises modem in accordance with the prior art and with
the present invention;
[0022] FIG. 3 shows the placement of downstream and upstream signal
spectra relative to HAM bands in accordance with a preferred
embodiment of the present invention;
[0023] FIG. 4 is a block diagram of a transmitter in accordance
with the preferred embodiment of the present invention;
[0024] FIG. 5A is a block diagram of a receiver in accordance with
the preferred embodiment of the present invention;
[0025] FIG. 5B is a block diagram of an SSB down converter for use
in the receiver of FIG. 5A;
[0026] FIG. 6 shows the placement of downstream and upstream signal
spectra relative to HAM bands in accordance with an alternate
embodiment of the present invention;
[0027] FIG. 7 is a block diagram of a transmitter in accordance
with an alternate embodiment of the present invention;
[0028] FIG. 8 is a block diagram of a receiver in accordance with
an alternate embodiment of the present invention; and
[0029] FIG. 9 is a block diagram of a receiver in accordance with
yet another alternate embodiment of the present invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
[0030] The present invention can best be understood with the aid of
FIG. 2, showing a block diagram of part of a telecommunications
access system, in which a telco modem 21 exchanges downstream and
upstream data with a customer premises modem 22 across a copper
twisted pair 23. The telco modem 21 may be located within a line
card at an optical network unit, which may exchange data with a
host digital terminal (not shown). The generic structure
illustrated in FIG. 2 is used both in the prior art and in the
present invention.
[0031] At the telco modem 21, a downstream digital data stream 24
destined for the customer is modulated in accordance with a
modulation format (e.g., 16-QAM) by a transmitter 25, producing a
transmitted modulated downstream signal 26 that is subsequently
output onto the twisted pair 23 by a hybrid 27. The hybrid 27 also
serves to separate a received modulated upstream signal 28 arriving
on the twisted pair 23, outputting it to a receiver 29 which
demodulates the received modulated upstream signal 28 and decodes
an upstream digital data stream 30 embedded therein.
[0032] Similarly, the customer premises modem 22 also comprises a
hybrid 31, used for extracting a received modulated downstream
signal 32 arriving on the twisted pair 23 and equally for sending a
transmitted modulated upstream signal 33 towards the telco modem
21. The received modulated downstream signal 32 travels to a
receiver 34, where it is demodulated and from which an embedded
downstream digital data stream 35 is recovered. The transmitted
modulated upstream signal 33 is produced by a transmitter 36 which
modulates a customer-generated upstream digital data stream 37
according to a suitable modulation format, such as 16-QAM.
[0033] Although the preferred modulation format is indeed QAM
having 16 constellation points, it is equally suitable to utilize
other modulation schemes, such as carrierless amplitude and phase
(CAP) modulation and binary (BPSK) and quadrature phase-shift
keying (QPSK). Furthermore, the use of different orders of
modulation is equally suitable, simply resulting in a different
ratio of the number of bits carried per symbol.
[0034] As already discussed, depending on the data rate
requirements in either direction of traffic flow, the transmitted
modulated downstream signal 26 and the transmitted modulated
upstream signal 33 may each straddle one or more amateur radio
bands. For example, if the required downstream data rate is
approximately 20 Mbps, and if the transmitter 25 employs 16-QAM,
then the resulting downstream spectrum will have a 3 dB width of 5
MHz. This spectral width necessarily straddles 2 HAM bands, which
leads to the aforementioned tradeoff between interference reduction
and modem complexity. However, if the carrier frequency and data
rate of the transmitted modulated downstream signal 26 are chosen
in accordance with the present invention, performance can be
improved while minimizing modem complexity.
[0035] Specifically, FIG. 3 shows a spectral allocation scheme in
accordance with the present invention, in which the spectrum of the
transmitted downstream modulated signal 26 (the "downstream
spectrum", 41) is placed such that the carrier frequency f.sub.C
falls substantially midway between HAM bands 13 and 14, more
precisely at 8.6375 MHz. This contrasts with the spectral
allocation scheme of FIG. 1, in which the downstream spectrum 11
was not strategically placed with respect to the HAM bands. Another
difference is that the downstream spectrum 41 has a spectral width
f.sub.S substantially equal to twice the difference between the
center frequencies of the interfering HAM bands, i.e., 5.95
MHz.
[0036] More generally, the present invention requires that the
carrier frequency f.sub.C and symbol rate f.sub.S obey the
following mathematical relationship:
f.sub.C=1/2(f.sub.H1+f.sub.H2)
f.sub.S=2*.vertline.f.sub.H2-f.sub.H1.vertline.
[0037] where f.sub.H1 and f.sub.H2 are the center frequencies of
any two adjacent narrow interference bands, in this case HAM bands.
From the above, it follows that f.sub.H1=f.sub.C-f.sub.S/4 and
f.sub.H2=f.sub.C-- f.sub.S/4. For advantageous operation of the
present invention, it is not required that the equalities be exact;
rather a 10% error margin in the above equations still provides
sufficiently acceptable operation of the invention.
[0038] It is emphasized that there is a strong correspondence
between the symbol rate f.sub.S and center frequencies
f.sub.H1,f.sub.H2 of the interfering HAM bands. In fact, for each
pair of HAM bands, there is a preferred symbol rate f.sub.S as
calculated above, which may or may not be equal to a standardized
downstream or upstream bandwidth. In the example considered in FIG.
3, a signal having the characteristics of the downstream spectrum
41 will deliver a bit rate of 4*5.95=23.8 Mbps if each symbol
contains 4 bits of information (as is the case in 16-QAM). This
shows that the present invention is capable of delivering a high
data rate using a single QAM carrier, i.e., with a simple
modem.
[0039] FIG. 3 also shows placement of the spectrum of the
transmitted modulated upstream signal 33 ("the upstream spectrum",
44) in such a way that it is nestled between the 80-meter and
160-meter HAM bands 15,16. Assuming that the upstream spectrum 44
carries data that is modulated using 16-QAM, and allowing for an
excess bandwidth of 20% for practical Nyquist filters and an
additional 25% guard band for frequency division duplexing (FDD)
filters, the maximum delivered data rate for this particular
positioning of the upstream spectrum will be on the order of
4*(3.5-2).div.(1.2.times.1.5)=4.0 Mbps, which is usually sufficient
for upstream applications. It is to be understood that any higher
upstream bandwidth demands can be met by placing the upstream
spectrum 44 so that it straddles two HAM bands in the manner of
downstream spectrum 41.
[0040] Moreover, the present invention does not exclude the
possibility of transporting downstream or upstream traffic using
multiple carriers in either or each direction. Added carriers may
indeed be used to increase the capacity of a system already having
downstream and/or upstream spectra whose respective carrier
frequencies and symbol rates are subject to the above mathematical
relationships.
[0041] Inventive placement of the downstream and/or upstream
spectra allows drastic simplifications in the corresponding
transmitter and receiver, which are now structurally and
functionally described. Without loss of generality, it is assumed
that the transmitter 25 in the telco modem 21 of FIG. 2 sends a
transmitted modulated downstream signal 26 having a carrier
frequency f.sub.C and a symbol rate f.sub.S obeying
f.sub.C=(f.sub.H1+f.sub.H2)/2 and
f.sub.S=2*.vertline.f.sub.H2-f.sub.H1.v- ertline. for a pair of
adjacent narrowband interferers having respective center
frequencies f.sub.H1 and f.sub.H2.
[0042] With reference to FIG. 4, there is shown a transmitter 25 in
accordance with the preferred embodiment of the present invention,
comprising a QAM encoder 251 for accepting and encoding the
downstream digital data stream 24. The QAM encoder 251 produces a
pair of digital signals 258A,B which lead to a respective pair of
substantially identical digital notch (or "band elimination")
filters 252A,B. The filters 252A,B accept the digital signals
258A,B and produce respective filtered signals 259A,B that are fed
to respective Nyquist filters 253A,B prior to entering a quadrature
modulator 254. The quadrature modulator 254 then modulates the
signals 257A,B onto a carrier, producing the transmitted downstream
modulated signal 26, which is fed to the hybrid 27 and relayed in
differential mode across the twisted pair of wires 23A,B.
[0043] In operation, the downstream digital data stream 24 enters
the QAM encoder 251 at the rate of 4*f.sub.S (for a 16-QAM system).
The transmitter 25 is designed to have the appropriate carrier
frequency f.sub.C for the QAM encoder 251 according to the
above-stated mathematical relationship. The QAM encoder 251 then
produces, at the symbol rate f.sub.S, the two digital signals
258A,B which are known in the modulation art as the baseband
"in-phase" and "in-quadrature" signals.
[0044] The digital notch filters 252A,B operate at baseband and
therefore have coefficients that are complex numbers in general.
The reason for this is that the conventional role of the notch
filters is to provide notching around the two HAM frequencies which
are, in the general case, asymmetrically placed about the carrier
f.sub.C at passband, or about DC at baseband.
[0045] However, in the present invention, the desired notches are
located symmetrically about the carrier frequency at passband, or
on either side of zero frequency at baseband, i.e., which makes it
feasible to use real numbers for the taps of the filters 252A,B.
Furthermore, since digital filters have a natural tendency to notch
(or "dip") at fractional multiples of the sampling (or symbol) rate
f.sub.S, the desired placement of a notch at f.sub.S/4
advantageously leads to simpler, i.e., shorter, notch filters
253A,B.
[0046] One side-effect of producing a symmetrically placed notch
(using real-valued coefficients for the notch filters 252A,B) is
that the width of the notch on either side of zero frequency cannot
be independently controlled. Since the notch at the "positive"
frequency covers the HAM band centered about f.sub.H2 and the notch
at the "negative" frequency depletes the signal in the HAM band
surrounding f.sub.H1, the required notch width will be different
for each notch. In order to achieve satisfactory performance,
therefore, it is preferable to design the filters 252A,B so that
they apply a baseband notch which is at least as wide as the wider
interfering HAM band.
[0047] The path through the inventive transmitter 25 is completed
by the quadrature modulator 254, which accepts the in-phase and
in-quadrature outputs 257A,B of the Nyquist filters 253A,B and
creates the transmitted downstream modulated signal 26 therefrom.
The quadrature modulator 254 must have the appropriate carrier
frequency f.sub.C which, as stated, preferably lies mid-way between
the center frequencies of the two interfering HAM bands, i.e.,
f.sub.C=1/2(f.sub.H1+f.sub.H2)
[0048] The transmitted downstream modulated signal 26 then leaves
the transmitter 25, passes through the hybrid 27 and begins its
journey along the twisted pair 23A,B as a differential signal. If
f.sub.H1=7.15 MHz and f.sub.H2=10.125 MHz, then the downstream
spectrum of the transmitted downstream signal 26 will be positioned
as the downstream spectrum 41 in FIG. 3, although there will very
little signal content at (and around) the centers of HAM bands 13
and 14.
[0049] The above discussion of the inventive transmitter 25 has
emphasized the removal of signal content around two interfering HAM
bands. However, amateur radio transmissions occurring in these same
bands are capable of seriously corrupting the signal travelling
along the twisted pair, and the received modulated downstream
signal 32 may end up having a downstream spectrum characterized by
intermittent periods of strong frequency content centered about (or
located around) f.sub.H1 and f.sub.H2. This demands notch filtering
at the receiver 34 in the customer premises modem 22 which, as
previously discussed, generally results in increased modem
complexity. However, the present invention proposes a simplified
receiver for use in a system in which the transmitted (and
received) signal straddles two HAM bands in the above manner having
reference to FIG. 3.
[0050] According to the present invention, therefore, FIG. 5A shows
a receiver 34 for accepting the received modulated downstream
signal in the form of a twisted-pair signal 32A,B and a
differential signal 32C extracted therefrom by the hybrid 31. The
differential signal 32C is fed to a summer 341, where two
interference-cancellation signals 358A,B are added to the
differential signal 32C to produce an interference-reduced signal
351. The interference-cancellation signals 358A,B originate from an
adaptive interference control mechanism, preferably comprising a
common mode detector 347, a pair of band-pass filters 366A,B, a
pair of vector modulators 343A,B and an AIC control block 360. It
is to be understood that the present invention may employ other
suitable interference cancellation techniques.
[0051] In the preferred AIC mechanism, the common mode detector 347
extracts a common mode signal 359 from the twisted pair signal
32A,B. (Although the twisted pair signal will preferably be
transmitted in differential mode, interference may manifest itself
as a common mode signal affecting both wires of the twisted pair.)
The common mode signal 359 is fed to two band-pass filters 366A,B,
each of which has a pass band centered about a different
frequency.
[0052] The band-pass filters 366A,B then feed respective signals
369A,B to respective vector modulators 343A,B, which then apply
amplitude and phase changes to the signals 369A,B, thereby
producing the interference cancellation signals 358A,B. The
required amplitude and phase changes are fed by the AIC control
block 360 via control signals 365A,B. The AIC control block derives
signals 365A,B from interference estimates 357A,B provided by a
single sideband (SSB) down converter 349. Each interference
estimate 357A,B provides a measure of the interference remaining in
one of the two interfering HAM bands.
[0053] Continuing along the main signal path in the receiver 34,
the interference-reduced signal 351 enters a variable gain
amplifier (VGA, 342), which is controlled by a control signal 352
to produce a level-controlled signal 356 leading to a quadrature
demodulator 344. The quadrature demodulator 344 produces two
demodulated signals, namely, an in-phase signal 353A and an
in-quadrature signal 353B, which are fed to respective identical
Nyquist filters 345A,B. The quadrature demodulator 344 contains an
automatic gain control (AGC) function which controls the VGA 342
via control signal 352 that is a function of the difference between
an estimate of the combined power of the demodulated signals 353A,B
and a desired value.
[0054] The Nyquist filters 345A,B select the desired signal
contained in the demodulated signals 353A,B, rejecting out-of-band
signals and producing respective baseband demodulated signals
354A,B that enter respective identical notch filters 346A,B. The
notch filters 346A,B then attempt to remove any remaining
radio-frequency interference in the baseband demodulated signals
354A,B, providing respective filtered demodulated signals 355A,B to
a decision-feedback equalizer (DFE, 348).
[0055] The DFE 348 is a known component, essentially comprising a
linear transversal equalizer section followed by a non-linear
feedback section. These sections may be implemented as respective
digital filters whose parameters, usually in the form of
multiplicative coefficients, are adjusted by an adaptive algorithm
internal to the DFE 348. The DFE strives to eliminate any residual
inter-symbol interference still present in the filtered demodulated
signals 355A,B and produces both an internal control signal for
adapting its taps as well as the digital data stream 35 ideally
containing the exact digital data transmitted by the HDT.
[0056] It is to be noted that the baseband demodulated signals
354A,B also establish a feedback control path by virtue of being
connected to the SSB down converter 349. As was introduced earlier,
the SSB, down converter 349 respectively provides the interference
estimates 357A,B to the AIC control block 360 based on the power
contained in respective HAM bands f.sub.H1,f.sub.H2. FIG. 5B shows
a suitable embodiment of the SSB down converter 349, which accepts
the baseband demodulated signals 354A,B and passes each signal
354A,B through a respective sine multiplier 3491A,B and cosine
multiplier 3492A,B. The sine multipliers 3491A,B are fed by a sine
wave at frequency f.sub.S/4 and the cosine multipliers 3492A,B are
fed by a cosine wave at frequency f.sub.S/4.
[0057] The output of sine multiplier 3491A is added to the output
of cosine multiplier 3492B at a summer 3493A, whose output is
subsequently fed to a low-pass filter 3494A. Similarly, the output
of sine multiplier 3491A is subtracted from the output of cosine
multiplier 3492B at a summer 3493B feeding a low-pass filter 3494B.
The outputs of sine multiplier 3491B and cosine multiplier 3492A
are similarly arranged at summers 3493C,D, which feed respective
low-pass filters 3494C,D.
[0058] The low-pass filters 3494A,B,C,D then remove any energy from
their input signals and supply residual interference signals to
respective rectifiers 3495A,B,C,D. The output of rectifier 3495A is
combined with the output of rectifier 3495D at a summer 3596A to
produce an estimate of the power of the interference surrounding
f.sub.H1, which is fed to the AIC control block 360 as the
interference estimate 357A. An estimate of the power of the
interference surrounding f.sub.H2 is similarly obtained by adding
together the outputs of rectifiers 3585B and 3595C, forming the
interference estimate 357B that is fed to the AIC control block
360.
[0059] Referring back to FIG. 5A, it is to be understood that
analog-to-digital (A/D) conversion is to be performed at some point
in the receiver 34. Preferably, such conversion will be performed
by an A/D converter placed at the output of the VGA 342, although
it is equally suitable to to provide a pair of converters accepting
the demodulated signals 353A,B at the output of the quadrature
demodulator 344 or at any other point.
[0060] It is also to be considered that while preferred placement
of the notch filters 346A,B is in the baseband domain, it is also
suitable to perform these operations at passband. However, the
savings in terms of reduced computational complexity with respect
to the prior art are not as significant as when filtering is
performed at baseband, as in the preferred embodiment of FIG.
5A.
[0061] In receiver operation, the band-pass filters 366A,B filter
the common mode signal 359 into separate non-overlapping signals
369A,B representing interference in the two HAM bands. Each vector
modulator 343A,B independently vector modulates the corresponding
signal according to a respective amplitude and phase adjustment fed
via the corresponding control signal 365A,B from the AIC control
block 360, thereby producing the interference-cancellation signals
358A,B which are added to the differential-signal 32C by the summer
341. The AIC control block 360 executes an algorithm to determine
the required amplitude and phase parameters based on the
interference estimates 357A,B obtained from the SSB down converter
349.
[0062] It has been observed that with the aid of an AIC mechanism
as described herein, narrowband radio-frequency interference can be
reduced by up to 30 dB in each band. Naturally, other AIC
techniques may also be used, which may or may not yield superior
performance than the embodiment in FIG. 5A.
[0063] Continuing along the path through the receiver 34, the gain
of the level-controlled signal 356 output by the VGA 342 is
adjusted according to the power (or energy or magnitude) level of
the demodulated signals 353A,B as demodulated by the quadrature
demodulator 344. This ensures that a relatively constant signal
gain is maintained. The quadrature demodulator is 344 a component
known and used in the art for producing the in-phase and
in-quadrature demodulated signals 353A,B from the (quadrature
modulated) level-controlled signal 356.
[0064] The Nyquist filters 345A,B provide the first step in
filtering the demodulated signals 353A,B, by eliminating any
spectral content outside the range of interest, which is located in
the baseband domain from DC to half the symbol rate. At this point,
it is useful to remind the reader that in accordance with the
present invention, any HAM radio interference will appear in the
baseband demodulated signals 354A,B at DC plus-or-minus f.sub.S/4.
The notch filters 346A,B then apply a notch symmetrically disposed
about DC in order to eliminate the remaining interference.
[0065] This symmetry allows implementation of the notch filters
346A,B with real-valued coefficients, as was the case in the
transmitter of FIG. 4. Again, the width of the notch cannot be
independently controlled and therefore it is preferable for the
symmetric baseband notch applied by the notch filters 346A,B to be
at least as wide as the wider of the two HAM bands centered about
f.sub.H1 and f.sub.H2. Another feature that the notch filters
346A,B share with the notch filters 252A,B in the transmitter 25 of
FIG. 4 is that the placement of a notch at f.sub.S/4 is easy to
achieve with a small number of taps, due to the natural tendency of
the frequency response of a digital filter to exhibit notches at
fractional intervals of the sampling frequency.
[0066] The DFE 348 is the final forward link in the receiver chain
and makes decisions about the transmitted symbols based on the
received symbols as output by notch filters 346A,B. It is important
to note that while the DFE 348 is capable of eliminating much of
the inter-symbol interference in a signal corrupted by a number of
stable, narrow interference bands, its coefficients must reconverge
when the interference changes bands dynamically. This is often the
case with intermittent HAM radio transmissions, which would lead to
frequent readaptation, and possibly divergence of the DFE
coefficients causing signal outages. For this reason, the presence
of the notch filters 346A,B is preferred, in order to eliminate
frequency content in the interference bands regardless of whether
or not HAM transmissions are currently taking place in those
bands.
[0067] The baseband demodulated signals 354A,B also form a feedback
path leading to the SSB down converter 349, which attempts to
estimate the amount of interference remaining in the in-phase
baseband demodulated signal 354A and in the in-quadrature baseband
demodulated signal 354B. The in-phase component of the interference
at a frequency f.sub.I can be represented as:
354A=A.sub.X(t) cos (2.pi.(f.sub.I-f.sub.C)t),
[0068] whereas the in-quadrature component can be represented
as
354B=A.sub.Y(t) sin (2.pi.(f.sub.I-f.sub.C)t),
[0069] where A.sub.X and A.sub.Y are the respective amplitudes of
the in-phase and in-quadrature baseband demodulated signals 354A,B
and t is a measure of time.
[0070] The sine and cosine multipliers 3491A,B and 3492A,B will
further demodulate these signals, bringing the interference to DC
when they operate at f.sub.S/4. This convenient sampling rate
requires the calculation of only three (real) values for the sine
and cosine multiplicands, i.e., 0, 1 or -1, which has the
implication that the signals 354A,B passing through the sine and
cosine multipliers 3491A,B and 3492A,B are simply dropped, passed
through or inverted.
[0071] The outputs of the sine and cosine multipliers have their
phases adjusted by the summers 3593A,B,C,D, which, after low-pass
filtering, provide the following signals at points 3597A,B,C,D: 1
3497 A = 1 - sgn ( f I - f H1 ) 2 A A ( t ) sin ( 2 .PI. ( f H1 - f
I ) t ) 3497 B = 1 + sgn ( f I - f H2 ) 2 A B ( t ) cos ( 2 .PI. (
f I - f H2 ) t ) 3497 C = 1 + sgn ( f I - f H2 ) 2 A C ( t ) sin (
2 .PI. ( f I - f H2 ) t ) 3497 D = 1 - sgn ( f I - f H1 ) 2 A D ( t
) cos ( 2 .PI. ( f I - f H1 ) t )
[0072] where sgn(x) is the signum function of x and A.sub.A through
A.sub.D are respective interference amplitudes. It is clear from
the above that the interference level at points 3497A and 3497D
will be very close to zero when fI is very close to f.sub.H1.
Therefore, measuring the power of signals 3497A and 3497D, and
summing these measurements as is done by summer 3496A, provides an
estimate of the residual interference surrounding f.sub.H1 (namely,
interference estimate 357A). Similarly, the interference level at
points 3497B and 3497C will be very close to zero when fI is very
close to f.sub.H2. These two signals are rectified and combined by
summer 3496B, thereby providing an estimate of the residual power
surrounding f.sub.H2, namely interference estimate 357B.
[0073] From the above, it can be concluded that by carefully
selecting f.sub.C and f.sub.S, a high-data-rate signal whose
spectrum straddles two HAM bands is made not to cause interference
to HAM radio operators and at the same time is made immune to their
transmissions. Furthermore, the implementational benefits include a
simplified transmitter and receiver having short baseband filters
with real-valued coefficients.
[0074] In some applications, the bandwidth of the signal intended
to be transmitted may be on the order of 5 MHz or less, in which
case it is possible to align the signal spectrum so that only one
amateur band is straddled, leading to even more radical
simplifications in the transmitter and receiver. An inventive
frequency allocation scheme in accordance with such an alternate
embodiment of the present invention is shown in FIG. 6.
[0075] In the example of FIG. 6, the downstream spectrum 42 now has
a carrier frequency f.sub.C equal to (or within about 10% of) the
center frequency of HAM band 14, in this case 7.15 MHz, and
occupies a spectral region between 4 MHz and 10 MHz supporting a
symbol rate f.sub.S of 6.div.1.5=4.0 MHz (after accounting for
guard bands). It is noted that f.sub.S is not constrained to a
single value, but rather can take on any value less than the above
calculated value in the case of HAM band 14.
[0076] In a 16-QAM system, a 6 MHz downstream bandwidth enables the
delivery of 4*(6.div.1.5)=16 Mbps, which is adequate in many
instances, illustrating that the alternate embodiment of the
present invention is just as useful than the preferred embodiment,
if not more so. It is to be understood, of course, that this
alternate spectral positioning technique may also be applied to the
upstream spectrum 44, although in this example, it continues to lie
between HAM bands 15 and 16.
[0077] FIG. 7 shows the corresponding simplifications to the
transmitter 25, which now comprises two real-coefficient baseband
high-pass filters 255A,B (instead of the two real-coefficient
baseband notch filters of FIG. 4). The high-pass filters 255A,B
simply attenuate frequencies around DC (at baseband), which
translates into removing frequency content around the carrier
frequency once the signals 259A,B are Nyquist filtered by
respective Nyquist filters 253A,B and quadrature modulated by the
quadrature modulator 254. Trading a band-pass filter for a
high-pass filter usually results in halving the computational
complexity, as only half the coefficients are generally required to
achieve the same spectral sharpness.
[0078] Aside from straightforward filtering, another way to produce
a notch around the carrier frequency is for the QAM encoder 251 to
provide digital signals 258A,B that have been encoded using a
partial response filter with a null at DC, for instance, a class 4
partial response filter. Subsequent to modulation by the quadrature
modulator 254, the transmitted downstream modulated signal 26 will
contain a "natural" notch at the carrier frequency.
[0079] The simplifications to the receiver are most evident when
described with reference to FIG. 8, in which is shown a receiver 34
for use with a system having a downstream spectrum centered about a
HAM band. The receiver 34 now comprises only one vector modulator
343 which deals with only one interferer, and therefore is capable
of accepting the (unfiltered) common mode signal 359 from the
common mode detector 347. The AIC control block 360 now provides
only one control signal 365, which is calculated by an algorithm
that relies on a single interference estimate 357 from a simple
power estimator 350.
[0080] The VGA 342, quadrature demodulator 344 and Nyquist filters
345A,B remain identical to the components in FIG. 5A. However, the
band-pass filters 346A,B of FIG. 5A have been replaced by optional
high-pass filters 340A,B, owing to the fact that the baseband
demodulated signals 354A,B will require a notch at DC to eliminate
the interference due to a single HAM band. The high-pass filters
340A,B are symmetrical, have real coefficients, and are also
optional, since the DFE 348 is capable of compensating for the
filtered demodulated signals 355A,B having been corrupted by a
single source of narrowband interference, as previously
discussed.
[0081] Moreover, the SSB down converter of FIGS. 5A and 5B has been
replaced in the receiver 34 of FIG. 8 by a very simple power
estimator 350 for removing any data present in the baseband
demodulated signals 354A,B by first passing them through respective
low-pass filters 3501A,B. (Since the residual interference occurs
from a single source at DC in the baseband domain, a second
demodulation phase is not required, and no separation of the
interferers is required.) The power of the in-phase interference is
measured by a rectifier 3502A connected to the output of low-pass
filter 3501A. Similarly, the in-quadrature interference power is
measured by a rectifier 3502B-connected to the output of the
low-pass filter 3501B. The output of each rectifier 3502A,B is then
supplied to a summer 3503, which forms the interference estimate
357 fed to the AIC control block 360.
[0082] Upon closer inspection of the receiver in FIG. 8, it is
noted that the AIC control block 360 requires only one interference
estimate 357 (as opposed to the AIC block of FIG. 5A, which needed
two such estimates). Since only a single power (or energy or
magnitude) estimate is required, it may in fact be taken from any
component which already generates a similar error signal. Such a
signal is generated by the quadrature demodulator 344, for example,
which provides the VGA 342 with a control signal 352 that will be
larger when there is more interference and smaller when there is
less interference. Also, the DFE 348 internally generates a
suitable control signal that may be tapped and brought to the AIC
control block.
[0083] These two control signals may both be used as an error
signal by the AIC control block, as is illustrated in FIG. 9. In
this alternate embodiment of the present invention, it is proposed
to supply the AGC control signal 352 from the quadrature
demodulator 344 and a DFE control signal 362 from the DFE 348 to
the AIC control block 360. The AIC control block 360 is assumed
capable of switching between relying on the control signal 352 or
on the control signal 362. It is also assumed that the control
signal 362 enables the AIC control block 362 to control adaptation
of the DFE coefficients.
[0084] In operation, there are two scenarios to consider, namely,
either the interfering HAM band is in use or it is not. When indeed
hit by interference, the AIC control block 360 begins by freezing
the DFE taps (via control signal 362), and calculates the
appropriate magnitude and phase shift to be applied by the vector
modulator 343, based on the control signal 352 from the quadrature
modulator 344. Meanwhile, the DFE 4348 attempts to eliminate
inter-symbol interference and will be unstable until the AIC
mechanism has somewhat reduced the interference in the differential
signal 32C.
[0085] After a stable operating point has been reached, that is to
say, after acquisition of the DFE 348, the AIC control block 360
switches to the DFE control signal 362 as a more sensitive estimate
of the remaining interference, which allows the AIC mechanism to
refine its suppression of HAM radio interference. The AIC control
block 360 will then converge sufficiently to reduce the strength of
the interference to less than the strength of the signal by a
certain amount of decibels. The AIC mechanism can provide up to 30
dB of interference suppression over an interference band of up to
approximately 10% of the carrier frequency.
[0086] The control algorithm 360 will subsequently un-freeze the
DFE coefficients via control signal 362, permitting them to
adaptively adjust in the normal way, using, e.g., a
least-mean-square algorithm. The DFE 348 will then create a notch
with its forward taps and strive to eliminate inter-symbol
interference with its feedback taps. When there are 16 forward taps
and 16 feedback taps, it has been simulated and confirmed through
experiment that additional interference suppression of 30 dB can be
achieved by the DFE, for a total of 60 dB when combined with the
effect of the AIC block 343.
[0087] If, on the other hand, the HAM band is not in use,
interference suppression by the AIC block 343 is neither desired
nor achieved. In this case, the AIC control block 360 will detect
an extremely low level of interference. The control algorithm 360
will simply instruct the DFE 348 via control signal 362 to
un-freeze the DFE coefficients, allowing them to adapt in the usual
way. In this scenario, it might also be desirable to cease
adaptation of the AIC block altogether.
[0088] It is to be noted that when the high-pass filters 340A,B are
omitted from the receiver 34 in FIG. 9, the only filtering stage in
the receiver is formed by the adaptable DFE coefficients. In cases
where HAM interference is intermittent, it is expected that
frequent readaptation of these coefficients will be required each
time there is a change in the amateur radio transmission
characteristics.
[0089] Interestingly, however, it has been observed that there is a
tendency for the DFE taps to remain constant, even after HAM band
interference has disappeared. That is to say, adaptation for the
HAM interference appears to reach a stable local optimum.
Nevertheless, since long term stability of the real part of the DFE
taps is not fully known, it is considered preferable to freeze the
real part of the DFE coefficients once a satisfactory operating
condition is reached so as to avert excessive readaptation periods
with a higher degree of certainty, while allowing the imaginary
part of the coefficients to adapt to changes in cable
characteristics, such as temperature.
[0090] While the preferred embodiment and several alternate
embodiments of the present invention have been described and
illustrated, it is to be understood that variations in the design
may be made. For example, the modulation format used in the
inventive transmitter and receiver may be a different from the
16-QAM modulation scheme described herein. Other suitable
modulation types include different levels of QAM or different
modulation formats altogether, such as CAP, BPSK or differential
schemes based on any of these formats.
[0091] Moreover, as a general comment having regard to the
inventive spectral allocation schemes depicted in FIGS. 3 and 6, it
should be appreciated that the invention applies to the
transmission and reception of signals in both directions of traffic
flow, i.e., upstream and downstream. Furthermore, use of the
methods disclosed herein does not preclude the parallel use of
other transmission techniques to enhance bandwidth capacity.
[0092] For example, it is feasible to transmit a first portion of a
downstream spectrum straddling two HAM bands in accordance with the
preferred embodiment of FIG. 3 and to transmit a second portion of
the downstream spectrum located between two other HAM bands. At the
same time, the upstream spectrum may also be divided into two
portions, one of which may rest between two HAM bands and the other
of which may straddle one HAM band in accordance with the alternate
embodiment of FIG. 6.
[0093] In view of the above description and illustrations,
therefore, the scope of the invention is only to be limited by the
claims appended hereto.
* * * * *