U.S. patent application number 10/060728 was filed with the patent office on 2003-07-31 for quadrature vestigial sideband digital communications method.
This patent application is currently assigned to The Aerospace Corporation. Invention is credited to Mitchell, Gregory S., Poklemba, John J..
Application Number | 20030141938 10/060728 |
Document ID | / |
Family ID | 27610076 |
Filed Date | 2003-07-31 |
United States Patent
Application |
20030141938 |
Kind Code |
A1 |
Poklemba, John J. ; et
al. |
July 31, 2003 |
QUADRATURE VESTIGIAL SIDEBAND DIGITAL COMMUNICATIONS METHOD
Abstract
A quadrature vestigial sideband (QVSB) communication system
provide bandwidth efficient data transmission using cross coupled
data signaling during both transmit and receive having controlled
intersymbol interference. The QVSB modem includes cross coupled arm
transmit and receive data filtering on both of the I&Q channels
providing a bandwidth efficient QVSB spectra. A quadrature
crosstalk maximum likelihood sequence estimator implements a
Viterbi decoding algorithm for providing estimated data sequence
outputs. The receiver is a coherently aided demodulator
synchronized by a synchronization loop providing time and phase
references using the estimated data sequence outputs.
Inventors: |
Poklemba, John J.;
(Ijamsville, MD) ; Mitchell, Gregory S.;
(Bethesda, MD) |
Correspondence
Address: |
Derrick M. Reid
Patent Attorney
The Aerospace Corporation
P.O. Box 92957 (M1/040)
Los Angeles
CA
90009-2957
US
|
Assignee: |
The Aerospace Corporation
Los Angeles
CA
|
Family ID: |
27610076 |
Appl. No.: |
10/060728 |
Filed: |
January 30, 2002 |
Current U.S.
Class: |
332/103 |
Current CPC
Class: |
H04L 27/02 20130101 |
Class at
Publication: |
332/103 |
International
Class: |
H03C 003/00 |
Claims
What is claimed is:
1. A method of modulating a carrier for transmitting a quadrature
vestigial single sideband (QVSB) signal for communicating I symbols
and Q symbols, the method comprising the steps of, cross arm
filtering the I and Q symbols for generating i filter responses and
i.sub.H filter responses from the I symbols and for generating
q.sub.H filter responses and q filter responses from the Q symbols,
combining the i and q.sub.H filter responses into an I channel
overlapping filter signal and the q and i.sub.H filter responses
into a Q channel overlapping filter signal, and quadrature
modulating a carrier by the I and Q channel overlapping filter
responses into a modulated inphase signal and a modulated
quadrature signal, the Q channel overlapping filter response
modulates the carrier shifted ninety degrees from the carrier
modulating the I channel overlapping filter responses for
modulating the carrier in quadrature, summing the modulated inphase
signal and a modulated quadrature signal into the QVSB signal, and
transmitting the QVSB signal for communicating the I and Q symbols
over a single sideband.
2. The method of claim 1 wherein cross arm filter step, the i and
i.sub.H filter responses are near Hilbert transform pair filtered
responses, and the q and q.sub.H filter responses are near Hilbert
transform paired filtered responses.
3. The method of claim 1 wherein, the i and i.sub.H filter
responses are near Hilbert transform pair filtered responses, the q
and q.sub.H filter responses are near Hilbert transform paired
filtered responses, the i and i.sub.H filter responses are selected
from the group consisting of raised cosine, jump, and smoothed jump
filtered responses, and the q and q.sub.H filter responses are
selected from the group consisting of raised cosine, jump, and
smoothed jump filtered responses.
4. The method of claim 1 wherein, the I and Q symbols have a symbol
time, and the QVSB signal transmitted in quadrature has cross
coupled intersymbol interference extending from the symbol time of
one of the I and Q symbols into an adjacent symbol times of an
adjacent one of the Q and I symbols, respectively.
5. The method of claim 1 further comprising the step of, the
mapping data into the I and Q symbols for providing the I and Q
symbols as independent data sets having a constellation of
detection levels.
6. The method of claim 1 wherein the cross arm filtering step and
the combining step comprises the steps of, Hilbert transform
filtering the I symbols into the i filter response, inverse Hilbert
transform filtering the Q symbol into the q.sub.H filter response,
inverse Hilbert transform filtering the I symbols into the i.sub.H
filter response, Hilbert transform filtering the Q symbols into the
q filter response, cross coupled summing the i and q.sub.H filter
responses into the I channel overlapping filter responses as
i+q.sub.H I channel overlapping filter responses, and cross coupled
subtracting the i.sub.H and q filter responses into the Q channel
overlapping filter responses as q-i.sub.H Q channel overlapping
filter responses.
7. A method of demodulating a transmitted quadrature vestigial
sideband (QVSB) signal modulating a carrier in quadrature by I and
Q overlapping filter responses respectively from I and Q symbols,
the method comprising the steps of, receiving the transmitted QVSB
signal as a received QVSB signal, splitting the received QVSB
signal into an I channel QVSB signal and a Q channel QVSB signal,
coherent demodulating the I channel QVSB signal by a replicated
carrier and the Q channel QVSB signal by a ninety degree phase
shifted replicated carrier for respectively generating the I and Q
overlapping filter responses, cross arm filtering the I and Q
overlapping filter responses for generating +{tilde over (q)}.sub.H
and -{tilde over (q)}+.sub.H match filter responses from the I
overlapping filter responses and for generating +{tilde over
(q)}.sub.H and {tilde over (q)}-.sub.H matched filter responses
from the Q overlapping filter responses, and combining both of the
+{tilde over (q)}.sub.H matched filter responses into an I channel
response signal and the -{tilde over (q)}+.sub.H and {tilde over
(q)}-.sub.H matched filter responses into a Q channel response
signal.
8. The method of claim 7 wherein, the +{tilde over (q)}.sub.H and
-{tilde over (q)}+.sub.H match filter responses are near Hilbert
transform pair filtered responses, and the +{tilde over (q)}.sub.H
and {tilde over (q)}-.sub.H matched filter responses are near
Hilbert transform pair filtered responses.
9. The method of claim 7 wherein, the +{tilde over (q)}.sub.H and
-{tilde over (q)}+.sub.H match filter responses are near Hilbert
transform pair filtered responses, the +{tilde over (q)}.sub.H and
{tilde over (q)}-.sub.H matched filter responses are near Hilbert
transform pair filtered responses, the +{tilde over (q)}.sub.H and
-{tilde over (q)}+.sub.H match filter responses are filter
responses are selected from the group consisting of raised cosine,
jump, and smoothed jump filtered responses, and +{tilde over
(q)}.sub.H and {tilde over (q)}-.sub.H match filter responses are
filter responses are selected from the group consisting of raised
cosine, jump, and smoothed jump filtered responses.
10. The method of claim 7 wherein the I channel overlapping
response signal is an i+q.sub.H overlapping response signal and the
Q channel overlapping response signal is a q+i.sub.H overlapping
response signal, the cross arm filtering step and combining step
comprises the steps of, Hilbert transform filtering the i+q.sub.H
overlapping response signal into an +{tilde over (q)}.sub.H match
filter response, inverse Hilbert transform filtering the q-i.sub.H
overlapping response signal into an +{tilde over (q)}.sub.H matched
filter response, inverse Hilbert transform filtering the i+q.sub.H
overlapping response signal into an -{tilde over (q)}+.sub.H match
filter response, Hilbert transform filtering the q-i.sub.H
overlapping response signal into a {tilde over (q)}-.sub.H match
filter response, cross coupled summing the +{tilde over (q)}.sub.H
matched filter responses into the I channel response signal as an
{tilde over ({tilde over (i)})}=2[+{tilde over (q)}.sub.H] I
channel response signal, and cross coupled subtracting the -{tilde
over (q)}+.sub.H and {tilde over (q)}-.sub.H matched filter
responses into the Q channel response signal as a {tilde over
({tilde over (q)})}=2[{tilde over (q)}-.sub.H] Q channel response
signal.
11. The method of claim 7 further comprising the steps of,
detecting the I and Q channel responses for generating a
synchronized timing signal for carrier tracking the replicated
carrier and the ninety degree phase shifted carrier for coherent
demodulation of the I and Q channel overlapping response
signals.
12. The method of claim 7 further comprising the step of, detecting
from the I and Q symbols from the I and Q channel response signals
and mapping the I and Q symbols into data.
13. A method of communicating I and Q symbols, the method
comprising the steps of, cross arm filtering the I and Q symbols
for generating i filter responses and i.sub.H filter responses from
the I symbols and for generating q.sub.H filter responses and q
filter responses from the Q symbols, combining the i and i.sub.H
filter responses into an I channel overlapping filter signal and
the q and q.sub.H filter responses into a Q channel overlapping
filter signal, and quadrature modulating a carrier by the I and Q
channel overlapping filter responses into a modulated inphase
signal and a modulated quadrature signal, the Q channel overlapping
filter response modulates the carrier shifted ninety degrees from
the carrier modulating the I channel overlapping filter responses
for modulating the carrier in quadrature, combining the modulated
inphase signal and a modulated quadrature signal into a quadrature
vestigial single sideband (QVSB), transmitting the QVSB signal for
communicating the I and Q symbols over a single sideband, receiving
the transmitted QVSB signal as a received QVSB signal, splitting
the received QVSB signal into an I channel QVSB signal and a Q
channel QVSB signal, coherent demodulating the I channel QVSB
signal by a replicated carrier and the Q channel QVSB signal by a
ninety degree phase shifted replicated carrier for respectively
generating the I and Q overlapping filter responses, cross arm
filtering the I and Q overlapping filter responses for generating
+{tilde over (q)}.sub.H and -{tilde over (q)}+.sub.H match filter
responses from the I overlapping filter responses and for
generating +{tilde over (q)}.sub.H and {tilde over (q)}-.sub.H
matched filter responses from the Q overlapping filter responses,
and combining the +{tilde over (q)}.sub.H matched filter responses
into an I channel response signal and -{tilde over (q)}+.sub.H and
{tilde over (q)}-.sub.H matched filter responses into a Q channel
response signal.
14. The method of claim 13 wherein, the i and i.sub.H filter
responses are near Hilbert transform pair filtered responses, the q
and q.sub.H filter responses are near Hilbert transform paired
filtered responses, the i and i.sub.H filter responses are selected
from the group consisting of raised cosine, jump, and smoothed jump
filtered responses, and the q and q.sub.H filter responses are
selected from the group consisting of raised cosine, jump, and
smoothed jump filtered responses, the +{tilde over (q)}.sub.H and
-{tilde over (q)}+.sub.H match filter responses are near Hilbert
transform pair filtered responses, the +{tilde over (q)}.sub.H and
{tilde over (q)}-.sub.H matched filter responses, are near Hilbert
transform pair filtered responses, the +{tilde over (q)}.sub.H and
-{tilde over (q)}+.sub.H match filter responses are filter
responses are selected from the group consisting of raised cosine,
jump, and smoothed jump filtered responses, and the +{tilde over
(q)}.sub.H and {tilde over (q)}-.sub.H match filter responses are
filter responses are selected from the group consisting of raised
cosine, jump, and smoothed jump filtered responses.
15. The method of claim 13 wherein, the i and i.sub.H filter
responses are near Hilbert transform pair filtered responses, the q
and q.sub.H filter responses are near Hilbert transform paired
filtered responses, the +{tilde over (q)}.sub.H and -{tilde over
(q)}+.sub.H match filter responses are near Hilbert transform pair
filtered responses, the +{tilde over (q)}.sub.H and {tilde over
(q)}-.sub.H matched filter responses are near Hilbert transform
pair filtered responses, and all of the near Hilbert transform pair
filter responses are generated from identical cross arm filtering
selected from the group consisting of raised cosine, jump, and
smoothed jump filtering.
16. The method of claim 13 further comprising the steps of, the
mapping data into the I and Q symbols for providing the I and Q
symbols as independent data sets having a constellation of
detection levels, sampling the I and Q channel responses into I and
Q discrete sample values at discrete times within a set of
constellation amplitude values, and estimating the data from the I
and Q discrete sample values using the I and Q discrete sample
values.
17. The method of claim 13 further comprising the steps of, the
mapping original data into the I and Q symbols for providing the I
and Q symbols as independent data set having a constellation of
detection levels prior to modulation, and quantizing the I and Q
channel response signals for estimating the original data.
18. The method of claim 13 further comprising the step of, carrier
varying the replicated carrier and ninety degree phase shifted
replicated carrier for synchronization to the symbol times for
coherent demodulation of the received QVSB signal.
19. The method of claim 13 further comprising the step of,
amplitude varying the amplitude of the received signal for constant
amplitude demodulation of the received QVSB signal.
20. The method of claim 13 further comprising the step of, phase
varying for phase synchronizing the replicated carrier and the
ninety degree phase shifted carrier for coherently demodulating the
received QVSB signal.
Description
FIELD OF THE INVENTION
[0001] The invention relates to the field of quadrature modulation
communication systems. More particularly, the present invention
relates to vestigial sideband modulation communications systems
using cross coupled independent data stream modulated on a common
carrier in quadrature.
BACKGROUND OF THE INVENTION
[0002] The rapid, worldwide expansion of communications services
underscores the importance of bandwidth conservation. With
increased demands for cellular and personal communications services
within a finite radio frequency spectrum, there is an
ever-increasing contention for bandwidth. Cellular services are
growing at a geometric rate. Microcellular sites are being
advocated to handle the increased demand through localized
frequency reuse, and hundreds of low earth orbit and medium earth
orbit satellites will support the increasing demand for bandwidth
over the next decade. In digital video communications, high
definition television (HDTV) transmits at 21.5 Mbits/s with a
greatly improved picture quality that must be compatible with the
existing 6.0 MHz channel bandwidth allocation. This requires a
bandwidth efficiency of greater than 3.0 bits/s/Hz. Additionally,
data throughputs in communications have also been increasing at an
exponential rate. Existing bandwidth allocations are typically
shared among different services. A review of the current frequency
allocations reveals that the majority of bands exhibit sharing of
multiple services, such as, fixed and mobile satellite services and
earth exploration satellites. A natural consequence of this sharing
is increasing interference. With the bandwidth being a finite
resource, there are increasing demands for this finite bandwidth
resources creating a need to develop general purpose practical
bandwidth efficiency communication techniques.
[0003] Digital data has been transmitted using double sideband
(DSB) or quadrature double sideband (QDSB) techniques.
Occasionally, single sideband (SSB) formats have been used, and
more recently two vestigial sideband (VSB) formats have been
selected as the standards for off the air and cable HDTV. DSB
signaling is the simplest and most straight forward means to
transmit analog or digital information on a carrier, such as, when
using AM and FM methods. SSB is employed when the bandwidth is at a
premium, such as, when multiplexing terrestrial telephone channels.
VSB is used when requiring a controlled component of energy at the
carrier frequency, such as, in TV and HDTV communications.
[0004] One of the most useful ways to assess bandwidth efficiency
is to make use of the Shannon channel capacity bound that provides
an upper limit on the signaling rate R.sub.s for error free
transmission over an arbitrary channel. Modern digital modulation
techniques are compared to the Shannon channel capacity bound to
provide a performance overview. When the maximum signaling rate is
normalized by the required transmission bandwidth, a measure of the
bandwidth efficiency of the modulation method is obtained in units
of bits/s/Hz. This normalized performance benchmark is known for
many of the widely used modulation formats. Unfiltered digital data
typically has a sin(x)/(x) frequency response with significant
sidelobe content over a bandwidth wider than the data symbol rate.
The Nyquist technique is used to transmit digital data within a
limited bandwidth without intersymbol interference. Intersymbol
interference (ISI) is eliminated when the response magnitude
through a transmission channel has vestigial symmetry about the
half amplitude point that occurs at a frequency equal to half the
symbol rate with the communication channel providing a linear phase
response. When the magnitude response of the channel transmission
function has vestigial symmetry about the half amplitude point that
occurs at a frequency equal to half the symbol rate, and when the
transmission function has linear phase, data can be communicated
without ISI. The bandwidth efficiency has been calculated assuming
transmission at the minimum Nyquist bandwidth R.sub.s/2. A
E.sub.b/N.sub.o scale is used to derive a bit error ratio (BER),
for example, 10.sup.-6, during data communications.
[0005] The single sideband and quadrature single sideband (QSSB)
modulation format data points have exactly twice the bandwidth
efficiency of the corresponding double sideband counterparts where
the effect of quadrature channel crosstalk can be rendered
negligible. The quadrature channel crosstalk is inherent in QSSB
transmission in which independent data is placed on quadrature
carriers. The crosstalk degrades performance and has been a major
problem in QSSB communications. The DSB techniques diverge from the
bound as the number of bits/s/Hz or bandwidth efficiency is
increased, whereas ideally transmitted QSSB formats run parallel to
the bound. This divergence is due to the redundancy in transmitting
two sideband replicas. As the bandwidth efficiency of the channel
is increased, QSSB potentially offers a progressively larger
advantage over DSB transmission. In particular, when a six bit/s/Hz
efficiency is needed, a conventional phase shift keying (PSK) may
be used, such as 64-PSK. The DSB scheme would be required as
compared to an 8-PSK QSSB format. The DSB scheme requires 18 dB
more signal power to achieve the same BER. In general, the number
of signal levels needed with DSB techniques is the square of that
required with an equivalent QSSB format. These large discrepancies
in signal to noise ratio (SNR) and number of signal levels leave
considerable margin for non-ideal SSB signaling due to crosstalk.
SSB uses half the bandwidth of conventional DSB yielding twice the
bandwidth efficiency. Because of the sharp cutoff characteristics
at one of the SSB band edges, vestigial sideband method is often
used to realize a more gradual rolloff. The VSB method is not as
bandwidth efficient as the SSB method, but generally leads to a
more practical solution with controlled crosstalk. Conventional VSB
filtering uses inphase and quadrature arm filters in both the
transmit modulators and receive demodulators. The data stream is
reinforced in the receiver by transmitting the data stream through
both quadrature and inphase channels. The quadrature arm odd
responses combine to yield a even response. The VSB is slightly
less bandwidth efficient than SSB, but is more controlled and
easier to implement. VSB or SSB frequency spectra transform to
inphase and quadrature impulse response with even and odd time
symmetry, respectively.
[0006] In practice, during VSB modem transmit and receive filtering
modeling, matched filtering is employed so that the square root of
the Nyquist frequency response is apportioned equally to the
transmitter and receiver as opposed to a full response. In general,
impulse responses with equally spaced axis crossings will only
occur after passing through matched sets of transmit and receive
filters. The critical filtering for a Nyquist band limited VSB
transmitters and receivers is employed in HDTV. In the HDTV system,
a single data channel is communicated through both inphase and
quadrature (I&Q) channels respectively having a h.sub.i arm
filter and a h.sub.q arm filter in both of the transmitter and
receiver for communicating i(t) and i.sub.H(t) signals. The
transmit square root h.sub.i and h.sub.q arm filters have an even
and odd impulse response relationship but neither has equally
spaced axis crossings. The H subscript is used to denote the odd
impulse responses that are similar to the Hilbert transform of the
even impulse responses. The Hilbert transform j*sgn(f) has an
abrupt 90.degree. phase transition in the frequency domain. The
Hilbert transform is used to realize the precipitous sideband
rejection in SSB via the phase shift generation method. The
sideband rejection in VSB is more gradual. Sideband rejection is
typically realized through a combination of I&Q channel
amplitude mismatch in conjunction with a Hilbert phase shift
discontinuity and hence similar to the Hilbert transform.
[0007] In the receiver, both the I&Q filter output responses
are approximately the same with each having even time symmetry. The
I-channel response is related to the Q-channel response. The
Q-channel response is approximately the same at the I-Channel
response because it is roughly the cascade of two Hilbert transform
90.degree. phase shifted versions of the I-channel response that
which is merely an inverted version of the I-channel response. The
Q-channel output then is summed with the I-channel to improve the
detection SNR by 3.0 dB. There is no ISI problem when Nyquist
filtering is used in conjunction with VSB data transmission as is
evidenced by I-channel responses in a 25% raised cosine VSB eye
diagram.
[0008] Vestigial sideband is defined such that when its spectrum is
downconverted to baseband, the inner transition regions of its
positive and negative frequency image bands overlap and are
complementary so as to sum to unity with proper phasing. VSB is a
good compromise between DSB and SSB because VSB approaches SSB in
bandwidth efficiency, but does not require an infinitely sharp
transition band. HDTV will be transmitted digitally using trellis
coded 8-ary VSB and 16-ary VSB formats, for terrestrial and cable
distribution, respectively. These VSB formats require 8-ary and
16-ary amplitude levels in their baseband modulating waveforms. To
facilitate VSB signaling, a common digital data stream modulates
two quadrature carriers where the impulse response pairs are
orthogonal and correlated. Quadrature SSB (QSSB) and QVSB are more
complex than SSB and VSB because each of the inphase (I) and
quadrature (Q) baseband modulating channels contain the
superposition of an independent pair of data streams having
interference and crosstalk that must be controlled. The advantage
of QSSB/QVSB over SSB/VSB is a doubling of the information carrying
capacity. The disadvantages are greater implementation complexity,
and a typically reduced noise margin due to crosstalk.
[0009] There are fundamentally two methods of generating SSB/VSB,
that is, the quadrature phase shift method and sideband filtering
method. In this study, the phase shift method is favored over the
filtering approach, because precise control over the modulating
waveshapes will be necessary; and this precision is best achieved
with digital signal processing techniques. The phase shift approach
is shown analytically for an SSB modulator output, s(t) by an SSB
modulator output equation
s(t)=i(t)cos(.omega..sub.ct).+-.(t)sin(.omega..sub.ct), where the
baseband message waveform, i(t) and the Hilbert transform (t)
modulate quadrature carriers. The minus sign on the Hilbert
component yields upper sideband (USB), whereas a plus sign gives
lower sideband (LSB). VSB can also be represented in this manner,
but the inphase and quadrature components are not strictly Hilbert
transforms.
[0010] To conserve bandwidth using SSB/VSB modem baseband filtering
modeling, digital modulation techniques are filtered prior to
transmission. To maximize the detection SNR, the receive filtering
is matched to the transmit filter. The critical baseband filtering
for a band limited SSB/VSB modem uses a single data stream with
single arm filtering for VSB modulation. The modulated output
signal is generated according to modulator output equation. To
facilitate SSB transmission, the inphase filters with the i
subscripts and the quadrature filters with the q subscripts must be
a Hilbert transform pair h.sub.q and H.sub.q, such that,
h.sub.q(t)=(1/.pi.t)*h.sub.i(t) and H.sub.q(f)=jsgn(f)H.sub.i(- f).
The symbol * is the convolution operator, and (t) is the time
domain variable and (f) is the frequency domain variable. The
Hilbert transform pair h.sub.i and h.sub.q are orthogonal by
definition, and with a perfectly balanced structure, complete
cancellation of one of the sidebands results. When h.sub.i(t) has
even symmetry, h.sub.q(t) would have odd. From the frequency
response definition in H(f), the cascaded response of any two
quadrature filters is the negative of the inphase filter responses,
for example h.sub.q*h.sub.q=-h.sub.i. Because the noises in the I/Q
detection arms are uncorrelated, and the signal components are
perfectly negatively correlated, combining the I&Q filtered
outputs yields a 3.0 dB improvement in the detection SNR. The term
{tilde over (h)}(t) is the Hilbert transform h(t). The double tilde
term {tilde over ({tilde over (h)})}(t) is the Hilbert transform of
{tilde over (h)}(t). Subtracting the double tilde impulse response
{tilde over ({tilde over (h)})}.sub.i from the inphase counterpart
acts as constructive interference where h.sub.q(t)={tilde over
(h)}.sub.i(t) and h.sub.i(t)={tilde over (h)}.sub.q(t) and {tilde
over ({tilde over (h)})}.sub.i(t)=-h.sub.i(t). For VSB
transmission, the h.sub.i and h.sub.q filter pairs are not strictly
Hilbert transforms of one another, but have vestigial symmetry
about the half power points in the frequency domain. This type of
VSB modem is used in HDTV, where i(t) has eight or sixteen
detection levels.
[0011] For memoryless Nyquist filtering, the Nyquist family of
filters are evaluated for applicability in achieving bandwidth
efficient transmission with minimal degradation in SNR performance
due to ISI. Ideal rectangular and raised cosine filtering have been
used for Nyquist filtering. Nyquist impulse responses are sinc
based waveshapes with even time symmetry and equally spaced zero or
axis crossings at integer multiples of the data symbol time. As a
result, responses from adjacent data symbols do not interfere at
the detection sampling instants. The impulse responses with equally
spaced axis crossings are realized when the frequency response has
vestigial symmetry about the half amplitude transmission points.
The most concentrated distribution of signal bandwidth in the
frequency domain is the ideal rectangular spectrum using ideal
rectangular filtering. The magnitude for an SSB version of the
ideal rectangular spectrum for the minimum Nyquist bandwidth
R.sub.s/2 can be considered on a frequency axis normalized by the
data symbol rate. The analytic signal is used so that SSB frequency
response is at baseband. The ideal rectangular spectrum represents
the sharpest cutoff extreme of the Nyquist filtering including the
raised cosine filtering. The inphase impulse response corresponding
to the SSB rectangular spectrum is the sinc function, and the
quadrature impulse response is a raised cosine with both decaying
at 1/t. Because the two impulses responses are Hilbert transform
pairs, the quadrature term h.sub.q(t) will have odd symmetry
because the quadrature term is odd and equivalent to 1/.pi.t
convolved with the even sinc function. The inphase and quadrature
transform pair is given by
h.sub.i(t)=sin(.pi.R.sub.st)/.pi.R.sub.st and
h.sub.q=(1-cos(.pi.R.sub.st))/.pi.R.sub.st.
[0012] When the SSB spectrum is band limited to half the data
symbol rate, the corresponding inphase impulse response will have
equally spaced axis crossings at integer multiples of the symbol
time T.sub.s and the quadrature impulse response will be zero at
.+-. even multiples of the data symbol time. The quadrature impulse
response has a 1/.pi.t symmetry. In a QSSB scheme, the quadrature
impulse response component from one channel will overlay the
inphase component of the other channel. Hence, in the case of an
ideal rectangular SSB quadrature impulse response pair, the
quadrature component will contribute ISI at .+-. odd multiples of
the data symbol time. Although the envelope of the ISI only falls
off as 1/t the ISI dispersion does not diverge because for random
data sequences with half of the ISI positive and half negative
resulting in significant cancellation. When the bandwidth of the
rectangular SSB spectrum is doubled to R.sub.s, the resulting
quadrature impulse response pairs are zero at all adjacent symbol
integer multiples where there will be no ISI at the detection
sampling points. However, to achieve this perfect isolation, the
same bandwidth as DSB is required.
[0013] A widely used Nyquist filter realization is the raised
cosine, which has a sinusoidally shaped transition band. The
frequency response for a raised cosine filter is defined by H(f).
The raised cosine H(f) equation is a frequency response equation
that defines the VSB magnitude response. The corresponding impulse
response is defined as h(t). The rolloff factor is 0<r<1 and
the half amplitude frequency is f.sub.h=R.sub.s/2. A closed form
expression for the time domain Hilbert transform of the impulse
response has not yet been found. The H(f) frequency response and
h(t) time domain impulse response equations are used to model the
raised cosine filter. 1 H ( f ) = { 1 , f ( 1 - r ) f h f 1 1 2 { 1
- sin [ ( f / f h - 1 ) 2 r ] } , f 1 f f h 0 , f ( 1 + r ) f h f 0
h ( t ) = sin ( R s t ) cos ( R s t ) ( R s t ) [ 1 - ( 2 rR s t )
2 ]
[0014] A Nyquist frequency response is known for a 20% raised
cosine filter in a VSB channel. In practice, this filter can be
closely approximated, but not realized exactly because of the
perfectly flat passband and stopband. In addition, the stopband
also has infinite attenuation. The corresponding impulse response
pair for the 20% square root raised cosine VSB response can be
generated by means of an FFT, and the impulse responses are very
similar to the ideal rectangular filter pair except ideal
rectangular filter pair have more ringing due to an abrupt
transition band. The even response has equally spaced axis
crossings, and the odd response has zeros at .+-. odd multiples of
the data symbol interval.
[0015] Employing conventional raised cosine family filtering for
QSSB or QVSB transmission would result in crosstalk that reduces
the intersignal distances thereby degrading the BER performance.
Partial response signaling has been used for SSB transmission.
However, partial response signaling has not been extended to QSSB.
The well known class-4 (1-D.sup.2) system has no DC content and is
characterized by a half sine wave magnitude response of a total
bandwidth R.sub.s/2. The (1-D.sup.2) moniker implies that for each
data symbol input, the PR filter outputs the difference of a data
modulated sinc pulse with a two symbol delayed version. For this
case, the Hilbert response does not have equally spaced axis
crossings at .+-. even multiples of the data symbol time. In
analyzing the band limiting pulse shapes for Nyquist equally spaced
axis crossings, the cross correlation of the I/Q filter pairs
should be zero at the detection sampling instants. In a typical
modem, matched filters that are the square root of the Nyquist
frequency transmittance function, are placed in the modulator and
demodulator. The transmit output that only passes through the
square root impulse response will generally not have equally spaced
axis crossings resulting in ISI.
[0016] A restricted type of QVSB signaling has previously been
disclosed in 1985. The QVSB system had two I&Q inphase and
quadrature data channels modulated in quadrature by a carrier in
the receiver and demodulated in quadrature in the receiver. There
were no arm filters in the I&Q channels in the transmitter or
receiver. At the output of the QVSB transmitter and at the input of
the receiver were disposed bandpass raise cosine filters for VSB
communication. The QVSB system operated only for very soft rolloff
spectra of restricted bandwidth efficiency range with substantial
degradations in signal to noise ratio (SNR) due to crosstalk. The
QVSB data transmission used Nyquist filters from the raised cosine
filtering to band limit the signal. Nyquist filtering is widely
used to eliminate intersymbol interference in conventional digital
data transmission schemes. However, in the QVSB system, Nyquist
filtering exhibits quadrature crosstalk and ISI in both channels.
The QVSB system has crosstalk between the inphase and quadrature
(I&Q) channels in a controlled form similar to intersymbol
interference in partial response systems. The QVSB system could use
a maximum likelihood sequence estimator (MLSE) to remove the ISI
based on a Viterbi algorithm. The QVSB system could employ digital
data feedback in the synchronization loops. These techniques are
taught in U.S. Pat. No. 4,419,759, entitled Concurrent Carrier and
Clock Synchronization for Data Transmission Systems, and U.S. Pat.
No. 4,472,817, entitled Non-PLL Concurrent Carrier and Clock
Synchronization. The QVSB system can behave like partial response
systems where precoding could be used to avoid error propagation.
However, the QVSB system precoder did not exploit the correlation
information in the received samples. Consequently, the Viterbi
probabilistic MLSE decoder showed a marked improvement over
preceding. The QVSB system achieved a bandwidth efficiency of 2.3
bits/s/Hz for a 75% raised cosine rolloff passband. This is double
the rate of 1.14 bits/s/Hz for QPSK transmission with a
corresponding rolloff passband. A digital SNR E.sub.b/N.sub.o
penalty of approximately 2.1 dB at a bit error ratio (BER) of
10.sup.-5 was experienced as a result of the crosstalk. At a
bandwidth efficiency of 3.0 Bits/s/Hz, the BER performance degraded
by about an additional 5.0 dB due to the increased crosstalk.
[0017] Nyquist filtering during VSB data transmission for QVSB
signaling can be analyzed using an eye diagram. An eye diagram is
an overlay of the time response for all possible data sequences.
The eye diagram highlights the effects of ISI. For the case of
binary data, the Nyquist filtered waveforms that make up the eye
diagram are typically bipolar. Hence, a threshold is set at zero
and samples are taken in the center, at the maximum eye opening.
Sample values above zero are detected as positive ones and samples
below zero are detected as negative ones, that is, digital ones and
zeros. Nyquist filtering does eliminate ISI at integer symbol time
multiples. Hence, it is known that digital data may be transmitted
without ISI when the channel filter response satisfies the Nyquist
criterion. The best linear channel detection performance is
obtained by matching the transmit and receive filter responses. The
best known Nyquist filters are the raised cosine filters. For
example, a VSB full raised cosine frequency response with a 25%
rolloff rate would have corresponding inphase and quadrature
impulse responses. These impulse responses correspond to the
overall Nyquist channel response when a single data one is
transmitted. Opposite polarity impulse responses would be used when
a data zero is transmitted. To facilitate VSB, complementary
impulse responses with even and odd time symmetry are needed in the
quadrature channels. The impulse response horizontal axis marks are
spaced such that adjacent symbol responses are centered at integer
symbol time multiples. The tails from adjacent symbol impulse
responses will overlap. However, for the inphase impulse response,
there is no ISI at integer symbol time multiples. Therefore, data
sequences can be symbol by symbol detected without any degradation
in SNR performance. The quadrature impulse response has ISI only at
odd symbol time multiples. The restricted QVSB system achieved a
good BER performance using a 100% raised cosine filter. The
performance for the 75% and 50% cases was substantially degraded,
and solutions do not converge below 50% rolloff. These and other
disadvantages are solved or reduced using the invention.
SUMMARY OF THE INVENTION
[0018] An object of the invention is to provide bandwidth efficient
communications using quadrature vestigial sideband signaling.
[0019] Another object of the invention is to generate bandwidth
efficient I&Q channel waveshapes that exhibit minimal
intersymbol interference and crosstalk.
[0020] Yet another object of the invention is to generate bandwidth
efficient I&Q channel waveshapes that exhibit minimal
intersymbol interference and crosstalk with reduced bit error
rates.
[0021] The present invention is a method for transmitting digital
data in a bandwidth efficient manner using a quadrature vestigial
sideband (QVSB) signaling. The method can be used in data
communication systems. The QVSB method may double the capacity of
comparable conventional formats by placing overlapping independent
data on each of two carriers in phase quadrature using cross
coupled arm filters. The data overlap is necessary to achieve QVSB
spectral occupancy.
[0022] The method eliminates as much of the crosstalk as desired in
progressive steps. The method is realized by modulating transmit
and demodulating receive hardware architectures, the later of which
preferably including a quadrature crosstalk maximum likelihood
sequence estimator (QCMLSE) specifically designed to support QVSB
signaling within I and Q channel crosstalk. Using various
combinations of filtering and higher level signaling
constellations, the method can provide as high a bandwidth
efficiency within signal processing technology permits with
relatively little degradation in the signal to noise ratio
(SNR).
[0023] A normalized channel capacity versus SNR for the QVSB
implementation can be derived from models of the QVSB structure
within a linear additive white Gaussian noise channel at perfect
synchronization. Over a very broad range of raised cosine filter
rolloffs, 4-ary QVSB achieves the same capacity as conventional
16-ary quadrature double sideband (QDSB), with up to 2.0 dB less
required SNR at a BER=10.sup.-5, and up to 5.5 dB less required SNR
for 16-ary QVSB. The implementation works down to 0% rolloff that
is equivalent to the ideal rectangular brick-wall filter response.
In addition to the raised-cosine family, jump filters can be used
to yield better capacity performance improvements. The performance
is better at higher BERs, such as 10.sup.-4 and 10.sup.-3. The
method can be augmented by forward error correction coding.
[0024] Operation with a 4-ary rectangular constellation over the
complete range of Nyquist spectral rolloff characteristics has been
achieved up to and including the 25% raised cosine response with
graceful SNR degradation. Thus, the method is robust with greater
bandwidth efficiency that can be realized via sharper rolloff.
M-ary QVSB signaling achieves twice the capacity of M-ary QDSB
signaling that is equivalent to the capacity of M.sup.2-ary QDSB.
In addition, M-ary QVSB attains the bandwidth efficiency with
several dB less SNR than required for QDSB. Due to the percent
rolloff definition for raised cosine filters, the same percent
rolloff for QVSB and QDSB results in a transition band that is half
as wide for QVSB and hence the factor of two in the bandwidth
efficiency. QVSB spectral shaping enables all significant
intersymbol interference (ISI) beyond the adjacent symbols of the
crosstalk to be eliminated. Hence, the complexity of the QCMLSE
decoder, that increases geometrically versus the number of
additional ISI points, is reduced.
[0025] The method achieves more bandwidth efficient data
transmission using QVSB signaling. Modulator and demodulator
hardware structures implementing the method enable improved
bandwidth efficient communications. These modem structures include
SNR efficient synchronization loops that will substantially
outperform brute force squaring circuitry. The method preferably
relies upon transmit and receive data filtering, specialized QVSB
spectra generation, the QCMLSE Viterbi decoding, and a coherently
aiding demodulator synchronization loop.
[0026] The symbol integer spaced zeros in the quadrature impulse
response as well as the inphase response are preferably realized by
jump filtering. The ISI removal at 6 (.+-.3) symbols can be
realized through multiplication of the impulse responses by a time
domain cosine waveform with a 6.+-.3 symbol period. When the time
domain cosine multiplication is performed on a sin(x)/(x) response,
the corresponding ideal rectangular single jump spectra is shifted
up and down resulting in the double jump spectrum. The ideal
rectangular bandwidth is expanded by 1/6th. Similarly, the ISI at
.+-.5, .+-.7 symbols can be eliminated by further multiplication
and spectral shifting, resulting in the quadruple jump and octal
jump spectra, respectively. A less complex alternative, that also
results in greatly improved ISI, can be realized by smoothing the
transitions of the double jump spectrum. The ISI at .+-.1 symbols
can not be removed because a doubling of the bandwidth would
result. Hence, whenever a data pulse is transmitted, the pulse will
be subjected to and dominated by controlled ISI from the adjacent
symbols in the opposing quadrature channel and this ISI will be of
approximate relative magnitude .+-.0.5. The QCMSLE provides for
effective Viterbi decoding that minimizes the effect of the
controlled ISI for improved bandwidth efficient communications.
These and other advantages will become more apparent from the
following detailed description of the preferred embodiment.
BRIEF DESCRIPTION OF THE DRAWINGS
[0027] FIG. 1 is a schematic of a quadrature vestigial sideband
(QVSB) modulator and demodulator system.
[0028] FIG. 2A is a 25% raised cosine QVSB eye diagram of the
filtered output of a quadrature channel. FIG. 2B is an octal jump
QVSB eye diagram of the filtered output of a quadrature
channel.
[0029] FIG. 3A is a frequency domain plot of a double jump spectra
of quadrature arm filters.
[0030] FIG. 3B is a frequency domain plot of a quadruple jump
spectra of quadrature arm filters.
[0031] FIG. 3C is a frequency domain plot of an octal jump spectra
of quadrature arm filters.
[0032] FIG. 3D is a frequency domain plot of a smoothed double jump
spectra of quadrature arm filters.
[0033] FIG. 4 is a block diagram of a generalized QVSB
receiver.
[0034] FIG. 5 is a block diagram of an analog feedback QVSB
receiver.
[0035] FIG. 6 is a block diagram of a digital feedback QVSB
receiver.
[0036] FIG. 7A is a digital data feedback sequence diagram.
[0037] FIG. 7B is a digital derivative data feedback sequence
diagram.
[0038] FIG. 7C is a digital Hilbert data feedback sequence
diagram.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
[0039] An embodiment of the invention is described with reference
to the figures using reference designations as shown in the
figures. Referring to FIG. 1, a quadrature vestigial sideband
(QVSB) modulator receives D.sub.m input data 10 that is mapped by a
quadrature data mapper 12 to generate independent inphase (I)
symbols I.sub.S and quadrature (Q) symbols Q.sub.S. A set of m
possible data signals D.sub.m 10 is mapped into I&Q symbol
values according to a constellation of modulation amplitude levels.
In the general form, the symbols I.sub.S and Q.sub.S may be any two
independent data sets. The mapper 12 receives the input data stream
D.sub.m 10 and maps the data stream 10 into m-ary symbol steam
according to an arbitrary m-ary modulation alphabet, for example, a
4-ary quadrature phase shift keying modulation alphabet. The symbol
stream is then split by projection into the I and Q constellation
axis for providing the I.sub.S and Q.sub.S symbol streams, at which
point, the I.sub.S and Q.sub.S symbol streams are statistically
independent symbol streams. Overlapping inphase and quadrature
filter responses are used to facilitate QVSB signaling. To achieve
overlapping filter responses, the I symbols I.sub.S and Q symbols
Q.sub.S are respectively communicated to modulator cross coupled
arm filters 14 and 16, and 18 and 20. The arm filters 14, 16, 18,
and 20 are near Hilbert transform filters and are grouped in
transform pairs. A first pair of near Hilbert transforms being the
h.sub.i arm filter 14 and the h.sub.q arm filter 16. A second pair
of near Hilbert transforms being the h.sub.q arm filter and the
h.sub.i arm filter 20. The h.sub.i designation designates an
inphase filter and the h.sub.q designation designates a quadrature
filter. An h.sub.i and h.sub.q filters form a Hilbert transform
pair implemented as transform filter pairs. The I symbols I.sub.S
are transformed into an i(t) filter response by the modulator
h.sub.i(t) arm filter 14 and into an i.sub.H(t) filter response by
the modulator h.sub.q(t) arm filter 16. The Q symbols Q.sub.S are
transformed into a q(t) filter response by the modulator h.sub.i(t)
arm filter 20 and into a q.sub.H(t) filter response by a modulator
h.sub.q(t) arm filter 18. The h.sub.i(t) arm filter 14 and the
h.sub.q(t) arm filter 16 are preferably a near Hilbert transform
pair operating on the I symbols I.sub.S. The h.sub.i(t) arm filter
20 and h.sub.q(t) arm filter 18 are preferably a near Hilbert
transform pair operating on the Q symbols Q.sub.S. The two h.sub.i
and the two h.sub.q arm filters 14 and 18, and and 20 are
respectively disposed in I and Q channels as band limiting filters.
The i(t) and q.sub.H(t) filter responses of respective arm filters
14 and 18 are summed by a modulator I channel summer 22 for
overlapping the i(t) and q.sub.H(t) filter responses into an
i(t)+q.sub.H(t) overlapped filter response of the I channel. The
i.sub.H(t) and q(t) filter responses of respective arm filters 16
and 20 are subtracted by a modulator Q channel summer 24 for
overlapping the i.sub.H(t) and q(t) filter responses into a
q(t)-i.sub.H(t) overlapped filter response of the Q channel. The
I&Q channel overlapping filter response signals i(t)+q.sub.H(t)
and q(t)-i.sub.H(t) in the QVSB modulator are baseband frequency
signals. Accurately defined filtering by filters 14, 16, 18 and may
be realized using digital finite impulse response filters. As such,
the arm filters 14, 16, 18 and 20 can be implemented using digital
signal processing chips, tapped delay lines with coefficient
multipliers, multiplier accumulators, or table lookups, all not
shown but well known. The use of a lookup table is possible when
the number of signal levels are deterministic and few, and the
amount of samples during digital filtering is small. With digital
filtering, D/A conversion and spectral replication removal, analog
filtering would typically be inserted before quadrature carrier
modulation.
[0040] The filter responses of filters 14, 16, 18 and 20 are
combined by the summers 22 and 24 prior to modulating quadrature
carriers. The summers 22 and 24 respectively generate the I channel
i(t)+q.sub.H(t) overlapped filter response and the Q channel
q(t)-i.sub.H(t) overlapped filter response as respective modulator
I&Q channel signals for I&Q channel quadrature modulation
of a carrier signal for transmission. A modulator carrier
oscillator 26 provides a modulator carrier signal .omega..sub.C to
a .phi. modulator ninety degree phase shifter 28 that in turn
provides a modulator cosine carrier signal cos(.omega..sub.Ct) to a
modulator I channel mixer 30 and provides a modulator sine carrier
signal sin(.omega..sub.Ct) to an modulator Q channel mixer 32 for
respectively modulating the cosine and sin I&Q channel carriers
by the modulator I channel signal and the modulator Q channel
signal for generating modulated I&Q channel quadrature signals.
The modulated I&Q channel quadrature signals from the mixers 30
and 32 are orthogonal quadrature signals. The modulated I&Q
channel quadrature signals from the mixers 30 and 32 are summed
together by a modulator quadrature summer 34 for providing a
modulated I&Q quadrature signal that is filtered by a modulator
band pass filter (BPF) 36 providing an intermediate frequency (IF)
quadrature vestigial sideband (QVSB) signal 38. The IF QVSB signal
could be further upconverted into a radio frequency (RF) QVSB
signal. When the I.sub.S and Q.sub.S symbol signals are passed
through near Hilbert transform pairs, approximately half of the
frequency spectra is canceled for spectra cancellation resulting in
the vestigial sideband transmitted signal 38 that is transmitted
with a gradual sideband rejection. The transmitted signal 38 is an
s(t) quadrature vestigial sideband signal defined as
s(t)=[i(t)+q.sub.H(t)]cos(.omega..sub.Ct)-[i.s-
ub.H(t)-q(t)]sin(.omega..sub.Ct).
[0041] Slightly less than one half of the signal spectrum of the
transmitted signal is cancelled when the transform filter pairs 14
and 16, and 18 and 20, are near Hilbert transform pairs. The signal
38 is a transmitted quadrature signal 38 that can be either an
upper or lower vestigial single sideband signal. For example, the
IF QVSB signal 38 may be an upper sideband signal (USB).
[0042] The QVSB demodulator receives the transmitted signal 38 as a
QVSB received signal that is filtered by a demodulator, band pass
filter (BPF) 40 and then translated to baseband by a demodulator I
channel mixer 42 and a demodulator Q channel mixer 44 respectively
using a demodulator cosine carrier signal cos(.omega..sub.Ct) and a
demodulator sine carrier signal sin(.omega..sub.Ct). A phase
synchronized demodulation carrier signal .omega..sub.C is provided
by a carrier synchronizer 46. The demodulation carrier signal
.omega..sub.C is phase shifted by a .phi. demodulator ninety degree
phase shifter 48 generating the demodulator cosine and sine carrier
signals respectively communicated to demodulator I&Q channel
mixers 42 and 44. The received signal 38 is demodulated by the
mixers 42 and 44 respectively using the cosine and sine carrier
signals into the respective I&Q channel overlapping filter
response signals i(t)+q.sub.H(t) and q(t)-i.sub.H(t) in the QVSB
demodulator. The modulator I&Q channel overlapping filter
response signals i(t)+q.sub.H(t) and q(t)-i.sub.H(t) in the QVSB
demodulator are demodulated into baseband frequency signals. The
demodulator I&Q channel overlapping filter response signals
i(t)+q.sub.H(t) and q(t)-i.sub.H(t) are then respectively
communicated to demodulator cross coupled arm filters 50 and 52,
and 54 and 56. The demodulator h.sub.i(t) arm filter 50 and the
demodulator h.sub.q(t) arm filter 52 are a near Hilbert transform
filter pair. The demodulator h.sub.i(t) arm filter 54 and the
demodulator h.sub.q(t) arm filter 56 are a near Hilbert transform
filter pair. The QVSB demodulator h.sub.i(t) arm filters 50 and 56
are preferably identical to the QVSB modulator h.sub.i(t) arm
filters 14 and 20, and each other. The QVSB demodulator h.sub.q(t)
arm filters 16 and 18 are preferably identical to the QVSB
modulator h.sub.q(t) arm filter filters 52 and 54, and each other.
The preferred arm filter matching is for matched transmitter and
receiver modem filtering. The demodulator h.sub.i(t) arm filter 50
provides a matched filter response (t)+{tilde over (q)}.sub.H(t).
The demodulator h.sub.q(t) arm filter 52 provides a matched filter
response -{tilde over (q)}(t)+.sub.H(t). The demodulator h.sub.i(t)
arm filter 56 provides a matched filter response -{tilde over
(q)}(t)-.sub.H(t). The demodulator h.sub.i(t) arm filter 54
provides a matched filter response {tilde over (q)}.sub.H(t)+(t).
The matched filter responses (t)+{tilde over (q)}.sub.H(t) from the
arm filters 50 and 54 are summed by a demodulator I channel summer
58 to provide an I channel response signal {tilde over ({tilde over
(i)})}(t) that is equal to 2[(t)+{tilde over (q)}.sub.H(t)]. The
matched filter response signal -{tilde over (q)}(t)-.sub.H(t) from
the arm filter 52 is subtracted from the matched filter response
signal {tilde over (q)}(t)-.sub.H(t) from the arm filter 56 to
provide a Q channel response signal {tilde over ({tilde over
(q)})}(t) that is equal to 2[{tilde over (q)}(t)-.sub.H(t)]. The
I&Q channel response signals {tilde over ({tilde over (i)})}(t)
and {tilde over ({tilde over (q)})}(t) are communicated to a
quadrature crosstalk maximum likelihood sequence estimator (QCMLSE)
62 that provides a data estimate {circumflex over (D)}.sub.m 64 of
the original input data D.sub.m 10. The QCMLSE also provides
necessary {circumflex over (Q)}.sub.k-n and .sub.k-n delayed
synchronization signals for controlling the carrier synchronizer 46
in a closed synchronization loop. In the preferred form, the
synchronization signals {circumflex over (Q)}.sub.k-n and .sub.k-n
represent the estimated data {circumflex over (D)}.sub.m 64 but
delayed in time for proper synchronized coherent demodulation in
the demodulator. The closed synchronization loop comprise the
mixers 42 and 44, arm filters 50, 52, 54, and 56, summers 58 and
60, the QCMLSE 62, carrier synchronizer 46 and the phase shifter 48
for coherent demodulation of the received signal 38.
[0043] The QCMLSE 62 receives the {tilde over ({tilde over (i)})}
and {tilde over ({tilde over (q)})} I&Q channel responses where
the {tilde over ({tilde over (i)})} response has a reconstituted
inphase component of the I.sub.S symbol set plus an undesirable
{tilde over (q)}.sub.H quadrature crosstalk component, and where
the {tilde over ({tilde over (q)})} response has a reconstituted
{tilde over (q)} quadrature component of the Q.sub.S symbol set
plus an undesirable .sub.H inphase crosstalk component. The QCMLSE
62 operates upon the {tilde over ({tilde over (i)})} and {tilde
over ({tilde over (q)})} responses, and more particularly upon the
inphase component and {tilde over (q)} quadrature component to
estimate the data estimate {circumflex over (D)}.sub.m. By adding
two positive or {tilde over (q)} for the I&Q channels, the
inphase and quadrature or {tilde over (q)} components are enhanced
by doubling as indicated by the factor of two in the {tilde over
({tilde over (i)})} (t)=2[(t)+{tilde over (q)}.sub.H(t)] and {tilde
over ({tilde over (q)})}(t)=2[{tilde over (q)}(t)-.sub.H(t)]
I&Q channel responses. The doubling of the and {tilde over (q)}
components provides enhanced SNR of the I&Q channel signals,
but the doubling addition also serves to double the unwanted
Hilbert transform crosstalk components {tilde over (q)}.sub.H and
.sub.H. However, the QCMLSE 62 can accurately estimate the data
estimate {circumflex over (D)}.sub.m 64 from the and {tilde over
(q)} components even in the presence of these unwanted Hilbert
transform crosstalk components {tilde over (q)}.sub.H and .sub.H.
The unwanted crosstalk components {tilde over (q)}.sub.H and .sub.H
will have a crosstalk signal structure based upon the I.sub.S and
Q.sub.S symbol sequence. This crosstalk signal structure is
superimposed upon the quadrature signal structure of desired and
{tilde over (q)} components. The use of the crosstalk signal
structure and the quadrature signal structure of the channel
response {tilde over ({tilde over (i)})} and {tilde over ({tilde
over (q)})} provides a composite signal structure having adequate
detection distances to generate the data estimate {circumflex over
(D)}.sub.m from the I&Q channel responses {tilde over ({tilde
over (i)})} and {tilde over ({tilde over (q)})} using a soft
decision Viterbi based decoding process in the QCMLSE 62.
[0044] The QVSB modem, including the QVSB modulator of a QVSB
transmitter and the QVSB demodulator of a QVSB receiver, is
characterized by the cross coupled arm filters that are near
Hilbert transform pairs in both the transmitting QVSB modulator and
the receiving QVSB demodulator. The near Hilbert transform pairs
are used for creating quadrature cross coupled vestigial sideband
signaling and function to provide a sufficient demodulating
detection distance for QCMLSE estimation of the data estimate
{circumflex over (D)}.sub.m 64. The arm filters cross couple signal
components between the I&Q channels so that the spectrum of the
transmitted and receive signal 38 is more compact within a given a
bandwidth for improved bandwidth efficiency but requiring the use
of both I&Q channels.
[0045] The h.sub.i and h.sub.q impulse responses are even and odd
time functions, respectively. The Hilbert transform arm filters
h.sub.i and h.sub.q provide complimentary channel waveforms that
are superimposed by the summers 22 and 24. When a symbol set, such
as the inphase I.sub.S or Q.sub.S, is filtered through a pair
cascaded identical matched filters, such as the modulator
h.sub.i(t) arm filter 14 and the demodulator h.sub.i(t) arm filter
50, the signal becomes a matched filtered signal, such as the (t)
component response of the arm filter 50. The matched filter
responses (t) and {tilde over (q)}(t) are respectively provided on
the I&Q channel for subsequent {circumflex over (D)}.sub.m data
detection or estimation.
[0046] The desired and {tilde over (q)} components can be
respectively separated from the undesired crosstalk {tilde over
(q)}.sub.H and .sub.H components by the quadrature data estimator
62 because the I.sub.S and Q.sub.S symbol signals have been passed
through cascaded cross coupled arm filters that are chosen to
minimize the intersymbol interference (ISI).
[0047] The bandwidth efficiency of vestigial sideband (VSB)
transmission is effectively doubled when independent data streams
are placed on the I&Q filtering arms during QVSB signaling. The
QVSB baseband filtering will result in crosstalk ISI in both
channels because each I&Q channel has both a conventional even
impulse response as well as near Hilbert odd impulse response
component. Intersymbol interference is the superposition of time
overlapping impulse responses, which last longer than a symbol
interval, of adjacent symbols on one another. The arm filters are
chosen so that at multiple symbol time detection points, the
intersymbol interference is zero. The Hilbert components have ISI
at odd symbol multiples. The minimum separation detection distance
between the transmitted signals will be adversely affected by the
crosstalk ISI. The arm filters are designed to minimize this
crosstalk ISI. In the QVSB demodulator, contributions from both
I&Q channels are summed so as to generate and increase the
detection SNR by 3 dB. The cross coupling of the I.sub.S and
Q.sub.S symbols inject cross coupled components serving as ISI.
However, in the QVSB demodulator, the ISI crosstalk persists in the
I&Q channel signals providing symbol detection ambiguity that
can be substantially reduced by a Viterbi decoding algorithm. The
unwanted components .sub.H and {tilde over (q)}.sub.H are produced
in a predetermined manner with a predetermined crosstalk signal
structure to aid in soft decision Viterbi decoding. The type of arm
filter used can increase the detection distance for improved
Viterbi decoding.
[0048] Referring to FIGS. 1, 2A, 2B, 3A, 3B, 3C, and 3D, various
types of filters could be used as the arm filters. The arm filters
provide crosstalk signal structures and quadrature signal structure
possessing adequate detection distances for Viterbi decoding.
However, the demodulated I and Q transformed {tilde over ({tilde
over (i)})} and {tilde over ({tilde over (q)})} channel signal will
then possess ambiguously overlapping detection distances unsuitable
for conventional symbol Viterbi decoding. To improve the detection
distances, the arm filters 14, 16, 18, 20, 50, 52, 54, and 56, are
modified to have as many zero values or equally spaced axis
crossings as possible, at multiples of the symbol time. The impulse
responses and zero equal axis crossings can be realized through
selecting suitable h.sub.i and h.sub.q arm filters.
[0049] Raised cosine filtering does not provide equal zero axis
crossing resulting in small detection distances and hence poor
Viterbi decoding. When specialized Nyquist filtering is used in the
QVSB modem, crosstalk ISI results at odd integer multiple symbol
time intervals. The superposition of impulse responses for all
possible transmitted sequences can disadvantageously result in
substantial eye closure as shown for the most raised cosine
responses. The bandwidth efficiency for the 25% raised cosine
filter response is 1.6 symbol/s/Hz. The raised cosine filter, for
example, could use smooth rolloff filtering, such as with a 100%
raised cosine filter for improved detection distances. However, the
bandwidth efficiency for the 100% raised cosine filter is reduced
to 1.0 symbol/s/Hz.
[0050] A preferable way to minimize crosstalk ISI is the use of
jump spectra filtering. For example, an SSB rectangular filter,
that is effectively a 0% raised cosine filter, or a single jump
spectra filter, exhibits crosstalk ISI in the quadrature response
at odd symbol time multiples. The crosstalk at third symbol time
can be eliminated by multiplying the impulse responses by a cosine
signal of frequency R.sub.s/6, where R.sub.S is the symbol rate.
This multiplication has the effect of shifting the frequency
response up and down by R.sub.s/6 thereby creating a double jump
spectrum. The ISI at the fifth and seventh symbol times can
likewise be eliminated by multiplying by R.sub.s/10 and R.sub.s/14,
respectively. Hence, the number of jumps in the jump spectra can be
increased by multiples of two through additional stages of
multiplication. The bandwidth efficiencies for the two, four, and
eight jump spectra are 1.5 symbol/s/Hz, 1.3 symbol/s/Hz, and 1.2
symbol/s/Hz, respectively. The penalty of eliminating the crosstalk
ISI is progressively less bandwidth efficiency.
[0051] The effect of the crosstalk ISI can be minimized using
smoothed jump spectra arm filtering to smooth the transitions of
the double jump spectra. Crosstalk ISI minimization using smoothed
double jump spectra arm filtering is simpler to implement than four
or eight jump spectra filtering. The filter smoothing has the
effect of substantially softening the ISI at the fifth and seventh
symbol times. The eye openings for the spectra of octal jump
filtering are wide reflecting large detection distances for
improved Viterbi decoding with increased bandwidth efficiency,
compared against the 25% raised cosine spectral filtering having
small detection distances for the same bandwidth efficiency. The
arm filtering selected should be optimized to increase the
detection distances for improved Viterbi decoding while minimizing
the excess bandwidth required for improved bandwidth
efficiency.
[0052] The octal jump QVSB eye diagram for octal jump filtering has
nearly the same eye openings detection distances as does double
jump filtering with 100% smoothing with about the same bit error
rate (BER) and channel capacity bandwidth performance. For QVSB
transmission, the choice of filtering can be optimized for the
application. Regardless of the choice, the number of distinct
amplitude levels of the I&Q channel responses {tilde over
({tilde over (i)})} and {tilde over ({tilde over (q)})} is greater
than that required for quadrature double sideband transmission. The
number of distinct amplitude levels that must be used is a function
of the signal constellation. Any number of distinct amplitude
levels may be used during QVSB transmission depending on the
desired BER and channel capacity. As the channel capacity increases
with improved bandwidth efficiency with raised cosine filtering,
the distinct amplitude levels progressively smear together as with
the 25% raised cosine spectra. As the smearing increases, the
detection distance decreases resulting in poorer detection. For a
given transmitted power, the more distinct amplitude signal levels
for greater bandwidth efficiency, the less is the detection
distance, and hence a trade off exists between bandwidth efficiency
and BER. However, the crosstalk ISI that causes the additional
levels in the QVSB modem is controlled under the crosstalk signal
structure and the additional crosstalk amplitude levels can be to a
large extent subsequently removed during data detection using
Viterbi decoding in the sequence estimator 62. The unwanted .sub.H
and {tilde over (q)}.sub.H components have crosstalk signal
structure defined by the arm filtering providing additional
amplitude levels. The h.sub.i and h.sub.q arm filtering is chosen
so that the set of all possible distinct amplitude levels are few
in number so that data estimation can reliably estimate the data
sequence using Viterbi decoding techniques in the estimator 62.
[0053] Referring to FIGS. 1 through 4, and more particularly to
FIG. 4, the adverse effect of the controlled intersymbol
interference of the transmitted signal 38 received by a QVSB
demodulator 66 is removed by the quadrature crosstalk maximum
likelihood sequence estimator (QCMLSE) 62 that employs a Viterbi
algorithm for generating the estimated data {circumflex over
(D)}.sub.m 64. The QCMLSE 62 provides the delayed estimate
{circumflex over (D)}.sub.m 64 of the most likely transmitted
symbol sequence from parallel sets of noisy I&Q channel samples
i.sub.k and q.sub.k of the {tilde over ({tilde over (i)})} and
{tilde over ({tilde over (q)})} I&Q channel response signals
from the demodulator 66. The I&Q delayed estimates {circumflex
over (D)}.sub.m 64 are generated by the QCMLSE 62 while further
generating {circumflex over (Q)}.sub.k-n and .sub.k-n quadrature
synchronization signals for coherent demodulation in the QVSB
demodulator 66. The {circumflex over (Q)}.sub.k-n and .sub.k-n
quadrature synchronization signals can be used to drive a
demodulator carrier synchronizer such as carrier synchronizer 46
during coherent demodulation. The QCMLSE 62 operates on paired
i.sub.k and q.sub.k samples that contain the unwanted crosstalk
.sub.H (t) and {tilde over (q)}.sub.H(t) ISI components from the
other respective I or Q channel.
[0054] The QCMLSE 62 provides the delayed estimates {circumflex
over (D)}.sub.m 64 by determining the most likely transmitted
symbol sequence from the values of the parallel sets of noisy
i.sub.k and q.sub.k samples. The QCMLSE 62 is a modified version of
a conventional maximum likelihood sequence estimator that would
otherwise process the inphase and quadrature channels independently
rather than jointly based on structured soft decision Viterbi
decoding techniques. The cross coupling between the channels in the
QVSB modulator and QVSB demodulator requires that the channels be
processed jointly rather than independently by the QCMLSE 62. The
QCMLSE 62 therefore operates jointly on the {tilde over ({tilde
over (i)})} and {tilde over ({tilde over (q)})} channel response
signals that are part of a multilevel alphabet determined by the
number of distinct amplitude values.
[0055] For 4-ary QVSB {tilde over ({tilde over (i)})} and {tilde
over ({tilde over (q)})} channel response signals with crosstalk
ISI at the first symbol time, the QCMLSE 62 is a 64-state machine
with each state corresponding to the binary contents of a pair of
three symbol delay registers 70 and 72 having a possible 64-states
(4.sup.3). Each state represents one of several possible distinct
amplitude levels according to the modulation constellation being
used. For 4-ary QVSB, the number of levels may be five in-phase and
five quadrature amplitude levels corresponding to twenty five total
amplitude pairs as shown in FIG. 2B. These twenty five amplitude
combinations map to the 64-states in a trellis. Therefore, some of
these 64-states correspond to ambiguous amplitude pairs. In
principle, any number of states could be used as long as the
computing capacity is available as the number of calculations grows
geometrically with the number of delayed samples and the size of
the modulation alphabet. For example, a 9-ary signaling alphabet
used with these quadrature crosstalk filters would require a
trellis having 9.sup.3 states, that is 729 states. That is, the
estimated sequence must be chosen from a pool of M.sup.N sequences
where M is the sequence alphabet size, and N is the sequence
length. The number of states is related to the size of the signal
alphabet M and the length L of the channel according to
S=M.sup.L-1. In general, the estimated sequence {circumflex over
(D)}.sub.m must be chosen from the set s of M.sup.N sequences.
[0056] The task of finding the best estimate of the I&Q
sequence from the space of all possible sequences is equivalent to
searching for the best path through the trellis based on the
minimization of distance metrics. The Viterbi decoder algorithm
provides a very efficient means for searching this trellis for the
best path sequence using Euclidean distance minimization techniques
that maximize the probability of receiving a particular sequence
hypothesis given some observable inputs that has been obscured by
additive Gaussian noise. This minimization is known as maximum
likelihood sequence estimation and is based on conventional Viterbi
decoding as is well known by those skilled in the art.
[0057] At each symbol time k, the state of the QVSB MLSE
corresponds to one of sixty four possible hypothesis or estimates
of the last three inphase and quadrature symbols. That is, with
I.sub.k as the inphase symbol at time k and Q.sub.k as the
quadrature symbol at time k, the value of the state S.sub.k, at
time k, is [I.sub.k, I.sub.k-1, I.sub.k-2, Q.sub.k, Q.sub.k-1,
Q.sub.k-2]. The transition to a different S.sub.k+1 state at time
k+1, is [I.sub.k+1, I.sub.k, I.sub.k-1, Q.sub.k+1, Q.sub.k,
Q.sub.k-1], and is generated from new input data I.sub.k+1 and
Q.sub.k+1 into the shift registers 70 and 72 and by the shifting of
the data [I.sub.k, I.sub.k-1, Q.sub.k, Q.sub.k-1] through shift
registers 70 and 72. The transition from one state at time k to
another state at time k+1 corresponds to a particular estimate or
hypothesis of the symbol at time k+1, and this estimate is
[I.sub.k-1, Q.sub.k-1]. The space of all possible state transitions
[S.sub.1, S.sub.2, . . . , S.sub.n, . . . ], known in the art as
the modulation trellis, is an equivalent representation of all
possible data sequences.
[0058] The function of the QMLSE 62 is to reduce the total number
of possible transmitted sequences down to the most likely sequences
by using maximum likelihood techniques to reduce the number of
paths through the trellis to the most likely paths such as sixty
four most likely paths. The decoding trellis may be represented by
a 64-state machine with each state defined by the elements in the
delay registers 70 and 72. The QCMLE 62 includes a local distance
metrics loop for generating local metric distances. The local loop
includes the input shift registers 70 and 72, a history next state
generator 74, a minimum distance recursive calculator 76, a best
64-of-256 paths selector 78, and a 64-path state history updater
80. The shift registers 70 and 72 are used to generate all
allowable states that transit through the trellis.
[0059] At time n, the shift registers 70 and 72 respectively
receive as input all possible hypothesis estimates I.sub.n and
Q.sub.n from the updater 80. Shifting the possible hypothesis
estimates I.sub.n and Q.sub.n through the registers 70 and 72 to
the right places the registers 70 and 72 in one of the 64-states.
The shift registers 70 and 72 generate all possible hypothesis
states S.sub.k transiting through four allowable state transitions
through the trellis. The hypothesis states S.sub.k are passed to
the path history next state generator 74. The path history next
state generator receives permissible trellis transitions from the
64-path state history updater 80 and the sequence of 64-states
S.sub.k from the registers 70 and 72 and generates sets of local
symbol estimates [I.sub.k, Q.sub.k]. The sets of local symbol
estimates [I.sub.k, Q.sub.k] from the generator 74 and I&Q
channel samples i.sub.k and q.sub.k from the demodulator 66 are
received by the minimum distance recursive calculator 76 that
recursively calculates minimum distance metrics for generating sets
of the most likely symbols from the observed samples q.sub.k and
i.sub.k and the sets of local symbol estimates [I.sub.k, Q.sub.k]
from the path history next state generator 74.
[0060] For a QVSB implementation, a local metric calculation is
based on three consecutive i.sub.k and q.sub.k samples at k symbol
time of the {tilde over ({tilde over (i)})} and {tilde over ({tilde
over (q)})} channel response signals, that is, six samples, because
each channel symbol is affected by crosstalk from the adjacent
symbols in the other respective channel. The QMLSE 62 operates on
the transmitted inphase and quadrature signals that will take on
discrete amplitude values at sample times that may be approximated
by a small number of discrete points. These discrete points can be
visualized as the center of the amplitude clusters corresponding to
the sample time in the eye diagrams. At any given k representing
some multiple of the symbol time, these discrete amplitude values
are a known function of the transmitted inphase and quadrature data
sequences I.sub.S and Q.sub.S.
[0061] The sets of most likely symbols from minimum distance
recursive calculator 76 are received by the best 64-of-256 paths
selector 78 that prune improbable paths through the trellis to
generate the best 64-paths, that is, the most probable 64-paths
through the trellis. The path pruning process that is performed by
the best 64-of-256 paths selector 78 is accomplished by simply
sorting the distance calculations performed by minimum distance
recursive calculator 76 and choosing those paths corresponding to
the minimum distances.
[0062] The best 64-paths from the selector 78 is fed to the 64-path
state history updater 80 for updating the local history of the best
64-paths through the trellis. The 64-path state history updater 80
stores the trellis and allowable trellis transition information
that is communicated to the path history next state generator 74
upon initialization of the QCMLSE 62. During operation, the 64-path
state history updater 80 updates the trellis with new sequence
estimates as the updater 80 generates all possible hypothesis
estimates I.sub.k and Q.sub.k.
[0063] The updated 64-path history is fed to updater 80, which
generates the hypothesis estimates I.sub.k and Q.sub.k These
estimates are shifted through the delay registers 70 and 72 and
used by the path history next state generator 74 to generate the
sets of local symbol estimates [I.sub.k, Q.sub.k]. These local
estimates are subsequently communicated to the minimum distance
calculator 76.
[0064] The most probable paths from the updater 80 in combination
with the previous states from the delay registers 70 and 72 are fed
to the generator 74 for providing a most probable indication of the
next state. The most probable next state and current samples
q.sub.k and i.sub.k are used by the calculator 76 for computing the
minimum distance metrics. The local distance metric loop 70, 72,
74, 76, 78 and 80 recursively computes the minimum metric distance
of the trellis while receiving the observed samples q.sub.k and
i.sub.k. In operation, the best path of the trellis stored in the
updater 80 is maintained as the calculator 76 generates the minimum
local distance metrics. In the local distance metric loop, a
pruning decision is made by the selector 78 upon surviving trellis
paths that are pruned from the total number of possible paths. This
pruning decision is impacted by the constraint that only one fourth
of the total number of available states can lead into the current
state that in turn can only proceed to one-fourth of the total
possible next available states. The trellis path with the smallest
combined metric is then selected by following each path back a
sufficient number of symbols through the trellis paths thereby
minimizing the BER.
[0065] The recursively calculated distance metrics from the
calculator 76 are fed to a global distance metric processor
including the extended path distance metric updater 82, a path
distance metric eliminator 84 and a delayed data symbol decision
processor 85. The minimum local distance metrics from the
calculator 76 are received by the extended path distance metric
updater 82. The local distance metrics from the calculator 76 are
added to running global distance metrics in the updater 82. The
updater 82 modifies distance metrics of the most likely symbol
sequence. The path distance metric eliminator 84 reduces the number
of most likely sequences. Each local metric calculation by the
calculator 76 is based on three consecutive q.sub.k and i.sub.k
samples, that is, six samples because each channel symbol is
affected by crosstalk from the adjacent symbols in the other
channel. The global distance metrics are reduced by the eliminator
84 and a decision is made by the processor 85 as to the most
probable, that is, an estimate of the current symbol so as to
provide estimated symbol sequences {circumflex over (Q)}.sub.k and
.sub.k at k symbol times of the input symbol sequence I.sub.S and
Q.sub.S. The processor 85 introduces a multiple constraint length
delay to the symbol time for synchronized coherent demodulation.
The estimated symbol sequences {circumflex over (Q)}.sub.k and
.sub.k generated by the processor 85 are fed into a timing buffer
86 that combines the estimated symbol sequences {circumflex over
(Q)}.sub.k and .sub.k into the estimated data {circumflex over
(D)}.sub.m of the D.sub.m input data. The timing buffer 86 also
provides the estimated sequences {circumflex over (Q)}.sub.k and
.sub.k as synchronizing timing signals communicated to the QVSB
demodulator 66 for coherent demodulation.
[0066] In the global processor 80, 82, 84, and 85, a symbol
decision is made by the processor 85 for generating the estimated
symbol sequences {circumflex over (Q)}.sub.k and .sub.k. The best
path selection process in the global processor 80, 82, 84 and 85
provides bursts of optimally detected symbol sequences {circumflex
over (Q)}.sub.k and .sub.k. The timing buffer 86 is necessary to
regulate a feedback flow of uniform I&Q channel timing signals
{circumflex over (Q)}.sub.k and .sub.k to the QVSB demodulator for
synchronization of the coherent demodulation. Hence, The QCMLSE 62
receives the I&Q channel samples and provides data estimate
{circumflex over (D)}.sub.m that are synchronously generated along
with the I&Q channel signal {circumflex over (Q)}.sub.k and
.sub.k for coherent demodulation through close loop
synchronization.
[0067] Referring to FIGS. 1 through 6, and more particularly to
FIGS. 5 and 6, an analog or a digital version of the QVSB
demodulator offer enhanced operations to process the received
signal 38 with coherent synchronized demodulation. Amplitude level
control, phase coherent recovered carrier control and symbol timing
reference control are preferred synchronized operations necessary
to make reliable symbol decisions. These synchronization operations
enable efficient coherent recovery of the transmitted symbol
sequences where the symbol decisions by the QCMLSE 62 and the
synchronized demodulator 66 are integrated together. The radio
frequency or intermediate frequency QVSB received signal 38 is an
input to the demodulator 66. The preferred USB IF signal 38 is
passed through a bandpass filter 100 to an automatic gain control
(AGC) amplifier 102 providing a filtered amplified received signal
to the I&Q demodulator mixers 42 and 44. The mixers 42 and 44
downconvert the filter amplified received signal to baseband,
respectively, using coherently recovered cosine
cos(.omega..sub.Ct+{circumflex over (.theta.)}t) and sine
sin(.omega..sub.Ct+{circumflex over (.theta.)}t) carriers from the
phase shifter 48. The mixer 42 provides the demodulated I channel
overlapping response i(t)+q.sub.H(t) signal to the h.sub.i arm
filter 50, and the h.sub.q arm filter 52. The mixer 44 provides
demodulation Q channel overlapping response q(t)-i.sub.H(t) signal
to the h.sub.i arm filter 56 and the h.sub.q arm filter 54. The
filters 50 and 54 provide the (t)+{tilde over (q)}.sub.H(t) signals
that are summed by summer 58 to providing the I channel signal
{tilde over ({tilde over (i)})}(t)=2[(t)+{tilde over
(q)}.sub.H(t)]. The filter 52 provide the overlapping response
-{tilde over (q)}(t)+.sub.H(t) signal and filter 56 provides the
overlapping response {tilde over (q)}(t)-.sub.H(t) signal to the
summer 60 that providing the Q channel signal {tilde over ({tilde
over (q)})}(t)=2[{tilde over (q)}(t)-.sub.H (t)]. The {tilde over
({tilde over (i)})} is sampled by a I channel sampling switch 104
as the {tilde over ({tilde over (q)})} is sampled by a q channel
sampling switch 106 for respectively providing the i.sub.k and
q.sub.k sampled input signals to the QCMLSE 62. In some
implementation with sufficiently low BER, quantizers 108 and 110
can respectively quantize the i.sub.k and q.sub.k sampled signals
for providing synchronized estimates .sub.k and {circumflex over
(Q)}.sub.k for coherent demodulation using an estimated symbol rate
clocking signal {circumflex over (R)}.sub.S. Alternatively, the
QCMLSE 62 can generate the quadrature synchronization {circumflex
over (Q)}.sub.k-n and .sub.k-n signals to timing switches 112 and
114 for respectively providing the synchronized estimates .sub.k
and {circumflex over (Q)}.sub.k.
[0068] In the analog demodulator, the estimate .sub.k is coupled to
a recovery I channel h.sub.i filter 116 and cross coupled to a
recovery Q channel h.sub.q filter 118, and, the estimate
{circumflex over (Q)}.sub.k is cross coupled to a recovery I
channel transform h.sub.q filter 120 and is coupled to a recovery Q
channel h.sub.i filter 122. Responses of filters 116 and 120 are
summed by summer 124 providing an estimated I channel overlapping
filter response (t)+{circumflex over (q)}.sub.H(t) Responses of
filters 118 and 122 are subtracted by summer 126 providing an
estimated Q channel overlapping filter response {circumflex over
(q)}(t)-.sub.H(t).
[0069] In both the analog version and the digital version, the
demodulator I&Q channel overlapping filter response signals
i(t)+q.sub.H(t) and q(t)-i.sub.H(t) respective from demodulator
mixers 42 and 44 are communicated to respective I&Q channel
amplitude recovery mixers 128 and 130 through respective I&Q
channel delays 132 and 134 providing delayed i(t)+q.sub.H(t) and
q(t)-i.sub.H(t). These two delayed i(t)+q.sub.H(t) and
q(t)-i.sub.H(t) signals are also respectively communicated to
I&Q channel symbol rate recover mixers 136 and 142 and to
I&Q channel symbol phase recovery mixers 140 and 146. The
mixers 128, 130, 136, 142, 140 and 146 are four quadrant
multipliers for generating .DELTA..tau. rate, .DELTA..theta. phase
and .DELTA.A amplitude error signals for improved coherent
demodulation.
[0070] In the analog version, the estimated filter responses
(t)+{circumflex over (q)}.sub.H(t) and {circumflex over
(q)}(t)-.sub.H(t) respectively from summers 124 and 126 are
communicated mixers 128, 146 and differentiator 138, and to mixers
130, 140 and differentiator 144. The differentiators 138 and 144 in
turn respectively provide respective estimated differentiated
I&Q channel signals {circumflex over ({dot over
(i)})}(t)+{circumflex over ({dot over (q)})}.sub.H(t) and
{circumflex over ({dot over (q)})}(t)-{circumflex over ({dot over
(i)})}.sub.H(t) that are respectively communicated to the rate
recovery mixers 136 and 142. The output of the phase recovery
mixers 140 and 146 are subtracted by a phase recovery summer 148
providing the .DELTA..theta. phase error signal to a phase recovery
loop filter 150 that drives the recovery carrier VCO 151 that in
turn drives the phase shifter 48. The outputs of the rate recovery
mixers 136 and 142 are summed by a rate recovery summer 152 provide
the rate error signal .DELTA..tau. to a rate loop filter 154 that
drives a rate clock VCO 156 providing the estimated symbol rate
signal {circumflex over (R)}.sub.S. An amplitude reference source
160 provides the amplitude reference A.sub.ref signal that is
subtracted from the sum of the outputs of the recovery amplitude
mixers 128 and 130 by an amplitude recovery summer 158 providing
the amplitude error signal .DELTA.A. The amplitude error signal is
fed through an amplitude loop filter 162 that provides a DC signal
for adjusting the gain of the AGC amplifier 102. The digital
version of the improved demodulator has a similar structure to the
analog version, including the mixers 128, 130, 140, 142, 146, 148,
delays 132 and 134, summers 148, 152, and 158, loop filters 150,
154, and 162, reference source 160, VCOs 151 and 156. However, in
the digital version, I&Q recovery delays 164 and 166
respectively replace the filters 116 and 122 of the analog version,
and, digital Hilbert transform filters 168 and 170 respectively
replace the filters 122 and 120 of the analog version, and, digital
summers 172 and 174 respectively replace summer 124 and 126 of the
analog version, and, digital differentiators 178 and 184
respectively replace differentiator 138 and 144 in the analog
version, with the addition of I&Q digital delays 176 and 180,
and 182 and 186. The I&Q channel digital delays 176 and 180,
and 182 and 186 respectively delay I&Q estimated signals
.sub.k+{circumflex over (Q)}.sub.k.sup.H and {circumflex over
(Q)}k-.sub.k.sup.H respectively to mixers 128 and 146, and 130 and
140 for proper synchronized timing during coherent recovery
demodulation. The digital differentiators 178 and 184 respectively
provide estimated differentiated signals {circumflex over ({dot
over (I)})}.sub.k+{circumflex over ({dot over (Q)})}.sub.k.sup.H
and {circumflex over ({dot over (Q)})}.sub.k-{circumflex over ({dot
over (I)})}.sub.k.sup.H to the rate recovery mixers 136 and 142.
The analog and digital demodulator similarly operate to provide
three concurrent recovery closed loops using phase rate and
amplitude summers 148, 152, and 158 providing the .DELTA..theta.
phase, .DELTA..tau. rate, and .DELTA.A amplitude error signals for
enabling coherent demodulation.
[0071] In close loop operation, the estimate signals, {circumflex
over (Q)}.sub.k, .sub.k, (t)+{circumflex over (q)}.sub.H(t),
{circumflex over (q)}(t)-.sub.H(t), {circumflex over ({dot over
(i)})}(t)+{circumflex over ({dot over (q)})}(t), {circumflex over
({dot over (q)})}(t)-{circumflex over ({dot over (i)})}.sub.H(t),
.sub.k+{circumflex over (Q)}.sub.k.sup.H, {circumflex over
(Q)}.sub.k-.sub.k.sup.H, {circumflex over ({dot over
(I)})}.sub.k+{circumflex over ({dot over (Q)})}.sub.k.sup.H, and
{circumflex over ({dot over (Q)})}.sub.k-{circumflex over ({dot
over (I)})}.sub.k.sup.H, are integrally synchronized, to maintain
the symbol rate {circumflex over (R)}.sub.S of the rate VCO 156,
the amplitude of the filter amplitude received signal of the
amplifier 102, and the carrier reference of the carrier VCO 151.
The delay of delays 132 and 134 matches the time required to
generate correlated data feedback of the estimate signals
(t)+{circumflex over (q)}.sub.H(t), {circumflex over
(q)}(t)-.sub.H(t), {circumflex over ({dot over
(i)})}(t)+{circumflex over ({dot over (q)})}.sub.H(t), {circumflex
over ({dot over (q)})}(t)-{circumflex over ({dot over
(i)})}.sub.H(t), .sub.k+{circumflex over (Q)}.sub.k.sup.H,
{circumflex over (Q)}.sub.k-.sub.k.sup.H, {circumflex over ({dot
over (I)})}.sub.k+{circumflex over ({dot over (Q)})}.sub.k.sup.H,
and {circumflex over ({dot over (Q)})}.sub.k-{circumflex over ({dot
over (I)})}.sub.k.sup.H. In this manner, like received symbols are
multiplied together. The QCMLSE 62 operates to detect the estimated
symbols {circumflex over (Q)}.sub.k and .sub.k, and, hence the
demodulator need to have delays for a sufficient number of data
symbol times using internal delays to have negligible impact on the
bit error ratio. As a result, the bandwidth of the three tracking
loops through summers 148, 152 and 158 has to be relatively narrow
compared to the symbol rate {circumflex over (R)}.sub.S, for
demodulation stability. For applications where a narrow tracking
bandwidth is not appropriate, as well as for most acquisition
situations, the QCMLSE 62 can be bypassed through the quantizers
108 and 110 where the sample values are effectively detected symbol
by symbol with an incumbent sacrifice in SNR. Hence, the QCMLSE 62
or the quantizers 108 and 110 feed post detected estimated data
symbol samples {circumflex over (Q)}.sub.k and .sub.k to drive the
three phase, rate and amplitude synchronized tracking loops.
[0072] The demodulators provide enhanced amplitude tracking,
carrier tracking, and symbol rate tracking under feedback control
operation of the demodulator to a steady state condition. The
purpose of the filters 116, 118, 120 and 122, as well as the
digital replacements of the digital version, is to recreate
effectively noiseless correlation waveforms to remove the data
modulation from the received signal 38. In this manner, the phase,
rate, and amplitude tracking loops can operate on the fundamental
components of received signal amplitude, carrier phase, and symbol
timing.
[0073] In the analog version, for amplitude synchronized feedback
using the automatic gain control synchronization loop, the tracking
loop error signal .DELTA.A varies the amplitude of the input signal
38 from the filter 100 to maintain a constant value. The delayed
mixer outputs of delays 132 and 134 are multiplied using mixer 128
and 130 by fed back estimated replica waveforms effectively
yielding a signal squared operation. Both of the I&Q channel
overlapping filter signals i(t)+q.sub.H(t) q(t)-i.sub.H(t) are
combined by recovery amplitude summer 158 to improve the tracking
loop SNR by 3.0 dB. The result is lowpass filtered to yield a DC
component proportional of the received signal to the difference
between the received signal level and a reference value .DELTA.A.
When the difference is zero, the nominal automatic gain control
level is attained.
[0074] For phase recovery, the estimated outputs of summers 124 and
126 are multiplied using mixers 140 and 136 and added by summer 148
to provide the phase error signal .DELTA..theta.. These multiplied
components are combined and filtered at baseband for driving the
phase loop filter 150 at baseband. The carrier phase recovery
feedback is fed into the multipliers 140 and 146 as cross coupled
from the opposite I&Q channels, and the multiplier outputs are
differenced by summer 148 to form a phase detector providing the
phase .DELTA..theta.. In this manner, most of the data pattern
noise is subtracted out, and a component proportional to the sine
of the phase difference is passed through the loop filter 150.
After the data modulation removal, lockup proceeds as in a
conventional phase lock loop, such that when the sine of the phase
error is zero, carrier phase tracking has reached steady state,
when the drive signal to the carrier voltage controlled oscillator
151 remains unchanged.
[0075] For symbol rate recovery, the data feedback is different
than the other two amplitude and phase tracking loops. The delayed
mixer outputs of delays 132 and 134 are multiplied in rate recovery
mixers 136 and 142 by the time derivatives of differentiators 138
and 144 of the estimated +{circumflex over (q)}.sub.H and
{circumflex over (q)}-.sub.H feedback providing the estimated
derivative signals {circumflex over ({dot over (i)})}+{circumflex
over ({dot over (q)})}.sub.H and {circumflex over ({dot over
(q)})}-{circumflex over ({dot over (i)})}.sub.H. The derivative is
typically a maximum when the nonderivative multiplier input goes
through a zero crossing. Hence, leading or lagging zero crossings
are converted to either positive or negative going DC offsets that
drive the rate loop to center the symbol timing rate. When the zero
crossings are centered, the net DC offset is zero, and the symbol
timing rate has reached a steady state value.
[0076] Referring to FIGS. 1 through 7, and more particularly to
FIGS. 6 and 7, the digital version of the demodulator can be
realized using digital equivalent components. The digital
embodiment includes QVSB synchronization with digital data feedback
that enables a digital signal processing embodiment of the QVSB
modem. In many applications, all of the hardware except the mixers
and phase shifters can be implemented with digital signal
processing technology. On the transmit side, D/A converters, not
shown, would typically be placed between the summed QVSB data
filters and the mixers. On the receive side, A/D converters, not
shown, would reside in the complementary position between the
mixers and the cross coupled h.sub.i and h.sub.q filters 50, 52,
54, and 56. Also, D/A converters, not shown, would be needed before
the VCO 151, or afterwards when a numerically controlled oscillator
(NCO) is used instead.
[0077] Referring generally to all of the figures, there are several
ways in which QVSB data demodulation and decoding may be realized
in hardware including the data feedback structure. For example, the
QCMLSE 62 may not be used for synchronization, but rather the
quantizers 108 and 110 are used to generate feedback estimates
{circumflex over (Q)}.sub.k and .sub.k. This quantization
estimation approach simplifies the hardware and removes some of the
additional delay from the tracking loops. The digital feedback
variations can be created from the QCMLSE data feedback whose
{circumflex over (Q)}.sub.k-n and .sub.k-n signals are discrete
amplitude estimates of the transmitted amplitude levels. In the
absence of filtering, the {circumflex over (Q)}.sub.k-n and
.sub.k-n estimates are not smooth, but digitized boxcar waveshapes
of fixed amplitude over the duration of the sample rate could be
used. The .DELTA./.DELTA.t time derivatives of differentiators 178
and 184 are also discrete amplitude estimates proportional to the
level differences in the non-derivative I.sub.k and Q.sub.k. The
{circumflex over ({dot over (Q)})}.sub.k and {circumflex over ({dot
over (I)})}.sub.k time derivative of the {circumflex over
(Q)}.sub.k and .sub.k estimates must also be advanced by half a
symbol time relative to the nonderivatives, to establish the proper
timing relationship between the incoming signals and the fed back
time derivatives, and hence the use of delays 176, 180, 182 and
186, as shown in FIG. 7.
[0078] The digital version preferably uses the digitized Hilbert
transforms 170 and 168. The Hilbert transforms 168 and 170 are
appropriate for SSB and VSB use because the h.sub.q quadrature
filter is approximately the Hilbert transform of the h.sub.i
inphase filter. The Hilbert transforms may be realized as a table
lookup because the dominant IR samples from the h.sub.q quadrature
filter can be limited to the .+-.1 symbol positions. Thus, a pulse
of inphase feedback generates two quadrature feedback pulses of
alternating polarity, at the adjacent symbol positions, with
approximately half the inphase pulse amplitude. Hence, delays 164
and 166 are used to time align the {circumflex over (Q)}.sub.k and
.sub.k estimates with the transform filter signal from the digital
Hilbert transforms 170 and 166. Because the Hilbert transform
output must be delayed at least one symbol time, the other waveform
inputs to the synchronization loop multipliers must be delayed
accordingly using the added delays 176, 178, 180 and 186.
[0079] To reduce the number of data filters in the modulator and
demodulator in an equivalent manner, the even time filter responses
of the h.sub.i filters could be pushed back through the cross
coupling nodes where the h.sub.i and h.sub.q paths split to the
I&Q channels. Then, after the split, the h.sub.i path would be
a wire and the h.sub.q path would have a 90.degree. broadband phase
shifter. Yet another equivalent realization would be to move the
baseband data filters 36 and 40 to IF/RF, on either or both the
transmit and receive sides, and to use simple lowpass roofing
filters at baseband.
[0080] The preferred embodiments may be subject to bursts of symbol
errors due to the QCMLSE processing. The QVSB modem implementation
can be enhanced with concatenated or turbo coding with interleaving
for improved performance. Similarly, a useful extension might also
include trellis-coded modulation. The preferred embodiments use
transmit and receive baseband filtering with overlaid quadrature
signals to modulate and demodulate QVSB signals. The use of jump
spectra minimizes crosstalk and ISI during QVSB transmission. Also,
due to sensitivity to the channel transmission amplitude and phase
dispersion, active or passive equalization circuitry may be
employed.
[0081] The system is general and could be used in various
applications where bandwidth efficient digital data transmission is
important. QVSB signaling can be readily implemented with currently
available components at data rates below 100 Mbits/s. The 4-ary
QVSB can be readily extended to 8-ary and 16-ary signaling having
bandwidth efficiencies that are equivalent to 64-ary and 256-ary
conventional modulation techniques, respectively. A 4-ary QVSB
achieves the same bandwidth efficiency of conventional 16-ary QDSB
data transmission, but it requires less SNR. In a linear channel,
QVSB is substantially more efficient in bandwidth and SNR than
conventional data transmission techniques. Those skilled in the art
can make enhancements, improvements, and modifications to the
invention, and these enhancements, improvements, and modifications
may nonetheless fall within the spirit and scope of the following
claims.
* * * * *