U.S. patent application number 09/992925 was filed with the patent office on 2003-05-08 for system and method for regulating multiple outputs in a dc-dc converter.
Invention is credited to Bourdillon, Laurence.
Application Number | 20030086279 09/992925 |
Document ID | / |
Family ID | 25538891 |
Filed Date | 2003-05-08 |
United States Patent
Application |
20030086279 |
Kind Code |
A1 |
Bourdillon, Laurence |
May 8, 2003 |
SYSTEM AND METHOD FOR REGULATING MULTIPLE OUTPUTS IN A DC-DC
CONVERTER
Abstract
A multiple output flyback converter provides independently
controlled high-bandwidth secondary side voltage regulation for
rapid response to small signal changes at the multiple outputs on
the secondary side. Each of the multiple output circuits includes a
fast local feedback loop to perform rapid and precise secondary
side voltage regulation. The fast local secondary side feedback
loops compensate for small changes in the load on the order of 1 to
5 percent. By providing each output circuit with independent
secondary-side control, the last or final voltage output is
controlled from the primary side. A further feature of the
inventive circuit is that the input circuit of the converter
includes an active clamp circuit for recovering both energy in the
leakage inductance and additionally recovers residual energy in the
magnetic field of the secondary winding of the transformer at the
end of each energy cycle.
Inventors: |
Bourdillon, Laurence; (New
Milford, CT) |
Correspondence
Address: |
Michael E. Marion
U.S. Philips Corporation
580 White Plains Roads
Tarrytown
NY
10591
US
|
Family ID: |
25538891 |
Appl. No.: |
09/992925 |
Filed: |
November 5, 2001 |
Current U.S.
Class: |
363/21.12 |
Current CPC
Class: |
H02M 3/33576 20130101;
H02M 1/009 20210501; H02M 3/33561 20130101 |
Class at
Publication: |
363/21.12 |
International
Class: |
H02M 003/335 |
Claims
What is claimed is:
1. A DC-DC converter for providing independent secondary side
voltage regulation for a plurality of loads comprising: a power
transformer having at least a primary winding and a secondary
winding; an input circuit coupled to said primary winding for
coupling an input DC voltage to said primary winding; a plurality
of output channels coupled to said secondary winding, each of said
output channels having a secondary side switching element,
secondary side control means for alternately turning on and off
said secondary side switching elements for providing said
independent secondary side voltage regulation.
2. The DC-DC converter circuit of claim 1 wherein said input
circuit includes a primary side switching element, a storage
capacitor and a switching element.
3 The DC-DC converter circuit of claim 1, where said each of said
output channels further includes a rectifier diode and a smoothing
filter capacitor.
4. The DC-DC converter of claim 1, wherein said input circuit
comprises an active clamp circuit.
5. The DC-DC converter of claim 4, wherein said active clamp
circuit is adapted to recover energy in each switching cycle of
said secondary side switching elements.
6. The DC-DC converter of claim 5, wherein the energy comprises
leakage energy and excess energy stored in said secondary winding
during each switching cycle.
7. The DC-DC converter of claim 1, wherein said secondary side
control means comprises a plurality of feedback loops each
associated with a respective one of said plurality of output
channels, said plurality of feedback loops being adapted to
maintain a secondary side voltage indicative of a respective
secondary side switching element independently of a primary side
switching element voltage.
8. The DC-DC converter of claim 7, wherein each of said plurality
of feedback loops comprise a pulse width modulation (PWM) control
unit coupled to an output node and coupled to respective said
secondary side switching elements for generating the PWM signal in
response to the output voltage.
9. The DC-DC converter of claim 1, wherein the DC-DC converter is
of flyback type.
10. The DC-DC converter of claim 1, wherein the DC-DC converter is
of forward type.
11. The DC-DC converter of claim 1, wherein the primary side
switching element is A MOSFET transistor with an integral reverse
diode.
12. The DC-DC converter of claim 1, wherein the primary side
switching element is a bipolar junction transistor with an integral
reverse diode.
13. The DC-DC converter of claim 1, wherein during a charging cycle
said input DC voltage charges said power transformer with an amount
of energy sufficient to provide an independent regulated DC power
level to respective said plurality of loads and to additionally
provide a residual amount of energy to compensate for fluctuations
in said plurality of loads.
14. A method for providing independent secondary side voltage
regulation for a plurality of loads, said method comprising the
steps of: providing a DC-DC converter for providing independent
secondary side voltage regulation for a plurality of loads
comprising; providing a power transformer having at least a primary
winding and a secondary winding; providing an input circuit coupled
to said primary winding for coupling an input DC voltage to said
primary winding; providing a plurality of output channels coupled
to said secondary winding, each of said output channels having a
secondary side switching element; and providing secondary side
control means for alternately turning on and off said secondary
side switching elements for providing said independent secondary
side voltage regulation.
Description
BACKGROUND OF THE INVENTION
[0001] 1. Technical Field of the Invention
[0002] The present invention relates generally to the field of
multiple output dc-dc converters. More particularly, the present
invention relates to the regulation of multiple outputs in flyback
converters.
[0003] 2. Description of the Related Art
[0004] Conventional switching power supplies commonly include a
power transformer and one or more power switches for alternately
coupling an unregulated DC or rectified AC voltage across a primary
winding of the power transformer in a series of voltage pulses.
These pulses are converted into a series of voltage pulses across
one or more secondary windings of the power transformer and then
rectified and filtered to provide one or more output DC voltages.
The output voltage or voltages of the power converter are commonly
regulated by controlling the relative amount of time that the power
switch is on (i.e., the duty cycle).
[0005] One common type of switching power supply is the flyback
power converter, which is an isolated version of the buck-boost
converter. A flyback power converter works by cyclically storing
energy in a coupled inductor, and then dumping this stored energy
into a load. By varying the amount of energy stored and dumped per
cycle, the output power can be controlled and regulated. A
switching transistor connected in series with the primary winding
of the power transformer normally provides such a switching
function. That is, the on-time and off-time of this power switch
controls the amount of energy stored in the coupled magnetic field.
When the power switch is on, current flows through the primary
winding of the power transformer, and energy is stored in the
transformer. When the power switch is off, the stored energy is
transferred out into a secondary circuit by means of current
flowing out of one or more secondary windings of the power
transformer. Note that the secondary current does not flow in the
power transformer at the same time that the power switch is on and
the primary current is flowing. The reason for this is that in a
conventional flyback power converter, the primary-secondary winding
polarity is chosen such that a rectifier when coupled to the
secondary winding will prevent conduction of current in the
secondary winding when the power switch is on.
[0006] Flyback power converters are advantageous at lower power
levels over other switching power converters due to the fact that
they are generally simpler, they require a reduced number of
components, and they allow multiple regulated outputs to be
available from a single supply. Common applications for flyback
converters are AC adapters, which may, for example, deliver an
output voltage in the range of between 9 VDC to 180 VDC at power
levels of 20 to 100 Watts, drawing power from a rectified AC mains,
which may vary between 85 VAC to 410 VAC for universal mains
input.
[0007] Flyback converters are generally operated in one of two
modes, a first mode, referred to as discontinuous conduction mode,
well known in the art, in which the energy stored in the
transformer is totally coupled to the output load before the next
energy cycle, generally resulting in the secondary current reaching
zero before the next drive cycle. The second mode of operation is
referred to as continuous conduction mode, whereby the next energy
cycle begins before all stored magnetic energy is released from the
transformer, and therefore before the secondary current reaches
zero. Discontinuous conduction mode is more common than continuous
conduction mode because relatively simple control circuitry can be
used to maintain output voltage regulation by varying the frequency
and/or on-time of the power switch to accommodate heavy or light
load conditions.
[0008] The control circuitry for either DCM or CCM operation is
somewhat similar. In the case of CCM, the loop bandwidth must
generally be much lower to avoid instability due to the right hand
plane zero (RHPZ) from the output diode. Because of its higher
bandwidth capabilities, DCM is often preferred. For high current
applications, CCM is generally preferable because the
root-mean-square (RMS) currents (hence losses) are lower.
[0009] FIG. 1 illustrates a conventional active clamp flyback
converter having multiple secondary outputs and an active snubber
network on the primary side to protect the main switch S.sub.M. In
a conventional flyback converter, all of the outputs are `on`
simultaneously. The ratio of the output voltages on the different
windings is determined by the winding ratio and the actual relative
voltage levels are determined by the `on` time of the main
switch.
[0010] FIG. 2 is an illustration of another conventional flyback
converter having sequenced multiple secondary outputs sourced from
a single secondary winding. The circuit of FIG. 2 further includes
a passive resistor-capacitor-diode (RCD) snubber network formed by
capacitor C.sub.s, resistor R.sub.s, and Diode DS on the primary
side to protect the main switch S.sub.M. Such networks are well
known in the art.
[0011] The circuit of FIG. 2 operates in accordance with a simple
sequential pulse-width modulation (PWM) scheme for controlling the
`on` time of the respective outputs. PWM schemes are well known in
the art. A drawback associated with the circuit configuration of
FIG. 2 concerns certain limitations imposed on the output voltage
control.
[0012] In particular, if the converter operates in a continuous
mode (DCM), then all of the energy stored in the transformer T
during the primary switch S.sub.m `on` time must be transferred to
the secondary outputs 211, 213, 215 during each switching
cycle.
[0013] In the circuit, each of the outputs on the secondary side is
locally controlled with the exception of the last or final output.
The final output (e.g., output 215) is indirectly controlled by the
`on` time of the primary side switch S.sub.M. That is the `on` time
of the primary switch S.sub.M must be of such duration to guarantee
that there is sufficient energy stored in the transformer during
the charging cycle to maintain a voltage V.sub.3 across R.sub.3 at
a pre-determined level, irrespective of the `on` time of switch
S.sub.3. It is noted that irrespective of how long a time switch
S.sub.3 is maintained in the `on` state, if there is insufficient
energy stored in the transformer in the first instance, maintaining
switch S.sub.3 `on` for a longer duration cannot correct for an
initially insufficient amount of energy stored in the transformer
to maintain the third output at the proper voltage level. For
precise secondary side control, this is not a desirable limitation.
Additionally, this restriction imposes a further drawback requiring
that there be a "smart" controller on the primary side.
[0014] It is therefore an object of the present invention to
provide an improved power converter having each of a plurality of
outputs regulated exclusively from the secondary side
quasi-independent of the `on` time of the primary switch.
SUMMARY OF THE INVENTION
[0015] In accordance with the present invention, a multiple output
flyback converter is provided. The converter provides independently
controlled high gain-bandwidth secondary side voltage regulation to
compensate for rapid small signal changes at the multiple outputs
on the secondary side. In a preferred embodiment, the converter
further includes a low gain-bandwidth slow loop feedback controller
connecting the secondary and primary sides for responding more
slowly to large signal changes.
[0016] The converter according to the present invention includes a
transformer, or coupled inductors, having a primary winding and a
secondary winding. The secondary winding includes first and second
terminals. An input circuit is connected to the primary winding for
coupling an input DC voltage to the primary winding. The input
circuit includes an active clamp circuit for recovering both energy
in the leakage inductance, as is performed in prior art circuits,
and in addition, recover residual energy in the magnetic field of
the secondary winding of the transformer at the end of each energy
cycle.
[0017] In accordance with one embodiment, multiple identically
constructed output circuits are connected in parallel to the first
and second terminals of the secondary winding. Each output circuit
includes a switch connected in series with a diode, an energy
storage capacitor connects the diode/switch pair to the return
lead. Each output circuit further includes a fast feedback loop
(i.e., secondary side controllers) which maintains a regulated
output voltage. The fast feedback loop compensates for small
changes in the load on the order of 1 to 5 percent.
[0018] The multiple output flyback converter of the present
invention, in addition to providing secondary side controllers for
maintaining a regulated output voltage, also requires a primary
controller for controlling the primary switch `on` time to supply
the output requirements.
[0019] An advantage of the circuit of the present invention is that
with secondary-side control, the accuracy and bandwidth of the
primary loop can be relaxed. The primary controller is responsible
for maintaining the required amount of energy `overdrive` to allow
small-signal control of the last secondary winding output. This can
be accomplished in a number of different ways. One way is to
monitor the current level in the last secondary winding at the time
of turn-off via an opto-coupler. Alternatively, the primary
controller could monitor secondary turn-off by the voltage change
on the drain of the primary switch.
[0020] Another advantage of the circuit of the present invention is
that small load changes are rapidly and precisely compensated for
exclusively on the secondary side via fast local feedback loops for
each output. This feature overcomes the limitation of the prior art
in which the last or final output is controlled from the primary
side.
[0021] A further advantage of the circuit of the present invention
is that the local feedback loops on the secondary side provide more
precise voltage control than can be achieved with prior art
configurations.
[0022] A still further advantage of the circuit of the present
invention is that excess or residual energy stored in the secondary
winding of the coupled winding is returned to the source via the
primary circuit via the active clamp network. A still further
advantage of the circuit of the present invention is that
soft-switching of primary side switches is achieved via the active
clamp.
BRIEF DESCRIPTION OF THE DRAWINGS
[0023] The foregoing features of the present invention will become
more readily apparent and may be understood by referring to the
following detailed description of an illustrative embodiment of the
present invention, taken in conjunction with the accompanying
drawings, where:
[0024] FIG. 1 is a circuit diagram illustrating a conventional
active clamp flyback converter having multiple secondary outputs
and an active snubber network;
[0025] FIG. 2 is a circuit diagram illustrating a conventional
active clamp flyback converter having sequenced multiple secondary
outputs sourced from a single secondary winding;
[0026] FIG. 3 is a circuit diagram illustrating an active clamp
flyback converter in accordance with an embodiment of the present
invention;
[0027] FIG. 4a illustrates the open and closed states of switches
of the circuit of FIG. 3 during one inductor charge/discharge
cycle; and
[0028] FIGS. 4b-4d illustrate exemplary energy storage curves
during one inductor charge/discharge cycle for the circuit of FIG.
3.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0029] FIG. 3 shows in schematic detail a preferred embodiment of
the invention including a multiple output converter of flyback type
300, in which isolation transformer T1 primary winding nP connects
from positive node P9 to node M9. While secondary winding nS1
provides converter input to a plurality of similarly constructed
output circuits 311, 313 and 315. Each output circuit includes
conventional flyback converter components, including, for example,
with reference to output circuit 311, switch S1 coupled to the
anode of rectifying diode D1 and output filter capacitor C1 is
connected in parallel with diode D1 and load resistor R1.
[0030] A positive terminal of the main power source V.sub.SRC is
coupled in parallel to a first terminal of capacitor C.sub.A P7 and
a first terminal of isolation transformer T1 P9. A second terminal
of the main power source is coupled to the ground M. Switch SA is
coupled at Intermediate node M9 to one terminal of power switch
S.sub.M. Capacitor C.sub.A is coupled at a second terminal to
switch S.sub.A at node M7. Capacitor C.sub.A and switch S.sub.A
collectively form an active clamp circuit.
[0031] Further, it will be apparent to one of ordinary skill in the
art that the device of the present invention could have additional
loads and switches coupled to nodes P11 and M11. The schematic of
FIG. 3 shows only three loads, i.e., loads R1, R2 and R3, in an
exemplary embodiment of the invention and in no way limits the
number of loads the device of the present invention can
accommodate. It will be apparent to those skilled in the art that
the device of the present invention can provide output voltages to
multiple loads and modification by adding additional loads may be
made to the embodiment chosen for illustration without departing
from the spirit and scope of the invention.
[0032] Referring now to the output circuits 311, 313 and 315. The
output voltages V1, V2 and V3 supplied to the loads R1, R2 and R3,
respectively, are formed by integrating a proportional amount of
inductor current in the capacitors C.sub.1, C.sub.2 and C.sub.3,
respectively. Energy from the main power source, V.sub.SRC is
stored in the transformer T1 during the transformer charging phase.
Supplying a charge to the output capacitors C.sub.1, C.sub.2 and
C.sub.3 at the loads occurs in the discharge phase.
[0033] The configuration of the switches S.sub.1, S.sub.2 and
S.sub.3 in the present invention determines whether the circuit is
charging or discharging. During the transformer charging phase
during which the main power source, V.sub.SRC charges isolation
transformer T1, all of the switches are in the open state, except
for the switch S.sub.M. Once the transformer T1 is sufficiently
charged, the switch S.sub.M is opened. In actual operation, the
first switch in the switching sequence, e.g., switch S1, is usually
turned on somewhat ahead of the primary switch S.sub.M being turned
off (opened). In general, there may often be some overlap between
the ON and OFF times of the sequential controlled switches. This
occurs for two reasons, first, the flyback converter should not be
left "unloaded" (i.e., all outputs OFF), and second, soft-switching
can be achieved with some overlap.
[0034] During the discharge cycle, only one switch is in a closed
state at any given time so that the load coupled to the closed
switch will receive the maximum charge from the inductor. Feedback
loops 302, 304 and 306 (i.e., fast loop controllers) are
responsible for coordinating and implementing the time duration for
the open and closed states for the switches S1, S2 and S3 to insure
that a predetermined voltage level is maintained at each
output.
[0035] FIG. 4a illustrates the open and closed states of the
switches S.sub.M, S.sub.1, S.sub.2 and S.sub.3 for the circuit of
FIG. 3 during one inductor charge/discharge cycle. A full cycle is
illustrated between the time period TP1 to time period TP6 where
the transformer T1 has gone through one full charging and
discharging cycle.
[0036] Charging Phase
[0037] While primary switch S.sub.M is closed, the isolation
transformer T1 is in a charging phase. This charging phase is
denoted by the time period TP1 to TP2 during which time the primary
winding nP charges in accordance with the portion of the energy
curve labeled "primary charging phase" in FIG. 4c. Likewise, while
primary switch S.sub.M is open, the isolation transformer T1 is in
a discharging phase as denoted by the time period TP2 to TP6 and
further in accordance with the portions of the energy curve labeled
{Sec1, Sec2, Sec3}. During the time period TP2 to TP6, only one
load is being charged at a time, and the loads R.sub.1, R.sub.2 and
R.sub.3 can receive different amounts of charging time based on
instantaneous load as measured by the respective feedback loops
312, 314 and 316.
[0038] That is, each of the output circuits 311, 313 and 315
requires a charging time sufficient to ensure that the final output
is being properly maintained. To do so, feedback loops 302, 304 and
306 detect the instantaneous voltage level and adjust the ON/OFF
times of respective switches S.sub.1, S.sub.2 and S.sub.3 to
maintain a predetermined voltage level. As an example, the feedback
loop 302 associated with output circuit 311 feeds back some measure
of the voltage V.sub.1 across load R.sub.1 to control switch
S.sub.1 so as to maintain the output voltage V.sub.1 at a
predetermined level. The time required to properly maintain the
voltage at a predetermined level is shown as the time period TP2 to
TP3. Similarly, the feedback loop 304 associated with output
circuit 313 feeds back some measure of the voltage V.sub.2 across
load R.sub.2 to control switch S.sub.2 and so on for the third or
more output circuits.
[0039] Discharge Phase
[0040] Time period TP2 defines the start of the discharge phase. At
this time, switch S.sub.M changes to an open state and switch
S.sub.1 changes to a closed state. For the time period TP2 to TP3,
the switch S.sub.1 remains closed and the load R.sub.1 is charged
as illustrated by the portion of the energy curve labeled Sec 1 of
FIG. 4b. The slope of the current decay is a function of the
inductance and the voltage across R.sub.1.
[0041] For the time period TP3 to TP4, the switch S.sub.2 is in a
closed state, and the switch S.sub.1 is in an open state which
results in the load R.sub.2 being charged as illustrated in the
portion of the energy curve labeled Sec 2 in FIG. 4b.
[0042] For the time periods TP4 to TP5, the switch S.sub.3 is in a
closed state, and switches S.sub.1 and S.sub.2 are in an open state
which results in the load R.sub.3 being charged as illustrated in
the portion of the energy curve labeled Sec 3 of FIG. 4b.
[0043] With reference to FIG. 3, there is also shown a slow loop
feedback controller 320 connecting the secondary and primary
circuits. It is contemplated that the slow loop controller may
serve one of two functions: (1) to perform low-bandwidth coarse
control that attempts to compensate for large signal changes at the
output. This process would occur over a much longer duration (i.e.,
hundreds of cycles) as compared with the fast loop controllers
which compensate within a few cycles. The slow-loop controller
generally operates as a standard feedback loop by sensing the
output voltage and controlling the ON times of the primary side
switches accordingly, (2) Use as a synchronizing signal to
synchronize the primary and secondary circuits. This may be
required because in certain energy cycles, the final output circuit
stabilizes sooner than in other cycles. The primary circuit must
have some way of knowing that the final secondary circuit has
stabilized or that the last secondary circuit in the sequence has
turned off.
[0044] In a preferred mode of operation, the circuit is operated to
compensate for load variations on the order of 1 to 5 percent. If
additional compensation is required, then such compensation may be
provided via the slow loop controller, illustrated in FIG. 3 as
element 320. As shown, the slow loop controller 320 bridges the
secondary and primary circuits to control primary side switches
S.sub.A and S.sub.M to thereby supply the output requirements.
While the slow loop controller 320 will compensate for larger
signal changes, as contrasted with the fast loops (e.g., 302, 304,
306) which can compensate for smaller instantaneous load
changes.
[0045] An important aspect of the present invention is that precise
and rapid small signal, secondary side control is performed for
each output circuit (e.g., 311, 313, 315) via the associated
fast-loop feedback circuits (e.g., 302, 304, 306). The rapid
secondary side control rapidly compensates for instantaneous load
changes.
[0046] In performing precise and rapid secondary side control, the
operation of the circuit of FIG. 3 results in a small amount of
residual energy at the end of each energy cycle, as shown in FIGS.
4b-4d (see returned energy areas E1, E2, E3). The residual energy
is returned back to the source in each cycle via the active clamp
circuit in the primary.
[0047] The method of performing secondary side control in
accordance with the method of the invention will now be described
with reference to FIGS. 4b through 4d.
[0048] Referring initially to FIG. 4b, the area labeled "residual
energy EI" represents energy which remains in the secondary winding
nS1 of the transformer T1 at the end of one representative energy
cycle. In each energy cycle, the energy stored in the secondary
winding nS1 during the primary charging phase is used to stabilize
the voltages at each of the respective output circuits (e.g., 311,
313, 315).
[0049] In each cycle, subsequent to each of the respective output
voltages being stabilized, there will typically remain some unused
(residual) energy after the final (e.g., third) output has been
stabilized. The average amount of residual energy at the end of a
cycle can be adjusted via the `on` time of primary switch S.sub.M.
That is, by maintaining the `on` time of primary switch S.sub.M
`on` for a longer time, a larger amount of residual energy will
result. A certain amount of residual energy is desired to provide
compensation for rapidly responding to instantaneous load changes
on the secondary side. The manner in which the residual energy is
used to compensate for the instantaneous load changes is now
described.
[0050] FIGS. 4b through 4d illustrate three cases which require
different amounts of residual energy to compensate for load
fluctuations.
[0051] FIG. 4b represents a baseline case wherein at time TP5, the
third or final output voltage level V.sub.3 stabilizes leaving an
amount of residual energy whose area is defined as E1.
[0052] FIG. 4c represents a case wherein the third or final output
voltage level V.sub.3 stabilizes at a later time TP5 than the
baseline case (i.e., TP5'>TP5). This delta in stabilization time
is indicated as delta-t1 in FIG. 4c which results in the use of
additional residual energy, E2>E1. It is noted, however, that if
the stabilization time exceeds the time T.sub.bound, then there
will be an insufficient amount of residual energy available to
provide compensation. That is, T.sub.bound represents an upper
limit in terms of compensation whereby all available residual
energy is consumed at this time.
[0053] FIG. 4d represents a case where the third or final output
voltage level V.sub.3 stabilizes at an earlier time TP5"
(TP5"<TP5) than described in the baseline case. This case
represents the situation where the instantaneous change in the load
requires less energy than that required in the baseline case (i.e.,
E1<E3). This delta in stabilization time from the baseline is
indicated by delta-t2 in FIG. 4d.
[0054] As shown in the illustrative examples, the falling edge of
the Sec 3 `on` time in FIG. 4a can oscillate to the left (e.g.,
FIG. 4d) or right (e.g., FIG. 4c) to compensate for small
instantaneous variations in the load. Such small signal changes are
compensated for by the fast loop 306 as long as the load change
requiring additional energy does not require more energy than is
made available in the "residual energy" region by design, as
indicated by T.sub.bound in the illustrative example.
[0055] It is noted that the circuit may be alternately designed
such that no "residual energy" is provided for in each cycle. In
this case, the falling of the Sec 3 `on` time would nominally occur
at the point defined as T.sub.bound in FIG. 4b. Compensation would
then be unipolar in the sense that compensation can only be
provided for load changes requiring less than the nominal amount of
residual energy, as illustrated in FIG. 4d.
[0056] An important aspect of the present invention is that an
active clamp circuit is used in the primary to recover the residual
energy. Conventionally, an active clamp has been used in prior art
circuits to recover non-coupled leakage inductor energy and provide
soft-switching of the primary switches. The inventive circuit uses
the active clamp for this purpose and to recover the residual
energy. In recovering the residual energy, there will be some loss
due to dissipation in the clamp switch, S.sub.A. The path by which
the residual energy is recovered and transferred back to the source
in each cycle is through the primary winding nP, switch S.sub.A and
back to the source, V.sub.SRC.
[0057] Although the invention has been described with a certain
degree of particularity, various modifications may be made to the
embodiments disclosed herein. It is, therefore, to be understood
that the above description should not be construed as limiting, but
merely as an exemplification of the various embodiments.
* * * * *