U.S. patent application number 10/186306 was filed with the patent office on 2003-01-23 for time domain radio transmission system.
Invention is credited to Barnes, Mark A., Fullerton, Larry W..
Application Number | 20030016157 10/186306 |
Document ID | / |
Family ID | 27573964 |
Filed Date | 2003-01-23 |
United States Patent
Application |
20030016157 |
Kind Code |
A1 |
Fullerton, Larry W. ; et
al. |
January 23, 2003 |
Time domain radio transmission system
Abstract
A time domain communications system wherein a broadband of
time-spaced signals, essentially monocycle-like signals, are
derived from applying stepped-in-amplitude signals to a broadband
antenna, in this case, a reverse bicone antenna. When received, the
thus transmitted signals are multiplied by a D.C., replica of each
transmitted signal, and thereafter, they are, successively, short
time and long time integrated to achieve detection.
Inventors: |
Fullerton, Larry W.;
(Brownsboro, AL) ; Barnes, Mark A.; (Madison,
AL) |
Correspondence
Address: |
TIME DOMAIN CORPORATION
7057 OLD MADISON PIKE
HUNTSVILLE
AL
35806
US
|
Family ID: |
27573964 |
Appl. No.: |
10/186306 |
Filed: |
June 28, 2002 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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10186306 |
Jun 28, 2002 |
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09419806 |
Oct 18, 1999 |
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09419806 |
Oct 18, 1999 |
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08335676 |
Nov 8, 1994 |
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08335676 |
Nov 8, 1994 |
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07846597 |
Mar 5, 1992 |
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5363108 |
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07846597 |
Mar 5, 1992 |
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07368831 |
Jun 20, 1989 |
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07368831 |
Jun 20, 1989 |
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07192475 |
May 10, 1988 |
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07192475 |
May 10, 1988 |
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06870177 |
Jun 3, 1986 |
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4743906 |
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06870177 |
Jun 3, 1986 |
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06677597 |
Dec 3, 1984 |
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4641317 |
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10186306 |
Jun 28, 2002 |
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PCT/US90/01174 |
Mar 2, 1990 |
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10186306 |
Jun 28, 2002 |
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PCT/US89/01020 |
Mar 10, 1989 |
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PCT/US89/01020 |
Mar 10, 1989 |
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07010440 |
Feb 3, 1987 |
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4813057 |
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Current U.S.
Class: |
342/27 ; 342/1;
342/21; 375/E1.001 |
Current CPC
Class: |
H01Q 21/061 20130101;
H04B 1/7183 20130101; G01S 13/0209 20130101; G01S 13/18 20130101;
H04B 1/71637 20130101; H04L 27/103 20130101; H01Q 9/28 20130101;
H01Q 9/40 20130101; H04B 1/7174 20130101; G01S 7/292 20130101; G01S
7/282 20130101; H04B 14/026 20130101; H04B 1/69 20130101 |
Class at
Publication: |
342/27 ; 342/21;
342/1 |
International
Class: |
G01S 013/04; G01S
007/282 |
Claims
Having thus described our invention, we claim:
1. A wideband electromagnetic system comprising; generating means
for generating stepped amplitude signals; transmitting means
responsive to said stepped amplitude signals and including a
wideband antenna for transmitting wideband burst signals, said
antenna comprising at least one dipole and wherein each dipole is
substantially of the same length and includes two generally
triangular elements characterized by having a like broad base and
narrow apex; and receiving means for detecting signals derived from
transmitted burst signals.
2. A system as set forth in claim 1 wherein said receiving means
comprises coherent detection means responsive to times of
initiation of said burst signals.
3. A system as set forth in claim 1 wherein said coupling is to the
bases of said antenna elements.
4. A light modulation system comprising; a source of narrow band
light; an elongated optical channel having an entrance end for
receiving light from said source and an exiting end and having a
refractive index variable by an electrical field; conductive means
extending along said optical channel for applying an electrical
field across said optical channel; and signal means for generating
a modulation signal and applying said signal to said conductive
means in a region adjacent to said exiting end of said optical
channel and thereby creating said electrical field therein; whereby
light from said light source passing from said entrance end to said
exit end of said optical channel is affected by said electrical
field, whereby the frequency of light passing through said optical
channel is varied in frequency as a function of said signal.
5. A system as set forth in claim 4 wherein said conductive means
comprises first and second elongated conductors positioned on
opposite sides of said optical channel.
Description
CROSS-REFERENCE OF RELATED APPLICATIONS
[0001] This application is a continuation-in-part of application
Ser. No. 08/335,676, filed on Nov. 8, 1994, which is a
continuation-in-part of application Ser. No. 07/846,597, filed on
Mar. 5, 1992, which is a continuation of application Ser. No.
07/368,831, filed on Jun. 20, 1989; which is a continuation-in-part
of application Ser. No. 07/192,475, filed on May 10, 1988; which is
a continuation-in-part of application Ser. No. 06/870,177, filed on
Jun. 3, 1986, now U.S. Pat. No. 4,743,906; which is a
continuation-in-part of application Ser. No. 06/677,597, filed on
Dec. 3, 1984, now U.S. Pat. No. 4,641,317.
[0002] This application is also a continuation-in-part of
International Application No. PCT/US90/01174, filed on Mar. 2,
1990, which is a continuation-in-part of International Application
No. PCT/US89/01020, filed on Mar. 10, 1989. Said PCT Application
No. PCT/US89/01020 is also a continuation-in-part of U.S.
application Ser. No. 07/010,440, filed on Feb. 3, 1987, now U.S.
Pat. No. 4,813,057.
[0003] This above-named prior patent applications and patents are
hereby incorporated by referenced.
FIELD OF THE INVENTION
[0004] This invention relates generally to radio systems wherein
time-spaced, essentially monocycle-like signals are created from DC
pulses and transmitted into space wherein the resulting energy
bursts are dispersed in terms of frequency to where the spectral
density essentially merges with ambient noise, and yet information
relating to these bursts is recoverable.
BACKGROUND OF THE INVENTION
[0005] Radio transmissions have heretofore been largely approached
from the point of view of frequency channelling. Thus, coexistent
orderly radio transmissions are permissible by means of assignment
of different frequencies or frequency channels to different users,
particularly as within the same geographic area. Essentially
foreign to this concept is that of tolerating transmissions which
are not frequency limited. While it would seem that the very notion
of not limiting frequency response would create havoc with existing
frequency denominated services, it has been previously suggested
that such is not necessarily true, and that, at least
theoretically, it is possible to have overlapping use of the radio
spectrum. One suggested mode is that provided wherein very short
(on the order of one nanosecond or less) radio pulses are applied
to a broadband antenna which ideally would respond by transmitting
short burst signals, typically comprising three or four polarity
lobes, which comprise, energywise, signal energy over essentially
the upper portion (above 100 megacycles) of the most frequently
used radio frequency spectrum, that is, up to the mid-gigahertz
region. A basic discussion of impulse effected radio transmission
is contained in article entitled "Time Domain Electromagnetics and
Its Application," Proceedings of the IEEE, Volume 66, No. 3, March
1978. This article particularly suggests the employment of such
technology for baseband radar, and ranges from 5 to 5,000 feet are
suggested. As noted, this article appeared in 1978, and now, 16
years later, it is submitted that little has been accomplished by
way of achieving commercial application of this technology.
[0006] From both a theoretical and an experimental examination of
the art, it has become clear to the applicant that the lack of
success has largely been due to several factors. One is that the
extremely wide band of frequencies to be transmitted poses very
substantial requirements on an antenna. Antennas are generally
designed for limited frequency bandwidths, and traditionally when
one made any substantial change in frequency, it became necessary
to choose a different antenna or an antenna of different
dimensions. This is not to say that broadband antennas do not, in
general, exist; however, applicant has reviewed many types
including bicone, horn, and log periodic types and has determined
that none provided a practical antenna which will enable impulse
radio and radar usage to spread beyond the laboratory. Of the
problems experienced with prior art antennas, it is to be noted
that log periodic antennas generally produce an undesired frequency
dispersion. Further, in some instances, elements of a dipole type
antenna may be configured wherein there is a DC path between
elements, and such is not operable for employment in applicant's
transmitter.
[0007] A second problem which has plagued advocates of the
employment of impulse or time domain technology for radio is that
of effectively receiving and detecting the presence of the wide
spectrum that a monocycle burst produces, particularly in the
presence of high levels of existing ambient radiation, presently
nearly everywhere. Ideally, a necessary antenna would essentially
evenly reproduce the spectrum transmitted, and the receiver it
feeds would have special properties which enable it to be utilized
despite the typically high noise level with which it must compete.
The state of the art prior to applicant's entrance generally
involved the employment of brute force detection, i.e., that of
threshold or time threshold gate detection. Threshold detection
simply enables passage of signals higher than a selected threshold
level. The problem with this approach is obvious that if one
transmits impulse generated signals which are of sufficient
amplitude to rise above ambient signal levels, the existing radio
services producing the latter may be unacceptably interfered with.
For some reason, perhaps because of bias produced by the wide
spectrum of signal involved, e.g., from 50 mHz to on the order of 5
gHz or ever higher, the possibility of coherent detection has been
thought impossible.
[0008] Accordingly, it is an object of this invention to provide an
impulse or time domain (or baseband) transmission system which
attacks all of the above problems and to provide a complete impulse
time domain transmission system which, in applicant's view,
eliminates the known practical barriers to its employment, and,
importantly, its employment for all important electromagnetic modes
of radio, including communications, telemetry, navigation and
radar.
SUMMARY OF THE INVENTION
[0009] With respect to the antenna problem, applicant has
determined a truly pulse-responsive antenna which translated an
applied DC impulse into essentially a monocycle. It is a dipole
which is completely the reverse of the conventional bat wing
antenna and wherein two triangular elements of the dipole are
positioned with their bases closely adjacent but DC isolated. They
are driven at near adjacent points on the bases bisected by a line
between apexes of the two triangular elements. This bisecting line
may mark a side or height dimension of the two triangular elements.
Alternately, a monopole configuration is employed.
[0010] As a further consideration, power restraints in the past
have been generally limited to the application of a few hundred
volts of applied signal energy to the transmitting antenna. Where
this is a problem, it may be overcome by a transmitter switch which
is formed by a normally insulating crystalline structure, such as
diamond material sandwiched between two metallic electrodes, which
are then closely coupled to the elements of the antenna. This
material is switched to a conductive, or less resistive, state by
exciting it with an appropriate wavelength beam of light,
ultraviolet in the case of diamond. In this manner, no metallic
triggering communications line extends to the antenna which might
otherwise pick up radiation and re-radiate it, adversely affecting
signal coupling to the antenna and interfering with the signal
radiated from it, both of which tend to prolong the length of a
signal burst, a clearly adverse effect.
[0011] With respect to a radio receiver, a like receiving antenna
is typically employed to that used for transmission as described
above, although a single antenna and transmit-receive switch may Be
substituted. Second, a locally generated, coordinately timed
signal, to that of the transmitted signal, is either detected from
the received signal, as in communications or telemetry, or received
directly from the transmitter, as, for example, in the case of
radar. Then, the coordinately timed signal, typically including a
basic half cycle, or a few, up to 10 half cycles, of signal, is
mixed or multiplied by a factor of 1 (as with sampling or gating of
the received signals), or ideally, as where the coordinately
locally generated signal is curved, the factor is greater than one,
giving rise to amplification in the process of detection, a
significant advantage. Thus, the modulation on a signal, or
position of a target at a selected range, as the case may be, is
determined. Such a detection is further effected by an integration
of the detected signal, with enhanced detection being accomplished
by both a short term (first) and long term (second) integration. In
this latter process, individual pulse signals are, first,
integrated only during their existence to accomplish short term
integration, and following this, the resultant short term
integration signals are long term integrated by integrating a
selected number of these and particularly by a method which omits
the noise signal content which occurs between individual pulse
signals, thereby effecting a very significant increase in
signal-to-noise ratio.
[0012] It is acknowledged that coherent detection of analog signals
has been effected by the employment of coincidence detection,
followed by only long term detection, but it is submitted that such
coherent detection did not contemplate the local generation of a
signal but contemplated storing of a portion of a transmitted
signal which was then phase coordinated with the incoming signal,
which on its face presents an essentially impossible task where
there is the detection of a ultra wideband frequency pulse as in
the present case.
[0013] Further, transmitted burst signals may be varied in time
pattern (in addition to a modulation pattern for communications or
telemetry). This greatly increases the security of the system and
differentiates signals from nearly, if not all, ambient signals,
that is, ambient signals which are not synchronous with transmitted
burst signals. This also enables the employment of faster
repetition rates with radar which would, absent such varying or
dithering, create range ambiguities as between returns from
successive transmission and therefore ranges. Burst signals are
signals generated when a stepped, or near stepped, voltage change
is applied to an impulse-responsive antenna as illustrated and
discussed herein.
[0014] As still a further feature of this invention, the repetition
rate of burst signals may be quite large, say, for example, up to
100 mHz, or higher, this enabling a very wide frequency dispersion;
and thus for a given overall power level, the energy at any one
frequency would be extremely small, thus effectively eliminating
the problem of interference with existing radio frequency based
services.
[0015] As still a further feature of this invention, moving targets
are detected in terms of their velocity by means of the employment
of a bandpass filter, following mixing and double integration of
signals.
[0016] As a still further feature of the invention, when employed
in this latter mode, two channels of reception are ideally employed
wherein the incoming signal is multiplied by a selected range, or
timed, locally generated signal in one channel, and mixing the same
incoming signal by a slightly delayed, locally generated signal in
another channel, delay being on the order of one-fourth to one-half
the time of a monocycle. This accomplishes target differentiation
without employing a separate series of transmissions.
[0017] As still another feature of this invention, multiple
radiators or receptors would be employed in an array wherein their
combined effect would be in terms of like or varied-in-time of
sensed (or transmitted) output, to thereby accent either a path
normal to the face of the antenna or to effect a steered path
offset to a normal path accomplished by selected signal delay
paths.
[0018] As still another feature of this invention, radio antenna
elements would be positioned in front of a reflector wherein the
distant between the elements and reflector is in terms of the time
of transmission from an element or elements to reflector and back
to element(s), typically up to about three inches, this being with
tip-to-tip dimension of elements of somewhat below nine inches up
to approximately nine inches.
BRIEF DESCRIPTION OF THE DRAWINGS
[0019] FIG. 1 is a combination block-schematic diagram of an
intelligence time domain transmission system.
[0020] Fig. 1a is a schematic diagram of an alternate form of the
output stage of the transmitter shown in FIG. 1.
[0021] FIG. 2 is a block diagram of a time domain receiver as
contemplated by this invention.
[0022] FIG. 2a is a block diagram of a single antenna system for
transmitting and receiving.
[0023] FIG. 3 is a set of electrical waveforms illustrative of
aspects of the circuitry shown in FIGS. 1 and 1b.
[0024] FIG. 4 is a set of electrical waveforms illustrating aspects
of operation of the circuitry shown in FIG. 2.
[0025] FIG. 5 is an electrical block diagram illustrative of a
basic radar system constructed in accordance with this
invention.
[0026] FIGS. 6a-6e and 7 illustrate the configuration of an antenna
in accordance with the invention.
[0027] FIGS. 8 and 9 diagrammatically show an antenna array.
[0028] FIG. 9a shows a side view of an alternate antenna array.
[0029] FIG. 9b shows a frontal view of the alternate antenna
array.
[0030] FIGS. 10-15 illustrate different switching assemblies as
employed in the charging and discharging of antennas to effect
signal transmission.
[0031] FIG. 16 illustrates a radar system particularly for
employment in facility surveillance, and FIG. 17 illustrates a
modification of this radar system.
[0032] FIGS. 18, 18A and 19 illustrate the general arrangement of
transmission and receiving antennas for three-dimensional location
of targets.
[0033] FIG. 20 is a schematic illustration of a modified portion of
FIG. 1 illustrating transmission and reception of time domain type
sonic signals.
[0034] FIG. 21 is a schematic illustration of an alternate portion
of FIG. 1 illustrating both the employment of like time domain
signals and a like modulation system adapted to produce broadband
modulated light signals from the output of a conventional narrow
band laser.
[0035] FIG. 22 is an illustration of an optical frequency
modulator.
[0036] FIG. 23 is an illustration of an optical frequency
demodulator.
DESCRIPTION OF THE PREFERRED EMBODIMENT
[0037] Referring to FIG. 1, and initially to transmitter 10, a base
frequency of 100 kHz is generated by oscillator 12, typically being
a crystal controlled oscillator. Its output, a pulse signal, is
applied to -4 divider 14 to provide at its output a 25-kHz (0 to 5
volts) pulse signal shown in waveform A of FIG. 3. Further
alphabetic references to waveforms will simply identify them by
their letter identity and will not further refer to the figure,
which will be FIG. 3. The 25-Khz output is employed as a general
transmission-signal.
[0038] The output of -4 divider 14 is employed as a signal base and
as such is supplied through capacitor 20 to pulse position
modulator 22. Pulse position modulator 22 includes in its input an
RC circuit consisting of resistor 24 and capacitor 26 which convert
the square wave input to an approximately triangular wave as shown
in waveform B, it being applied across resistor 25 to the
non-inverting input of comparator 28. A selected or reference
positive voltage, filtered by capacitor 27, is also applied to the
non-inverting input of comparator 28, it being supplied from
+5-volt terminal 31 of DC bias supply 30 through resistor 32.
Accordingly, for example, there would actually appear at the
non-inverting input a triangular wave biased upward positively as
illustrated by waveform C.
[0039] The actual conduction level of comparator 28 is determined
by an input signal supplied through capacitor 36, across resistor
37, to the inverting input of comparator 28, as biased from supply
30 through resistor 38 and across resistor 32. The combined signal
input bias is illustrated in waveform D.
[0040] Four alternate intelligence inputs are provided for
comparator 28. With switch 39 open, switch 39a open, 39b switched
to an alternate position from that shown, and switch 39c open,
there is simply an amplified output of microphone 34 applied to the
inverting input of comparator 28.
[0041] A second type of operation is achieved by simply closing
switch 39, with the result being that the signal input to
comparator 28 would be the sum, appearing across resistor 41, of
the microphone signal and the signal output of signal generator 33.
For example, signal generator-33 would provide a known sequence of
analog or binary signals. This combination would result in an
encoded or dithered signal. As in the first instance, the combined
signal would be provided to comparator 28. Third, switch 39 would
be open, switch 39a open, switch 39b in the indicated position, and
switch 39c closed. In this posture, the amplified microphone signal
would be provided to A-D converter 34a which would digitize the
microphone signal. The digitized microphone signal is then fed to
parallel-to-serial converter 34b, and then the resulting digitized
serial version of the signal is fed through switch 39c to
comparator 28.
[0042] Finally, the circuit configuration may be changed with
switch 39 open, switch 39a closed, switch 39b in the indicated
position, and switch 39c open. In this configuration, digital data
from digital source 29 is fed to parallel-to-serial converter 29a,
which converts the data to serial form and provides it as an input
to comparator 28. In all cases, the signal to be transmitted is fed
through capacitor 36 and across resistor 37 to the inverting input
of comparator 28. The output of generator 33 may also be used to
impose a dither on the inputs to comparator 28 wherein the signal
from microphone 24 is digitized when the intelligence signal
emanates from digital source 29.
[0043] In operation, with one of the signals described above
present at the inverting input of comparator 28, and by virtue of
the combination thus described, the output of comparator 28 would
rise to a positive saturation level when a triangular signal 40
(waveform C) is of a higher value than the effective modulation
signal 42 and drop to a negative saturation level when modulation
signal 42 is of a greater value than the triangular wave signal 40.
The output signal of comparator 28 is shown in waveform F, and the
effect is to vary the turn-on and turn-off of the pulses shown in
this waveform as a function of the input signal. Thus, there is
effected a pulse position modulation from any one of the alternate
input amplitude signals. Where a dither signal is employed, it
enables an added discrete pattern of time positions to be included
with a transmitted signal, thus requiring that to receive and
demodulate it, the dither signal must be accurately reproduced.
This provides an element of security.
[0044] With respect to the output signal of comparator 28, we are
interested in employing a negative going or trailing edge 44 of it,
and it is to be noted that this trailing edge will vary in its time
position as a function of the signal modulation. This trailing edge
of the waveform, in waveform F, triggers "on" mono, or monostable
multivibrator, 46 having an "on" time of approximately 50
nanoseconds, and its output is shown in waveform G. For purposes of
illustration, while the pertinent leading or trailing edges of
related waveforms are properly aligned, pulse widths and spacings
(as indicated by break lines, spacings are 40 microseconds) are not
related in scale. Thus, the leading edge of pulse waveform G
corresponds in time to the trailing edge 44 (waveform F), and its
time position within an average time between pulses of waveform G
is varied as a function of the input modulation signal to
comparator 28.
[0045] The output of mono 46 is applied through diode 48 across
resistor 50 to the base input of NPN transistor 52 operated as a
triggering amplifier. It is conventionally biased through resistor
54, e.g, 1.5K ohms, from +5-volt terminal 31 of 5-volt power supply
30 to its collector. Capacitor 56, having an approximate
capacitance of 0.01 mf, is connected between the collector and
ground of transistor 52 to enable full bias potential to appear
across the transistor for its brief turn-on interval, 50
nanoseconds. The output of transistor 52 is coupled between its
emitter and ground to the primary 58 of trigger transformer 60.
Additionally, transistor 52 may drive transformer 60 via an
avalanche transistor connected in a common emitter configuration
via a collector load resistor. In order to drive transformer 60
with a steep wave front, an avalanche mode operated transistor is
ideal. Identical secondary windings 62 and 64 of trigger
transformer 60 separately supply base-emitter inputs of NPN
avalanche, or avalanche mode operated, transistors 66 and 68 of
power output stage 18. Although two are shown, one or more than two
may be employed when appropriately coupled.
[0046] With avalanche mode operated transistors 66 and 68, it has
been found that such mode is possible from a number of types of
transistors not otherwise labeled as providing it, such as a
2N2222, particularly those with a metal can. The avalanche mode
referred to is sometimes referred to as a second breakdown mode,
and when transistors are operated in this mode and are triggered
"on" their resistance rapidly goes quite low (internally at near
the speed of light), and they will stay at this state until
collector current drops sufficiently to cut off conduction (at a
few microamperes). Certain other transistors, such as a type
2N4401, also display reliable avalanche characteristics.
[0047] As illustrated, impulse antenna 200 having antenna elements
204 and 206 is charged by a DC source 65 through resistors 67 and
69 to an overall voltage which is the sum of the avalanche voltage
of transistors 66 and 68 as discussed above. Resistors 67 and 69
together have a resistance value which will enable transistors 66
and 68 to be biased as described above. Resistors 71 and 73 are of
relatively low value and are adjusted to receive energy below the
frequency of cut-off of the antenna. In operation, when a pulse is
applied to the primary 58 of pulse transformer 60, transistors 66
and 68 are turned "on," effectively shorting, through resistors 71
and 73, antenna elements 204 and 206. This action occurs extremely
fast, with the result that a signal is generated generally as shown
in pulse waveform G (but somewhat rounded). Antenna 200
differentiates the pulse G to transmit essentially a monocycle of
the general shape shown in waveform H. The illustrated
configuration of antenna 200, and a feature of this invention, is
further described below.
[0048] FIG. 1a illustrates an alternate embodiment of a transmitter
output stage. It varies significantly from the one shown in FIG. 1
in that it employs a light-responsive avalanche transistor 63,
e.g., a 2N3033. Similar components are designated with like
numerical designations to that shown in FIG. 1, but with the suffix
"a" added. Transistor 63 is triggered by laser diode or fast
turn-on LED (light emitting diode) 61, in turn driven by NPN
avalanche transistor 52a generally operated as shown in FIG. 1. By
employment of a light-activated avalanche or other avalanche mode
operated semiconductor switches (now existing or soon appearing),
or a series of them connected in series, it appears that the
voltage for power source 65 or 65a may be elevated into the
multi-kilovolt range, thus enabling a power output essentially as
high as desired. In this respect, and as a particular feature of
this invention, a light-triggered, gallium arsenide, avalanche mode
operated switch would be employed.
[0049] Referring back to FIG. 1, the output of monocycle producing
antenna 200, with elements 204 and 206, is typically transmitted
over a discrete space and would typically be received by a like
broadband antenna, e.g., antenna 200 of a receiver at a second
location (FIG. 2).
[0050] FIG. 2 illustrates a radio receiver which is particularly
adapted to receive and detect a time domain transmitted signal. In
addition, it particularly illustrates a system for detecting
intelligence which has been mixed with a particular offset or
dither signal, analog or digital, such as provided by binary
sequence "A" producing generator 33 shown in FIG. 1. It will thus
be presumed for purposes of description that switch 39 of FIG. 1 is
closed and that the signal transmitted by transmitter 10 is one
wherein intelligence signals from microphone 34 are combined with
the output of binary sequence "A" of generator 33, and thus that
the pulse position output of transmitter 10 is one wherein pulse
position is a function of both intelligence and offset or dither
signals. Thus, the transmitted signal may be described as a pulse
position modulated signal subjected to changes in pulse position as
effected by a time offset pattern of the binary sequence "A."
[0051] The transmitted signal from transmitter 10 is received by
antenna 200 (FIG. 2), and this signal is fed to two basic circuits,
demodulation circuit 222 and template generator 224. In accordance
with this system, a replica of the transmitted signal, waveform H
FIG. 3, is employed to effect detection of the received signal,
basic detection being accomplished in multiplier or multiplying
mixer 226. For maximum response, the template signal, reproduced as
waveform T1 in FIG. 4, must be applied to mixer 226 closely in
phase with the input, as will be further described. It will differ
by a magnitude not perceptible in the waveforms of FIG. 4 as a
function of modulation, effecting swings of approximately 200
picoseconds, typically for a 1-nanosecond pulse. To accomplish such
near synchronization, template generator 234 employs a crystal
controlled but voltage controlled oscillator 227 which is operated
by a control voltage which synchronizes its operation in terms of
the received signal.
[0052] Oscillator 227 operates at a frequency which is
substantially higher than the repetition rate of transmitter 10,
and its output is divided down to the operating frequency of 25 Khz
by frequency divider 230, thus equal to the output of divider 14 of
transmitter 10.
[0053] In order to introduce a pattern of dither corresponding to
that provided by binary sequence "A" generator 33, a like generator
228 provides a binary changing voltage to programmable delay
circuit 232 which applies to the signal output of divider 230 a
delay pattern corresponding to the one effected by binary sequence
"A" generator 33 of FIG. 1 when added to intelligence modulation.
Thus, for example, this might be four 8-bit binary words standing
for the numerals 4, 2, 6, and 8, the same pattern having been
generated by binary sequence "A" generator 33 and transmitted by
transmitter 10. It is further assumed that this is a repeating
binary pattern. Thus, programmable delay 232 will first delay a
pulse it receives from divider 230 by four units. Next, the same
thing would be done for the numeral 2, and so on, until the
four-numeral sequence has been completed. Then, the sequence would
start over. In order for the two binary sequence generators to be
operated in synchronization, either the start-up time of the
sequence must be communicated to the receiver, or else signal
sampling would be for a sufficient number of signal input pulses to
establish synchronization by operation of the synchronization
system, as will be described. While a repeatable sequence is
suggested, it need not be such so long as there is synchronization
between the two generators, as by transmission of a sequence start
signal and the provision in the receiver of means for detecting and
employing it.
[0054] Either programmable delay 232 or a second delay device
connected to its output would additionally provide a general
circuit delay to take care of circuit delays which are inherent in
the related circuitry with which it is operated, as will be
described. In any event, the delayed output of delay 232, which is
a composite of these, will be provided to the input of template
generator 234, and it is adapted to generate a replica of the
transmitted signal, illustrated in FIG. 4 as waveform T1.
Differential amplifier 246 basically functions to provide a DC
voltage as needed to apply a correction or error signal to
oscillator 227 as will enable there to be provided to mixer 226
replica signal Ti exactly in phase with the average time of input
signal Ea.
[0055] In order to generate the nearest signal, the input signal Ea
is multiplied by two spaced, in time, replicas of the template
signal output of template generator 234. The first of these,
indicated as T1, is multiplied in mixer 236 by input signal Ea in
mixer 238. As will be noted in FIG. 4, T2 is delayed from signal T1
by delay 240 by a period of essentially one-half of the duration of
the major lobe P of template signal T1.
[0056] The output of mixer 236 is integrated in integrator 242, and
its output is sampled and held-by sample and hold unit 244 as
triggered by delay 232. The output of sample and hold unit 244, the
integral of the product of the input signal Ea and T1, is applied
to the non-inverting input of differential amplifier 246 similarly,
the output of mixer 238 is integrated by integrator 249 and sampled
and held by sample and hold 250 as triggered by delay 232, and the
integrated product of the input signal Ea and template signal T2 is
applied to the inverting input of differential amplifier 246.
[0057] To examine the operation of differential amplifier 246, it
will be noted that if the phase of the output of oscillator 227
should advance, signals T1 and Ea applied to mixer 236 would become
closer in phase, and their product would increase, resulting in an
increase in input signal to the non-inverting input of differential
amplifier 246, whereas the advance effect on template signal T2
relative to the input signal Ea would be such that their
coincidence would decrease, causing a decrease in the product
output of mixer 238 and therefore a decreased voltage input to the
inverting input of differential amplifier 246. As a result, the
output of differential amplifier 246 would be driven in a positive
direction, and this polarity signal would be such as to cause
oscillator 227 to retard. If the change were in the opposite
direction, the result would be such that higher voltages would be
applied to the inverting input than to the non-inverting input of
differential amplifier 246, causing the output signal to decrease
and to drive oscillator 227 in an opposite direction. In this
manner, the near average phase lock is effected between the input
signal Ea and template signal Ta which is directly employed in the
modulation of the input signal. The term "near" is used in that the
output of differential amplifier 246 is passed through low pass
filter 253 before being applied to the control input of oscillator
227. The cut-off frequency of low pass filter 253 is set such that
it will take a fairly large number of pulses to effect phase shift
(e.g., 10 Hz to perhaps down to 0.001 Hz). As a result, the
response of oscillator 227 is such that it provides an output which
causes waveform T1 and thus waveform Ta to be non-variable in
position with respect to modulation effect. With this limitation in
mind, and in order to obtain a synchronous detection of the input
signal, the output T1 of template generator 234 is delayed by a
period equal to essentially one-fourth the period P of the major
lobe of the template and input signal, and this is applied as
signal Ta with the input signal Ea to multiplying mixer 226. As
will be noted, the resulting delayed signal, Ta, is now near
synchronization with the input signal Ea, and thus the output of
multiplier 226 provides essentially a maximum signal output. When
there is simply no transmitted signal, or a noise signal, at the
signal input of mixer 226, there would be between input signals Ea
an elapsed time of exactly 40 milliseconds shown in FIG. 4, and a
quite minimum deviation in output would appear from mixer 226.
[0058] The signal output of mixer 226 is integrated in integrator
250, and the output signal is multiplied by a factor of 0.5 by
amplifier 252. Then this one-half voltage output of amplifier 252
is applied to the inverting input of comparator 254, and this
voltage represents one-half of the peak output of integrator 250.
At the same time, a second output of integrator 250 is fed through
delay 256 to the non-inverting input of comparator 254, delay being
such as required for stabilization of the operation of amplifier
252 and comparator 254 in order to obtain an effective comparison
signal level that will be essentially free of the variable
operation of these two units. The output of comparator 254
represents an essentially precise time market which varies with the
position of input signal Ea. It is then fed to the reset input of
flip-flop 258, a set input being provided from the output of delay
232 which represents, because of low pass filter 253, an averaged
spacing between input signals, thus providing a reference against
which the modulation controlled time variable output signal of
comparator 254 may be related. It is related by virtue of the
output of delay 232 being provided as the set input of flip-flop
258. Thus, for example, the output of flip-flop 258 would rise at a
consistent time related to the average repetition rate as
essentially dictated by low pass filter 253. Thus, the output of
flip-flop 258 would be brought back to zero at a time which
reflected the intelligence modulation on the input signal. Thus, we
would have a pulse height of a constant amplitude, but with a pulse
width which varied directly with modulation. The output of
flip-flop 258 is then fed through low pass filter 260, which
translates the signal from pulse width demodulation to amplitude
signal modulation, which is then reproduced by loudspeaker 262 with
switch A in the upper position.
[0059] Where the intelligence transmission is in digital form,
switch A is moved to the lower position wherein the output of LP
filter 260 is fed to the non-inverting input of comparator 261a, a
potential being applied to the inverting input sufficient to block
the transition of comparator 261a from an off state to an on state
absent a significant "1" binary signal. Assuming that the digital
signal is a converted analog signal and the signal is
representative of an analog voice input as shown in FIG. 1, switch
B will be positioned in the indicated position wherein the output
of comparator 261a is fed to D-A converter 261b, and the thus
derived analog signal is fed via switch C in the lower position to
loudspeaker 262.
[0060] In the event that the digital transmission is derived from
another digital source, such as illustrated by digital source 29 in
FIG. 1, which might be a computer, switch B is switched from its
shown position to its lower position, wherein the output of
comparator 261a is fed via serial-to-parallel converter 261d to
digital register 261c, such as another digital computer or a
digital computer terminated by a monitor. Thus, in this
configuration, purely transmitted digital signals would be
processed in purely digital form. In this case, switch C would be
moved to its upper position as no signal is being transmitted to
it.
[0061] While the generation and detection of digital signals have
been described in terms of binary encoding, it is to be appreciated
that multi-level encoding might be employed and detected wherein
discretely positioned bits would be represented by different
effected delays and encoded in this manner.
[0062] Assuming that binary sequence generator 33 of transmitter 10
and binary sequence "A" generator 228 for the receiver are operated
essentially in synchronization, the effect of the time position
dither effected by generator 33 of transmitter 10 will have no
dislocating effect on the signal.
[0063] As suggested above, in order to ensure synchronization, some
form of signaling between the transmitter and receiver as to the
starting of the binary sequence generator, generator 33, is
required. This may be done by an auxiliary transmitter or by a
decoding arrangement wherein there would be provided at the
conclusion of, say, one sequence of binary sequence generator 33, a
start signal for binary sequence generator 228 of the receiver.
Absent this, in the free running mode, there would be effected
synchronization by the operation of template generator 234 which,
for short codes, and with relatively low noise levels, would be
relatively short; and for longer codes, or instances where noise
was a significant problem, longer codes would be required for
synchronization. Where needed, a receiving station might transmit
back to the original transmitting station an acknowledgment that
synchronization has been achieved.
[0064] From the foregoing, it should be appreciated that applicant
has provided both an inexpensive and practical time domain system
for communications. While a system has been described wherein a
single short pulse, for example, a nanosecond, is transmitted at a
repetition rate such that 40 microseconds is between pulses, the
invention contemplates that a group of pulses might be sent which
would be separated by the longer period. Thus, for example, an
8-bit set might be transmitted as a group wherein there was simply
room between the pulses to detect their multi-position shifts with
modulation. By this arrangement, it is to be appreciated that
intelligence information transmitted would be increased by up to
256 times, or the immunity from noise could be substantially
improved by this technique and related ones.
[0065] FIG. 2a illustrates the employment of a single antenna 200
for both transmitting and receiving. Thus transmitter 18 (FIG. 1)
provides an output to antennas 200 through transmit/receive switch
TR, being arranged such that bias supply B is normally connected as
illustrated in FIG. 1 to the antenna elements and a switch of the
transmitter discharges bias on the antenna element to effect
transmission of the signal. Switch TR supplies a signal received by
antenna 200 to receiver 222 on a time sharing basis. In one version
of the present invention, the transmit repetition rate is raised
from that earlier described to 10 megahertz. In such case, as an
example, switch TR would be controlled, by means not shown, to
enable transmission from transmitter 18 for 12 microseconds. Then,
after a few microseconds, depending on range of transmission,
antenna elements 200 would be connected in the RECEIVE mode for 12
microseconds.
[0066] FIG. 5 particularly illustrates a radar system of the
present invention for determining range. Impulse-responsive, or
impulse, antenna 200, or antenna 201 as shown in FIG. 6a, of
transmitter 239 FIG. 5 comprises triangular elements A and B with
closely spaced bases. A dimension of a base and a dimension normal
to the base of each element is approximately 4 inches and is
further discussed and illustrated with respect to FIGS. 6 and 7.
Typically, a reflector would be used as illustrated in FIG. 8.
Alternately, as shown in FIG. 6a, a base is reduced to 2 inches
wherein the elements are halved as shown in FIG. 6a.
[0067] Significantly, however, the length of path from a feed point
to an edge is the same in both cases.
[0068] The transmitter is basically controlled by control 310. It
includes a transmit sequence, or rate, control portion 312 which
determines the timing of transmitted signal bursts, at, for
example, 10,000 bursts per second, in which case transmit sequence
control 312 generates an output at 10,000 Hz on lead 314.
Oscillator 316 is operated at a higher rate, for example, 20
Mhz.
[0069] The signal output of transmit sequence control 312 is
employed to select particular pulse outputs of oscillator 316 to be
the actual pulse which is used as a master pulse for controlling
both the output of transmitter 239 and the timing of receiver
functions, as will be further described. In order to unambiguously
and repetitively select an operative pulse with low timing
uncertainty from oscillator 316, the selection is one and some
fraction of an oscillator pulse interval after an initial signal
from sequence control 312. The selection is made via a control
sequence employing D-type flip-flops 318, 320, and 322. Thus, the
transmit sequence control pulse on lead 314 is applied to the clock
input of flip-flop 318. This causes the Q output of flip-flop 318
to transition to a high state, and this is applied to a D input of
flip-flop 320. Subsequently, the output of oscillator 316 imposes a
rising edge on the clock input of flip-flop 320. At that time, the
high level of the D input of this flip-flop is transferred to the Q
output. Similarly, the Q output of flip-flop 320 is provided to the
D input of flip-flop 322, and the next rising edge of the pulse
from oscillator 316 will cause the not Q output of flip-flop 322 to
go low and thus initiate the beginning of the transmit-receive
cycle.
[0070] For the transmit mode, the not Q output of flip-flop 322 is
fed as an input to analog programmable delay 313 and to counter
315. Counter 315, for example, would respond to the not Q outputs
of flip-flop 322 and count up to a selected number, for example,
356, and recycle to count again. Its binary output would be fed as
an address to memory unit 317, ROM or RAM, which would have stored,
either in numerical address order, or randomly selected order, a
number. As a result, upon being addressed, a discrete output number
would be fed to D/A converter unit 321. D/A converter unit 321
would then provide an analog signal output proportional to the
input number. This -output is employed to sequentially operate
programmable delay unit 313 for delays of pulses from flip-flop 322
by an amount proportional to the signal from D/A converter 321. The
range of delays would typically be up to the nominal timing between
pulses, in this case, up to 300 nanoseconds, and practically up to
99 nanoseconds. The delayed output of programmable delay unit 313
is then fed to fixed delay unit 324, which provides a fixed delay
of 200 nanoseconds to each pulse that it receives. The thus delayed
pulses are then fed to trigger generator 323. Trigger generator
323, e.g., an avalanche mode operated transistor, would provide a
sharply rising electrical output at the 10,000 Hz rate or a like
response of light output, e.g., by laser, depending upon the
transmitter to be driven. In accordance with one feature of this
invention, trigger generator 323 would be an ultraviolet laser. In
any event, a pulse of trigger generator 323 is fed to and rapidly
turns "on" a switch, for example, diamond 335, which, for example,
may again be an electrically operated or light operated switch,
such as a diamond switch in response to the ultraviolet laser
triggering device via fiber optic 327. Importantly, it must be
capable of switching in a period of a nanosecond or less. t is then
switched "on" to discharge elements A and B of antenna 00, having
earlier been charged from power source B through resistors R1 and
R2, source B being, for example, 100 to 5,000 volts.
[0071] Conformal impulse antenna 200 or 201 (FIG. 6a) is turned
"on" or turned "off," or successively both, by switch assembly 319
which applies stepped voltage changes to the antenna. It responds
by transmitting essentially short burst signals each time that it
is triggered. These burst signals are then transmitted into space
via directional versions of antenna 200 as illustrated in FIGS. 8
and 9, or simply by an omni-directional antenna as shown by antenna
200 in FIG. 1 or 201 in FIG. 6a.
[0072] Signal returns from a target would be received by receiver
326, typically located near or together with transmitter 239, via
receiving antenna 200a, which would, for example, be like a
transmitting antenna 200. The received signals are amplified in
amplifier 328 and fed to mixer 330, together with a signal from
template generator 332, driven by delay line 336, which is timed to
produce signals, typically half cycles in configuration, and
corresponding in time to the anticipated time of arrival of a
signal from a target at a selected range.
[0073] Mixer 330 functions to multiply the two input signals, and
where there are coincidence signals, timewise and with like or
unlike polarity coincident signals, there is a significant and
integratable output, indicating a target at the range. A mixer and
the following circuitry may be reused for later arriving signals
representative of different range, this range or time spacing being
sufficient to complete processing time for reception and
integration at a range as will be described. Additional like
mixtures and following circuitry sets may be employed to fill in
the range slots between that capable for one set.
[0074] Since the goal here is to determine the presence or absence
of a target based on a number of signal samplings as effected by
integration, where a true target does not exist, the appearance of
signals received by mixer 330 corresponding to the time of receipt
of signals from template generator 332 will typically produce
signals which vary not only in amplitude, but also in polarity. It
is to be borne in mind that the present system determines
intelligence, not instantaneously, but after a period of time,
responsive to a preponderance of coherent signals over time, a
facet of time domain transmission. Next, it is significant that the
template generator produce a template signal burst which is no
longer than the effecting signal to be received and bear a
consistent like or opposite polarity relationship in time with it.
As suggested above, received signals which do not bear this
relation to the template signal will be substantially attenuated.
As one signal, the template signal is simply a one polarity burst
signal. Assuming that it maintains the time relationship described,
effective detection can be effected.
[0075] For purposes of illustration, we are concerned with looking
at a single time slot for anticipated signal returns following
signal bursts from transmitting antenna 200 or 201. Accordingly,
template generator 332 is driven as a function of the timing of the
transmitter. To accomplish this, coarse delay counter 335 and fine
delay programmable delay line 336 are employed. Down counter 335
counts down the number of pulse outputs from oscillator 316 which
occur subsequent to a control input of lead 338, the output of
programmable delay unit 313. A discrete number of pulses thereafter
received from oscillator 316 is programmable in down counter 335 by
an output X from load counter 341 on lead 340 of control 310, a
conventional device wherein a binary count is generated in control
310 which is loaded into down counter 335. As an example, we will
assume that it is desired to look at a return which occurs 175
nanoseconds after the transmission of a signal from antenna 200. To
accomplish this, we load into down counter 335 the number "7,"
which means it will count seven of the pulse outputs of oscillator
316, each being spaced at 50 nanoseconds. So there is achieved a
350-nanosecond delay in down counter 335, but subtracting 200
nanoseconds as injected by delay unit 324, we will have really an
output of down counter 335 occurring 150 nanoseconds after the
transmission of a burst by transmitting antenna 200 or 201. In
order to obtain the precise timing of 175 nanoseconds, an
additional delay is effected by programmable delay line 336, which
is triggered by the output of down counter 335 when its seven count
is concluded. It is programmed in a conventional manner by load
delay 342 of control 310 of lead Y and, thus in the example
described, would have programmed programmable delay line 336 to
delay an input pulse provided to it by 25 nanoseconds. In this
manner, programmable delay line 336 provides a pulse output to
template generator 332, 175 nanoseconds after it is transmitted by
transmitting antenna 200. Template generator 332 is thus timed to
provide, for example, a positive half cycle or square wave pulse to
mixer 330 or a discrete sequence or pattern of positive and
negative excursions.
[0076] The output of mixer 330 is fed to analog integrator 350.
Assuming that there is a discrete net polarity likeness or
unlikeness between the template signal and received signal during
the timed presence of the template signal, analog integrator 350,
which effectively integrates over the period of template signal,
will provide a discrete voltage output. If the signal received is
not biased with a target signal imposed on it, it will generally
comprise as much positive content as negative content on a time
basis; and thus when multiplied with the template signal, the
product will follow this characteristic, and likewise, at the
output of integrator 350, there will be as many discrete products
which are positive as negative. On the other hand, with target
signal content, there will be a bias in one direction or the other,
that is, there will be more signal outputs of analog integrator 350
that are of one polarity than another. The signal output of analog
integrator 350 is amplified in amplifier 352, and then,
synchronously with the multiplication process, discrete signals
emanating from analog integrator 350 are discretely sampled and
held by sample and hold 354. These samples are then fed to A/D
converter 356 which digitizes each sample, effecting this after a
fixed delay of 40 nanoseconds provided by delay unit 358, which
takes into account the processing time required by sample and hold
unit 354. The now discrete, digitally calibrated positive and
negative signal values are fed from A/D converter 356 to digital
integrator 362, which then digitally sums them to determine whether
or not there is a significant net voltage of one polarity or
another, indicating, if such is the case, that a target is present
at a selected range. Typically, a number of transmissions would be
effected in sequence, for example, 10, 100, or even 1,000
transmissions, wherein the same signal transmit time of reception
would be observed, and any signals occurring during like
transmissions would then be integrated in digital integrator 362,
and in this way enable recovery of signals from ambient,
non-synchronized signals which, because of random polarities, do
not effectively integrate.
[0077] The output of digital integrator 362 would be displayed on
display 364, synchronized in time by an appropriate signal from
delay line 336 (and delay 358) which would thus enable the time or
distance position of a signal return to be displayed in terms of
distance from the radar unit.
[0078] FIGS. 6 and 7 illustrate side and front views of an antenna
200. As is to be noted, antenna elements A and B are triangular
with closely adjacent bases, and switch 335 connects close to the
bases of the elements as shown. As an example, and as described
above, it has been found that good quality burst signals can be
radiated from impulses having a stepped voltage change occurring in
one nanosecond or less wherein the base of each element is
approximately 4 inches, and the height of each element is
approximately the same. Alternately, the antenna may be, as in all
cases, like that shown in FIG. 6a where antenna 201 is sliced in
half to have a base dimension of 2 inches. Either of the antennas
illustrated in FIG. 6, 8, or 6a may be employed as antennas in any
of the figures.
[0079] To further illustrate the antennas of this invention,
reference is made to FIGS. 6c-6f, showing monopole antennas.
[0080] FIGS. 6cand cl illustrates a monopole consisting of antenna
elements 7b and ground plane g. As will be noted, it is fed by
coaxial cable wherein the outer conductive cover C is connected to
ground plane g and the center conductor L to the center of antenna
element 7b. The distance between ground plane g and base region of
element 7b is exxagerated and in fact, in the center element 7b is
about 1 millimeter from ground plane g. It is to be noted that the
base of element 7b slopes up on each side at an angle of about 15
degrees. By virtue of this slope, the impedance at the feed point
is about 50 ohms, a desirable value. The monopole version lends
itself to a more compact arrangement. FIG. 6c illustrates a
modification of the antenna assembly shown in FIG. 6b where one
side of the antenna, being antenna 7c, omits one-half of the
antenna element of FIG. 6b. It is fed as described with with
respect to FIG. 6b.
[0081] As a second feature it employs a second ground plane, g2.
The second ground plane is approximately one inch below the second
ground plane g1. It has been found that by the addition of the
second ground plane member g2 that the frequency response of the
antenna assembly, with a one and one-half inch height of element 7c
and accordingly having a mid frequency of approximately 2
gigahertz, which is based on this dimension representing a one-half
wavelength, that a noticable notch decrease in response at about
900 megahertz occurs. This coincides with a substantial amount of
spectrum usage by other services and thus tends to reduce
interference.
[0082] FIGS. 6e and 6f illustrate the folding of the antenna shown
in FIG. 7b. This, of course, reduces the space required for antenna
element 6c. It is to be noted that the dimension of the antennas as
illustrated in FIGS., 6c-6f are of reduced size with respect to
certain antennas earlier discussed with the center frequency of
operation moved upward from from 600-700 megahertz to about 2
gigahertz. FIG. b2 illustrates an antenna control system for
employing a single antenna for both transmitting and receiving,
this being for a radar configuration. Thus, transmitter 239 (FIG.
5) provides a transmit pulse through transmit/receive switch TR1 to
antenna elements 200 and then switch TR1 switches to a second mode
wherein the antenna elements 200 are coupled to receiver 226 for a
period of time sufficient to receive an echo signal from a target
at a selected range. Thereafter, the transmit, followed by RECEIVE
mode would be repeated. Transmitting antenna bias, for charging
elements 200, would occur after the discrete receiving period and
thereafter the process of transmitting and receiving would be
repeated.
[0083] FIGS. 8 and 9 diagrammatically illustrate an antenna
assembly wherein a multiple, in this case, 12, separate antenna
element sets, for example, as antenna 200, are employed, each being
spaced forward of a metal reflector 200b by a distance of
approximately 3 inches, for a nine-inch tip-to-tip antenna element
dimension. The antennas are supported by insulating standoffs 200c,
and switches 335 (transmitting mode) are shown to be fed by
triggering sources 323 which conveniently can be on the back side
of reflector 200b, and thus any stray radiation which might tend to
flow back beyond this location to a transmission line is
effectively shielded. The multiple antennas may be operated in
unison, that is, all of them being triggered (in the case of a
transmitter) and combined (in the case of a receiver) with like
timing, in which case the antenna would have a view or path normal
to the antenna array or surface of reflector 200b as a whole.
Alternately, where it is desired to effect beam steering, the
timing by combination, or triggering devices (receiving or
transmitting), would be varied. Thus, for example, with respect to
reception, while the outputs of all of the antennas in a column
might be combined at a like time point, outputs from other columns
might be delayed before a final combination of all signals. Delays
can simply be determined by lead lengths, and, in general, multiple
effects are achievable in almost limitless combinations.
[0084] Alternately, antenna elements may be arranged in an end-fire
format wherein each element is driven with or without a reflector.
They may be arrayed as illustrated in FIGS. 9a and 9b wherein four
end-fire unit Y1, Y2, Y3, and Y4 are employed and positioned in
front of a common reflector R1. Alternately, the reflector may be
omitted, and further alternately, an absorber may be positioned
behind the array.
[0085] FIG. 10 diagrammatically illustrates a transmitting switch
wherein the basic switching element is an avalanche mode operated
transistor 400, the emitter and collector of which are connected
through like resistors 402 to antenna elements A and B of antenna
200, the resistors being, for example, 25 ohms each (for an antenna
as shown in FIG. 6a, it would be doubled). In the time between the
triggering "on" of avalanche transistor 400, it is charged to a DC
voltage, e.g., 150 volts, which is coordinate with the avalanche
operating point of transistor 400. Charging is effected from (+)
and (-) supply terminals through like resistors 404 to antenna
elements A and B. The primary of pulse transformer 408 is supplied
a triggering pulse, as from trigger circuit 323 of FIG. 5, and its
secondary is connected between the base and emitter of transistor
400. Typically, the transmission line for the triggering pulse
would be in the form of a coaxial cable 410. When triggered "on,"
transistor 400 shorts antenna elements A and B and produces a
signal transmission from antenna 200 (or antenna 200a).
[0086] FIG. 11 illustrates a modified form of applying a charging
voltage to antenna elements A and B, in this case, via a constant
current source, and wherein the charging voltage is supplied across
capacitor 507 through coaxial cable 412, which also supplies a
triggering voltage to transformer 408, connected as described
above. For example, the (+) voltage is supplied to the inner
conductor of coaxial cable 412, typically from a remote location
(not shown). This voltage is then coupled from the inner conductor
of the coaxial cable through the secondary of pulse transformer 408
and resistor 414, e.g., having a value of 1K ohms, to the collector
of a transistor 416 having the capability of standing the bias
voltage being applied to switching transistor 400 (e.g., 150
volts). The (+) voltage is also applied through resistor 418, for
example, having a value of 220K ohms, to the base of transistor
416. A control circuit to effect constant current control is formed
by a zener diode 420, across which is capacitor 422, this zener
diode setting a selected voltage across it, for example, 7 volts.
This voltage is then applied through a variable resistor 424 to the
emitter of transistor 416 to set a constant voltage between the
base and emitter and thereby a constant current rate of flow
through the emitter-collector circuit of transistor 416, and thus
such to the antenna. Typically, it is set to effect a full voltage
charge on antenna 200 in approximately 90% of the time between
switch discharges by transistor 400. The thus regulated charging
current is fed through resistors 406 to antenna elements A and B.
In this case, discharge matching load resistors 402 are directly
connected between transistor 400 and antenna elements A and B as
shown.
[0087] FIG. 12 illustrates the employment of a light responsive
element as a switch, such as a light responsive avalanche
transistor 423, alternately a bulk semiconductor device, or a bulk
crystalline material such as diamond, would be employed as a
switch, there being switching terminals across, on opposite sides
of, the bulk material. The drive circuit would be similar to that
shown in FIG. 10 except that instead of an electrical triggering
system, a fiber optic 426 would provide a light input to the light
responsive material, which would provide a fast change from high to
low resistance between terminals to effect switching.
[0088] FIG. 13 bears similarity to both FIGS. 11 and 12 in that it
employs a constant current power source with light responsive
switching element 423, such as a light responsive transistor, as
shown. Since there is no coaxial cable for bringing in triggering
signals, other means must be provided for bias voltage. In some
applications, this may simply be a battery with a DC-to-DC
converter to provide the desired high voltage source at (+) and (-)
terminals.
[0089] FIGS. 14 and 15 illustrate the employment of multiple
switching elements, actually there being shown in each figure two
avalanche mode operated transistors 450 and 452 connected
collector-emitter in series with resistors 402 and antenna elements
A and B. AS will be noted, separate transformer secondary windings
of trigger transformer 454 are employed to separately trigger the
avalanche mode transistors. The primary winding of a transformer
would typically be fed via a coaxial cable as particularly
illustrated in FIG. 10. Antenna elements A and B (either 200 or
201) are charged between occurrences of discharge from (+) or (-)
supply terminals, as shown.
[0090] FIG. 15 additionally illustrates the employment of a
constant current source as described for the embodiment shown in
FIGS. 11 and 13. Actually, the system of feeding the constant
current source through coaxial cable as shown in FIG. 11 can
likewise be employed with the circuitry shown in FIG. 14.
[0091] Referring to FIG. 16, there is illustrated a radar system
particularly intended for facility surveillance, and particularly
for the detection of moving targets, typically people. Transmitter
500 includes a 16-Mhz clock signal which is generated by signal
generator 501. This signal is then fed to -16 divider 502 to
provide output signals of 1 Mhz. One of these 1-Mhz outputs is fed
to 8-bit counter 504 which counts up to 256 and repeats. The other
1-mHz output of -16 divider 502 is fed through a programmable
analog delay unit 506 wherein each pulse is delayed by an amount
proportional to an applied analog control signal. Analog delay unit
506 is controlled by a magnitude of count from counter 504, which
is converted to an analog voltage proportional to this count by D/A
converter 509 and applied to a control input of analog delay unit
506.
[0092] By this arrangement, each of the 1-mHz pulses from -16
divider 502 is delayed a discrete amount. The pulse is then fed to
fixed delay unit 508 which, for example, delays each pulse by 60
nanoseconds in order to enable sufficient processing time of signal
returns by receiver 510. The output of fixed delay unit 508 is fed
to trigger generator 512, for example, an avalanche mode operated
transistor, which provides a fast rise time pulse. Its output is
applied to switch 514, typically an avalanche mode operated
transistor as illustrated in FIG. 10 or 11. Antenna 200 (or 201) is
directly charged through resistors 503 from a capacitor 507 (FIG.
11) which generally holds a supply voltage provided at the (+) and
(-) terminals.
[0093] Considering now receiver 510, antenna 513, identical with
antenna 200 or 201, receives signal returns and supplies them to
mixer 514. Mixer 514 multiplies the received signals from antenna
513 with locally generated ones from template generator 516.
Template generator 516 is triggered via a delay chain circuitry of
analog delay unit 506 and adjustable delay unit 518, which is set
to achieve generation of a template signal at a time corresponding
to the sum of delays achieved by fixed delay 508 and elapsed time
to and from a target at a selected distance. The output of mixer
514 is fed to short-term analog integrator 520 which discretely
integrates for the period of each template signal. Its output is
then fed to long-term integrator 522 which, for example, may be an
active low pass filter and integrates over on the order of 50
milliseconds, or, in terms of signal transmissions, up to, for
example, approximately 50,000 such transmissions. The output of
integrator 522 is amplified in amplifier 524 and passed through
adjustable high pass filter 526 to alarm 530. By this arrangement,
only AC signals corresponding to moving targets are passed through
the filters and with high pass filter 526 establishing the lower
velocity limit for a target and integrator-low pass filter 522
determining the higher velocity of a target. For example, high pass
filter 526 might beset to pass signals from targets at a greater
velocity than 0.1 feet per second and integrator-low pass filter
522 adapted to pass signals representing targets moving less than
50 miles per hour. Assuming that the return signals pass both such
filters, alarm visual would be operated.
[0094] FIG. 17 illustrates a modification of FIG. 16 for the
front-end portion of receiver 510. As will be noted, there are two
outputs of antenna 200, one to each of separate mixers 650 and 652,
mixer 650 being fed directly an output from template generator 618,
and mixer 652 being fed an output from template generator 618 which
is delayed 0.5 nanosecond by 0.5 nanosecond delay unit 654. The
outputs of mixers 650 and 652 are then separately integrated in
short-term integrators 656 and 658, respectively. Thereafter, the
output of each of these short-term integrators is fed to separate
long-term integrators 660 and 662, after which their outputs are
combined in differential amplifier 664. The output of differential
amplifier 664 is then fed to high pass filter 526 and then to alarm
530, as discussed above with respect to FIG. 16. Alternately, a
single long-term integrator may replace the two, being placed after
differential amplifier 664.
[0095] By this technique, there is achieved real time
differentiation between broad boundary objects, such as trees, and
sharp boundary objects, such as a person. Thus, assuming that in
one instance the composite return provides a discrete signal and
later, for example, half a nanosecond later, there was no change in
the scene, then there would be a constant difference in the outputs
of mixers 650 and 652. However, in the event that a change
occurred, as by movement of a person, there would be changes in
difference between the signals occurring at the two different
times,SPACE HERE and thus there would be a difference in the output
of differential amplifier 664. This output would then be fed to
high pass filter 526 (FIG. 16) and would present a discrete change
in the signal which would, assuming that it met the requirements of
high pass filter 526 and integrator-low pass filters 660 and 662
(FIG. 17, be signalled by alarm 530.
[0096] In terms of a system as illustrated in FIG. 16, it has been
able to detect and discriminate very sensitively, sensing when
there was a moving object within the bounds of velocities described
and within the range of operation, several hundred feet or more.
For example, movement of an object within approximately a 1-foot
range of a selected perimeter of measurement is examinable, leaving
out sensitivity at other distances which are neither critical nor
desirable in operation. In fact, this feature basically separates
the option of this system from prior systems in general as it
alleviates their basic problem: committing false alarms. Thus, for
example, the present system may be positioned within a building and
set to detect movement within a circular perimeter within the
building through which an intruder must pass. The system would be
insensitive to passersby just outside the building. On the other
hand, if it is desirable to detect people approaching the building,
or, for that matter, approaching objects inside or outside the
building, then it is only necessary to set the range setting for
the perimeter of interest. In general, walls present no barrier. In
fact, in one test, an approximately 4-foot thickness of stacked
paper was within the perimeter. In this test, movement of a person
just on the other side of this barrier at the perimeter was
detected.
[0097] While the operation thus described involves a single
perimeter, by a simple manual or automatic adjustment, observations
at different ranges can be accomplished. Ranges can be in terms of
a circular perimeter, or, as by the employment of a directional
antenna (antenna 200 with a reflector) or yagi-type array, effect
observations at a discrete arc.
[0098] FIG. 18 illustrates an application of applicant's radar to a
directional operation which might cover a circular area, for
example, from 20 to 30 feet to several thousand feet in radius. In
this illustration, it is assumed that there is positioned at a
selected central location a transmit antenna, in this case,
oriented vertically as a non-directional, or omni-directional,
antenna 700. There are then positioned at 120 points around it like
received antennas 702, 704, and 706. An antenna 700, e.g., as
previously described, is powered by a trigger switch transmitter
707. Assuming that a single signal burst is transmitted from
transmit antenna 700, it would be radiated around 360 and into
space. At some selected time as discussed above, receivers 708,
710, and 711 would be supplied a template signal as described above
to thus, in effect, cause the receivers to sample a signal echo
being received at that precise instant. This process would be
repeated for incrementally increasing or deceasing times, and thus
there would be stored in the memory's units 712, 714, and 716
signals representative of a range of transit times. Then, by
selection of a combination of transit times for each of the
receivers, in terms of triangularizations, it is possible to select
stored signals from the memory units representative of a particular
location in space. For surveillance purposes, the result of signals
derived from one scan and a later occurring scan would be digitally
subtracted, and thus there an object at some point within the range
of the unit has moved to a new location, there will then be a
difference in the scan information. This thus would signal that
something may have entered the area. This process in general would
be controlled by a read-write control 718 (FIG. 18a) which would
control the memory's units 712, 714, and 716 and would control a
comparator 720 which would receive selected values X, Y, and Z from
memory units 712, 714, and 716 to make the subtraction. Display
722, such as an oscilloscope, may be employed to display the
relative position of an object change with respect to a radar
location.
[0099] FIG. 19 illustrates an application of applicant's invention
to a radar system wherein there is one transmitting antenna, e.g.,
antenna 200, located in a discrete plane position with respect to
the direction of observation, three receiving antennas spaced in a
plane parallel to the first plane, and a fourth receiving antenna
positioned in a third plane. Thus, responsive to transmitter or
transmitter switch 802, radiation from transmitting antennas 200,
which is reflected by a target, is received by the four receiving
antennas at varying times by virtue of the difference in path
length. Because of the unique characteristic of applicant's system
in that it can be employed to resolve literally inches, extreme
detail can be resolved from the returns. Control 800 directs a
transmission by a transmitter 802, which supplies a signal burst to
transmitting antenna 200. Signal returns are received by antennas
806, 808, and 810 and are located, for example, in a plane
generally normal to the direction of view and separate from the
plane in which transmit antenna 200 is located. A fourth receiving
antenna 812 is located in still a third plane which is normal to
the direction of view and thus in a plane separate from the plane
in which the other receiving antennas are located. By virtue of
this, there is provided means for locating, via triangularization,
a target in space, and thus there is derived sufficient signal
information to enable three-dimensional information displays. The
received signals from receivers 811, 814, 816, and 818 are
separately supplied to signal processor and comparator 820, which
includes a memory for storing all samples received and in terms of
their time of receipt. From this data, one can compute position
information by an appropriate comparison as well as target
characteristics, such as size and reflectivity, and can be
displayed on display 822.
[0100] FIG. 20 illustrates a portion of a radar system generally
shown in FIG. 5 except that the pulse output of switch 335 is
applied through an impedance matching device, i.e., resistor 900,
to wideband sonic transducer 902. Sonic transducer 902 is a known
structure, it being, for example, constructed of a thin
piezoelectric film 904 on opposite sides of which are coated
metallic films 906 and 908 as electrodes. The energizing pulse is
applied across these plates. Impedance matching is typically
required as switch 335 would typically supply a voltage from a
relatively low impedance source whereas sonic transducer 902
typically would have a significantly higher impedance. The sonic
output of sonic transducer 902, a wide frequency band, on the order
of at least three octaves, would typically be attached to an
impedance transformer for the type of medium into which the sonic
signal is to be radiated; for example, transducer 902 would attach
to a low impedance material 903, such as glass, in turn mounted on
a support 905 (for example, the hull of a ship).
[0101] An echo or reflection from a target of the signal
transmitted by sonic transducer 902 would be received by a
similarly configured sonic transducer 910, and its output would
then be coupled via plates 912 and 914 to amplifier 28 and thence
onto mixer 330 as illustrated in FIG. 5 wherein operation would be
as previously described.
[0102] FIG. 21 illustrates a broadband light transmitter. With
respect to a first version, with switches 929 and 929a in the
indicated positions, a pulse as from switch 335 (FIG. 5) triggers a
conventional laser 922 operating, for example, in a conventional
narrow frequency mode at approximately 700 nanometers to provide
such an output to a narrow band to wideband light converter
assembly consisting of light modulator 924 and a dispersive medium
926. The output of laser 922 is applied to one end 928 of a fiber
optic 923 having a variable refractive index as a function of an
applied voltage and, in this case, for example, having a thickness
dimension on the order of 2 millimeters and a length dimension of
approximately 1 meter. The fiber optic is positioned between two
elongated metallic or otherwise conductive plates 930 and 932. A
modulating voltage from signal generator 934, for example, a ramp
voltage, is applied across the plates adjacent to the exiting end
of fiber optic 923 and terminated by resistor 939 as a load and
ground. Plate 932 is grounded at both ends to prevent destructive
reflections. Generator 934 typically would be triggered also by
switch 335 to create, in this example, a ramp voltage which would
effect a traveling wave from right to left along the plates and
thus along the enclosed fiber optic, opposing the traveling light
pulse from left to right. As a result, there is effected a light
output at end 936 which varies, changing from the initial
wavelength of the input light pulse to a higher or lower frequency,
and this, in effect, creates a chirp-type pulse. It is then
supplied to a dispersive material 926 such as lead glass, with the
result that at its output, the resultant light pulse is converted
to a quite short duration pulse having a wide broadband spectrum of
frequencies, or white or near white light output. Emitted beam 938
then travels outward, and upon striking a target, a reflection is
reflected back to optical mixer 940 which is also supplied a laser
output pulse from laser 922 (e.g., by a beam splitter), in turn
triggered by a selectably variable delay line 942, being delayed in
terms of selected range. As a result, optical mixer 940 multiplies
the two input signals, a template signal and a received signal, and
provides a multiplied output to integrator 950, and the signals are
then processed as generally described with respect to FIG. 5.
[0103] It is believed of perhaps greater significance that light
modulator 924, a light frequency modulator, has many other
applications, particularly as an intelligence modulator of a laser
beam.
[0104] FIG. 22 illustrates a modification of the transmitter shown
in FIG. 20, illustrating the technique of frequency modulation
multiplexing of a plurality of intelligence signals. In this case,
the same optical assembly 924 is illustrated as in FIG. 20, leaving
out signal generator 934 and switch 335. Further, the dispersive
material 926 would not be needed. Thus, there is provided to plate
930 a plurality of frequency modulated multiplexed signals in place
of a radar type signal. Two frequency modulation signals are
illustrated, and with respect to one of them, it would take this
form. An IF source 941 would generate a first intermediate
frequency signal, typically being small with respect to the
frequency of the laser beam itself. Its output would be fed to
frequency modulator 942 which would then frequency modulate the
applied IF frequency over a desired frequency deviation, typically
depending upon the bandwidth of the intelligence signal applied to
it, and it would be supplied as a first intelligence signal as
shown. Thus, the output of frequency modulator 942 would be
provided as one input to plate 930 of the light modulator 924,
being applied across summing resistor 944. As an illustration of
multiplexing, a second IF frequency would be generated by IF source
946 at a different frequency than that generated by IF source 941,
and it would be applied to frequency modulator 948, which in turn
would receive a second intelligence signal. As a result, frequency
modulator 948 would provide a selected frequency deviation of the
IF frequency applied to it, and its output would also be provided
to light modulator 924 across summing resistor 944. The combined
outputs of modulators 942 and 948 would then be transmitted by
optical modulator 924.
[0105] Referring now to FIG. 23, which shows a receiver for the
transmitter shown in FIG. 22, the signal output 938 of optical
modulator 924 would be received in the receiver by optical detector
982 which would provide an electrical output to mixer 984 to which
is also applied the two IF frequencies generated in FIG. 22, one by
a local oscillator 986 and the other by oscillator 988. As a
result, mixer 984 provides an output, being the first IF frequency
modulation and a second frequency modulation, these being applied
separately to signal discriminators 990 and 992 to thus provide
typical analog outputs of the two modulations effected by the
system shown in FIG. 22. Of course, where digital signals are
involved, accordingly, the output of signal discriminators 990 and
992 would provide discrete outputs representative of the modulated
levels for digital signals, either being of the multi-level type or
binary type.
[0106] Of course, in a typical installation, there could be many,
many separate signal discriminators, each providing a frequency
modulated output of one set of intelligence. Thus in the system
just described, there is provided a frequency modulated multiplex
system which not only can carry many, many different signals, but
also is quite cheap to construct, certainly much cheaper than the
present system of high-speed digital communications.
* * * * *