U.S. patent application number 09/821988 was filed with the patent office on 2002-10-03 for power supply circuit and method.
Invention is credited to Basso, Christophe, Goyhenetche, Philippe, Omet, Dominique.
Application Number | 20020140501 09/821988 |
Document ID | / |
Family ID | 25234796 |
Filed Date | 2002-10-03 |
United States Patent
Application |
20020140501 |
Kind Code |
A1 |
Goyhenetche, Philippe ; et
al. |
October 3, 2002 |
Power supply circuit and method
Abstract
An integrated switching mode power supply (10) has a follower
device (59) providing a supply voltage (V.sub.BOOT) to a node (70)
of the power supply. A driver circuit operates in response to an
input signal (V.sub.CONTROL) and has an output (40) for providing a
drive signal (V.sub.DRIVE) that bootstraps the node to a potential
greater than the supply voltage.
Inventors: |
Goyhenetche, Philippe;
(Fonsorbes, FR) ; Omet, Dominique; (Toulouse
cedex, FR) ; Basso, Christophe; (Toulouse Cedex,
FR) |
Correspondence
Address: |
Robert D. Atkins
ON Semiconductor
Patent Administration Dept - MD A230
P.O. Box 62890
Phoenix
AZ
85082-2890
US
|
Family ID: |
25234796 |
Appl. No.: |
09/821988 |
Filed: |
April 2, 2001 |
Current U.S.
Class: |
327/589 ;
363/59 |
Current CPC
Class: |
H03K 17/063 20130101;
H02M 7/538 20130101; H02M 1/0006 20210501 |
Class at
Publication: |
327/589 ;
363/59 |
International
Class: |
H02M 007/00; H02M
003/18 |
Claims
What is claimed is:
1. An integrated circuit, comprising: a follower device coupled for
providing a supply voltage to a node of the integrated circuit; and
a driver circuit operating in response to an input signal and
having an output for providing a drive signal that bootstraps the
node to a potential greater than the supply voltage.
2. The integrated circuit of claim 1, further comprising a
capacitor coupled between the node and the output of the drive
circuit.
3. The integrated circuit of claim 2, wherein the follower device
comprises a transistor having a conduction electrode coupled to the
node, and a charge on the capacitor turns off the transistor as the
drive signal increases.
4. The integrated circuit of claim 3, wherein the transistor has a
control electrode coupled to a supply terminal of the integrated
circuit, further comprising a diode coupled to the supply terminal
for limiting the supply voltage.
5. The integrated circuit of claim 4, wherein the transistor
comprises a bipolar transistor having an emitter coupled to the
node and a base coupled to the supply terminal.
6. The integrated circuit of claim 1, further comprising a clamping
device that breaks down to limit a potential on the node to a
predefined level.
7. The integrated circuit of claim 6, wherein the clamping device
comprises a first diode coupled between the node and a supply
terminal of the integrated circuit.
8. The integrated circuit of claim 7, wherein the first diode
comprises an avalanche diode.
9. The integrated circuit of claim 1, wherein the driver circuit
includes a first transistor having a conduction electrode coupled
to the output and a control electrode coupled to the node.
10. The integrated circuit of claim 1, further comprising a
semiconductor package for housing the driver circuit and the
follower device.
11. A power supply, comprising: a driver having an input for
amplifying an input signal for producing an output signal
(V.sub.DRIVE) at an output; a capacitor coupled to a node of the
driver for bootstrapping the output signal; and a first diode
coupled to the node for breaking down to limit a magnitude of the
output signal.
12. The power supply of claim 11, wherein the first diode has an
anode coupled to a terminal of the power supply and an cathode
coupled to the node.
13. The power supply of claim 11, further comprising a follower
device for providing a supply voltage to the node.
14. The power supply of claim 13, further comprising a second diode
having a cathode coupled to a control electrode of the follower
device and an anode coupled to the terminal of the power
supply.
15. A method of amplifying, comprising the steps of: bootstrapping
a node in response to an input signal to produce a drive signal;
and avalanching a clamping device to limit the drive signal to a
predefined level.
16. The method of claim 15, wherein the step of bootstrapping
includes the step of charging a capacitance with the drive
signal.
17. The method of claim 16, further comprising the step of biasing
the node through a follower device.
18. The method of claim 17, wherein the step of bootstrapping
includes the step of turning off the follower device with the
capacitance.
19. The method of claim 16, wherein the step of avalanching
includes the step of avalanching a diode to discharge the
capacitance.
20. The method of claim 15, further comprising the step of
switching a coil current with the drive signal.
Description
BACKGROUND OF THE INVENTION
[0001] The present invention relates in general to semiconductor
devices and, more particularly, to switch mode power supplies used
in battery chargers.
[0002] The global nature of technology creates a demand for
"universal" devices that can operate in most if not all countries.
For example, portable devices such as notebook computers and
digital cameras can operate in multiple countries because they are
powered from batteries rather than a local alternating current (AC)
power source. However, the battery chargers used to recharge the
batteries do operate from a local AC power source, and therefore
often do not operate in multiple countries due to different AC
power standards. Worldwide, AC power is provided at a voltage level
ranging from about eighty volts root mean square (RMS) to about
three hundred sixty volts RMS.
[0003] A battery charger typically includes a power supply whose
integrated circuits and other electrical components process the
incoming AC power to produce a direct current (DC) supply voltage
for charging a battery. However, many of the power supplies'
components cannot function over the necessary voltage range. A
charger configured to operate at two hundred volts RMS may incur
damage if used in a country providing three hundred volts RMS. On
the other hand, if the AC power provides only eighty volts RMS, the
components may not receive enough voltage to function correctly,
which can damage the battery.
[0004] Most previous power supplies function over a limited voltage
range, and therefore can operate in only one country. Chargers
using these power supplies have a high cost because manufactures
use different designs and/or components for each country, thereby
losing the economy of scale. Other battery chargers can operate in
multiple locations but require external controls such as
user-operated switches to select an AC voltage appropriate for the
location. Such chargers have lower design costs but are susceptible
to damage if a user inadvertently selects the wrong voltage level.
The external controls are inconvenient for the user and also
increase the component count, which increases the fabrication cost
of the charger.
[0005] Hence, there is a need for a circuit and method of providing
a supply voltage which can operate from a wide range of supply
voltages without incurring damage while reducing the manufacturing
cost by reducing the number of external user controls.
BRIEF DESCRIPTION OF THE DRAWINGS
[0006] FIG. 1 is a schematic diagram of a battery charger; and
[0007] FIG. 2 is a schematic diagram of a driver circuit of the
battery charger.
DETAILED DESCRIPTION OF THE DRAWINGS
[0008] In the figures, elements having the same reference numbers
have similar functionality.
[0009] FIG. 1 is a schematic diagram of a battery charger 10
coupled for charging a battery 8. Battery charger 10 includes an
alternating current (AC) to direct current (DC) converter 12, a
transformer 14, diodes 15-16, capacitors 17-18, an optoisolator 20,
a control circuit 22, a drive circuit 24, a transistor 26 and a
resistor 28. Battery charger 10 functions as a power supply which
has a terminal 30 for coupling to a wall outlet (not shown) to
receive operating power from a power source designated as AC
voltage V.sub.AC. Depending on a country's local power standards,
voltage V.sub.AC can have a value ranging from about eighty volts
root-mean-square (RMS) to about three hundred sixty volts RMS at a
frequency from fifty to sixty hertz. This range incorporates most
if not all of the power standards in effect worldwide so that
battery charger 10 is considered to be a "universal" battery
charger. An output terminal 32 provides a DC charging voltage
V.sub.CH for recharging battery 8.
[0010] Converter 12 has an input coupled to terminal 30 for
receiving AC voltage V.sub.AC and an output coupled to a node 31
for providing a DC voltage V.sub.P. Converter 12 includes a
standard diode bridge network and a filter capacitor for producing
voltage V.sub.P as a rectified and filtered voltage. Voltage
V.sub.P essentially operates at a peak value of voltage V.sub.AC,
and therefore ranges from about one hundred ten volts to over five
hundred volts in accordance with the range of voltage V.sub.AC.
[0011] Transformer 14 has a primary winding 34, a secondary forward
winding 36 and a secondary flyback winding 38. A switched current
I.sub.P flows through primary winding 34 as transistor 26 switches.
Current I.sub.P induces AC voltages V.sub.S1 and V.sub.S2 across
secondary windings 36 and 38, respectively. Voltages V.sub.S1, and
V.sub.S2 are rectified by diodes 15-16 and filtered by capacitors
17-18 to produce rectified and filtered DC signals V.sub.CH and
V.sub.CC on terminals or nodes 32 and 33, respectively. Voltage
V.sub.S1 has a value determined by the turns ratio of secondary
winding 36 to primary winding 34, and therefore has a range as
broad as that of AC voltage V.sub.AC. In one embodiment, voltage
V.sub.S1 has a value between about eight volts and about forty
volts.
[0012] Signal V.sub.S1 is in phase with primary voltage V.sub.P,
while signal V.sub.S2 is out of phase, so diodes 15 and 16 are
forward biased, and therefore conduct, on alternate cycles. That
is, when voltage V.sub.S1 is positive to forward bias diode 15,
voltage V.sub.S2 is negative to reverse bias diode 16, and vice
versa. Hence, when transistor 26 turns on and current I.sub.P flows
through primary winding 34, an induced current flows through
forward winding 36 but not flyback winding 38. When transistor 26
turns off, I.sub.P is zero and no current flows through forward
winding 36, but energy stored in primary winding 34 on the previous
cycle induces a current flow in flyback winding 38. In effect,
energy is transferred to forward winding 36 when transistor 26 is
on and to flyback winding 38 when transistor 26 turns off. Since
voltage V.sub.CC provides the supply voltage for control circuit 22
and drive circuit 24, the opposing phase relationship of secondary
windings 36 and 38 ensures that the value of V.sub.CC is
substantially unaffected by a high current flowing through node 32.
As a result, battery charger 10 continues to function properly even
if a high current from a shorted or discharged battery forces node
32 near ground potential.
[0013] Voltage V.sub.S2 is a regulated voltage whose value is set
by the type of battery which battery charger 10 is designed to
recharge. Regulation is achieved by a feedback loop from node 32
through optoisolator 20 to a node 35 at an input of control circuit
22. Optoisolator 20 includes a light emitting diode and a
phototransistor for optically coupling information regarding the
level of voltage V.sub.S2 from node 32 to a node 35 as feedback
signal V.sub.FB. Voltage V.sub.S2 has a value ranging from about
six volts to about nine volts, depending on the type of battery
being recharged. In one embodiment, voltage V.sub.S2 is regulated
at six volts. In accordance with safety standards, optoisolator 20
provides at least four thousand volts of electrical isolation
between nodes 32 and 35 to achieve safe operation of battery
charger 10.
[0014] Control circuit 22 comprises a microcontroller that is
programmed to control the recharging cycle of a battery. A first
feedback input receives feedback signal V.sub.FB1 on node 35 to
indicate the level of charging voltage V.sub.CH. Control circuit 22
includes circuitry to generate a first reference signal and a first
comparator for comparing V.sub.FB1 to the first reference signal.
The result of the comparison is processed to produce a pulse width
modulated control signal V.sub.CONTROL at an output at a node 39 to
maintain voltage V.sub.S2 at the desired amplitude. A second
feedback input is coupled to a node 41 to receive feedback signal
V.sub.FB2, which indicates the current flow through transistor 26
and resistor 28. Control circuit 22 further includes circuitry to
generate a second reference signal and a second comparator for
comparing the second reference to V.sub.FB2. When the current
flowing through resistor 28 reaches a predetermined level, a pulse
of V.sub.CONTROL may be truncated to limit the current flowing
through transistor 26.
[0015] Drive circuit 24 operates as an amplifier that has an input
coupled to node 39 for receiving V.sub.CONTROL pulses and an output
at a node 40 for producing a drive signal V.sub.DRIVE. The
component pulses of drive signal V.sub.DRIVE swing from a low logic
level of approximately zero volts to a high logic level of between
7.2 and fifteen volts. A supply terminal operates from voltage
V.sub.CC, which ranges between about eight and about forty volts.
In one embodiment, drive circuit 24 is formed on an integrated
circuit substrate for housing in a semiconductor package 25.
[0016] Transistor 26 is configured as an n-channel enhancement mode
metal-oxide-semiconductor field effect transistor (MOSFET). Drive
signal V.sub.DRIVE is coupled to the gate of transistor 26 for
switching current through primary winding 34. The gate electrode
has a breakdown voltage of twenty volts or less, so it is necessary
that V.sub.DRIVE pulses be limited to a lower amplitude. Transistor
26 is configured as a high current device and therefore has a high
effective gate capacitance. In one embodiment, the gate capacitance
is at least one nanofarad.
[0017] Current through transistor 26 develops a voltage V.sub.FB2
across resistor 28 which is fed back to control circuit 22 to set a
current limit through transistor 26. In one embodiment, resistor 28
has a value of one ohm to set a maximum current through transistor
26 of two-hundred fifty milliamperes.
[0018] FIG. 2 is a schematic diagram showing drive circuit 24 in
further detail, including predrivers 52 and 54, diodes 56-57,
transistors 58-63, a capacitor 64 and a resistor 65. Drive circuit
24 operates as a push-pull amplifier receiving control signal
V.sub.CONTROL at node 39 and producing drive signal V.sub.DRIVE at
node 40. Operating power is supplied by voltage V.sub.CC at node
33.
[0019] Diode 56 is configured as an avalanche diode that avalanches
or breaks down when voltage V.sub.CC is greater than about ten
volts. The avalanching clamps the potential on a node 66 at ten
volts to avoid subjecting low voltage components of drive circuit
24 to high voltage damage. The low voltage components operate from
internal supply voltages V.sub.PD and V.sub.BOOT which are derived
from the node 66 potential and produced at the emitters of
transistors 58-59. V.sub.PD and V.sub.BOOT have a value of about
7.2 volts when V.sub.CC is eight volts, increasing to about 9.2
volts when V.sub.CC is ten volts and clamped at 9.2 volts when
V.sub.CC is greater than ten volts. By clamping V.sub.PD and
V.sub.BOOT at 9.2 volts, components of drive circuit 24 can be made
smaller, which reduces the die size and manufacturing cost.
[0020] Resistor 65 is selected to provide about one microamperes of
base current to transistors 58-59 and about nine microamperes of
breakdown current through diode 56 when V.sub.CC operates at eight
volts. When V.sub.CC has a higher value, additional current is
shunted through diode 56.
[0021] Transistors 58-59 comprise NPN bipolar transistors operate
as follower devices or emitter followers to provide V.sub.PD and
V.sub.BOOT on nodes 68 and 70, respectively, with a low emitter
impedance. In one embodiment, transistor 59 has a base-emitter
breakdown of about eight volts. Alternatively, transistors 58-59
may be MOSFETs operating as source followers.
[0022] Predrivers 52 and 54 operate as amplifiers that boost the
current capability of V.sub.CONTROL pulses so that drive signal
V.sub.DRIVE has fast switching transitions when driving the high
gate capacitance of transistor 26. Transistors 60-61 function as an
inverter stage so that the gates of transistors 62-63 are driven
with opposite polarities. That is, the signal at the gate of
transistor 62 has the opposite polarity as the signal at the gate
of transistor 63 to ensure that transistor 62 is turned on when
transistor 63 is turned off, and vice versa.
[0023] Transistors 62-63 comprise n-channel MOSFETS coupled to
function as a push-pull output stage. Transistors 62-63 have
conduction thresholds of about one volt and are selected to have a
size adequate to drive the capacitance of transistor 26 while
switching with a short transition time. Transistors 62-63 have
relatively thin gate oxides which are specified to break down if
the voltages on their respective gates exceeds eighteen volts. To
further reduce positive V.sub.DRIVE voltage transitions, the drive
signal of transistor 62 is bootstrapped with capacitor 64.
[0024] The operation of the output stage is described as follows.
Assume that V.sub.CC is greater than ten volts, so the potential of
node 66 is clamped at ten volts. Voltages V.sub.PD and V.sub.BOOT
initially operate at about 9.2 volts. Further assume that
V.sub.CONTROL is logic high, so nodes 70 and 74 are driven to a
potential of about 9.2 volts and node 72 is at ground potential.
Hence, transistor 63 is turned on while transistor 62 is turned
off. Drive voltage V.sub.DRIVE is at ground potential, charging
capacitor 64 through transistors 59 and 63 to a potential of about
9.2 volts. In one embodiment, capacitor 64 has a capacitance of ten
picofarads.
[0025] When V.sub.CONTROL goes to a logic low, node 74 is at ground
potential to turn off transistor 63 and turn on transistor 60. As
the potential of node 72 increases to about one volt, transistor 62
turns on to pull node 40 high to begin a positive V.sub.DRIVE
transition. The charge on capacitor 64 causes V.sub.BOOT to rise,
turning off transistor 59 by reverse biasing its base-emitter
junction. Node 72 is pulled more positive through transistor 60,
which increases the gate drive of transistor 62 to reduce the
positive going transition time of V.sub.DRIVE.
[0026] Diode 57 is selected to have an avalanche or breakdown
voltage of fifteen volts in order to avoid breaking down the
emitter-base junction of transistor 59. Hence, during a positive
transition, as V.sub.BOOT rises above its 9.2 volt level to fifteen
volts, diode 57 avalanches, discharging capacitor 64 and clamping
the potential of node 70. This clamping effectively limits the
reverse bias of the emitter-base junction of transistor 59 to about
five volts, thereby avoiding a damaging breakdown of transistor 59.
Clamping has the further advantage of protecting the gate of
transistor 62 from a thin oxide breakdown. Clamping further limits
the positive swing of V.sub.DRIVE to about fourteen volts to avoid
and excessive voltage that could damage transistor 26. Note that
the avalanche current through diode 57 is effectively limited to
the discharging current of capacitor 64. Hence, no direct current
flows through diode 57, so drive circuit 24 operates at a low
power.
[0027] When V.sub.CONTROL goes to a logic high, node 70 is at a
potential of about 9.2 volts, turning on transistor 61 and pulling
node 72 to ground potential to turn off transistor 62. Node 74 is
at a potential of 9.2 volts, which turns on transistor 63 and
drives V.sub.DRIVE to ground potential. Transistor 59 turns on as
V.sub.BOOT is pulled down through capacitor 64 and its low emitter
impedance maintains node 70 at a potential of 9.2 volts. Capacitor
64 is charged to 9.2 volts through transistors 59 and 63 to end the
cycle.
[0028] During standard operation, transistor 59 has a low emitter
impedance to maintain node 66 at a constant potential for reducing
switching noise. During bootstrap operation, the emitter-base
junction of transistor 59 reverse biases to provide a high
impedance to allow charge stored on capacitor 64 to turn off
transistor 59 and bootstrap node 70 to a voltage higher than 9.2
volts.
[0029] By now it should be appreciated that the present invention
provides an integrated switching power supply circuit with fast
switching transitions and a wide operating voltage range. A
follower device provides a supply voltage to a node of the power
supply circuit, and a driver circuit operates in response to an
input signal to providing a drive signal that bootstraps the node
above the supply voltage. A clamping device limits the value and
range of the potential on the node to reduce component size and
prevent damage to power supply components.
* * * * *