U.S. patent application number 10/134241 was filed with the patent office on 2002-09-19 for processes for preparation of marek's disease virus using continuous avian cell lines.
Invention is credited to Rong, Sing, Sheppard, Michael G..
Application Number | 20020132337 10/134241 |
Document ID | / |
Family ID | 22339565 |
Filed Date | 2002-09-19 |
United States Patent
Application |
20020132337 |
Kind Code |
A1 |
Rong, Sing ; et al. |
September 19, 2002 |
Processes for preparation of marek's disease virus using continuous
avian cell lines
Abstract
An amplifier distortion reduction system obtains a distortion
signal from the amplifier output and feeds the distortion signal
back to the input side of the amplifier to cancel with the
distortion produced at the amplifier output. For example, a signal
to be amplified by an amplifier is received on a main signal path.
The amplifier produces an amplified output with a non-distortion
spectrum and a distortion spectrum. A sample of the amplified
output is produced from the main signal path and placed on a
feedback path. On the feedback path, the distortion spectrum is
obtained from the sample amplified output. The distortion spectrum
is phase and/or amplitude adjusted to produce the distortion
signal. The distortion signal is placed onto the main signal path
at the input side of the amplifier with the signal to be amplified
to destructively combine with the distortion produced from the
amplifier in amplifying the signal to be amplified.
Inventors: |
Rong, Sing; (Old Lyme,
CT) ; Sheppard, Michael G.; (North Stonington,
CT) |
Correspondence
Address: |
Kohn & Associates
Suite 410
30500 Northwestern Hwy.
Farmington Hills
MI
48334
US
|
Family ID: |
22339565 |
Appl. No.: |
10/134241 |
Filed: |
April 29, 2002 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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10134241 |
Apr 29, 2002 |
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09443800 |
Nov 19, 1999 |
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6410297 |
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60111627 |
Dec 9, 1998 |
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Current U.S.
Class: |
435/235.1 ;
424/204.1; 435/349 |
Current CPC
Class: |
A61P 31/22 20180101;
A61P 35/00 20180101; C12N 2710/16343 20130101; A61K 2039/525
20130101; A61P 31/16 20180101; C12N 2710/16361 20130101; C12N 7/00
20130101; C12N 2710/16334 20130101; C12N 2710/16351 20130101; A61P
31/20 20180101; A61P 31/12 20180101; A61K 39/12 20130101; A61K
39/255 20130101; C12N 15/86 20130101 |
Class at
Publication: |
435/235.1 ;
435/349; 424/204.1 |
International
Class: |
A61K 039/12; C12N
007/00; C12N 005/06 |
Claims
1. A method of producing an amplified signal, said method
comprising the steps of: amplifying an input signal on a signal
path by an amplifier on said signal path to produce said amplified
signal with distortion; obtaining a distortion signal on a feedback
path from said amplified signal with distortion; and placing said
distortion signal from said feedback path onto said signal path
prior to said amplifier to reduce distortion produced from said
amplifier.
2. The method of claim 1 comprising the steps of: sampling an input
signal to be amplified onto a carrier cancellation path; sampling
said amplified signal with distortion to produce a sample amplified
signal with distortion on a feedback path; combining said sample
input signal on said carrier cancellation path with said sample
amplified signal with distortion on said feedback path.
3. The method of claim 2 comprising the steps of: phase shifting of
said sample input signal on said carrier cancellation path prior to
said step of combining; and attenuating said sample amplified
signal with distortion prior to said step of combining.
4. The method of claim 3 comprising the step of: responding to an
output of said step of combining by adjusting said steps of phase
shifting said sample input signal and attenuating said sample
amplified signal.
5. The method of claim 4 comprising the steps of: sampling said
output of said step of combining; frequency down-converting said
output sample to baseband; low pass filtering said baseband output
sample to produce a residual non-distortion signal; generating a
control signal from said residual non-distortion signal using a
voltage differential amplifier and integrator.
6. The method of claim 4 wherein said input signal has multiple
carriers and said step of responding comprises the step of:
responding to said portion of said output corresponding to the
lowest carrier frequency of said multiple carriers.
7. The method of claim 1 comprising the steps of: phase shifting
and amplifying a distortion spectrum from said amplified signal
with distortion to produce said distortion signal; injecting said
distortion signal on said signal path before said amplifier to
cancel distortion generated by said amplifier in amplifying said
input signal to be amplified.
8. The method of claim 7 comprising the steps of: sampling said
amplified signal with distortion to produce a sample amplified
signal with distortion on said feedback path; sampling said sample
amplified signal with distortion to produce an output sample;
responding to said output sample to control said phase shifting and
amplifying said distortion spectrum.
9. The method of claim 8 wherein said step of responding comprising
the steps of: attenuating said output sample; frequency
down-converting said output sample to baseband; filtering said
output sample using a low pass filter and a high pass filter to
produce a residual distortion signal; and providing said residual
distortion signal to a voltage differential amplifier and
integrator to control said steps of phase shifting and amplifying
said distortion signal.
10. The method of claim 8 wherein said input signal has multiple
carriers and said step of responding comprises the step of:
responding to said portion of said output sample corresponding to
the highest carrier frequency of said multiple carriers.
11. The method of claim 1 comprising the steps of: sampling said
amplified signal with distortion on said signal path to produce a
sample amplified signal with distortion on said feedback path; and
filtering a non-distortion spectrum from said sample amplified
signal with distortion.
12. The method of claim 11 comprising the steps of: down-converting
said sample amplified signal with distortion on said feedback path
to an intermediate frequency (IF); producing an IF image of the
distortion spectrum portion on said feedback path by low pass and
high pass filtering of said IF sample amplified signal with
distortion; and up-converting said IF image of the distortion
spectrum portion on said feedback path to radio frequency (RF) and
recover a RF distortion spectrum.
13. The method of claim 11 comprising the steps of: down-converting
said sample amplified signal with distortion on said feedback path
to a baseband frequency; producing a baseband image of the
distortion spectrum portion on said feedback path by low pass and
high pass filtering of said baseband sample amplified signal with
distortion; and up-converting said image of said baseband
distortion spectrum portion on said feedback path to radio
frequency (RF) to recover a RF distortion spectrum.
14. An amplifier distortion reduction system, said system
comprising: an amplifier on a signal path adapted to amplify an
input signal on said signal path to produce an amplified signal
with distortion; and distortion feedback circuitry adapted to
obtain a distortion signal from said amplified signal with
distortion and to place said distortion signal on said signal path
prior to said amplifier to reduce distortion from said
amplifier.
15. The system of claim 14 wherein said distortion feedback
circuitry comprising: a splitting device adapted to place a sample
of said input signal to be amplified onto a carrier cancellation
path; a coupler on said signal path adapted to produce a sample of
said amplified signal with distortion on a feedback path; a
combining device adapted to combine said input signal sample on
said carrier cancellation path with said sample amplified signal
with distortion on said feedback path.
16. The system of claim 14 comprising: a coupler adapted to produce
on a feedback path a sample of said amplified signal with
distortion on said signal path; and said feedback distortion
circuitry includes a filtering arrangement adapted to reduce a
non-distortion spectrum of said sample amplified signal with
distortion on said feedback path.
17. The system of claim 16 wherein said feedback distortion
circuitry comprising: a down-converter adapted to down convert said
sample amplified signal with distortion on said feedback path; said
filtering arrangement adapted to reduce a down-converted
non-distortion spectrum of said down-converted amplified signal
with distortion to leave a down-converted distortion spectrum on
said feedback path; and an up-converter adapted to up-convert said
distortion spectrum on said feedback path.
18. The system of claim 17 wherein said down-converter is adapted
to down-convert said sample amplified signal with distortion on
said feedback path to an intermediate frequency; and said
up-converter is adapted to up-convert said distortion spectrum on
said feedback path to radio frequency.
19. The system of claim 17 wherein said down-converter is adapted
to down-convert said sample amplified signal with distortion on
said feedback path to a baseband frequency; and said up-converter
is adapted to up-convert said distortion on said feedback path to
produce said distortion spectrum on said feedback path at radio
frequency
Description
BACKGROUND OF THE INVENTION
[0001] 1. Field of The Invention
[0002] This invention relates to a radio frequency (RF) signal
amplification system producing an amplified signal with reduced
distortion.
[0003] 2. Description of Related Art
[0004] The relation between the output and the input of a RF power
amplifier may be linear or non-linear depending on the amplitude of
its input signal. "Linear" refers to the amplifier gain and phase
shift being constant. "Non-linear" means both gain and phase shift
are not constant. A distorted spectrum will be generated at the
output of an amplifier if it operates in non-linear region. For
example, if an input signal is a two single frequency carrier, an
amplifier operating non-linearly produces multiple intermodulation
(IMD) components at its output. If the input signal is a modulated
channel with a definite bandwidth, the spectrum shape will be
distorted (expanded, or what can be referred to as spectral
regrowth) at the output.
[0005] Here, amplifier non-linearity is viewed from a spectral
point of view. As such, no matter the type of input signal to be
amplified, whether single frequency carriers or modulated channels,
amplifier distortion will be described by spectral regrowth
(including intermodulation components or distorted spectrum shape).
The amplifier output waveform (spectrum) will be divided into
"distortion spectrum" and "non-distortion" spectrum. The latter is
the part with the same shape as the input signal and the former is
the regrown spectrum.
[0006] Spectral regrowth directly raises the ACPR (adjacent channel
power suppression ratio) and raises interference to adjacent
channels. In most of wireless communication systems there are
strong limitations to ACPR, and therefore, RF amplifiers have to be
linearized (to reduce spectral regrowth) to meet communication
requirements.
[0007] The simplest way to keep a radio frequency (RF) amplifier
working linearly is to keep its output power much lower than its
P.sub.1dB (1 dB output power suppression point) level.
Unfortunately, in this case, the amplifier efficiency is very low.
For example, a base station system requires ACPR of -56 dBc. If an
RF power amplifier has P.sub.1dB=40 dBm and IP.sub.3 (third order
intercept point) of -50 dBm, this amplifier can only operate to
output 22 dBm to satisfy the ACPR requirement. This output level is
18 dB lower than its rating level. In this case, the amplifier
efficiency will be very low (5% or less).
[0008] Various linearization methods are used to enable the use of
more cost-effective and more power efficient amplifiers while
maintaining an acceptable level of linearity. For example,
predistortion techniques are commonly used to improve the
performance of RF power amplifiers. Predistortion techniques
distort the input signal prior to amplification by taking into
account the transfer function characteristics for the amplifier.
Digital predistortion techniques can linearize RF amplifiers
effectively, but the circuit is complicated and costly.
Furthermore, predistortion systems have to cover a wide bandwidth
which is much wider than the bandwidth of the input signal, and
therefore, there is difficulty in getting fast digital processing
processors for real time linearization in case of wideband
applications.
[0009] Feed-forward correction is another approach for
linearization of RF power amplifiers. The basics of this technique
is to cancel distortion on the output side of the amplifier. There
are normally two additional loops in such kind of circuits. One
loop is to obtain the amplifier distortion and the other loop is to
amplify the distortion signal. The amplified distortion signal is
fed forward to the output side of the amplifier and is used to
cancel the distortion in the delayed main amplified signal. Because
the cancellation is taken at the output side of the amplifier, and
the fed forward distortion signal is coupled to the main signal
path through a coupler, the power needed for the feed forward
distortion signal is quite large. As such, an additional power
amplifier for the distortion signal is needed. The need for the
high power distortion amplifier (error amplifier) will considerably
reduce the total efficiency of the amplifier module, and the error
amplifier itself will introduce additional non-linearity
problems.
[0010] Thus, other improved amplifier linearization techniques are
desired, especially one which can reduce distortion over a wide
frequency band of operation and maintain a reasonably high
efficiency.
SUMMARY OF THE INVENTION
[0011] The present invention is an amplifier distortion reduction
system which obtains a distortion signal from the amplifier output
and feeds the distortion signal back to the input side of the
amplifier to cancel with the distortion produced at the amplifier
output. For example, a signal to be amplified by an amplifier is
received on a main signal path. The amplifier produces an amplified
output with a non-distortion spectrum and a distortion spectrum. A
sample of the amplified output is produced from the main signal
path and placed on a feedback path. On the feedback path, the
distortion spectrum is obtained from the sample amplified output.
The distortion spectrum is phase and/or amplitude adjusted to
produce the distortion signal. The distortion signal is placed onto
the main signal path at the input side of the amplifier with the
signal to be amplified to destructively combine with the distortion
produced from the amplifier in amplifying the signal to be
amplified.
BRIEF DESCRIPTION OF THE DRAWINGS
[0012] Other aspects and advantages of the present invention may
become apparent upon reading the following detailed description and
upon reference to the drawings in which:
[0013] FIG. 1 shows an embodiment of the amplifier distortion
reduction system using distortion feedback according to principles
of the present invention;
[0014] FIG. 2 shows an embodiment of the control loops used in an
embodiment of the amplifier distortion reduction system using
distortion feedback according to principles of the present
invention;
[0015] FIG. 3 shows an amplifier output spectrum without radio
frequency (RF) distortion feedback cancellation;
[0016] FIG. 4 shows an amplifier output spectrum with RF distortion
feedback cancellation according to principles of the present
invention;
[0017] FIG. 5 shows a multiple carrier embodiment of the amplifier
distortion reduction system using distortion feedback according to
principles of the present invention;
[0018] FIGS. 6a and 6b show embodiments of the control loops used
in an embodiment of a multiple carrier amplifier distortion
reduction system using distortion feedback according to principles
of the present invention;
[0019] FIG. 7 shows another embodiment of the amplifier distortion
reduction system using distortion feedback with conversion through
an intermediate frequency (IF) band according to principles of the
present invention;
[0020] FIG. 8 shows a multiple carrier embodiment of the amplifier
distortion reduction system using distortion feedback with
conversion through an intermediate frequency (IF) band according to
principles of the present invention; and
[0021] FIG. 9 shows an embodiment of the amplifier distortion
reduction system using distortion feedback with conversion through
a baseband frequency band according to principles of the present
invention.
DETAILED DESCRIPTION
[0022] Illustrative embodiments of an amplifier distortion
reduction system and method using distortion feedback to reduce
distortion according to the principles of the present invention are
described below. The described feedback amplifier system uses RF
distortion spectrum (or signal) cancellation to reduce distortion.
A distortion spectrum is obtained from the sampled output signal of
the main RF amplifier. In certain embodiments, the amplifier
distortion reduction system using distortion feedback, which can be
referred to as a feedback amplifier system, obtains on a feedback
path the distortion spectrum from the output of the power amplifier
by subtracting the non-distortion spectrum (which is a sample of
the input signal to the power amplifier) from a sample of the
output spectrum of the power amplifier. In other embodiments, the
distortion spectrum is obtained on the feedback path by filtering
out the non-distortion spectrum portion from a sample of the
amplifier output.
[0023] The distortion spectrum is properly phase shifted and
amplitude adjusted, then the resulting distortion signal is fed to
the input side of the main RF amplifier, combined with the
continuous input signal to be amplified. The amplified distortion
signal will cancel with the distortion produced from amplifying the
input signal. Because the distortion signal is typically much lower
in power than the signal to be amplified, the nonlinearity of the
amplifier to the distortion signal can be neglected. Moreover,
since the distortion signal is of low power, the system can be
implemented to avoid using extra power consuming devices on the
feedback path, and the main amplifier can have higher power
efficiency. This distortion reduction architecture can be of
relatively simple construction. Since the amplifier distortion
reduction system using distortion feedback does not depend on
characteristics of the signal to be amplified, the system is
flexible and can be applied to a variety of wireless
communications. Accordingly, it can be applied to multi-channel
wideband systems without the need for high speed digital processors
although this system can be used along with other distortion
reduction systems.
[0024] FIG. 1 shows a general block diagram of a distortion
reduction system 80 in which distortion feedback is injected back
to the input of the amplifier to reduce distortion produced from
the amplifier. In the amplifier architecture 80, an input signal 82
is received by the distortion reduction system 80 on a main signal
path 84. The input signal 82 is to be amplified by an amplifier 86
on the main signal path 84. After amplifying the input signal 82,
the amplifier 86 produces an amplified output signal 88 with
distortion 90. A sample 92 of the amplifier output is obtained, for
example by having a coupler 94 provide a sample 92 of the amplified
output signal 88 onto a feedback path 96.
[0025] Distortion feedback circuitry 98 receives the sample
amplifier output 92 on the feedback path 96 and produces a
distortion signal using the sample amplifier output 92. The
resulting distortion signal is placed on the main signal path 84,
for example by a combiner 104, with the input signal to be
amplified. After the amplifier 86 amplifies the input signal, the
distortion generated by the amplifier 86 in amplifying the input
signal is canceled by the amplified distortion signal, thereby
reducing the distortion generated by the amplifier 86. The total
distortion at the output of the amplifier 86 is reduced because the
feedback distortion signal is produced with the proper phase and
amplitude to destructively combine with the distortion generated by
the amplifier 86 to the input signal. Ideally, the feedback
distortion signal is produced such that the amplified feedback
distortion signal will have the same amplitude but a 180 degree
phase shift to the distortion signal generated by the amplifier 86
to the input signal 82.
[0026] Depending on the embodiment, the distortion feedback
circuitry 98 can be implemented in different ways. In FIG. 1, a
sample of the input signal is placed onto a carrier cancellation
path 122, for example by coupler 124 on the main signal path 84. To
produce the distortion signal, the distortion spectrum of the
amplifier output is isolated on the feedback path 96. In this
embodiment, a combiner 126 combines the sample amplifier output 92
(with non-distortion and distortion spectrums) on the feedback path
96 with an input signal sample 128 on the carrier cancellation path
122. The input signal sample 128 on the carrier cancellation path
122 is the inverse of the non-distortion spectrum of the amplifier
output 92 on the feedback path. As such, the input signal sample
128 on the carrier cancellation path 122 combines with the
non-distortion spectrum of the amplifier output 92 on the feedback
path 96 to cancel the non-distortion spectrum on the feedback path
96, leaving the distortion spectrum on the feedback path 96.
[0027] In certain embodiments, a power divider 130 at the output of
the combiner 126 on the feedback path 96 provides a signal
representative of the signal remaining on the feedback path 96
after the combiner 126. A variable phase and/or amplitude adjuster
132 responsive to the signal representative of the remaining
carrier signal on the feedback path 96 can adjust the relative
phase and/or amplitude between the input signal sample 128 on the
carrier cancellation path 122 and the sample amplified output
signal to improve cancellation of the non-distortion spectrum on
the feedback path 96. Depending on the embodiment, a phase and/or
amplitude adjuster can be located on different paths to provide a
desired amplitude and/or phase relationship between combining
signals on the different paths to get the desired cancellation.
[0028] A distortion signal 102 is produced from the distortion
spectrum left on the feedback path 96. The distortion signal 102 is
coupled onto the main signal path 84 at the input side of the
amplifier 86. The distortion signal 102 has a phase and amplitude
such that the distortion signal 102 on the main signal path
combines to cancel or at least reduce the distortion generated from
amplifying the input signal to produce the amplified output signal
with reduced distortion at the output of the amplifier 86. A
variable phase and/or amplitude adjuster 136 on the feedback path
96 is responsive to the residual distortion spectrum in the
amplified output signal on the main signal path 84 to adjust the
phase and/or amplitude of the distortion signal on the feedback
path 96 to improve cancellation of the distortion generated by the
amplifier 86. In this embodiment, a power divider 138 obtains a
representation or sample of the sample amplified output signal on
the feedback path 96.
[0029] In a particular implementation of the feedback amplifier
architecture of FIG. 1, the power amplifier stage 86 is an
amplifier module normally with 40 dB (or larger) gain and is
operating with just a few dB (or 0 dB) back-off to its P.sub.1dB
point to get high efficiency (30%, typically). As such, there are
inter-modulation components or distortion 90 at its output if no
linearization is implemented, as shown on the figure for a two
single tone input signal 82. The output power (non distortion
spectrum 88 and distortion spectrum 90) of the amplifier 86 is
sampled by directional coupler 94 which has normally a 30 dB
coupling attenuation. The sample amplified output signal 92
contains the distortion spectrum and the non-distortion spectrum
(which is just a linear image of the transmitter (TX) input signal
82).
[0030] The sample amplified output signal is fed to a variable
attenuator 142 where its amplitude is adjusted to be equal to the
amplitude of the input signal sample on the carrier cancellation
path 122. After the variable attenuator 142, the sample amplified
output signal is fed to the combiner 126 where it is combined with
the phase shifted (by phase shifter 127) input signal sample on the
carrier cancellation path 122. After the combiner 126, the
non-distortion spectrum is cancelled, leaving the distortion
spectrum on the feedback path 96. The distortion spectrum is fed to
the following variable amplifier stage 146 where it is so amplified
that the total loop attenuation from the directional coupler 94 at
the output of the main amplifier 86 to the combiner 104 will be
equal to the total gain of the main amplifier 86. The signal out
from this variable amplifier 146 is fed to a phase shifter 148
where its phase is shifted by 180.degree.-Q, where Q is a constant
and is the total phase shift generated by all the components in the
feedback loop (they should all operate in linear region), including
the phase shift (which is a constant, as will be explained)
generated by the amplifier to the feedback distortion signal. The
distortion signal out from the phase shifter 148 is fed to an
isolator 150 and then is fed back to the input side of the main
amplifier 86. The feedback distortion signal has the proper
amplitude and phase, after being amplified by the main amplifier
86, to cancel with the distortion generated by the main amplifier
86 to the input signal from the transmitter (TX).
[0031] In this case, an assumption is implied that the distortion
generated by the amplifier stage 86 to the feedback distortion
signal can be neglected. This is normally true because the feedback
distortion signal is normally 20 dB lower than the input signal
from the transmitter if the power amplifier stage 86 is operating
at its P.sub.1dB level or lower output level. The reason is that,
normally, the IP.sub.3 (the output third intercept point) is 10 dB
higher than P.sub.1dB for RF power amplifiers. So, if the output
power is at the P.sub.1dB level, the distortion (the maximum
distortion component is the third order inter-modulation) is 20 dB
or more lower than P.sub.1dB of the output power of the amplifier.
Thus, at its input, the feedback signal is 20 dB lower than the
transmitter signal. In this case, the main amplifier 86 operates in
its linear region to the feedback distortion signal and the phase
shift generated by the main amplifier 86 to the feedback signal is
a constant.
[0032] Note, the power within the feedback loop is about 20 dB
lower than the power from the transmitter and can be at least 60 dB
lower than the output power of the power amplifier. So, the
feedback circuit consumes only little power and will not affect the
total efficiency of the system.
[0033] For wide band applications (the power amplifier amplifies a
wide band RF signal), the phase shifters used in this invention
have to have a wide frequency band, namely, its phase shift under
fixed control signal level has to keep constant or varies less than
1 degree over a wide frequency range (typically, over 50 MHz for 10
MHz RF channel band-width). Such wide band phase shifters are
available by using vector modulators.
[0034] FIG. 2 shows the block diagram for the two control loops,
control loop-1 (154) and control loop-2 (156) for use with an
embodiment of the amplifier distortion reduction system of FIG. 1.
The control loop-1 (154) maintains the non-distortion spectrum
cancellation. The signal out from the combiner 126 is branched into
the control loop-1 (154) using power divider 158 and down converted
to baseband signal by a mixer 160 in loop-1 (154). A local
oscillator (LO) 162 produces an LO frequency signal which is the
same as the carrier frequency of the transmitter carrier. Then the
baseband signal 163 is fed to a low pass filter (LPF) 164 which has
a sharp cut-off edge at 0.5*Bw, where Bw is the RF channel
band-width. This LPF 164 will also reject the LO leakage from the
mixer 160 and LO harmonics leakage. The signal out from the LPF 164
is the residual non-distortion spectrum (but frequency shifted to
baseband) 165, and the distortion spectrum is rejected. The
residual non-distortion spectrum power is further converted to DC
voltage by an RF to DC converter 166 and fed to an VDA &
integrator (VDA: voltage differential amplifier) 168. The output
polarity switch point of the integrator is controlled by a
reference signal. The output of the integrator is used to control
the phase shifter 127 (FIG. 1) and the variable attenuator 142
(FIG. 1) to maintain the cancellation of the non-distortion
spectrum at the output of the combiner 126.
[0035] The control loop-2 (156) maintains cancellation of the
distortion spectrum. The sample amplified output on the feedback
path 96 is branched to this loop by coupler or power divider 138
(FIG. 1) and attenuated (as shown by attenuator 170). The branched
signal is down-converted to a baseband signal 171 by a mixer 172.
This baseband signal is fed to a LPF 174 and a high pass filter
(HPF) 176. The LPF 174 is a low order filter and is for the
rejection of the LO leakage. This LPF 174 has its cut-off frequency
a distance from the distortion spectrum (for example, a cut-off
frequency at 0.5*Fc, where Fc is the carrier frequency). The HPF
176 is a high order filter with a sharp cut-off edge at 0.5*Bw,
where Bw is the RF channel band-width. After the HPF 176, only the
frequency shifted, residual distortion spectrum 178 is passed.
Then, the power of the residual distortion spectrum is converted to
DC voltage by RF to DC converter 80 and is fed to a VDA &
integrator 182. The output of the integrator is used to control the
phase shifter 148 (FIG. 1) and the variable amplifier 146 (FIG. 1)
to maintain the cancellation of the distortion spectrum.
[0036] Note, in this embodiment, the pick-up of the residual
non-distortion spectrum and the pick-up of the residual distortion
spectrum is done in baseband frequency range. This has the
advantage that the implementation of the high order LPF (164) in
loop 1 and the HPF (176) in loop 2 is more convenient and more
accurate in baseband than in IF band and even better than in the RF
band. However, other embodiments are possible.
[0037] A simulation was carried out for the feedback amplifier of
FIG. 1. FIG. 3 shows the amplifier output spectrum for the case
without RF distortion feedback cancellation, and FIG. 4 shows the
amplifier output for the circuit of FIG. 1 with RF distortion
feedback cancellation. In this example, the input signal to the
amplifier are two single tone signals, which are the two central
signals (183) shown in FIGS. 3 and 4 along with the 3.sup.rd order
(184) and 5.sup.th order (185) inter-modulation components. As can
be seen in FIGS. 3 and 4, when the RF distortion feedback
cancellation is applied, both the 3.sup.rd (down by 30 dB) and the
5.sup.th order (down by 70 dB) inter-modulation components are
considerably suppressed.
[0038] FIG. 5 shows an embodiment 190 of the distortion reduction
system using distortion feedback for a multi-carrier amplifier (or
an amplifier which amplifies several wide-band RF channels at the
same time). In this example, multiple carriers from a plurality of
transmitters 192a-c are combined by a combiner 194 to be amplified
by the amplifier 86. No matter how many channels are operating, the
non-distortion spectrum can be obtained from the output of the
combiner 194, while the mixed spectrum is obtained from the output
of the power amplifier 86. So, the basic method for linearization
of one carrier amplifier is applicable to multi-carrier cases. For
multi-carrier cases, two null loops (control loop 196 for
maintaining cancellation of the non-distortion spectrum and control
loop 198 for maintaining cancellation of the distortion spectrum)
are implemented to avoid inter-channel interference.
[0039] FIG. 6a shows the flow diagram of the control loop 196, and
FIG. 6b shows the flow diagram for the control loop 198. Only one
channel is monitored by control loop 196, and in this embodiment,
that channel is the channel with the lowest carrier frequency. In
control loop 196, one branch output of the power divider 130 (FIG.
5) is frequency converted to an intermediate frequency (IF) using
mixer 200 and a LO generator 202. The resulting frequency converted
signal 203 is provided to a band pass filter 204 to pick up the
residual non-distortion spectrum of the considered channel. In this
embodiment, the cut-off edges of the band pass filter 204 are set
at F.sub.IF-0.5*Bw and F.sub.IF, where Bw is the RF channel
bandwidth and F.sub.IF is the IF frequency. As such, the
non-distortion spectrum at the lower frequency side of the lowest
frequency carrier is picked up. The frequency shifted, residual
non-distortion spectrum 205 is further converted to DC voltage by
an RF to DC converter 206 and fed to an VDA & integrator (VDA:
voltage differential amplifier) 208. The output polarity switch
point of the integrator is controlled by a reference signal. The
output of the integrator is used to control the phase shifter 127
(FIG. 5) and the variable attenuator 142 (FIG. 5) to maintain the
cancellation of the non-distortion spectrum at the output of the
combiner 126 (FIG. 5).
[0040] Control loop 198 monitors only one channel for maintaining
cancellation of distortion spectrum, and in this embodiment, the
channel is the one with the highest carrier frequency. The sample
signal from the output of the main amplifier 86 is branched to this
loop by a power divider 138 (FIG. 5) and attenuated (as shown by
attenuator 210). The sample is down converted to a baseband signal
212 by mixer 214 and LO generator 216, which in this embodiment is
set to be equal to the highest carrier frequency. As such, the
distortion spectrum at the high frequency side of the highest
frequency carrier will be picked up in the loop 198 for monitoring
purposes. This baseband signal is fed to a LPF 218 and a high pass
filter (HPF) 220. The LPF 218 is a low order filter and is for
rejection of the LO leakage. This LPF 218 has its cut-off frequency
a distance from the distortion spectrum (for example, at 0.5*Fc,
where Fc is the carrier frequency). The HPF 220 is a high order
filter with a sharp cut-off edge at 0.5*Bw, where Bw is the RF
channel band-width. After the HPF 220, only the frequency shifted,
residual distortion spectrum is passed. Then, the power of the
residual distortion spectrum is converted to DC voltage by RF to DC
converter 222 and is fed to a VDA & integrator 224. The output
of the integrator is used to control the phase shifter 148 (FIG. 5)
and the variable amplifier 146 (FIG. 5) to maintain the
cancellation of the distortion spectrum.
[0041] Note, in control loop 196, the LO frequency is set to
F.sub.CL-F.sub.IF, where F.sub.CL is the lowest carrier frequency
and F.sub.IF is in the low IF band, for example 70 MHz. The output
power of the main amplifier 86 is limited so that the distortion
spectrum of each channel will not spread for more than one RF
channel bandwidth. Under the above conditions, the control loop 198
will pick up the distortion spectrum at the high frequency side of
the carrier for the channel with the highest carrier frequency, and
control loop 196 will pick up the non-distortion spectrum at the
low frequency side of the carrier for the channel with the lowest
carrier frequency. The inter-channel interference will not be
picked up by either control loop. Therefore, a correct adaptation
is maintained. In case there is a dominant channel in the
multi-channel case this channel should be selected by the two
control loops. Alternative embodiments are possible.
[0042] FIG. 7 shows a block diagram of another embodiment of the
feedback amplifier system using a different implementation for the
distortion feedback circuitry 98 which produces a distortion signal
on the feedback path 96 and feeds back the distortion signal to the
input side of the amplifier 86 to reduce the distortion produced at
the output of the amplifier 86. In this embodiment, the distortion
signal is obtained from the amplifier output by sampling the
amplifier output and filtering out the non-distortion spectrum from
the sample amplified output signal to leave the distortion spectrum
on the feedback path 96. In the particular implementation of FIG.
7, where like reference numerals correspond to analogous
components, the amplifier distortion reduction system 230 picks up
the distortion spectrum from the output side of the amplifier 86.
In this embodiment, the coupler 94 places a representation or
sample of the amplified output signal onto the feedback path 96. In
this embodiment, the sample on the feedback path is frequency
down-converted into an IF signal by down-converter 232. The
non-distortion spectrum is filtered out by a filter block 234, such
as a filter-equalizer combination in the IF band. The frequency
shifted distortion spectrum 235 is then up-converted to RF band
again by up-converter 236 and fed into the input side of the
amplifier by the combiner 104. An amplitude and/or phase shifter
238 adjusts the amplitude and/or phase of the distortion signal
such that the distortion signal is inverse in phase and equal in
amplitude to the distortion to be generated by the amplifier 86
when amplifying the input signal. The advantage of the down-up
conversion is that the filter will perform better in the IF band,
and the implementation of the filters and/or equalizers in the IF
band is easier than in the RF band. Embodiments are possible where
the distortion spectrum remains at RF on the feedback path 96.
[0043] In a particular implementation of the feedback amplifier
architecture of FIG. 7, the power amplifier module normally with 40
dB (or larger) gain and is operating with just a few dB (or 0 dB)
back-off to its P.sub.1dB point to get high efficiency (30%,
typically). As such, there is inter-modulation distortion generated
at the amplifier output. If no linearization is implemented,
spectral regrowth is seen at the output for a raised cosine
filtered spread spectrum input. The output power is sampled by the
directional coupler 94 which has normally a 30 dB coupling
attenuation. The sampled RF signal is further attenuated by about
40 dB or more by attenuator 252 to keep the following mixer working
in such a state that its high harmonic terms are very low. The
sampled RF signal is converted to an IF signal 254 centered at FIF
by the down-conversion mixer 232, which is just a balanced mixer
and not necessary being a demodulator, and the LO generator 256.
The IF signal is fed to a low pass filter 234. This low pass filter
234 has a sharp cut-off edge which is at F.sub.IF-0.5* Bw where Bw
is the RF channel bandwidth. The passed spectrum out from the LPF
234 is the distortion spectrum at the low frequency side of the
carrier. The local oscillator signal leakage is also rejected by
the LPF 234. The distortion spectrum out from the LPF 234 is fed to
an equalizer 258.
[0044] Because the LPF 234 has a sharp cut-off edge, its phase
shift versus frequency is not linear, especially near the cut-off
frequency. The purpose of the equalizer 258 is to linearize the
non-linear phase shift versus frequency characteristics generated
by the LPF 234 so that the phase shift versus frequency
characteristics of the distortion spectrum after the equalizer 258
is linearly calibrated (in other words, the delay is a constant).
This phase shift linearization is used in this embodiment,
otherwise the picked-up distortion spectrum may not be made phase
reversed uniformly compared to the distortion spectrum at the
output of the amplifier. Note, the equalizer is in the IF frequency
range (typically 70 MHz) which is easier to implement.
[0045] The distortion spectrum 235 out from the equalizer 258 is
up-converted to RF band at the up-converter 236. The full
distortion spectrum at both sides of the carrier is recovered.
Note, the input signal level to the mixer 236 is still very low so
the high harmonic modulation components generated by the
up-converter 236 can be neglected.
[0046] The distortion spectrum centered at the carrier frequency is
fed to a RF band pass filter 260. The purpose of the band pass
filter (BPF) is to reject the local oscillator signal leakage from
the up converter 236 and its harmonics leakage. The cut-off edges
of the BPF 260 can be designed a distance from the frequency range
of the distortion spectrum. As such, another equalizer need not be
inserted after the BPF to linearize the group delay generated by
the BPF at its cut-off edges. The RF distortion spectrum 262 out
from the BPF 260 is amplified by amplifier 264 to a level so that
the total loop attenuation from the directional coupler 94 to the
isolator 266 is equal to the linear gain of the power amplifier
stage 86. The amplified distortion spectrum is phase shifted at a
phase shifter 268 by 180.degree.0-Q, where Q is the total phase
shift for the feedback signal to travel through the whole feedback
loop, except the phase shifter 268 itself, from the directional
coupler 94 through the down-up conversion loop to the output of the
isolator 266. Q includes the phase shift generated by the main
power amplifier 86 to the feedback signal (which is a constant as
will be explained below). Because the phase shift versus frequency
character of the distortion spectrum is already linear at the input
of the phase shifter 268, such a constant phase shift over the
whole distortion spectrum is possible.
[0047] The phase shifted distortion spectrum is fed to the isolator
266, and then is fed back to the input side of the power amplifier
stage 86 through the power combiner 104. The feedback distortion
spectrum 270 with the appropriate amplitude and phase, after being
amplified by the power amplifier stage 86, will cancel the
distortion generated by the amplifier 86 to the input signal from
the transmitter. In this embodiment, the distortion generated by
the amplifier stage 86 to the distortion signal can be neglected
because the main amplifier 86 operates in its linear region to the
distortion signal, and the phase shift generated by the main
amplifier 86 to the distortion signal is a constant.
[0048] Note, in this example, the maximum power within the feedback
loop is about 20 dB lower than the power from the transmitter and
at least 60 dB lower than the output power of the power amplifier
86. So, the linearization circuit consumes little power and will
not affect the total efficiency of the system. For wide band
applications (the power amplifier operates with a wide band RF
input spectrum), the phase shifter used in this example has a wide
frequency band in that its phase shift under a fixed control signal
level has to keep constant or varies less than 1 degree over a wide
frequency range (typically, over 50 MHz for 10 MHz RF channel
band-width). Such wide band phase shifters are available by using
vector modulators.
[0049] FIG. 8 shows an embodiment of the amplifier distortion
reduction system in which multiple RF channels are amplified at the
same time. In this embodiment, where like reference numerals
indicate analogous components, two RF channels are to be amplified
where each RF channel has a carrier frequency modulated by a
separate data series. These two RF channels may be neighboring
channels or not. In FIG. 8, two RF signals from two transmitters, X
(302) and Y (304), are combined together by a power combiner 306
and fed to the power amplifier 86. Each transmitter 302 and 304 has
its own carrier and data input. For convenience of description,
carrier frequency Fcx of transmitter X (302) is assumed to be lower
than the carrier frequency Fcy of transmitter Y (304). In this
embodiment, there are two linearization feedback loops, block X
(308) and block Y (310). Block X (308), which is for the low
frequency carrier channel, is analogous to what is described for
FIG. 7.
[0050] Block Y (310), which is for the high frequency carrier
channel, has a LPF 312 and a HPF 314, where the LPF 312 is to
reject the LO generator 316 leakage and a sharp cut-off edge design
is not necessary for it. The HPF 314 is to pick up the distortion
spectrum at the high frequency side of the high frequency carrier
and a sharp cut-off frequency at FIF +0.5*Bw is used. The two
feedback loops have same IF frequency (for example, 70 MHz), and
the RF LO frequency Lx and Ly are selected based on the input
channel carrier frequency.
[0051] The coupler 94 provides the full spectrum sample from the
amplifier 86 to a power divider 328 which provides the full
amplifier output spectrum sample to each loop corresponding to a
carrier. In this embodiment, the two feedback loops 308 and 310
receive the full amplified spectrum. The feedback loop 308 will
only pick up the distortion spectrum at the low frequency side of
the carrier from the low frequency channel. Then, the full
distortion spectrum originating from the transmitter X (302) will
be recovered and phase shifted and fed back to the amplifier stage
86 for cancellation. The feedback loop block Y (310) will only pick
up the distortion spectrum at the high frequency side of the
carrier from the high frequency carrier channel. Then, the full
distortion spectrum originating from the transmitter Y (304) will
be recovered and phase shifted and fed back to the amplifier stage
86 for cancellation. In this way, there will not be inteference
between the two loops 308 and 310 if the distortion spectrum
regrowth does not spread more than one RF channel bandwidth. As
such, for the two carrier example, the saturation of the amplifier
should be limited such that the distortion spectrum regrowth will
not spread more than the RF channel bandwidth.
[0052] The distortion feedback cancellation scheme where the
distortion signal is obtained by filtering the non-distortion
spectrum can also be implemented by down to baseband conversion of
a the sample of the amplifier output to obtain the distortion
signal. In FIG. 9, where like reference numerals indicate analogous
components, an amplifier distortion reduction system 330 is
described for one carrier by using down to baseband conversion. A
sample of the amplifier output spectrum attenuated by an attenuator
332 is down-converted to baseband using a mixer 334 and a local
oscillator generator 336 (RF-LO-X). The baseband spectrum is fed to
a filter combination. The filter combination now is a low pass
filter 338 in series with a high pass filter 340. Note, the
baseband high pass filter 340 has a sharp cut-off edge at 0.5*Bw,
where Bw is the RF channel band width. The purpose of this filter
340 is to pick up the distortion spectrum at the high frequency
side of the carrier. The baseband low pass filter 338 is just for
rejecting the LO leakage and does not need to be of a sharp cut-off
edge design, and therefore, can be a low order design. Note, the
full distortion spectrum at both sides of the non-distortion
spectrum is recovered after the up-converter 234. Then, the
recovered distortion spectrum is phase shifted and amplified to
produce the distortion signal.
[0053] There is an additional branch in the circuit, which is made
of attenuater B (342), phase shifter A (344) and power combiner
346, to cancel possible LO leakage from the up-converter 234. The
LO leakage at its harmonic frequencies is rejected by the RF low
pass filter 260, which is again a low order design and its cut-off
edge is a distance from the distortion spectrum. This scheme is a
little more complicated than the scheme described in FIG. 7 but has
the advantage that the implementation of the high order HPF 340 in
baseband is more convenient than in the IF band.
[0054] In addition to the embodiments described above, alternative
configurations or implementations of the amplifier distortion
reduction system according to the principles of the present
invention are possible which omit and/or add components and/or use
variations or portions of the described system. The amplifier
distortion reduction system has been described as amplifying single
or multiple channels. The number and nature of the channels may
vary, such as TDMA (30 KHz channel bandwidth), Global System for
Mobile Communications (GSM)(200 KHz channel bandwidth), CDMA (1.28
MHz channel bandwidth), wideband CDMA (5 MHz channel bandwidth),
and Universal Mobile Telecommunication Systems (UMTS)(3.84 MHz
channel bandwidth), depending on the application as does the
particular implementation or configuration of the amplifier
distortion reduction system using distortion feedback. Depending on
the implementation, the relative gain and/or phase circuitry,
equalizer circuitry and/or filter arrangement(s) can be positioned
in different locations and/or paths within the amplifier distortion
reduction system or implemented in different configurations or
arrangements. Additionally, particular implementations of the
feedback distortion circuitry are described, but alternative
configurations are possible.
[0055] Furthermore, the amplifier distortion reduction system has
been described using a particular operation without digital
conversion, but it should be understood that the amplifier
distortion reduction system and portions thereof can be implemented
in application specific integrated circuits, software-driven
processing circuitry, firmware, programmable logic devices,
hardware or other arrangements of discrete components and/or in the
digital domain as would be understood by one of ordinary skill in
the art with the benefit of this disclosure. What has been
described is merely illustrative of the application of the
principles of the present invention. Those skilled in the art will
readily recognize that these and various other modifications,
arrangements and methods can be made to the present invention
without strictly following the exemplary applications illustrated
and described herein and without departing from the spirit and
scope of the present invention.
Sequence CWU 1
1
8 1 2023 DNA Marek's disease virus 1 acggtctctg aacaagacgg
gcgataatat tagccatgtt tcgcatagcc gtacctcccg 60 ttctctcctg
attatttgaa aatgataaag tagccgtttt attacaagct atatgattcc 120
tcaaatccgt tacgttagca gacgcctttc cactgcgtcg ttgtatatgt atcgtgtttg
180 tattatgacg ttttaaaatt ttatgagtgt cagttatccg tgctttatag
tcagacgcgg 240 tcgccaatat agagcatagt ctatgaaaat cagtcactat
gtgccttttc tttaggcaca 300 tcacatgtag aacagacagt tttcgtcttg
ctacaaatac taacattgga caaataacga 360 tacaatctga tccttgaggc
gcaatttgcc caatcagaga tttggaatcc aataactgct 420 ttatgccggt
gagtctttgt tcatgtttac tgcgtgtctt caggttacga gaaaatttgc 480
aagtttttag ttctagaatg acgcatactc catcacagcc tacttcccac aaatcacgag
540 gcaacttaaa catgcaaata caatccggtc tacgtcgttc taggtttact
tcgaagacca 600 atcgaaaatc cgtcaactgt ttaaatacat ctaataccat
gaccttccca aaaattttgg 660 caaagcttct ccccggccaa tcatacacct
gagatcctag acacatcgct tctgcataaa 720 gccgtttgta aaagcgatcg
tgacatcgaa caccagccgc taaacgtcgc tttctaagga 780 cattcgtatt
tacatgccgt ttgaaatttc gagtgctact aacctgtctg cgatatcttt 840
tgagtacgtt cttctctccc attgaacatg tcggagccac aatcgtggtc ggtaatggca
900 tctcagatga catctgcaca gctcatacgt gtatacctcg atggatcaat
gggtataggt 960 aaaacgtcaa tgttgaatga gataccgacg cactctttaa
tgggagtacc cgtactaaag 1020 gttttcgaac ctatgaaata ctggcggtat
tattttactg atttggtcac gaccgtaaat 1080 gatacatgtg atcgtcgtcg
caggggagag ttttctttat ttcaatctag catgattgta 1140 acagctttac
aatcaaagtt tgcagatccc tatcttgtat ttcatgagcg cttatcgtcg 1200
aagtgtcatc gcataacagg aacacgtggc aatccatcgc ttatattaat tctagatcga
1260 catcccatat ccgctaccgt atgttttccc attgctcgac atttaactgg
agattgttcc 1320 ttggagatgc taattagtat gataataagg ttgccccagg
aaccgccagg atgcaacttg 1380 gtgattgtcg atctacatga cgaaaaggag
catgttagcc gtctatcttc acggaatagg 1440 accggcgaga aaacagatct
actaatgctc agggcactta atgcagtgta ttcctgttta 1500 gtagacacta
ttatgtacgc aaatcatatt tgtccctaca gtaaggatga atgggaatct 1560
gaatggttgg atctaccatg gtttgataca tctttggcca caacgtttat aaacgaacct
1620 cgtactgatt atcgcggtag tagggtgtca ttacaccata cgcttttagc
gatatttaag 1680 cggcgagaat tatgtgccga agatggtagc ttatcaacaa
cgcatgcatg gatattgtgg 1740 ggattattaa tgaaactgcg gaacattaac
gtcgaacgat ttaatattac tggcctgtcc 1800 acaacaaagt gtgtagaatc
gttcatggat actatgtcgg agagattggt aacacatagt 1860 agctggaatg
atgccttcga gattgaagct gatgtactag cctataataa agagatggct 1920
atgtaaaact acccattcat atcgcgcttc tataattagc ttgcccacat cacaatgatg
1980 cggcaatatt gacttatatt aagatagtaa tttggcgtcc tta 2023 2 2236
DNA Marek's disease virus 2 taaataaaga actttgggaa taacaagcta
tgtatagaat ttatttcgcg tgaagatttt 60 tcccaagtcc gatcacattt
caggtattac agcggtaata gatccatgca ttatgagggt 120 ttgacgtatt
atctcgatta agaacatatt gtaatacacc cactgtttct caaacgagtg 180
tctatcaatg atataataca ttgatgtatc gactataata cccccaatgt tcaaaggctg
240 ataaaactga tatatctatg gccgcgcata gcaattctgc cgtatcttct
cccaccgatt 300 ctcgtaacgc gacgtctatg ggatcaatgt ctttatatag
accgtctaga ataagagcca 360 gtttacgtat cttgaggtcc tgtatagatt
ttggtgcaga tgtttctgcc acatccaata 420 aagtagtctc gtctgcaaag
gctgatggac taagaactcc atgttgctct tccaatgaag 480 aagtccagtt
cacaactaat ttcagtaacc atgccaagaa ataaaatcct ctgaataaac 540
tgtttgtttc tgcaagacaa gtcggcatgg agtaggcatt ccccctcaat ggtagaggta
600 tgatgatcgc acaactcgcg aattaagtca taacaatttg gcagacgatt
taatatatgt 660 atatactgaa gcaacaaaaa cttctgactg ggcgatatat
ttttgttttc tggtccaact 720 ccaacaaaca tggatgcgtg tcttccaaat
aaagcatttg aaatcatccc caactcactt 780 tgtataattt ccaggtcgga
ttgagatcca ttctccgtat aagattagaa tttaaattga 840 gcatgttcat
attaaaaacc gagtctactt tccagaagat ttcccataac ttatttagag 900
aagtagaggg tatacaagag ctggtatcgc aactccatat cttaaatacg ggtggtatat
960 atttgatttg taccaaagaa ctgaaccgag catgtttttt ccgttgtact
ggatttgttt 1020 gttactgttc acgttcaatt taccccggct ccagccgtca
tatcccatgc gcgttgcaca 1080 gtcgtcgtgt ttgcagcttt ctttgctgta
actataacat cgactcgcct gccgaatatc 1140 tctgatgata atgcttctct
aggagtggga atgccatcaa ataatccttc aacgaggtca 1200 ctcaaagact
taggtaattc agtcaatctt gcacaagtta gcacaaatgc atcacgactg 1260
cactcatata ctaaatctga atatatgtcc gtgattatag ggaattcggg tatatgaatt
1320 gtacgatcat gtggaaaatc gtatgcggcc tgtatcgtta acccagaaat
tgcatttgtc 1380 ggtaccatat actttgctat atccggatca tacgtttcca
gacagagaag cccacaaagc 1440 tcacgttcac tgcatatacc atcacgactt
aacacagcta tactatcgat gaacaattca 1500 tcttcatcgg aagaaaaagc
ccacttcata cctctgcgaa gtaattctcg gcgaacatga 1560 gctgccaatg
gtttggactg accaccacgt agaaccaacc caatttttgc gagctctggt 1620
aataccatca tctatacagc ctgcctacag caaaaaacaa ccgccgcaaa aaaatacctt
1680 tatatcccat tccgatacat aaaactggac attctataac gaaaacatgt
ccgtatttaa 1740 tatccattga ctgtcctctc tggacgtaac ctatatcact
gtagcgcaaa tccaatcctt 1800 gataacagca ttgcgttaat cactgggtgc
acggattaac gtgtacgtat ttactgtcgc 1860 gtcatatgaa cgacaatgag
cttgggtatg cagctcgtca ttgaacgcca tttgtggcaa 1920 agcaataagg
gtctcagacc atcacattat tcgacgaatt gtactacata ggccacccct 1980
tgtttaacta tgtcaagcat ggatttggat actatgtcaa cagaagctaa tgaatatacc
2040 atccccctca tgaattgatg atggacgatc ggatacatgc gaaaactctt
gggtcgtatt 2100 gaccactatc tgaggaatta gattgggatg atattatgca
ctttctctta tttaggcgat 2160 atattttaca atccaacagc tatgacatac
atcctcaaat cacccgtatg tttactcttt 2220 ggctatctac tttgtc 2236 3 31
DNA Artificial Sequence Description of Artificial Sequence Primer 3
gggggtacca agugcattgg atggctacat a 31 4 33 DNA Artificial Sequence
Description of Artificial Sequence Primer 4 ggggctagct taaagatcgt
cgtacaggct caa 33 5 18 DNA Artificial Sequence Description of
Artificial Sequence Primer 5 aagatttttc ccaagtcc 18 6 18 DNA
Artificial Sequence Description of Artificial Sequence Primer 6
tcgtcgaata atgtgatc 18 7 59 DNA Artificial Sequence Description of
Artificial Sequence pCR 3.1 vector 7 taatacgact cactataggg
agacccaagc tggctagcgt ttaaacttaa gcttggtac 59 8 54 DNA Artificial
Sequence Description of Artificial Sequence pCR 3.1 vector 8
tcgagtctag agggcccgtt taaacccgct gatcagcctc gactgtgcct tcta 54
* * * * *