U.S. patent application number 09/775636 was filed with the patent office on 2002-04-18 for soft-switched full-bridge converters.
Invention is credited to Jang, Yungtaek, Jovanovic, Milan M..
Application Number | 20020044461 09/775636 |
Document ID | / |
Family ID | 24618527 |
Filed Date | 2002-04-18 |
United States Patent
Application |
20020044461 |
Kind Code |
A1 |
Jang, Yungtaek ; et
al. |
April 18, 2002 |
Soft-switched full-bridge converters
Abstract
A family of soft-switched, full-bridge pulse-width-modulated (FB
PWM) converters provides zero-voltage-switching (ZVS) conditions
for the turn-on of the bridge switches over a wide range of input
voltage and output load. The FB PWM converters of this family
achieve ZVS with the minimum duty cycle loss and circulating
current, which optimizes the conversion efficiency. The ZVS of the
primary switches is achieved by employing two magnetic components
whose volt-second products change in the opposite directions with a
change in phase shift between the two bridge legs. One magnetic
component always operates as a transformer, where the other
magnetic component can either be a coupled inductor, or uncoupled
(single-winding) inductor. The transformer is used to provide
isolated output(s), whereas the inductor is used to store the
energy for ZVS.
Inventors: |
Jang, Yungtaek; (Apex,
NC) ; Jovanovic, Milan M.; (Cary, NC) |
Correspondence
Address: |
VENABLE
Post Office Box 34385
Washington
DC
20043-9998
US
|
Family ID: |
24618527 |
Appl. No.: |
09/775636 |
Filed: |
February 5, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
09775636 |
Feb 5, 2001 |
|
|
|
09652869 |
Aug 31, 2000 |
|
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Current U.S.
Class: |
363/17 |
Current CPC
Class: |
H02M 1/0058 20210501;
Y02B 70/10 20130101; H02M 3/33569 20130101 |
Class at
Publication: |
363/17 |
International
Class: |
H02M 003/335 |
Claims
We claim:
1. A soft-switched, constant-frequency, full-bridge power converter
with phase-shift modulation comprising: an input power source; a
first and second bridge leg each comprising a pair of
serially-connected controllable switching devices adapted to
connect across said input power source, each of said controllable
switching devices comprising a switch, an antiparallel diode
coupled across said switch and a capacitor coupled across said
switch; a first and second magnetic device each having a plurality
of windings formed around a corresponding magnetic core; said first
and second magnetic device coupled to said first and second bridge
leg in an arrangement so that when corresponding switches in said
first and second bridge leg are open and closed in phase the
volt-second product of said windings of said first magnetic device
is maximal and the volt-second product of said windings of said
second magnetic device is minimal, and when corresponding switches
in said first and second bridge leg are open and closed in
antiphase the volt-second product of said windings of said first
magnetic device is minimal and the volt-second product of said
windings of said second magnetic device is maximal; a plurality of
capacitors coupled to said windings of said first and second
magnetic devices to prevent their saturation by providing a
volt-second balance of said windings; an output circuit for
coupling a load.
2. A power converter as in claim 1 wherein said first magnetic
device is arranged as a transformer having primary and secondary
windings, and wherein said second magnetic device is arranged as a
coupled inductor having two windings that are connected in series,
and wherein the magnetizing inductance of said coupled inductor is
selected so that the energy stored in said magnetizing inductance
is large enough to substantially discharge said output capacitance
of each of said switching devices that is about to be turned on so
that voltage across said each of said switching devices at the
moment of turn-on is substantially reduced in the entire current
range of said load.
3. A power converter as in claim 2 wherein said output circuit is
coupled to said secondary winding of said transformer.
4. A power converter as in claim 1 wherein said second magnetic
device is arranged as a transformer having primary and secondary
windings, and wherein said first magnetic device is arranged as an
inductor, and wherein the inductance of said inductor is selected
so that the energy stored in said inductor is large enough to
substantially discharge said output capacitance of said each of
said switching devices that is about to be turned on so that
voltage across said each of said switching devices that is about to
be turned on at the moment of turn-on is substantially reduced in
the entire current range of said load.
5. A power converter as in claim 4 wherein said output circuit is
coupled to said secondary winding of said transformer.
6. A power converter as in claim 1 further comprising a plurality
of resistors for precharging said plurality of capacitors
immediately after said power source is applied to said power
converter so that said plurality of capacitors provide required
voltages for maintaining volt-second products of said windings of
said first and second magnetic devices during a start-up
period.
7. A power converter as in claim 1 wherein said output circuit is
the full-wave rectifier.
8. A power converter as in claim 1 wherein said output circuit is
the current doubler.
9. A power converter as in claim 1 wherein said output circuit
comprises a filter.
Description
CROSS REFERENCE TO RELATED APPLICATION
[0001] This is continuation-in-part of patent application Ser. No.
09/652,869, filed on Aug. 31, 2000.
BACKGROUND OF THE INVENTION
[0002] 1. Field of the Invention
[0003] This invention relates to isolated dc/dc converters, and
more particularly, to the constant-frequency, isolated dc/dc
full-bridge converters that operate with ZVS of the primary-side
switches in a wide range of input voltage and load current.
[0004] 2. Description of the Prior Art
[0005] The major factors hindering the operation of conventional
("hard-switched") pulse-width-modulated (PWM) converters at higher
switching frequencies are circuit parasitics such as semiconductor
junction capacitances, transformer leakage inductances, and
rectifier reverse recovery. Generally, these parasites introduce
additional switching losses and increase component stresses, and,
consequently, limit the maximum frequency of operation of
"hard-switched" converters. To operate converters at higher
switching frequencies and, eventually, achieve higher power
densities, it is necessary to eliminate, or at least reduce, the
detrimental effects of parasitics without a degradation of
conversion efficiency. The most effective approach in dealing with
parasitics is to incorporate them into the operation of the circuit
so that the presence of parasitics does not affect the operation
and performance of the circuit. Generally, this incorporation of
parasitics can be accomplished by two techniques: the resonant
techniques and constant-frequency PWM soft-switching
techniques.
[0006] The common feature of the resonant techniques is the
employment of a resonant tank that is used to shape the current and
voltage waveforms of the semiconductor switch (es) to create
conditions for either zero-current turn-off, or zero-voltage
turn-on. However, zero-current switching (ZCS), or zero-voltage
switching (ZVS) in resonant-type converters is achieved at the
expense of increased current and/or voltage stresses of
semiconductors compared to the stresses in the corresponding
"hard-switched" topologies. In addition, the majority of resonant
topologies need to circulate a significant amount of energy to
create ZCS or ZVS conditions, which increases conduction losses.
This strong trade-off between the switching-loss savings and
increased conduction losses may result in a lower efficiency and/or
larger size of a high-frequency resonant-type converter compared to
its PWM counterpart operating at a lower frequency. This is often
the case in applications with a wide input-voltage range. In
addition, variable frequency of operation is often perceived as a
disadvantage of resonant converters. As a result, although resonant
converters are used in a number of niche applications such as those
with pronounced parasitics, the resonant technique has never gain a
wide acceptance in the power-supply industry in high-frequency
high-power-density applications.
[0007] To overcome some of the deficiencies of the resonant
converters, primarily increased current stresses and conduction
losses, a number of techniques that enable constant-frequency PWM
converters to operate with ZVS, or ZCS have been proposed. In these
soft-switching PWM converters that posses the PWM-like square-type
current and voltage waveforms, lossless turn-off or turn-on of the
switch (es) is achieved without a significant increase of the
conduction losses. Due a relatively small amount of the circulating
energy required to achieve soft switching, which minimizes
conduction losses, these converters have potential of attaining
high efficiencies at high frequencies.
[0008] One of the most popular soft-switched PWM circuit is the
soft-switched, full-bridge (FB) PWM converter shown in FIG. 1(a),
which is discussed in the article "Design Considerations for
High-Voltage High-Power Full-Bridge Zero-Voltage-Switched PWM
Converter," by J. Sabate et al., published in IEEE Applied Power
Electronics Conf. (APEC) Proc., pp. 275-284, 1990. This converter
features ZVS of the primary switches at a constant switching
frequency with a reduced circulating energy. The control of the
output voltage at a constant frequency is achieved by the
phase-shift technique. In this technique the turn-on of a switch in
the Q.sub.3-Q.sub.4 leg of the bridge is delayed, i.e., phase
shifted, with respect to the turn-on instant of the corresponding
switch in the Q.sub.1-Q.sub.2 leg, as shown in FIG. 1(b). If there
is no phase-shift between the legs of the bridge, no voltage is
applied across the primary of the transformer and, consequently,
the output voltage is zero. On the other hand, if the phase shift
is 180.degree., the maximum volt-second product is applied across
the primary winding, which produces the maximum output voltage. In
the circuit in FIG. 1(a), the ZVS of the lagging-leg switches
Q.sub.3 and Q.sub.4 is achieved primarily by the energy stored in
output filter inductor L.sub.F. Since the inductance of L.sub.F is
relatively large, the energy stored in L.sub.F is sufficient to
discharge output parasitic capacitances C.sub.3 and C.sub.4 of
switches Q.sub.3 and Q.sub.4 in the lagging leg and to achieve ZVS
even at very light load currents. However, the discharge of the
parasitic capacitances C.sub.1 and C.sub.2 of leading-leg switches
Q.sub.1 and Q.sub.2 is done by the energy stored in leakage
inductance L.sub.LK of the transformer because during the switching
of Q.sub.1, or Q.sub.2 the transformer primary is shorted by the
simultaneous conduction of rectifiers D.sub.1 and D.sub.2 that
carry the output filter inductor current. Since leakage inductance
L.sub.LK is small, the energy stored in L.sub.LK is also small so
that ZVS of Q.sub.1 and Q.sub.2 is hard to achieve even at
relatively high output currents. The ZVS range of the leading-leg
switches can be extended to lower load currents by intentionally
increasing the leakage inductance of the transformer and/or by
adding a large external inductance in series with the primary of
the transformer. If properly sized, the external inductance can
store enough energy to achieve ZVS of the leading-leg switches even
at low currents. However, a large external inductance also stores
an extremely high energy at the full load, which produces a
relatively large circulating energy that adversely affects the
stress of the semiconductor components, as well as the conversion
efficiency.
[0009] In addition, a large inductance in series with the primary
of the transformer extends the time that is need for the primary
current to change direction from positive to negative, and vice
verse. This extended commutation time results in a loss of duty
cycle on the secondary of the transformer, which further decreases
the conversion efficiency. Namely, to provide full power at the
output, the secondary-side duty-cycle loss must be compensated by
reducing the turns ratio of the transformer. With a smaller
transformer's turns ratio, the reflected output current into the
primary is increased, which increases the primary-side conduction
losses. Moreover, since a smaller turns ratio of the transformer
increases the voltage stress on the secondary-side rectifiers, the
rectifiers with a higher voltage rating that typically have higher
conduction losses may be required.
[0010] Finally, it should be noted that one of the major
limitations of the circuit in FIG. 1(a) is a severe parasitic
ringing at the secondary of the transformer during the turn-off of
a rectifier. This ringing is cased by the resonance of the
rectifier's junction capacitance with the leakage inductance of the
transformer and the external inductance, if any. To control the
ringing, a heavy snubber circuit needs to be used on the secondary
side, which may significantly lower the conversion efficiency of
the circuit.
[0011] The ZVS range of the leading-leg switches in the FB ZVS-PWM
converter in FIG. 1(a) can be extended to lower load currents
without a significant increase of the circulating energy by using a
saturable external inductor instead of the linear inductor, as
described in the article "An Improved Full-Bridge
Zero-Voltage-Switched PWM Converter Using a Saturable Inductor," by
G. Hua et al., published in IEEE Power Electronics Specialists'
Conf Rec., pp. 189-194, 1991, and in U.S. Pat. No. 5,132,889,
"Resonant-Transition DC-to-DC Converter," by L. J. Hitchcock et.
al., issued on Jul. 21, 1992. However, even with the modifications,
the performance of these converters is far from optimal.
[0012] An FB ZVS-PWM converter that achieves ZVS of the primary
switches in the entire load and line range with virtually no loss
of secondary-side duty cycle and with minimum circulating energy
was described in patent application Ser. No. 09/652,869, filed Aug.
31, 2000 by Jang and Jovanovi and assigned to the assignee of this
application. This converter, shown in FIG. 2, employs a
primary-side coupled inductor to achieve a wide-range ZVS. The two
windings of the coupled inductor are connected in series and their
common terminal is connected to one end of the primary winding of
the transformer, which has the other end of the primary winding
connected to the ground. The other two terminals of the coupled
inductor are connected to the midpoint of the two bridge legs
through a corresponding blocking capacitor. The secondary side can
be implemented with any type of the full-wave rectifier such, for
example, the full-wave rectifier with a center-tap secondary, the
full-wave rectifier with current doubler, or the full-bridge
full-wave rectifier. The output voltage regulation in the converter
is achieved by employing a constant-frequency phase-shift control
as in the circuit in FIG. 1(a).
[0013] The circuit in FIG. 2 utilizes the energy stored in the
magnetizing inductance of the coupled inductor to discharge the
capacitance across the switch that is about to be turned on and,
consequently, achieve ZVS. By properly selecting the value of the
magnetizing inductance of the coupled inductor, the primary
switches in the converter in FIG. 2 can achieve ZVS even at no
load. This feature is quite different from the characteristics of
the conventional FB ZVS where the capacitances of the lagging-leg
switches are discharge by the energy stored in the output filter
inductor, whereas the discharge of the capacitances of the
leading-leg switches is done by the energy stored in the leakage
inductance of the transformer or external inductance. Because in
the circuit in FIG. 2 the energy required to create ZVS conditions
at light loads does not need to be stored in the leakage
inductance, the transformer leakage inductance can be minimized. As
a result, the loss of the duty cycle on the secondary-side is
minimized, which maximizes the turns ratio of the transformer and,
consequently, minimizes the conduction losses. In addition, the
minimized leakage inductance of the transformer significantly
reduces the secondary-side ringing caused by the resonance between
the leakage inductance and junction capacitance of the rectifier,
which greatly reduces the power dissipation of a snubber circuit
that is usually used to damp the ringing.
[0014] In this invention, the concept employed to achieve ZVS of
the primary switches in the converter in FIG. 2 is generalized. The
generalized concept is used to derive a family of FB ZVS converters
with the same characteristics.
SUMMARY OF THE INVENTION
[0015] The present invention discloses a family of isolated,
constant-frequency, phase-shift-modulated FB ZVS-PWM converters
that provide ZVS of the bridge switches in a wide range of input
voltage and load current. Generally, the converters of this family
employ two transformers that are connected to the bridge legs so
that a change in the phase shift between the two legs of the bridge
increases the volt-second product on the windings of one
transformer and decreases the volt-second product on the windings
of the other transformer. By connecting a load circuit to the
secondary winding(s) of one transformer and by regulating the
output of the load circuit, the energy stored in a properly
selected magnetizing inductance of the other transformer can be
used for creating ZVS conditions. Specifically, as the load current
and/or input voltage decreases, the phase shift between the bridge
legs changes so that the volt-second product on the windings of the
transformer connected to the load also decreases. At the same time,
the volt-second product on the windings of the other transformer
increases, which increases the energy stored in the magnetizing
inductance of the transformer. Therefore, since available energy
for ZVS stored in the magnetizing inductance increases as the load
current and/or input voltage decreases, the circuits of the present
invention can achieve ZVS in a very wide range of input voltage and
load current, including no load.
[0016] Since the energy used to create the ZVS condition at light
loads is not stored in the leakage inductances of the transformer,
the transformer's leakage inductances can be minimized, which also
minimizes the duty-cycle loss on the secondary side of the
transformer. As a result, the converters of this invention can
operate with the largest duty cycle possible, thus minimizing both
the conduction loss of the primary switches and voltage stress on
the components on the secondary side of the transformer, which
improves the conversion efficiency. Moreover, because of the
minimized leakage inductances, the secondary-side parasitic ringing
caused by a resonance between the leakage inductances and the
junction capacitance of the rectifier is also minimized so that the
power dissipation of a snubber circuit usually required to damp the
ringing is also reduced.
[0017] The circuits of the present invention can be either
implemented as dc/dc converters, or dc/ac inverters. If implemented
as dc/dc converters, any type of the secondary-side rectifier can
be employed such, for example, the full-wave rectifier with a
center-tap secondary winding, full-wave rectifier with current
doubler, or a full-bridge full-wave rectifier. In addition, in some
embodiments of the present invention, the transformer that is not
connected to the load circuit reduces to a single winding
inductor.
BRIEF DESCRIPTION OF THE DRAWINGS
[0018] FIG. 1 shows the conventional full-bridge ZVS-PWM converter:
(a) circuit diagram of power stage; (b) gate-signal timing
diagrams. (prior art).
[0019] FIG. 2 shows the improved full-bridge ZVS-PWM converter with
wide ZVS range.
[0020] FIG. 3 shows a generalized embodiment of the full-bridge
ZVS-PWM converter of this invention.
[0021] FIG. 4 shows the control timing diagrams of the switches and
the voltages across the primary windings of transformers TX and TY
(voltages v.sub.AB and v.sub.CO, respectively).
[0022] FIG. 5 shows a simplified circuit diagram of the converter
in FIG. 3 when output Y is regulated.
[0023] FIG. 6 shows the key current and voltage waveforms of the
circuit in FIG. 5.
[0024] FIG. 7 shows a simplified circuit diagram of the converter
in FIG. 3 when output X is regulated.
[0025] FIG. 8 shows the key current and voltage waveforms of
circuit in FIG. 7.
[0026] FIG. 9 is another generalized embodiment of the full-bridge
ZVS-PWM converter of this invention.
[0027] FIG. 10 is another generalized embodiment of the circuit in
FIG. 3 obtained by splitting the primary winding of transformer
TY.
[0028] FIG. 11 is another generalized embodiment of the circuit in
FIG. 9 obtained by splitting the primary winding of transformer
TY.
[0029] FIG. 12 shows the implementation of the dc/dc FB ZVS-PWM
converter derived from the circuit in FIG. 3 when output Y is
regulated.
[0030] FIG. 13 shows implementation of dc/dc FB ZVS-PWM converter
derived from the circuit in FIG. 9 when output Y is regulated.
[0031] FIG. 14 shows the implementation of the dc/dc FB ZVS-PWM
converter derived from the circuit in FIG. 11 when output X is
regulated.
[0032] FIG. 15 shows the implementation of the dc/dc FB ZVS-PWM
converter derived from the circuit in FIG. 9 when output X is
regulated.
[0033] FIG. 16 shows the implementation of the dc/dc FB ZVS-PWM
converter derived from the circuit in FIG. 3 when output X is
regulated.
[0034] FIG. 17 shows the implementation of a high-power dc/dc
converter that employs two FB ZVS-PWM converters that share the
same current-doubler rectifier. Each FB ZVS-PWM converter is
derived from the circuit in FIG. 3 by regulating output Y.
[0035] FIG. 18 shows a pre-charging circuit for capacitor C.sub.B1
for the circuit implementation on FIG. 15.
[0036] FIG. 19 shows a pre-charging circuit for capacitors C.sub.B1
and C.sub.B2 for the circuit implementation in FIG. 2.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT OF THE
INVENTION
[0037] FIG. 3 shows one of the generalized embodiments of the
isolated, phase-shift-controlled FB ZVS-PWM converter of this
invention. The circuit in FIG. 3, employs two transformers TX and
TY, which have their respective secondary outputs connected to two
output circuits X and Y. Generally, in the dc/dc implementations of
the converter in FIG. 3, each output circuit X and Y includes a
rectifier, low-pass filter, and load, whereas in the dc/ac
(inverter) applications each output circuit X and Y consists only
of a combination of a load and filter. Two constant voltage sources
V.sub.1 and V.sub.2, connected in series with the primary winding
of transformer TX, are employed to provide the volt-second balance
on the windings of both transformers so that the transformers do
not saturate.
[0038] Generally, the volt-second products of the windings of
transformers X and Y in the circuit in FIG. 3 are dependent on the
phase-shift between the turn-on instances of the corresponding
switches in bridge legs S.sub.1-S.sub.2 and S.sub.3-S.sub.4, as
illustrated in FIG. 4. Namely, for the zero phase shift, i.e., when
switches S.sub.1 and S.sub.2 and their corresponding switches
S.sub.3 and S.sub.4 are turned on and off in unison (D=0 in FIG.
4), voltage v.sub.AB across the primary of transformer TX is zero.
As a result, for the zero phase shift, the volt-second product of
the primary winding of transformer TX is also zero. At the same
time, since voltage v.sub.AC across winding AC must be equal to
voltage v.sub.CB across winding CB because windings AC and CB have
the same number of turns, and since v.sub.AB=v.sub.AC+v.sub.CB=0,
it follows that v.sub.AC=v.sub.BC=0. As a result, voltage vco
across the primary winding of transformer TY is V.sub.IN/2, i.e.,
the volt-second product of the primary winding of this transformer
is maximal. Similarly, when switches S.sub.1 and S.sub.2 and their
corresponding switches S.sub.3 and S.sub.4 are turned on and off in
antiphase, i.e., with a 180.degree. phase shift (D=1 in FIG. 4),
the volt-second product on the primary of transformer TX is
maximal, whereas the volt-second product of the primary winding of
transformer TY is zero (minimal). Because the output voltages of
output circuits X and Y are directly proportional to the
volt-second products of the corresponding primary windings, the
circuit in FIG. 3 delivers power to outputs X and Y in a
complementary fashion. Specifically, for zero phase shift (D=0),
the maximum power is delivered to output Y, whereas no power (or
minimal power) is delivered to output X. For 180.degree. phase
shift (D=1), the maximum power is delivered to output X, whereas no
power is delivered to output Y.
[0039] Because the incremental changes of the delivered power to
outputs X and Y with phase-shift changes are in opposite
directions, the circuit in FIG. 3 cannot simultaneously regulate
both outputs if constant-frequency control is employed.
Nevertheless, the property of the circuit to deliver power to
outputs X and Y in the complementary fashion makes the circuit
ideal for implementing ZVS of the primary switches in a wide range
of input voltage and load current. Namely, as already explained,
the conventional FB ZVS-PWM converter has difficulties achieving
ZVS of the leading-leg switches. Specifically, as the load
decreases the energy available for discharging the capacitance of
the leading-leg switch that is about to be turned on, which is
stored in the leakage inductance of the transformer and any
seriesly connected external inductance, is decreasing as the load
decreases. If in the converter in FIG. 3 one output is regulated,
the energy in that output will decease as the load decreases. At
the same time, the energy stored in the magnetizing inductance of
the associated transformer will also decrease because a lighter
load requires a smaller volt-second product on the primary winding
of the transformer. However, the energy in the other, unregulated,
output circuit and in the magnetizing inductance of the
corresponding transformer will increases because of an increased
volt-second product on the primary of the transformer. This
increased energy in the unregulated output circuit and in the
magnetizing inductance of its transformer can be used to create the
ZVS condition for the primary switches at lighter loads, including
no load.
[0040] To facilitate the analysis of the operation of the circuit
in FIG. 3, FIG. 5 shows its simplified circuit diagram when output
Y is regulated. In the simplified circuit in FIG. 5, it is assumed
that only energy stored in the magnetizing inductance of
transformer TX of the unregulated output is used to create the ZVS
condition. Because no energy stored in output circuit X is used to
create the ZVS condition, output circuit X and the associated
secondary of transformer X are not shown in FIG. 5. In fact, since
in the circuit in FIG. 5 only the primary windings of transformer
TX are used, transformer TX operates as a coupled inductor.
Generally, this simplification does not have a significant effect
on the operation of the circuit. Namely, if energy stored in output
circuit X is used for ZVS in addition to the energy stored in the
magnetizing inductance of transformer TX, the only effect of output
circuit X is to increase the total available energy that can be
used for creating the ZVS condition. However, due to a reduced
component count, the implementation in FIG. 5 is preferred in
practice.
[0041] The further simplified the analysis, it is assumed that the
resistance of the conducting semiconductor switches is zero,
whereas the resistance of the non-conducting switches is infinite.
In addition, the leakage inductances of both transformers are
neglected since their effect on the operation of the circuit is not
significant. Finally, the magnetizing inductance of transformer TY
of the regulated output is also neglected since it does not have a
significant effect on the operation of the circuit (although the
energy stored in this inductance could be used to assist ZVS at
heavier loads). However, the magnetizing inductance of transformer
TX, which operates as a coupled inductor, and output capacitances
of primary switches C.sub.1-C.sub.4 are not neglected in this
analysis since they play a major roll in the operation of the
circuit. Consequently, in FIG. 5, transformer TX is modeled as an
ideal transformer with magnetizing inductance L.sub.MX connected
across the series connection of primary windings AC, whereas
transformer TY is modeled only by an ideal transformer with turns
ratio n.sub.Y. It should be noted that magnetizing inductance
L.sub.MX of transformer TX represents the inductance measured
between terminals A and B.
[0042] With reference to FIG. 5, the following relationships
between currents can be established:
i.sub.PY=i.sub.PX1+i.sub.PX2, (1)
N.sub.PYi.sub.PY=N.sub.SYi.sub.SY, (2)
i.sub.1=i.sub.PX1+i.sub.MX (3)
i.sub.2=i.sub.PX2-i.sub.MX (4)
[0043] Since the number of turns of winding AC and winding CB of
transformer TX are the same, it must be that
i.sub.PX1=i.sub.PX2. (5)
[0044] Substituting Eq. (5) into Eqs. (1)-(4) gives 1 i PX 1 = i PX
2 = i SY 2 n Y , ( 6 ) i 1 = i SY 2 n Y + i MX , ( 7 ) i 2 = i SY 2
n Y - i MX , ( 8 )
[0045] where N.sub.Y=N.sub.PY/N.sub.SY is the turns ratio of
transformer TY.
[0046] As can be seen from Eqs. (7) and (8), currents of both
bridge legs i.sub.1 and i.sub.2 are composed of two components:
load-current component i.sub.SY/2n.sub.Y and magnetizing-current
component i.sub.MX. The load-current component is directly depended
on the load current, whereas the magnetizing current does not
directly depend on the load, but rather on the volt-second product
across the magnetizing inductance. Namely, a change of the
magnetizing current with a change in the load current occurs only
if the phase shift is changed to maintain the output regulation.
Usually, the change of the phase shift with the load change is
greater at light loads, i.e., as the load decreases toward no load,
than at heavier loads. Since in the circuit in FIG. 5 the phase
shift increases as the load approaches zero, the volt-second
product of L.sub.MX also increases so that the circuit in FIG. 5
exhibits the maximum magnetizing current at no load, which makes
possible to achieve ZVS at no load.
[0047] Because magnetizing current i.sub.MX does not contribute to
the load current, but flows between the two bridge legs, as seen in
FIG. 5, it represents a circulating current. Generally, this
circulating current and its associated energy should be minimized
to reduce losses and maximize the conversion efficiency. Due to an
inverse dependence of the volt-second product of L.sub.MX on the
load current, circuit in FIG. 5 circulates less energy at the full
load than at a light load, and, therefore, features ZVS in a wide
load range with a minimum circulating current.
[0048] To further understand the operation of the circuit in FIG.
5, FIG. 6 shows its key current and voltage waveforms when the
circuit is implemented as a dc/dc converter. The waveforms in FIG.
6 are obtained based on the analysis described in patent
application Ser. No. 09/652,869 that assumes that output circuit Y
comprises a low-pass LC filter, which has a large filter inductance
L.sub.F so that during a switching cycle the reflected load current
into the primary of transformer TY is constant, as shown in
waveform (k) in FIG. 6. As can bee seen from waveforms (m) and (n)
in FIG. 6, for all four primary switches S.sub.1 through S.sub.4
the magnitude of the current flowing trough the switch at the
turn-off moment is the same, i.e., 2 i 1 ( T 1 ) = i 2 ( T 4 ) = i
1 ( T 7 ) = i 2 ( T 10 ) = i PY 2 + i MX , ( 9 )
[0049] where, I.sub.MX is the amplitude of the magnetizing current
i.sub.MX.
[0050] According to Eq. (9), the commutation of the switches in
both legs, during which the capacitance of the turned-off switch is
charging (voltage across the switch is increasing) and the
capacitance of the switch that is about to be turned on is
discharging (voltage across the switch is decreasing), is done by
the energy stored by both primary current i.sub.PY and magnetizing
current i.sub.MX. While the commutation energy contributed by
magnetizing current i.sub.MX is always stored in magnetizing
inductance L.sub.MX of transformer TX, the commutation energy
contributed by current i.sub.PY nis stored either in the filter
inductance (not shown in FIG. 5) of output circuit Y, or leakage
inductances (not shown in FIG. 5) of transformers TX and TY.
Specifically, for leading-leg switches S.sub.1 and S.sub.2, the
commutation energy contributed by i.sub.PY is stored in
output-filter inductor L.sub.F, whereas for lagging-leg switches
S.sub.3 and S.sub.4 it is stored in the leakage inductance of the
transformers. Since it is desirable to minimize the leakage
inductance of transformer TY to minimize the secondary-side
parasitic ringing, the energy stored in its leakage inductances is
relatively small, i.e., much smaller than the energy stored in
output-filter inductance. As a result, in the circuit in FIG. 3, it
is easy to achieve ZVS of leading-leg switches S.sub.1 and S.sub.2
in the entire load range, whereas ZVS of the lagging-leg switches
S.sub.3 and S.sub.4 requires a proper sizing of the magnetizing
inductance L.sub.MX since at light loads almost entire energy
required to create the ZVS condition of lagging-leg switches
S.sub.3 and S.sub.4 is stored in the magnetizing inductance.
[0051] A similar analysis can be performed by assuming that output
X of the circuit in FIG. 3 is regulated. A simplified circuit
diagram when output X is regulated is shown in FIG. 7. In the
simplified circuit in FIG. 7, it is assumed that only energy stored
in the magnetizing inductance of transformer TY of the unregulated
output is used to create the ZVS condition. Because no energy
stored in output circuit Y is used to create the ZVS condition,
output circuit Y is not shown in FIG. 7. Furthermore, because of
the absence of output circuit Y, transformer TY operates with the
open secondary winding, i.e., only the primary winding of the
transformer is involved in the operation of the circuit. Therefore,
in the circuit in FIG. 7, transformer TY operates as an inductor.
In the simplified circuit in FIG. 7, this inductor is modeled by
inductance L.sub.MY. Also, in FIG. 7, the magnetizing inductance of
transformer TX is neglected because it has no important roll in the
operation of the circuit. Generally, this simplification does not
have a significant effect on the operation of the circuit. Namely,
if energy stored in output circuit Y is used for ZVS in addition to
the energy stored in the magnetizing inductance of transformer TY,
the only effect of output circuit Y is to increase the total
available energy that can be used for creating ZVS condition.
However, due to a reduced component count, the implementation in
FIG. 7 is preferred in practice.
[0052] With reference to FIG. 7, the following relationships
between currents can be established:
N.sub.PXi.sub.1-N.sub.PXi.sub.2-N.sub.SXi.sub.SX=0 (10)
i.sub.MY=i.sub.1+i.sub.2 (11)
[0053] Solving Eqs. (10) and (11) for i.sub.1 and i.sub.2 gives 3 i
1 = i MY 2 + i SX 2 n X , ( 12 ) i 2 = i MY 2 - i SX 2 n X , ( 13
)
[0054] where n.sub.X-N.sub.PX/N.sub.SX is the turns ratio of
transformer TX.
[0055] As can be seen from Eqs. (12) and (13), currents of both
bridge legs i.sub.1 and i.sub.2 are composed of two components:
load-current component i.sub.SX/2n.sub.X and magnetizing-current
component i.sub.MY/2. The load-current component is directly
depended on the load current, whereas the magnetizing current does
not directly depend on the load, but rather on the volt-second
product across the magnetizing inductance. Namely, a change of the
magnetizing current with a change in the load current occurs only
if the phase shift is changed to maintain the output regulation.
Usually, the change of the phase shift with the load change is
greater at light loads, i.e., as the load decreases toward no load,
than at heavier loads. Moreover, since in the circuit in FIG. 7 the
phase shift decreases as the load approaches zero, the volt-second
product of L.sub.MY also increases so the circuit in FIG. 7
exhibits the maximum magnetizing current at no load, which makes
possible ZVS at no load.
[0056] As can be seen from FIG. 7, magnetizing current i.sub.MY
does not contribute to the load current because one-half of this
current flows through primary windings AC and CB of transformer X
in opposite directions. Therefore, current i.sub.MY represents a
circulating current that should be minimized. Due to an inverse
dependence of the volt-second product of L.sub.MY on the load
current, the circuit in FIG. 7, likewise the circuit in FIG. 5,
circulates less energy at full load than at light load, and,
therefore, features ZVS in a wide load range with a minimum
circulating current.
[0057] FIG. 8 shows key current and voltage waveforms of the
circuit in FIG. 7, when the circuit is implemented as a dc/dc
converter. The waveforms in FIG. 8 are obtained by assuming that
output circuit X comprises a low-pass LC filter, which has a large
filter inductance L.sub.F so that during a switching cycle the
reflected load current into the primary of transformer TX is
constant, as shown in waveform (k) in FIG. 8. As can be seen from
waveforms (m) and (n) in FIG. 8, for all four primary switches
S.sub.1 through S.sub.4 the magnitude of the current flowing trough
the switch at the turn-off moment is the same, i.e., 4 i 2 ( T 1 )
= i 1 ( T 4 ) = i 2 ( T 7 ) = i 1 ( T 10 ) = i SX 2 n X + I MY 2 ,
( 14 )
[0058] where, I.sub.MY is the amplitude of the magnetizing current
i.sub.MY.
[0059] However, it should be noted that opposite from the
implementation in FIG. 5, in the implementation in FIG. 7 the
energy for creating the ZVS condition of the lagging-leg switches
are S.sub.3 and S.sub.4 is stored in the output filter inductor,
whereas the energy for creating the ZVS condition of leading-leg
switches are S.sub.1 and S.sub.2 is stored in the leakage
inductances of transformer TX and inductance L.sub.MY. Therefore,
in the circuit in FIG. 7, it is harder to achieve ZVS of the
leading-leg switches than the legging-leg switches. In fact, since
almost all energy for zero-voltage commutation of leading-leg
switches S.sub.1 and S.sub.2 is stored in inductance L.sub.MY, to
achieve ZVS of the leading-leg switches in a wide load range
requires a proper sizing of the magnetizing inductance
L.sub.MY.
[0060] Other generalized embodiments of the isolated,
phase-shift-controlled FB ZVS-PWM converter of this invention are
shown in FIGS. 9, 10, and 11. The operation and characteristics of
the generalized circuits in FIGS. 9, 10, and 11, are identical to
those of the circuit in FIG. 3. In fact, the circuit in FIG. 9 is
obtained by shifting of voltage sources V.sub.1 and V.sub.2 from
the respective primaries of transformer TX into the primary of
transformer TY. Since this circuit transformation does not change
any of the circuit's branch currents and node voltages, it also
does not change the waveforms of the circuit. Circuits in FIGS. 10
and 11 are obtained from the circuits in FIGS. 3 and 9,
respectively, by splitting the primary winding of transformer Y.
Since this transformation also does not change any of the circuit's
branch currents and node voltages, the operation of all generalized
circuits shown in FIGS. 3, 9, 10, and 11 is identical.
[0061] According to the generalized embodiments shown in FIGS. 3,
9, 10, and 11, a number of FB ZVS-PWM converter circuits can be
derived. FIGS. 12 through 17 shows some examples of these circuits
implemented as dc/dc converters. It should be noted that other
implementations, or variations of the shown implementations are
possible. Specifically, the presented generalized circuits and
their implementations can be implemented as dc/ac inverters, as
well.
[0062] The circuit in FIG. 12 is derived from the circuit in FIG. 3
by implementing output circuit Y with the current-doubler
rectifier. Transformer TX of the unregulated output is implemented
as coupled inductor L.sub.C, whereas voltage sources V.sub.1 and
V.sub.2 are implemented with capacitors C.sub.B1 and C.sub.B2,
respectively. Namely, if capacitors C.sub.B1 and C.sub.B2 are large
enough so that the resonant frequency of the series resonant
circuit formed by these capacitors and the magnetizing inductance
of L.sub.C is much smaller than the switching frequency than the
voltage across capacitors is constant and equal to V.sub.IN/2. It
also should be noted that the circuit in FIG. 12 could be also
implemented with other types of the secondary-side rectifier
circuit such, for example, the full-wave rectifier, as shown in
FIG. 2.
[0063] FIG. 13 shows an embodiment of the circuit in FIG. 9. In
this embodiment voltage source V.sub.1 is implemented by splitting
the rail voltage with two capacitors C.sub.B1. Theoretically,
capacitor C.sub.B, which serves to prevent the saturation of
transformer TX if the switching waveforms of the bridge legs are
not identical, is not necessary. However, it is always used in
practice. Generally, the voltage across capacitor C.sub.B is small
(close to zero) since this capacitor only takes on the voltage
difference caused by a mismatching of the bridge legs, which is
usually small.
[0064] FIG. 14 shows the implementation of the FB ZVS-PWM converter
according to the circuit in FIG. 11 when Y is regulated output,
whereas FIG. 15 shows the circuit in FIG. 11 when X is the
regulated output. Both embodiments employ capacitor C.sub.B1 to
implement source V.sub.1. It should be noted that the circuit in
FIG. 14 uses coupled inductor L.sub.C to store energy for ZVS,
whereas inductor L in the circuit in FIG. 15 is uncoupled. In both
circuits, voltage source V.sub.1 can also be implemented by
splitting the rail as it was done in FIG. 13.
[0065] Finally, FIGS. 16 and 17 show two more implementations of
the FB ZVS-PWM converter. The circuit in FIG. 16 is derived from
the generalized circuit in FIG. 3 by regulating output X. The
circuit in FIG. 17, which is suitable for high-power applications
with a high input voltage, employs two FB ZVS-PWM converters as in
FIG. 12 that share the same current-doubler rectifier. In this
circuit, switch pairs Q.sub.1-Q.sub.6, Q.sub.2-Q.sub.5,
Q.sub.3-Q.sub.8, and Q.sub.4-Q.sub.7 are turned on and off
simultaneously.
[0066] As already explained, in the circuits of this invention, it
is more difficult to achieve ZVS of the switches in one bridge leg
than in the other because the energy available for creating the ZVS
condition in the two legs is different. Generally, the ZVS
condition is harder to create for the switches that are in the
bridge leg which utilizes the energy stored in the magnetizing
inductance of the transformer in the unregulated output and energy
stored in the leakage inductances of the transformers. To achieve
ZVS this energy must be at least equal the energy required to
discharge the capacitance of the switch which is about to be turned
on (and at the same time charge the capacitance of the switch that
just has been turned off). At heavier load currents, ZVS is
primarily achieved by the energy stored in the leakage inductances
of transformers TX and TY. As the load current decreases, the
energy stored in the leakage inductances also decreases, whereas
the energy stored in the magnetizing inductance of the transformer
of the unregulated output increases so that at light loads this
magnetizing provides an increasing share of the energy required for
ZVS. In fact, at no load, this magnetizing inductance provides the
entire energy required to create the ZVS condition. Therefore, if
the value of the magnetizing inductance of the transformer in the
unregulated output is selected so that ZVS is achieved at no load
and maximum input voltage V.sub.IN(max), ZVS is achieved in the
entire load and input-voltage range.
[0067] Neglecting the capacitances of the transformer's windings,
magnetizing inductance L.sub.MX necessary to achieve ZVS of
legging-leg switches in the implementations where output Y is
regulated is 5 L MX 1 32 Cf S 2 , ( 15 )
[0068] whereas, magnetizing inductance L.sub.MY required to achieve
ZVS of leading-leg switches in the implementations where output X
is regulated is 6 L MY 1 128 Cf S 2 ( 16 )
[0069] where C is the total capacitance across the primary switches
(parasitic and external capacitance, if any) in the corresponding
legs.
[0070] As can be seen from FIG. 5, current i.sub.MX flowing through
magnetizing inductance L.sub.MX introduces a current asymmetry in
the two bridge legs. Namely, since in the circuits of this
invention that have output Y regulated i.sub.1=i.sub.2+2i.sub.MX
(as can be derived from Eqs. (3)-(5)), leading leg S.sub.1-S.sub.2
always carries a higher current than lagging leg S.sub.3-S.sub.4.
On the other hand, for the circuits of this invention that have
output X regulated, as for example that shown in FIG. 7, both legs
carry the same current. Furthermore, if the current imbalance in
the circuits with regulated output Y is significant, i.e., if
current i.sub.2 in lagging lag S3-S.sub.4 is significantly lower
than current i.sub.1 in leading leg S.sub.1-S.sub.2, different size
switches can be selected for the two legs, which may reduce the
cost of the implementation without sacrificing the circuit
performance.
[0071] Finally, it should be noted that in the circuits of this
invention the parasitic ringing on the secondary side is
significantly reduced because these circuits do not require
increased leakage inductances of the transformers, or a large
external to store the required energy for ZVS of the lagging-leg
switches. Since the transformers in the circuits of this invention
can be made with small leakage inductances, the secondary-side
ringing between the leakage inductances of the transformers and the
junction capacitance of the rectifier can be greatly reduced. Any
residual parasitic ringing can be damped by a small (low-power)
snubber circuit.
[0072] The control of the circuits of this invention is the same as
the control of any other constant-frequency FB ZVS converter. In
fact, any of the integrated phase-shift controllers available on
the market can be used implement the control of the proposed
circuit. However, it should be noted that in the circuits with
regulated output Y the maximum output voltage (volt-second product)
is obtained when the bridge legs are operated in phase (0.degree.
phase shift), whereas the maximum output voltage (volt-second
product) for the circuits with regulated output X occurs when the
bridge legs are operated in antiphase (180.degree. phase shift).
This difference in the control characteristics of the two circuit
implementations has a minor effect on the control loop design since
a simple control-signal inversion in the voltage control loop
solves the problem.
[0073] Because voltage sources V.sub.1=V.sub.IN/2 and
V.sub.2=V.sub.IN/2 in FIGS. 3, 9, 10, and 11 are implemented with
capacitors C.sub.B1 and C.sub.B2, respectively, as shown in FIGS. 2
and 12 through 17, it is necessary to pre-charge these capacitors
to V.sub.IN/2 before the start-up moment. Namely, without
pre-charging the voltages of the capacitors are zero, which causes
a volt-second imbalance on the windings of the transformers during
the start-up. This volt-second imbalance may lead to the saturation
of the transformers, which produces excessive currents in the
primary that may damage the switches. FIGS. 18 and 19 show examples
of pre-charging circuits. FIG. 18 shows a pre-charging circuit
implemented with resistors Rc for the circuit shown in FIG. 15,
whereas FIG. 19 shows a pre-charging circuit implementation for the
circuit in FIG. 2. It should be noted that many other
implementations of the pre-charging circuit are possible for any
circuit of this invention.
[0074] It also should be noted the above detailed descriptions are
provided to illustrate specific embodiments of the present
invention and are not intended to be limiting. Numerous variations
and modifications within the scope of this invention are possible.
The present invention is set forth in the following claims.
* * * * *