U.S. patent application number 09/865537 was filed with the patent office on 2002-04-18 for synchronous motor control device, electric motor vehicle control device and method of controlling synchronous motor.
This patent application is currently assigned to Hitachi, Ltd.. Invention is credited to Kaneko, Satoru, Masaki, Ryoso.
Application Number | 20020043953 09/865537 |
Document ID | / |
Family ID | 17576629 |
Filed Date | 2002-04-18 |
United States Patent
Application |
20020043953 |
Kind Code |
A1 |
Masaki, Ryoso ; et
al. |
April 18, 2002 |
Synchronous motor control device, electric motor vehicle control
device and method of controlling synchronous motor
Abstract
A synchronous motor control system includes a synchronous motor
1, an inverter 3 and a controller 4 wherein a current differential
detecting unit 13 detects a variation of a motor current when the
three phases of the motor 1 is short circuited by the inverter 3,
namely at the moment when a carrier wave in a PWM signal generator
9 assumes maximum or minimum value, in a calculating unit 14 a
phase .gamma. from .alpha. axis of a stationary coordinate system
to a three phase short circuited current differential vector is
calculated, a phase .delta. is estimated from d axis to the three
phase short circuited current differential vector by making use of
d axis current id and q axis current iq on d-q axes coordinate
system in the controller 4, thereafter the magnetic pole position
.theta. with respect to .alpha. axis is calculated from the phases
.gamma. and .delta., based on thus calculated magnetic pole
position .theta., d-q axes control units 11, 7 and 8 are
constituted to control the synchronous motor, thereby a highly
reliable control system for the motor which permits a detection of
the magnetic pole position without affecting a state of applied
voltage thereto while performing a usual PWM control with a low
cost controller.
Inventors: |
Masaki, Ryoso; (Hitachi-shi,
JP) ; Kaneko, Satoru; (Naka-gun, JP) |
Correspondence
Address: |
CROWELL & MORING, LLP
Intellectual Property Group
P.O. Box 14300
Washington
DC
20044-4300
US
|
Assignee: |
Hitachi, Ltd.
|
Family ID: |
17576629 |
Appl. No.: |
09/865537 |
Filed: |
May 29, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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|
09865537 |
May 29, 2001 |
|
|
|
09409992 |
Sep 30, 1999 |
|
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|
6281656 |
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Current U.S.
Class: |
318/700 |
Current CPC
Class: |
H02P 6/18 20130101; Y02T
10/7005 20130101; Y02T 10/72 20130101; B60L 2220/14 20130101; B60L
2220/18 20130101; Y02T 10/70 20130101; Y02T 10/64 20130101; Y02T
10/7077 20130101; B60L 2220/16 20130101; Y02T 10/7072 20130101;
B60L 50/16 20190201; B60L 15/20 20130101; B60L 50/60 20190201; Y02T
10/7275 20130101; B60L 50/51 20190201; Y02T 10/643 20130101; B60L
15/025 20130101 |
Class at
Publication: |
318/700 |
International
Class: |
H02P 001/46; H02P
003/18; H02P 005/28; H02P 007/36 |
Foreign Application Data
Date |
Code |
Application Number |
Sep 30, 1998 |
JP |
10-276946 |
Claims
1. A synchronous motor control device comprising: a synchronous
motor; an inverter which drives said synchronous motor; and a
controller which detects a magnetic pole position of said
synchronous motor based on a variation amount of a motor current
when said synchronous motor is put in a short circuited state and
outputs a control signal to control said synchronous motor based on
the detected magnetic pole position, wherein said inverter controls
said synchronous motor based on the control signal.
2. A synchronous motor control device comprising: a synchronous
motor; an inverter which drives said synchronous motor; and a
controller which detects a magnetic pole position of said
synchronous motor based on a variation direction of a motor current
when said synchronous motor is put in a short circuited state and
generates a control signal to control said synchronous motor based
on the detected magnetic pole position, wherein said inverter
controls said synchronous motor based on the control signal.
3. A synchronous motor control device comprising: a synchronous
motor; an inverter which drives said synchronous motor and a
controller which generates a control signal, wherein said inverter
drives said synchronous motor based on the control signal generated
by said controller; and said controller which detects a variation
direction of a motor current when said synchronous motor is in a
short circuited state, sets a d-q axes coordinate system while
assuming the magnetic flux direction of a rotor of said synchronous
motor as d axis and an axis orthogonal to the d axis as q axis,
detects a d axis current and a q axis current on the d-q axes
coordinate system, calculates the magnetic pole position of said
synchronous motor based on the detected variation direction of the
motor current, the detected d axis current and the detected q axis
current and generates the control signal depending on the
calculated magnetic pole position.
4. A synchronous motor control device comprising: a three phase
synchronous motor; a controller which generates PWM signals based
on three phase voltage command values; an inverter which drives
said three phase synchronous motor with the PWM signals; and
wherein said controller which detects a motor current in
synchronism with a PWM signal of a phase having an intermediate
voltage command value among the three phase voltage command values,
detects the magnetic pole position of said three phase synchronous
motor based on the detected current and determines the three phase
voltage command values depending on the detected magnetic pole
position.
5. A synchronous motor control device according to claim 1, wherein
the short circuited state of said synchronous motor is a state that
all of the phases thereof are short circuited.
6. A synchronous motor control device according to claim 2, wherein
the short circuited state of said synchronous motor is a state that
all of the phases thereof are short circuited.
7. A synchronous motor control device according to claim 3, wherein
the short circuited state of said synchronous motor is a state that
all of the three phases thereof are short circuited.
8. A synchronous motor control device according to claim 1, wherein
said controller detects the variation amount or the variation
direction of the motor current at the moment when said synchronous
motor is placed in a two phase short circuited state which is
generated when said inverter controls said synchronous motor
through a PWM control.
9. A synchronous motor control device according to claim 2, wherein
said controller detects the variation amount or the variation
direction of the motor current at the moment when said synchronous
motor is placed in a two phase short circuited state which is
generated when said inverter controls said synchronous motor
through a PWM control.
10. A synchronous motor control device according to claim 3,
wherein said controller detects the variation amount or the
variation direction of the motor current at the moment when said
synchronous motor is placed in a two phase short circuited state
which is generated when said inverter controls said synchronous
motor through a PWM control.
11. A synchronous motor control device according to claim 1,
wherein said controller detects the variation amount or the
variation direction of the motor current at the moment when said
synchronous motor is placed in a three phase short circuited state
which is generated when said inverter controls said synchronous
motor through a PWM control.
12. A synchronous motor control device according to claim 2,
wherein said controller detects the variation amount or the
variation direction of the motor current at the moment when said
synchronous motor is placed in a three phase short circuited state
which is generated when said inverter controls said synchronous
motor through a PWM control.
13. A synchronous motor control device according to claim 1,
wherein said controller obtains the variation amount or the
variation direction of the motor current under a short circuited
state representing the same when said synchronous motor is placed
in a three phase short circuited state from a variation amount or
variation direction of the motor currents at the time of a
plurality of different two phase short circuited states of said
synchronous motor.
14. A synchronous motor control device according to claim 2,
wherein said controller obtains the variation amount or the
variation direction of the motor current under a short circuited
state representing the same when said synchronous motor is placed
in a three phase short circuited state from a variation amount or
variation direction of the motor currents at the time of a
plurality of different two phase short circuited states of said
synchronous motor.
15. A synchronous motor control device according to claim 3,
wherein said controller obtains the variation amount or the
variation direction of the motor current under a short circuited
state representing the same when said synchronous motor is placed
in a three phase short circuited state from a variation amount or
variation direction of the motor currents at the time of a
plurality of different two phase short circuited states of said
synchronous motor.
16. A synchronous motor control device according to claim 1,
wherein said controller comprises prolonging means which prolongs
the three phase short circuited state.
17. A synchronous motor control device according to claim 2,
wherein said controller comprises prolonging means which prolongs
the three phase short circuited state.
18. A synchronous motor control device according to claim 3,
wherein said controller comprises prolonging means which prolongs
the three phase short circuited state.
19. A synchronous motor control device according to claim 16,
wherein said prolonging means prolongs the three phase short
circuited state through a two phase switching operation.
20. A synchronous motor control device according to claim 17,
wherein said prolonging means prolongs the three phase short
circuited state through a two phase switching operation.
21. A synchronous motor control device according to claim 18,
wherein said prolonging means prolongs the three phase short
circuited state through a two phase switching operation.
22. A synchronous motor control device according to claim 1,
wherein said controller judges an abnormality in the synchronous
motor control device through comparison of a first motor speed
obtained from a variation state of the detected magnetic pole
position with a second motor speed obtained from the variation
amount of the motor current.
23. A synchronous motor control device according to claim 2,
wherein said controller judges an abnormality in the synchronous
motor control device through comparison of a first motor speed
obtained from a variation state of the detected magnetic pole
position with a second motor speed obtained from the variation
amount of the motor current.
24. A synchronous motor control device according to claim 3,
wherein said controller judges an abnormality in the synchronous
motor control device through comparison of a first motor speed
obtained from a variation state of the detected magnetic pole
position with a second motor speed obtained from the variation
amount of the motor current.
25. A synchronous motor control device according to claim 4,
wherein said controller judges an abnormality in the synchronous
motor control device through comparison of a first motor speed
obtained from a variation state of the detected magnetic pole
position with a second motor speed obtained from the variation
amount of the motor current.
26. A synchronous motor control device comprising: a synchronous
motor; a magnetic pole position detector which detects a magnetic
pole position of said synchronous motor; a controller which
controls said synchronous motor depending on the magnetic pole
position detected by said magnetic pole position detector; and an
inverter which drives said synchronous motor based on a signal from
said controller, wherein said controller determines the magnetic
pole position of said synchronous motor based on a variation amount
or a variation direction of a motor current when said synchronous
motor is placed in a short circuited state and detects an
abnormality in said magnetic pole position detector or the
controller through comparison of the magnetic pole position
detected by the magnetic pole position detector with the magnetic
pole position obtained from the variation amount or the variation
direction of the motor current.
27. A synchronous motor control device according to claim 26,
wherein when an abnormality in said magnetic pole position detector
is detected, said controller controls said synchronous motor based
on the magnetic pole position obtained from the variation amount or
the variation direction of the motor current.
28. An electric motor vehicle control device comprising: a
synchronous motor which drives the electric motor vehicle; a
magnetic pole position detector which detects a magnetic pole
position of said synchronous motor; a controller which controls
said synchronous motor depending on the magnetic pole position
detected by said magnetic pole position detector; and an inverter
which drives said synchronous motor based on a signal from said
controller, wherein said controller determines the magnetic pole
position of said synchronous motor based on a variation amount or a
variation direction of a motor current when said synchronous motor
is placed in a short circuited state and detects an abnormality in
said magnetic pole position detector or the controller through
comparison of the magnetic pole position detected by the magnetic
pole position detector with the magnetic pole position obtained
from the variation amount or the variation direction of the motor
current.
29. An electric motor vehicle control device according to claim 28,
wherein when an abnormality in said magnetic pole position detector
is detected, said controller controls said synchronous motor based
on the magnetic pole position obtained from the variation amount or
the variation direction of the motor current.
30. An electric motor vehicle control device according to claim 28,
wherein when said controller detects an abnormality, said
controller stops the electric motor vehicle.
31. An electric motor vehicle control device according to claim 30,
wherein after stopping the electric motor vehicle, said controller
restarts a drive of the electric motor vehicle by making use of a
normal magnetic pole position among the magnetic pole position
obtained from magnetic pole position detector and the magnetic pole
position obtained from the motor current through said controller
determined as normal.
32. A synchronous motor control method comprising: a first step of
detecting a variation direction of a motor current when a
synchronous motor is in a short circuited state; a second step of
setting a d-q axes coordinate system while assuming the magnetic
flux direction of a rotor of the synchronous motor as d axis and an
axis orthogonal to the d axis as q axis; a third step of detecting
a d axis current and a q axis current on the d-q axes coordinate
system; a fourth step of calculating the magnetic pole position of
the synchronous motor based on the detected variation direction of
the motor current, the detected d axis current and the detected q
axis current; and a fifth step of controlling the synchronous motor
depending on the calculated magnetic pole position.
33. A synchronous motor control method according to claim 32,
wherein when a difference between the magnetic pole position on the
d-q axes coordinate system set in said second step and the magnetic
pole position calculated in said fourth step is in a predetermined
range, the synchronous motor is controlled based on the magnetic
pole position calculated in said fourth step.
34. A synchronous motor control device comprising: a synchronous
motor; a controller; an inverter which drives said synchronous
motor based on an output from said controller; and wherein said
controller includes a first detector unit which detects a magnetic
pole position of said synchronous motor based on a variation amount
or a variation direction of a motor current when said synchronous
motor is placed in a short circuited state and a control unit which
generates the output based on the detected magnetic pole position,
and further includes a second detector unit which detects a
magnetic pole position at the time when a rotating speed of said
synchronous motor is in a predetermined low range including a
rotation stoppage state, wherein said controller controls the
synchronous motor based on an output from said second detector unit
when a rotating speed of said synchronous motor is in a
predetermined low range including a rotation stoppage state and
controls based on an output from said first detector unit when a
rotating speed of said synchronous motor is in a range larger than
the predetermined low range.
35. A synchronous motor control device according to claim 34,
wherein said second detector unit includes a signal generating
means which generates a signal for detecting a current used for
calculating the magnetic pole position and a polarity
discriminating unit which discriminates whether the calculated
magnetic pole position is N pole direction or S pole direction.
36. A synchronous motor control device according to claim 35,
wherein said controller sends out PWM pulses to said inverter so
that a current flows through said synchronous motor based on the
output from said signal generating means, calculates a magnetic
pole position by detecting the current of the synchronous motor
based on the output from said signal generating means and controls
the synchronous motor.
37. A synchronous motor control device according to claim 35,
further comprising a current control unit which controls a torque
of said synchronous motor, wherein said polarity discriminating
unit applies to said current control unit a current command for
discriminating polarity in d axis direction and discriminates
polarity of the magnetic pole based on differences in response
characteristics of said synchronous motor.
38. A synchronous motor control device according to claim 35,
wherein said polarity discriminating unit discriminates polarity of
the magnetic pole based on the torque of said synchronous motor
which is generated depending on the calculated magnetic pole
position and rotating direction of a rotor shaft of said
synchronous motor.
39. A synchronous motor control device according to claim 34,
further comprising a mechanism which prevents rotation of said
synchronous motor, wherein when the rotation of said synchronous
motor is prevented by said mechanism, a current based on an output
current from said second detector unit is supplied to said
synchronous motor.
40. An electric motor vehicle control device comprising the
synchronous motor control device according claim 34.
41. An electric motor vehicle control device comprising the
synchronous motor control device according claim 35.
42. An electric motor vehicle control device comprising the
synchronous motor control device according claim 36.
43. An electric motor vehicle control device comprising the
synchronous motor control device according claim 37.
44. An electric motor vehicle control device comprising the
synchronous motor control device according claim 38.
45. An electric motor vehicle control device comprising the
synchronous motor control device according claim 39.
46. An electric motor vehicle control device according to claim 40,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
rotation of a driving wheel of said electric motor vehicle is
prevented including a state wherein a brake for said electric motor
vehicle is activated.
47. An electric motor vehicle control device according to claim 41,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
rotation of a driving wheel of said electric motor vehicle is
prevented including a state wherein a brake for said electric motor
vehicle is activated.
48. An electric motor vehicle control device according to claim 42,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
rotation of a driving wheel of said electric motor vehicle is
prevented including a state wherein a brake for said electric motor
vehicle is activated.
49. An electric motor vehicle control device according to claim 43,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
rotation of a driving wheel of said electric motor vehicle is
prevented including a state wherein a brake for said electric motor
vehicle is activated.
50. An electric motor vehicle control device according to claim 44,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
rotation of a driving wheel of said electric motor vehicle is
prevented including a state wherein a brake for said electric motor
vehicle is activated.
51. An electric motor vehicle control device according to claim 45,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
rotation of a driving wheel of said electric motor vehicle is
prevented including a state wherein a brake for said electric motor
vehicle is activated.
52. An electric motor vehicle control device according to claim 40,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
an operation range of said electric motor vehicle is in parking
range.
53. An electric motor vehicle control device according to claim 41,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
an operation range of said electric motor vehicle is in parking
range.
54. An electric motor vehicle control device according to claim 42,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
an operation range of said electric motor vehicle is in parking
range.
55. An electric motor vehicle control device according to claim 43,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
an operation range of said electric motor vehicle is in parking
range.
56. An electric motor vehicle control device according to claim 44,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
an operation range of said electric motor vehicle is in parking
range.
57. An electric motor vehicle control device according to claim 45,
wherein the current supply to said synchronous motor based on the
output from said second detector unit is started under a state when
an operation range of said electric motor vehicle is in parking
range.
Description
BACKGROUND OF THE INVENTION
[0001] 1. Field of the Invention
[0002] The present invention relates to a control device which
controls a synchronous motor including a reluctance motor, and a
method of controlling a synchronous motor including a reluctance
motor, and an electric motor vehicle control device using the
same.
[0003] 2. Conventional Art
[0004] In order to control such as speed and torque of a
synchronous motor it is necessary to detect or estimate its
magnetic pole position, and thus the speed and torque of the
synchronous motor can be controlled through a current control or a
voltage control thereof based on the detected or estimated magnetic
pole position.
[0005] Conventionally, the magnetic pole position was detected by a
position detector. However, recently a method of controlling a
synchronous motor while estimating the magnetic pole position, in
that a control method with magnetic pole position sensorless has
been proposed which is different from the conventional method of
detecting the magnetic pole position by making use of a position
sensor.
[0006] For example, Takeshita, Ichikawa et al. "Control of Salient
Type Brushless DC Motor with Sensorless Based on Estimation of
Speed Electromotive Force" (Collected Papers of Japanese Electrical
Engineers Society Vol. 117-D, No. 1, 1997) proposes a method of
performing speed control of a motor while estimating a speed
electromotive force by making use of a motor model.
[0007] Further, JP-A-8-205578 (1996) discloses a method of
detecting a salient pole characteristic of a synchronous motor
based on a correlation of ripple components of a voltage vector
applied to the synchronous motor through a pulse width modulation
control (hereinafter referred to as PWM control) and of the
corresponding motor current vector.
[0008] The art disclosed in the above paper is a method of
estimating the magnetic pole position based on a difference between
a current calculated on the control model and an actual motor
current flowing therethrough, and has a feature that a control
system can be formed only through control calculations in a
controller.
[0009] Further, since the art disclosed in JP-A-8-205578 (1996)
uses general PWM signals which control a voltage of the synchronous
motor, the method has an advantage that no additional signals for
detecting the magnetic pole position are required.
[0010] Further, with the method of estimating magnetic pole
position based on a difference between a current calculated from a
control model and an actual motor current flowing therethrough,
there was an unsolved problem that once the synchronous motor steps
out on any causes, the synchronous motor may be brought into an
out-of-control condition.
[0011] On the other hand, with the art disclosed in JP-A-8-205578
(1996) the magnetic pole position of the synchronous motor can
always be detected by its salient pole characteristic, therefore,
the synchronous motor is never brought into an out-of-control
condition.
[0012] However, with the method of detecting the magnetic pole
position of a synchronous motor through its salient pole
characteristic, it is necessary to detect a correlation between the
motor current state and the applied voltage every time when the PWM
signals change.
[0013] Namely, it is necessary to detect the motor current state
and to grasp the applied voltage state at least six times for one
cycle of a carrier wave, for this reason there arose a problem that
the calculation speed can not catch up with, if a controller of
high performance is not used.
SUMMARY OF THE INVENTION
[0014] An object of the present invention is to provide a
synchronous motor control device which can be produced with low
cost.
[0015] Another object of the present invention is to provide a
highly reliable synchronous motor control system.
[0016] One of the measures according to the present invention is to
calculate, namely to estimate a magnetic pole position of the
synchronous motor based on a variation amount or a variation
direction of a motor current when the synchronous motor is put in a
short circuited state and to control the synchronous motor based on
the calculated magnetic pole position.
[0017] Other measures according to the present invention will be
explained in the Detailed Description of the Preferred
Embodiments.
BRIEF DESCRIPTION OF THE DRAWINGS
[0018] FIG. 1 is a block diagram showing one embodiment of the
present invention in which a magnetic pole position of a cylinder
type synchronous motor is detected by making use of a current
differential circuit ;
[0019] FIG. 2 is a circuit diagram of the inverter 3 in FIG. 1
;
[0020] FIG. 3 is a time chart showing a relation between a carrier
wave signal, three phase voltage command values and PWM signals,
and a fetching timing of an inverter current in the embodiment
shown in FIG. 1;
[0021] FIG. 4 is a flow chart when detecting magnetic pole position
in the embodiment shown in FIG. 1;
[0022] FIG. 5 is a block diagram showing another embodiment of the
present invention in which the magnetic pole position is calculated
by detecting motor currents when two phases of a cylinder type
synchronous motor is under a short circuited condition ;
[0023] FIG. 6 is a time chart showing a relation between a carrier
wave signal, three phase voltage command values and PWM signals,
and a fetching timing of an inverter current in the embodiment
shown in FIG. 5;
[0024] FIG. 7 is a flow chart when detecting magnetic pole position
in the embodiment shown in FIG. 5;
[0025] FIG. 8 is a Table showing arithmetic expressions for
calculating current difference values when two phases being short
circuited and phases of current differential vectors when three
phases being short circuited in steps 115 and 116 in FIG. 7;
[0026] FIG. 9 is a block diagram showing still another embodiment
of the present invention in which the magnetic pole position of a
salient type synchronous motor is detected by making use of the
inverter currents while prolonging the three phase short circuited
interval ;
[0027] FIG. 10 is a time chart showing a relation between a carrier
wave signal, three phase voltage command values and PWM signals,
and a fetching timing of an inverter current in the embodiment
shown in FIG. 9;
[0028] FIG. 11 is a flow chart when detecting magnetic pole
position with a high accuracy in the embodiment shown in FIG.
9;
[0029] FIG. 12 is a block diagram showing a further embodiment of
the present invention which comprises a magnetic pole position
sensor for controlling a salient type synchronous motor for an
electric motor vehicle and a magnetic pole position detecting means
which detects the magnetic pole position thereof based on the
inverter currents when two phases being short circuited;
[0030] FIG. 13 is a flow chart when detecting the magnetic pole
position of the salient pole type synchronous motor by making use
of the inverter currents when two phases being short circuited in
the embodiment shown in FIG. 12;
[0031] FIG. 14 is a Table showing arithmetic expressions for
calculating current difference values when two phases being short
circuited and phases of current differential vectors when three
phases being short circuited in steps 136 and 137 in FIG. 13;
[0032] FIG. 15 is a flow chart when performing an abnormality
judgement of the magnetic pole position in the embodiment shown in
FIG. 12;
[0033] FIG. 16 is a block diagram showing a still further
embodiment of the present invention which includes a self diagnosis
function of false detection in magnetic pole position in a salient
pole type synchronous motor having a magnetic pole position
detecting means detecting the magnetic pole position by making use
of inverter currents when two phases being short circuited;
[0034] FIG. 17 is a vector diagram showing an exemplary relation
between a current vector, a differential current vector and
magnetic pole position, in other words, d axis in a synchronous
motor;
[0035] FIG. 18 is a vector diagram showing a relation between a
differential current vector when two phases being short circuited
and a differential current when three phases being short circuited
in the cylinder type synchronous motor shown in FIG. 9;
[0036] FIG. 19 is a vector diagram showing a relation between
differential current vectors which are generated by a voltage
applied on a axis of a salient pole type synchronous motor;
[0037] FIG. 20 is a vector diagram showing a relation between a
differential current vector when two phases being short circuited
and a differential current when three phases being short circuited
in the salient pole type synchronous motor shown in FIG. 16;
[0038] FIG. 21 is a diagram of a synchronous motor control system
showing another embodiment according to the present invention;
[0039] FIG. 22 is a diagram showing a possible region in which
detection accuracy of the magnetic pole position reduces;
[0040] FIG. 23 is a diagram showing a structure of a calculating
unit 52 in FIG. 21;
[0041] FIG. 24 is a diagram showing a structure of a current
command value generating unit 6 in FIG. 21;
[0042] FIG. 25 is a diagram showing a magnetic characteristic of a
synchronous motor;
[0043] FIG. 26 is a diagram showing a d axis characteristic of a
synchronous motor;
[0044] FIG. 27 is a flow chart showing a processing sequence for
detecting a magnetic pole position during the time when a
synchronous motor is started; and
[0045] FIG. 28 is a flow chart showing a processing sequence for
discriminating polarity of magnetic pole based on torque generating
direction and rotating direction of a rotor shaft of a synchronous
motor.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0046] Hereinbelow, an embodiment of the present invention will be
explained with reference to FIG. 1.
[0047] FIG. 1 is a block diagram of a motor control device in which
a cylinder type synchronous motor 1 is driven by DC energy from a
battery 2. The DC voltage of the battery 2 is inverted by an
inverter 3 into a three phase AC voltage, which is applied to the
cylinder type synchronous motor 1. The inverter 3 is controlled
based on an output of a controller 4.
[0048] The output of the controller 4 is determined based on the
following calculation result. Although the controller 4 in FIG. 1
is illustrated in a functional block diagram, the controller 4 can
be realized not only by a hardware but also by a software. A
differential circuit 12, a current detector unit 10 as well as a
PWM signal generating unit 9, which will be explained later, use
partly an input/output circuit of a computer. For example, the
input/output circuit is such as an analogue/digital converter and a
pulse output circuit, and through their use all of the functions
can be performed by software programs.
[0049] Namely, at first a current command value generating unit 6
determines a d axis current command value idr and a q axis current
command value iqr with respect to a torque command value .tau.r to
be generated from the motor 1. Further, the torque command value
.tau.r is issued to the current command value generating unit 6
from a control device or a control program which is in a higher
hierarchy with respect to the controller 4.
[0050] The d axis is a direction in the magnetic pole or the
magnetic fluxes, the q axis is electrically orthogonal direction to
the d axis, and d axis and q axis in combination constitute d-q
axes coordinate system. When a rotor with magnets of a motor
rotates, the d-q axes coordinate system also rotates, therefore, a
phase of the d-q axes coordinate system from a stationary
coordinate system, in that .alpha.-.beta. axes coordinate system,
is assumed as .theta.. Namely, an object of the present embodiment
is to detect the phase .theta. of the magnetic pole (hereinbelow,
referred to as magnetic pole position .theta.) based on inverter
currents.
[0051] FIG. 17 shows a vector diagram illustrating one exemplary
relation between coordinate systems and currents therein. If the d
axis current and the q axis current can be controlled according to
the command values, the synchronous motor 1 can generate a torque
coincident with the torque command value .tau.r. The value of the
torque command .tau.r is commanded either directly to the current
command value generating unit 6 or indirectly via a speed control
calculating circuit (not showing). Signals respecting the values of
a U phase current iu and a V phase current iv from current sensors
5a and 5b are sent to a current detecting unit 10 and are detected
by the current detecting unit 10 at a current detection timing P1
which will be explained later. The detected current values are
respectively converted by a coordinate system converting unit 11
into a d axis current id and a q axis current iq for the d-q axes
coordinate system.
[0052] In the present embodiment, the currents detected by the
current detecting unit 10 are two phase currents iu and iv of U
phase and V phase, this is because W phase current iw can be
determined by the U and V phase currents iu and iv and the
detection of W phase current iw is omitted. Of course, all of the
three phase currents can be detected.
[0053] A current control unit 7 calculates a d axis current
deviation between the d axis current command value idr and the d
axis current id and a q axis current deviation between the q axis
current command value iqr and the q axis current, and performs a
proportion and integration calculation for the respective
deviations to determine a d axis voltage command value Vdr and a q
axis voltage command value Vqr.
[0054] A voltage setting unit 8, which receives the d axis voltage
command value Vdr and the q axis voltage command value Vqr,
calculates three phase voltage command values Vur, Vvr and Vwr for
the stationary coordinate system based on a magnetic pole position
.theta. and outputs the same to a PWM signal generating unit 9.
[0055] The PWM signal generating unit 9 calculates three phase PWM
pulses Pup, Pvp, Pwp, Pun, Pvn and Pwn and outputs the same to the
inverter 3.
[0056] FIG. 2 shows a relation between the circuit connection
diagram of the inverter 3 and the PWM pulses therefor. For example,
when the PWM pulse Pup is high, a switching element Sup is turned
on, and when the Pup is low, the switching element Sup is turned
off.
[0057] Further, the PWM pulses Pup and Pun are generally in an
opposite relation with regard to high and low state. However, in
order to prevent a power source short circuiting, a short circuit
preventing interval is provided which keeps the both PWM pulses in
a low state, when the state of the PWM pulses are inverted.
[0058] Processing contents performed in the PWM signal generating
unit 9 are explained with reference to a timing chart as shown in
FIG. 3. Through comparison of the wave forms of the respective
phase voltage command values Vur, Vvr and Vwr with triangular wave
shaped carrier waves, three phase PWM pulses Pup, Pvp and Pwp are
obtained. Further, an illustration of the above mentioned short
circuit preventing interval is omitted in the drawing for simplify
the explanation.
[0059] Namely, when the PWM pulses Pup, Pvp and Pwp are in high in
FIG. 3, the switching elements Sup, Svp and Swp in upper arms in
FIG. 2 are respectively placed in an on state, and the switching
elements Sun, Svn and Swn in lower arms therein are respectively
placed in an off state. When the PWM pulses Pup, Pvp and Pwp are
low, the switching elements Sun, Svn and Swn are respectively in an
on state and the switching elements Sup, Svp and Swp are
respectively in an off state.
[0060] As will be seen from FIG. 3, when the voltage command values
of the respective phases are in a predetermined range including
maximum value and minimum value of the carrier waves, there exists
an interval in which three phases either in the upper arms or in
the lower arms are in a short circuited condition. When the
detection use pulse P1 is designed to be generated when the carrier
wave reaches to its maximum value and to its minimum value, the
detection use pulse P1 is resultantly generated when the three
phases of the synchronous motor are in a short circuited state.
[0061] Further, it is known that when the current detection unit 10
is designed to detect the currents of the respective phases when
the detection use pulse P1 is generated, the detected instantaneous
current values substantially correspond to respective average
current values of the concerned phases.
[0062] Still further, the short circuited state of respective phase
windings in the synchronous motor exists not only at a moment of
the maximum value and the minimum value of the carrier waves as
shown in FIG. 3 but also exists in a predetermined range including
the same. The predetermined range is represented by a pulse width
among PWM pulses Pup, Pvp and Pwp having the narrowest pulse
interval and by an interval between the most wide pulse and the
adjacent pulse thereto. Timing t1 appears in a width range of pulse
Pvp, timing t2 appears between two successive pulses Pup, timing t3
appears in a width range of pulse Pvp, timing t4 appears between
another successive two pulses Pup, timing t5 appears in a width
range of pulse Pwp and timing t6 appears between still another two
successive pulses Pup. Still further, the timings t1 through t6
represent moments either the maximum value or the minimum value of
the carries waves. As has been explained above during a
predetermined interval including the moments of the respective
maximum and minimum values the short circuited state of the phase
winding is caused and which is repeated. In order to take out a
current flowing through the windings under a short circuited state
thereof, the pulse p1 is produced. It is sufficient when the pulse
p1 is generated at the predetermined interval. The method according
to the present embodiment in which the detection use pulses are
generated at the timings of the maximum value and the minimum value
of the carrier waves shows advantages such as that the detection
use pulses are easy to produce and a possibility of erroneous
operation is reduced, because the detection use pulses are
generated at substantially the center period of the short circuited
state.
[0063] Now, an important principle of the present embodiment as
shown in FIG. 1 is explained.
[0064] A current differential circuit 12 is inputted of signals
representing such as the U phase current iu and the V phase current
iv and outputs differential current values piu and piv obtained by
differentiating or affine differentiating the input current
values.
[0065] These differentiated current values such as piu and piv are
inputted into a detection unit 13 and are held until the detection
use pulse P1 is generated, and thereafter are outputted. Namely,
the current differential values piu and piv are detected at the
timing of the pulses p1, in other words are fetched into a
calculating unit 14.
[0066] The calculating unit 14 which calculates a magnetic pole
position performs the processings as shown in the flow chart in
FIG. 4 to determine the magnetic pole position .theta..
[0067] At first, in step 101 the differentiated current values piu
and piv when the three phases are short circuited, are inputted
into the calculating unit 14.
[0068] In step 102, a phase .gamma. of a differentiated current
vector pis, when the three phases are short circuited, is
calculated and determined.
[0069] In FIG. 17, phase relations of the differentiated current
vector pis with respect to other vectors are illustrated. From the
differentiated current values piu and piv when the three phases are
short circuited an .alpha. axis differentiated current value
pi.alpha. and .alpha. .beta. axis differentiated current value
pi.beta. can be determined.
[0070] When the U phase axis coincides with the .alpha. axis, the
.alpha. axis differentiated current value pi.alpha. and the .beta.
axis differentiated current value pi.beta. are respectively
obtained by the following arithmetic formulas;
pi.alpha.=({square root}{square root over (3/2)})piu (1)
pi.beta.=(1/{square root}{square root over (2)})(piu-2piv) (2)
[0071] Subsequently, the phase .gamma. is calculated based on the
thus determined values pi.alpha. and pi.beta. by making use of the
relations illustrated in FIG. 17.
[0072] In step 103, the magnetic pole position .theta. is
determined according to the following arithmetic formula;
.theta.=.gamma.+.pi./2 (3)
[0073] One of the feature of the present embodiment is our
discovery that a relation between the magnetic pole position
.theta. and the phase .gamma. of the three phase short circuited
current is approximately expressed by the above arithmetic formula
(3) of which ground will be explained below.
[0074] Fundamental operation of a synchronous motor in d-q axes
coordinate system are expressed by the following arithmetic
formulas, wherein p=d/dt and .omega. represents a rotating angular
speed of the motor;
Vd=(R+pLd)id-.omega.Lqiq (4)
Vd=(R+pLq)iq+.omega.(Ldid+.phi.) (5)
[0075] When a synchronous motor is placed in a three phase short
circuited state, the applied voltage of the synchronous motor
stands Vd=Vq=0, therefore, the condition of the synchronous motor
when the three phases are short circuited is expressed by the
following arithmetic formulas;
pid=(.omega.Lqiq-Rid)/Ld (6)
piq=-{.omega.(Ldid+.phi.)+Riq}/Lq (7)
[0076] The differentiated current vector in the stationary
.alpha.-.beta. axes coordinate system is a sum of the
differentiated current vector in d-q axes coordinate system and a
differentiated current vector generated through the rotation of the
d-q axes coordinate system at an angular speed .omega., therefore,
a d axis differentiated current value pids and a q axis
differentiated current value piqs seen from the .alpha.-.beta. axes
coordinate system are respectively expressed by the following
arithmetic formulas;
pids={.omega.(Lq-Ld)iq-Rid}/Lq (8)
piqs=-{.omega.(Ld-Lq)id+.phi.)+Rid}/Lq (9)
[0077] Accordingly, the phase .delta. of the differentiated current
vector when three phase are short circuited with respect to d axis,
namely the magnetic pole position .theta., is expressed by the
following arithmetic formula;
tan
(.delta.)=piqs/pids=-Ld[.omega.{(Ld-Lq)id+.phi.}+Rid]/[Lq{.omega.(Lq-L-
d)iq-Rid}] (10)
[0078] In the present embodiment, since the cylinder type
synchronous motor 1 is used, a condition Ld=Lq is given, therefore,
the above arithmetic equation (10) is modified as follows;
tan (.delta.)=Ld(.omega..phi.+Riq)/(LqRid) (11)
[0079] When id<0, the phase .delta. is approximated by the
following arithmetic formula;
.delta..apprxeq.-.pi./2 (12)
[0080] For this reason, the calculation according to the arithmetic
formula (3) is performed in step 103.
[0081] When the motor angular speed .omega. is low, the error based
on the approximation (12) increases, therefore, the phase .delta.
can be obtained asymptotically based on the arithmetic formula (11)
of which method will be explained later in connection with other
embodiments.
[0082] As has been explained above, through a simple calculation in
the calculating unit 14 as shown in FIG. 1 the magnetic pole
position .theta. can be determined. When coordinate conversions are
performed in the voltage setting unit 8 and in the coordinate
conversion unit 11 based on the thus determined magnetic pole
position .theta., the motor is controlled to generate a required
torque corresponding to a torque command value.
[0083] Accordingly, the present embodiment is characterized by the
fact that the magnetic pole position of a cylinder type synchronous
motor can be detected through a comparatively simple calculation
only with the provision of current sensors without using a
mechanical magnetic pole position sensor such as a resolver and
encoder which directly measures the rotating position of the
magnetic pole of the cylinder type synchronous motor. For this
reason the control device is produced with a low cost.
[0084] Further, even if the synchronous motor steps out on any
causes, the synchronous motor is never brought into an
out-of-control condition, because the magnetic pole position can
always be detected.
[0085] Moreover, the present embodiment is characterized by the
fact that in parallel with a usual PWM control since a sensorless
control system is constructed only by making use of information
obtained during the performance of the PWM control, noises and
torque ripple of the synchronous motor are reduced in comparison
with the conventional method of detecting the magnetic pole
position by applying detection use additional signals.
[0086] FIG. 5 is a block diagram of another embodiment for a
cylinder type synchronous motor in which the magnetic pole position
is detected without using a current differential circuit. Like FIG.
1 embodiment, the present embodiment is also realized not only by
electric circuits but also by computer softwares.
[0087] Major different points of the present embodiment from that
shown in FIG. 1 embodiment are that the current differential
circuit 12 is not used, the current detection timing is modified by
an introduction of a detection use pulse P2 and a different
processing other than that in the calculating unit 14 as shown in
FIG. 1 is performed in the calculating unit 15. An important
feature of the present embodiment is that the three phase short
circuited current is not directly detected.
[0088] Now, the detection use pulse P2 which controls detection
timing of the current detector unit 10 is explained with reference
to FIG. 6. FIG. 6 shows the same state of PWM signals as that shown
in FIG. 3, however, the current detection use pulses P2 as shown in
FIG. 6 is different from the current detection use pulses P1 as
shown in FIG. 3 in the following points.
[0089] With respect to respective phases of a 180.degree.
conduction type three phase inverter as shown in FIG. 2, either the
switching element in the upper arm or the switching element in the
lower arm is usually placed in an on state and the other is placed
in an off state. For this reason, at least two phases among the
three phases are always short circuited.
[0090] FIG. 6 illustrates such interval. For example, in a time
section from time t(n-2) to time t(n-1) the switching elements Svn
and Swn in the lower arms of V phase and W phase are placed in an
on state, and the V phase and the W phase of the synchronous motor
1 are short circuited.
[0091] Further, in a time section from the time t(n-1) to time t(n)
the U phase and the V phase are short circuited through the upper
arms thereof.
[0092] As will be seen from the above, the 180.degree. conduction
type inverter has two modes of two phase short circuited state
during one cycle of the carrier wave.
[0093] As illustrated in FIG. 6, the detection use pulses P2 are
generated at the moment when the modes of two phase short circuited
state charges over.
[0094] The PWM signal generating unit 9 is designed to produce the
detection use pulses P2 in synchronism with the PWM signal which is
generated from the phase having the second largest voltage command
value, namely, the intermediate voltage command value (with regard
to duration time, in other words pulse width), among the voltage
command values for three phases.
[0095] The current detection unit 10 fetches, for example, signals
representing two phase current values, a U phase current iu and a V
phase current iv outputted from the current sensors 5a and 5b every
time when the detection use pulses P2 are generated.
[0096] The U phase and V phase currents obtained at such timings
are inputted from the current detection unit 10 to the calculating
unit 15, in which the processings as shown in FIG. 7 are performed.
A U phase average current value iua and a V phase average current
value iva calculated in the calculating unit 15 are outputted to
the coordinate system converting unit 11, and a magnetic pole
position .theta. also calculated therein is outputted respectively
from the calculating unit 15 to the voltage setting unit 8 and to
the coordinate system converting unit 11 to perform substantially
the same operation as that of FIG. 1 embodiment.
[0097] The flow chart of FIG. 7 illustrating processing contents
performed in the calculating unit 15 is now explained.
[0098] A U phase average current iua(n) and a V phase average
current iva(n) are calculated in step 112 by making use of a V
phase current iu(n) and a V phase current iv(n) at time t(n)
inputted from the current sensors 5a and 5b to the current detector
unit 10 based on the detection use pulse P2 in step 111. An average
of the U phase current iu(n-1) at time t(n-1) and the U phase
current iu(n) at timing t(n) substantially corresponds to the value
of V phase current iu at time t5 in FIG. 3. The processing in step
112 is performed because the U phase current at the generation
timing of the current detection use pulses P1 substantially
corresponds to the average value thereof.
[0099] In the next step 113, current difference values or
differential values of respective phases between time t(n-1) and
time t(n) are calculated.
[0100] In step 114, a two phase short circuited mode Msc is judged
which two phases are in a two phase short circuited state in the
time section from time t(n-1) to time t(n). In the present
instance, it is understood that the upper arms of U phase and V
phase are short circuited from FIG. 6, which is judged in step 114
to determine that the two phase short circuited mode Msc(n) is "U-V
phase short circuited". Further, the previous two phase short
circuited mode Msc(n-1) in the time section from time t(n-2) to
time t(n-1) is "V-W phase short circuited".
[0101] In step 115, the calculation of the short circuited current
difference values is performed by making use of the arithmetic
formulas shown in the table in FIG. 8 to determine a short
circuited current difference value pisc of short circuiting
axis.
[0102] Now, the short circuited current difference value pisc of
short circuiting axis is explained. In FIG. 8, when V-W phase short
circuited, the short circuiting axis corresponds to .beta. axis,
when W-U phase short circuited, the short circuiting axis
corresponds to .beta.' axis, and when U-V phase short circuited,
the short circuiting axis corresponds to .beta." axis.
[0103] For example, when converting three phase voltages into
.alpha.-.beta. axes coordinate system, in that to coincide the U
phase axis with .alpha. axis, .beta. axis voltage V.beta. can be
expressed by the following arithmetic formula;
V.beta.=(Vv-Vw)/({square root}{square root over (2)}) (13)
[0104] wherein, when V-W phases are in a short circuited state,
Vv=Vw, therefore V.beta.=0. Namely, it is understood that the
.beta. axis is in a short circuiting state, accordingly the very
axis is called as a short circuiting axis.
[0105] Likely, when W-U phases are short circuited, .beta.' axis
which is formed by rotating the .beta. axis by 120.degree. assumes
the short circuiting axis, and when U-V are short circuited,
.beta." axis which is formed by rotating the .beta. axis by
240.degree. assumes the short circuiting axis.
[0106] In case of a cylinder type synchronous motor, the short
circuited current difference value pisc of the short circuiting
axis coincides with the short circuiting axis component of the
three phase short circuited current differential vector pis. FIG.
18 shows a vector diagram representing these relations.
[0107] The reason why such vector diagram stands is explained by
developing the arithmetic formulas (4) and (5).
[0108] The .alpha. axis current differential value pia and .beta.
axis current differential value pi.beta. are respectively expressed
as follows from the arithmetic formulas (4) and (5) 1 pi = [ ( L0 -
L1 cos2 ) V - ( L1 sin2 ) V + k1 ( ) i + k2 ( ) i + k3 ( ) ] / ( L0
2 - L1 2 ) ( 14 ) pi = [ - ( L1 sin2 ) V + ( L0 + L1 cos2 ) V + k4
( ) i + k5 ( ) i + k6 ( ) ] / ( L0 2 - L1 2 ) ( 15 )
[0109] wherein, L0=(Ld+Lq)/2, L1=(Ld-Lq)/2, and k1(.theta.),
k2(.theta.), k3(.theta.), k4(.theta.), k5(.theta.) and k6(.theta.)
respectively represent functions relating to magnetic pole position
.theta..
[0110] In case of a cylinder type synchronous motor, since L1=0, it
is understood that the .beta. axis current differential value
pi.beta. affects no influence on the .alpha. axis voltage
V.alpha..
[0111] When V-W phases are in a short circuited state, only the
.alpha. axis voltage V.alpha. is applied depending on the state of
the U phase voltage Vu for the .alpha. axis current differential
value pi.alpha., however, the .beta. axis current differential
value pi.beta. is invariable at the time when V.alpha.=0, moreover,
since V-W phases are in a short circuited state, V.beta.=0,
therefore, this implies that the above .beta. axis current
differential value pi.beta. coincides with the .beta. axis current
differential value pi.beta. under the three phases being short
circuited. With the hitherto explanation it will be understood that
the vector relation as illustrated in FIG. 18 stands.
[0112] Likely, when W-U phases are short circuited, a .beta.' axis
current differential value pi.beta.' becomes identical as the
.beta.' axis component of the three phase short circuited current
differential vector pis. Accordingly, when a current differential
value or a current difference value of a short circuiting axis
under a two phase short circuited state is detected, the phase
.gamma. of the three phase short circuited current differential
value can be calculated based on the vector diagram illustrated in
FIG. 18.
[0113] When determining the phase .gamma. of the three phase short
circuited current differential value based on the two phase short
circuited current mode Msc(n) and the two phase short circuited
previous mode Msc(n-1), the calculation method thereof varies
depending on the combination of the short circuited modes.
[0114] For this reason, in step 116 the phase .gamma. of the three
phase short circuited differential vector is determined by making
use of one of separate arithmetic formulas each determined
depending on the short circuited modes as illustrated in FIG.
8.
[0115] In step 117, a magnetic pole position .theta. is obtained in
the same manner as in step 103 in FIG. 4.
[0116] The present embodiment as has been explained above has an
advantage that a highly accurate detection of the magnetic pole
position can be achieved with a limited current fetching, since the
direction of the current differential vector under a three phase
short circuited state can be determined, namely calculated by the
current variation amount or difference value under the two phase
short circuited state having a comparatively long duration
time.
[0117] Further, the present embodiment uses no differential
circuit, therefore, provides an advantage that a comparatively low
cost controller having a high noise resistance can be realized.
[0118] FIG. 9 is a block diagram of still another embodiment in
which the present invention is applied to a salient pole type
synchronous motor. Like the previous embodiments, the controller 10
can be realized by electrical circuits as well as by computer
softwares. The embodiment as shown in FIG. 9 differs from the
previous embodiments as shown in FIGS. 1 and 5 in connection with
the provision of a two phase switching calculation unit 18 and of
current detection use pulses P3 and P4 generated from the PWM
signal generating unit 9 and a processing method in the magnetic
pole position calculating unit 17.
[0119] The processing content of the two phase switching
calculating unit 18 is explained with reference to the time chart
illustrated in FIG. 10.
[0120] The two phase switching implies a method in which while
inhibiting switching of one phase among three phase PWM signals,
the same sinusoidal currents as in the three phase switching are
produced.
[0121] With FIG. 10, a method of causing a sinusoidal current like
a three phase switching is explained, while, for example,
preventing switching of U phase. V phase or W phase other than the
U phase can be likely used as the switching prevention phase.
[0122] In FIG. 10, an additional voltage V0 is forcedly applied so
that the U phase voltage command value Vur always assumes the same
value as the maximum value of the carrier wave. Thereby, the U
phase PWM signal Pup is always in a high state and the switching
element Sup keeps an on state.
[0123] For V phase voltage command value Vvr and W phase voltage
command value Vwr are determined by adding the additional voltage
V0 to the respective usual command values, and depending thereon
the respective PWM signals Pvp and Pwp are generated.
[0124] When a same voltage is added to all of the phase voltages,
no influence is caused to their line voltages, the current flowing
through the synchronous motor 1 is identical when no additional
voltage is applied, which is a well known two phase switching
method. When employing this method, the three phase short circuited
state for one time is prolonged in comparison with the instance in
FIG. 3 as will be seen from FIG. 10.
[0125] The detection use pulses P3 and P4 generated from the PWM
signal generating unit 9 are also illustrated in FIG. 10.
[0126] The detection use pulses P3 are designed to be generated in
synchronism with the maximum values of the carrier waves and are
used so as to obtain the average current values ius and iva of the
respective phases in the current detection unit 10 as shown in FIG.
9.
[0127] Further, the current detection use pulse P4 are designed to
be generated in synchronism with the start and end of the prolonged
three phase short circuited state. A current detecting unit 27 in
FIG. 9 is inputted U phase current iu and V phase current iv in
response to the current detection use pulses P4.
[0128] These detected current values are inputted into the
calculating unit 17 where the processings as shown by the flow
chart in FIG. 11 are performed to determine the magnetic pole
position .theta..
[0129] The processings as shown in FIG. 11 are performed in the
following manner. In step 121, current difference values piu, piv
and piw of the respective phases are calculated by making use of U
phase current iu(n-1) and V phase current iv(n-1) at the start time
t(n-1) of the three phase short circuited state and U phase current
iu(n) and V phase current iv(n) at the end time t(n), of which
processing is similar to that in step 113 in FIG. 7.
[0130] In the next step 122, the phase .gamma. of the three phase
short circuited current differential vector is calculated by making
use of the current difference values piu, piv and piw, of which
processing is also similar to that in step 102 in FIG. 4.
[0131] In the following steps, it is assumed that a magnetic pole
position which is used for control in the controller 4 at that
moment is .theta.' and an actual magnetic pole position of the
synchronous motor 1 is .theta.. Further, it is also assumed that d
axis current and q axis current calculated based on the magnetic
pole position .theta.' used in the controller 4 are respectively
id' and iq', and the actual d axis current and q axis current of
the synchronous motor 16 are respectively id and iq.
[0132] In step 123, the d axis current id' and q axis current iq'
are calculated by making use of the magnetic pole position .theta.'
and of the average current values iua and iva inputted from the
current detecting unit 10.
[0133] In step 124, a calculation according to the arithmetic
formula (10) is performed by making use of id' and iq' instead of
id and iq to determine the phase .gamma. from the magnetic pole
position, in other words d axis, to the three phase short circuited
current differential vector.
[0134] When the motor angular speed .omega. exceeds a predetermined
value, the phase .gamma. can be determined according to the
following approximate formula;
tan (.delta.).apprxeq.-Ld{(Ld-Lq)id+.phi.}/{Lq(Lq-Ld)iq} (16)
[0135] In step 125, the magnetic pole position .theta. is
determined according to the following arithmetic formula by making
use of the phase .gamma. obtained in step 122;
.theta.=.gamma.-.delta. (17)
[0136] The above relation is illustrated by the vector diagram in
FIG. 17.
[0137] In step 126, it is judged whether the magnetic pole position
.theta. is determined in step 125 is substantially coincident with
the magnetic pole position .theta.' used for determining id' and
iq' in step 123. If the both are not coincident, processings from
the step 123 to the step 125 are again repeated to calculate the
magnetic pole position .theta..
[0138] When the magnetic pole position .theta.' used in the
controller 4 differs from the actual magnetic pole position
.theta., id' and iq' do not coincide with id and iq, therefore, an
error is caused in the phase .delta.. However, every time when
performing the processings from step 123 to step 125, the error
decreases and the magnetic pole position .theta.' used in the
controller 4 converges to the actual magnetic pole position
.theta., which is judged in step 126 and if the calculation of the
magnetic pole position substantially converges, the processings
end.
[0139] Further, since it is estimated that the above calculation
converges in a few times, for example two or three times, number of
calculations, for example two times calculation, can be used to
terminate the calculation instead of judging the calculation result
of the magnetic pole position to ascertain the convergence.
[0140] Still further, in view of a relation between sampling time
for detecting the magnetic pole position and motor angular speed,
the step 126 can be omitted and a method of detecting the magnetic
pole position by samplings of a few times can be employed.
[0141] As has been explained above, when detecting the magnetic
pole position of a salient pole type synchronous motor, the
magnetic pole position has to be calculated by making use of the d
axis current id' and q axis current iq' including errors. A feature
of the present embodiment is the provision of an argorism which can
converge the errors, therefore, the present embodiment has an
advantage in which a magnetic pole position sensorless control
system for a salient pole type synchronous motor can be constructed
by making use of current variation under a three phase short
circuited state.
[0142] In the present embodiment, the variation range of the
currents during the three phase short circuited period can be
enlarged by incorporating a method of prolonging the three phase
short circuited period such as the two phase switching method. For
this purpose a method of detecting the magnetic pole position
having a high noise resistance which can directly measure the three
phase short circuited current differential vector without using a
differential circuit can be realized by simple software
processings.
[0143] FIG. 12 is a further embodiment of the present invention
applied to a salient pole type synchronous motor including a highly
reliable system suitable for application to an electric motor
vehicle in which the magnetic pole position is detected in view of
the two phase short circuited state. The controller 4 of the
present embodiment can likely realized not only by electric
circuits but also by computer software programs. Like the previous
embodiments, the respective blocks of the controller 4 can be
understood as representing the corresponding processing functions
of the computer software programs.
[0144] A difference of the present embodiment as shown in FIG. 12
from the embodiment applied to a cylinder type synchronous motor as
shown in FIG. 5 is the processings in a calculating unit 20.
[0145] Further, the salient pole type synchronous motor 16 is
mechanically coupled to wheel tires 24 and 25 of the electric motor
vehicle to drive the same.
[0146] Further, in order to enhance reliability of the electric
motor vehicle, a magnetic pole position sensor 23 is provided which
is designed to detect directly and mechanically the magnetic pole
position of the motor.
[0147] At first, the operation of the calculating unit 20 will be
explained. The processings performed therein are illustrated in
FIG. 13.
[0148] Processings performed from step 131 to step 134 are the same
as performed from step 111 to step 114 in FIG. 7.
[0149] A salient pole related correction phase .epsilon. calculated
in step 135 is a correction amount which is required to take into
account of an influence of the salient pole related
characteristic.
[0150] As will be seen from the arithmetic formula (15), in case of
the salient pole type synchronous motor 16, since L1.noteq.0, the
.beta. axis current differential value pi.beta. varies depending on
the .alpha. axis voltage V.alpha.. For this reason, the .beta. axis
current differential value pi.beta. takes a different value from
the .beta. axis component of the three phase short circuited
current differential vector.
[0151] FIG. 19 illustrates a .alpha. axis current differential
value pi.alpha.1 and a .beta. axis current differential value
pi.beta.1 and a current differential pi1 compounded therefrom.
[0152] When assuming that the axis which coincides with the current
differential vector pi1 is x axis, and another axis orthogonal
thereto is y axis, the y axis component of the current differential
vector pi1 is always zero regardless to the a axis voltage
V.alpha.. For this reason, the y axis component of the current
differential vector pi1 coincides with the y axis component of the
three phase short circuited current differential vector pis, which
is called as the salient pole related correction phase
.epsilon..
[0153] Therefore, in case of the salient pole type synchronous
motor, the y axis current differential value or difference value
which advances from the .beta. axis by the salient pole related
correction phase .epsilon. is detected instead of the .beta. axis
current differential value pi.beta..
[0154] Actually, since there are three types of two phase short
circuited states, when assuming that the salient pole related
correction phases for V-W phases short circuited, W-V phases short
circuited and U-V phases short circuited are respectively
represented as .epsilon.1, .epsilon.2 and .epsilon.3 and the axes
determined thereby are respectively represented as y axis, y' axis
and y" axis, the salient pole related correction phases .epsilon.1,
.epsilon.2 and .epsilon.3 are respectively expressed as follows
from the arithmetic formulas (14) and (15);
tan (.epsilon.1)=-(L1 sin 2.theta.)/(L0-L1 cos 2.theta.) (18)
tan (.epsilon.2)=-{L1 sin (2.theta.-4.pi./3) }/{L0-L1 cos
(2.theta.-4.pi./3)} (19)
tan (.epsilon.3)=-{L1 sin (2.theta.-2.pi./3) }/{L0-L1 cos
(2.theta.-2.pi./3)} (20)
[0155] Accordingly, in step 135 calculation according to one of the
three arithmetic formulas (18), (19) and (20) is performed
depending on the two phase short circuited state to determine a
salient pole related correction phase .epsilon..
[0156] The magnetic pole position .theta. used in these
calculations is one in the controller 4 and includes an error,
therefore, if required, an accurate magnetic pole position can be
determined and used while converging the same according to the
processings as shown in FIG. 11.
[0157] In step 136, a corrected short circuited current difference
value for a short circuited axis, in that either y axis, y' axis
and y" axis, is calculated based on the current difference values
piu(n) and piv(n) by making use of the Table in FIG. 14 to
determine a short circuited current difference pisc for the
converted short circuiting axis.
[0158] The short circuiting axis as has already been explained is
an axis of which current differential value or difference value
direction is not affected by the .alpha. axis voltage.
[0159] In the next step 137, the calculation mode is altered
depending on the current and previous two phase short circuited
states as illustrated in FIG. 14, and a phase .gamma. of the three
phase short circuited current differential vector is determined by
making use of one of three arithmetic formulas shown in FIG.
14.
[0160] FIG. 20 shows an example of vector diagrams which is
determined according to the arithmetic formulas in FIG. 14.
[0161] The processings performed from step 138 to step 140 are
identical to those performed from step 123 to step 125 in FIG. 11
which takes into account of the phase from the magnetic pole
position of the salient pole type synchronous motor 16 to the
current differential vector.
[0162] As will be understood from the above, when the calculating
unit 20 is used, the magnetic pole position even for salient pole
type synchronous motor 16 can be detected only by detecting a
current under the two phase short circuited state thereof.
[0163] In the electric motor vehicle driving system as illustrated
in FIG. 12, a signal from the magnetic pole position sensor 23 is
inputted into a magnetic pole position detecting unit 21 to thereby
detect a magnetic pole position .theta.1.
[0164] The magnetic pole position abnormality detecting unit 22 is
inputted of the magnetic pole position .theta.1 from the magnetic
pole position detecting unit 21 and of the magnetic pole position
.theta. from the magnetic pole position calculating unit 20, and
performs the processings as illustrated in FIG. 15.
[0165] The magnetic pole position calculating unit 20, the magnetic
pole position detecting unit 21 and the magnetic pole position
abnormality detecting unit 22 in combination constitute a magnetic
pole position and current calculating unit 19.
[0166] In step 142 the magnetic pole positions .theta. and .theta.1
inputted in step 141 are compared to judge whether the difference
thereof is in a predetermined normality range.
[0167] When it is judged as normal, the magnetic pole .theta.1 is
stored as an output use magnetic pole position .theta.2 in step
143, and in step 144 the output use magnetic pole position .theta.2
is outputted to the current setting unit 8 and to the coordinate
system conversion unit 11.
[0168] When it is judged in step 142 that the comparison results of
the two magnetic pole positions is outside the predetermined normal
range, the processing of once stopping the electric motor vehicle
is performed in step 145.
[0169] In step 146, it is judged whether the rotation of the
synchronous motor once stopped. When the stoppage is judged, a
drive of the electric motor vehicle is permitted in step 147 by
making use of a normal magnetic pole position within a safety
speed. The safety speed drive implies that the vehicle drives under
a speed limit of 40 km/h or 50 km/h through control of a proper
control device (not shown).
[0170] The advantage of the present embodiment is that a highly
reliable electric motor vehicle control device can be provided with
the introduction of the magnetic pole position sensor 23 and
through the determination of the magnetic pole position based on
the magnetic pole position sensor 23 as well as through the
determination of the magnetic pole position by the calculation
processing of the motor currents.
[0171] In particular, the present embodiment is suitable when the
size reduction of the motor through use of reluctance torque and
light weighting of the electric motor vehicle are intended.
[0172] FIG. 16 is a block diagram of a still further embodiment of
the present invention including a magnetic pole position sensorless
control system which detects the magnetic pole position only with a
current sensor, and moreover has a self diagnosis function to
determine abnormality in the detecting characteristics.
[0173] A feature of the present embodiment as shown in FIG. 16 with
respect to the embodiment as shown in FIG. 12 is the provision of a
self diagnosis unit 26.
[0174] The magnetic pole position calculating unit 20 performs a
calculation of detecting a motor angular speed .omega. in addition
to the processings as illustrated in FIG. 13.
[0175] In the previous embodiments, the magnetic pole position
.theta. is obtained by detecting phase .gamma. of the current
differential vector, and information relating to the magnitude of
the current differential vector is neglected.
[0176] Herein, the motor angular speed .omega. is determined as
follows by making use of the arithmetic formulas (6) and (7);
.omega.=(Ldpid+Rid)/Lqiq (21)
.omega.=-(Lqpiq+Riq)/(Ldid+.phi.) (22)
[0177] The arithmetic formulas (21) and (22) can be modified into
simple arithmetic formulas by neglecting the resistance R. The thus
calculated motor angular speed .omega. is outputted to the self
diagnosis unit 26.
[0178] Further, the magnetic pole position .theta. obtained in step
140 in FIG. 13 is also inputted into the self diagnosis unit
26.
[0179] In the self diagnosis unit 26 it is judged through
comparison of the variation of the magnetic pole position with the
motor angular speed .omega. whether any abnormality in the system
occurs.
[0180] If it is judged there is an abnormality, the self diagnosis
unit 26 outputs an abnormality diagnosis signal Se to stop the
operation of the sensorless control system.
[0181] In the present embodiment as has been explained above,
through estimation of the motor angular speed by making use of a
plurality of independent variables of the current differential
vectors, a self diagnosis function is provided without using an
additional sensor.
[0182] The present invention is applicable to a reluctance motor
other than the synchronous motor by making use of its salient pole
related characteristic.
[0183] Further, in the above embodiments an influence due to
rotation of the motor rotor during a sampling period on the
calculation of the magnetic pole position is neglected for the sake
of explanation simplicity, however, such influence can be taken
into account for the calculation of the magnetic pole position.
[0184] In the above embodiments, an application to an electric
motor vehicle is exemplified. However, the present invention is
also applicable to a magnet motor which is presently sensorlessly
controlled by making use of 120.degree. conduction type inverter so
as to obtain a sensorless control system with small torque ripple
and low noises controlled by a 180.degree. conduction type
inverter.
[0185] The embodiments as has been explained in connection with
FIGS. 1 through 20 are preferable for a control in which the
synchronous motor 1 is rotated above a predetermined rotating
speed, for example more than 800 rpm. When the motor is rotated
under a low speed including stoppage thereof, it is sometimes
necessary to detect the magnetic pole position with further higher
accuracy which will be explained later. An embodiment which permits
the detection of the magnetic pole position with higher accuracy
under a low speed including stoppage of the motor will be explained
with reference to FIG. 21. An important difference of FIG. 21
embodiment from FIG. 1 embodiment is the provision of a calculating
unit 52 which is detectable of the magnetic pole position when the
synchronous motor is rotated under a low speed including stoppage
thereof. The controller 4 of the present embodiment can likely
realized not only by electric circuits but also by computer
software programs. Like the previous embodiments, the respective
blocks of the controller 4 can be understood as representing the
corresponding processing functions of the computer software
programs.
[0186] The processings performed in the calculating unit 14 for
detecting the magnetic pole position are the same as those
explained in connection with FIG. 1. Further, the operations of the
current command value generating unit 6, the current control unit
7, the coordinate system conversion unit 8, the PWM signal
generating unit 9, the current detection unit 10, the coordinate
system conversion unit 11 and the inverter 3 are also fundamentally
the same as those explained in connection with FIG. 1. The
processing method performed for calculating the magnetic pole
position of the synchronous motor based on the variation amount of
the motor current or the variation direction thereof when the
synchronous motor 3 is placed in a three phase short circuited
state is again explained.
[0187] In the present calculating method of the magnetic pole
position, the magnetic pole position is calculated and determined
based on the variation amount of the motor current or the variation
direction thereof under a three phase short circuited state which
is free from an influence of the applied voltage from the inverter
3. In order to detect the variation amount of the motor current or
the variation direction thereof under the three phase short
circuited state, pulses P1 are generated which control the timing
under the three phase short circuited state as has been explained
in connection with FIG. 3. As typical examples, the pulses P1 are
generated at the timings of the maximum value, the minimum value or
the both values of the carrier waves as shown in FIG. 3, and in
synchronism with these pulses P1 the current values or the
differential values thereof are detected. Namely, the pulses P1 are
generated from the PWM signal generating unit 9 and the variation
amount of the motor is detected by using the pulses P1 as signals,
in other words as triggers, representing the three phase short
circuited state of the motor. The relation of the vector namely the
differential vectors of the motor current variation amount detected
at the time when the three phases are short circuited is the same
as that explained in connection with FIG. 17. In FIG. 17, the
magnetic pole position to be detected is the phase .theta. between
.alpha. axis of the stationary coordinate system and d axis of the
rotary coordinate system, namely the phase .theta. is expressed by
the following arithmetic formula (17);
.theta.=.gamma.-.delta. (17)
[0188] wherein .gamma. represents a phase of motor current
differential vector pis, when the three phases are short circuited
with respect to .alpha. axis, and .delta. is a phase of the motor
current differential vector pis with respect to d axis.
[0189] The calculation of the phase .gamma. is performed in the
following manner;
[0190] At first the motor currents iu and iv are differentiated in
the differential circuit 12, and further in synchronism with the
detection use pulses P1 generated when the motor is under a short
circuited state the differential values piu and piv of the motor
currents when the motor is short circuited are fetched at the
current differential value detector unit 13. Further, these motor
current differential values piu and piv when the motor is short
circuited are converted into current differential values pi.alpha.
and pi.beta. for .alpha.-.beta. axes according to the following
arithmetic formulas (1) and (2), thereafter, the phase .gamma. is
calculated according to the following arithmetic formula (23);
pi.alpha.=({square root}{square root over (3/2)})piu (1)
pi.beta.=(1{square root}{square root over (/2)})(piu+2piv) (2)
.gamma.=tan.sup.-1(pi.beta./pi.alpha.) (23)
[0191] In the above, although pi.alpha. and pi.beta. are determined
by making use of the two phase current differential values piu and
piv, the three phase current differential values piu, piv and piw
can be used for the calculation. In the present embodiment the
differential circuit is used for calculating the current
differential values when the three phases are short circuited,
however, when the incorporation of such differential circuit is
impossible, instead of the current differential values current
variation rates can be used which are determined by calculating
current variation amounts in the three phase short circuited
interval time and by dividing the current variation amounts by the
respective short circuited interval times. Still further, if the
short circuited interval times are extremely short and the
calculation of the current variation rates in the three phase short
circuited interval is impossible, the current variation rates in
the three phase short circuited interval can be calculated by
making use of two phase short circuited intervals which is
absolutely longer than the three phase short circuited interval. As
has been explained above in detail, the phase .gamma. of the motor
current differential vector pis when the three phases are short
circuited with respect to .alpha. axis is determined.
[0192] The calculation of the phase .delta. of the motor current
differential vector pis with respect to d axis is performed in the
following manner
[0193] At first, the fundamental behavior of a synchronous motor in
d-q axes of rotary coordinate system is expressed by the following
arithmetic formulas (4) and (5) as has been explained
previously;
Vd=(R+pLd)id-.omega.Lqiq (4)
Vd=(R+pLq)iq+.omega.(Ldid+.phi.) (5)
[0194] wherein, Vd and Vq are d-q axes voltages, Ld and Lq are d-q
axes inductances, R is a winding resistance, .omega. is motor
angular velocity, .phi. is main fluxes of magnetic field and P is
d/dt. In the above arithmetic formulas, when the three phases are
short circuited, the applied voltage in d-q axes assumes zero,
therefore, the above fundamental arithmetic formulas are modified
as follows;
pid=(.omega.Lqiq-Rid)/Ld (6)
piq=-{.omega.(Ldid+.phi.)+Riq}/Lq (7)
[0195] The differentiated current vector in the stationary
.alpha.-.beta. axes coordinate system is expressed by a sum of the
differentiated current vector in d-q axes coordinate system as
expressed by the arithmetic formulas (6) and (7) and a
differentiated current vector generated through the rotation of the
d-q axes coordinate system at an angular speed .omega., therefore,
a d axis differentiated current value pids and a q axis
differentiated current value piqs seen from the .alpha.-.beta. axes
coordinate system are respectively expressed by the following
arithmetic formulas;
pids={.omega.(Lq-Ld)iq-Rid }/Ld (8)
piqs=-{.omega.(Ld-Lq)id+.phi.)+Riq}/Lq (9)
[0196] Accordingly, the phase .delta. of the motor current
differential vector pis with respect to d axis is expressed by the
following arithmetic formula (24) 2 = tan - 1 ( piqs / pids ) = tan
- 1 [ - Ld { ( ( Ld - Lq ) id + ) + Riq } / { Lq ( ( Lq - Ld ) iq -
Rid ) } ] ( 24 )
[0197] Resultantly, with the arithmetic formulas (17), (23) and
(24) the magnetic pole position can be determined. Wherein, as the
.omega. included in the arithmetic formula (24), an estimated
angular velocity value determined by a variation amount of a phase
estimation value. Further, in an operating region where the angular
velocity is sufficiently large and component R is negligible, an
influence of .omega. is eliminated.
[0198] Above is the outline of the position detecting method of
detecting the magnetic pole position of a synchronous motor based
on the variation amount of the motor current or the variation
direction thereof when the three phases of the synchronous motor
are short circuited. The present method can be applicable not only
to a salient pole type synchronous motor but also to a cylindrical
type synchronous motor.
[0199] In the present position detecting method, since the current
variation amount under the motor short circuited state which is
caused during usual PWM operation, estimation of the magnetic pole
position is enabled up to a high speed operation range without
applying any specific estimation signals.
[0200] However, in the following operating region which will be
explained hereinbelow, the position estimation accuracy only with
the present detection method will possibly be reduced. FIG. 22
shows the above referred to operation range. Under the rotation
speed of zero, namely when the rotation of the motor 1 is stopped,
the torque of the motor is also zero. The motor rotation stoppage
state, for example, under the condition when the motor 1 is going
to start, no motor current yet flows. Therefore, the current
variation amount when short circuited can not be detected even if
intended to perform the PWM control. Further, even after the motor
is started, when the rotation speed thereof is low and the torque
thereof is small, the current value is also small, therefore, the
above indicated measurement is also difficult. Resultantly, the
detection accuracy of the magnetic pole position is also reduced.
Still further, even in a region near the zero speed where a current
flows, the direction of magnetic fluxes can be detected by making
use of the current variation amount at the time of short circuited
state, it is impossible to discriminate its polarity, in that
whether the detected direction is in N pole or S pole because of
extremely small effect of the induced voltage.
[0201] For the above mentioned operating region, for example, below
800 rpm a detecting unit 52 is provided according to the present
embodiment so as to enhance the detection accuracy of the magnetic
pole position. The structure and operation of the detecting unit 52
are explained with reference to FIG. 23. The detecting unit 52
primarily includes a signal generating unit 54 which generates
signals used for estimation of the magnetic pole position and a
polarity discrimination unit 56.
[0202] The signal generating unit 54 which generates the signals
used for estimation of the magnetic pole position generates a
current command value idhr used for position detection so as to
cause a current enough to permit the detection of a current
variation amount when the motor is short circuited to provide the
same to the current command value generating unit 6, when detecting
the magnetic pole position in the region where no current is
flowing or in the region where the current value is small, for
example, during the starting-up of the motor 1.
[0203] The structure of the current command value generating unit 6
is shown in FIG. 24. In the embodiments as explained in connection
with FIGS. 1 through 20 as well as in FIG. 21 embodiment, the
torque command Tr is inputted into the current command value
generating unit 6 from an upper hierarchy control unit with respect
to the controller 4. The current command value generating unit 6
includes a torque control unit 63 and calculates current commands
idr and iqr for d-q axes so that the synchronous motor 1 generates
a torque according to the command. Usually, the torque command and
the motor speed (in FIG. 24 illustration thereof is omitted) are
used as its inputs, and proper idr and iqr which achieve the
maximum efficiency at the moment concerned are calculated through
such as a map retrieval thereof. The usual operating state as
referred to above implies, for example, a state where the motor 1
is rotating at a higher speed than the predetermined rotating speed
of 800 rpm.
[0204] As has been explained in the above, when the synchronous
motor 1 is rotating at a low speed less than 800 rpm such as during
starting-up of the synchronous motor 1, the current command idhr
for detecting the magnetic pole position is outputted from the
signal generating unit 54 and is applied to an adder 65. The
current command idhr is added to a current command idr1 in d axis
direction at the adder 65, and the added result is applied as d
axis current command idr from the current command value generating
unit 6 to the current control unit 7. In the present embodiment the
current command idhr for detecting the magnetic pole position is
applied to the current command in d axis direction. This is to
prevent generation of useless torque due to application of idhr.
Namely, if a current in q axis direction is zero, no torque is, in
principle, generated even if a current in d axis direction is
applied. Of course, the magnetic pole position can be detected,
even when a current command iqhr in q axis direction or current
commands idhr and iqhr in d and q axes directions are applied.
Accordingly, when a torque generation is acceptable for the total
control operation a current command in q axis direction can be
added for the detection of the magnetic pole position. Further, the
current command can be applied only for a short period
corresponding to the measuring time.
[0205] Further, the current command idhr to be applied and used for
the position estimation can be either AC signals or DC signals.
When AC signals are used for the current command idhr, an averaged
torque assumes zero which is to be generated due to deviation
between a detected position value which is caused in the initial
period of the estimating operation and the magnetic pole position
of the synchronous motor. Thereby, an influence of torque variation
is suppressed.
[0206] As has been explained hitherto, under the condition that the
speed of the synchronous motor is zero and no current flows
therethrough, for example, when starting rotation of the motor from
its stand still condition, the position detection making use of the
current variation amount when the motor is short circuited can be
used by applying the signals used for position estimation. However,
the detected position indicates direction of fluxes, it is unclear
whether the direction is in N pole direction or in S pole
direction. This is because that the angular velocity .omega. of the
motor is zero and no influence of an induced voltage is
affected.
[0207] In order to resolve the above task, FIG. 23 embodiment is
provided with the polarity discriminating unit 56 for
discriminating whether the calculated position is N pole or S pole,
namely discriminating polarity of the magnetic pole. Although the
discrimination methods used for the polarity discriminating unit 56
are not limited, a method which uses the magnetic saturation
characteristic of a synchronous motor is effective. Hereinbelow, as
an example of polarity discriminating methods the method of using
the magnetic saturation characteristic will be explained.
[0208] With regard to the magnetic characteristic of a synchronous
motor, since the rotor thereof includes permanent magnets, there
exist magnetic fluxes even when d axis current id representing the
magnetic flux axis is zero as illustrated in FIG. 25. Due to this
magnetic characteristic, the d axis inductance Ld shows the
characteristic as illustrated in FIG. 26. From FIG. 26 it is
understood that there exist a plurality of regions having different
magnitudes depending on differences of positive or negative of d
axis current id, in that regions as indicated by hatchings in FIG.
26. Accordingly, if an AC signal having an biasing component which
causes d axis current id within the hatched regions as illustrated
in FIG. 26, the response in the current control system varies
depending on the magnitude of the inductance Ld and the polarity
difference between N pole and S pole reflects in the amplitude
difference of the motor current, thereby the polarity
discrimination of the magnetic pole is enabled through the
measurement of the magnitude of the motor current amplitude, which
is an example of the polarity discrimination methods.
[0209] Now, the processing sequence of the above polarity
discriminating method is explained with reference to FIG. 23. At
first, from the signal generating unit 54 which generates signals
for detecting the magnetic pole position an AC signal having a DC
biasing component is applied to the d axis current command.
Subsequently, a detected d axis current id is inputted into the
polarity discriminating unit 56 in which an amplitude of the AC
component in the d axis current id is measured and the polarity of
the current position setting value .theta..LAMBDA. is
discriminated.
[0210] If the discrimination result is as N pole, the position
setting value .theta..LAMBDA. for the control system is used as it
is for the control. On the other hand, if the discrimination result
is as S pole, the position setting value .theta..LAMBDA. is
corrected to N pole by adding or subtracting 180.degree. to and
from the position setting value .theta..LAMBDA.. In this polarity
discriminating method a current is flowed in d axis direction to
the extent that magnetic saturation is caused in the synchronous
motor, therefore, if a slight error is caused in the position
setting value .theta..LAMBDA., a torque in a predetermined
direction is induced in the motor. Accordingly, the magnetic pole
position detecting operation during start-up period is performed
under a condition that the rotation of the rotary shaft is
prevented with a provision of a lock mechanism which temporarily
and mechanically prevents rotation of the rotary shaft and the
rotor of the synchronous motor possible caused by the induced
torque. For example, as illustrated in FIG. 21, a lock mechanism 74
is provided which prevents rotation of a rotor 70 or a rotary shaft
72 of the motor 1. The lock mechanism 74 corresponds to a usual
brake mechanism and temporarily prevents their rotation in response
to a signal from the detector unit 52. Further, the lock mechanism
74 is activated during the rotation start-up period, and since the
torque at that period is small, a lock mechanism with a simple
structure can achieve the object concerned. The rotation preventing
operation by the lock mechanism 74 is released by a signal from the
detector unit 52 after completing the detection of the magnetic
pole position and the polarity thereof for starting rotation.
[0211] Now, the processing sequence of detecting the magnetic pole
position at zero motor speed during the motor start-up period which
is performed by the detector unit 52 is explained with reference to
FIG. 27 flow chart. In FIG. 27, letter S implies a step, namely a
processing sequence. At first, in step S 30, it is judged whether
the rotation of the motor rotary shaft is prevented by the lock
mechanism. If the rotation of the rotary shaft is prevented, a
signal idhr used for detecting the magnetic pole position is
outputted from the polarity discriminating unit 56 to apply the
same to the adder 65 for performing adding calculation to the
current command idr1 from the current command value generating unit
6 in step S 31.
[0212] The above explanation relates to the instance in which the
lock mechanism for preventing motor rotation during motor start-up
period, namely rotation start-up period is provided. If no such
lock mechanism is provided, the step S 30 is omitted.
[0213] When the lock mechanism is provided and is not activated, a
command for activating the lock mechanism is issued in step S 30 or
the actuation thereof is waited until a command from another
control unit is issued. When the lock mechanism is activated, the
process moves to step S 31.
[0214] In step S 32, a short circuited current which is induced
during a PWM control is detected and a detection value
.theta..LAMBDA. is calculated through the magnetic pole position
detecting method using the short circuited current variation amount
as has been explained above. Further, in step S 33, it is judged
whether the detected value .theta..LAMBDA. obtained in step S 32
and representing the magnetic pole position is in N pole direction
or S pole direction. When the judgement result reveals as N pole,
the detected value .theta..LAMBDA. is determined corresponding to
the current magnetic pole position in step S 34. Contrary thereto,
when it is judged as S pole in step S 33, the detected value
.theta..LAMBDA. is corrected by adding or subtracting 180.degree.
so as to determine the current magnetic pole position in step S 35.
Thereafter, the motor drive control is initiated in step S 36 by
making use of the obtained detection value of the magnetic pole
position.
[0215] The above is the processing sequence during the motor
start-up period which makes use of the magnetic pole position
detecting method at zero motor rotation speed. If the detection
accuracy of the magnetic pole position only with the magnetic pole
position detecting method using the short circuited current
variation of the motor is insufficient, a highly accurate magnetic
pole position detection can be performed through the above
explained processings. In the present embodiment, although the
locking of the rotary shaft of the motor is determined as the
condition for initiating the detecting operation of the magnetic
pole position, it is possible to detect the magnetic pole position
even if the rotary shaft is not locked as has been explained
previously.
[0216] Now, the operation of the calculating unit 52 is explained
under the condition that the rotating speed of the motor is zero
but the motor current is flowing therethrough. This operating
environment corresponds to the instance that the load torque at the
time of rotation stoppage, namely the motor torque which is
necessary to drive a device to be driven by the motor is larger
than that now being generated by the motor and further the magnetic
pole position setting value in the controller is erased or become
inaccurate due to influences such as noises. In such instance,
since the motor current is flowing, the detection of the magnetic
pole position by making use of the motor short circuited current
variation as has been explained above is possible. However, since
the motor speed is zero, it is necessary to perform the polarity
discrimination. In this instance, since the motor rotary shaft is
not locked and the motor is being driven, the following polarity
discrimination method is more preferable than the polarity
discrimination method which makes use of the magnetic saturation
characteristic as has been explained above.
[0217] The above referred to polarity discrimination method is one
in which the polarity is discriminated from the motor torque
generating direction and the motor shaft rotating direction. The
operation, namely the processing contents of the present
discrimination method will be explained with reference to FIG. 28
flow chart. At first, in step S 40 a predetermined desired torque
is generated by using the detection value .theta..LAMBDA. which is
obtained based on the motor short circuited current variation as
the setting value for the magnetic pole position. Thereafter, in
step S 41 the motor shaft rotating direction and the generated
torque direction are compared, and if the both directions coincide,
the setting value for the magnetic pole position is determined as
the current magnetic pole position in step S 42. On the other hand,
if the motor shaft rotating direction and the generated torque
direction do not coincide each other, it is judged that the
polarity is opposite in step S 41, and in step S 43 the setting
value is corrected by 180.degree. to determine the current magnetic
pole position. Further, in step S 44 the torque command is
increased. The above operation is repeated up to a predetermined
number of times and if the number of coincidence of the motor shaft
rotating direction and the generated torque direction exceeds
successively the predetermined number of times, the polarity
discrimination is ended. The above is the processing sequence of
the polarity discrimination method of the setting value for the
magnetic pole position based on the torque generating direction and
the motor rotation shaft rotating direction. In the present method,
an application of specific signals used for polarity discrimination
is unnecessary.
[0218] In the above, the magnetic pole position detecting method
when motor speed is zero is explained which incorporates the
magnetic pole position detecting method making use of the motor
short circuited current variation amount. The present method can be
applied when the motor speed is extremely low so that an induced
voltage thereby is also extremely low. For example, the present
method is effective for a rotating speed less than 800 rpm.
[0219] Further, the present method can be applied as it is to a
driving device for an electric vehicle including an electric motor
vehicle and a hybrid car having both a motor and an engine. In the
driving device for an electric motor vehicle and a hybrid car, if
the detection through application of a signal used for position
detection or the polarity discrimination making use of the magnetic
saturation characteristic is performed under the condition that a
brake therefor is activated or the operation range is set at the
parking range, vibration or displacement of the vehicle due to
unnecessary torque generation is avoided.
[0220] According to the above embodiments the magnetic pole
position detection is enabled over the entire operating range of a
synchronous motor without being affected by the applied voltage
states while performing a usual PWM control with a low cost
controller.
[0221] Still further, according to the embodiment as explained in
connection with FIGS. 21 through 28, an accurate control can be
performed from the motor stand still period.
[0222] Still further, according to the embodiments as explained in
connection with FIGS. 1 through 28, since the magnetic pole
position of a synchronous motor is determined based on the motor
current variation amount or the variation direction thereof, the
magnetic pole position detection can be realized without providing
a position detector while performing a usual PWM control.
[0223] Moreover, when a magnetic pole detector is provided, through
the comparison between the magnetic pole position detected by the
magnetic pole detector and the magnetic pole position determined by
the motor current an abnormality of the magnetic pole position
detector can be sensed while performing a usual PWM control.
[0224] According to the present invention, a low cost synchronous
motor driving system with an excellent control performance can be
provided.
[0225] Further, according to the present invention, in case when a
usual mechanical magnetic pole position sensor is used, an
abnormality of such sensor is detected, and a highly reliable
synchronous motor driving system can be provided.
* * * * *