U.S. patent application number 09/444032 was filed with the patent office on 2002-02-07 for isolated dual converter having primary side internal feedback for output regulation.
Invention is credited to TELEFUS, MARK D..
Application Number | 20020015315 09/444032 |
Document ID | / |
Family ID | 23763211 |
Filed Date | 2002-02-07 |
United States Patent
Application |
20020015315 |
Kind Code |
A1 |
TELEFUS, MARK D. |
February 7, 2002 |
ISOLATED DUAL CONVERTER HAVING PRIMARY SIDE INTERNAL FEEDBACK FOR
OUTPUT REGULATION
Abstract
The present invention comprises an isolated dual power supply
having an internal feedback loop. The dual power supply comprises a
modulated switched power converter generating an internal regulated
voltage output coupled to an alternating current tank. The
alternating current tank has a first and a second power switch that
are alternately switched ON at a 50% duty cycle. When the first
switch is ON, the internal regulated voltage output is coupled to a
storage capacitor and a primary winding of a transformer in the
alternating current tank wherein the storage capacitor is charged
and a current flows in a first direction through the primary
winding. Alternatively, when the second switch is ON, the charged
storage capacitor discharges and a current flows in a second
direction opposite to the first direction through the primary
winding. In this fashion, the transformer becomes a linear device,
allowing the use of internal feedback to control the external
output of the transformer.
Inventors: |
TELEFUS, MARK D.; (Orinda,
CA) |
Correspondence
Address: |
LYON & LYON LLP
633 WEST FIFTH STREET
SUITE 4700
LOS ANGELES
CA
90071
US
|
Family ID: |
23763211 |
Appl. No.: |
09/444032 |
Filed: |
November 19, 1999 |
Current U.S.
Class: |
363/15 |
Current CPC
Class: |
H02M 1/008 20210501;
H02M 3/3376 20130101; H02M 1/007 20210501 |
Class at
Publication: |
363/15 |
International
Class: |
H02J 001/00 |
Claims
I claim:
1. A power converter, comprising: a modulated switching power
converter having a power switch for regulating an internal voltage
output; and an alternating current tank coupled to the internal
voltage output, the alternating current tank having a storage
capacitor and a transformer having a primary winding, a first
switch that, when ON, couples the storage capacitor to the internal
voltage output wherein the storage capacitor is charged and a first
current flows in a first direction through the primary winding, a
second switch that, when ON, permits the charged storage capacitor
to discharge, wherein a second current flows in a second direction
opposite to the first direction through the primary winding, and
50% duty cycle control circuitry for alternately switching ON and
OFF the first and second switches, the first switch being ON when
the second switch is OFF and the second switch being ON when the
first switch is OFF, the 50% duty cycle control circuitry
alternately switching the first and second switches such that the
ON and OFF times of each switch are substantially equal, whereby a
voltage produced across the primary winding is linearly related to
a voltage produced across a secondary winding of the
transformer.
2. The power converter of claim 1, wherein the modulated switching
power converter is a pulse width modulated boost power
converter.
3. The power converter of claim 2, wherein the storage capacitor is
in series with the primary winding.
4. The power converter of claim 3, wherein the storage capacitor is
a resonant capacitor, the resonant capacitor and a leakage
inductance of the primary winding forming a series resonant tank
having a resonant frequency less than or equal to a period of the
50% duty cycle control circuitry, whereby the first and second
currents are each half-wave quasi-sinusoidal currents.
5. The power converter of claim 4, wherein the voltage across the
secondary winding is regulated by adjusting the pulse width
modulation of the power switch in response to sensing a peak of the
first and second currents.
6. The power converter of claim 5, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the internal
voltage output.
7. The power converter of claim 6, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the voltage
across the secondary winding through an isolator.
8. The power converter of claim 7, wherein the isolator is an
optoisolator.
9. The power converter of claim 2, wherein the storage capacitor is
a resonant capacitor in parallel with the primary winding, the
resonant capacitor and a leakage inductance of the primary winding
forming a parallel resonant tank having a resonant frequency less
than or equal to a period of the 50% duty cycle control circuitry,
whereby when the first switch is ON, a first half-wave
quasi-sinusoidal voltage is impressed across the primary winding,
and when the second switch is ON, a second half-wave
quasi-sinusoidal voltage is impressed across the primary winding,
the second half-wave quasi-sinusoidal voltage being antipodal to
the first half-wave quasi-sinusoidal voltage.
10. The power converter of claim 9, wherein the voltage across the
secondary winding is regulated by adjusting the pulse width
modulation of the power switch in response to sensing the first and
second currents.
11. The power converter of claim 1, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the internal
voltage output.
12. The power converter of claim 11, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the voltage
across the secondary winding through an isolator.
13. The power converter of claim 12, wherein the isolator is an
optoisolator.
14. The power converter of claim 1, wherein the pulse width
modulated switching power converter is a buck power converter.
15. The power converter of claim 14, wherein the storage capacitor
is in series with the primary winding.
16. The power converter of claim 15, wherein the storage capacitor
is a resonant capacitor, the resonant capacitor and a leakage
inductance of the primary winding forming a series resonant tank
having a resonant frequency less than or equal to a period of the
50% duty cycle control circuitry, whereby the first and second
currents are each half-wave quasi-sinusoidal currents.
17. The power converter of claim 16, wherein the voltage across the
secondary winding is regulated by adjusting the pulse width
modulation of the power switch in response to sensing a peak of the
first and second currents.
18. The power converter of claim 17, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the internal
voltage output.
19. The power converter of claim 18, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the voltage
across the secondary winding through an isolator.
20. The power converter of claim 19 wherein the isolator is an
optoisolator.
21. The power converter of claim 14, wherein the storage capacitor
is a resonant capacitor in parallel with the primary winding, the
resonant capacitor and a leakage inductance of the primary winding
forming a parallel resonant tank having a resonant frequency less
than or equal to a period of the 50% duty cycle control circuitry,
whereby when the first switch is ON, a first half-wave
quasi-sinusoidal voltage is impressed across the primary winding,
and when the second switch is ON, a second half-wave
quasi-sinusoidal voltage is impressed across the primary winding,
the second half-wave quasi-sinusoidal voltage being antipodal to
the first half-wave quasi-sinusoidal voltage.
22. The power converter of claim 21, wherein the voltage across the
secondary winding is regulated by adjusting the pulse width
modulation of the power switch in response to sensing the first and
second currents
23. The power converter of claim 22, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the internal
voltage output.
24. The power converter of claim 23, wherein the voltage across the
secondary winding is further regulated by adjusting the pulse width
modulation of the power switch in response to sensing the voltage
across the secondary winding through an isolator.
25. The power converter of claim 24, wherein the isolator is an
optoisolator.
26. A circuit for use in a power supply, comprising a power switch
coupled to a voltage input; a pulse width modulator for controlling
a duty cycle of the power switch; a diode having its anode coupled
to the voltage input; a first switch coupled to the cathode of the
diode and to a voltage output; a second switch coupled to the
voltage output; and control circuitry for alternately switching ON
and OFF the first and second switches, the first switch being ON
when the second switch is OFF and vice versa, the control circuitry
alternately switching the first and second switches such that the
ON and OFF times of each switch are substantially equal.
27. The circuit of claim 26, wherein the power switch, the diode,
the pulse width modulator, the first switch, the second switch and
the control circuitry are integrated on the circuit.
28. A power converter, comprising: a modulated switching power
converter having a power switch for regulating an internal voltage
output; a plurality of alternating current tanks coupled to the
internal voltage output, each alternating current tank in the
plurality of alternating current tanks having a storage capacitor
and a transformer having a primary winding, a first switch that,
when ON, couples the storage capacitor to the internal voltage
output wherein the storage capacitor is charged and a first current
flows in a first direction through the primary winding, a second
switch that, when ON, permits the charged storage capacitor to
discharge, wherein a second current flows in a second direction
opposite to the first direction through the primary winding, and
50% duty cycle control circuitry for alternately switching ON and
OFF the first and second switches, the first switch being ON when
the second switch is OFF and the second switch being ON when the
first switch is OFF, the 50% duty cycle control circuitry
alternately switching the first and second switches such that the
ON and OFF times of each switch are substantially equal, whereby a
voltage produced across the primary winding is linearly related to
a voltage produced across a secondary winding of the transformer;
and a clock coupled to the plurality of the alternating current
tanks, wherein the first switches are switched synchronously with
each other and the second switches are switched synchronously with
each other.
29. The power converter of claim 28, wherein the modulated
switching power converter is a boost power converter.
30. The power converter of claim 29, wherein within each of the
alternating current tanks, the storage capacitor is in series with
the primary winding.
31. The power converter of claim 30, wherein within each of the
alternating current tanks, the storage capacitor is a resonant
capacitor, the resonant capacitor and a leakage inductance of the
primary winding forming a series resonant tank having a resonant
frequency less than or equal to a period of the 50% duty cycle
control circuitry, whereby the first and second currents are each
half-wave quasi-sinusoidal currents.
32. The power converter of claim 29, wherein within each of the
alternating current tanks, the storage capacitor is a resonant
capacitor in parallel with the primary winding, the resonant
capacitor and a leakage inductance of the primary winding forming a
parallel resonant tank having a resonant frequency less than or
equal to a period of the 50% duty cycle control circuitry, whereby
when the first switch is ON, a first half-wave quasi-sinusoidal
voltage is impressed across the primary winding, and when the
second switch is ON, a second half-wave quasi-sinusoidal voltage is
impressed across the primary winding, the second half-wave
quasi-sinusoidal voltage being antipodal to the first half-wave
quasi-sinusoidal voltage.
33. The power converter of claim 28, wherein the modulated
switching power converter is a pulse width modulated buck power
converter.
34. The power converter of claim 33, wherein within each of the
alternating current tanks, the storage capacitor is in series with
the primary winding.
35. The power converter of claim 34, wherein within each of the
alternating current tanks, the storage capacitor is a resonant
capacitor, the resonant capacitor and a leakage inductance of the
primary winding forming a series resonant tank having a resonant
frequency less than or equal to a period of the 50% duty cycle
control circuitry, whereby the first and second currents are each
half-wave quasi-sinusoidal currents.
36. The power converter of claim 33, wherein within each of the
alternating current tanks, the storage capacitor is a resonant
capacitor in parallel with the primary winding, the resonant
capacitor and a leakage inductance of the primary winding forming a
parallel resonant tank having a resonant frequency less than or
equal to a period of the 50% duty cycle control circuitry, whereby
when the first switch is ON, a first half-wave quasi-sinusoidal
voltage is impressed across the primary winding, and when the
second switch is ON, a second half-wave quasi-sinusoidal voltage is
impressed across the primary winding, the second half-wave
quasi-sinusoidal voltage being antipodal to the first half-wave
quasi-sinusoidal voltage.
37. A method of generating power, comprising: generating an
internal regulated voltage output through a modulated switching
power converter; and then coupling a storage capacitor and a
primary winding of a transformer to the internal regulated voltage
output during a first period, wherein the storage capacitor is
charged and a current flows in a first direction through the
primary winding; and then uncoupling the charged storage capacitor
and the primary winding from the internal regulated voltage output
during a second period, wherein the charged storage capacitor
discharges and a current flows in a second direction opposite that
of the first direction through the primary winding, the first
period being equal to the second period.
38. The method of claim 37, wherein the modulated switching power
converter is a pulse width modulated power converter.
39. The method of claim 38, wherein the storage capacitor and the
primary winding are in series.
40. The method of claim 39, wherein the storage capacitor and the
primary winding are in parallel.
41. The method of claim 37, wherein the modulated switching power
is a pulse width modulated buck converter.
42. The method of claim 41, wherein the storage capacitor and the
primary winding are in series.
43. The method of claim 42, wherein the storage capacitor and the
primary winding are in parallel.
Description
[0001] This invention pertains generally to the field of power
conversion and more particularly to a pulse width modulated
switching power supply with linear feedback control.
BACKGROUND OF THE INVENTION
[0002] Compact and efficient power supplies are an increasing
concern to users and manufacturers of electronics. Pulse width
modulated (PWM) switching power supplies offer both compactness and
efficiency in a number of different topologies. Boost and buck PWM
switching power supply topologies are efficient, but do not isolate
the power input from the power output. Other topologies, such as
the flyback, do isolate the power input from the power output by
using a transformer. In such topologies, however, feedback from the
secondary (power output) side of the transformer is required to
adjust the pulse width modulation of the power switch. To properly
compensate the feedback from the secondary requires extra
components and often involves expensive re-design, depending upon
the particular application.
[0003] In contrast to a PWM switching power supply, soft-switched
converters possess resonant elements to generate sinusoidally
varying resonant voltages and/or currents that help reduce
switching losses. Notably, in a particular form of soft-switched
converter, a resonant transition converter, LC elements coupled to
two power switches that turn on and off only at either zero current
states or zero voltage states during a sinusoidally varying
resonant current or voltage waveform minimizes switching stress and
loss. In general, the behavior of these resonant waveforms depends
on the values of the inductance and capacitance within the resonant
tank as well as values of the DC input and output voltages.
Accordingly, considerable research has been conducted on the
relationship between these values and the resonant waveforms.
Researchers have discovered that to maintain a constant output
voltage independent of the output current from such series or
parallel resonant converters requires, for example, frequency
modulation of the switching elements within the resonant
converters. See, e.g., U.S. Pat. Nos. 4,796,173, 5,448466,
4,017,784, 4,727,469, and 4,757,432. However, because of the
nonlinear loading within a resonant tank, analysis and design of
the feedback control systems for these converters is difficult and
cumbersome.
[0004] There is a need in the art for an improved PWM switching
power supply that combines the simplicity and ease of design
provided by a PWM switching power supply yet provides the stress
and loss advantages of a resonant converter without requiring
adjustment, such as through frequency modulation, of the properties
of the resonant converter. Further, there is a need in the art for
a PWM switching power supply that isolates the input and outputs
through a transformer without requiring feedback from the secondary
side of the transformer, thereby easing design and reducing the
component count. In addition, there is a need in the art for an
improved resonant converter and methods of controlling such
converters that avoid the complexity of prior art methods involving
frequency control, magnetic flux control, or impedance adjustment
of resonant converters. The present invention addresses these needs
by providing, in one embodiment, a resonant converter whose output
is regulated by a DC input voltage that is in turn is adjusted
accordingly by the output current or voltage status.
SUMMARY OF THE INVENTION
[0005] In accordance with one aspect of the present invention, a
power converter comprising a PWM switching power supply coupled to
an alternating current tank is provided. The PWM switching boost
power supply includes a power switch for regulating an internal
voltage output. The alternating current tank comprises a first and
a second switch, control circuitry for controlling the first and
second switches, a transformer having a primary winding and a
secondary winding, and a storage capacitor. The control circuitry
alternatively switches the first switch ON when the second switch
is OFF and switches the second switch ON when the first resonant
switch is OFF, wherein the control circuitry alternatively switches
the first and second resonant switches such that the ON and OFF
times of each switch are substantially equal. When the first switch
is ON, the storage capacitor is coupled to the internal voltage
output such that the storage capacitor is charged and a current
flows in a first direction through the primary winding.
Alternatively, when the second switch is ON, the charged storage
capacitor discharges such that a current flows in a second
direction, opposite to the first direction, through the primary
winding.
[0006] Because the ON and OFF times of the first and second
switches are substantially equal, an output voltage produced by the
secondary winding is linearly related to the internal output
voltage and the current through the primary. By sensing the current
through the primary and adjusting a duty cycle of the power switch
accordingly, the present invention regulates the output voltage
without the need for a feedback loop from the isolated secondary.
In addition, the duty cycle of the power switch may be adjusted in
response to directly sensing the internal output voltage through a
voltage divider or the like.
[0007] In accordance with another aspect of the invention, the PWM
switching power supply may be a boost converter, a buck converter,
or a buck/boost converter. The storage capacitor of the alternating
current tank may be a resonant capacitor either in series or in
parallel with a leakage inductance of the primary, forming a series
resonant tank or a parallel resonant tank, respectively.
[0008] In accordance with a still further aspect of the invention,
a power converter comprising a modulated switching power supply
having a power switch for regulating an internal voltage output
coupled to a plurality of alternating current tanks is provided.
The storage capacitor in each of the alternating current tanks may
be a resonant capacitor either in series or parallel with a leakage
inductance of the primary winding, forming a series resonant tank
or a parallel resonant tank, respectively, as described herein. A
clock coupled to the plurality of alternating current tanks permits
the first and second switches to be switched synchronously with
each other. The output voltage from each secondary winding is
combined in parallel for application to a load. An intelligent
switch may be coupled between the modulated switching power supply
and the plurality of alternating current tanks wherein a given
alternating current tank is coupled to the internal voltage output
through the intelligent switch only when required to support a
required voltage across the load.
[0009] In accordance with a still further aspect of the present
invention, methods of generating DC or AC power are provided. In
one embodiment, the method comprises generating an internal
regulated voltage output using a modulated switching power supply.
The internal regulated voltage output is coupled to a storage
capacitor and a primary winding of a transformer wherein the
storage capacitor is charged and a first current flows in a first
direction through the primary winding during a first period. The
internal regulated voltage output is then decoupled from the
storage capacitor and the primary winding wherein the charged
storage capacitor discharges and a second current flows in a second
direction opposite the first direction through the primary winding
during a second period, the first period being equal to the second
period.
[0010] Other aspects and advantages of the present invention are
disclosed by the following description and figures.
DESCRIPTION OF FIGURES
[0011] The various aspects and features of the present invention
may be better understood by examining the following figures:
[0012] FIG. 1 illustrates a modulated switching power supply
coupled to an alternating current tank according to one embodiment
of the invention.
[0013] FIG. 2 illustrates a prior art PWM boost power
converter.
[0014] FIG. 3 is illustrates an alternating current tank in a
series resonant tank configuration according to one embodiment of
the invention.
[0015] FIG. 4 is a graphical representation of the switch signals
and the quasi-sinusoidal resonant currents in the circuit of FIG.
3.
[0016] FIG. 5 is a schematic diagram of a power supply comprising a
boost power converter coupled to an alternating current tank in a
series resonant configuration according to one embodiment of the
invention.
[0017] FIG. 6 is a schematic diagram of the power supply of FIG. 5,
wherein portions of the power supply are incorporated into an
integrated circuit.
[0018] FIG. 7 is a schematic diagram illustrating the integrated
circuit of FIG. 6, incorporated into an AC input power supply.
[0019] FIG. 8 is a block diagram illustrating a modulated switched
power converter coupled to a plurality of alternating current tanks
according to one embodiment of the invention.
[0020] FIG. 9 illustrates a prior art PWM buck power converter.
[0021] FIG. 10 is a schematic diagram of an alternating current
tank in a parallel resonant tank configuration according to one
embodiment of the invention.
DETAILED DESCRIPTION
[0022] Turning now to FIG. 1, one embodiment of the power converter
1 of the present invention is illustrated. In this embodiment, a
modulated switching power converter 5 couples to an alternating
current tank 20. Because of these two stages, the resulting power
converter 1 has a "dual converter" topology. The power converter 1
receives an unregulated DC voltage input, V.sub.in, which may be
generated by a rectifier 7 operating on an unregulated AC input, to
generate a regulated AC or DC output voltage, V.sub.O.
[0023] Within the power converter 1, the modulated switching power
converter 5 receives the unregulated DC input voltage, V.sub.in,
and produces an internal regulated voltage output, V.sub.int.
Because this regulated voltage output V.sub.int is internal to the
power converter 1, it may be considered "preregulated" as compared
to the regulated output voltage, V.sub.O. As used herein, a
"modulated" switching power converter may include any suitable form
known in the art, for example, a pulse width modulated (PWM)
switched power converter. Because PWM switching power converters
have desirable output ripple and noise properties, they will be
illustrated in the following embodiments of the invention. That is
not to imply, however, that other types of modulated switching
power converters such as frequency modulated switching power
converters could not be used to generate the internal regulated
voltage output.
[0024] An alternating current tank 20 couples to V.sub.int to
generate an isolated voltage output, V.sub.O, through a transformer
(illustrated in FIG. 3). As will be described herein, because the
of the unique configuration of the alternating current tank 20, the
transformer becomes a linear device, permitting the use of a
strictly internal feedback loop within the alternating current tank
20, if desired.
[0025] The modulated switching power converter 5 may be of any type
known in the art such as a boost, buck, buck/boost, flyback,
half-bridge, forward, push-pull, or full-bridge switching power
converter. Turning now to FIG. 2, a boost converter 6 is
illustrated. The boost converter 6 comprises a power switch Q1
(typically a field effect transistor (FET)) coupled to a boost
inductor 8, a boost diode 10, and a storage capacitor 12. A pulse
width modulator 14 adjusts a duty cycle of the power switch Q1 in
response to sensing an internal output voltage, V.sub.int. The
relationship between the input voltage, V.sub.in, and V.sub.int may
be approximated as
V.sub.int=V.sub.in*(T/t.sub.off)
[0026] where T is the switching period and t.sub.off is the off
time of the power switch Q1. Inspection of FIG. 2 reveals that the
ground of V.sub.int is not isolated from that of V.sub.in.
[0027] Referring now to FIG. 3, in the present invention, an
alternating current tank 20 couples to V.sub.int to generate an
output voltage that is isolated from V.sub.in. The alternating
current tank 20 includes a storage capacitor 34 that is either in
series or in parallel with the primary winding 22 of a transformer
24. A first and a second switch (illustrated here as switches Q2
and Q3, respectively) couple to the primary winding 22 and storage
capacitor 34. As will be further explained, the alternating current
tank 20 generates an alternating current through the primary
winding 22 by switching switches Q2 and Q3 ON and OFF at a 50% duty
cycle. The alternating current (AC) thus induced through the
secondary winding 26 may be rectified or used as AC. In the
embodiment of the alternating current tank illustrated in FIG. 3, a
rectifier 27 on the secondary side of the transformer rectifies the
current through the load. The rectifier 27 may be either a
full-wave or half-wave rectifier as is known in the art. In one
embodiment, the rectifier 27 comprises a center tapped secondary
winding 26 coupled to diodes 29 and 28 and output capacitor 30 to
form a full wave rectifier such that current is unidirectional
through the load.
[0028] As will be described herein, the alternating current tank 20
may be in either a series resonant tank or a parallel resonant tank
configuration. Such configurations have the storage capacitor 34
and the primary winding 22 in series or parallel, respectively, as
described above. However, in such configurations, the value of a
capacitance of the storage capacitor 34, a leakage inductance 36 of
the primary winding 22, and the period of the 50% duty cycle used
to operate switches Q2 and Q3 are such that resonant waveforms are
generated. Because these configurations permit zero-transition
switching of switches Q2 and Q3, which reduces stress and loss,
they will be described with respect to the series and parallel
embodiments of the alternating current tanks described herein. That
is not to imply, however, that a non-resonant alternating current
tank 20 is not included within the scope of the invention.
[0029] The operation of the alternating current tank 20 of FIG. 3
in a series resonant tank configuration occurs as follows. 50% duty
cycle control circuitry 32 controls a first resonant switch Q2 and
a second resonant switch Q3 such that when Q2 is ON, Q3 is OFF.
Conversely, control circuitry 32 turns Q2 OFF when Q3 is ON. When
Q2 is on, the internal output voltage is coupled to the
series-connected storage capacitor 34 and primary winding 22,
thereby charging the storage capacitor 34 and inducing a half-wave
quasi-sinusoidal current in a first direction through the primary
winding 22 and the resonant tank circuit formed by the storage
capacitor 34 and the leakage inductance 36 of the primary winding
(represented separately from the primary winding for illustration
purposes). Conversely, when Q3 is on, series-connected storage
capacitor 34 and the primary winding 22 are uncoupled from the
internal voltage output such that the charged storage capacitor 34
discharges and a half-wave quasi-sinusoidal current is induced in a
second direction, opposite to that of the first direction, through
the primary winding 22 and the resonant tank circuit formed by the
storage capacitor 34 and the linkage inductance 36.
[0030] The control circuitry 32 operates the resonant switches Q2
and Q3 at substantially a 50% duty cycle such that the ON time
equals the OFF time of each resonant switch. Referring now to FIG.
4, the relationship between the ON and OFF times of Q2 and Q3, the
full-wave quasi-sinusoidal current induced in the primary winding
22, I.sub.PR, and the voltage, Vc, across the storage capacitor 34
is illustrated. As can be seen from inspection of FIG. 4, the
resonant switches Q2 and Q3 are turned ON and OFF when the current
I.sub.PR is zero, hence the denotation of a "zero-current" resonant
converter. In this fashion, switching losses are minimized.
Moreover, because the ON and OFF times of each resonant switch are
equal, the primary winding 22 is effective excited by a full-wave
sinusoidal current. During the time Q2 is ON, the voltage, Vc,
across the storage capacitor charges to a maximum value. During the
time Q3 is ON, the voltage V.sub.c discharges to zero. Note that
there will be ordinarily some dead time (not illustrated) wherein
Q2 has turned OFF but Q3 has not yet turned ON. In addition, the
resonant frequency of the series resonant tank formed by the
storage capacitor and the leakage inductance of the primary winding
must be chosen such that the half-wave sinusoidal current waveform
can be completed during the time when the resonant switches are ON.
Thus, the current I.sub.PR is not a true full-wave sinusoid but
rather a full-wave quasi-sinusoid. Nevertheless, the departure of
I.sub.PR from a true sinusoidal wave is minimal.
[0031] The period T.sub.p of the 50% duty cycle for each of the
switches Q2 and Q3 is controlled the control circuitry 32. In one
embodiment, control circuitry 32 may comprise a high voltage half
bridge driver with oscillator, model L6571A or L6571B from
SGS-Thomson Microelectronics. The period T.sub.p determined by the
control circuitry must be related to the period of the
quasi-sinusoidal resonant current. As can be seen from inspection
of FIG. 4, T.sub.p must be greater than the period of the
quasi-sinusoidal resonant current (I.sub.PR) of the series resonant
tank circuit formed by the linkage inductance 36 and the storage
capacitor 34 so that each half cycle of the resonant current may
finish during the ON time of its corresponding switch Q2 or Q3. For
example, during the time Q2 is ON, I.sub.PR must cycle from zero,
through a maximum, and back to zero again. If the resonant
frequency f.sub.r of the tank circuit was too slow, the
quasi-sinusoidal resonant current I.sub.PR would not be able to
finish a half cycle during this time.
[0032] Consider the following example. If T.sub.p is set at 20
.mu.s, then each half period (i.e., the time Q2 or Q3 is ON) is 10
.mu.s. Thus, to assure completion of a half wave of the resonant
current I.sub.PR, the resonant half period should be less than this
time, for example 8 .mu.s. Such a half period gives a resonant
frequency f.sub.r of 55.6 KHZ. For a series (and also a parallel)
resonant tank circuit, the resonant frequency (in Hz) is given
by
f.sub.r=1/(2.pi.*sqrt(L.sub.RC.sub.R)
[0033] where L.sub.R is the value of the leakage inductance and
C.sub.R is the value of the resonant capacitance. Inspection of
this equation indicates that to increase the resonant frequency
f.sub.r, the value of the (in this case, resonant) storage
capacitor C.sub.R should be minimized. This leads to a design
choice, because the output power of the primary winding 22 may be
approximated as
P.sub.primary=(C.sub.R*V.sup.2*f.sub.r)/2
[0034] where V is the voltage across the primary winding 22 and
P.sub.primary is the output power of the primary winding 22. Note
that the contribution to P.sub.primary from the leakage inductance
may be neglected due because the leakage inductance is typically
quite small compared to the mutual inductance of the primary
winding. Thus, if the mutual inductance is a few milliH, the
leakage inductance will be a few .mu.H.
[0035] Inspection of the equation for P.sub.primary reveals that
the output power is increased if the value of the resonant storage
capacitor C.sub.R is increased. However, if the value of C.sub.R is
increased too much, then f.sub.r will be too slow to allow the
resonant current to complete a half cycle during the times when
either Q2 or Q3 is ON. Thus, tradeoffs should be made between the
switching period T.sub.p, the resonant (storage) capacitance and
the desired output power.
[0036] It may be shown that the voltage output of the full wave
rectifier 27 of FIG. 3, V.sub.O, is approximated by
V.sub.O=V.sub.int/2N
[0037] where N is the turn ratio between the primary winding 22 and
the secondary winding 26. It follows that the resonant current
through the primary, I.sub.PR, may be approximated by
I.sub.PR=(V-N V.sub.O-(I.sub.O*ESR/N))/Z.sub.O;
Z.sub.O=sqrt(L.sub.R/C.sub- .R)
[0038] where V is the voltage impressed across the series-connected
resonant tank circuit formed by the storage capacitor 34 (C.sub.R)
and leakage inductance 36 (L.sub.R) of the primary winding 22,
I.sub.O is the current through the load, ESR is the equivalent
series resistance seen by the secondary winding 26, and Z.sub.O is
the impedance of the resonant tank circuit formed by the leakage
impedance 36 (L.sub.R) and the storage capacitor 34 (C.sub.R).
Thus, I.sub.PR will have a peak absolute value that is also
linearly related to V.sub.int. This demonstrates one of the
advantages of the present invention--i.e., no external feedback is
necessary from the secondary side 26 of the transformer 24.
Instead, a single internal (primary side) feedback loop may be
utilized because the peak values of I.sub.PR are linearly related
to the output voltage. Thus, the often onerous task of compensating
feedback from the secondary side of the transformer may be
eliminated.
[0039] The internal feedback loop of the present invention wherein
the PWM modulated switching power converter is a boost power
converter and the alternating current tank is in a series resonant
tank configuration may be implemented as shown in FIG. 5. In this
embodiment of the invention, the peak current through the primary
winding, I.sub.peak is sensed by coupling the voltage across the
sense resistor 40 to an error amplifier 45. The switches Q1, Q2,
and Q3 are implemented through semiconductor FET transistors. Note
that in an alternate embodiment, the sense resistor 40 could have
been placed in series with the primary winding 22, such that
I.sub.peak could be sensed in each half cycle of the
quasi-sinusoidal current flowing through the primary winding 22. In
such an embodiment, however, the sensed voltage would be bipolar,
alternating in polarity with each half cycle. To use a conventional
error amplifier 45 with this bipolar signal would require
rectification. Thus, it is preferred to sense I.sub.peak only in
the half cycle when switch Q3 is ON by placing the sense resistor
in series with switch Q3.
[0040] The voltage across the sensing resistor 40 is coupled to the
differential amplifier 45 (error amplifier) where it is compared to
a reference voltage V.sub.R and produces an error signal for
inputting to the pulse width modulator 14. In one embodiment, the
pulse width modulator 14 would receive only this error signal to
control the output voltage V.sub.O across the load, or
equivalently, the current through the load. Such embodiments would
thus have only a current feedback control loop. This is sufficient
because of the linear relationship between I.sub.peak and V.sub.O.
However, as illustrated in the prior boost converter of FIG. 2, the
pulse width modulation may also be varied by directly sensing the
internal voltage output V.sub.int, thereby providing a voltage
feedback control loop. Thus, in the embodiment of FIG. 5, V.sub.int
is sensed through the voltage divider formed by resistors 50 and
55. The sensed internal voltage and the error signal from the
differential amplifier 45 are summed in summing circuit 60 and the
combined signal forms the input for the pulse width modulator 14.
Notably, both voltage and current feedback are used in the internal
feedback loop of FIG. 5: I.sub.peak from the primary and the sensed
voltage from V.sub.int. Such an arrangement provides an
advantageous degree of control over the output voltage V.sub.O.
However, either form of feedback may be used alone. Thus, the pulse
width modulator could be responsive only to the sensed internal
voltage as in the prior art boost converter of FIG. 2.
[0041] As can be seen from the equation for the primary current,
I.sub.PR, given herein, the effects of load losses are reflected in
the value of I.sub.PR through the (I.sub.O*ESR/N) term. Thus,
sensing the primary current and adjusting the PWM accordingly does
provide load compensation. In the circuit of FIG. 5, load
compensation will not occur unless rises above V.sub.R. Prior to
this point, the internal voltage output, V.sub.int, is controlled
by the voltage feedback provided through the voltage divider formed
by resistors 55 and 50. When I.sub.PR rises such that V.sub.R is
exceeded across the sensing resistor 40, the differential amplifier
45 will produce an output signal. This signal will dominate over
that produced by the voltage divider such that the PWM modulator 14
is adjusted largely by just the current feedback.
[0042] Moreover, the present invention does not exclude the use of
an external feedback loop coupled through the use of optoisolators,
or other isolation means, as implemented in conventional flyback
converters and the like. Indeed, an embodiment of the present
invention may have solely an external feedback loop as is known in
the art and still possess advantageous properties because of the
efficiencies inherent when an alternating current tank couples to a
PWM switched converter. Regardless of the type of feedback, the
characteristics of the resonant tank circuit within the alternating
current tank 20 remains constant: no adjustment in the switching
speed of Q2 and Q3 need be made. Thus, unlike prior art resonant
converters, power factor correction and regulation of the
alternating current tank 20 is controlled through adjusting the PWM
of the switching power supply 5, not by internal adjustments of the
resonant tank.
[0043] Portions of the circuit of FIG. 5 may be packaged into a
single integrated circuit. Turning now to FIG. 6, in one embodiment
of the invention, an integrated circuit 70 having six leads is
illustrated. Contained within integrated circuit 70 are the boost
converter diode 10, the three switching transistors Q1, Q2, and Q3,
the pulse width modulator 14, the control circuitry 32, the error
amplifier 45, the summer 60, the sensing resistor 40, and the
voltage divider formed by resistors 55 and 50 all coupled as
described with respect to FIG. 5. Components that are difficult to
construct on a semiconductor substrate may be located externally to
the integrated circuit 70.
[0044] The boost inductor 8 is external to the circuit 70 and
couples to a Vin lead. The boost diode 10 couples to the external
boost capacitor 12 through a Vboost lead. A Vout lead allows the
resonant switches Q2 and Q3 to couple the resonant current I.sub.PR
to the externally located storage capacitor 34 and transformer 24.
The transformer 24 has a primary winding 22 and secondary winding
24, as discussed with respect to FIGS. 2 and 5. It is to be noted
that resonant switch Q2 couples to the high internal voltage within
the integrated circuit 70. Thus, a high voltage capacitor 75
coupled to V.sub.out and to the control circuitry 32 through a lead
Vhigh allows the control circuitry to efficiently switch on the
resonant switch Q2. A V.sub.cc lead and a ground lead (Gnd)
complete the integrated circuit 70.
[0045] As shown in FIG. 6, a conventional V.sub.cc generation
circuit 80 couples to the V.sub.in lead. Although those of ordinary
skill in the art will appreciate that many configurations could be
used for this circuit, V.sub.cc generation circuit 80 is shown
comprising the zener diodes 90, the capacitors 95 and 96 and diode
100. Such a circuit provides a dependable voltage V.sub.cc for use
in powering the components of the integrated circuit 70. Moreover,
Vcc generation circuit 80 provides a convenient means to control
V.sub.R. It can be shown that the amount of current drawn into the
V.sub.cc input pin will depend upon the value of the capacitor 95.
The higher the value of capacitor 95, the greater the amount of
current flowing in the V.sub.cc input pin. Sensing circuitry (not
illustrated) within the integrated circuit 70 senses a voltage
produced by the current through the V.sub.CC input pin. A voltage
inverter (not illustrated) within the integrated circuit 70 inverts
the sensed voltage to produce V.sub.R. Thus, the value of is
V.sub.R is inversely proportional to the value of capacitor 95.
[0046] V.sub.CC is distributed to the components needing power in
the integrated circuit 70 using a conventional network (not
illustrated). Note that the integrated circuit 70 includes both
types of feedback inputs discussed previously, i.e., the peak
resonant current I.sub.peak through sensing resistor 40, and the
internal voltage output V.sub.int through the voltage divider
formed by resistors 50 and 55. In alternate embodiments of the
integrated circuit 70, only one of these feedback inputs could be
utilized to affect the pulse width modulator 14. In addition,
integrated circuit 70 could be modified to accept feedback from the
external side of the transformer 24, as discussed with respect to
conventional flyback power converters.
[0047] Referring now to FIG. 7, the integrated circuit 70 of FIG. 6
may be implemented to form an AC input power supply. An AC power
source is coupled to a conventional full wave rectifier 110 to
produce a rectified input for coupling to the boost inductor 8,
which in turn couples to the Vin lead of the integrated circuit 70.
The remaining leads are coupled as described with respect to FIG.
6. In addition, the rectifier 27 on the secondary side of the
transformer 24 and the load are illustrated. It is to be noted that
the AC input power supply of FIG. 7 may be modified to include
feedback from the secondary side of the transformer 24 through the
use of optoisolators or the like.
[0048] The embodiments of the invention described to this point
have comprised a single "front end" (the modulated switching power
converter) coupled to a single "back end" (the alternating current
tank). The inventor has discovered that additional power and
linearity may be provided by coupling a modulated switching power
converter to a plurality of alternating tanks as illustrated in
FIG. 8. In this embodiment, the modulated switching power converter
5 generates an internal regulated voltage output as in the
previously described embodiments. A plurality of alternating
current tanks 20 are coupled in parallel to the internal regulated
voltage output. This plurality of alternating current tanks 20 may
be "hardwired" to the internal regulated voltage output such that
each alternating current tank 20 within the plurality is always
connected to the internal voltage output and constantly operating.
Alternatively, an optional regulating switch 120 may couple between
the plurality of alternating current tanks 20 and the modulated
switching power converter 5. The regulating switch 120 couples a
given alternating current tank 120 to the internal voltage output
on an as-required basis. Regulating switch thus includes logic
coupled to a feedback loop. The logic controls active switches,
such as FETs, within the regulating switch 120 that couple a given
alternating current tank to the internal voltage output. The
voltage output from each of the alternating current tanks 20 within
the plurality may be rectified or used as an AC output before being
coupled in parallel to the load. A clock (not illustrated) may
couple to each of the alternating current tanks 20 to permit the
switches within the alternating current tank to operate
synchronously with each other. As discussed herein, the modulated
switching power converter may be of any type known in the art, for
example, a boost or a buck converter. The plurality of alternating
current tanks may have their storage capacitor either in series or
in parallel with the primary winding of their respective
transformers. As discussed herein, the modulated switching power
converter may be of any type known in the art, for example, a boost
or a buck converter. The plurality of alternating current tanks may
have their storage capacitor either in series or in parallel with
the primary winding of their respective transformers.
[0049] Regardless of whether a single alternating current tank or a
plurality of alternating current tanks are used, the "front end" of
the power converter may be a buck power supply. Turning now to FIG.
9, a buck power converter 125 is illustrated. The buck converter
125 comprises a power switch Q1 (typically a field effect
transistor (FET)) coupled to a buck inductor 130, a buck diode 135,
and a storage capacitor 12. A pulse width modulator 14 adjusts a
duty cycle of the power switch Q1 in response to sensing an
internal output voltage, V.sub.int. The relationship between the
input voltage, V.sub.in, and V.sub.int may be approximated as
V.sub.int=V.sub.in*(.sub.on/T)
[0050] where T is the switching period and t.sub.on is the ON time
of the power switch Q1. Inspection of FIG. 9 reveals that, just as
with the boost power converter of FIG. 2, the ground of V.sub.int
is not isolated from that of V.sub.in.
[0051] Just as the "front end" may be either a boost or a buck
converter, the "back end" alternating current tank may have its
storage capacitor either in series or in parallel with the primary
winding of its transformer. Turning now to FIG. 10, an alternating
current tank 20 is illustrated in which storage capacitor 34 is in
parallel with the primary winding 22. The remainder of the
alternating current tank 20 is as described with respect to FIG. 3.
The inventor has discovered that in such a configuration, it is
preferable that a parallel resonant tank circuit be formed between
the storage capacitor 34 and the leakage inductance 36 of the
primary winding 22. As is known from the duality of parallel vs.
series resonant tanks, rather than exciting a quasi-sinusoidal
current through the primary winding as discussed with respect to
FIG. 4, the alternating current tank of FIG. 10 has a
quasi-sinusoidal voltage excited through the primary. An
alternating current, is, of course, still present in the primary
winding 22. However, the alternating current has a square waveform
instead of a quasi-sinusoidal waveform. As illustrated, a rectifier
27 coupled to the secondary winding 26 is used to rectify the
current through the load. If desired, however, the rectifier could
be omitted, resulting in an alternating current flowing through the
load. It is to be noted that, in certain high power applications,
the leakage inductance of the transformer may be too small for
efficient resonant performance. In such circumstances, an
additional inductor (not illustrated) arranged in parallel with the
storage capacitor 34 might be required for efficient resonant
performance.
[0052] Specific examples of the present invention have been shown
by way of example in the drawings and are herein described in
detail. It is to be understood, however, that the invention is not
to be limited to the particular forms or methods disclosed, but to
the contrary, the invention is to broadly cover all modifications,
equivalents, and alternatives encompassed by the scope of the
appended claims.
* * * * *