U.S. patent application number 09/802484 was filed with the patent office on 2002-01-31 for global positioning system receiver capable of functioning in the presence of interference.
Invention is credited to Heinzl, Johann, Jacobsen, Gary, Shenoi, Kishan, Yang, Jining.
Application Number | 20020012411 09/802484 |
Document ID | / |
Family ID | 26890406 |
Filed Date | 2002-01-31 |
United States Patent
Application |
20020012411 |
Kind Code |
A1 |
Heinzl, Johann ; et
al. |
January 31, 2002 |
Global positioning system receiver capable of functioning in the
presence of interference
Abstract
Systems and methods are described for a GPS receiver capable of
functioning in the presence of interference. A method includes
detecting an interfering signal including: tuning a band pass
filter over a frequency range; and at each of a plurality of
incremental frequencies: computing a set of band pass filter
coefficients; sending the set of band pass filter coefficients to a
digital filter; repeatedly transforming an analog-to-digital
converter output having a quantization level in excess of 2 bits
into a band pass filter output with the digital filter to obtain a
plurality of samples; computing an average of the plurality of
samples; and comparing the average to a threshold to detect peaks
that exceed a threshold. An apparatus, comprising: an analog radio
frequency circuit; an analog-to-digital converter coupled to the
analog radio frequency circuit, the analog-to-digital converter
providing a quantization level in excess of 2 bits; a digital
filter coupled to the analog-to-digital converter; and a digital
circuit coupled to the digital filter.
Inventors: |
Heinzl, Johann;
(Forestville, CA) ; Jacobsen, Gary; (San Jose,
CA) ; Yang, Jining; (Sunnyvale, CA) ; Shenoi,
Kishan; (Saratoga, CA) |
Correspondence
Address: |
FULBRIGHT & JAWORSKI L.L.P.
A REGISTERED LIMITED LIABILITY PARTNERSHIP
600 CONGRESS AVENUE, SUITE 2400
AUSTIN
TX
78701
US
|
Family ID: |
26890406 |
Appl. No.: |
09/802484 |
Filed: |
March 9, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
60194798 |
Apr 5, 2000 |
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Current U.S.
Class: |
375/350 ;
375/E1.023 |
Current CPC
Class: |
H03J 1/0008 20130101;
H03J 3/08 20130101; H04B 1/7102 20130101; G01S 19/21 20130101; G01S
19/37 20130101 |
Class at
Publication: |
375/350 |
International
Class: |
H04B 001/10 |
Claims
What is claimed is:
1. A method, comprising detecting an interfering signal including:
tuning a band pass filter over a frequency range; and at each of a
plurality of incremental frequencies: computing a set of band pass
filter coefficients; sending said set of band pass filter
coefficients to a digital filter; repeatedly transforming an
analog-to-digital converter output having a quantization level in
excess of 2 bits into a band pass filter output with said digital
filter to obtain a plurality of samples; computing an average of
said plurality of samples; and comparing said average to a
threshold to detect peaks that exceed a threshold.
2. The method of claim 1, wherein said digital filter is
implemented using a field programmable gate array (FPGA) or
application specific integrated circuit (ASIC).
3. The method of claim 1, wherein transforming includes computing a
summation of absolute values of the band pass filter output that
are a function of said band pass filter coefficients.
4. The method of claim 1, wherein transforming includes computing a
summation of the squares of the band pass filter output that are a
function of said band pass filter coefficients.
5. The method of claim 1, wherein tuning includes tuning the band
pass filter in increments equal to approximately half of a band
pass filter bandwidth.
6. The method of claim 1, wherein said analog-to-digital converter
output has a quantization level of from 8 to 10 bits.
7. The method of claim 1, wherein the interfering signal includes
continuous wave jamming.
8. The method of claim 1, further comprising computing a frequency
of said interfering signal.
9. The method of claim 1, further comprising attenuating said
interfering signal by a notch filter centered at said frequency of
said interfering signal from an input X.sub.n to said notch filter
to obtain an output y.sub.n from said notch filter that contains a
useful portion of the input x.sub.n.
10. The method of claim 9 wherein said interfering signal is
substantially removed by the notch filter and resulting signal is
used as an input to the digital part of a GPS receiver.
11. The method of claim 10, wherein a result of subtracting the
intermediate output u.sub.n from the input x.sub.n is mapped to a
2-bit digital representation by assigning to each sample at the
output of the notch filter: a magnitude bit of "1" (or "0") if the
magnitude of the output of the notch filter is greater than a
reference level; or a magnitude bit of "0" (or "1") if the
magnitude of the notch filter not greater than said reference
level; and a sign bit of "1" (or "0") if the polarity of the output
of the notch filter is positive; or a sign bit of "0" (or "1") if
the polarity of the output of the notch filter is negative.
12. The method of claim 10, wherein the output of the notch filter
is transformed with a .DELTA..SIGMA.M circuit to provide an
improvement with regard to in-band signal-to-noise.
13. The method of claim 10, wherein the output of the notch filter
is provided to a multi-bit correlator as a multi-bit input.
14. The method of claim 13, wherein the multi-bit input to the
multi-bit correlator includes from 8 to 16 bits.
15. The method of claim 10, further comprising calibrating a code
phase alignment with a timing offset that is a function of a
frequency of said notch filter.
16. A computer program, comprising computer or machine readable
program elements translatable for implementing the method of claim
1.
17. A global positioning system (GPS) receiver including the method
of claim 1.
18. An electronic media, comprising a program for performing the
method of claim 1.
19. An apparatus, comprising the electronic media of claim 18.
20. A process of receiving a global positioning system signal,
comprising utilizing the apparatus of claim 19.
21. An apparatus, comprising: an analog radio frequency circuit; an
analog-to-digital converter coupled to said analog radio frequency
circuit, said analog-to-digital converter providing a quantization
level in excess of 2 bits; a digital filter coupled to said
analog-to-digital converter; and a digital circuit coupled to said
digital filter.
22. The apparatus of claim 21, wherein said digital filter includes
a field programmable gate array.
23. The apparatus of claim 21, further comprising a selector
located between 1) a) said digital filter and b) said analog radio
frequency circuit and 2) said digital circuit.
24. The apparatus of claim 21, wherein said digital filter includes
a notch filter.
25. The apparatus of claim 21, wherein said digital filter includes
a band-pass filter.
26. The apparatus of claim 21, wherein said analog-to-digital
converter provides a quantization level of from 8 to 10 bits.
27. The apparatus of claim 21, wherein said digital circuit
includes a multi-bit correlator.
28. The apparatus of claim 21, wherein said analog radio frequency
circuit composes a chip.
29. A method of receiving a global positioning system signal which
comprises utilizing the apparatus of claim 21.
30. An integrated circuit, comprising the apparatus of claim
21.
31. A circuit board, comprising the integrated circuit of claim
30.
32. A computer, comprising the circuit board of claim 31.
33. A network, comprising the computer of claim 32.
34. An electronic media, comprising: a computer program having a
sequence of instructions for detecting an interfering signal
including: tuning a band pass filter over a frequency range; and at
each of a plurality of incremental frequencies: computing a set of
band pass filter coefficients; sending said set of band pass filter
coefficients to a digital filter; repeatedly transforming an
analog-to-digital converter output having a quantization level in
excess of 2 bits into a band pass filter output with said digital
filter to obtain a plurality of samples; computing an average of
said plurality of samples; and comparing said average to a
threshold to detect peaks that exceed a threshold.
35. An apparatus, comprising the electronic media of claim 34.
36. A method, comprising deploying the electronic media of claim
34.
37. A computer program comprising computer program means adapted to
perform the steps of detecting an interfering signal including:
tuning a band pass filter over a frequency range; and at each of a
plurality of incremental frequencies: computing a set of band pass
filter coefficients; sending said set of band pass filter
coefficients to a digital filter; repeatedly transforming an
analog-to-digital converter output having a quantization level in
excess of 2 bits into a band pass filter output with said digital
filter to obtain a plurality of samples; computing an average of
said plurality of samples; and comparing said average to a
threshold to detect peaks that exceed a threshold when said program
is run on a computer.
38. A computer program as claimed in claim 37, embodied on a
computer-readable medium.
39. A kit for retrofitting a receiver, comprising: an analog radio
frequency circuit; an analog-to-digital converter coupled to said
analog radio frequency circuit, said analog-to-digital converter
providing a quantization level in excess of 2 bits; a digital
filter coupled to said analog-to-digital converter; and a digital
circuit coupled to said digital filter.
40. The kit of claim 39, further comprising instructions.
Description
CROSS-REFERENCE TO RELATED APPLICATION
[0001] This application is a continuation-in-part of, and claims a
benefit of priority under 35 U.S.C. 119(e) and/or 35 U.S.C. 120 of
copending U.S. Ser. No. 60/194,798, filed Apr. 5, 2000, the entire
contents of which are hereby expressly incorporated by reference
for all purposes.
BACKGROUND OF THE INVENTION
[0002] 1. Field of the Invention
[0003] The invention relates generally to the field of global
positioning system (GPS) receivers. More particularly, the
invention relates to a GPS receiver capable of functioning in the
presence of interference such as continuous wave (CW) jamming. The
invention thus relates to a GPS receiver of the type that can be
termed CW jamming robust.
[0004] 2. Discussion of the Related Art
[0005] GPS receivers are most often considered as tools for
location or navigation. However, GPS receivers can also be used to
provide time. In particular, the GPS constellation is monitored and
controlled by the Department of Defense so that a GPS receiver can
be used to extract time traceable to the US Naval Observatory
(USNO) which in turn tracks UTC (Universal Coordinated Time) that
is maintained in Paris as the international time standard.
Consequently a GPS receiver can be used to discipline a local
oscillator to provide accurate frequency and can be used to provide
a time signal that tracks UTC. We refer to time extracted in such a
manner as "GPS time."
[0006] Symmetricom Inc. (San Jose, Calif.) provides a series of
products based on GPS for providing accurate time and frequency.
For example, Symmetricom products are used by telephone companies
to provide accurate timing (frequency) signals. Wireless telephone
operators that deploy CDMA (code divisional multiple access)
cellular telephony are familiar with Symmetricom timing units that
are installed in the base station to provide accurate frequency as
well as accurate time, embodied, for example, in a signal
designated as 1-PPS (for one pulse per second). In practice the
interval between pulses can be something other than 1 second, but
the pulse position is related in an absolute sense to GPS time. For
example, CDMA base stations often require a pulse corresponding to
every even second of GPS time.
[0007] When a receiver is deployed, there is often little
flexibility as to where the antenna may be mounted. In the case of
a cellular base station, the GPS antenna is mounted in reasonable
proximity to the cellular antenna. Now the signal level received at
the antenna from the GPS satellites is extremely small, of the
order of -130 dBm (10.sup.-13 milliwatt) and thus the GPS receiver
is quite susceptible to interfering signals. The nominal bandwidth
(one-sided) of the GPS spectrum is 1.1 MHz centered at the RF
carrier which is transmitted at the L1 frequency (1.57542 GHz). An
interfering signal outside this band can, in principle, be removed
by the filters that delineate the GPS band, either at the L1
frequency itself or in combination with filters at a suitable
intermediate frequency (IF). Using (analog) filters in this manner
to reduce the impact of interfering signals is well known.
[0008] There have been some techniques proposed to date that have
addressed the problem of interfering signals that lie within the
GPS bandwidth. References [6 through 9] address methods proposed
for improved performance in the presence of jamming CW
interference. The jamming signal could well be inband. Generally
speaking, these methods address the mitigation of the impact of the
jamming signal on the delay-locked-loop used for code tracking.
[0009] One approach is to use an increased number of bits in the
ADC (A/D converter) and rely on the correlation process to mitigate
the impact of the jammer. This approach is based on the premise
that the correlator will distinguish between the Gold code and a CW
(tone), which indeed it does, provided that the power level of the
jammer is not so large as to send the ADC into a saturation mode.
By increasing the dynamic range of the ADC (with the concurrent
increase in number of bits) one would expect jamming immunity.
[0010] A second approach that has been proposed is to use an
antenna array and develop a synthetic antenna pattern that has a
null in the direction from which the CW signal is being received.
This method is quite complex and expensive.
[0011] A third approach is indicated in Ref. [7]. The method
described therein is one of "cancellation" rather than "removal."
That is, a "replica" (or as close to a replica as is feasible) of
the jamming signal is generated and is subtracted from the received
signal. This cancellation method is of value when the jamming
signal is indeed a very narrow-band CW signal, essentially a pure
tone. For wider applicability, multiple CW "replicas" can be
generated and subtracted in the event that there are more than one
jamming signals. This method has a severe drawback in the sense
that if the replica generated is not indeed identical and
180-degrees out-of-phase from the jamming signal, one could
actually be adding interference.
SUMMARY OF THE INVENTION
[0012] There is a need for the following embodiments. Of course,
the invention is not limited to these embodiments.
[0013] One embodiment of the invention is based on a method,
comprising: detecting an interfering signal including: tuning a
band pass filter over a frequency range; and at each of a plurality
of incremental frequencies: computing a set of band pass filter
coefficients; sending said set of band pass filter coefficients to
a digital filter; repeatedly transforming an analog-to-digital
converter output having a quantization level in excess of 2 bits
into a band pass filter output with said digital filter to obtain a
plurality of samples; computing an average of said plurality of
samples; and comparing said average to a threshold to detect peaks
that exceed a threshold. Another embodiment of the invention is
based on an apparatus, comprising: an analog radio frequency
circuit; an analog-to-digital converter coupled to said analog
radio frequency circuit, said analog-to-digital converter providing
a quantization level in excess of 2 bits; a digital filter coupled
to said analog-to-digital converter; and a digital circuit coupled
to said digital filter. Another embodiment of the invention is
based on an electronic media, comprising: a computer program having
a sequence of instructions for detecting an interfering signal
including: tuning a band pass filter over a frequency range; and at
each of a plurality of incremental frequencies: computing a set of
band pass filter coefficients; sending said set of band pass filter
coefficients to a digital filter; repeatedly transforming an
analog-to-digital converter output having a quantization level in
excess of 2 bits into a band pass filter output with said digital
filter to obtain a plurality of samples; computing an average of
said plurality of samples; and comparing said average to a
threshold to detect peaks that exceed a threshold. Another
embodiment of the invention is based on a computer program,
comprising: computer program means adapted to perform the steps of
detecting an interfering signal including: tuning a band pass
filter over a frequency range; and at each of a plurality of
incremental frequencies: computing a set of band pass filter
coefficients; sending said set of band pass filter coefficients to
a digital filter; repeatedly transforming an analog-to-digital
converter output having a quantization level in excess of 2 bits
into a band pass filter output with said digital filter to obtain a
plurality of samples; computing an average of said plurality of
samples; and comparing said average to a threshold to detect peaks
that exceed a threshold when said program is run on a computer.
[0014] The embodiments described above relate to the detection of
an interfering signal. It is easily shown that by using a notch
filter instead of a band pass filter, the interfering (CW) signal
can be removed, thereby minimizing the impact of the interfering
signal on the performance of the GPS receiver. Furthermore, the
structure of the notch filter and structure of the band pass filter
are closely related so that computing a set of notch filter
coefficients based on a set of band pass filter coefficients, and
vice versa, is quite straightforward.
[0015] These, and other, embodiments of the invention will be
better appreciated and understood when considered in conjunction
with the following description and the accompanying drawings. It
should be understood, however, that the following description,
while indicating various embodiments of the invention and numerous
specific details thereof, is given by way of illustration and not
of limitation. Many substitutions, modifications, additions and/or
rearrangements may be made within the scope of the invention
without departing from the spirit thereof, and the invention
includes all such substitutions, modifications, additions and/or
rearrangements.
BRIEF DESCRIPTION OF THE DRAWINGS
[0016] The drawings accompanying and forming part of this
specification are included to depict certain aspects of the
invention. A clearer conception of the invention, and of the
components and operation of systems provided with the invention,
will become more readily apparent by referring to the exemplary,
and therefore nonlimiting, embodiments illustrated in the drawings,
wherein like reference numerals (if they occur in more than one
view) designate the same elements. The invention may be better
understood by reference to one or more of these drawings in
combination with the description presented herein. It should be
noted that the features illustrated in the drawings are not
necessarily drawn to scale.
[0017] FIG. 1 illustrates a schematic block diagram of conventional
GPS signal generation, appropriately labeled "Prior Art."
[0018] FIG. 2 illustrates a schematic block diagram of the
fundamental features of a conventional GPS receiver, appropriately
labeled "Prior Art."
[0019] FIG. 3 illustrates a schematic generic block diagram of a
conventional GPS receiver architecture, appropriately labeled
"Prior Art."
[0020] FIG. 4 illustrates a schematic block diagram of a generic
GPS receiver, appropriately labeled "Prior Art."
[0021] FIG. 5 illustrates a schematic block diagram of conventional
GPS chip-set interconnections (signal), approximately labeled
"Prior Art."
[0022] FIG. 6 illustrates a schematic block diagram of a
configuration for providing an anti-jamming feature, representing
an embodiment of the invention.
[0023] FIG. 7 illustrates positioning of a notch filter response to
eliminate continuous wave interference, representing an embodiment
of the invention.
[0024] FIG. 8 illustrates a notch filter pole/zero configuration in
the Z plane, representing an embodiment of the invention.
[0025] FIG. 9 illustrates a flow diagram depicting implementation
of a notch filter, representing an embodiment of the invention.
[0026] FIG. 10A illustrates a schematic block diagram of an
implementation of a bandpass filter, representing an embodiment of
the invention.
[0027] FIG. 10B illustrates a signal flow graph for the bandpass
filter shown in FIG. 10A.
[0028] FIG. 11A illustrates a schematic block diagram of a
quantizer (word length reduction), representing an embodiment of
the invention.
[0029] FIG. 11B illustrates the input-output characteristic
associated with the quantizer depicted in FIG. 11 A.
[0030] FIG. 12 illustrates a schematic block diagram of a
.DELTA..SIGMA.M configuration for wordlength reduction,
representing an embodiment of the invention.
[0031] FIG. 13 illustrates a schematic block diagram of a multi-bit
correlator architecture, representing an embodiment of the
invention.
DESCRIPTION OF PREFERRED EMBODIMENTS
[0032] The invention and the various features and advantageous
details thereof are explained more fully with reference to the
nonlimiting embodiments that are illustrated in the accompanying
drawings and detailed in the following description. Descriptions of
well known components and processing techniques are omitted so as
not to unnecessarily obscure the invention in detail. It should be
understood, however, that the detailed description and the specific
examples, while indicating preferred embodiments of the invention,
are given by way of illustration only and not by way of limitation.
Various substitutions, modifications, additions and/or
rearrangements within the spirit and/or scope of the underlying
inventive concept will become apparent to those skilled in the art
from this detailed description.
[0033] Within this application several publications are referenced
by Arabic numerals within brackets. Full citations for these, and
other, publications may be found at the end of the specification
immediately preceding the claims after the section heading
References. The disclosures of all these publications in their
entireties are hereby expressly incorporated by reference herein
for the purpose of indicating the background of the invention and
illustrating the state of the art.
[0034] The below-referenced U.S. patent(s), and U.S. patent
application(s) disclose embodiments that were satisfactory for the
purposes for which they are intended. The entire contents of U.S.
patent application Ser. No. 60/194,798, filed Apr. 5, 2000 are
hereby expressly incorporated by reference herein for all
purposes.
[0035] Our method uses digital filters to remove the jamming
signal. It is quite robust in the sense that the nature of the
jamming signal is not constrained and there is just a small penalty
paid if we try to "remove" a jamming signal that is not indeed
present.
General Background
[0036] The GPS signal has two distinct components. These are often
considered in terms of the coding scheme and availability. The
"P-code" signal was designed for military applications and is not
generally available for use by common (commercial) receivers. The
"C/A-code" is available for general use. Commercial GPS receivers
are universally based on the C/A-code and the "P-code" signal
appears as "noise." In this discussion we shall concentrate on the
C/A version exclusively.
[0037] The signal transmitted by a GPS satellite comprises a Gold
code with superimposed data. The base-band signal is modulated onto
the RF carrier at the L1 frequency (1.57542 GHz). The chip rate of
the Gold code (C/A code) is 1.023 Mchips/sec and the period of the
Gold code is 1023 chips. Thus, the period of the Gold code is 1
msec. The underlying data rate is 50 bps (20 msec per data-bit).
That is, each data-bit is transmitted over 20 periods of the Gold
code.
[0038] A rudimentary block diagram of the GPS signal generation is
depicted in FIG. 1. The Gold code generator 110, where each SV
(Satellite Vehicle) is assigned a unique Gold code, provides the
periodic signal of 1023 chips at the chip rate of 1.023 Mchips/sec.
This can be viewed as a two-level signal (binary valued) with the
two values logically equivalent to a "Digital-0" and "Digital-1"
(or Logic-0 and Logic-1). The binary data at 50 bps is modulated on
the Gold code. The action is simple in that a data-bit of "1"
inverts the Gold code whereas a data-bit of "0" does not (obviously
the role of "1" and "0" for the data-bits can be reversed with no
loss of generality) and this is equivalent to an EXCLUSIVE-OR
(EXOR) logic operation. This composite signal (which is the
"base-band" signal) is modulated onto the RF carrier for
transmission. In the satellite, the data, chip, and carrier are
phase and frequency synchronous. That is, the data signal changes
at a chip boundary; the chip boundary coincides with a
zero-crossing of the carrier; and all timing signals
(data-bit-clock, chip-clock, and carrier-frequency) are derived
from the same source.
[0039] Furthermore, the actual data is formatted into "words,"
"sub-frames," and "frames." Each "word" comprises 30 bits (600
msec); each "subframe" consists of 10 words (300 bits, 6 sec.);
each frame of 5 subframes or 1500 bits (50 words, 1500 bits, 30
sec.).
[0040] The notion of "acquiring GPS" has multiple connotations. At
the lowest level is "code-phase acquisition" wherein the receiver
has established the "start-of-code" (the 1-msec period) for the SV
being acquired (tracked). The receiver can be considered to be in
"chip-synch" with the SV. Chip-synch is necessary in order to
achieve "data-bitsynch," considered next. "Data-bit-synch" wherein
the receiver has identified the start of a data-bit-period (which
is 20 msec). Data-bit-synch is important for demodulating the data.
The data pattern is examined to establish "frame-synch."
Frame-synch is necessary for interpretation of the data.
[0041] Gold codes have good auto-correlation and cross-correlation
properties. This behavior is one of the reasons Gold codes were
chosen as the foundation of the GPS signal structure. The
correlation properties of Gold codes dictate the structure of the
GPS receiver, the underlying principle of which is depicted in FIG.
2. The incoming RF signal from the antenna 210 is down-converted to
base-band to provide the "I" (in-phase) and "Q` (quadrature)
components. A correlation operation is performed (on both I and Q
components) by multiplying the incoming (base-band version) signal
by a replica of the appropriate Gold code and integrating. The
integration is performed over 1-msec intervals (the period of the
Gold code). The magnitude of this (complex, i.e., I and Q)
correlation is indicative of the presence or absence of the chosen
Gold code in the received signal. In particular, if the chosen Gold
code is absent from the received signal, the correlation value will
always be small (i.e., the SV that uses this Gold code is not
visible). If the chosen Gold code is present (i.e., the SV is
visible) and the replica generated locally is phase-aligned with
Gold code in the received signal, then the correlation (magnitude)
value will be high.
[0042] In a GPS receiver, several such correlators operate in
parallel, each using a different Gold code. That is, multiple
satellites can be viewed simultaneously. In practice, one must take
into account the fact that the SV and the receiving antenna are
moving relative to one another. The satellites orbit the earth
every 12 hours and the receiving antenna, which can be viewed as a
point on the earth's surface, is completing one rotation every 24
hours. Thus, a given satellite is not always in view and, further,
when in view is moving from one horizon to the opposite horizon.
This movement introduces a Doppler shift that is manifest in terms
of a shift in carrier frequency as well as a shift in code
frequency. Further, this Doppler effect is not constant but changes
with relative position of the SV with respect to the receiver
antenna. To account for this apparent "drift" a GPS receiver takes
the form depicted in FIG. 2. The effective carrier frequency is
"adjusted" using an NCO 220 and the effective chip-rate is adjusted
using an NCO 230 as well.
[0043] In a GPS receiver 300 the assembly of NCOs and correlator is
often called a "channel" and with this terminology a "12-channel"
receiver will have 12 parallel assemblies of the form shown in FIG.
3 where a generic block diagram of a GPS receiver is depicted. The
correlators provide one (complex) correlation value every
millisecond. A processor unit 310 reads these correlation values to
extract the GPS data (if present); the processor 310 also controls
the NCOs for adjusting carrier and code rates and also controls the
"code-phase." The code-phase is simply the notion of the start of
the code (the first chip of the 1023 that comprise one period).
Each satellite is at a different distance from the receiver antenna
and thus the arrival of the first chip of the Gold code from the
different satellites will be different. The processor 310 controls
the code generators of the different channels to keep the local
replica of the code aligned with the received code and this action
is equivalent to controlling the code-phase for that channel.
[0044] Several chip-sets for implementing GPS receivers are
available. All such chip-sets follow the rudimentary schematic
depicted in FIG. 4. The signal from an antenna 410 is amplified
using a low noise amplifier (LNA) and down-converted to a suitable
intermediate frequency (IF). The signal is then converted into a
digital format using a suitable sampling rate and wordlength (i.e.,
using an A/D converter 430). All available chip-sets assume a very
coarse quantization, typically 1-bit (2 levels) or 2-bits (four
levels) or 1.5-bits (3 levels). After conversion, the effective IF
(i.e., down-converted carrier frequency) is typically
(approximately) one quarter the sampling frequency with a small
offset. This down-conversion and A/D conversion is often achieved
in the "analog chip" of the chip-set. The digital signal comprising
the sampled and quantized GPS signal (at IF) is fed to the "digital
chip" of the chip-set. The subsequent down-conversions,
correlations, etc. are performed using hardware in the
digital-chip. The digital chip also includes a processing element
(i.e., a programmable micro-processor) for performing the
computations related to acquisition, tracking, data-demodulation,
navigation, etc., associated with a GPS receiver. In some
implementations the "digital-chip" is split into the GPS specific
part and a general-purpose processor, usually of the RISC type.
[0045] Whereas the discussion here is applicable in principle to
all chip-sets, for specificity we shall assume we are using the
readily commercially available chip-set from Mitel Semiconductor
Inc. (see Ref. [2]).
[0046] Methods for Acquiring GPS
[0047] The underlying principle of GPS signal acquisition is the
notion of correlation. Consider the situation where the "signal" is
a particular Gold code, {g(k); k=0,1, . . . , 1022}, sent
continually. Assume that the down-conversion has translated the GPS
signal from L1 frequency down to base-band and that we have taken
(complex) samples at the chip rate (1.023 MHz). The sampled
(complex) signal can be written as
w(k)=.alpha..multidot.g(k-M)+s(k) (2.1.1)
[0048] where s(k) is the noise component (from a variety of
sources), and .alpha. is a complex number representative of the
amplitude and phase of the "signal" component (a phase angle can be
introduced because the phase of the equivalent local oscillator is
not the same as the equivalent carrier). We assume that the sample
index, k, is based on some reference in the receiver and the
incoming Gold code "start-of-code" is offset from this time
reference by M chips. The signal power is thus
.vertline..alpha..vertline..sup.2. Assuming the noise power is
.sigma..sup.2, the raw signal-to-noise ratio (SNR) is given by 1
SNR = 20 log 10 ( ) dB (2.1.2)
[0049] At the receiver we can generate a replica of the Gold code
with a (programmable, under processor control) offset, say
{g(k-K)}. This locally generated signal is correlated with the
incoming signal in "chunks" of N=1023 samples (the period of the
Gold code), or essentially we generate 1-msec samples of the
correlation between the local replica and the incoming signal. This
operation can be written as (with the index n representing samples
taken at 1 kHz rate which is 1 msec intervals) 2 x ( n ) = k = ( n
- 1 ) N k = nN = 1 g ( k - K ) w ( k ) = k = ( n - 1 ) N k = nN - 1
g ( k - K ) ( g ( k - M ) + s ( k ) ) = R gg ( K - M ) + k = ( n -
1 ) N k = nN - 1 g ( k - K ) s ( k ) (2.1.3)
[0050] where R.sub.gg(k) is the autocorrelation function of the
Gold code. The autocorrelation function of the Gold code exhibits a
peak (of value N) at zero lag and is small elsewhere. Thus, if the
Gold code is present, and we have chosen the offset of the local
replica appropriately (i.e., when K=M), the correlation samples
{x(n)} will exhibit a large "dc" value equal to N.alpha.. The two
components of x(n) can be viewed as "signal" (the first part) and
"noise" (the second part). Assuming that the additive noise {s(k)}
is white noise, then the variance (i.e., power) of the noise
component of {x(n)} is N.sigma..sup.2. The power of the signal
component is N.sup.2.alpha..sup.2. The post-correlation SNR is thus
given by 3 SNR = 20 log 10 ( ) + 10 log 10 ( N ) dB (2.1.4)
[0051] Correlation has thus provided an increase in SNR of roughly
30 dB provided that the controlled offset, K, is equal to the
actual offset, M. Thus, even if the raw SNR is "small," so small
that presence of the Gold code signal is buried in the noise and
may not be discernible by a simple observation of the sample
values, the correlation operation makes detection viable. In GPS
literature the signal-to-noise ratio is often considered in terms
of "C over N" or "C/N" where the noise power is considered in a 1
Hz bandwidth. The post-correlation SNR, considering the implied
sampling rate is 1 kHz, is roughly 30 dB smaller than the "C/N"
number. Typical values for the C/N number, assuming the antenna is
roof-top mounted and has a clear, unobstructed, view of the sky
(and hence the satellites) are in the 40 to 45 dB range for
satellites that are roughly overhead.
[0052] The received signal will contain components from all the
visible satellites. However, each satellite uses a different Gold
code and Gold codes have the nice property that the
cross-correlation between different Gold codes, for any offset, is
small, typically (N/10), worst-case, and thus the correlation
process suppresses signals from satellites other than the one whose
Gold code replica is being employed by the correlator.
[0053] The notion of "code-phase acquisition" is equivalent to
establishing the correct offset M. The conventional method for
acquiring code-phase involves searching over all 1023 possible
values for K. That is, if we are searching for a particular
satellite identified with a particular Gold code, {g(k)}, we
compute the strength, X(K), of the correlation of this Gold code
with the input signal {w(k)} for various values of offset, K, and
compare this value with a threshold, T. If X(K)>T for some value
of K, we declare acquisition of code-phase; if X(K)<T for all
1023 values of K, then we declare that we cannot acquire code-phase
for that satellite at that time.
[0054] To establish the threshold, T, we compute the strength of
the correlation (at some arbitrary offset), say Y, of the received
input signal using a Gold code that does not correspond to any
satellite in the GPS constellation, say {h(k)}, and set the
threshold to be somewhat greater than Y. Since we are dealing with
signals that are noise-like, or random, in nature, the estimate of
strength is obtained by averaging over a suitable number of
milliseconds, say L. That is, 4 Y = 1 L n = 0 n = ( L - 1 ) k = ( n
- 1 ) N k = nN - 1 h ( k - K 1 ) w ( k ) 2 T = A Y { A is typically
2 or 3 } X ( K ) = 1 L n = 0 n = ( L - 1 ) k = ( n - 1 ) N k = nN -
1 g ( k - K ) w ( k ) 2 compare X ( K ) T (2.1.5)
[0055] In practice there could be another unknown, namely the
Doppler shift (or sum of the actual Doppler and the frequency
"error" of the local oscillator). Consequently we do this search
process for all values of K over a grid of Doppler shifts. The
local oscillator can be of high quality and the range of Doppler
shifts to account for the frequency bias error of the local
oscillator can be small. The choice of L is typically 4 to 10.
Using larger values of L reduces the variability of the strength
estimate but has the drawback of increasing the search (and thus
acquisition) time. Also, L cannot be made indefinitely large since
the Doppler shift could change over the (long) interval, in which
case the averaging to reduce variability is contraindicated.
[0056] The signal-to-noise ratio influences our ability to detect
the presence of the Gold code and thus our ability to acquire
code-phase-synch. In particular, we can show that 5 Y N 2 X ( M ) N
2 2 + N 2 X ( M ) Y 1 + N 2 2 (2.1.6)
[0057] Assuming that the factor, A, used to determine the threshold
is about 3, then the post-correlation SNR needs to be above 3 dB
corresponding to a "C/N" number of about 33 dB. In practice there
are also other factors that come into play and as a consequence of
which the "C/N" number for reliable detection (using the
conventional scheme described above) is somewhat greater than 33
and is closer to 36 or 37. These other factors include non-ideal
behavior of components such as band-pass filters in the analog
front-end, the A/D converter, down-conversion mixers, amplifier
noise, and so on, all of which tend to reduce the correlation peak
(relative to the noise floor).
[0058] The Code Tracking Loop
[0059] The notion of acquiring code-phase is to establish a
starting point for determining the start of the code in the
received signal. The function of the Code-Tracking Loop is to
automate the mechanism of "holding-on" to this condition. Because
of the motion of the satellites relative to the receiver, the
effective distance between the receiver and the satellite is
continuously changing and hence the code-phase (i.e., apparent
start of code relative to some reference time in the receiver) is
not constant. The Code-Tracking Loop functions to continuously
adjust the local code-phase so that there is match between local
code replica and incoming Gold code from the satellite. This topic
is not discussed in detail here except to mention the
"discriminator" used to assess how much the local code-phase should
be altered so as to keep in alignment with the received Gold
code.
[0060] So far the notion of acquisition was tied to finding the
code-phase for which the correlation (power) was maximized. This
code-phase is referred to as the "prompt." The function of the Code
Tracking Loop is to maintain the code-phase such that the prompt
correlation is (always) maximized. To do this, correlation is also
evaluated at a code-phase corresponding to a half-chip prior, or
"early," and a half-chip delayed, or "late." If the receiver is
indeed tracking the incoming Gold code, then the correlation value
for "prompt" should be larger than the correlation values for
either "early" or "late." Furthermore, if the receiver is indeed in
alignment with the incoming Gold code, then, under ideal
conditions, the correlation value for early and late will be equal.
If they are not equal, then the difference provides an "error
signal" suitable for controlling the NCO associated with the Code
Generator. The control signal derived from the difference (i.e.,
the "error") thus modifies the code-phase such that the "early"
equals "late" condition is satisfied. We still need to ensure that
the prompt correlation value is still large and is larger than
"early" as well as "late."
[0061] Denoting by I.sub.p, I.sub.e, and I.sub.L the real part of
the correlation values for "prompt," "early" and "late," and by
Q.sub.p, Q.sub.e, and Q.sub.L the corresponding imaginary parts,
the following "discriminator" is suggested: 6 = ( I e - I L ) + j (
Q e - Q L ) I p + jQq p = ( I e - I L ) I p + ( Q e - Q L ) Q p I p
2 + Q p 2 + j - ( I e - I L ) Q p + ( Q e - Q L ) I p I p 2 + Q p 2
(2.2.1)
[0062] It can be shown that when good code-lock is achieved, the
discriminator is zero (or close to zero). Further, it can be shown
that the real-part of the right hand side of Eq. (2.2.1.DELTA. is
often adequate for generating a control signal for adjusting the
code-loop NCO.
[0063] Some comments regarding the discriminator are in order.
First, if we can assume that the effective frequency response
between the transmitter (satellite) and the receiver (correlator)
can be modeled as an "ideal" filter, one whose magnitude response
is flat and phase response is linear-phase over the bandwidth of
the principal signal, then the cross-correlation between the local
Gold code replica and the received signal is equivalent to the
auto-correlation of the Gold code itself, scaled by ".alpha." which
takes into account the sign of the data-bit, the flat gain of the
transmission medium, including any filters in the down-conversion
process. Of course there may be an additive noise term as well.
This is depicted in Eq. (2.1.3). Then if we are in code-phase
alignment, we can write
I.sub.p+jQ.sub.p=.alpha..multidot.R.sub.gg(0)+.gamma..sub.p
I.sub.e+jQ.sub.e=.alpha..multidot.R.sub.gg(-.tau.)+.gamma..sub.e
I.sub.L+jQ.sub.L=.alpha..multidot.R.sub.gg(+.tau.)+.gamma..sub.L
(2.2.2)
[0064] In Eq. (2.2.2) the noise terms are referred to by ".gamma.."
The time quantity ".tau." in the equation refers to the implied
delay between the correlation phases. Typically .tau. is equal to
one-half chip but we could choose any fraction of a chip between 0
and 1. Because the autocorrelation function is symmetric, and if we
can ignore the noise component, we can see that when we are in
code-phase alignment that the discriminant, .DELTA., will be zero.
If we are not in code-phase alignment, and we have an offset, say,
.epsilon., then the prompt, early, and late correlation values are
given by
I.sub.p+jQ.sub.p=.alpha..multidot.R.sub.gg(.epsilon.)+.gamma..sub.p
I.sub.e+jQ.sub.e=.alpha..multidot.R.sub.gg(-.tau.+.epsilon.)+.gamma..sub.e
I.sub.L+jQ.sub.L=.alpha..multidot.R.sub.gg(+.tau.+.epsilon.)+.gamma..sub.L
(2.2.3)
[0065] and, again assuming that the noise terms are negligible, the
discriminant .DELTA. is not zero. Since the autocorrelation
function, R.sub.gg(.tau.), is symmetric, the imaginary part of
.DELTA. will be principally noise and the real part will be
monotonically related to the offset, .epsilon., and thus can be
used to generate the control signal for the code NCO.
[0066] If the transmission path cannot be modeled as an "ideal"
filter, then the correlation between the received signal and the
local Gold code replica is no longer related to the autocorrelation
function, R.sub.gg(.tau.), but must be modified to account for the
effective transfer function of the (entire) transmission medium
between the transmitter and the correlator (receiver). Denote this
by R.sub.gg'(.tau.). Then Eq. (2.2.3) gets modified to
I.sub.p+jQ.sub.p=.alpha..multidot.R.sub.gg'(.epsilon.)+.gamma..sub.p
I.sub.e+jQ.sub.e=.alpha..multidot.R.sub.gg'(-.tau.+.epsilon.)+.gamma..sub.-
e
I.sub.L+jQ.sub.L=.alpha..multidot.R.sub.gg'(+.tau.+.epsilon.)+.gamma..sub.-
L (2.2.4)
[0067] Since the cross-correlation function R.sub.gg'(.tau.) is not
necessarily symmetric, then the discriminant .DELTA. is not zero
when we are in perfect code-phase alignment and, further, is zero
when we are offset by some amount, .eta.. The impact on the code
tracking loop is to settle in at an offset which is between 0 and
.eta..
[0068] Impact of Interference
[0069] The analysis provided here assumes that any signal component
present in the GPS band can effectively be modeled as additive
white noise. If there is a strong interfering component which is
coherent, such as a tone, then the analysis is not quite applicable
anymore. Whereas it is true that the action of correlation of input
with the code tends to suppress all signals that are of a form
other than the Gold code in question, this suppression is not
perfect. An attenuation of interference relative to the expected
Gold code is large but often not enough. Even if we assume that the
correlation process introduces a 30 dB discrimination between Gold
code and any other signal, an interfering signal of just -100 dBm
would provide a post-correlation level comparable with that of the
Gold code (assuming a -130 dBm GPS signal level).
[0070] Now consider a tone jammer at a frequency outside the GPS
spectral region. The (analog) filters associated with the RF
down-conversion chain would have to attenuate this signal to a
level well below -100 dBm. When the interference is additive white
noise, integrating over longer intervals helps to improve the
post-correlation SNR. In the case of coherent jammers,
unfortunately, this "nice" property of longer integration intervals
does not hold. Integrating over one code period provides the (bulk
of the) SNR improvement and extending the integration over multiple
code periods provides little improvement.
[0071] When the tone jammer is "in-band" then the analog filters
associated with the RF down-conversion provide no benefit. The only
"defense" is the action of the correlator, and that provides just
about 30 dB of SNR improvement. Furthermore, this improvement is
achievable only if the analog-to-digital converter (ADC) does not
introduce significant overload-related components. Early GPS
receivers used a 1-bit ADC, essentially providing as the digital
word just the polarity of the analog signal being converted. This
coarse quantization introduces a lot of conversion noise but,
provided the analog signal is representative of just GPS signal and
additive white noise, the ADC noise is reasonably white and thus
"suppressed" by the action of the correlator (30 dB SNR improvement
for each code period). Investigations have indicated that if the
sole defense against jamming is the SNR improvement afforded by the
correlator, then there is an improvement in performance if the ADC
word-length is increased.
[0072] From the viewpoint of complexity of the digital
implementation of the correlator, most chip-sets assume that ADC
output wordlength is very limited, typically 1 (2-level) or 1.5
(3-level) or 2 (4-level) bits wide. Clearly, the smaller we make
the word-length, the more beneficial it is from the viewpoint of
circuit complexity, albeit at the expense of receiver performance.
In Ref. [3] a 2-bit adaptive scheme is analyzed and found to be
"adequate." The MITEL chip-set uses just such a conversion scheme
to convert from analog to digital format. However, there is a limit
to jamming immunity afforded by these "conventional" methods. Field
experience has indicated that this level of immunity is inadequate.
In particular, complaints of GPS receivers "losing lock," "going in
and out of lock condition," and other such (similar sounding)
complaints were received regarding GPS receivers deployed in a
variety of locations. In most such cases, when faulty circuitry was
eliminated as the root cause, the reason for such anomalous
behavior was traced to the presence of strong CW interference.
The Anti-jamming Principle and Method
[0073] The methodology developed and described herein for combating
the deleterious impact of CW jamming can be partitioned into four
components. First is the notion of detection. Namely, the scheme
provides for detecting the presence of a potential jammer. Second
is the elimination, or significant attenuation, of this jamming
component while preserving the bulk of the true GPS signal. Third,
is the notion of calibration. By introducing a signal processing
component into the signal path, the code tracking loop settles in
at an offset. This offset can be estimated theoretically in terms
of the offset at which the discriminant is zero. However, the
theoretical analysis cannot take into account all the "non-ideal"
effects of components and so on. Thus, we calibrate the receiver to
see, for a particular design, what the correction term needs to be.
Fourth, the method is appropriate as an "add-on" so that existing
GPS chip-sets can take advantage of the anti-jamming scheme.
[0074] CW Detection and Removal Configuration
[0075] For purposes of specificity, we shall consider the Mitel GPS
chip-set. Variations to accommodate other chip-sets are
straightforward. The arrangement in FIG. 5 indicates how the Mitel
chip-set is deployed. For clarity, only the signal lines that are
of interest have been shown. The "RF chip" 500 handles the analog
section including the down-conversion. The RF (radio frequency)
chip 500 provides as one output 510 the actual "baseband" IF
(intermediate frequency) analog signal as well as a second output
520 which is the output of an ADC that converts the baseband IF
analog signal to digital. The ADC is built into the RF chip 500. In
conventional designs the digital output of the ADC is connected
from the RF chip to the digital chip 540 and the analog output is
either left floating or properly terminated to avoid reflections or
extraneous signal injection. The digital chip 540 contains the bulk
of the receiver including the correlators and processor. In FIG. 5
we indicate that the processor can interface with external
peripherals via the "processor interface" 550.
[0076] To implement the anti-jamming feature, an analog-to-digital
converter 610 that provides a quantization level in excess of 2
bits can be used. The (high-precision) ADC output 620 is fed to the
device that implements the digital filters associated with the
anti-jamming feature. In the current implementation this is
achieved using an FPGA (Field Programmable Gate Array) 630 that
operates as a peripheral under control of the receiver processor
located within the digital chip 540. The interconnection is
depicted in FIG. 6 (again not all signal lines are shown). The
interconnection includes an output 640 from the FPGA (Field
Programmable Gate Array) 630. A selector 650 can be coupled between
the digital chip and the FPGA 630. Clearly the additional circuitry
can be provided in a manner that the digital chip 540 can accept
the digital IF from either the RF chip 500 or the FPGA 630.
Consequently the anti-jamming feature can be incorporated and left
unused if it is known that there is no CW interference present.
[0077] Detection and Removal of CW Interference
[0078] Within the FPGA digital notch filters are implemented which
position transmission zeros at the precise frequency at which we
have CW interference. This is the means by which the interference
is removed.
[0079] In FIG. 7 we indicate the presence of a CW interfering
signal 710 present in the Digital IF (output of the ADC). The CW
(continuous wave) interfering signal is located at f.sub.1. The
frequency range shown is between 0 (dc) and 0.5 f.sub.s, where
f.sub.s is the sampling rate used to generate the digital IF
signal. The GPS spectrum has a nominal width of approximately 1 MHz
and is centered at a "digital IF" of f.sub.o. In keeping with
normal digital signal processing concepts, the frequency range
depicted is adequate to completely describe the situation since the
signals are all real-valued. In the Mitel chip-set implementation
the sampling frequency is (40/7) MHz, or approximately 5.71 MHz.
The analog down-conversion centers the signal at (approximately)
4.3 MHz and because of the inherent aliasing associated with
sampling, the digital IF (i.e., f.sub.o) is approximately 1.41
MHz.
[0080] From the theory of digital filter design (see, for example,
Ref. [4]) we can achieve the requisite notch filter characteristic
with a digital filter that has a transfer function of zero at the
desired notch frequency and a corresponding transfer function pole
close to the zero. For example, the following second order section
(unscaled) will provide the requisite behavior. 7 H ( z ) = 1 - 2
cos ( ) z - 1 + z - 2 1 - 2 r cos ( ) z - 1 + r 2 z - 2 where r is
close to 1, typically 0.9, and f 1 = 2 f s = frequency of notch
(3.2.1)
[0081] Digital Filters for Jamming Identification and
Cancellation
[0082] This filter was selected to have the minimum possible number
of numerical operations (additions, multiplications) and the
divisions for scaling to be implemented as shifts. To eliminate a
narrow-band jamming signal the invention can use a notch filter. To
determine the intensity and frequency of the jamming signal the
invention can use a band pass filter.
[0083] Clearly, it should be possible to change the center
frequency of the band pass filter over the entire digital IF
frequency range plus a guard band. In order to be able to compare
signal levels at different frequencies, the band pass filter gain
should be independent of its center frequency. The band pass filter
bandwidth was chosen to be a small fraction of the GPS signal
bandwidth; we found 50 KHz to be adequate. A reasonable tuning
range is from 100 KHz to 2.8 MHz (100 KHz is twice the band pass
filter bandwidth and 2.8 MHz is approximately half of the sampling
frequency, which is equal to 40 MHz/7).
[0084] The Notch Filter
[0085] In order to introduce a zero in the frequency response we
must have at least a pair of conjugate zeros on the unit circle in
the z plane. This sets the numerator of the filter to a
second-degree polynomial in z of the form:
N.sub.zero(z)=z.sup.2+a.sub.1.multidot.z+1 (3.2.2)
[0086] Referring to FIG. 8, an easy way to control the notch filter
bandwidth is to introduce two poles 810 close to the zeros and
inside the unit circle for a stable filter. Then the denominator of
the filter is also a second-degree polynomial in z of the form:
D(z)=z.sup.2+c.sub.1.multidot.z+c.sub.2 (3.2.3)
[0087] where 0<c.sub.2<1 and it determines the bandwidth of
the notch filter. The notch filter transfer function is given by: 8
N ( z ) = z 2 + a 1 N z + 1 z 2 - b 1 N z - b 2 N = 1 - 2 cos ( 0 )
z - 1 + z - 2 1 - 2 cos ( 0 ) z - 1 + 2 z - 2 (3.2.4)
[0088] The angular frequency, .omega., and z are related by:
z=e.sup.j.omega.T.sub..sub.s=e.sup.j.theta. (3.2.5)
[0089] where T.sub.s is the sampling period.
[0090] With f.sub.1 the notch filter center frequency, f.sub.B the
notch filter 3 dB bandwidth and f.sub.s(f.sub.s=1/T.sub.s) the
sampling frequency, the notch filter coefficients are given by: 9 0
= 2 f 1 f s a 1 N = - 2 cos ( 0 ) b 1 N = 2 cos ( 0 ) = 1 - f B f s
b 2 N = - 2 (3.2.6)
[0091] The corresponding difference equation relating the filter
input x(nT.sub.S) and filter output y(nT.sub.S) is:
y.sub.n=b.sub.1N.multidot.y.sub.n-1+b.sub.2N.multidot.y.sub.n-2+x.sub.n+a.-
sub.1N.multidot.x.sub.n-1+x.sub.n-2 (3.2.7)
[0092] Where y.sub.n=y(nT.sub.S) and x.sub.n=x(nT.sub.S).
[0093] The flow diagram depicted in FIG. 9 shows one possible
implementation of the filter defined by the equation above. In FIG.
9, z.sup.-1 represents a time delay of one sampling period and
a.sub.0=1; a.sub.1=a.sub.1N; a.sub.2=1; b.sub.1=b.sub.1N;
b.sub.2=b.sub.2N. Thus, signal transformation by the notch filter
can result in an output y.sub.n that is substantially unchanged,
except at approximately the notch filter center frequency f.sub.1
where the signal strength can be substantially zero.
[0094] To compute the output y.sub.n, we need the input x.sub.n and
the previous values x.sub.n-1, x.sub.n-2, y.sub.n-1, y.sub.n-2 that
must be stored. Every time the filter coefficients are changed it
is a good idea to initialize the stored values to zero.
[0095] The Band Pass Filter
[0096] Referring to FIG. 10A, the band pass filter can be
implemented by subtracting the output y.sub.n of the notch filter
N(z) from its input x.sub.n as in an operation 1010 to obtain the
band pass filter output u.sub.n. Thus, a band pass filter output
can be obtained that that is, within the defined upper and lower
frequency limits, substantially zero, except at approximately the
notch filter center frequency f.sub.1 where the signal strength can
be substantially the same as the input. The output y.sub.n contains
the useful part of x.sub.n where the (narrow-band) interference has
been substantially removed. The output u.sub.n contains only the
(narrow-band) interfering signal.
[0097] The filter from x.sub.n to u.sub.n will be a band pass with
same center frequency and bandwidth as the notch filter from
x.sub.n to y.sub.n.
[0098] The band pass filter transfer function is given by: 10 B ( z
) = 1 - N ( z ) = a 1 B z - 1 + a 2 B z - 2 1 - b 1 B z - 1 - b 2 B
z - 2 (3.2.8)
[0099] With f.sub.o, the band pass filter center frequency, f.sub.B
the band pass filter 3 dB bandwidth and f.sub.s(f.sub.s=1/T) the
sampling frequency, the band pass filter coefficients are given by:
11 0 = 2 f 0 f s a 1 B = 2 ( 1 - ) cos ( 0 ) a 2 B = 2 - 1 = 1 - f
B f s b 1 B = 2 cos ( 0 ) b 2 B = - 2 (3.2.9)
[0100] Both the band pass and the notch filters have the same
general form, that is, the ratio of two second-degree polynomials
in z. Referring to FIG. 10B, the same flow diagram used for the
notch filter (see FIG. 9) can be used to implement the band pass
filter by setting the coefficients as indicated below.
[0101] Where z.sup.-1 represents a time delay of one sampling
period and a.sub.0=0; a.sub.1=a.sub.1B; a.sub.2=a.sub.2B;
b.sub.1=b.sub.1B; b.sub.2=b.sub.2B. Note that since a.sub.0=0 we
could ignore the first delay and replace x.sub.n-1 by x.sub.n and
x.sub.n-2 by x.sub.n-1 saving one storage element.
[0102] Detection of a Jamming Signal
[0103] To detect the presence of a jamming signal the
receiver-microprocessor tunes a band pass filter over the required
frequency range in increments equal to, for example, approximately
half of the band pass filter bandwidth (25 KHz). At each frequency
increment, the band pass filter coefficients are computed and sent
to the FPGA from the receiver processor.
[0104] Ideally we should compute the band pass filter output power
at each frequency, but this would require the summation of the
values squared of the band pass filter output. To avoid a
multiplication (square), the sum of the absolute values of the
filter output can be used instead. This procedure will also detect
the concentration of energy at the center frequency of the band
pass filter.
[0105] The band pass filters and the summation (about 1000 terms)
of the absolute values of its output are implemented in the FPGA.
Then samples of this sum are passed to the receiver-microprocessor
where an average is computed and used as a measure of the "energy"
level. This average is then analyzed as a function of frequency to
search for peaks exceeding a preset threshold, which is determined
by the noise level without a jamming signal. Finally using the band
pass filter center frequency and the knowledge of the various local
oscillator frequencies in the GPS receiver, the frequency of the
jamming signal is established.
[0106] Filtering the Jamming Signal
[0107] To reduce the effect of a jamming signal, a notch filter is
tuned to the jamming signal frequency at the digital IF frequency,
which was determined by the detection process described above. The
notch filter coefficients are computed by the
receiver-microprocessor and sent to the FPGA, where the filter is
implemented.
[0108] Generating the Output Digital Format (Version 1)
[0109] The output of the notch filter, computed by the FPGA, is
sent to the receiver-microprocessor, which uses these values to
compute a reference-level. The receiver-microprocessor sends this
computed reference-level back to the FPGA. The FPGA uses this
reference-level to map the output of the notch filter to Zero (0)
or One (1) depending on if its absolute value is smaller or larger
than the reference-level. To complete the operation, the FPGA sends
the sign of the notch filter output and the resultant Zero (0) or
One (1) to the correlator. That is, the digital output mimics the
output of the RF chip.
[0110] The digital IF is the result of sampling the 4.309 MHz
analog IF signal at a sampling rate of (40/7)MHz (see pages 10 and
11 of the Global Positioning Products Handbook GP2000 Designer's
Guide, Ref [2]).
[0111] Number of Bits (Scaling for Finite Wordlength)
[0112] In practice the sampled signal and the filter coefficients
have to be represented by a finite number of bits. For the notch
filter to work properly (remove the jamming signal) we will need to
use an ADC with 8 to 10 bits of precision to sample the 4.309 MHz
analog IF signal.
[0113] By proper choice of coefficients the difference equation for
both the notch filter and band pass filter equation can be written
as:
1.multidot.y.sub.n=b.sub.1.multidot.y.sub.n-1+b.sub.2.multidot.y.sub.n-2+a-
.sub.0.multidot.x.sub.n+a.sub.1.multidot.x.sub.n-1+a.sub.2.multidot.x.sub.-
n-2 (3.2.10)
[0114] This difference equation can be scaled so that the
coefficients can be approximated by integers. Using K to represent
the number of bits and considering that -2<b.sub.2<2 and that
one bit is used for the sign then 1 is scaled as:
1.fwdarw.S=2.sup.K-2 (3.2.1 1)
[0115] Scaling the difference equation we have:
S.multidot.y.sub.n=S.multidot.b.sub.1.multidot.y.sub.n-1+S.multidot.b.sub.-
2.multidot.y.sub.n-2+S.multidot.a.sub.0.multidot.x.sub.n+S.multidot.a.sub.-
1.multidot.x.sub.n-1+S.multidot.a.sub.2.multidot.x.sub.n-2
(3.2.12)
[0116] Solving for y.sub.n we have: 12 y n = B 1 y n - 1 + B 2 y n
- 2 + A 0 x n + A 1 x n - 1 + A 2 x n - 2 S (3.2.13)
[0117] The division by S can be implemented by a shift since S is a
power of two.
[0118] Next we have to approximate the filter coefficients by their
nearest integers (round as indicated below).
[0119] Using the formulas below the receiver-microprocessor
computes the filter coefficients and sends them to the FPGA.
[0120] Band Pass Filter: 13 0 = 2 f 0 f s a 1 B = 2 ( 1 - ) cos ( 0
) a 2 B = 2 - 1 = 1 - f B f s b 1 B = 2 cos ( 0 ) b 2 B = - 2 S = 2
K - 1 B 1 b = Round ( S b 1 B ) = INT ( S b 1 B + .5 ) B 2 b =
Round ( S b 2 B ) = INT ( S b 2 B + .5 ) A 0 b = 0 A 1 b = Round (
S a 1 B ) = INT ( S a 1 B + .5 ) A 2 b = - ( B 2 B + S ) ( preserve
A 2 b + B 2 b = - S ) (3.2.14)
[0121] Notch Filter (Same Frequency Parameters as the Band Pass
Filter):
B.sub.1N=B.sub.1B
B.sub.2N=B.sub.2B
A.sub.0N=S
A.sub.1N=-(A.sub.1B+B.sub.1B)
A.sub.2N=S (3.2.15)
[0122] Generating the Output Digital Format (Version 2)
[0123] In describing generating the output digital format (Version
1), we described one method for converting the multi-bit signal in
the digital filters to the 2-bit format that the correlators in the
digital chip require. This previously described method was adaptive
in the sense that the processor could set the thresholds, but
otherwise that previously described method could be considered
"brute-force" and quite direct. The wordlength reduction to 2-bits
introduces quite a large quantization noise power.
[0124] In this section we describe a method based on
Delta-Sigma-Modulation or .DELTA..SIGMA.M (See Ref. [4], Chapter
8). .DELTA..SIGMA.M is a technique that shapes the quantization
noise power such that the noise spectrum is low where the desired
signal has most of its power.
[0125] The rudimentary form of word-length reduction implied in
describing generating the output digital format (Version 1) is
depicted in FIGS. 11A and 11B. The action of the quantizer 1100 is
simply to reduce the wordlength from M bits, representative of the
wordlength used in the implementation of the digital filters, to
2-bits as required by the correlators in the digital chip. This is
achieved by first deciding the sign (+or -) and second deciding
whether the magnitude is greater than a threshold, T (the threshold
for the conversion must not be confused with the threshold for
detection). The numerical values associated with the 2-bit code are
.+-.0.5T and .+-.1.5T. The receiver processor assigns the threshold
T based on the strength of the signal to achieve an adaptive
quantizer behavior. A rough algorithm is to adjust the threshold T
such that about 1/3 of the samples (approximately) correspond to
.+-.1.5T.
[0126] This form of quantization can be modeled as the addition of
"white noise" (this is the common assumption used when dealing with
quantizers) and thus the quantization noise power is uniformly
distributed over frequency. Using .DELTA..SIGMA.M techniques we can
shape the quantization noise power spectrum to be small within the
spectral region of the GPS signal, albeit large outside the signal
bandwidth. The effective filtering action of the correlator can
reduce the out-of-band noise but cannot really affect in-band
noise. Consequently this spectral shaping provides enhanced
performance. The .DELTA..SIGMA.M structure is depicted in FIG.
12.
[0127] The action of the .DELTA..SIGMA.M is to establish the error
made when a sample value is quantized and feed this error back to
influence subsequent samples. By embedding a quantizer 1200 in a
feedback loop in this manner we achieve the result of error
spectrum shaping. The notion of a register 1210 in FIG. 12 is to
store intermediate signals. The register 1210 contents during the
interval nT.sub.s through (n+1)T.sub.s, namely one cycle of the
sampling clock are associated with the value of the sample at the
n-th sampling epoch. With this in mind, the following equations
define the operation of the .DELTA..SIGMA.M structure in FIG.
12.
e(n)=w(n)-b(n)
b(n)=Q[w(n)]
w(n+1)=x(n)-e(n-1) (3.2.16)
[0128] Eq. (3.2.16) can be, with some algebra, reduced to
b(n+1)=x(n)+(e(n+1)+e(n-1)) (3.2.17)
[0129] which implies that the structure introduces a one-sample
delay and the quantization noise, e(n), appears in the output as
though it had been filtered by a first order all-zero transfer
function given by
H.sub.e(z)=1+z.sup.-2 (3.2.18)
[0130] From this we can tell that the quantization noise power has
increased by a factor of two (3 dB) but the error spectrum has a
zero at (f.sub.s/4). For the Digital IF signal, the sampling rate
is 5.714 MHz and the center frequency of the GPS component is 1.405
MHz, roughly one-fourth the sampling frequency.
[0131] Consequently this scheme provides for improvement in
signal-to-inband-quantization noise. In particular, (see Ref. [4])
we can estimate the improvement as 8.7 dB. That is, even though the
total quantization noise power is increased by a factor of 2, a
small fraction (0.0668) lies within the range 1.405.+-.0.5 MHz
which is the nominal bandwidth of the GPS signal.
[0132] Estimate of Timing Offset Caused by 1 Notch
[0133] In the previous discussion of the code tracking loop we
indicated that if there was a deviation from "ideal" in the
transmission path, the discriminant, D, would be zero at an offset
as opposed to perfect code-phase alignment. The actual timing
offset observed would be less than this theoretical offset. The
theoretical offset can indeed be estimated and a computer program
to do just that was generated. The results are shown in the Table
below. The calculations assume that there is one notch filter. The
frequency of this notch is varied over the range of from 500 kHz
below to 500 kHz above the nominal center frequency (which we refer
to as L1). The offset for each of the 32 possible Gold codes was
computed and we provide the average and standard deviation over
these 32 possibilities for each notch frequency. The time units are
nanoseconds.
[0134] (Average is over 32 SVs; variance=square of standard-dev.
over 32 SVs)
1 Notch Freq. Offset Offset (kHz from L1) (Average) (Variance)
-.5000E + 03 -.4303E + 02 .7803E + 00 -.4900E + 03 -.4144E + 02
.2996E + 01 -.4800E + 03 -.3953E + 02 .6249E + 01 -.4700E + 03
-.3741E + 02 .1043E + 02 -.4600E + 03 -.3503E + 02 .1547E + 02
-.4500E + 03 -.3253E + 02 .2025E + 02 -.4400E + 03 -.2966E + 02
.2541E + 02 -.4300E + 03 -.2663E + 02 .3273E + 02 -.4200E + 03
-.2344E + 02 .3643E + 02 -.4100E + 03 -.2003E + 02 .4091E + 02
-.4000E + 03 -.1634E + 02 .4410E + 02 -.3900E + 03 -.1256E + 02
.4675E + 02 -.3800E + 03 -.8594E + 01 .4918E + 02 -.3700E + 03
-.4438E + 01 .4993E + 02 -.3600E + 03 -.3125E + 01 .5172E + 02
-.3500E + 03 .4563E + 01 .5056E + 02 -.3400E + 03 .9313E + 01
.5046E + 02 -.3300E + 03 .1416E + 02 .4969E + 02 -.3200E + 03
.1906E + 02 .4731E + 02 -.3100E + 03 .2416E + 02 .4488E + 02
-.3000E + 03 .2919E + 02 .4053E + 02 -.2900E + 03 .3438E + 02
.3692E + 02 -.2800E + 03 .3953E + 02 .3450E + 02 -.2700E + 03
.4463E + 02 .3086E + 02 -.2600E + 03 .4963E + 02 .2705E + 02
-.2500E + 03 .5444E + 02 .2531E + 02 -.2400E + 03 .5916E + 02
.2213E + 02 -.2300E + 03 .6372E + 02 .2020E + 02 -.2200E + 03
.6831E + 02 .1834E + 02 -.2100E + 03 .7269E + 02 .1728E + 02
-.2000E + 03 .7694E + 02 .1775E + 02 -.1900E + 03 .8106E + 02
.1706E + 02 -.1800E + 03 .8500E + 02 .1681E + 02 -.1700E + 03
.8888E + 02 .1442E + 02 -.1600E + 03 .9241E + 02 .1343E + 02
-.1500E + 03 .9588E + 02 .1192E + 02 -.1400E + 03 .9909E + 02
.1052E + 02 -.1300E + 03 .1020E + 03 .9843E + 01 -.1200E + 03
.1047E + 03 .9663E + 01 -.1100E + 03 .1069E + 03 .1075E + 02
-.1000E + 03 .1090E + 03 .1091E + 02 -.9000E + 03 .1107E + 03
.1128E + 02 -.8000E + 03 .1123E + 03 .1215E + 02 -.7000E + 03
.1134E + 03 .1237E + 02 -.6000E + 03 .1145E + 03 .1281E + 02
-.5000E + 03 .1154E + 03 .1242E + 02 -.4000E + 03 .1160E + 03
.1241E + 02 -.3000E + 03 .1165E + 03 .1262E + 02 -.2000E + 03
.1169E + 03 .1268E + 02 -.1000E + 03 .1171E + 03 .1165E + 02 .0000E
+ 00 .1172E + 03 .1171E + 02 .1000E + 03 .1171E + 03 .1165E + 02
.2000E + 03 .1169E + 03 .1268E + 02 .3000E + 03 .1165E + 03 .1262E
+ 02 .4000E + 03 .1160E + 03 .1241E + 02 .5000E + 03 .1154E + 03
.1242E + 02 .6000E + 03 .1145E + 03 .1281E + 02 .7000E + 03 .1134E
+ 03 .1237E + 02 .8000E + 03 .1123E + 03 .1215E + 02 .9000E + 03
.1107E + 03 .1128E + 02 .1000E + 03 .1090E + 03 .1091E + 02 .1100E
+ 03 .1069E + 03 .1075E + 02 .1200E + 03 .1047E + 03 .9663E + 01
.1300E + 03 .1020E + 03 .9843E + 01 .1400E + 03 .9909E + 02 .1052E
+ 02 .1500E + 03 .9588E + 02 .1192E + 02 .1600E + 03 .9241E + 02
.1343E + 02 .1700E + 03 .8888E + 02 .1442E + 02 .1800E + 03 .8500E
+ 02 .1681E + 02 .1900E + 03 .8106E + 02 .1706E + 02 .2000E + 03
.7694E + 02 .1775E + 02 .2100E + 03 .7269E + 02 .1728E + 02 .2200E
+ 03 .6831E + 02 .1834E + 02 .2300E + 03 .6372E + 02 .2020E + 02
.2400E + 03 .5916E + 02 .2213E + 02 .2500E + 03 .5444E + 02 .2531E
+ 02 .2600E + 03 .4963E + 02 .2705E + 02 .2700E + 03 .4463E + 02
.3086E + 02 .2800E + 03 .3953E + 02 .3450E + 02 .2900E + 03 .3438E
+ 02 .3692E + 02 .3000E + 03 .2919E + 02 .4053E + 02 .3100E + 03
.2416E + 02 .4488E + 02 .3200E + 03 .1906E + 02 .4731E + 02 .3300E
+ 03 .1416E + 02 .4969E + 02 .3400E + 03 .9313E + 01 .5046E + 02
.3500E + 03 .4563E + 01 .5056E + 02 .3600E + 03 -.3125E - 01 .5172E
+ 02 .3700E + 03 -.4438E + 01 .4993E + 02 .3800E + 03 -.8594E + 01
.4918E + 02 .3900E + 03 -.1256E + 02 .4675E + 02 .4000E + 03
-.1634E + 02 .4410E + 02 .4100E + 03 -.2003E + 02 .4091E + 02
.4200E + 03 -.2344E + 02 .3643E + 02 .4300E + 03 -.2663E + 02
.3273E + 02 .4400E + 03 -.2966E + 02 .2541E + 02 .4500E + 03
-.3253E + 02 .2025E + 02 .4600E + 03 -.3503E + 02 .1547E + 02
.4700E + 03 -.3741E + 02 .1043E + 02 .4800E + 03 -.3953E + 02
.6249E + 01 .4900E + 03 -.4144E + 02 .2996E + 01 .5000E + 03
-.4303E + 02 .7803E + 00
[0135] An Enhanced Correlator Scheme
[0136] As mentioned before, most GPS chip-sets available today use
ADCs that provide limited precision quantization, between 2 and 4
levels. As a consequence, part of the jamming elimination circuitry
described above involved reduction of wordlength to that expected
by the (original) design of the GPS chip-set correlator. It can be
an advantageous to have an enhanced correlator, one that would
accept a multi-bit (for example 8 or 10 bits) input directly. The
architecture of such a multi-bit correlator is depicted in FIG. 13
which indicates just the "in-phase" part of the correlation
operation. For the "quadrature" part of the correlation operation,
the "cosine" generator 1310 can be replaced by a "sine"
generator.
[0137] In most chip-sets, the final down-conversion is done
digitally, multiplying the real-valued signal from the ADC by the
complex carrier of frequency f.sub.0 (see FIG. 7) to generate a
complex signal wherein the nominal "carrier frequency" is zero
(i.e., dc). In the Mitel implementation f.sub.0 is approximately
1.41 MHz. To avoid the use of a full-blown digital multiplier, this
complex carrier signal is quantized very coarsely, retaining just 1
or 1.5 or 2 bits for each of the cosine and sine parts of the
signal. In the Mitel chip the sinusoids are quantized to 4 levels
(effectively 2 bits) that, in a normalized sense, correspond to
numerical values of .+-.1 and .+-.2.
[0138] Recognizing that the Gold-code values can be treated as
.+-.1, and the quantized complex carrier can be treated as having
values of .+-.1 and .+-.2 for the real (i.e., "in-phase") and
imaginary (i.e., "quadrature") parts, the implied product of input
signal samples with the carrier and the code can be implemented
very simply as indicated in FIG. 13. In particular, the barrel
shifter shifts by 0 bits or 1 bit to get a magnitude multiplication
factor of 1 or 2. Assuming that the representation of numbers uses
the 2-s-complement scheme, multiplication by -1 is easily
accomplished by inverting all the bits as indicated by the "X-OR
block" together with controlling the carry-in bit of the subsequent
adder module. The "accumulate-and-dump" operation corresponding to
the integration over one code-period is accomplished by the
accumulator which is cleared at the start of the Gold-code period
and the value at the end of the period is loaded into the
"Correlation Register."
[0139] The use of a multi-bit correlator has two fundamental
advantages. First, if the jamming is not severe, then the notch
filters can be bypassed and the ADC values fed directly to the
correlator. This reduces the, albeit small, deleterious impact of
the notch filters altering the signal and we rely on the ability of
the correlation procedure itself to mitigate the impact of the
jamming signal. Second, if the jamming is severe enough to warrant
the inclusion of the notch filters in the signal path, having the
multi-bit correlator obviates the need to reduce the word-length of
the filter outputs to 1 or 2 bits.
[0140] A System Viewpoint of Anti-jamming
[0141] The approach to anti-jamming described herein has been
two-pronged. First is the detection mechanism. Second is the
removal mechanism. Both can be under software control. The
invention can include a system called "GPS TRACS" which provides a
graphical user interface ("GUI") so that an operator, or network
management system, can obtain information on the performance of the
GPS receiver in general. A special screen has been developed for
observing and displaying the nature and impact of interference.
[0142] The detection mechanism is "always on." The center frequency
of the band-pass filter is swept across the GPS band and thus we
have a continually updated view of the interference situation. Two
modes of operation are provided in the event that significant
jamming power is detected at any particular frequency. These are
the "manual" and "automatic" modes of operation. In the automatic
mode, if a significant jammer is detected, a notch-filter is
invoked to remove the interfering signal. In the manual mode,
operator intervention is called for in order to invoke the
instantiation of a notch filter, with the system determining the
appropriate frequency.
[0143] The invention can also be included in a kit. The kit can
include some, or all, of the components that compose the invention.
The kit can be an in-the-field retrofit kit to improve existing
systems that are capable of incorporating the invention. The kit
can include software/firmware and/or hardware for carrying out the
invention. The kit can also contain instructions for practicing the
invention. The components, software, firmware, hardware and/or
instructions of the kit can be the same as those used in the
invention.
[0144] The term approximately, as used herein, is defined as at
least close to a given value (e.g., preferably within 10% of, more
preferably within 1% of, and most preferably within 0.1% of). The
term substantially, as used herein, is defined as at least
approaching a given state (e.g., preferably within 10% of, more
preferably within 1% of, and most preferably within 0.1% of). The
term coupled, as used herein, is defined as connected, although not
necessarily directly, and not necessarily mechanically. The term
deploying, as used herein, is defined as designing, building,
shipping, installing and/or operating. The term means, as used
herein, is defined as hardware, firmware and/or software for
achieving a result. The term program or programmable or
programming, as used herein, is defined as a sequence of
instructions designed for execution on a computer system. A
sequence of instructions, may include a subroutine, a function, a
procedure, an object method, an object implementation, an
executable application, an applet, a servlet, a source code, an
object code, a shared library/dynamic load library and/or other
sequence of instructions designed for execution on a computer
system. The terms including and/or having, as used herein, are
defined as comprising (i.e., open language). The terms a or an, as
used herein, are defined as one or more than one. The term another,
as used herein, is defined as at least a second or more.
Advantages of the Invention
[0145] The invention can be cost effective and advantageous for at
least the following reasons. The anti-jamming principle and methods
described above can be summarized into the following key
innovations:
[0146] The invention has application with existing chip-sets. The
methods for detecting and eliminating jamming signals described
here are appropriate for incorporation into GPS receivers using
currently available chip-sets.
[0147] The invention permits CW detection and removal of jamming.
The method is based on applying a digital band-pass filter, with
programmable center frequency, so as to detect the presence and
strength of jamming signals (if any). A notch filter, with
programmable notch frequency, can be invoked to remove the jamming
signal. Whereas the detection is an ongoing process, the notch
filter remains in place until removed. The programming of frequency
and insertion/removal of notch filter(s) is under software
control.
[0148] The invention can include calibration. The insertion of a
notch filter introduces a shift in the (time) position of the 1-PPS
signal. We have shown the bounds of the maximum expected "error"
and it is possible to calibrate the receiver to compensate for the
"error."
[0149] The invention can include enhanced quantization. Existing
chip-sets use correlators that expect the ADC word length to be 1
or 2 bits. As part of the signal processing chain in the (external
to the chip-set) circuitry (i.e., in the FPGA) we include an
enhanced quantizer of the .DELTA..SIGMA.M type to reduce the impact
of quantization noise. In particular, the quantization noise is
shaped so that the majority of the quantization noise power lies
outside the bandwidth occupied by the Gold-code signal of GPS.
[0150] The invention can include an enhanced correlator. For future
designs of GPS correlators, we have described a multi-bit
correlator architecture. This would obviate the need for the coarse
quantization of the notch filter output and, further, if the
jamming power is small, the multi-bit correlator by itself would
provide the jamming immunity.
[0151] The invention can include system operation in an automatic
mode. As part of the control software, the notion of automatic
versus manual introduction of notch filters provides a valuable
tool for the system designer. If the GPS receiver must operate in
an autonomous manner, with little intervention by an operator, the
automatic mode is appropriate. By allowing a manual mode, the
operator is provided flexibility regarding the provisioning of the
notch filters.
[0152] All the disclosed embodiments of the invention disclosed
herein can be made and used without undue experimentation in light
of the disclosure. Although the best mode of carrying out the
invention contemplated by the inventor(s) is disclosed, practice of
the invention is not limited thereto. Accordingly, it will be
appreciated by those skilled in the art that the invention may be
practiced otherwise than as specifically described herein.
[0153] Further, the individual components need not be combined in
the disclosed configurations, but could be combined in virtually
any configuration. Further, variation may be made in the steps or
in the sequence of steps composing methods described herein.
[0154] Further, although the GPS receiver described herein can be a
separate module, it will be manifest that the GPS receiver may be
integrated into the system with which it is (they are) associated.
Furthermore, all the disclosed elements and features of each
disclosed embodiment can be combined with, or substituted for, the
disclosed elements and features of every other disclosed embodiment
except where such elements or features are mutually exclusive.
[0155] It will be manifest that various substitutions,
modifications, additions and/or rearrangements of the features of
the invention may be made without deviating from the spirit and/or
scope of the underlying inventive concept. It is deemed that the
spirit and/or scope of the underlying inventive concept as defined
by the appended claims and their equivalents cover all such
substitutions, modifications, additions and/or rearrangements.
[0156] The appended claims are not to be interpreted as including
means-plus-function limitations, unless such a limitation is
explicitly recited in a given claim using the phrase(s) "means for"
and/or "step for." Subgeneric embodiments of the invention are
delineated by the appended independent claims and their
equivalents. Specific embodiments of the invention are
differentiated by the appended dependent claims and their
equivalents.
REFERENCES
[0157] [1] Understanding GPS Principles and Applications, Elliott
D. Kaplan (Ed.), Artech House Publishers, Boston/London, 1996.
ISBN: 0-89006-793-7.
[0158] [2] GEC Plessey Semiconductors, Global Positioning Products
Handbook, August 1996.
[0159] [3] Frank Amoroso and Jacob Bricker, "Performance of the
Adaptive A/D Converter in Combined CW and Gaussian Interference,"
IEEE Transaction on Communications, VOL. COM-34, No. 3, March
1986.
[0160] [4] Kishan Shenoi, Digital Signal Processing in
Telecommunications, Prentice-Hall Inc, Upper Saddle River, N.J.,
1995. ISBN: 0-13-096751-3.
[0161] [5] D. Moulin, M. N. Solomon, T. M. Hopkinson, P. T.
Capozza, J. Psilos, "High-Performance RF-to-Digital Translators for
GPS Anti-Jam Applications."
[0162] [6] R. L. Fante, J. J. Vaccaro, "Enhanced Anti-Jam
Capability for GPS Receivers."
[0163] [7] M. Zhodzishsky, D. Chemiavsky, A. Kirsanov, M. Vorobiev,
V. Prasolov, A. Zhdanov, J. Ashjaee, "In-Band Interference
Suppression for GPS/GLONASS."
[0164] [8] S. V. Lyusin, L. B. Sazonov, I. G. Khazanov, A. S.
Komarov, "Combined GPS/GLONASS Receiver with high Antijamming
Performance."
[0165] [9] J. M. Lopez-Almansa, P. A. Pablos, "Measurement Error
and Protection Envelopes in the Presence of Interference for GNSS
Receivers."
[0166] [10] The Electrical Engineering Handbook, CRC Press,
(Richard C. Dorf et al. eds., 1993).
* * * * *