U.S. patent application number 09/886233 was filed with the patent office on 2001-10-25 for mosfet mixer for low supply voltage.
Invention is credited to Chien, Hwey-Ching.
Application Number | 20010033193 09/886233 |
Document ID | / |
Family ID | 24197990 |
Filed Date | 2001-10-25 |
United States Patent
Application |
20010033193 |
Kind Code |
A1 |
Chien, Hwey-Ching |
October 25, 2001 |
MOSFET mixer for low supply voltage
Abstract
A MOSFET operating as a mixer has its drain biased at the knee
of the I.sub.D vs V.sub.DS characteristic. A local oscillator
voltage is applied to the gate and a RF signal voltage is applied
to the drain through a singled-ended source follower. The nonlinear
curvature at the knee produces a beat frequency current. This mixer
requires less supply voltage, and results in more conversion gain
and less feed-through of the RF input signal than the Gilbert
multiplier. Conversely, the RF voltage can be applied to the gate
and the local oscillator voltage can be applied to the drain.
Inventors: |
Chien, Hwey-Ching; (San
Diego, CA) |
Correspondence
Address: |
Hung Chang Lin
8 Schindler Court
Silver Spring
MD
20903
US
|
Family ID: |
24197990 |
Appl. No.: |
09/886233 |
Filed: |
June 22, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
09886233 |
Jun 22, 2001 |
|
|
|
09550638 |
Apr 17, 2000 |
|
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Current U.S.
Class: |
327/355 |
Current CPC
Class: |
H03D 2200/0084 20130101;
H03D 7/1441 20130101; H03D 2200/0043 20130101; G06G 7/14 20130101;
H03D 7/1433 20130101; H03D 7/125 20130101 |
Class at
Publication: |
327/355 |
International
Class: |
G06G 007/12 |
Claims
1. A mixer circuit using MOS field effect transistors (MOSFET)
having a threshold voltage Vth, a drain characteristic with an
ohmic region where the drain current I.sub.D increases with
increasing drain-to-source voltage V.sub.DS, and current saturation
region where the drain current is constant with increasing V.sub.DS
for a fixed gate-to-source voltage, comprising: a first MOSFET
having a first source, a first gate and a first drain; a ground
supply voltage applied to said first source; a first dc quiescent
gate voltage applied to said first gate; a first radio frequency
(RF) voltage V1 superimposed on said first dc quiescent gate
voltage; a second MOSFET operating as a single ended source
follower, having a drain connected to a dc power supply, a source
dc-connected to to said first drain of said first MOSFET, and a
gate: applied with a dc gate voltage such that the dc
drain-to-source voltage of the said first MOSFET is equal to a
transition (knee) voltage V.sub.knee between said ohmic region and
said current saturation region below said current saturation region
of the drain characteristic of the said first MOSFET, and
superimposed with a second RF voltage V2; and a means to sense the
drain current of said second MOSFET having a frequency component
equal the beat frequency of said V1 and said V2.
2. The mixer circuit as described in claim 1, wherein said
V.sub.knee is equal to said dc quiescent gate voltage minus said
threshold voltage.
3. The mixer circuit as described in claim 1, wherein said means to
sense is a resistor.
4. The mixer circuit as described in claim 1, wherein said means to
sense is an inductor.
5. The mixer circuit as described in claim 4 further comprising a
capacitor in parallel with aid inductor to form a parallel resonant
circuit, resonant at said beat frequency.
6. The mixer circuit as described in claim 1, wherein said first
MOSFET and said second MOSFET are N-channel field effect
transistors and said dc supply voltage is positive.
7. A mixer circuit as describe in claim 1, wherein said means to
sense is a current meter.
8. A mixer circuit as described in claim 1, wherein said first RF
voltage is an incoming signal voltage and said second RF voltage is
a local oscillator voltage of a superheterodyne radio receiver.
9. A mixer circuit as described in claim 1, wherein said second RF
signal is an incoming signal voltage and said first RF signal is a
local oscillator voltage of a superheterodyne receiver.
10. A mixer circuit as described in claim 1, wherein the first
MOSFET and the second MOSFET are of equal size, the dc drain
voltage of the second MOSFET is at least one threshold voltage less
than the dc gate voltage, and the dc gate voltage of said second
MOSFET is equal to twice the dc gate-to-source of said first MOSFET
minus the threshold voltage (Vt) of said second MOSFET so that the
dc drain-to-source voltage of said first MOSFET is automatically
biased to the V.sub.knee.
11. A mixer circuit as described in claim 10, wherein the dc
gate-to-source of said first MOSFET is obtained by dividing the dc
gate supply voltage of said second MOSFET into one half and adding
one half of a threshold voltage.
12. A mixer circuit as described in claim 1, wherein the first
MOSFET and second MOSFET are of equal size, the dc drain voltage of
the second MOSFET is at least one threshold volage less than the de
gate voltage, and the dc gate voltage of said second MOSFET is
equal to [1+{square root}(K1/K2)] times the de gate voltage of said
first MOSFET minus ({square root}(K1+K2) times the threshold
voltage of said second MOSFET, where K1 and K2 are the
transconductance parameters of the first MOSFET and the second
MOSFET, respectively, so that the dc drain-to-source voltage of
said first MOSFET is automatically biased the knee voltage
V.sub.knee.
Description
[0001] This is a continuation-in-part patent application of U.S.
patent application Ser. No. 09/550,638 filed Apr. 17, 2000, now
abandoned.
BACKGROUND OF THE INVENTION
[0002] (1) Field of Invention
[0003] This invention relates to mixers using MOSFETs, in
particular the down converter of a superheterodyne radio
receiver
[0004] (2) Description of the Related Art
[0005] In a conventional radio receiver, the incoming radio
frequency is mixed with a local oscillator (LO) signal to produce a
beat frequency, which is the intermediate frequency (IF). The IF is
then amplified and filtered to attenuate other unwanted
signals,
[0006] A popular mixer circuit is the Gilbert multiplier. Since
MOSFETs are widely used in circuit designs today, an MOSFET version
of the Gilbert multiplier is shown in FIG. 1. Basically, a
differential amplifier with a differential pair N2 and N3 is fed
from a current source N1. The differential gain of the differential
amplifier is proportional to the transconductance gm of N2 and N3.
This transconductance varies as the square root of the dc drain
current of N2 and N3, which is controlled by the drain current of
N1. The dc drain current I.sub.D1of N1 is controlled by the dc
gate-to-source voltage V.sub.GS1 of N1 and has a square law
relationship with the gate-to-source voltage i.e.
I.sub.D1.sup.OC(V.sub.GS1-Vt).sup.2, where Vt is the threshold
voltage of N1. When a local oscillator signal V.sub.LO of frequency
f.sub.LO is applied differentially to N2 an N3 (i.e. V.sub.LO+ and
V.sub.LO- respectively),and a radio frequency signal V.sub.rf of
frequency f.sub.rf is applied to the gate of N1, the output current
of the differential amplifier is equal to V.sub.LO*gm, and the gm
is proportional to Vrf.sup.*(V.sub.GS1-Vt). When the V.sub.rf is
multiplied by V.sub.LO, a beat frequency
f.sub.if=f.sub.rf.+-.f.sub.LO intermediate frequency signal
V.sub.if is produced.
[0007] While the Gilbert multiplier is widely used in the past, it
has a number of drawbacks for low voltage and low power
applications. In modern CMOS technology, the tendency is to use a
low supply voltage V.sub.DD: for instance 25 V for 0.25 .mu.m
technology and 1.8 V for 0.18 .mu.m technology. In the Gilbert
mixer, the current source is operating in the current saturation
region of the V.sub.DS vs I.sub.D V-I characteristic N1' in FIG. 2
to obtain a higher transconductance and has a square law relation
with V.sub.rf. Therefore the drain voltage V.sub.DS1 for the
current source N1 is larger than the knee voltage V.sub.D1' of the
N1' V-I characteristic curve. That knee voltage V.sub.D1' is equal
to V.sub.GS1-Vt.
[0008] Similarly, the differential pair N2 and N3 also must have
its drain voltage higher than the knee voltage, i.e.
V.sub.D2>2(V.sub.GS1-Vt) as shown by the dotted V-I
characteristic of N2 in FIG. 2. If a resistor is used as a load,
another voltage drop VL will be added to V.sub.DS1 to be supplied
by the power supply V.sub.DD. These three stacks of voltages,
V.sub.DS1, V.sub.DS2 and VL, dictate that the supply voltage cannot
be made very low. For a typical threshold voltage of 0.6 V, there
is hardly any "head room" for signal voltage swing.
[0009] Lee et al disclosed in U.S. Pat. No. 6,194,947 a mixer
structure which is basically a Gilbert mixer having a differential
pair fed from a current source with its shortcomings.
[0010] Sakusabe disclosed in U.S. Pat. No. 5,789,963, FIGS. 1 &
9 ", a mixer operating with a drain to source voltage V.sub.DS in
the current saturation region of a MOSFET without claiming the
exact V.sub.DS. The RF signal is injected to the drain of the mixer
by AC coupling (i.e. through a coupling capacitor). The AC coupling
requires many additional components such the coupling capacitor Ca
and other components such as Z1-Z8 and capacitors C1-C5 as shown in
Sakusabe's FIG. 1
[0011] Another drawback of Sakusabe's mixer is that the gate of the
mixer FET2 must be adjusted to set the quiescent drain voltage to
the current saturation region for different operating currents. It
is desirable to set the set the quiescent operating point (i.e.
V.sub.DS) automatically for different operating currents.
SUMMARY OF THE INVENTION
[0012] An object of this invention is to design a MOSFET mixer
which requires a lower supply voltage than the Gilbert mixer or
similar structure. Another object of this invention is to reduce
the power consumption of the MOSFET mixer. Still another object of
this invention is to provide a high conversion gain of the mixer. A
further object of this invention is to set the operating point of
the mixer at its optimum conversion gain automatically.
[0013] These objects are achieved by mixing the RF signal and the
local oscillator signal at the knee of the output V.sub.DS-I.sub.D
characteristic of a MOSFET by dc coupling. At the knee, the
characteristic has the sharpest curvature. The nonlinearity
produces a maximum beat frequency signal. For implementation, a
mixer MOSFET is biased at the knee of the V.sub.DS-I.sub.D
characteristic. The LO (or RF) signal voltage V.sub.LO (or
V.sub.rf) is applied at the gate of the mixer MOSFET, and the RF
(or LO) signal voltage V.sub.rf(or V.sub.LO) is injected at the
drain of the mixer MOSFET. Then a beat frequency drain current is
produced. Specifically, the gate of a single-ended mixer MOSFET is
fed with a local oscillator signal and the drain of the mixer is dc
coupled to a single-ended source follower with the gate fed from a
radio frequency signal or vise versa.
BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS
[0014] FIG. 1 shows a prior art Gilbert mixer.
[0015] FIG. 2 shows the output V-I characteristic of MOSFETs.
[0016] FIG. 3 shows the basic circuit of the present invention.
[0017] FIG. 4 shows the basic circuit with an inductive load.
[0018] FIG. 5 shows a dc biasing circuit for automatically biasing
the mixer MOSFET to the drain knee voltage.
DETAILED DESCRIPTION OF THE INVENTION
[0019] The basic circuit of the present invention is shown in FIG.
3. Two N-channel MOSFETs N2' and N1' are connected in series. The
pull-down NMOS N1' has its source grounded and its drain D1
connected to the source of the pull-up NMOS N2. The drain D2 of N2'
is connected through a load R.sub.L to the positive power supply
V.sub.DD. The RF voltage V.sub.rf is applied to the gate of N1',
which has a dc bias voltage V.sub.G1. The drain D1 of the N1' is
set at a quiescent voltage V.sub.knee at the knee of the
V.sub.DS-I.sub.D characteristic of N1'. This voltage is
V.sub.knee=(V.sub.G1-Vt), where Vt is the threshold voltage of N1'.
The appropriate V.sub.knee is set by choosing the appropriate dc
gate voltage V.sub.G2 of N2'.
[0020] A LO signal voltage V.sub.LO is applied to the gate of N2',
and an RF voltage is applied to the gate of N1'. At the knee of the
V.sub.DS-I.sub.D characteristic of N1', the dc current:
I.sub.D=(K/2)[2(V.sub.GS-Vt)V.sub.DS-V.sub.DS.sup.2] (1)
[0021] where K is a transconductance parameter
[0022] The slope is:
.DELTA.I.sub.D/.DELTA.V.sub.DS=K(V.sub.GS-Vt-V.sub.DS) (2)
[0023] Let a sine wave RF signal V.sub.rf sin .omega..sub.rft be
applied to the gate of N1' and a sine wave LO signal V.sub.LO sin
.omega..sub.LOt be applied to the gate of N2'. Due to source
follower action, .DELTA.VDS=VLO sin .omega..sub.LOt. Then equation
(2) can be written as:
.DELTA.I.sub.D=K[(V.sub.GS+V.sub.rf sin
.omega..sub.rft)-Vt-V.sub.DS]V.sub- .LO sin
.omega..sub.LOt=KV.sub.LO sin .omega..sub.LO t.sup.*V.sub.rf sin
.omega..sub.rft+K(V.sub.GS-Vt-V.sub.DS).sup.*V.sub.LO sin
.omega..sub.LOt (3)
[0024] The first term on the right side of equation (3) is a cross
product term which yields a beat frequency signal V.sub.if
[0025] V.sub.if=KV.sub.LOV.sub.rf[ cos
(.omega..sub.rf-.omega..sub.LO)t]- /2 (4)
[0026] The sum frequency KV.sub.LOV.sub.rf[ cos
(.omega..sub.rf+.omega..su- b.LO)t]/2 term in equation (3) can be
filtered out in the IF amplifier. The dc term
(V.sub.GS-Vt-V.sub.DS) is equal to zero at the knee.
[0027] The foregoing analysis also holds true if the local
oscillator voltage V.sub.LO and the RF voltage V.sub.rf are
interchanged.
[0028] For comparison with the Gilbert multiplier, it is assumed
that I.sub.D1 is the same in both cases, and all three MOSFETs N1,
N2 and N3 in the Gilbert mixer as shown in FIG. 1 are the same.
Then, the dc currents I.sub.D2=I.sub.D3=I.sub.D1/2. The ac
differential output current is:
I.sub.d2=2gm.sub.2.sup.*V.sub.LO/2 (5)
[0029] where gm.sub.2 is the transconductance of N2 and
gm.sub.2=(2K I.sub.D2).sup.1/2=(KI.sub.D1).sup.1/2 (6)
[0030] Combining (5) and (6), the ac output current becomes:
I.sub.d2=V.sub.LO(KI.sub.D1).sup.1/2 (7)
[0031] With the RFvoltage V.sub.g1 at the gate of N1,
I.sub.D1=K(V.sub.GS1+V.sub.g1-Vt).sup.2/2 (8)
[0032] Let V.sub.g1=V.sub.rf sin .omega..sub.rft and
V.sub.LO(t)=V.sub.LO sin .omega..sub.LOt (9)
[0033] Combine equations (7), (8) and (9)
I.sub.d2=K V.sub.LOf sin .omega..sub.LOt.sup.*(V.sub.rf sin
.omega..sub.rft+V.sub.GS1-Vt)/2.sup.1/2 (10)
[0034] Note that the cross product term KV.sub.rf sin
.omega..sub.rft.sup.*V.sub.LO sin .omega..sub.LOt/2.sup.1/2 of
equation (10) is 1/2.sup.1/2 that of the cross product in equation
(3). Therefore the present invention shown in FIG. 3 has 2.sup.1/2
times the conversion gain of the Gilbert mixer.
[0035] In addition, the present invention has the advantage of a
lower V.sub.DS1=V.sub.GS1-Vt, while the Gilbert mixer must use a
V.sub.DS1>V.sub.GS1-Vt. Furthermore, the Gilbert's differential
pair must be operating in the current saturation region to obtain
high transconductance, i.e. N2 and N3 also require high drain
voltage V.sub.DS2>V.sub.GS2-Vt. On the other hand, N2' of the
present invention acts like a transmission gate, and the dc drain
to source voltage V.sub.DS2' is equal to the voltage drop across
the on resistance Ron of N2'. This voltage drop
V.sub.DS2'(.apprxeq.I.sub.D2'Ron) [is] can be less than
(V.sub.GS2'-Vt) as shown by the load line of N2' in FIG. 2 and can
be made low by reducing Ron (i.e. large width to length ratio of
the gate of N2'). This reduction in V.sub.DS1' and V.sub.DS2' for
FIG. 3 translates into a lower supply voltage V.sub.DD, as indicted
by V.sub.D2(min) for the Gilbert mixer and V.sub.D2' for the
present invention. The characteristic of N2' in FIG. 2 assumes that
N2' is twice as wide as N1' and equal to the total width of N2 and
N3 of the Gilbert mixer.
[0036] Note also the RF component V.sub.rf sin
.omega..sub.rft(V.sub.GS1-V- t-V.sub.DS) in equation (3) for the
present invention is equal to zero, because (V.sub.GS1-Vt-V.sub.DS)
is equal to zero at the knee. Conversely in equation (10) for the
Gilbert mixer, the RF component V.sub.rf sin
.omega..sub.rft(V.sub.GS1-Vt) is not equal to zero, and must be
filtered out to avoid intermodulation. On the other hand, , the RF
component in the present invention is equal to zero and need not be
filtered. The elimination of filtering the RF component is an
advantage. Although the image frequency
(.omega..sub.rf+.omega..sub.LO) is still present, this frequency is
far away from the beat frequency (.omega..sub.rf-.omega..sub- .LO)
and can be filtered out more easily than the lower RF signal
.omega..sub.rf. Thus, there is less feed-through of the input RF
signal for the present invention than the Gilbert mixer, and the
ratio of the magnitudes of the image frequency to the beat
frequency is no worse than the Gilbert mixer.
[0037] In the foregoing analysis, it is assumed that the RF signal
V.sub.rf is applied to the gate of N1' and the local oscillator
voltage V.sub.LO is applied to the gate of N2'. From equation (4),
the beat frequency IF voltage is proportional the product
V.sub.rf.sup.*V.sub.LO. Therefore, from a theoretical standpoint
where the voltage gain the source follower N2' is assumed to be
unity, the conversion gain should be the same when V.sub.LO is
applied to N1' and V.sub.rf is applied to N2'. In practice,
however, the gain of the source follower N2' is less than unity.
When V.sub.rf is applied to the gate of N2', the RF signal at the
source of N2' is less than the signal at the gate. In either case,
the noise output should be the same. As a result, the
signal-to-noise ratio at the output of the mixer can be better for
the case where the RF signal is applied to N1' and the local
oscillator voltage is applied to N2'.
[0038] While the load device shown in FIG. 3 is a resistor with a
voltage drop, the power supply voltage V.sub.DD can further be
reduced by using an inductive load device as shown in FIG. 4. The
inductance L can be connected in parallel with a capacitor to form
a tank circuit resonant at the beat or intermediate frequency
f.sub.if.
[0039] Besides the lower supply voltage and reduced RF feedthrough,
the single-ended mixer of the present invention also has some other
advantages over the balanced structure of the Gilbert circuit. A
balanced structure requires differential RF and LO inputs and IF
outputs. Most applications only provide single-ended RF and LO
inputs and preferably single-ended IF output. As a result, the user
need to provide single-to differential buffer amplifiers for RF and
LO, and differential-to-single-- ended buffer for the IF.
Otherwise, differential filters are needed. Buffer amplifiers
consume power while differential filters cost more and sometimes
not available.
[0040] In the present invention, the dc knee voltage V.sub.knee of
the first mixer MOSFET N1', V.sub.G1-Vt, is equal to the dc source
voltage V.sub.S2 of the source follower N2', i.e.
V.sub.S2=V.sub.G1-Vt. For equal size N1' and N2' and equal source
currents and V.sub.D2.gtoreq.V.sub.G2-V- t,
V.sub.G1-Vt=V.sub.G2-V.sub.S2-Vt. Combining these relations
yields
V.sub.G2=2V.sub.GS1-Vt. (11)
[0041] So long as this relationship is satisfied, the mixer N1' is
automatically biased to the knee without adjustment, regardless of
the drain or source current of the mixer MOSFET N1'. FIG. 5 shows a
circuit for satisfying this dc requirement. In this circuit, the
supply voltage V.sub.G2 is divided into one half by a voltage
divider of two equal R1+R2. The divided voltage V.sub.G2/2 is added
with a divided threshold voltage Vt/2 across the MOSFET N4 in diode
connection, which has a threshold voltage Vt from drain to source,
to constitute the dc gate bias V.sub.G1. The dc drain voltage
V.sub.D2 can be lower than (V.sub.G2-Vt) as explained previously.
As can be seen from FIG. 5, only a low supply voltage V.sub.G2 less
than twice the V.sub.G1 is needed. The RF voltage and local
oscillator voltage can readily be superimposed on VG1 and VG2 (not
shown). For unequal size N1' and N2', another condition similar to
eq.(11) can be derived as as follows: V.sub.G2=[1+{square
root}(K1/K2)] V.sub.G1.sup.-[{square root}(K1/K2)]Vt (12)
[0042] where K1 and K2 are the transconductance parameters as
defined in eq.(1).
[0043] V.sub.G2 is the minimum dc supply voltage required to
operate the present invention, and is compared favorably to the
Gilbert mixer, because the dc drain voltage of N2 or N3 used for
the Gilbert mixer shown in FIG. 1 requires a higher drain voltage
than (V.sub.G2-Vt) for proper operation of N2 and N3 in the current
saturation region, the dc drain voltage of N1 requires a higher dc
drain voltage than N1' for proper operation in the current
saturation region, and the load R connected to V.sub.DD requires
additional voltage drop. Thus the required dc voltage supply
V.sub.DD for the Gilbert mixer is much higher than the dc supply
voltage of the present invention. If V.sub.G2 is used as the supply
voltage V.sub.DD for N2' and the dc drain voltage V.sub.D2 of N2'
need only be less than (VG.sub.2-Vt), then a large load as shown in
FIG. 3 can be inserted between V.sub.DD and V.sub.D2 to obtain more
conversion gain. More importantly, the present invention has a
higher conversion gain than the Gilbert mixer.
[0044] While the forgoing mixer is described using MOSFETs, similar
technique should be applicable to bipolar transistors.
[0045] While the preferred embodiments of this invention have been
described, it will be apparent to those skilled in the art that
various modifications may be made in the embodiments without
departing from the spirit of the present invention. Such
modifications are all within the scope of this invention.
* * * * *