U.S. patent application number 09/778854 was filed with the patent office on 2001-10-11 for integrated beamforming/rake/mud cdma receiver architecture.
Invention is credited to Bierly, Scott, Harlacher, Marc, Smarrelli, Robert, Weinberg, Aaron.
Application Number | 20010028675 09/778854 |
Document ID | / |
Family ID | 27391429 |
Filed Date | 2001-10-11 |
United States Patent
Application |
20010028675 |
Kind Code |
A1 |
Bierly, Scott ; et
al. |
October 11, 2001 |
Integrated beamforming/rake/mud CDMA receiver architecture
Abstract
A spread-spectrum demodulator architecture is presented which
utilizes parallel processing to accomplish rapid signal acquisition
with simultaneous tracking of multiple channels, while implementing
an integrated multi-element adaptive beamformer, Rake combiner, and
multi-user detector (MUD). A matched filter computational
architecture is utilized, in which common digital arithmetic
elements are used for both acquisition and tracking purposes. As
each channel is sequentially acquired by the parallel matched
filter, a subset of the arithmetic elements are then dedicated to
the subsequent tracking of that channel. Additionally, multiple
data inputs and delay lines are present, connecting the sampled
baseband data streams of numerous RF bands and antenna elements
with the arithmetic elements. The matched filter/despreader
processing is virtually independent of channel origin or
utilization; e.g., CDMA users, RF bands, beamformer elements, or
Rake Fingers. Integration of the beamformer weighting computation
with the demodulator results in substantial savings by sharing the
existing circuitry performing carrier tracking and AGC. An optimal
demodulator solution can be achieved through unified "space/time"
processing, by providing all observables (element snapshots, Rake
Fingers, carrier/symbol SNR/phase, etc.), for multiple channels, to
a single adaptive algorithm processor that can beamform, Rake, and
perform joint detection (MUD).
Inventors: |
Bierly, Scott; (Oak Hill,
VA) ; Harlacher, Marc; (Herndon, VA) ;
Smarrelli, Robert; (Oak Hill, VA) ; Weinberg,
Aaron; (Potomac, MD) |
Correspondence
Address: |
Law Office of Jim Zegeer
Suite 108
801 North Pitt Street
Alexandria
VA
22314
US
|
Family ID: |
27391429 |
Appl. No.: |
09/778854 |
Filed: |
February 8, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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09778854 |
Feb 8, 2001 |
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09707909 |
Nov 8, 2000 |
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60181571 |
Feb 10, 2000 |
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Current U.S.
Class: |
375/143 ;
375/E1.012; 375/E1.018; 375/E1.032 |
Current CPC
Class: |
H04B 1/7093 20130101;
H04B 2001/70706 20130101; H04B 2201/7071 20130101; H04B 1/708
20130101; H04B 1/7117 20130101 |
Class at
Publication: |
375/143 |
International
Class: |
H04K 001/00 |
Claims
What is claimed is:
1. A multiple frequency band, multiple channel radio receiver
comprising a front end circuitry for providing complex base band
samples for a plurality of frequency bands and multiple channels in
digital format, a parallel digital matched filter in which common
digital arithmetic elements are used for both acquisition and
tracking purposes connected to said front end circuitry, said
matched filter being arranged to perform a plurality of
simultaneous correlations of received spread spectrum signals
against selected replica offsets of a spreading sequence, said
parallel digital matched filter providing N slices with M stages
per slice and W bit data quantization, each slice being adapted to
perform 1/N of the acquisition computation and then is handed off
to become a dedicated tracking module for one channel,
respectively, and an integrated multi-element adaptive digital
beamformer combiner connected to said parallel digital matched
filter.
2. The multiple frequency band, multiple channel radio receiver
defined in claim 1 wherein said matched filter includes an N*M
stage data delay line composed of B distinct bands, each band being
composed of E distinct elements of 2*W bits each and a single
numerically controlled oscillator to serve as a control digital
frequency source matched to the expected chipping rate of the
incoming signal.
3. The multiple frequency band, multiple channel radio receiver
defined in claim 1 including a Rake Combiner for mitigating the
effects of multipath interference dominated communication channels
connected to said parallel digital matched filter.
4. A multiple frequency band, multiple channel radio receiver
comprising a front end circuitry for providing complex base band
samples for a plurality of frequency bands and multiple channels in
digital format, a matched filter in which common digital arithmetic
elements are used for both acquisition and tracking purposes
connected to said front end circuitry, said matched filter being
arranged to perform a plurality of simultaneous correlations of
received spread spectrum signals against selected replica offsets
of a spreading sequence, said parallel digital matched filter
providing N slices with M stages per slice and W bit data
quantization, each slice being adapted to perform 1/N of the
acquisition computation and then is handed off to become a
dedicated tracking module for one channel, respectively, and
integrated multi-element adaptive digital beamformer and Rake
Combiners connected to said matched filter.
5. The multiple frequency band multiple channel radio receiver
defined in claim 4 including a multi-channel demodulator connected
to said matched filter for simultaneously processing all bands,
elements and channels and Rake Fingers.
6. A multiple frequency band, multiple channel radio receiver
comprising a front end circuitry for providing complex base band
samples for a plurality of frequency bands and multiple channels in
digital format, a matched filter in which common digital arithmetic
elements are used for both acquisition and tracking purposes
connected to said front end circuitry, said matched filter being
arranged to perform a plurality of simultaneous correlations of
received spread spectrum signals against selected replica offsets
of a spreading sequence, said parallel digital matched filter
providing N slices with M stages per slice and W bit data
quantization, each slice being adapted to perform 1/N of the
acquisition computation and then is handed off to become a
dedicated tracking module for one channel, respectively, including
a multi-channel demodulator processor connected to said matched
filter for simultaneously processing all bands, elements and
channels, and to function as an integrated multi-element adaptive
digital beamformer combiner.
7. The multiple frequency band, multiple channel radio receiver
defined in claim 6 wherein said multi-channel demodulator processor
includes a defragmentation module to insure the maximum acquisition
capability of said receiver is maintained over time.
8. The multiple frequency band, multiple channel radio receiver
defined in claim 6 including a single clocking system synchronous
to a data sampling clock for generating G independent
NCO/PN-generators that produce PN chipping sequences whose average
rates can precisely track the various received signal chipping
rates, respectively.
9. The multiple frequency band, multiple channel radio receiver
defined in claim 6 wherein said multi-channel demodulator processor
is adapted to form a completely independent set of beams on each
single channel on each Rake Finger of each signal channel.
10. The multiple frequency band, multiple channel radio receiver
defined in claim 6 wherein as each channel is sequentially acquired
by said matched filter, and said demodulator processor assures that
common digital arithmetic elements are used both for acquisition
and tracking purposes, respectively.
11. The multiple frequency band, multiple channel radio receiver
defined in claim 6 wherein multiple data inputs and delay lines are
present and are available for processing at each arithmetic element
so that the matched filter/despreader processing is virtually
independent of channel origin (e.g. CDMA users, beamform element,
or Rake Fingers).
12. The multiple frequency band, multiple channel radio receiver
defined in claim 6 wherein there is a common NCO/PN-generator
within a common beamformer element set.
13. A multiple frequency band, multiple channel radio receiver
comprising a front end circuitry for providing complex base band
samples for a plurality of frequency bands and multiple channels in
digital format, a parallel digital matched filter in which common
digital arithmetic elements are used for both acquisition and
tracking purposes connected to said front end circuitry, said
matched filter being arranged to perform a plurality of
simultaneous correlations of received spread spectrum signals
against selected replica offsets of a spreading sequence, said
parallel digital matched filter providing N slices with M stages
per slice and W bit data quantization, each slice being adapted to
perform 1/N of the acquisition computation and then is handed off
to become a dedicated tracking module for one channel,
respectively, including a multi-channel demodulator processor
connected to said matched filter for simultaneously processing all
bands, elements and channels, and to function as an integrated
multi-element multi-user detector.
14. The multiple frequency band, multiple channel radio receiver
defined in claim 13 wherein said matched filter includes an N*M
stage data delay line composed of B distinct bands, each band being
composed of E distinct elements of 2*W bits each and a single
numerically controlled oscillator to serve as a control digital
frequency source matched to the expected chipping rate of the
incoming signal.
15. The multiple frequency band, multiple channel radio receiver
defined in claim 13 including a Rake Combiner for mitigating the
effects of multipath interference dominated communication channels
connected to said parallel digital matched filter.
16. The multiple frequency band multiple channel radio receiver
defined in claim 15 including a multi-channel demodulator connected
to said matched filter for simultaneously processing all bands,
elements and channels and Rake Fingers.
17. The multiple frequency band, multiple channel radio receiver
defined in claim 16 wherein said multi-channel demodulator
processor includes a defragmentation module to insure the maximum
acquisition capability of said receiver is maintained over
time.
18. The multiple frequency band, multiple channel radio receiver
defined in claim 13 including a single clocking system synchronous
to a data sampling clock for generating G independent
NCO/PN-generators that produce PN chipping sequences whose average
rates can precisely track the various received signal chipping
rates, respectively.
19. The multiple frequency band, multiple channel radio receiver
defined in claim 13 wherein said multi-channel demodulator
processor is adapted to form a completely independent set of beams
on each single channel on each Rake Finger of each signal
channel.
20. The multiple frequency band, multiple channel radio receiver
defined in claim 13 wherein as each channel is sequentially
acquired by said matched filter, and said demodulator processor
assures that common digital arithmetic elements are used both for
acquisition and tracking purposes, respectively.
21. The multiple frequency band, multiple channel radio receiver
defined in claim 13 wherein multiple data inputs and delay lines
are present and are available for processing at each arithmetic
element so that the matched filter/despreader processing is
virtually independent of channel origin (e.g. CDMA users, beamform
element, or Rake Fingers).
22. The multiple frequency band, multiple channel radio receiver
defined in claim 13 wherein there is a common NCO/PN-generator
within a common beamformer element set.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] The present application is the subject of provisional
application Ser. No. 60/181,571 filed Feb. 10, 2000 entitled
INTEGRATED BEAMFORMING/CDMA-RAKE RECEIVER ARCHITECTURE. This
application is also a continuation-in-part application of
application Ser. No. 09/707,909 filed Nov. 8, 2000.
[0002] Reference is also made to Weinberg et al application Ser.
No. 09/382,202 filed Aug. 23, 1999 and entitled MULTI-BAND,
MULTI-FUNCTION, INTEGRATED TRANSCEIVER which is incorporated herein
by reference.
BACKGROUND OF THE INVENTION
[0003] 1. Field of the Invention
[0004] This invention relates in general to wireless communication
receivers. In particular, it relates to the integration of multiple
signal types (CDMA, FDMA, CW, etc.), from multiple bands, with each
band and signal type potentially containing multiple user channels,
and a single receiver processing architecture with multiple antenna
elements per band for sequentially acquiring, and simultaneously
demodulating these multiple channels, utilizing jointly-optimized
advanced signal processing techniques of digital beamforming, Rake
multipath combining, and joint detection.
[0005] 2. Description of the Prior Art Matched Filtering
[0006] A matched-filter is typically employed in a spread-spectrum
demodulator to remove the effects of PN-spreading and allow the
carrier and modulating information to be recovered. The digital
implementation of a matched filter can be expressed as an
integrate-and-dump correlation process, which is of relatively
modest computational burden during signal tracking and
demodulation. However, it is computationally and/or time intensive
to acquire such a signal, where many such correlations must be
performed to achieve synchronization with the transmitted spreading
sequence. For each potential code-phase offset to be searched
(which typically number in the thousands), sufficient samples must
be correlated to ensure that the integrated SNR is sufficient for
detection. Performed one at a time, acquisition could easily take
several minutes to achieve in typical applications.
[0007] For applications requiring rapid signal acquisition (e.g.,
seconds), a highly parallel matched-filter structure may be used to
search many spreading code offsets simultaneously. Typically, this
computationally expensive apparatus would be underutilized once
acquisition is completed, during the much less demanding tracking
operation. If the same parallel matched filter is also used for
tracking purposes, only perhaps three of its numerous correlation
branches (perhaps hundreds) are useful in this instance.
Alternatively, it may be simpler to use a separate set of early,
on-time, and late integrate-and*dump correlators to take over once
acquisition is complete; in this case, the parallel matched filter
would go completely unused during tracking.
[0008] In implementations evidenced by the prior art, the
matched-filtering solution has generally fallen into one of several
classes:
[0009] 1. Slow acquisition by sequential traversal of the search
space using only the hardware required for tracking a signal;
dedicated hardware per channel.
[0010] 2. Rapid acquisition by parallel traversal of the search
space using a dedicated parallel matched filter, which is idle or
shut down when dedicated tracking hardware takes over; dedicated
hardware per channel.
[0011] 3. Either class 1 or 2, but multi-band and/or multi-channel,
using a loosely integrated but disparate collection of individual
processing resources.
[0012] Beamforming
[0013] Beamforming is a form of spatial filtering in which an array
of sensor elements are utilized with appropriate signal processing
to digitally implement a phased array antenna, for the purpose of
shaping the antenna response over time in a space-varying manner
(i.e., steering gain in some directions, and attenuation or nulls
in other directions). In a radio communications system, a signal
arriving at each element of an antenna array will arrive at
slightly different times, due to the direction of arrival with
respect to the antenna array plane (unless it has normal incidence
to the plane, in which case the signal will arrive at all elements
simultaneously). A phased array antenna achieves gain in a
particular direction by phase-shifting, or time-shifting, the
signal from each element, and then summing them in a signal
combiner. By choosing the relative phasing of each element
appropriately, coherence can be achieved for a particular direction
of arrival (DOA), across a particular signal bandwidth.
[0014] Digital beamforming is very analogous to this, except that
the signal on each antenna element is independently digitized, and
the phasing/combining operation performed mathematically on the
digital samples. Traditionally, digital beamforming is done on a
wideband signal, prior to despreading a CDMA waveform. This forces
the computationally intense beamforming to take place at a much
higher sampling rate, resulting in more mathematical operations per
second, and corresponding increased hardware cost (there are
examples addressing this shortcoming in the prior art, such as
Hanson et al., where beamforming is performed at baseband to avoid
this and other issues).
[0015] Furthermore, digital beamforming is traditionally done as a
separate process, independent of symbol demodulation, perhaps even
as a separate product from the demodulator. In addition to the
resulting inability to support advanced demodulation techniques
with this architecture, the cost of the beamforming function is
greater as a stand-alone function, compared to the incremental cost
of adding the capability to a demodulator. The largest
cost-component of beamforming is the complex multiplication of each
sample for each element with the beamforming weights. When combined
with the demodulator, the complex multiply can be absorbed into
computation already taking place for extremely low incremental cost
due to beamforming (there is, for example, an implementation of
beamforming using digital direct synthesis (DDS) functions in the
prior art, such as Rudish, et al.). Thus, whether stand-alone
beamformers merely point in the direction of the signal of
interest, or respond more adaptively to dynamic interference
conditions by null-steering, they still lack the ability to be
tightly coupled with potential advanced demodulation
techniques.
[0016] Rake Combining
[0017] Rake combining is a method of mitigating the effects of a
multipath interference dominated communications channel, as is
adaptive equalization. However, in a typical equalizer, the filter
time-span must correspond to the multipath delay spread, and
therefore tends to be limited to very close-in multipath, spanning
perhaps a few symbols. The Rake, however, exploits the properties
of CDMA signals (i.e., during despreading, all other codes become
uncorrelated, including copies of the desired code delayed by
greater than about half a chip, and are reduced to noise across the
entire spread bandwidth) that enables each multipath component
(offset by more than about half a chip) to be acquired, tracked,
and despread in isolation, and then coherently combined. Much like
beamforming, this coherent combining results in increased effective
antenna aperture and improved SNR, although using only a single
antenna element. This divide-and-conquer approach allows the Rake
to span an essentially arbitrary multipath delay spread, applying
computational resources based linearly on the number of desired
despreader branches, or "Fingers", desired, and not based on the
delay spread itself (although acquisition time, and thus dynamic
performance, is related to the actual delay spread, as this defines
the limits of what must be searched).
[0018] In the prior art, Rake combining is typically employed as a
dedicated function in a fixed CDMA receiver structure. Resources
are designed into the receiver to perform some fixed maximum number
of Rake Fingers, and those resources are tied up regardless of
whether those Fingers are actually utilized or not. What is needed
is a more flexible and generalized receiver architecture, which can
task resources on more of a demand basis, and furthermore treat
diversity information such as Rake Fingers as simply one of several
diversity inputs to be jointly optimized in a common process that
yields maximum advantage to each desired user signal.
[0019] What is needed is the ability to combine potential spatial
processing information with other dimensions of information and
diversity, both regarding the signal(s) of interest, and the
interference environment. To this end, what is needed is a receiver
architecture for efficiently processing spatial information
(antenna elements), temporal information (coherent signal multipath
components; i.e., Rake Fingers), and interference information
(noise power estimates, co-channel interfering symbol soft
decisions) jointly and efficiently.
SUMMARY OF THE INVENTION
[0020] The present invention applies approaches to achieve rapid
acquisition in a multi-band, multi-channel signal environment, by
sharing a homogeneous collection of digital processing elements.
This is done, in part, by taking maximum advantage of the
computational commonality between the acquisition and tracking
correlation processes. Furthermore, the mismatch in computational
demand between acquisition and tracking is exploited by creating a
multi-channel, multi-band integrated receiver. Since only a small
percentage of the computational resources are consumed by tracking
an individual channel, the remaining resources may be employed to
accelerate the acquisition of additional channels. As more
resources become dedicated to tracking, fewer remain for
acquisition; this has the effect of gradually reducing the number
of parallel code offsets that can be searched, gradually increasing
acquisition time. In many applications, such as a GPS receiver,
this is quite acceptable, as generally additional channels beyond
the first four are less urgent, and are used primarily for position
refinement, and back-up signals in the event that a channel is
dropped. These ideas are the subject of U.S. patent application
Ser. No. 09/707,909, filed Nov. 8, 2000, entitled
"Sequential-Acquisition, Multi-Band, Multi-Channel, Matched
Filter", and are preserved as features of the present
invention.
[0021] The present invention embodies various extensions to the
previously disclosed invention, wherein the multi-band capability
is evolved to support multiple antenna elements at a common band
(as well as other bands), to support digital beamforming; the
multi-channel capability is evolved to support multiple Rake
Fingers on a common channel (as well as other channels); and the
multi-channel demodulator capability is evolved to support
computationally efficient, simultaneous processing of all bands,
elements, channels, and Rake Fingers. The present invention thus
forms an architectural framework capable of hosting a variety of
algorithms for joint space-time optimization of individual user
channels in a multipath environment, as well as multi-user (joint)
detection of multiple user channels limited by co-channel
interference. By considering these capabilities together, rather
than as independent solutions to problems, considerable
efficiencies and improvements are realized by this invention, in
comparison to the prior art.
[0022] In the first aspect of the present invention, the
multi-datapath receiver architecture allows independent
automatic-gain control (AGC) between multiple input bands B or
elements E, minimizing inter-band/element interference, and
avoiding additive noise compared to schemes that combine the
bands/elements into a single signal and data stream.
[0023] To accomplish this, the present invention efficiently
processes multiple streams of W-bit complex sampled data (real data
is easily processed as well, by adding a complex-to-read conversion
to the front of the matched filter), so that multi-band or
multi-element receiver signals can be kept spectrally separated.
This concept, implemented using D data storage paths, supports D
bands and elements when shifting at the data sampling rate
(F.sub.samp); alternatively, the same D data storage paths can
support D*k bands and elements by multiplexing the
multi-band/multi-element streams and shifting the data at the
higher sampling rate of k*F.sub.samp.
[0024] In another aspect of the present invention, the parallel
acquisition correlator, or matched-filter, aids in rapid
pseudo-noise (PN)-acquisition by simultaneously searching numerous
possible PN-code alignments, as compared with a less
compute-intensive (but more time-intensive) sequential search.
Multiple channels of data may be co-resident in each band/element
and sampled data stream using Code Division Multiple Access (CDMA)
techniques, and multiple bands/elements and sampled data streams
share the common computation hardware in the Correlator. In this
way, a versatile, multi-channel receiver is realized in a
hardware-efficient manner by time-sequencing the available
resources to process the multiple signals, multiple antenna
elements, and multiple multipath components resident in the data
shift registers simultaneously.
[0025] In still another aspect of the present invention, the
matched filter is organized into N "Slices" of M-stages/Slice. Each
Slice is composed further of D data paths supporting multiple bands
B and/or antenna elements E. Each Slice can accept a code phase
hand-off the from the PN-Acquisition Correlator and become a
PN-tracking despreader by providing separate outputs for early,
on-time, and late correlations for each element (with spacing
depending on the sampling rate; typically half a chip). Slices are
handed-off for tracking in the same direction as data flows, and
correlation reference coefficients are shifted (for instance, left
to right)-this permits shifting data to be simultaneously available
for the leftmost Slices that are using the data for tracking, and
rightmost Slices that are using the data for acquisition. Each
Slice can choose between using and shifting the acquisition
reference coefficient stream to the right, or accepting the handoff
of the previous acquisition reference coefficient stream and using
it to track the acquired signal.
[0026] In still another aspect of the present invention, the
Acquisition correlator can integrate across all available Slices to
produce a single combined output, or the individual Slice
integrations can be selectively output for post-processing in the
case of high residual carrier offsets or high-symbol rates, where
the entire N*M-stage correlator width cannot be directly combined
without encountering an integration cancellation effect.
Alternatively, the Acquisition correlator can be configurable to
switch from coherent integration to non-coherent integration, by
taking the magnitude of I and Q partial integrations within the
summer tree, or Slices themselves, at a point appropriate for the
signal being acquired.
[0027] In yet another aspect, the present invention embodies a
Scaleable Acquisition Correlator, which when tracking a maximum of
G independent channels and/or Rake Fingers, can use the remaining
N-G Slices to search for new signals for fast re-acquisition of
dropped signals, and for continually searching the multipath
environment for Rake Fingers to track dynamic channel conditions.
Initially, Slices will be allocated sequentially (for instance,
from left to right), but after running for some time, with signals
alternately being acquired and dropped, the Slice allocation will
most likely become fragmented, resulting in inefficient use of the
Acquisition Correlator. This can be resolved by implementing a
de-fragmentation algorithm that swaps tracking Slices around
dynamically to maximize the number of contiguous rightmost Slices,
and thus optimize Acquisition.
[0028] In another aspect, the present invention contains G
independent numerically-controlled oscillator (NCO)-based PN-Code
Generators with almost arbitrary code rate tracking resolution (for
example, better than 0.0007 Hertz for a 32-bit NCO clocked at 3
MHz). All NCOs run using a single reference clock which is the same
clock that is used for all signal processing in the Matched-Filter
and Demodulator. Ultra-precise tracking of PN Code phase is
maintained in the G independent phase accumulators. Multi-channel
NCOS can in one embodiment be efficiently implemented by sharing
computational resources and implementing phase accumulation
registers in RAM, for the case when the processing rate is in
excess of the required NCO sampling rate. Note that while each
channel and Rake Finger requires its own PN-NCO, a single NCO is
shared across all elements when beamforming.
[0029] In still another aspect of the present invention, the
incoming wideband element data is made available to all Slices,
which allows each element to be independently despread for each
channel/Rake Finger using the core matched filter structure. As a
result, beamforming is easily performed at narrowband (despread)
sampling and processing rates, and with improved potential
precision. The present invention is an improvement over the prior
art, because in addition to the raw computational savings of
narrowband processing, the beamformer hardware is time-shared
across multiple elements, channels, and Rake Fingers for improved
computational efficiency.
[0030] In another aspect, the present invention allows the
Beamforming computation to be implemented with only additional
adders, due to integration with the demodulation carrier phase
rotation and the AGC scaling functions.
[0031] In yet another aspect, the present invention allows an
element snapshot memory to operate at narrowband sampling rates,
allowing an eased implementation for any snapshot operations
required.
[0032] In still another aspect, integration of the beamformer with
the demodulator in the present invention allows advanced adaptive
algorithms to be implemented that can be enhanced by the feedback
of post-demodulation metrics such as PN-SNR/phase,
carrier-SNR/phase, symbol-SNR/phase, as well as error control
decoding metrics.
[0033] In still another aspect of the present invention, the
integrated beamforming CDMA Rake receiver exploits both space and
time diversity aspects of a multi-path environment by assigning
Slices to each Rake Finger, and steering beams that individually
optimize along the line-of-sight (DOA) of each multipath reflection
(i.e., a potential beam for each Rake Finger).
[0034] In another aspect of the present invention, the integrated
multi-channel demodulator and Rake combiner make coherent complex
symbol data for each Rake Finger (potentially for multiple user
channels sharing the same frequency band), as well as individual
channels not being Raked, available to a single optimization
process. This allows the use of advanced multi-user detection (MUD)
algorithms (e.g., joint detection) to mitigate co-channel
interference that has not been suppressed by beamforming.
[0035] In yet another aspect, the present invention's Slice-based
data-flow computational architecture permits dynamic, flexible
allocation of resources between tracking of multiple input bands,
user channels, and Rake Fingers, and acquisition resources for
dropped/new channels and continuously monitoring Rake dynamics.
[0036] In another aspect, the matched-filter Slice architecture of
the present invention contains PN-tracking integrators (i.e.,
early, on-time, late) for each beamforming element. Furthermore,
after all elements are weighted and combined, the demodulator
architecture uses the combined early/on-time/late integrations to
maintain a single PN-tracking loop for each beamforming channel, or
Rake Finger.
[0037] In another aspect, the present invention allows each
beamforming channel, or Rake Finger, to combine data from all
elements and form a composite carrier and symbol discriminator that
allows all elements of that channel to be tracked with a single
carrier loop, and a single symbol loop.
[0038] In still another aspect, the present invention's
multi-channel architecture allows continuous on-line element
calibration capability to take place. Furthermore, calibration can
be performed independently on each user channel, and each Rake
Finger, closing the calibration loops individually to remove
essentially all bias terms.
BRIEF DESCRIPTION OF THE DRAWINGS
[0039] FIG. 1 is a generalized functional block diagram of the
multi-channel matched filter architecture, illustrating the
multiple input bands, the multiple NCO-based PN Generators, and the
division of the parallel matched filter into multiple Slices; the
matched filter can be seen to have an acquisition output, and a
tracking output which sequentially sends despread element data for
each channel and each channel's Rake Finger into the integrated
multi-channel beamforming demodulator and Rake combiner.
[0040] FIG. 2 is a generalized functional block diagram of the
matched filter Slice architecture (for the specific embodiment in
which RAM structures are utilized to form highly efficient data
storage cells, for the case of relatively low sampling rates); note
that each Slice shares a single PN chipping stream for despreading,
and contains "E" computation elements, corresponding to the number
of supported beamformer elements; each computation element is
shared across all "M" stages/Slice.
[0041] FIG. 3 is a functional block diagram showing an example
embodiment of the multi-channel, NCO-driven, PN code Generator,
using efficient RAM-based state machines.
[0042] FIG. 4 is an illustration of the sequential acquisition and
handoff to tracking in the matched filter, showing how multiple
antenna elements along with multiple signal bands and channels are
handled simultaneously, using an example embodiment and a time
sequence of resource allocation diagrams.
[0043] FIG. 5 is a dataflow diagram showing the complex arithmetic
calculations required to weight and combine all beamformer elements
for each despread sample coming from the matched filter.
[0044] FIG. 6 illustrates how the embodiment in FIG. 3 might
produce sequential despread outputs, corresponding to each band,
element, and channel (early, on-time, and late), as well as the
sequence of carrier NCO outputs and beamformer weight outputs that
might be produced during the tracking process; this figure also
shows graphically how these sequences flow through computation
elements to simultaneously accomplish both the carrier tracking and
beamforming functions (no Rake in this example).
[0045] FIG. 7 is an illustration of the sequential acquisition and
handoff to tracking in the matched filter (similar to FIG. 4), for
a different example embodiment containing beamforming and Rake
combining, using a time sequence of resource allocation
diagrams.
[0046] FIG. 8 is a functional block diagram of one embodiment of a
processing architecture for the integrated beamforming/Rake
multi-channel demodulator, illustrating: the manner in which
sequential data from the PN matched filter is processed to form PN,
carrier, AGC, and symbol tracking loops for each channel and Rake
Finger; the integration of the carrier tracking rotation and
beamforming functions; and the presentation of all channels and
Rake Fingers to a single integrated demodulator, which can host a
variety of algorithms capable of optimizing and combining
same-channel multipath (Rake Fingers) and joint detection of
multiple, potentially interfering, co-channel users.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)
[0047] The first aspect of the preferred embodiment relates to the
implementation of multiple channel, multiple frequency band
receivers. At any given point in time, the state of the art in
analog-to-digital conversion (A/D) chips, and subsequent digital
signal processing (DSP) technology for performing data
demodulation, will allow only a certain amount of frequency
spectrum (band) to be digitized into a single data stream. Within
that band, multiple user channels can coexist using various well
known multiple-access techniques such as FDMA, TDMA, CDMA, etc.
[0048] When additional channels of interest lie outside of the
frequency bandwidth that can be digitized into a single digital
band, and simultaneous reception is required from each band, then
multiple RF downconverters and A/Ds must be used to digitize
multiple bands. The present invention allows an arbitrary number of
such bands to be processed together in a unified computational
engine. In this embodiment, a pool of arithmetic processing
resources, or receiver channels, can be applied on demand to
various user signals, regardless of which band they originated in.
In this way, an almost arbitrary variety and amount of frequency
spectrum can be utilized, and an almost arbitrary number of user
channels of varying modulation type can be digitally extracted from
it.
[0049] There are several advantages of using this technique to
present multiple bands to a single receiver structure. Firstly, it
is well known that as wider bandwidths containing multiple and
various signals are received together, increasing analog fidelity
requirements are imposed. This is a significant limitation, in that
analog circuitry suffers from such problems as intermodulation
distortion (IMD), where multiple frequency sources interact to
produce distortion components. The present invention optimizes the
analog signal fidelity by digitizing downconverting and digitizing
each band, while preserving the advantages of a digital "software"
radio-namely, integrated, flexible, multi-channel demodulation
using DSP techniques.
[0050] Secondly, given an arbitrary RF and A/D dynamic range, it is
desirable to use automatic gain control (AGC) to capture the signal
of interest within the available amplitude range of both analog
circuitry and A/D converter. As wider bandwidths containing
multiple and various signals are digitized together, they must also
be subject to a common AGC process, which will be dominated by the
largest signals across all bands; this potentially decreases the
SNR of the smaller signals, due to A/D quantization noise. The
present invention optimizes the AGC process by allowing each band
to be treated separately.
[0051] Thirdly, other schemes to digitize a composite mix of
various frequency bands might use a technique of summing together
the signals after translation to non-overlapping adjacent
intermediate frequencies, allowing the use of a single A/D
converter. In this type of scheme, the limitations of the analog
circuitry will dictate that additive noise from each of the various
RF bands will somewhat degrade the signal-to-noise ratio (SNR) of
the resultant composite signal. The present invention optimizes the
SNR of each band by maintaining separate RF, IF, and digital signal
paths.
[0052] Fourthly, this aspect of the present invention directly
supports digital beamforming by utilizing this multi-band receiver
technique. Since digital beamforming involves the use of a
multi-element antenna, resulting in dedicated RF downconversion
paths for each element, the present invention allows beamforming to
be accommodated in a flexible, scalable fashion, by treating each
element in the same manner as if it were another signal band.
Naturally, the RF implementation can in fact be simplified in
comparison with the generic multi-band case, because a dedicated
Beamformer implementation could optimize the frequency synthesis
circuitry by using the same LO for each antenna element
downconverter. If the beamformer combiner is implemented after the
matched filter (despreader), the digital implementation is now able
to treat each element as though it were just another datapath for
another signal band.
[0053] The second aspect of implementing the preferred embodiment
relates to the architecture of the flexible computation core of the
digital matched filter. The architecture has been designed to
satisfy two different driving requirements: accelerated acquisition
of a single user channel, and simultaneous tracking of multiple
user channels. Referring to FIG. 1, front end circuitry FE provides
complex baseband samples for a plurality of frequency bands and
multiple channels in the radio spectrum to an (N*M) stage data
delay line (shown as being embodied by N distinct Slices), composed
of B distinct bands, each band composed of E distinct elements of
2*W bits each (W bits I, W bits Q complex data), contains a
sequence of samples of the bands of interest. It is well known that
the sampling rate must be chosen to satisfy the Nyquist criterion
to preserve the appropriate signal bandwidth of interest, and to
allow sufficient time resolution for acquisition and tracking;
generally two or more times the chipping rate for a spread spectrum
signal. The data is then shifted through the data delay lines at
the sampling rate.
[0054] For the purposes of acquisition, a single numerically
controlled oscillator (PN-NCO) is needed, to serve as a finely
controllable digital frequency source matched to the expected
chipping rate of the incoming signal. In conjunction with this,
during acquisition a single PN chip Generator is needed, to
reproduce the PN sequence of the incoming signal, at the rate
dictated by the PN-NCO. This PN Sequence is then presented to the
leftmost end of the data delay line (to the leftmost Slice), where
it is also shifted from left to right down a PN sequence delay line
(shown in more detail in FIG. 2). At appropriate time intervals,
the state of the PN sequence delay line is latched into a reference
correlation register. The computational logic within the Slices
then performs a correlation of the latched reference PN sequence
against the signal samples contained in the data delay line. Note
that in the example embodiment in FIG. 2, the Slice architecture is
illustrated with a RAM-based implementation, which is efficient for
low sampling rates with respect to the available processing rate.
Other embodiments of the present invention might utilize a
register-based architecture variant, which would allow for much
higher sampling rates (less than or equal to the processing rate);
registers are in that case used for all data shift-registers.
[0055] For each sample time, up to (N*M) multiplications (or N*M*E,
if elements are acquired in parallel) are performed of each data
sample with its corresponding reference PN chip (in some
applications, the stages are decimated prior to performing the
correlation, so that not all are tapped for computation); all of
these products are then summed into a single partial correlation
value by the Acquisition Summation Network shown in FIGS. 1-2,
which is then passed on to the demodulator circuitry for further
integration, beamforming and Rake Finger selection (depending on
the acquisition scheme chosen), thresholding and detection. Because
the data samples are shifted by one position at each sample time,
and the latched reference PN sequence is held in the same position
over a period of time (update period), each sequential partial
correlation within a given update period represents a different
potential alignment (code offset) between the reference PN sequence
and the received signal. In this way, over time a correlation is
performed for all possible code offsets, to within the nearest
fraction of a chip defined by the chosen sampling rate; the timing
of the latch update period, and the NCO/PN-Generator code phase,
are carefully controlled to determine the specific offset search
sequence. The post-processing circuit can perform additional
integrations for each code offset to achieve sufficient SNR to
enable detection at the correct offset.
[0056] At this point, the receiver can be said to have completed PN
acquisition, and the matched filter is able to go into PN tracking
mode. During tracking, the problem is substantially easier. If
there were no phase or frequency drift present, only the single
correctly aligned correlation sequence must be computed; that would
be a single multiply and sum per input sample. Since there are
phase and frequency drifts (i.e., the reference PN-NCO frequency
setting becomes incorrect over time) in typical applications, two
additional correlations must be computed as well, corresponding to
the code offsets that are slightly early and slightly late, with
respect to the currently tracked (on-time) code offset. These
correlations allow the PN phase and frequency drift to be observed
and tracked with the PN-NCO, using well known PN tracking loop
techniques. The early, on-time, and late correlations (or partial
correlations) are output via a separate signal path to the
demodulator tracking circuitry. So, where (N*M or N*M*E) multiplies
and sums must be computed for each input sample during acquisition,
only (3*E) multiply/sums must be computed for each sample during
tracking. Since there is motivation to choose (N*M) to be as large
as possible for rapid acquisition, this leaves a substantial
surplus of computational horsepower idle during tracking.
[0057] Thus, the primary nature of the second aspect of
implementing the present invention lies in the agility of the
computational structure in transitioning, one Slice at a time, from
being part of an acquisition correlation process as described
above, to being part of a tracking correlation process as described
above. For the multi-channel case, this involves adding additional
NCO/PN-Generator pairs corresponding to the desired number of
channels and Rake Fingers (shown as G in FIG. 1) to be
simultaneously tracked. Each of these creates a unique PN sequence,
at unique chipping rates, and presents them to unique Slices, from
left to right, as shown in FIG. 1.
[0058] Each combination of NCO/PN-Generator and Slice (matched up
from left-to-right) form the required computational capability for
tracking a single user signal, or single Rake Finger for a single
user signal. The rightmost unused NCO/PN-Generator pair, and all
rightmost unused Slices, form the available computational
capability for acquiring a new user signal, and for searching for
and acquiring the strongest Rake multipath components. The amount
of time required to acquire the new signal depends on the number of
correlation stages available, because that determines the number of
correlation samples that are integrated at each sample time. All of
this computation, for acquisition and tracking of multiple
channels, happens concurrently using the flexible computation
resources, and occurs transparently with respect to the multiple
bands and elements of sampled data that constantly stream through
the data delay lines. This entire process is illustrated in FIGS.
4, 6 and 7.
[0059] The third aspect of implementing the preferred embodiment
relates to the partial acquisition integration method. For the
problem of PN Acquisition, it would be ideal to integrate an
arbitrary number of correlation samples until the appropriate SNR
level is reached. However, this cannot be done in the presence of
residual carrier components due to unknown doppler and other
frequency offsets, which would cause integrations across complete
carrier cycles to cancel out. In a similar manner, integrations
across multiple data symbol transitions also causes cancellation.
These effects limit the useful size of the acquisition matched
filter, and would normally force much of the computational
capabilities to go unused (through masking-out of that portion of
the filter which exceeds the appropriate integration length). This
problem is mitigated in the present invention by allowing the
individual Slice partial integrations to be output to the
post-processing circuitry. Various methods can be used to combine
the partial integrations non-coherently into a complete integration
while mitigating the cancellation effects.
[0060] An alternative embodiment of the present invention
accomplishes this same goal by modifying the Slice architecture
slightly to incorporate the magnitude detection circuitry, or other
means of switching to non-coherent integration, directly into each
Slice. This would allow each Slice to be configurable to integrate
the appropriate amount of signal coherently, perform detection, and
allow the summer tree to perform non-coherent summation of each
Slice's output, passing that sum to the acquisition circuitry to
complete the integration/detection process.
[0061] In a fourth aspect of the present invention, the preferred
embodiment employs a defragmentation algorithm to ensure that the
maximum acquisition capability is maintained over time. This is
particularly important with the use of the Rake combiner
functionality, as multipath components can change rather
dynamically, depending on the channel environments; acquisition
resources will continually need to be available to monitor and
acquire them. The manner of sequential acquisition and, from left
to right in FIG. 1, allocation of Slices for tracking has been
described. In that initial context, the rightmost Slices are always
optimally utilized for acquisition; none are wasted. However, as
signals are dropped in a multiple channel tracking environment,
holes will develop where middle Slices are no longer tracking, but
cannot participate in acquisition in the normal fashion due to
isolation from the rightmost Slices.
[0062] This problem is mitigated in the present invention by
swapping out tracking Slices from right to left in order to
maintain contiguous unused rightmost Slices for acquisition. This
is done by initiallizing the NCO/PN-Generator of the unused (left)
Slice to run in offset-synchronism with the currently tracking
(right) Slice that is to be moved; offset, in the sense that
chipping frequency is identical, but code phase is advanced by an
appropriate amount to correspond with the relative difference in
received signal phase at the two Slices. In units of time, this is
basically the number of delay stages of offset between the two
Slices, divided by the sampling rate. At the known chipping rate,
this is easily converted to a code offset. After the handoff is
complete, the process is repeated until all tracking Slices are
packed to the left.
[0063] The fifth aspect of implementing the preferred embodiment
involves a method of using a single clocking system, synchronous to
the data sampling clock, to generate G independent
NCO/PN-Generators that produce PN chipping sequences whose average
rates can very precisely track the various received signal chipping
rates. Also, if the NCO processing clock is in excess of the
required NCO sampling rate, efficient RAM state storage and code
phase computational hardware can be time-shared for reduced
hardware size (if this is not the case, a more traditional
register-based embodiment of the NCO would be required). A block
diagram of this concept is shown in FIG. 3.
[0064] Because each NCO is operating at the NCO sampling rate
(perhaps equal to the data sampling rate), it can only make a
decision to advance to the next chip at those coarse sampling
intervals. Thus, even though the NCO phase accumulator knows when
to advance to the next chip to within fractions of a sampling
interval, it must incorrectly wait until the end of the sampling
interval to do so. However, this chip-jitter averages out in the
long term (as long as the NCO sampling rate is asynchronous to the
chipping rate); furthermore, because the NCO clocks are all
synchronous to the data sampling clocks, the jitter exactly
reflects the effective jitter that will be contained in the
received chip transitions. In other words, both the incoming signal
code phase, and the internal accumulated code phase will track very
precisely; since they are both asynchronously sampled by data/NCO
sampling clock, a common phase jitter will be superimposed onto
both, such that the jitter itself causes no additional processing
loss.
[0065] FIG. 3 shows an example 6-channel implementation of the
RAM-based PN-code Generator. In this example, it is assumed that
the processing clock is at least 6 times the desired NCO sampling
rate. So, within the time of each NCO sampling interval, the
computational resources may be cycled 6 times to produce new code
phases and PN chips for each of 6 channels or Rake Fingers. This
allows, for example, a single adder to compute for 6 phase
accumulators. The six fractional and integer code phases are stored
in RAM storage cells, and can be retrieved sequentially for
processing. The new code phases are then sequentially updated back
into the RAMs. Also, in this example, RAM is utilized to store the
entire PN sequence for each channel. Thus, arbitrary sequences can
be generated, and the phase accumulator circuitry merely plays back
the chips at the correct rate. Alternatively, specific PN sequence
generators could be constructed, with a slight modification of the
indicated block diagram. A specific implementation requires a
combination NCO/PN-Generator for each simultaneously tracked
channel or Rake Finger, plus an additional one for acquisition.
[0066] In the sixth aspect of the present invention, it can be seen
that the core matched filter architecture readily supports
beamforming through the despreading of multiple antenna elements
for each signal band independently for each user channel and Rake
Finger, and presenting the narrowband data to the demodulator for
weighting and combining. This is facilitated by first treating each
antenna element as if it were just another supported band, and
passing the digitized element samples into the multi-band matched
filter. Second, the Slice architecture could then be configured to
assign each channel-element to a unique Slice for despreading. Such
an embodiment would directly extend the previously disclosed
architecture (U.S. patent application Ser. No. 09/707,909, filed
Nov. 8, 2000) with essentially no change to the matched filter
itself, but would likely require a large number of Slices to
implement (#channels * #elements * #Rake-Fingers). This could also
have the side-effect of increasing the relative length of the data
delay lines (i.e., N*M, by increasing N).
[0067] An alternative embodiment, described here and illustrated in
FIG. 2, evolves the Slice architecture by despreading all
elements/channel or elements/Rake-Finger within a single Slice.
This has several advantages: in the first advantage, the single
PN-stream needed by each element is already available in the Slice;
in the second advantage, the time-aligned samples for each element
are all available (either from registers, or having been read from
RAM) in the same processing clock cycle for multiplication; and
third, the overall Slice count and delay line length may be reduced
to (#channels * #Rake-Fingers). So, in this embodiment, each Slice
requires E multipliers, and E early/on-time/late integrators. After
a configurable amount of integration within the Slice, the stream
of early/on-time/late partial integrations for each element are
multiplexed into a sequential stream and presented to the
multi-channel demodulator for further processing.
[0068] What is significant about this aspect of the present
invention is its ability to reduce the beamforming computational
burden proportionally to the matched filter decimation ratio, and
utilize that advantage by sequentially processing the despread
element samples in the demodulator. This allows the demodulator
hardware to be multiplexed to accomplish combined carrier tracking,
AGC, and beamformer weighting and combining, using minimal
additional hardware resources compared to a non-beamforming
demodulator. This advantage in beamforming, combined with the
similar ease with which Rake combiner capability is also added,
along with the possibility of joint processing of these functions
for improved receiver performance, represents a significant
improvement over implementations in the prior art.
[0069] The seventh aspect of the present invention is the sharing
of existing computational resources in the demodulator to perform
the actual beamforming weighting and combining functions. To form a
beam on a given channel/Rake Finger, each complex sample must be
multiplied by the appropriate complex weight (to cause the desired
rotation of the vector). After the weighting/rotations are
performed, the complex element samples can then be added
together-the summation will constructively combine energy for
signals arriving at the antenna array from the desired direction
(and sidelobes), and destructively cancel elsewhere.
[0070] A schematic dataflow diagram showing elements and
corresponding weights is shown in FIG. 5, along with mathematical
operations necessary to perform beamforming. In the prior art, the
physical implementation of this could take the form of anything
from literally implementing the diagram as shown (for sampling
rates equal to the processing rate), to a single complex multiplier
and two accumulators (for sampling rates much less than the
available processing rate). The innovation of the present invention
is illustrated from a high level in FIG. 1, where the entire
beamforming operation can be seen to be absorbed into computation
already taking place in the demodulator for carrier tracking
rotation, AGC scaling, and symbol integration. The only added
complexity is the additional scalar adder that rotates the carrier
NCO accumulated phase by the desired beamformer phase shift, prior
to using the phase to determining corresponding SIN/COS amplitude,
as well as additional multiply operations per sample due to the
elements (E times as many multiplies).
[0071] FIG. 6 illustrates this in more detail, along with an
example enumeration of the actual data operands that would flow
through this process (element data samples, carrier NCO samples,
and beamformer weights). Whereas in the conventional implementation
(FIG. 5) only complex multiplies and summation are required, this
technique is split into the beamformer rotation and the beamformer
scaling as separate operations. This is due to the manner in which
this technique works. Normally, multiplication of the complex
weight and the complex element sample simultaneously rotates and
scales the element I/Q vector. However, in the manner of this
invention (FIGS. 1 and 6), the existing carrier rotation circuitry
is exploited to serve the additional purpose of beamforming
rotation. This is done by computing the desired carrier rotation
for carrier tracking (which would be constant for each element for
a given channel/Rake Finger), but adding to that the additional
rotation desired (different for each element) for beamforming.
While this achieves the desired element rotation, the gain coming
out of the carrier NCO look-up table is fixed.
[0072] Thus, the scaling portion of the beamforming weighting
operation is not yet done, and must be performed in a subsequent
scalar multiply on each I and Q. Conveniently, just such a function
is already next in the demodulator dataflow-the AGC function. Since
software typically performs both the beamformer weight calculation
and the AGC scale factor, at a relatively slow rate, the multiply
is absorbed into a software operation to modify the AGC weight to
also include the beamformer weight, and the single pair of existing
scalar multipliers is used to serve both purposes.
[0073] In the eighth aspect of the present invention, snapshots of
element data may need to be captured for various processing to aid
the beamformer adaptive algorithms. By performing the beamforming
on despread, narrowband data, the snapshot memory functionality
benefits from the decimated sampling rate and potentially becomes
reduced in complexity. This may be a benefit manifested in reduced
implementation cost.
[0074] In the ninth aspect, the present invention enables advanced
adaptive beamforming algorithms to be implemented, through the full
integration and tight coupling of the beamformer with the
demodulation process. Typically, in the case of a beamformer that
is more standalone and distinct from demodulation, the beamformer
would be able to point to a known signal location (DOA), and
adaptively form nulls to mitigate powerful, readily measurable
sources of interference. This invention, however, makes much more
information available to the beamforming algorithm. By providing
demodulator metrics to the beamformer algorithm, a closed loop is
formed between demodulator performance (PN/carrier/symbol
SNR/phase), and the weight adaptation process. This facilitates the
use of algorithms that start with known information, such as signal
DOA and interference DOA, and iteratively find the best weights
that minimize the demodulator's prioritized, observable errors.
Even after the symbol demodulator, error-control decoding
performance metrics can also be fed back to the optimization
process.
[0075] In the tenth aspect, the present invention combines the
advantages of beamforming and Rake combining, yielding a result
that improves SNR (actually,
SINR-Signal-to-Interference-plus-Noise-Ratio, but the term SNR will
continue to be used for convenience) with respect to either
technique applied in isolation. By itself, beamforming is
advantageous because it simultaneously increases the effective
antenna aperture in the direction of the desired signal (as well as
sidelobes), while also decreasing the effective antenna aperture in
other directions, perhaps containing interferers. With an adaptive
algorithm, the beam can actually be formed to perfectly null out
detected interferers; typically, a compromise is actually made
between these extremes, balancing desired signal gain and
interfering signal attenuation.
[0076] The preferred embodiments of the present invention have the
advantage of being able to form a completely independent set of
beams on each signal channel, or on each Rake Finger of each signal
channel. This allows each user channel the luxury of all degrees of
freedom afforded by the available antenna elements to optimize its
SNR. Furthermore, this allows each multipath component to be
optimized independently as well. In many environments where
multipath is prevalent, each multipath component (reflection) is
likely to come from a different direction (DOA). Likewise, in those
same environments the interference signals are likely to be subject
to the same multipath conditions, causing interference power to be
distributed among different DOAs as well. Thus, each desired signal
multipath component really requires a completely different beam
pattern, in order to optimize its particular signal gain and
interference rejection situation.
[0077] It may be the case that a large multipath
component-desirable for combining-arrives from the same direction
as a strong interference signal. This may result in the situation
where a beam pattern cannot be generated that both passes the
desired signal, and excludes the interferer. This is an example of
the strength of the joint beamformer/Rake optimization capability
afforded by the present invention: an intelligent algorithm in the
demodulator can make use of the broad information presented to it
to select (or iteratively determine based on demodulator feedback)
the best combination of multipath elements and beam patterns to
optimize demodulator SNR. In this example, it may be necessary to
reject the multipath component from combining, and choose one with
better spatial isolation from interference.
[0078] In the eleventh aspect of the present invention, complex,
integrated, beamformed symbol data is maintained coherently for
each channel, and for each Rake Finger, and presented to a
single-point detection process within the demodulator as shown in
FIG. 8. Also available to this process is the integrated PN and
carrier phase error, as well as information about the beamformer
processing. Using this wealth of information, a single, unified,
combiner-detection algorithm can now be used to optimize the SNR of
each signal channel. Having already applied the benefits of
beamforming to spatially isolate each desired incident signal
component, and integrated/decimated each sampled stream to (or
near) the symbol rate, it is necessary at this point in the
demodulation process to form soft symbol decisions.
[0079] Although there are many ways in which this process could be
carried out, a few example methods will be outlined for clarity. In
the simplest method, samples for each Rake Finger are integrated to
the symbol level, and then delayed, weighted, and combined
according to various combining schemes well known in the
literature. Then, a soft decision is formed on the combined result
and output to the next process (e.g., error control decoding).
[0080] In a more complicated method, the process just described is
used to independently form soft decisions for each of multiple
signals sharing the same frequency channel (for example, co-channel
signals in a CDMA system). These independent soft decisions are
each corrupted by the combined co-channel interference of all of
the other channels, due to imperfect orthogonality of the spreading
codes. However, the character of the co-channel interference is now
somewhat understood, having just formed soft decision estimates of
each of the interferers. Thus, a process can now be followed to
subtract the effects of the estimated interference from each
signal, resulting in an improved estimated soft decision of each
channel. Naturally, since the procedure just followed results in
improved estimates of each interferer, it can be repeated, and in
fact applied iteratively until each soft decision is somewhat
cleansed of the effects of the co-channel interference. This
process, joint detection, is a form of multi-user detection (MUD),
and is one of many such techniques that is well described in the
technical literature.
[0081] In even more complicated methods, the MUD iterative process
can be combined with the Rake process, and perhaps even the
beamforming process, to afford an almost arbitrary level of
optimization to be achieved. What is novel in the present invention
is the scalable architecture which efficiently processes all the
available information, and makes it available to the demodulator to
enable such algorithms to be implemented. The present invention
allows spatial diversity (antenna elements), time diversity (Rake
Fingers), and intelligent processing of interference (beam
patterns, co-channel joint detection) information to be jointly
optimized in a common demodulation and detection process.
[0082] In the twelfth aspect, the present invention facilitates a
highly flexible, adaptable software radio architecture that allows
a fixed hardware structure of computational dataflow elements to be
tasked on a dynamic basis as needed. This allows the controlling
software to choose how the hardware resources are allocated between
the number of user channels, the number of channels supporting
Rake, the number of Rake Fingers per channel (could be different on
each channel), and the length of the acquisition matched filter.
Furthermore, the actual processing capacity of this architecture
really depends on the relationship between the extent of physical
resources actually committed (e.g., #multipliers/Slice, or
#Slices), the maximum processing clock rate supported by the
implementing technology, and the desired input data sampling
rate-therefore, an embodiment of this invention could allow the
aforementioned features to be further traded-off against input
sample rate. Note that yet another related consideration is the
relationship between the number of bands and elements, and the
number of datapaths that an embodiment implements-if there are more
elements and bands than datapaths, the effective sampling rate must
be increased to accommodate multiplexing. Thus, the present
invention enables the dynamic tradeoff of all these features and
issues, which represents a clear improvement over the prior art,
and which is much needed in the rapidly evolving and complex
wireless communications field.
[0083] In the thirteenth aspect, the preferred embodiment of the
present invention incorporates all elements for a given channel or
Rake Finger directly into the Slice architecture, rather than
treating each element as if it were an independent channel (as the
Rake Fingers are treated by the matched filter). While either of
these methods could be chosen to implement a specific embodiment
under this invention, there are advantages to the former method, as
illustrated in FIG. 2. Specifically, by incorporating the
despreading of all elements for a given channel or Rake Finger into
the same Slice, the integrations remain exactly time-aligned, which
is required for coherent beamforming (while this could certainly
still be satisfied in other embodiments, such as one element per
Slice, the timing and control problem is more complicated). The
dataflow control within the Slice is also simplified, because each
element sample is automatically available in parallel for
multiplication by the PN sequence. Furthermore, each element shares
the same PN-Generator in a direct and convenient fashion, because
the PN code phase misalignment on each element due to the angle of
arrival at the element array should in most cases be negligible
compared with the length of each chip.
[0084] In addition, another benefit to this aspect of the present
invention lies in the ability to obtain the benefit of beamforming
gain on the early and late error integrations. By rotating and
combining the early/late outputs from each element, a single,
coherent PN code phase error is generated, which is used to correct
a single PN-tracking loop for each channel or Rake Finger.
[0085] In the fourteenth aspect of the present invention, the
architecture of the preferred embodiment allows a single carrier
tracking loop and a single symbol tracking loop to track all of the
elements for a channel or Rake Finger. This is similar to the
single PN tracking loop just described. The reduction in the number
of tracking loops is useful in minimizing the demodulator
complexity.
[0086] In the fifteenth aspect, the present invention represents a
substantial improvement over implementations described in the prior
art, particularly with respect to the calibration problem. In the
ideal case, the geometry of the antenna array would be precisely
known, each antenna element would exhibit identical performance
characteristics, and the entire RF downconversion path through the
A/D converters would have identically matched delay and other
attributes. Furthermore, this perfection of array and element
pathways must be maintained over time, temperature, and other
variations. It is well known that this can only be achieved with
limited precision, and that the resulting mismatch errors will
severely degrade the ability to form a coherently phased beam. As a
result, element calibration is required.
[0087] To calibrate the array and element pathways, a commonly used
technique is to generate a reference signal at the center frequency
of the array, and couple this reference signal into the receiver
antenna elements such that a known and fixed angle of incidence
(DOA) is achieved. Resources are then allocated within the
beamformer hardware to measure the phase of the reference signal
after propagating through the entire RF pathway for each element.
In this way, the relative phase error for each element path can be
measured over time. Once the phase mismatches are known, a
calibration vector is formed which is embedded in the beamformer
weight calculations in such a way as to remove most of the effects
of the mismatch from the beamforming process. This calibration
vector must be updated periodically to keep up with changes in the
mismatch between elements. In some cases, this calibration method
is sub-optimal, because it ignores any dispersive effects of the
atmosphere that may slightly skew the signal arriving at each
element.
[0088] In the present invention, the calibration process is greatly
simplified and improved. In fact, due to the integration of the
beamformers with the demodulators for each channel and Rake Finger,
calibration can be done for each antenna element signal path in the
absence of any additional calibration signal, and without consuming
any additional matched filter or demodulator processing resources.
Furthermore, calibration can be performed independently on each
signal channel or Rake Finger, taking into account any atmospheric
distortions that may distinguish their different propagation
pathways. Finally, this process of calibration actually closes a
loop between the beamforming process, and the integrated carrier
error terms on a per element basis-the residual carrier phase error
resulting from imperfections in pointing the beamformer are
systematically forced to zero as a result. Thus, the complexity of
calibration is reduced, no additional hardware is required, and the
quality of calibration becomes nearly perfect, on a per-signal
basis.
[0089] The way that this calibration is implemented, at essentially
zero increase in cost, is once again due to the way the present
invention embeds all despread element information with the
demodulator, where the actual beamforming weighting and combining
occurs. In the normal operation of the present invention, the
element rotation complex multiplication occurs sequentially on an
element by element basis, followed by the scalar magnitude
multiplication, after which the products are combined by
integrating and dumping in an accumulator (see FIGS. 1 and 6). In
normal operation, only the combined weighted elements are
subsequently used for symbol demodulation, and only a single PN,
symbol, and carrier tracking loop is formed. As a result, the
individual rotated element samples are not needed. However, those
individual rotations are actually calculated, and can be saved and
integrated for an arbitrary amount of time to achieve an extremely
accurate accumulation of residual carrier phase error per element.
If the pointing were perfect, the integrated error would approach
zero. Any non-zero residue represents calibration error for that
element, which can now be corrected and fed back to the weights.
The only additional cost associated with this calibration process
is additional memory or registers to store the per-element
integrations, and an additional adder to perform the sequential
integration.
[0090] The previous description of the preferred embodiments is
provided to enable any person skilled in the art to make or use the
present invention. The various modifications to these embodiments
will be readily apparent to those skilled in the art, and the
generic principles defined herein may be applied to other
embodiments without the use of the inventive facility. Thus, the
present invention is not intended to be limited to the embodiments
shown herein but is to be accorded the widest scope consistent with
the principles and novel features disclosed herein.
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