U.S. patent application number 09/770853 was filed with the patent office on 2001-08-09 for self-calibrating pricision timing circuit and method for a laser range finder.
Invention is credited to Dunne, Jeremy G..
Application Number | 20010012104 09/770853 |
Document ID | / |
Family ID | 23483000 |
Filed Date | 2001-08-09 |
United States Patent
Application |
20010012104 |
Kind Code |
A1 |
Dunne, Jeremy G. |
August 9, 2001 |
Self-calibrating pricision timing circuit and method for a laser
range finder
Abstract
A highly precise range measurement instrument is made possible
through the use of a novel and efficient precision timing circuit
which makes use of the instruments internal central processing unit
crystal oscillator. A multi-point calibration function includes the
determination of a "zero" value and a "cal" value through the
addition of a known calibrated pulse width thereby providing the
origin and scale for determining distance with the constant linear
discharge of capacitor.
Inventors: |
Dunne, Jeremy G.;
(Littleton, CO) |
Correspondence
Address: |
William J. KUBIDA, Esq.
Hogan & Hartson, LLP
1200 17th Street, Suite 1500
Denver
CO
80202
US
|
Family ID: |
23483000 |
Appl. No.: |
09/770853 |
Filed: |
January 26, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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09770853 |
Jan 26, 2001 |
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09513596 |
Feb 25, 2000 |
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6226077 |
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09513596 |
Feb 25, 2000 |
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09234724 |
Jan 21, 1999 |
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6057910 |
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09234724 |
Jan 21, 1999 |
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08918396 |
Aug 26, 1997 |
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5880821 |
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08918396 |
Aug 26, 1997 |
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08717635 |
Sep 23, 1996 |
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5703678 |
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08717635 |
Sep 23, 1996 |
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08375941 |
Jan 19, 1995 |
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5574552 |
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Current U.S.
Class: |
356/5.1 |
Current CPC
Class: |
G04F 10/105 20130101;
G01S 17/10 20130101; G01S 17/14 20200101; G01C 3/08 20130101; G04F
10/10 20130101; G01S 7/497 20130101; G01S 7/483 20130101 |
Class at
Publication: |
356/5.1 |
International
Class: |
G01C 003/08 |
Claims
What is claimed is:
1. A laser range finder comprising: a laser transmitting section
for producing a series of transmitted laser pulses directable
towards a target and producing a plurality of returned laser pulses
at least partially reflected therefrom in response thereto; a laser
receiving section for receiving at least a portion of said
plurality of returned laser pulses; a central processing section
coupled to said laser transmitting and laser receiving sections for
determining a distance to said target based upon a time of flight
of said transmitted and returned laser pulses; a user viewable
display coupled to said central processing section for displaying
said determined distance to said target; and a user actuatable mode
switch coupled to said central processing section for indicating a
specified distance less than which returned pulses from an object
intervening between said laser range finder and said target will be
ignored in said distance determination.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] The present application is a continuation of co-pending U.S.
patent application Ser. No. 09/513,596, filed Feb. 25, 2000, which
is a continuation of U.S. patent application Ser. No. 09/234,724,
now U.S. Pat. No. 6,057,910, which is a continuation of U.S. patent
application Ser. No. 08/918,396, Now U.S. Pat. No. 5,880,821, which
is a continuation of U.S. patent application Ser. No. 08/717,635,
now U.S. Pat. No. 5,703,678, which is a continuation of U.S. patent
application Ser. No. 08/375,941, filed Jan. 19, 1995, now U.S. Pat.
No. 5,574,552. The present invention is also related to those
disclosed and claimed in U.S. Pat. No. 5,652,651 for: "Laser Range
Finder Having Selectable Target Acquisition Characteristics and
Range Measuring Precision"; and U.S. Pat. No. 5,612,779 for:
"Automatic Noise Threshold Determining Circuit and Method for a
Laser Range Finder", all filed concurrently herewith and assigned
to the assignee of the present invention, Laser Technology, Inc.,
Englewood, Colo., the disclosures of which are hereby specifically
incorporated by this reference.
BACKGROUND OF THE INVENTION
[0002] The present invention relates, in general, to the field of
distance or range measuring equipment. More particularly, the
present invention relates to a laser based range finder which may
be inexpensively produced yet provides highly accurate precision
range measurements of up to 1000 yards or more with a resolution of
less than 1 yard. The laser range finder herein disclosed has a
number of user selectable target acquisition and enhanced precision
measurement modes which may viewed on an in-sight display during
aiming and operation of the instrument. Extremely efficient
self-calibrating precision timing and automatic noise threshold
circuits incorporated in the design provide a compact, low-cost,
highly accurate and reliable ranging instrument for a multitude of
uses.
[0003] Laser based distance and range measuring equipment have been
used for a number of years to provide extremely accurate distance
measurements to a remote target or object. A representative
instrument is the Criterion.TM. 100 laser range finder developed
and marketed by Laser Technology, Inc., assignee of the present
invention. Although a highly accurate and reliable device, its
great distance ranging capability and inherent complexity
translates to a cost and form factor most suitable only for certain
specific applications. A need therefore exists for a laser based
range finder of perhaps more limited range, which can be
economically manufactured as a rugged, compact unit to provide
accurate distance measurement capabilities in other less stringent
types of applications.
SUMMARY OF THE INVENTION
[0004] Herein disclosed is a precise, yet accurate and reliable
laser range finder which may be economically produced and is
adapted to individual portable use in a unit potentially weighing
less than a pound with an on-board battery based power supply.
Moreover, the compact instrument herein provided has a number of
user selectable target acquisition operational modes which may be
invoked depending on the distance, type and reflectivity of the
target being sighted.
[0005] Through the use of an in-sight display, distance or range
information can be shown while the user may also view and select
the instrument's mode of operation through successive actuations or
a push button mode switch while simultaneously sighting the target
object. A precision mode of operation may also be invoked in which
an even more precise measurement to an object may be achieved
following an initial measurement together with the visual
indication of "precision flag" on the in-sight display.
[0006] A highly precise range measurement is made possible through
the use of a novel and efficient timing circuit which makes use of
the instrument's internal central processing unit crystal
oscillator. A likewise unique automatic noise threshold determining
circuit allows for instrument operation with a low signal-to-noise
ratio to optimize sensitivity and performance in conjunction with a
processor based pulse discrimination procedure which, nevertheless
assures accurate range measurements.
[0007] The unit herein disclosed can be utilized in a multitude of
endeavors including such recreational activities as golf where it
can be utilized to very accurately determine the distance to a flag
or pin as well as to trees and other natural objects. The
principles of the invention are further applicable to the design of
a laser based "tape measure" where ranges can be precisely measured
with resolutions of on the order of an inch or less.
[0008] Specifically disclosed herein is a self-calibrating,
precision timing circuit and method for determining a range to a
target based upon a flight time of a pulse toward the target. The
circuit comprises means for initially establishing first and second
reference voltage levels together with means for unclamping the
second reference voltage level and means for allowing the second
reference voltage level to then diminish at a first rate to the
first reference voltage level. Further provided are means for
storing a first reference time extending from the step of
unclamping until the first and second reference voltage levels are
determined to be equal. Means are also provided for then
re-establishing the first and second reference voltage levels
together with means for again unclamping the second reference
voltage level. Additional means are provided for increasing the
second reference voltage level at a second higher rate than the
first rate for a predetermined period of time to establish a third
reference voltage level together with means for then allowing the
third reference voltage level to diminish at the first rate to the
first reference voltage level at which time, a second reference
time extending from the step of again unclamping until the first
and third reference voltage levels are equal is additionally
stored. The first and second reference voltage levels are again
re-established and the second reference voltage level is further
unclamped. Means are provided for again increasing the second
reference voltage level at the second higher rate for a period of
time related to the flight time of the pulse to the target to
establish a fourth reference voltage level, together with means for
then allowing the fourth reference voltage level to diminish at the
first rate to the first reference voltage level. A third reference
time extending from the unclamping of the second reference voltage
level until the first and fourth reference voltage levels are equal
is then stored and the range to the target may be computed as
proportional to the quantity of the (third reference time minus the
first reference time) divided by the quantity of the (second
reference time minus the first reference time).
[0009] In a particular embodiment the establishing means may
comprise a transistor switch for coupling a capacitor to a source
of the second voltage while the unclamping means may comprise a
second transistor switch for decoupling the capacitor from the
second voltage source. The allowing means may comprise a third
transistor switch coupling a resistor to the capacitor to bleed off
the charge therefrom.
[0010] The means for increasing the second reference voltage level
may comprise means for applying a charge to the capacitor at the
second rate and the predetermined time period specified may be
determined by reference to a crystal oscillator. In a particular
embodiment, the second charging rate may be substantially 1000
times the first discharging rate.
DETAILED DESCRIPTION OF THE DRAWINGS
[0011] The foregoing and other features and objects of the present
invention and the manner of attaining them will become more
apparent and the invention itself will be best understood by
reference to the following description of a preferred embodiment
taken in conjunction with the accompanying drawings, wherein:
[0012] FIG. 1 is a simplified logic block diagram of a laser range
finder in accordance with the present invention illustrating the
significant functional aspects thereof, inclusive of a laser signal
transmitting and receiving section, central processing unit and the
precision timing and automatic noise threshold sections
thereof;
[0013] FIG. 2 is a detailed schematic diagram of the laser transmit
section of FIG. 1 illustrating, inter alia, the laser signal
producing diode and the associated driving and reference signal
producing circuitry;
[0014] FIG. 3 is an additional detailed schematic diagram of the
laser receive section of FIG. 1 illustrating, inter alia, the laser
signal receiving diode, transimpedance amplifier and the precision
comparator for establishing the V.sub.threshold and RX(OUT+)
signals for the precision timing and automatic noise threshold
circuits;
[0015] FIGS. 4 and 5 are further detailed schematic diagrams of the
precision timing section of the laser range finder of FIG. 1
illustrating the circuit nodes for establishing the voltages
V.sub.1 and V.sub.2 during the zero, calibration ("CAL") and laser
firing phases of operation;
[0016] FIG. 6 is an additional detailed schematic diagram of the
central proceeding unit ("CPU") portion of the laser range finder
of FIG. 1 illustrating the CPU, associated oscillator and the
in-sight liquid crystal display ("LCD") for displaying measured
distances to an operator of the laser range finder in addition to
the various signals for operative association with the precision
timing and automatic noise threshold sections thereof;
[0017] FIGS. 7A, 7B and 7C are individual graphic representations
of the voltages V.sub.1 and V.sub.2 of certain of the precision
timing section circuit nodes during the zero, calibration and laser
firing phases of operation from which the values Zero.sub.TIME,
CAL.sub.TIME and LASER.sub.TIME are derived to enable rapid and
accurate calculation of the distance to an object from the laser
range finder; and
[0018] FIG. 8 is a final detailed schematic diagram of the
automatic noise threshold section of the laser range finder of FIG.
1 illustrating the various components thereof as well as the
signals coupling the same to the laser receive section and CPU.
DESCRIPTION OF A PREFERRED EMBODIMENT
[0019] With reference now to FIG. 1, a logic block diagram of a
laser range finder 10 in accordance with the present invention is
shown. The laser range finder 10 includes, in pertinent part, a
main power supply unit ("PSU") 12 as operatively controlled by a
trigger switch 14. The main power supply unit 12 is coupled to a
high voltage ("HV") power supply unit 16 for supplying operating
power in conjunction with the main power supply unit 12 to a laser
transmit section 18.
[0020] The laser transmit section 18 activates a laser emitting
diode 20 for directing a laser signal toward an object in the
operation of the laser range finder 10. The laser transmit section
18 also supplies a /FIRE signal to the central processing unit
("CPU") section 28 as will be more fully described hereinafter.
[0021] The main power supply unit 12 also supplies operating power
to a laser receive section 22 which further has as an input a
signal generated by a laser receiving diode 24 as the laser signal
emitted from the laser emitting diode 20 is reflected from an
object back thereto. The laser receive section 22 supplies a
V.sub.threshold Signal and RX(OUT+) signal to an automatic noise
threshold section 36 and a precision timing section 34 both of
which will be described in more detail hereinafter.
[0022] The CPU section 28 receives as one input a signal from a
mode switch 26 by means of which an operator can change the
operating mode and functional operation of the laser range finder
10. An oscillator 30 supplies a clocking signal to the CPU section
28 as well as to the precision timing section 34. The CPU section
28 provides an output indicative of the distance from the laser
range finder 10 to an object as sighted through a viewing scope
thereof on an insight liquid crystal display ("LCD") 32.
[0023] The precision timing section 34 provides a number of signals
to the CPU section 28 including a TIMER and /RX DETECT signals as
shown and receives a RUN/CLAMP signal back therefrom. The CPU
section 28 provides a number of signals to the precision timing
section 34 including a HOLD OFF, NORM/CAL, /RESET, and a CAL DITHER
signal. The automatic noise threshold section 36 also receives a
number of inputs from the CPU section 28 including a number of
noise set ("INSET") signals and a REFLECTION MODE signal to
operatively control its function.
[0024] With reference additionally now to FIG. 2, the laser
transmit section 18 is shown in more detail. The laser transmit
section 18 receives a transmit ("TX") BIAS signal on supply line 50
of approximately 110 to 140 volts for application through resistor
52 to the emitter of transistor 54. The emitter of transistor 54 is
coupled to its base by means of a resistor 58 which also couples
the collector of transistor 56 to resistor 52. The emitter of
transistor 56 is connected to circuit ground on ground line 60. A
capacitor 62 couples the emitter of transistor 54 to the cathode of
the laser emitting diode 20 which has its anode also connected to
circuit ground 60. An additional diode 64 is coupled in parallel
with the laser emitting diode 20 having its anode connected to the
cathode of the laser emitting diode 20 and its cathode connected to
circuit ground 60. A resistor 66 is placed in parallel with the
laser emitting diode 20 and the diode 64.
[0025] A source of +5 volts is also received by the laser transmit
section 18 on supply line 68 through resistor 70. Resistor 70 is
coupled to the emitter of transistor 72 as well as to circuit
ground 60 through a capacitor 74. A resistor 76 couples the emitter
transistor 72 to its base which is coupled through resistor 78 to
line 80 for supplying a /FIRE signal to the CPU section 28 (shown
in FIG. 1).
[0026] An additional diode 82 has its anode connected to the
collector of transistor 72 and its cathode coupled to circuit
ground 60 through resistor 86. A capacitor 84 couples the cathode
of diode 82 to the common connected collector of transistor 54 and
base of transistor 56. The common connected collector of transistor
54 and base of transistor 56 is coupled through a voltage divider
network comprising resistor 88 and resistor 90 to circuit ground. A
resistor 92 coupled between resistor 88 and resistor 90 provides a
REF signal on line 94 for application to the precision timing
section 34 (shown in FIG. 1).
[0027] With reference additionally now to FIG. 3, the laser receive
section 22 is shown in more detail. The output signals of the laser
receive section 22 are the signals RX (OUT+) and V.sub.threshold
provided on lines 100 and 102 respectively for application to the
precision timing section 34 and automatic noise threshold section
36 as previously shown in FIG. 1. A source of +5 volts providing a
receive ("RX") BIAS signal is input to the laser receive section 22
from the HV power supply unit 16 on supply line 104. A low pass
filter network 106 comprising resistors 108 and 112 in conjunction
with capacitors 110 and 114 couples the supply line 104 to circuit
ground 60 to provide a bias signal to the cathode of the laser
receiving diode 24. The laser receiving diode 24 has its anode
connected to the base of transistor 118 which, in conjunction with
transistors 120,122, and 124 comprises a transimpedance amplifier
116 providing an output on node 126 which is capacitively coupled
to the "+" input of a precision comparator 134. A source of +5
volts is input to the laser receive section 22 from the main power
supply unit 12 (shown in FIG. 1) for input to the transimpedance
amplifier 116 through a low pass filter comprising resistor 130 and
capacitor 132. The +5 volt RX supply voltage is also coupled to the
V+ input of the precision comparator 134 through resistor 136 and
is coupled to circuit ground through capacitor 138. The "+" input
of the precision comparator 134 is connected between the plus 5
volt RX voltage source and circuit ground 60 through the node
intermediate resistor 142 and resistor 144.
[0028] The precision comparator 134 which may, in a preferred
embodiment, comprise a MAX 913 low power precision
transistor-transistor logic ("TTL") comparator available from Maxim
Integrated Products, Inc., Sunnyvale, Calif., has its "v", `LE" and
ground ("GND") inputs connected to circuit ground 60 as shown. A
capacitor 146 couples the "-" output of the precision comparator
134 to circuit ground 60 as shown. The "O+" output of the precision
comparator 134 is supplied through a resistor 148 to line 100 to
provide the Rx(OUT+) signal while the "-" output of the precision
comparator 134 is supplied through resistor 150 to line 102 to
provide the V.sub.threshold signal.
[0029] With reference additionally now to FIG. 4, a portion of the
precision timing section 34 (shown in FIG. 1) is illustrated. A CPU
clock ("CLK") signal is input to the precision timing section 34 on
line 152 to the CLK input of a serial in/parallel out shift
register 160 from the oscillator 30 as previously shown in FIG. 1.
An additional input to the shift register 160 is received on line
154 comprising a NORM/CAL signal from the CPU section 28 to the
data set B ("DSB") input thereof. The active low clear ("{overscore
(CLR)}") input and DSA input are held high as shown.
[0030] An additional input to the precision timing section 34 is
received from the CPU section 28 (shown in FIG. 1) on line 156
comprising a /RESET signal for input to the reset ("{overscore
(R)}") inputs of D type flip-flop 158 and flip-flop 162. The
{overscore (Q)} output of flip-flop 158 is supplied as one input to
an invertor comprising a portion of d NAND Schmitt trigger 168
through a low pass filter comprising resistor 164 and capacitor 166
as shown. The remaining input to the invertor 168 is connected to a
source of +5 volts.
[0031] A resistor 172 couples a source of +5 volts to the collector
of transistor 174 having its emitter coupled to circuit ground. The
collector terminal of transistor 174 is coupled through capacitor
170 to the input of the inverter 168 coupled to the {overscore (Q)}
output of flip-flop 158. Transistor 174 has its based coupled to
circuit ground through resistor 176 and receives a HOLD OFF signal
on node 178 received from the CPU section 28.
[0032] The flip-flop 158 receives an input to its CLK terminal on
line 94 comprising the REF output signal from the laser transmit
section 18 (shown in FIG. 1). Its data ("D") input is coupled to a
source of +5 volts and the Q1 output of the shift register 160 is
provided to the active low set ("{overscore (S)}") input as shown.
The Q output of flip-flop 158 is supplied as one input to a
transmit gate 204 having its other input coupled to the output of
an inventor comprising an additional HAND Schmitt trigger 202.
Inventor 202 has one input connected to a source of +5 volts and
another input connected to the Q output of flip-flop 162. Flip-flop
162 has its {overscore (S)} input coupled to the Q7 output of shift
register 160 and its D input connected to the output of inventor
168. The {overscore (Q)} output of flip-flop 162 is supplied on
line 184 to comprise a /RX DETECT signal for input to the CPU
section 28 (shown in FIG. 1). The flip-flop 162 has its CLK input
connected to line 100 for receiving the RX(OUT+) signal from the
laser receive section 22 (shown in FIG. 1) which is also supplied
as one input to NAND Schmitt trigger 180. The other input of HAND
Schmitt trigger 180 is connected to line 184 through resistor 182
and coupled to circuit ground through capacitor 186. The output of
Schmitt trigger 180 is supplied to the base electrode of transistor
200 which has its collector terminal coupled to circuit ground.
Line 196, comprising an analog-to-digital ("A/D") POWER CORRECTION
signal is supplied to the emitter terminal of transistor 200
through resistor 198 as well as to the collector terminal of
transistor 190 which is coupled to circuit ground through capacitor
194. The /RESET signal on line 156 is supplied to the base terminal
of transistor 190 through resistor 188. A source of +5 volts is
connected to the emitter of transistor 190 as well as through
resistor 192 to the base of transistor 190 to provide an operating
bias.
[0033] Referring additionally now to FIG. 5, the remaining portion
of the precision timing section 34 (shown in block form in FIG. 1)
is illustrated. The HOLD OFF signal output from CPU section 28 to
the precision timing section 134 is supplied on line 258 through
resistor 256 to node 178 for input to the base of transistor 174
(shown in FIG. 4).
[0034] The output of transmit gate 204 appearing on node 206 is
supplied through resistor 208 to the base terminal of transistor
210. A source of +5 volts is supplied to the emitter terminal of
transistor 210 through the series connection of resistor 216 and
resistor 222. The node intermediate resistors 216 and 222 is
coupled to circuit ground through the parallel combination of
capacitors 218 and 222 as well as to the output of comparator 236
through resistor 246 to provide a TIMER signal on line 250 for
input to the CPU section 28 as will be more fully described
hereinafter. The source of +5 volts is also connected to the base
terminal of transistor 210 through the series connection of
resistors 216 and 224. A V.sub.1 node 228 at the common connected
base of transistor 212 and emitter of transistor 214 is coupled
through a source of +5 volts through resistor 216 and resistor 226.
Node 228 is connected through resistor 230 to V.sub.2 node 232
which, in turn, is connected to circuit ground through resistor
240. A capacitor 238 couples V.sub.1 node 228 to circuit ground.
V.sub.2 node 232 is connected to the "-" input of comparator 236.
V.sub.1 node 228 is connected to line 254 from the CPU section 28
(shown in FIG. 1) to receive the CAL DITHER signal through resistor
252.
[0035] The collector terminal of transistor 210 is coupled to the
collector terminals of transistors 212 and 214 as well as to the
"+" terminal of comparator 236 which, in turn, is coupled to
circuit ground through capacitor 244. A {overscore (RUN)}/CLAMP
signal output from the CPU section 28 (shown in FIG. 1) is
furnished on line 260 through resistor 248 for input to the base
terminal of transistor 214.
[0036] With reference additionally now to FIG. 6, the CPU section
28 is shown in greater detail. The CPU section 28 comprises, in
pertinent part, a microcomputer 270 which may, in a preferred
embodiment, comprise a ST6240 device. An 8 megaHertz ("MHz")
crystal 274 forms a portion of the oscillator 30 for providing an
oscillator ("OSCIN") and oscillator out ("OSCOUT") signal to the
microcomputer 270 as well as supplying a CPU CLK signal on line 152
for input to the precision timing section 34 as previously
described. The VDD input of microcomputer 270 is coupled to a
source of +5 volts and the /RESET input thereof is held high
through pull up resistor 276 which is coupled to circuit ground
through capacitor 278. Output from the microcomputer 270 is taken
on a display bus 280 comprising the communication ("COM") lines COM
1-COM 4 and 516-528 lines for input to the LCD display 32.
[0037] An A/D LOW BATTERY signal, a TRIGGER signal, and a POWER
CONTROL signal are input to the microcomputer 270 on lines 284,
286, and 288 respectively. The A/D LOW BATTERY signal on line 284
is also supplied to the "-" input of comparator 296 which is
coupled to circuit ground through capacitor 304. The "+" input of
comparator 296 is coupled to a source of +5 volts through resistor
298 which is also coupled to circuit ground through the parallel
combination of resistor 300 and capacitor 302. The output of
comparator 296 appearing on line 306 provides a SHUTDOWN signal for
the laser range finder 10 in the event the onboard battery voltage
drops below a predetermined limit.
[0038] The microcomputer 270 supplies the HOLD OFF signal on line
258, the {overscore (RUN)}/CLAMP signal on line 260, the CAL DITHER
signal on line 254, the /RESET signal on line 156 and the NORM/CAL
signal on line 154 for input to the precision timing section 34 as
has been previously described. The microcomputer 270 receives as
outputs from the precision timing section 34 the /RX DETECT signal
online 184 and the TIMER signal on line 250. Additional inputs to
the microcomputer 270 are the /FIRE signal on line 80 from the
laser transmit section 18 (shown in FIG. 1) as well as the A/D
POWER CORRECTION signal on line 196 from the precision timing
section 34 (as shown in FIG. 4). A MODE input signal on line 294 is
received from the mode switch 26 which is otherwise held to a +5
volts through resistor 292. Microcomputer 270 supplies an NSET1 and
NSET2 signal on lines 308 and 310 respectively as well a REFLECTION
MODE signal on line 312 for input to the automatic noise threshold
section 36 (as shown in FIG. 1).
[0039] In overall operation, a reference signal (REF) on line 94 is
generated by the laser transmit section 18 (shown in FIG. 2) when
the laser range finder 10 is fired by placing a current pulse
through the laser emitting diode 20 in response to manual actuation
of the trigger switch 14. The REF signal on line 94 is derived from
the current placed through the laser emitting diode 20 and not from
the light pulse itself and is sufficiently precise for accurately
indicating the time of the laser firing. The REF signal is
ultimately input to the CLK input terminal of flip-flop 158, which
has its Q output coupled to the transmit gate 204, which then turns
on the current switch comprising transistor 210, and starts
charging the capacitor 244. When the receive pulse RX(OUT+) on line
100 comes back from the laser receive section 22 (shown in FIG. 3),
it triggers the flip-flop 162 at its CLK input. Flip-flop 162 has
its Q output coupled to the input of invertor 202 which then shuts
the transmit gate 204 off, stopping the current pulse. At this
point, a constant current sink discharges capacitor 244. In this
manner, capacitor 244 is charged up with a relatively large current
(on the order of 10 milliamps), and later discharged with a small
current (on the order of 10 microamps) applied over the entire
flight time of the laser pulse from its firing from the laser
emitting diode 20 to its reflection from a target back to the laser
receiving diode 24. Because the laser range finder 10 is intended
for a shorter maximum range than other laser based range finding
instruments, the use of this technique does not require a separate
counting oscillator followed by an interpolation operation and the
entire flight time is essentially stretched by a factor of 1000 and
then the stretched result is counted. By charging capacitor 244 at
a fast rate and then discharging it and then monitoring the time it
takes to discharge, the flight time is expanded so that the slower
clock in the CPU section 28 can then count it accurately. The
microcomputer 270 utilized in the CPU section 28 has a 1.5
microsecond resolution and, because the incoming flight times been
expanded by a factor of 1,000 on the input side to the precision
timing section 34, it is the equivalent of a 1.5 nanosecond
resolution, which corresponds to a measurement resolution for the
laser range finder 10 of on the order of nine inches. Therefore,
given that the laser range finder 10 is intended to be a one-yard
instrument with a nine-inch resolution, sufficient resolution is
provided to be able to measure distances up to a thousand yards to
a one-yard accuracy.
[0040] The precision timing section 34 of the laser range finder 10
has three distinct modes of operation including a zero calibration,
fixed pulse width calibration and laser measurement function as
will be more fully described hereinafter. The portion of the
precision timing section 34 comprising transistors 210, 214, and
212 (shown in FIG. 5) is the essence of the integrating flight time
expander. Transistor 210 functions as a current switch which is
turned on for the duration of the laser flight time in the laser
mode of operation and is also turned on for the duration of
whatever calibration pulse is placed into it during the calibrate
mode. In the latter instance, a calibration pulse is supplied by
the shift register 160 via flip-flop 158 and the start and end of
the calibration pulse is gated via transmit gate 204 to actually
turn the transistor 210 on and off in order to function as a
current source, typically sourcing 10 milliamps of current. It
should be noted that prior to turning transistor 210 on, transistor
214 must first be turned off and, when the system is in the reset
state ready to start the whole measurement sequence, transistor 210
is off. Transistor 212, which is the current sink in the system, is
always on, and typically sinks on the order of 10 microamps of
current. In the reset condition, transistor 214 is on, and that
clamps the voltage at the top plate of capacitor 244 to a voltage
level designated as V1 at node 228. A voltage V2 is defined as the
voltage at node 232 at the "-" input of comparator 236. It should
also be noted that a metal oxide semiconductor field effect
transistor ("MOSFET") may be utilized for transistor 244 and would
exhibit a much lower offset than the bipolar device shown. However,
due to the lower cost of bipolar transistors and the fact that any
offset cancels during the processing of the signal, a bipolar
transistor is entirely adequate for this purpose.
[0041] When transistor 214 is on, the voltage on the positive plate
of capacitor 244 is clamped to voltage V1, plus a fixed offset due
to the transistor 210, which is small and typically on the order of
50 millivolts. During the zero calibration function, transistor 214
is turned on by holding the {overscore (RUN)}/CLAMP signal on line
260 high, thereby applying a positive current to its base through
resistor 248. To initiate the zero calibration, the TIMER signal on
line 250 is asserted and supplied to the microcomputer 270 of the
CPU section 28. Utilizing the ST6240 unit shown in FIG. 6, when the
microcomputer TIMER pin is held high, the device is counting.
Conversely, the microcomputer stops counting when the pin is
allowed to go low. In operation, the output comparator 236,
determines whether or not the voltage at the top plate of capacitor
244 is greater or less than V2, and its output determines whether
the TIMER pin on the microcomputer 270 is high or low. In the
normal reset condition, the output of the comparator 236 is high,
which means the timer's active. In sequence, the microcomputer 270
initiates the TIMER function and then turns off transistor 214 by
lowering the control signal {overscore (RUN)}/CLAMP on line 260, to
unclamp capacitor 244. Capacitor 244 then starts discharging
towards zero due to the current being drained out of it via
transistor 212 at a rate of about ten microamps. When it has
discharge such that the charge removed drops the voltage V1 at node
228 to the level of V2, the output of the comparator 236 changes
state to stop the TIMER function. (In the particular embodiment
shown, V1 is typically on the order of 1.0 volts and V2 is about
0.9 volts.) The microcomputer 270 of the CPU section 28 now has a
count value that relates to the amount of time it takes for
capacitor 244 to discharge from V1 down to V2. This process is
repeated several times and the result is averaged. Typically ten
iterations may be performed with the results accumulated and an
average time computed.
[0042] As shown particularly with respect to FIG. 5, the CAL DITHER
signal on line 254 is applied to the base terminal of transistor
212 and is utilized during both the zero calibration and fixed
pulse width calibration times and incorporates a relatively high
value resistor 252. The CAL DITHER signal allows for the
introduction of a deliberately controlled change in the discharge
current in order that the resultant count will vary slightly such
that when the total counts are averaged together, a finer
resolution is produced than would be the case merely using a fixed
current to get the same count value. An adjustment of one part in
about a thousand is provided during the zero calibration and, fixed
pulse width calibration modes because the finite resolution of the
microcomputer 270 timer otherwise provides discreet timing
intervals of 1.5 nanoseconds which would only provide distance
measurement resolution of approximately one yard. In operation, the
zero calibration count in the microcomputer 270 will typically be
about 150 while in the fixed pulse width calibration mode it will
be on the order of 900. The flight time count during the laser mode
of operation can be anything from close to the zero calibration
value to about 4500.
[0043] For example, during the zero calibration mode, the count
value in the microcomputer 270 might be 150 but there is no way of
knowing just how close the count actually is to 149 to 151. By
utilizing the CAL DITHER signal to force the count over a couple of
count boundaries (for example: 150, 150, 150, 151, 151, 152) the
resolution of the counter may be effectively raised by a factor of
two without having to utilize additional fine counters. In the
embodiment shown, the resultant resolution is sufficient to
maintain calibration to plus or minus one yard over a range of one
thousand yards or less. Although implementations may vary the CAL
DITHER signal may be held high for five out of ten pulses and low
for the remainder to provide the foregoing resolution
enhancement.
[0044] Due to the fact that the actual laser flight time varies due
to noise in the laser pulses and variability in target aiming,
there is generally enough scatter in the measured laser flight time
such that it covers more than one clock boundary and so will
automatically average to a higher resolution through the use of the
precision timing section 34 without invoking the CAL DITHER
function in the laser mode of operation.
[0045] With reference additionally now to FIGS. 7A, 7B and 7C, the
operation of the precision timing section 34 is shown in the zero
calibration, fixed pulse width calibration and laser measurement
function modes of operation respectively. In its normal state, the
voltage on the top plate of capacitor 244 is clamped at V.sub.1,
and at a time T.sub.0, the precision timing section 34 will
initiate the TIMER by changing the output state of comparator 236
to the logic high state. After a very short fixed number of
instructions later shown as T.sub.1, the clamp transistor 214 will
be turned off and the voltage on capacitor 244 will begin
discharging slowly until that voltage crosses V.sub.2 at time
T.sub.3 when the output of comparator 236 will change state. In
essence, during the zero calibration process, transistor 210 is
never turned on thereby determining the timing conditions of what
would effectively be a zero flight time. Therefore, if there's no
charge current applied to capacitor 244, T.sub.3 - T.sub.0 zero is
the time that would be in the microcomputer 270 and the timer in
whatever units they operate, which is usually dependent on the CPU
section 28 crystal frequency. In the embodiment shown, the
microcomputer 270 utilizes an 8 MHz crystal and the internal timer
has a 1.5 microsecond resolution resulting in a count of about
150.
[0046] During the fixed pulse width calibration process (shown
particularly in FIG. 7B) at time T.sub.4, once again the
microcomputer 270 stops the TIMER and a short time later at T.sub.5
it releases the clamp. At T.sub.6, a known pulse width is applied
to the base terminal of transistor 21 which is precisely derived
from the main oscillator 30 as applied to the CLK input of the
shift register 160. The signal applied to the CLK input of the
shift register 160 directly tracks the main oscillator 30 and the
serial data input to the shift register 160 is a logic line 154
from the CPU section 28 designated NORM/CAL. When the NORM/CAL
signal is high, the precision timing section 34 is in its normal
mode of operation and, when it drops to a logic low state, the
fixed pulse width calibration function is initiated. Thereafter,
typically about fifty microseconds later, at time T.sub.6 the
NORM/CAL, signal on line 154 will be dropped low. It should be
noted that during both the zero and the fixed pulse width
calibration modes, the logic reset signal /RESET on line 156 is
held low, its active state. In the logic low state the two
flip-flops 158, 162 determine whether the input signal comes from
shift register 160 which generates the fixed pulse width or whether
it comes from the REF and RX(OUT+) signals and relates to an actual
laser flight time. The /RESET signal is generally held low at all
times during the fixed pulse width calibration process so that any
noise on the RX(OUT+) receive line 100 will not accidently clock
flip-flop 162 and therefore trigger the precision timing section 34
resulting in an indeterminate time period measurement invalidating
the calibration. The reset state for the Q outputs of flip-flops
158, 162 is low but is high for the {overscore (Q)} outputs.
Therefore, the {overscore (Q)} outputs can not be directly driven
with the reset circuit and must be driven off the {overscore (Q)}
outputs in both cases which introduces a small fixed offset delay
which must be accounted for later. As soon as the NORM/CAL signal
on line 154 is dropped low, which occurs, approximately 50
microseconds after the clamp has been released, the low signal
propagates through the shift register 160 precisely with the main
oscillator 30 clock. The Q0 output of the shift register 160 is the
first to be triggered but is not used because it is used to
synchronize with the incoming signal. The Q1 is then the first
output of the shift register 160 to be utilized and on every
positive edge of the clock the Zero signal that is applied into the
serial input will propagate one state of the shift register 160
from Q zero to Q7. Therefore, the Q1 output will go low first, and
as soon as that output goes low, the set line input {overscore (S)}
forces the Q output of flip-flop 158 to go high since the Q output
of flip-flop 162 is in the low state. As a result, logic level ones
appear at the two inputs of the transmit gate 204, which turns on
the current switch transistor 210. Exactly six clocks later, the
same thing happens with flip-flop 162 which has its {overscore (S)}
input coupled to the Q7 output of the shift register 160. As the Q
output of flip-flop 162 goes high, the output of the invertor 202
goes low, and the transmit gate 204 will be turned off. At this
point the count pulse will stop meaning that the fixed width pulse
feeding the current switching circuit at the output of the transmit
gate 204 is precisely six clock cycles. The time difference between
the (Q1 and Q7 outputs of the shift register 160 is exactly 750
nanoseconds when utilizing an 8 MHz oscillator 30 applied to its
CLK input. The invertor 202 adds an additional delay of about 10
nanoseconds for a total of delay of about 760 nanoseconds which
varies only slightly with temperature, perhaps one or two
nanoseconds, yet still provides sufficient precision for
measurements of less than one yard resolution.
[0047] Transistor 210 is then turned on for a period of time
between T.sub.6 and T.sub.7 to enable the capacitor 244 to charge
very rapidly and then discharge at the same rate as has been
previously shown with respect to FIG. 7A. As V1 reaches the level
of V2 the TIMER signal goes low at Time T.sub.8. The fifty
microsecond delay between the unclamping at T.sub.5 and T.sub.6 is
to allow the clamp transistor 214 to turn off fully since it is a
relatively inexpensive bipolar device. If a MOSFET were used
instead, its turn off would be virtually instantaneous and the
additional delay it introduced would not be a problem because the
microcomputer 270 couldn't issue the next instruction quickly
enough Utilizing a bipolar device, approximately 20 microseconds
are required for the discharge to become linear and the slope of
the discharge curve between T.sub.7 and T.sub.8 is then identical
to the slope from T.sub.1 to T.sub.3 in the zero calibration mode
except for the step due to the charging of capacitor 244. As a
consequence, the value of ZERO.sub.TIME equals T.sub.3 minus
T.sub.0 and the value of CAL.sub.TIME value equals the time due to
the CAL.sub.TIME value not due to the ZERO.sub.TIME value, which
is, T.sub.8 minus T4 minus the ZERO.sub.time value or, T.sub.8
minus T.sub.3.
[0048] In essence then, very small flight times are effectively
disregarded and the value of CAL.sub.TIME is known. Therefore, with
the zero calibration function and the addition of a known
calibrated pulse width, the time delay at zero is known together
with the time delay for the known pulse width providing the origin
and scale for determining distance with a constant linear discharge
of capacitor 244.
[0049] With particular reference additionally to FIG. 7C, the
operation of the precision timing section 34 is shown in the laser
measurement mode of operation. The laser measurement operation is
essentially the same as the fixed pulse width calibration mode
except that the NORMAL/CAL signal on line 154 to the shift register
160 is held high and the /RESET signal on line 156 is taken high at
time T.sub.9 to enable the flip-flops 158, 162 to trigger. At time
T.sub.10 the timer is started and at T.sub.11, (at precisely the
same relationship T.sub.11 minus T.sup.10 equals T.sub.5 minus
T.sub.4 equals T.sub.1 minus T.sub.0) the clamp is released. There
is normally a fifty microsecond wait and then the laser pulse is
fired when the microcomputer 270 asserts the /FIRE signal on line
80 to initiate the firing sequence. Upon firing the laser emitting
diode 20, the laser transmit section sends the REF signal on line
94 to the CLK input of flip-flop 158 of the precision timing
section 34. This opens the transmit gate 204 which turns on the
current source transistor 210, which, in turn, charges capacitor
244 at a known rate.
[0050] When the reflected laser pulse comes is detected by the
laser receiving diode 24 of the laser receive section 22 (shown in
FIG. 3), the RX(OUT+) signal on line 100 is directed to the CLK
input of flip-flop 162. The Q output signal of flip-flop 162 is
inverted by invertor 202 which turns off the transmission gate 204
so that the current source transistor 210 is on for the flight time
duration of the laser pulse to charge capacitor 244 to a level
determined by the timer during that flight time. The charge applied
to the capacitor 244 may be anything from just a few millivolts
(essentially zero distance and flight time) to up to two volts
(maximum range and flight distance) depending on the distance to
the target. Time T.sub.12 represents the firing of the laser as
indicated by the REF signal and T13 represents the receipt of the
reflected laser signal as indicated by the RX(OUT+) signal.
Transistor 210 is turned on at T.sub.12 and turned off at T.sub.13.
As a consequence, V.sub.1 will equal V.sub.2 at anytime between
T.sub.14A (minimum distance when T.sub.12 and T.sub.13 are
essentially coincident) and T.sub.14B (maximum range of the laser
range finder 10). Times T.sub.14A through T.sub.14B represent the
range of times (depending on the distance to the target) when the
value of V1 is discharged below the level of V2 and the comparator
236 output changes state stopping the timer.
[0051] The actual laser flight time LASER.sub.TIME (or
FLIGHT.sub.TIME) then equals T.sub.14A (or T.sub.14B) minus
T.sub.10 minus ZERO.sub.TIME or, T.sub.14 minus T.sub.13, The time
T.sub.8 has to be greater than T.sub.3, and T.sub.14 is greater
than or equal to T.sub.3. There is no theoretical limit on the
lower range of the laser range finder 10 and flight time (and
distance) can be measured down to zero due to its linearity. The
only factors in the near zero range are the time it takes
transistor 210 to turn on, the propagation time of the laser beam
and the various circuit gates, but since the time for each of these
factors is the same during calibration as during flight time, they
essentially cancel out. The precision timing section 34 can be
effectively utilized down to on the order of ten nanoseconds and
still remain perfectly linear. RANGE to a target is then a
constant, "k" times the quantity FLIGHT.sub.TIME ZERO.sub.TIME over
CAL.sub.TIME-ZERO.sub.TIME.
[0052] For each of the values: ZERO.sub.TIME, CAL.sub.TIME and
FLIGHT.sub.TIME values are accumulated and are expressed in time
units that derive from the very accurate crystal oscillator 30.
Typically, ten pulses may be utilized to establish the
ZERO.sub.TIME average, ten pulses to establish the CAL.sub.TIME
average and ten pulses to establish the minimum precision (or
rough) FLIGHT.sub.TIME range to the target. Another group of ten
through thirty laser pulse FLIGHT.sub.TIMES may be also averaged in
order to obtain a higher precision distance to a target as
indicated by a "precision flag" which may be displayed on the LCD
display 32 within the laser range finder 10 eyepiece. Nevertheless,
the actual values derived in these time expansions will, of course,
vary with time, temperature and aging and affects the gain of the
transistors, the leakages, as well as the value of the resistances
and capacitances. Initially the exact values of these effects are
completely unknown but, through the use of the zero and calibration
functions above-described, the zero problem has been eliminated,
and a crystal reference calibration has been provided for the
entire flight time without having to resort to a complicated
counter circuitry.
[0053] Another aspect of the precision timing section 34 is the
automatic set noise control and invertor 168 provides, in
conjunction with other circuit elements, a hardware hold off
function. Upon firing of the laser and receipt of the reference
signal REF on line 94 at the CLK input of flip-flop 158, a certain
time must elapse, as determined by the time constant of resistor
164 and capacitor 166, before the D input goes high. Until that
time, all noise pulses and/or early laser pulses on the clock line
are ignored. The purpose for this function is that, when the laser
fires, it generates unintended ground bounce and noise that may
prematurely trigger the receive flip-flop 162 rather than the real
laser return signal RX(OUT+). For that reason, a hold off period is
provided corresponding to the minimum range of the laser range
finder 10 and, as an example, considering a minimum range of about
twenty yards, the holdoff time is approximately 60 nanoseconds.
With a lower sensitivity laser range finder 10 utilized at shorter
ranges the function can be eliminated and it is clearly most useful
with a high sensitivity receiver where the noise from the firing
circuit determines an effective minimum range.
[0054] Transistor 174 provides an additional function and allows
the microcomputer 270 to extend the hold off range by asserting the
HOLD OFF signal on line 258. In this manner, the minimum range of
the laser range finder 10 may be extended out to, for example,
sixty or eighty yards, whatever is the desirable setting. This
microcomputer 270 hold off function may be implemented by the mode
switch 26 and would allow shooting through branches, twigs,
precipitation or other partial obstructions. By extending the hold
off range out beyond such partial obstructions, there is
insufficient back scatter from the obstructions to trigger the
precision timing section 34 and the measurement will be made to the
desired target instead of the intervening obstructions. This is
accomplished by not allowing flip-flop 162 to trigger until a set
timer period has elapsed. Transistor 174 is the switching device
utilized to allow setting of an extension to the hold off range and
gate 180 is used to determine the receive pulse width in
conjunction with the discharge rate of capacitor 194. This allows
the microcomputer 270, which has a built in analog-to-digital
("A/D") convertor, to determine the residual voltage on capacitor
194 and therefore derive a measure of the pulse width, (which is a
measure of the return signal power) and thus use an internal lookup
table to correct for that power variation and get a higher range
accuracy. When the logic reset signal /RESET on line 156 is low,
transistor 190 clamps capacitor 194 to the +5 volt rail. During the
laser measurement routine, the transistor 190 is turned off. When a
pulse subsequently arrives, that bit turns on transistor 200 and
the voltage in capacitor 194 will be discharged via resistor 198
for the duration of that pulse. The charge on capacitor 194 is then
digitized by the processor to determine the effect of incoming
power.
[0055] With reference additionally now to FIG. 8, the automatic
noise threshold section 36 of the laser range finder 10 is shown.
The automatic noise threshold section 36 receives the RX (OUT+)
signal from the laser receive section 22 (shown in FIG. 1) on line
100 for input thereto through resistor 314. Resistor 314 is
connected to the anode of diode 316 which has its cathode connected
to the "+" input of operational amplifier ("OpAmp") 318 forming a
V3 node 320. V3 node 320 is coupled to circuit ground through the
parallel combination of resistor 322 and capacitor 324. The output
of OpAmp 318 is coupled back to the "-" input thereof as well as to
line 1.02 for supplying the V.sub.threshold signal to the laser
receive section 22 (shown in FIG. 1). Line 102 is connected through
resistor 330 to the center tap of potentiometer 332 which has one
terminal thereof connected to a source of +5 volts through resistor
334 and another terminal thereof coupled to circuit ground through
resistor 336.
[0056] Lines 308 and 310 from the microcomputer 270 (shown in FIG.
6) are connected through resistors 338 and 334 respectively to line
102. Additionally, line 312 from microcomputer 270 is connected to
line 102 through resistor 342 as shown.
[0057] In operation, the automatic noise threshold section 36 in
conjunction with the CPU section 28 (shown in FIG. 6) provides a
simply implemented yet highly effective threshold adjustment to the
laser receive section 22 (shown in FIG. 3). As shown in FIG. 3, the
laser receiving diode 24 utilizes a high voltage source (of about
50 volts) supplied via a noise filtering network, comprising low
pass filter network 106, to bias it. The diode 24 responds with an
output current proportional to the incoming laser light which is
generally a short duration laser pulse producing a short current
pulse which is amplified by transistors 118,120, 122,124,
comprising the active circuit elements of a transimpedance
amplifier 116. The transimpedance amplifier 116 produces an output
voltage pulse proportional to the incoming laser pulse impinging on
the laser receiving diode 24. The output of the transimpedance
amplifier 116 is capacitively coupled to the "+" input of
comparator 134, which is a high speed comparator. When the laser
pulse input to the "+" input crosses a threshold determined by the
voltage on the "-" threshold pin, a positive output pulse is
produced.
[0058] To maximize performance, the threshold of the comparator 134
has to be set for maximum sensitivity in order to detect the
weakest possible laser pulse to get the maximum performance out of
the laser range finder 10. Conventional approaches include using
digital controls or a potentiometer to adjust the threshold.
However, these approaches have the down side that over time and
temperature changes the gain of the receiver will change with the
background noise generated by the background light rendering a
fixed threshold as less than an ideal solution.
[0059] The automatic noise threshold section 36 of FIG. 8 discloses
a circuit that automatically sets a threshold such that a constant
noise pulse firing rate is output from the detector comprising
resistor 314, diode 316, capacitor 324 and resistor 322. In
operation, when the threshold pin of the comparator 134 (FIG. 3) is
at a considerably higher voltage than the input pin, no noise
pulses will appear at the output due to the inherent amplifier and
optically generated noise. As the voltages on the threshold and
input pins are brought closer together, noise pulses will appear at
the output and, when the voltage level are nearly coincident, a
great deal of noise can be seen. In essence then, the automatic
noise threshold section 36 sets the noise pulse rate at that point
at which, given the right firmware algorithm, one can still acquire
the target and not be blinded by the noise. The higher the noise
that can be tolerated, and the closer the voltage levels at the
threshold and input pins of the comparator 134, the weaker the
laser pulse that can be detected. The automatic noise threshold
section 36 automatically adjusts that threshold level to maintain
constant noise pulse firing rate.
[0060] As shown in FIG. 8, this is accomplished by monitoring the
digital logic receive signal RX(OUT+) on line 100 that goes to the
receive flip-flop 162 (shown in FIG. 4). The detector monitors line
100 for the presence of noise pulses via a detector comprising the
aforementioned resistor 314, diode 316, capacitor 324 and resistor
322. The value of resistor 322 is typically considerably greater
than that of 314, on the order of a 150:1 ratio. The peak amplitude
of the noise pulses is typically at or near the logic threshold,
except for very narrow pulses where the comparator will not reach
full amplitude, however, the width of these pulses is going to vary
randomly because it depends on the noise signal that is being
detected. Moreover, the spacing of the noise pulses will also vary
at a random rate, but, for any given threshold setting, there will
be a fixed average rate. The average rate is dependent on the
threshold. Therefore, during the time the pulse is high, capacitor
324 charges via resistor 314 and diode 316 at a rate determined by
the high on the logic pulse, resistor 314 and whatever voltage is
still existing on capacitor 324.
[0061] Initially, capacitor 324 is charged as follows: once the
noise pulse terminates, the logic line goes back to zero. There is
a residual voltage on capacitor 324, diode 316 will be reverse
biased, and the discharge path is now via resistor 322. (As
previously described, the value for resistor 322 is chosen to
provide a relatively longer time constant, a factor of 150.) When
another pulse comes in, capacitor 324 will charge a bit more. What
will then happen is, quite rapidly, (i.e. within a few
milliseconds) the voltage across capacitor 324 stabilizes at a rate
that is proportional to the average firing rate. The reason for
having a large ratio between resistor 324 and resistor 322 is
because the noise pulses typically may average 50 nanoseconds wide,
and the averaged time between them to maximize the sensitivity of
the laser range finder 10 should be of the order of two
microseconds or so. As an example, if a 50% voltage were desired,
and the high state was occurring for 50 nanoseconds while the low
state average was occurring for one microsecond, a 20:1 ratio would
be produced. Nevertheless, the optimum ratio has been determined
empirically to be about 150:1 as previously described and is
related to average pulse widths (typically on the order of 30
nanoseconds in length) and pulse repetition rates (on the order of
4 microseconds) with a typical voltage level of 1.5 volts.
[0062] Op amp 318 is configured as a unity gain buffer, although it
need not be unity gain, with a voltage V3 at its "+" input pin on
node 320. The input is high impedance and the output is low
impedance in order to drive external circuitry. The voltage that is
derived at the output of the op amp 318 is then fed into a resistor
network comprising resistor 338, resistor 340, resistor 342 and
resistor 330. A summing node of the resistor network on line 102
goes to the threshold control to provide the signal V.sub.threshold
to the laser receive section 22 (shown in FIG. 3). Resistor 330 is
connected to the center tap of a potentiometer 332 so that the DC
voltage on the other end of resistor 330 can be controlled. It's
just a DC voltage.
[0063] In combination, the circuit comprises a feedback network
such that, if there are no noise pulses, then V3 is zero and
V.sub.threshold and drops to a low value. Initially,
V.sub.threshold will be higher, and the "-" input of comparator 134
(shown in FIG. 3) will be higher than the "+" input, forcing a
logic low on the output as the starting state. As the level of V3
on node 320 falls, the voltage level on the "-" pin of comparator
134 starts approaching the level of the signal from the
transimpedance amplifier 116 on the positive "+". When it
approaches the noise zone, noise pulses start appearing. As soon as
noise pulses start appearing, a charge appears on node 320, so V3
stops to charge up, and when the two match, that's the feedback
point, and it stops. Basically, the voltage on the threshold is set
at such a point that the noise firing rate maintains V3 at that
voltage which is necessary to maintain V.sub.threshold. Because
very small changes in V.sub.threshold make a very large change in
the noise firing rate, typically, a ten millivolt change in
V.sub.threshold will change the voltage V3 at node 320 by about a
volt. What is produced then, is a fairly high gain feedback loop,
such that Vthreshold will track very closely the noise firing rate
and V3 will stabilize very accurately and rapidly. This further
provide the capability to adjust the noise firing rate by
controlling the bias and forcing V3 to compensate. The voltage V3
at node 320 then represents the noise firing rate.
[0064] NSET1 line 308 and NSET2 line 310, are two control lines
from the microcomputer 28 such that when held low or high, adjusts
the noise rate to obtain the maximum range to different
reflectivity targets. If both lines 308 and 310 are taken high, V3
will drop to compensate to maintain a constant threshold noise.
Similarly, potentiometer 332 provides an adjustment such that the
threshold point may be set together with the level of V3.
Typically, the V3 point might be set equal to: 0.5, 1.0, 1.5 and
2.0 volts as desirable choices for the average noise firing rates.
As such, since resistor 338 is approximately twice the value of
resistor 340, four voltage combinations are obtained roughly
equally spaced in voltage by half a volt. Potentiometer 332 is used
to set the first voltage level to 0.5 or the last one to 2.0 while
the intervals are determined by the logic control lines 308 and 310
set NSET1 arid NSET2. Obviously, this approach could be extended,
four combinations provides adequate resolution in the particular
implementation of the laser range finder 10 described and shown.
When both lines 308 and 310 are high, there is a current injected
into the node comprising the V.sub.threshoId line 102, and to
compensate for that, V3 must drop, so less current flows through
resistor 326 and vice versa. V3 will follow these values, depending
on the permutations of logic high and low signals on the lines 308
and 310. Resistor 330 in used just to set where this whole block
resides while potentiometer 332 is used to establish the initial
set point. Since the noise characteristics from unit to unit will
vary somewhat, potentiometer 332 enables the setting of the initial
device characteristics.
[0065] Resistor 342 is of a considerably lower value than resistors
338 and 340 and its value is chosen such that, when the REFLECTOR
MODE signal on line 312 is asserted by being taken high, V3 will
drop to zero and will stay there because it cannot go below zero.
At this point, the feedback loop is saturated and is no longer
effective, so V.sub.threshold is no longer is stabilized. In
operation, line 312 will be pulled high by a considerable voltage,
on the order of 0.4 volts, such that it completely desensitizes the
laser receive section 22 so the laser range finder 10 will then
only respond to a retro reflector. In this mode of operation the
receiver is detuned and its noncooperative range drops from 500
yards down to about 30 or 40 yards, such that the laser range
finder 10 only latch onto a retro reflector or survey prism
comprising a high grade reflector that returns the laser energy
back to the source. Possible applications also include determining
the distance to a particular golf hole where a laser reflector is
attached to the pin and the signal might otherwise be actually
returned from trees behind or in front of the green in a more
sensitive mode of operation.
[0066] The essence of the automatic noise threshold section 36 is,
as previously described, a feedback loop comprising the detected
average noise firing rate forming a feedback loop that controls the
threshold. Use of this circuit has resulted in an addition of
almost 50% to the range of the laser range finder 10 compared to
attempting to manually act the threshold. By setting the noise
firing rate, noise pulses are being produced deliberately, all the
time and the only way for you to take advantage of that fact is by
implementing a firmware algorithm in the microcomputer 270 that
allows you to discriminate between noise pulses and laser return
pulses. What the algorithm does is, during the laser firing
process, on the first pulse that fires, it gets a laser pulse, and
it places it in a stack of pulses. For example, the stack may have
locations designated 0 through 9, to enable 10 pulses to be
maintained in the stack. The values of the FLIGHT.sub.TIME are
saved, corrected for power return, (the microcomputer 270
determines the power level of the return signal and corrects the
flight time for power return) and placed in one of the locations in
the stack. Upon receipt of the next pulse, the microcomputer 270
will then compare the next pulse with the remaining locations in
the stack. Initially, most of the locations will be empty, and
there will be no match. If no match is found, the microcomputer 270
puts the pulse in the stack and carries on, merely placing pulses
in the stack, and then when it gets to the top, it goes back and
overwrites the base, so you have a history of N number of pulses in
the stack. Any time a new pulse comes in, it compares the entire
stack for a match, where N=10, it searches the preceding ten pulses
for a match.
[0067] The reason for doing that is, since a high noise firing rate
has been deliberately set to get maximum sensitivity, many noise
pulses are going to have shown up, but the noise pulses will be of
random occurrence and the chance of a precision match is very low.
Because the tolerance can be set as any other firmware parameter, a
default value will be typically loaded that has been determined
empirically. As an example, a tolerance of a few nanoseconds may be
set for a match to be assumed to be a real target and not a noise
pulse. Utilizing the algorithm, the process continues, trying to
lock on the target until a match is achieved. The match need only
be two pulses within the preset tolerance (providing very
acceptable results) or, if higher sensitivity were desired, a match
of three through N may be specified, depending on the reliability
needed to guarantee a real target and not a noise pulse. In an
exemplary operation, the first pulse (pulse 0) could be the real
target, followed by eight noise pulses, and as long as the ninth
pulse is again the real target, the distance to the target can, be
accurately determined. The stack can be increased in size up to
whatever memory limit is available in the system, depending on how
far into the noise level the laser range finder 10 must work.
[0068] Having found a match, the average of the match values may
then be used to compare all subsequent pulses, rather than needing
to place them in a stack and only pulses that match up with that
initial match average will contribute to the measurement. If a
certain number of pulses elapse before another matching pulse is
received, it may be assumed that an accidental lock-on to noise has
been achieved and the process restart. By adjusting the various
parameters, a trade off can be made between the time it takes to
get a measurement to how far into the noise the laser range finder
to must work. Because the noise rate can set to whatever is desired
by means of the automatic noise threshold section 36, it is
possible to optimize the algorithm to provide the optimum
acquisition characteristics against time and against range.
[0069] The higher the value of V3, the more noise is coming out of
the receiver, and the more sensitive the laser receive section 22
is running. The probability of a noise pulse showing up is
proportional to the flight time, so given a very "black" target,
the maximum range will be less, but the maximum flight time is also
less, so a higher noise rate can be tolerated. Therefore, running
at a higher gain will provide the best range to a black target. On
the other hand, if the target's very reflective, a high gain is not
required, so the noise rate can be lowered, which then provides the
same probability of a noise pulse appearing over a longer flight
range, and therefore a quick acquisition on a bright white target
can be achieved. Thus, by depressing the mode switch 26, different
modes of operation of the laser range finder 10 can be selected. As
an example, one mode might be utilized to find the range to
reflective road signs out to a distance of 1000 yards or more.
Alternatively, aiming the laser range finder 10 at something like
wet black tree bark, might reduce the maximum range to only 350-400
yards and so a different operational mode might be selected which
would otherwise require a relatively long time to hit the road
sign, if ever, because there would always be a noise pulse in the
way. The mode switch 26 allows the setting of these variables to
maximize the range of the laser range finder 10, depending on the
target quality and a visual indication of the target quality
selected may be provided to the operator on the insight, LCD
display 32 wherein the first mode would correspond to the brightest
target or most reflective target, and the Nth mode would correspond
to the least reflective target.
[0070] While there have been described above, the principles of the
invention in conjunction with specific apparatus, it is to be
clearly understood that the foregoing description is made only by
way of example and not as a limitation on the scope of the
invention.
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