U.S. patent number RE48,563 [Application Number 16/575,017] was granted by the patent office on 2021-05-18 for method for transmitting data by using polar coding in wireless access system.
This patent grant is currently assigned to LG Electronics Inc.. The grantee listed for this patent is LG Electronics Inc.. Invention is credited to Bonghoe Kim, Dongyoun Seo.
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United States Patent |
RE48,563 |
Kim , et al. |
May 18, 2021 |
Method for transmitting data by using polar coding in wireless
access system
Abstract
The present invention relates to data transmission/reception
methods using a polar coding scheme, and devices for supporting
same. The method for transmitting data by using polar coding in a
wireless access system, according to one embodiment of the present
invention, may comprise the steps of deriving Bhattacharyya
parameters according to data bits input for finding noise-free
channels among equivalent channels; allocating data payloads
comprising data bits and cyclic redundancy check (CRC) bits to the
found noise-free channels; inputting the data payloads into a polar
encoder; and transmitting code bits output by the polar encoder,
wherein the CRC bits may be allocated to better noise-free
channels, among the noise-free channels indicated by the
Bhattacharyya parameters, than the data bits.
Inventors: |
Kim; Bonghoe (Seoul,
KR), Seo; Dongyoun (Seoul, KR) |
Applicant: |
Name |
City |
State |
Country |
Type |
LG Electronics Inc. |
Seoul |
N/A |
KR |
|
|
Assignee: |
LG Electronics Inc. (Seoul,
KR)
|
Family
ID: |
52483870 |
Appl.
No.: |
16/575,017 |
Filed: |
September 18, 2019 |
PCT
Filed: |
August 20, 2014 |
PCT No.: |
PCT/KR2014/007716 |
371(c)(1),(2),(4) Date: |
February 03, 2016 |
PCT
Pub. No.: |
WO2015/026148 |
PCT
Pub. Date: |
February 26, 2015 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
|
61867606 |
Aug 20, 2013 |
|
|
|
|
61941456 |
Feb 18, 2014 |
|
|
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Reissue of: |
14909901 |
Aug 20, 2014 |
9768915 |
Sep 19, 2017 |
|
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H03M
13/6356 (20130101); H03M 13/13 (20130101); H04L
1/00 (20130101); H04L 1/1861 (20130101); H03M
13/356 (20130101); H04L 1/1671 (20130101); H03M
13/2906 (20130101); H03M 13/6362 (20130101); H04L
1/0061 (20130101); H03M 13/09 (20130101); H03M
13/6525 (20130101); H04L 1/0068 (20130101); H03M
13/353 (20130101); H04L 1/0041 (20130101) |
Current International
Class: |
H03M
13/00 (20060101); H03M 13/35 (20060101); H03M
13/13 (20060101); H03M 13/09 (20060101); H04L
1/00 (20060101); H04L 1/18 (20060101); H04L
1/16 (20060101); H03M 13/29 (20060101) |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Erdal Arikan; "Channel Polarization: A Method for Constructing
Capacity-Achieving Codes for Symmetric Binary-Input Memoryless
Channels"; Jul. 2009; IEEE Transactions on Information Theory; vol.
55, No. 7; pp. 3051-3073. cited by examiner .
Erdal Arikan; "Channel Polarization: A Method for Constructing
Capacity-Achieving Codes"; Jul. 6-11, 2008; ISIT 2008, Toronto,
Canada; pp. 1173-1177. cited by examiner .
Chen et al.; "Improved Successive Cancellation Decoding of Polar
Codes"; Jan. 17, 2013; arXiv:1208.3598v2; pp. 1-9. cited by
examiner .
Bonik et al.; "A variant of list plus CRC concatenated polar code";
Jul. 19, 2012; arXiv:1207.4661v1; 4 pages. cited by examiner .
Tal et al.; "List Decoding of Polar Codes"; May 31, 2012;
arXiv:1206.0050v1; pp. 1-11. cited by examiner .
Tal et al.; "How to Construct Polar Codes"; Apr. 10, 2013;
arXiv:1105.6164v3; pp. 1-21. cited by examiner .
Kai Niu et al.: "Beyond Turbo Codes: Rate-Compatible Punctured
Polar Codes", 2013 IEEE International Conference on Communication
(ICC) Budapest Hungary, Jun. 9-13, 2013, see pp. 3423-3427; and
figure 1. cited by applicant .
Peyman Hesami, "Channel Polarization and Polar Codes: Capacity
Achieving", Tutorial of Information Theory Course, University of
Notre Dame, Dec. 9, 2009, see pp. 1-9. cited by applicant .
Kai Niu et al.: "CRC-Aided Decoding of Polar Codes", IEEE
Communications Letters, vol. 16, No. 10, Oct. 2012, see pp.
1668-1671. cited by applicant .
Bin Li et al., "An Adaptive Successive Cancellation List Decoder
for Polar Codes with Cyclic Redundancy Check", IEEE Communications
Letters, vol. 16, issue Dec. 12, 2012, see pp. 1-4. cited by
applicant .
Written Opinion of the ISA from PCT/KR2014/007716, dated Nov. 28,
2014. cited by applicant.
|
Primary Examiner: Wood; William H.
Attorney, Agent or Firm: Dentons US LLP
Parent Case Text
This application is .Iadd.a reissue of U.S. Pat. No. 9,768,915,
which issued Sep. 19, 2017 from U.S. patent application Ser. No.
14/909,901, filed Feb. 3, 2016, which is .Iaddend.a 35 USC
.sctn.371 National Stage entry of International Application No.
PCT/KR2014/007716 filed on Aug. 20, 2014, and claims priority to
U.S. Provisional Application Nos. 61/867,606 filed on Aug. 20, 2013
and 61/941,456 filed on Feb. 18, 2014, all of which are hereby
incorporated by reference in their entireties as if fully set forth
herein.
Claims
The invention claimed is:
1. A method for transmitting data by using polar coding in a
wireless access system, the method comprising: .[.inputting first
data bits to a polar encoder; deriving Bhattacharyya parameters in
accordance with the first data bits to discover noise free channels
from equivalent channels, wherein the equivalent channels are
defined for each of the first data bits;.]. allocating a data
payload including .[.second.]. data bits and cyclic redundancy
check (CRC) bits to .[.the discovered.]. noise free channels
.Iadd.among equivalent channels of a polar encoder.Iaddend.,
wherein the number of the .[.discovered.]. noise free channels
corresponds to a total number of bits of the data payload;
.Iadd.and .Iaddend. .[.performing rate matching for the data
payload based on a size of the polar encoder and a size of the data
payload; and.]. transmitting code bits output from the polar
encoder.[.through the noise free channels.]., wherein the CRC bits
are allocated to noise free channels .[.starting from a noise free
channel.]. having .[.a.]. lowest error probability among the
.[.discovered.]. noise free channels and the .[.second.]. data bits
are allocated to the other noise free channels among the
.[.discovered.]. noise free channels.[.according to the
Bhattacharyya parameters.]..
2. The method according to claim 1, further comprising:
transmitting .[.size.]. information indicating .[.the.]. .Iadd.a
.Iaddend.size of the data payload and information .[.on.].
.Iadd.regarding .Iaddend.a coding rate of the polar encoder to a
receiver.
3. The method according to claim 1.[., wherein the.]. .Iadd.further
comprising: performing .Iaddend.rate matching .[.is performed.].
.Iadd.for the data payload .Iaddend.by puncturing or repeating the
data payload .[.in accordance with.]. .Iadd.based on .Iaddend.the
size of the polar encoder.
4. The method according to claim 3, further comprising: generating
a generator matrix for the polar encoder by puncturing one or more
specific columns of a mother generator matrix based on .[.the.].
.Iadd.a .Iaddend.size of the data payload when the size of .[.the
code block of.]. the polar encoder is greater than the size of the
data payload when performing .Iadd.the .Iaddend.rate matching.
5. The method according to claim 4, wherein the one or more
specific columns are selected starting from a column having a
lowest weight value.
6. The method according to claim 4, wherein the one or more
specific columns are selected considering a minimum distance set
based on an index of a noise free channel to which the data bits
are allocated.
7. The method according to claim 4, wherein the one or more
specific columns are selected based on a priority according to a
column index.
8. The method according to claim 3, further comprising: generating
a generator matrix for the polar encoder by repeating one or more
specific columns of a mother generator matrix based on .[.the.].
.Iadd.a .Iaddend.size of the data payload when the size of .[.the
code block of.]. the polar encoder is smaller than the size of the
data payload when performing .Iadd.the .Iaddend.rate matching.
9. The method according to claim 8, wherein the one or more
specific columns are selected starting from a column having a
highest weight value.
.Iadd.10. A method for transmitting data by using polar coding at a
transmitting device in a wireless communication system, the method
comprising: allocating a payload of K bits to K predetermined input
bits among N input bits for a polar code of size N, where K is a
positive integer not greater than N; generating encoded bits based
on the N input bits and the polar code; and transmitting the
encoded bits to a receiving device, wherein the payload includes L
data bits and M cyclic redundancy check (CRC) bits, where L+M=K,
and L and M are positive integers, and wherein the M CRC bits are
allocated to M most reliable input bits among the K predetermined
input bits..Iaddend.
.Iadd.11. The method according to claim 10, further comprising:
allocating 0 to each of N-K input bits other than the K
predetermined input bits among the N input bits for the polar code,
wherein the K predetermined input bits have higher reliability than
the N-K input bits..Iaddend.
.Iadd.12. The method according to claim 10, further comprising:
transmitting information regarding K and information regarding a
code rate for the payload..Iaddend.
.Iadd.13. A transmitting device for transmitting data by using
polar coding in a wireless communication system, the transmitting
device comprising: a transmitter; at least one processor; and at
least one computer memory that is operably connectable to the at
least one processor and that has stored thereon instructions which,
when executed, cause the at least one processor to perform
operations comprising: allocating a payload of K bits to K
predetermined input bits among N input bits for a polar code of
size N, where K is a positive integer not greater than N;
generating encoded bits based on the N input bits and the polar
code; and transmitting, via the transmitter, the encoded bits to a
receiving device, wherein the payload includes L data bits and M
cyclic redundancy check (CRC) bits, where L+M=K, and L and M are
positive integers, and wherein the M CRC bits are allocated to M
most reliable input bits among the K predetermined input
bits..Iaddend.
.Iadd.14. The transmitting device according to claim 13, wherein
the operations further comprising: allocating 0 to each of N-K
input bits other than the K predetermined input bits among the N
input bits for the polar code, wherein the K predetermined input
bits have higher reliability than the N-K input bits..Iaddend.
.Iadd.15. The transmitting device according to claim 13, further
comprising: transmitting, via the transmitter, information
regarding K and information regarding a code rate for the
payload..Iaddend.
.Iadd.16. A receiving device for receiving data by using polar
coding in a wireless communication system, the receiving device
comprising: a receiver; at least one processor; and at least one
computer memory that is operably connectable to the at least one
processor and that has stored thereon instructions which, when
executed, cause the at least one processor to perform operations
comprising: receiving, via the receiver, encoded bits from a
transmitting device; and decoding the encoded bits to obtain a
payload of K bits based on a polar code of size N, wherein the
encoded bits are decoded based on a mapping relationship between
the payload of K bits and K predetermined input bits among N input
bits for the polar code, where K is a positive integer not greater
than N; wherein the payload includes L data bits and M cyclic
redundancy check (CRC) bits, where L+M=K, and L and M are positive
integers, and wherein the mapping relationship comprises: mapping
the M CRC bits to M most reliable input bits among the K
predetermined input bits..Iaddend.
.Iadd.17. The receiving device according to claim 16, wherein the
mapping relationship further comprises: mapping 0 to each of N-K
input bits other than the K predetermined input bits among the N
input bits for the polar code, wherein the K predetermined input
bits have higher reliability than the N-K input bits..Iaddend.
.Iadd.18. The receiving device according to claim 16, wherein the
operations further comprise: receiving, via the receiver,
information regarding K and information regarding a code rate for
the payload..Iaddend.
Description
TECHNICAL FIELD
The present invention relates to a wireless access system, and more
particularly, to a method for applying a polar coding scheme to a
wireless access system and a device for supporting the same. That
is, the present invention relates to a method for transmitting and
receiving data using a polar coding scheme and a device for
supporting the same.
BACKGROUND ART
Wireless access systems have been widely deployed to provide
various types of communication services such as voice or data. In
general, a wireless access system is a multiple access system that
supports communication of multiple users by sharing available
system resources (a bandwidth, transmission power, etc.) among
them. For example, multiple access systems include a Code Division
Multiple Access (CDMA) system, a Frequency Division Multiple Access
(FDMA) system, a Time Division Multiple Access (TDMA) system, an
Orthogonal Frequency Division Multiple Access (OFDMA) system, and a
Single Carrier Frequency Division Multiple Access (SC-FDMA)
system.
DISCLOSURE
Technical Problem
An object of the present invention is to provide methods for
efficiently transmitting and receiving data.
Another object of the present invention is to provide methods for
coding input data bits using a polar coding scheme for efficient
data transmission and reception.
Still another object of the present invention is to provide rate
matching methods for coding bits coded using a polar coding
scheme.
Further still another object of the present invention is to provide
various methods for configuring generator matrixes of a polar
encoder for applying a polar coding scheme.
Further still another object of the present invention is to provide
methods for applying HARQ when a polar coding scheme is
applied.
Further still another object of the present invention is to provide
devices for supporting the aforementioned methods.
Additional advantages, objects, and features of the invention will
be set forth in part in the description which follows and in part
will become apparent to those having ordinary skill in the art upon
examination of the following or may be learned from practice of the
invention. The objectives and other advantages of the invention may
be realized and attained by the structure particularly pointed out
in the written description and claims hereof as well as the
appended drawings.
Technical Solution
The present invention relates to a method for applying a polar
coding scheme to a wireless access system and a device for
supporting the same. That is, the present invention discloses a
method for transmitting and receiving data using a polar coding
scheme and devices for supporting the same.
In one aspect of the present invention, a method for transmitting
data by using polar coding in a wireless access system comprises
the steps of deriving Bhattacharyya parameters in accordance with
data bits input to discover noise free channels from equivalent
channels; allocating a data payload including data bits and cyclic
redundancy check (CRC) bits to the discovered noise free channels;
inputting the data payload to a polar encoder; and transmitting
code bits output from the polar encoder. At this time, the CRC bits
can be allocated to a noise free channel better than the data bits,
among the noise free channels indicated by the Bhattacharyya
parameters.
The method may further comprise the step of performing rate
matching for the data payload considering a size of the data
payload and a size of a code block of the polar encoder.
At this time, the method may further comprise the step of
generating a generator matrix used for the polar encoder by
puncturing a specific column of a mother generator matrix as much
as a size difference if the size of the code block of the polar
encoder is greater than the size of the data payload in the step of
performing rate matching.
In this case, the specific column may be selected as much as the
size difference from the column having the lowest weight value, and
the weight value may be set to the number of `1` among elements of
the mother generator matrix.
Otherwise, the specific column may be selected considering a
minimum distance set based on an index of a noise free channel to
which the data bits are allocated.
Otherwise, the specific column may be selected considering a
priority according to a column index.
The method may further comprise the step of generating a generator
matrix used for the polar encoder by repeating a specific column of
a mother generator matrix as much as a size difference if the size
of the code block of the polar encoder is smaller than the size of
the data payload in the step of performing rate matching.
At this time, the specific column may be selected as much as the
size difference from the column having the highest weight value,
and the weight value may be set to the number of `1` among elements
of the mother generator matrix.
The method may further comprise the step of transmitting size
information indicating the size of the data payload and information
on a coding rate of a code block of the polar encoder to a
receiver.
It is to be understood that both the foregoing general description
and the foll owing detailed description of the present invention
are exemplary and explanatory and are i ntended to provide further
explanation of the invention as claimed
Advantageous Effects
As is apparent from the above description, the embodiments of the
present invention have the following effects.
First of all, a polar coding scheme is applied to a wireless access
system, whereby data can be transmitted and received
efficiently.
Secondly, if a polar coding scheme is used, a generator matrix is
configured considering a weight value, a minimum distance and/or a
priority, whereby data matching can be performed in accordance with
the number of bits of a data payload.
Thirdly, new methods for performing HARQ are suggested during
application of a polar coding scheme.
It will be apparent to those skilled in the art that various
modifications and variations can be made in the present invention
without departing from the spirit or scope of the inventions. Thus,
it is intended that the present invention covers the modifications
and variations of this invention provided they come within the
scope of the appended claims and their equivalents.
BRIEF DESCRIPTION OF THE DRAWINGS
The accompanying drawings, which are included to provide a further
understanding of the invention, illustrate embodiments of the
invention and together with the description serve to explain the
principle of the invention.
FIG. 1 is a conceptual diagram illustrating physical channels used
in the embodiments and a signal transmission method using the
physical channels.
FIG. 2 is a diagram illustrating a structure of a radio frame for
use in the embodiments.
FIG. 3 is a diagram illustrating an example of a resource grid of a
downlink slot according to the embodiments.
FIG. 4 is a diagram illustrating a structure of an uplink subframe
according to the embodiments.
FIG. 5 is a diagram illustrating a structure of a downlink subframe
according to the embodiments.
FIG. 6 is a diagram illustrating an example of a component carrier
(CC) used in the embodiments of the present invention and carrier
aggregation used in an LTE-A system.
FIG. 7 is a diagram illustrating a subframe frame structure of an
LTE-A system according to cross carrier scheduling used in the
embodiments of the present invention;
FIG. 8 is a diagram illustrating physical mapping of a PUCCH format
into PUCCH RBs.
FIG. 9 is a diagram illustrating PUCCH formats 2/2a/2b in case of a
normal cyclic prefix.
FIG. 10 is a diagram illustrating PUCCH formats 2/2a/2b in case of
an extended cyclic prefix.
FIG. 11 is a diagram illustrating PUCCH formats 1a/1b in case of a
normal cyclic prefix.
FIG. 12 is a diagram illustrating PUCCH formats 1a/1b in case of an
extended cyclic prefix.
FIG. 13 is a diagram illustrating one of constellation mapping of
HARQ ACK/NACK for a normal CP.
FIG. 14 is a diagram illustrating joint coding performed for HARQ
ACK/NACK and CQI for an extended CP.
FIG. 15 is a diagram illustrating one of methods for multiplexing
SR and ACK/NACK signal.
FIG. 16 is a diagram illustrating constellation mapping of ACK/NACK
and SR for PUCCH formats 1/1a/1b.
FIG. 17 is a diagram illustrating methods of matching control
information with a physical resource region.
FIG. 18 is a diagram illustrating an example of a coding method
based on a dual RM scheme.
FIG. 19 is a diagram illustrating a method for interleaving output
code bits when dual RM described in FIG. 18 is used.
FIG. 20 is a diagram illustrating a first level channel combining
procedure performed by polar coding.
FIG. 21 is a diagram illustrating an Nth level channel combining
procedure performed by polar coding, wherein a size N of a code
block has a limit of 2N (n is a natural number).
FIG. 22 is a diagram illustrating an example of a procedure of
transmitting data from a transmitter through polar coding.
FIG. 23 is a diagram illustrating a device through which methods
described in FIG. 1 to FIG. 22 can be embodied.
BEST MODE FOR CARRYING OUT THE INVENTION
The embodiments of the present invention described in detail
hereinafter provides methods for transmitting CSI in a wireless
access system that supports a multi-connection mode in which a user
equipment is connected with two or more small cells and devices for
supporting the same.
The embodiments of the present disclosure described below are
combinations of elements and features of the present disclosure in
specific forms. The elements or features may be considered
selective unless otherwise mentioned. Each element or feature may
be practiced without being combined with other elements or
features. Further, an embodiment of the present disclosure may be
constructed by combining parts of the elements and/or features.
Operation orders described in embodiments of the present disclosure
may be rearranged. Some constructions or elements of any one
embodiment may be included in another embodiment and may be
replaced with corresponding constructions or features of another
embodiment.
In the description of the attached drawings, a detailed description
of known procedures or steps of the present disclosure will be
avoided lest it should obscure the subject matter of the present
disclosure. In addition, procedures or steps that could be
understood to those skilled in the art will not be described
either.
Throughout the specification, when a certain portion "includes" or
"comprises" a certain component, this indicates that other
components are not excluded and may be further included unless
otherwise noted. The terms "unit", "-or/er" and "module" described
in the specification indicate a unit for processing at least one
function or operation, which may be implemented by hardware,
software or a combination thereof. In addition, the terms "a or
an", "one", "the" etc. may include a singular representation and a
plural representation in the context of the present invention (more
particularly, in the context of the following claims) unless
indicated otherwise in the specification or unless context clearly
indicates otherwise.
In the embodiments of the present disclosure, a description is
mainly made of a data transmission and reception relationship
between a Base Station (BS) and a User Equipment (UE). A BS refers
to a terminal node of a network, which directly communicates with a
UE. A specific operation described as being performed by the BS may
be performed by an upper node of the BS.
Namely, it is apparent that, in a network comprised of a plurality
of network nodes including a BS, various operations performed for
communication with a UE may be performed by the BS, or network
nodes other than the BS. The term `BS` may be replaced with a fixed
station, a Node B, an evolved Node B (eNode B or eNB), an Advanced
Base Station (ABS), an access point, etc.
In the embodiments of the present disclosure, the term terminal may
be replaced with a UE, a Mobile Station (MS), a Subscriber Station
(SS), a Mobile Subscriber Station (MSS), a mobile terminal, an
Advanced Mobile Station (AMS), etc.
A transmitter is a fixed and/or mobile node that provides a data
service or a voice service and a receiver is a fixed and/or mobile
node that receives a data service or a voice service. Therefore, a
UE may serve as a transmitter and a BS may serve as a receiver, on
an UpLink (UL). Likewise, the UE may serve as a receiver and the BS
may serve as a transmitter, on a DownLink (DL).
The embodiments of the present disclosure may be supported by
standard specifications disclosed for at least one of wireless
access systems including an Institute of Electrical and Electronics
Engineers (IEEE) 802.xx system, a 3.sup.rd Generation Partnership
Project (3GPP) system, a 3GPP Long Term Evolution (LTE) system, and
a 3GPP2 system. In particular, the embodiments of the present
disclosure may be supported by the standard specifications, 3GPP TS
36.211, 3GPP TS 36.212, 3GPP TS 36.213, 3GPP TS 36.321 and 3GPP TS
36.331. That is, the steps or parts, which are not described to
clearly reveal the technical idea of the present disclosure, in the
embodiments of the present disclosure may be explained by the above
standard specifications. All terms used in the embodiments of the
present disclosure may be explained by the standard
specifications.
Reference will now be made in detail to the embodiments of the
present disclosure with reference to the accompanying drawings. The
detailed description, which will be given below with reference to
the accompanying drawings, is intended to explain exemplary
embodiments of the present disclosure, rather than to show the only
embodiments that can be implemented according to the invention.
The following detailed description includes specific terms in order
to provide a thorough understanding of the present disclosure.
However, it will be apparent to those skilled in the art that the
specific terms may be replaced with other terms without departing
the technical spirit and scope of the present disclosure.
Hereinafter, 3GPP LTE/LTE-A systems which are examples of a
wireless access system which can be applied to embodiments to the
present invention will be explained.
The embodiments of the present disclosure can be applied to various
wireless access systems such as Code Division Multiple Access
(CDMA), Frequency Division Multiple Access (FDMA), Time Division
Multiple Access (TDMA), Orthogonal Frequency Division Multiple
Access (OFDMA), Single Carrier Frequency Division Multiple Access
(SC-FDMA), etc.
CDMA may be implemented as a radio technology such as Universal
Terrestrial Radio Access (UTRA) or CDMA2000. TDMA may be
implemented as a radio technology such as Global System for Mobile
communications (GSM)/General packet Radio Service (GPRS)/Enhanced
Data Rates for GSM Evolution (EDGE). OFDMA may be implemented as a
radio technology such as IEEE 802.11 (Wi-Fi), IEEE 802.16 (WiMAX),
IEEE 802.20, Evolved UTRA (E-UTRA), etc.
UTRA is a part of Universal Mobile Telecommunications System
(UMTS). 3GPP LTE is a part of Evolved UMTS (E-UMTS) using E-UTRA,
adopting OFDMA for DL and SC-FDMA for UL. LTE-Advanced (LTE-A) is
an evolution of 3GPP LTE. While the embodiments of the present
disclosure are described in the context of a 3GPP LTE/LTE-A system
in order to clarify the technical features of the present
disclosure, the present disclosure is also applicable to an IEEE
802.16e/m system, etc.
1. 3GPP LTE/LTE-A System
In a wireless access system, a UE receives information from an eNB
on a DL and transmits information to the eNB on a UL. The
information transmitted and received between the UE and the eNB
includes general data information and various types of control
information. There are many physical channels according to the
types/usages of information transmitted and received between the
eNB and the UE.
1.1 System Overview
FIG. 1 illustrates physical channels and a general method using the
physical channels, which may be used in embodiments of the present
disclosure.
When a UE is powered on or enters a new cell, the UE performs
initial cell search (S11). The initial cell search involves
acquisition of synchronization to an eNB. Specifically, the UE
synchronizes its timing to the eNB and acquires information such as
a cell Identifier (ID) by receiving a Primary Synchronization
Channel (P-SCH) and a Secondary Synchronization Channel (S-SCH)
from the eNB.
Then the UE may acquire information broadcast in the cell by
receiving a Physical Broadcast Channel (PBCH) from the eNB.
During the initial cell search, the UE may monitor a DL channel
state by receiving a Downlink Reference Signal (DL RS).
After the initial cell search, the UE may acquire more detailed
system information by receiving a Physical Downlink Control Channel
(PDCCH) and receiving a Physical Downlink Shared Channel (PDSCH)
based on information of the PDCCH (S12).
To complete connection to the eNB, the UE may perform a random
access procedure with the eNB (S13 to S16). In the random access
procedure, the UE may transmit a preamble on a Physical Random
Access Channel (PRACH) (S13) and may receive a PDCCH and a PDSCH
associated with the PDCCH (S14). In the case of contention-based
random access, the UE may additionally perform a contention
resolution procedure including transmission of an additional PRACH
(S15) and reception of a PDCCH signal and a PDSCH signal
corresponding to the PDCCH signal (S16).
After the above procedure, the UE may receive a PDCCH and/or a
PDSCH from the eNB (S17) and transmit a Physical Uplink Shared
Channel (PUSCH) and/or a Physical Uplink Control Channel (PUCCH) to
the eNB (S18), in a general UL/DL signal transmission
procedure.
Control information that the UE transmits to the eNB is generically
called Uplink Control Information (UCI). The UCI includes a Hybrid
Automatic Repeat and reQuest Acknowledgement/Negative
Acknowledgement (HARQ-ACK/NACK), a Scheduling Request (SR), a
Channel Quality Indicator (CQI), a Precoding Matrix Index (PMI), a
Rank Indicator (RI), etc.
In the LTE system, UCI is generally transmitted on a PUCCH
periodically. However, if control information and traffic data
should be transmitted simultaneously, the control information and
traffic data may be transmitted on a PUSCH. In addition, the UCI
may be transmitted aperiodically on the PUSCH, upon receipt of a
request/command from a network.
FIG. 2 illustrates exemplary radio frame structures used in
embodiments of the present disclosure.
FIG. 2(a) illustrates frame structure type 1. Frame structure type
1 is applicable to both a full Frequency Division Duplex (FDD)
system and a half FDD system.
One radio frame is 10 ms (Tf=307200Ts) long, including equal-sized
20 slots indexed from 0 to 19. Each slot is 0.5 ms (Tslot=15360Ts)
long. One subframe includes two successive slots. An ith subframe
includes 2ith and (2i+1)th slots. That is, a radio frame includes
10 subframes. A time required for transmitting one subframe is
defined as a Transmission Time Interval (TTI). Ts is a sampling
time given as Ts=1/(15 kHzx2048)=3.2552.times.10-8 (about 33 ns).
One slot includes a plurality of Orthogonal Frequency Division
Multiplexing (OFDM) symbols or SC-FDMA symbols in the time domain
by a plurality of Resource Blocks (RBs) in the frequency
domain.
A slot includes a plurality of OFDM symbols in the time domain.
Since OFDMA is adopted for DL in the 3GPP LTE system, one OFDM
symbol represents one symbol period. An OFDM symbol may be called
an SC-FDMA symbol or symbol period. An RB is a resource allocation
unit including a plurality of contiguous subcarriers in one
slot.
In a full FDD system, each of 10 subframes may be used
simultaneously for DL transmission and UL transmission during a
10-ms duration. The DL transmission and the UL transmission are
distinguished by frequency. On the other hand, a UE cannot perform
transmission and reception simultaneously in a half FDD system.
The above radio frame structure is purely exemplary. Thus, the
number of subframes in a radio frame, the number of slots in a
subframe, and the number of OFDM symbols in a slot may be
changed.
FIG. 2(b) illustrates frame structure type 2. Frame structure type
2 is applied to a Time Division Duplex (TDD) system. One radio
frame is 10 ms (Tf=307200Ts) long, including two half-frames each
having a length of 5 ms (=153600Ts) long. Each half-frame includes
five subframes each being 1ms (=30720Ts) long. An ith subframe
includes 2ith and (2i+1)th slots each having a length of 0.5 ms
(Tslot=15360Ts). Ts is a sampling time given as Ts=1/(15
kHzx2048)=3.2552.times.10-8 (about 33 ns).
A type-2 frame includes a special subframe having three fields,
Downlink Pilot Time Slot (DwPTS), Guard Period (GP), and Uplink
Pilot Time Slot (UpPTS). The DwPTS is used for initial cell search,
synchronization, or channel estimation at a UE, and the UpPTS is
used for channel estimation and UL transmission synchronization
with a UE at an eNB. The GP is used to cancel UL interference
between a UL and a DL, caused by the multi-path delay of a DL
signal.
[Table 1] below lists special subframe configurations
(DwPTS/GP/UpPTS lengths).
TABLE-US-00001 TABLE 1 Normal cyclic prefix in downlink Extended
cyclic prefix in downlink UpPTS UpPTS Normal Extended Normal
Extended Special subframe cyclic prefix cyclic prefix cyclic prefix
cyclic prefix configuration DwPTS in uplink in uplink DwPTS in
uplink in uplink 0 6592 T.sub.s 2192 T.sub.s 2560 T.sub.s 7680
T.sub.s 2192 T.sub.s 2560 T.sub.s 1 19760 T.sub.s 20480 T.sub.s 2
21952 T.sub.s 23040 T.sub.s 3 24144 T.sub.s 25600 T.sub.s 4 26336
T.sub.s 7680 T.sub.s 4384 T.sub.s 5120 T.sub.s 5 6592 T.sub.s 4384
T.sub.s 5120 T.sub.s 20480 T.sub.s 6 19760 T.sub.s 23040 T.sub.s 7
21952 T.sub.s -- -- -- 8 24144 T.sub.s -- -- --
FIG. 3 illustrates an exemplary structure of a DL resource grid for
the duration of one DL slot, which may be used in embodiments of
the present disclosure.
Referring to FIG. 3, a DL slot includes a plurality of OFDM symbols
in the time domain. One DL slot includes 7 OFDM symbols in the time
domain and an RB includes 12 subcarriers in the frequency domain,
to which the present disclosure is not limited.
Each element of the resource grid is referred to as a Resource
Element (RE). An RB includes 12.times.7 REs. The number of RBs in a
DL slot, NDL depends on a DL transmission bandwidth. A UL slot may
have the same structure as a DL slot.
FIG. 4 illustrates a structure of a UL subframe which may be used
in embodiments of the present disclosure.
Referring to FIG. 4, a UL subframe may be divided into a control
region and a data region in the frequency domain. A PUCCH carrying
UCI is allocated to the control region and a PUSCH carrying user
data is allocated to the data region. To maintain a single carrier
property, a UE does not transmit a PUCCH and a PUSCH
simultaneously. A pair of RBs in a subframe are allocated to a
PUCCH for a UE. The RBs of the RB pair occupy different subcarriers
in two slots. Thus it is said that the RB pair frequency-hops over
a slot boundary.
FIG. 5 illustrates a structure of a DL subframe that may be used in
embodiments of the present disclosure.
Referring to FIG. 5, up to three OFDM symbols of a DL subframe,
starting from OFDM symbol 0 are used as a control region to which
control channels are allocated and the other OFDM symbols of the DL
subframe are used as a data region to which a PDSCH is allocated.
DL control channels defined for the 3GPP LTE system include a
Physical Control Format Indicator Channel (PCFICH), a PDCCH, and a
Physical Hybrid ARQ Indicator Channel (PHICH).
The PCFICH is transmitted in the first OFDM symbol of a subframe,
carrying information about the number of OFDM symbols used for
transmission of control channels (i.e. the size of the control
region) in the subframe. The PHICH is a response channel to a UL
transmission, delivering an HARQ ACK/NACK signal. Control
information carried on the PDCCH is called Downlink Control
Information (DCI). The DCI transports UL resource assignment
information, DL resource assignment information, or UL Transmission
(Tx) power control commands for a UE group.
1.2 Physical Downlink Control Channel (PDCCH)
1.2.1 PDCCH Overview
The PDCCH may deliver information about resource allocation and a
transport format for a Downlink Shared Channel (DL-SCH) (i.e. a DL
grant), information about resource allocation and a transport
format for an Uplink Shared Channel (UL-SCH) (i.e. a UL grant),
paging information of a Paging Channel (PCH), system information on
the DL-SCH, information about resource allocation for a
higher-layer control message such as a random access response
transmitted on the PDSCH, a set of Tx power control commands for
individual UEs of a UE group, Voice Over Internet Protocol (VoIP)
activation indication information, etc.
A plurality of PDCCHs may be transmitted in the control region. A
UE may monitor a plurality of PDCCHs. A PDCCH is transmitted in an
aggregate of one or more consecutive Control Channel Elements
(CCEs). A PDCCH made up of one or more consecutive CCEs may be
transmitted in the control region after subblock interleaving. A
CCE is a logical allocation unit used to provide a PDCCH at a code
rate based on the state of a radio channel. A CCE includes a
plurality of RE Groups (REGs). The format of a PDCCH and the number
of available bits for the PDCCH are determined according to the
relationship between the number of CCEs and a code rate provided by
the CCEs.
1.2.2 PDCCH Structure
A plurality of PDCCHs for a plurality of UEs may be multiplexed and
transmitted in the control region. A PDCCH is made up of an
aggregate of one or more consecutive CCEs. A CCE is a unit of 9
REGs each REG including 4 REs. Four Quadrature Phase Shift Keying
(QPSK) symbols are mapped to each REG. REs occupied by RS s are
excluded from REGs. That is, the total number of REGs in an OFDM
symbol may be changed depending on the presence or absence of a
cell-specific RS. The concept of an REG to which four REs are
mapped is also applicable to other DL control channels (e.g. the
PCFICH or the PHICH). Let the number of REGs that are not allocated
to the PCFICH or the PHICH be denoted by NREG. Then the number of
CCEs available to the system is NCCE(=.left
brkt-bot.N.sub.REG/9.right brkt-bot.) and the CCEs are indexed from
0 to NCCE-1.
To simplify the decoding process of a UE, a PDCCH format including
n CCEs may start with a CCE having an index equal to a multiple of
n. That is, given CCE i, the PDCCH format may start with a CCE
satisfying i mod n=0.
The eNB may configure a PDCCH with 1, 2, 4, or 8 CCEs. {1, 2, 4, 8}
are called CCE aggregation levels. The number of CCEs used for
transmission of a PDCCH is determined according to a channel state
by the eNB. For example, one CCE is sufficient for a PDCCH directed
to a UE in a good DL channel state (a UE near to the eNB). On the
other hand, 8 CCEs may be required for a PDCCH directed to a UE in
a poor DL channel state (a UE at a cell edge) in order to ensure
sufficient robustness.
[Table 2] below illustrates PDCCH formats. 4 PDCCH formats are
supported according to CCE aggregation levels as illustrated in
[Table 2].
TABLE-US-00002 TABLE 2 Number of Number Number of PDCCH format CCEs
(n) of REGs PDCCH bits 0 1 9 72 1 2 18 144 2 4 36 288 3 8 72
576
A different CCE aggregation level is allocated to each UE because
the format or Modulation and Coding Scheme (MCS) level of control
information delivered in a PDCCH for the UE is different. An MCS
level defines a code rate used for data coding and a modulation
order. An adaptive MCS level is used for link adaptation. In
general, three or four MCS levels may be considered for control
channels carrying control information.
Regarding the formats of control information, control information
transmitted on a PDCCH is called DCI. The configuration of
information in PDCCH payload may be changed depending on the DCI
format. The PDCCH payload is information bits. [Table 3] lists DCI
according to DCI formats.
TABLE-US-00003 TABLE 3 DCI Format Description Format 0 Resource
grants for the PUSCH transmissions (uplink) Format 1 Resource
assignments for single codeword PDSCH transmissions (transmission
modes 1, 2 and 7) Format 1A Compact signaling of resource
assignments for single codeword PDSCH (all modes) Format 1B Compact
resource assignments for PDSCH using rank-1 closed loop precoding
(mode 6) Format 1C Very compact resource assignments for PDSCH
(e.g. paging/ broadcast system information) Format 1D Compact
resource assignments for PDSCH using multi-user MIMO (mode 5)
Format 2 Resource assignments for PDSCH for closed-loop MIMO
operation (mode 4) Format 2A Resource assignments for PDSCH for
open-loop MIMO operation (mode 3) Format Power control commands for
PUCCH and PUSCH with 2-bit/ 3/3A 1-bit power adjustment Format 4
Scheduling of PUSCH in one UL cell with multi-antenna port
transmission mode
Referring to [Table 3], the DCI formats include Format 0 for PUSCH
scheduling, Format 1 for single-codeword PDSCH scheduling, Format
1A for compact single-codeword PDSCH scheduling, Format 1C for very
compact DL-SCH scheduling, Format 2 for PDSCH scheduling in a
closed-loop spatial multiplexing mode, Format 2A for PDSCH
scheduling in an open-loop spatial multiplexing mode, and Format
3/3A for transmission of Transmission Power Control (TPC) commands
for uplink channels. DCI Format 1A is available for PDSCH
scheduling irrespective of the transmission mode of a UE.
The length of PDCCH payload may vary with DCI formats. In addition,
the type and length of PDCCH payload may be changed depending on
compact or non-compact scheduling or the transmission mode of a
UE.
The transmission mode of a UE may be configured for DL data
reception on a PDSCH at the UE. For example, DL data carried on a
PDSCH includes scheduled data, a paging message, a random access
response, broadcast information on a BCCH, etc. for a UE. The DL
data of the PDSCH is related to a DCI format signaled through a
PDCCH. The transmission mode may be configured semi-statically for
the UE by higher-layer signaling (e.g. Radio Resource Control (RRC)
signaling). The transmission mode may be classified as single
antenna transmission or multi-antenna transmission.
A transmission mode is configured for a UE semi-statically by
higher-layer signaling. For example, multi-antenna transmission
scheme may include transmit diversity, open-loop or closed-loop
spatial multiplexing, Multi-User Multiple Input Multiple Output
(MU-MIMO), or beamforming. Transmit diversity increases
transmission reliability by transmitting the same data through
multiple Tx antennas. Spatial multiplexing enables high-speed data
transmission without increasing a system bandwidth by
simultaneously transmitting different data through multiple Tx
antennas. Beamforming is a technique of increasing the Signal to
Interference plus Noise Ratio (SINR) of a signal by weighting
multiple antennas according to channel states.
A DCI format for a UE depends on the transmission mode of the UE.
The UE has a reference DCI format monitored according to the
transmission mode configure for the UE. The following 10
transmission modes are available to UEs:
(1) Transmission mode 1: Single antenna port (port 0);
(2) Transmission mode 2: Transmit diversity;
(3) Transmission mode 3: Open-loop spatial multiplexing when the
number of layer is larger than 1 or Transmit diversity when the
rank is 1;
(4) Transmission mode 4: Closed-loop spatial multiplexing;
(5) Transmission mode 5: MU-MIMO;
(6) Transmission mode 6: Closed-loop rank-1 precoding;
(7) Transmission mode 7: Precoding supporting a single layer
transmission, which does not based on a codebook (Rel-8);
(8) Transmission mode 8: Precoding supporting up to two layers,
which do not based on a codebook (Rel-9);
(9) Transmission mode 9: Precoding supporting up to eight layers,
which do not based on a codebook (Rel-10); and
(10) Transmission mode 10: Precoding supporting up to eight layers,
which do not based on a codebook, used for CoMP (Rel-11).
1.2.3 PDCCH Transmission
The eNB determines a PDCCH format according to DCI that will be
transmitted to the UE and adds a Cyclic Redundancy Check (CRC) to
the control information. The CRC is masked by a unique Identifier
(ID) (e.g. a Radio Network Temporary Identifier (RNTI)) according
to the owner or usage of the PDCCH. If the PDCCH is destined for a
specific UE, the CRC may be masked by a unique ID (e.g. a cell-RNTI
(C-RNTI)) of the UE. If the PDCCH carries a paging message, the CRC
of the PDCCH may be masked by a paging indicator ID (e.g. a
Paging-RNTI (P-RNTI)). If the PDCCH carries system information,
particularly, a System Information Block (SIB), its CRC may be
masked by a system information ID (e.g. a System Information RNTI
(SI-RNTI)). To indicate that the PDCCH carries a random access
response to a random access preamble transmitted by a UE, its CRC
may be masked by a Random Access-RNTI (RA-RNTI).
Then the eNB generates coded data by channel-encoding the CRC-added
control information. The channel coding may be performed at a code
rate corresponding to an MCS level. The eNB rate-matches the coded
data according to a CCE aggregation level allocated to a PDCCH
format and generates modulation symbols by modulating the coded
data. Herein, a modulation order corresponding to the MCS level may
be used for the modulation. The CCE aggregation level for the
modulation symbols of a PDCCH may be one of 1, 2, 4, and 8.
Subsequently, the eNB maps the modulation symbols to physical REs
(i.e. CCE to RE mapping).
1.2.4 Blind Decoding (BD)
A plurality of PDCCHs may be transmitted in a subframe. That is,
the control region of a subframe includes a plurality of CCEs, CCE
0 to CCE NCCE,k-1. NCCE,k is the total number of CCEs in the
control region of a kth subframe. A UE monitors a plurality of
PDCCHs in every subframe. This means that the UE attempts to decode
each PDCCH according to a monitored PDCCH format.
The eNB does not provide the UE with information about the position
of a PDCCH directed to the UE in an allocated control region of a
subframe. Without knowledge of the position, CCE aggregation level,
or DCI format of its PDCCH, the UE searches for its PDCCH by
monitoring a set of PDCCH candidates in the subframe in order to
receive a control channel from the eNB. This is called blind
decoding. Blind decoding is the process of demasking a CRC part
with a UE ID, checking a CRC error, and determining whether a
corresponding PDCCH is a control channel directed to a UE by the
UE.
The UE monitors a PDCCH in every subframe to receive data
transmitted to the UE in an active mode. In a Discontinuous
Reception (DRX) mode, the UE wakes up in a monitoring interval of
every DRX cycle and monitors a PDCCH in a subframe corresponding to
the monitoring interval. The PDCCH-monitored subframe is called a
non-DRX subframe.
To receive its PDCCH, the UE should blind-decode all CCEs of the
control region of the non-DRX subframe. Without knowledge of a
transmitted PDCCH format, the UE should decode all PDCCHs with all
possible CCE aggregation levels until the UE succeeds in
blind-decoding a PDCCH in every non-DRX subframe. Since the UE does
not know the number of CCEs used for its PDCCH, the UE should
attempt detection with all possible CCE aggregation levels until
the UE succeeds in blind decoding of a PDCCH.
In the LTE system, the concept of Search Space (SS) is defined for
blind decoding of a UE. An SS is a set of PDCCH candidates that a
UE will monitor. The SS may have a different size for each PDCCH
format. There are two types SSs, Common Search Space (CSS) and
UE-specific/Dedicated Search Space (USS).
While all UEs may know the size of a CSS, a USS may be configured
for each individual UE. Accordingly, a UE should monitor both a CSS
and a USS to decode a PDCCH. As a consequence, the UE performs up
to 44 blind decodings in one subframe, except for blind decodings
based on different CRC values (e.g., C-RNTI, P-RNTI, SI-RNTI, and
RA-RNTI).
In view of the constraints of an SS, the eNB may not secure CCE
resources to transmit PDCCHs to all intended UEs in a given
subframe. This situation occurs because the remaining resources
except for allocated CCEs may not be included in an SS for a
specific UE. To minimize this obstacle that may continue in the
next subframe, a UE-specific hopping sequence may apply to the
starting point of a USS.
[Table 4] illustrates the sizes of CSSs and USSs.
TABLE-US-00004 TABLE 4 Number of candidates Number of candidates
PDCCH Number of in common search in dedicated search format CCEs
(n) space space 0 1 -- 6 1 2 -- 6 2 4 4 2 3 8 2 2
To mitigate the load of the UE caused by the number of blind
decoding attempts, the UE does not search for all defined DCI
formats simultaneously. Specifically, the UE always searches for
DCI Format 0 and DCI Format 1A in a USS. Although DCI Format 0 and
DCI Format 1A are of the same size, the UE may distinguish the DCI
formats by a flag for format0/format 1a differentiation included in
a PDCCH. Other DCI formats than DCI Format 0 and DCI Format 1A,
such as DCI Format 1, DCI Format 1B, and DCI Format 2 may be
required for the UE.
The UE may search for DCI Format 1A and DCI Format 1C in a CSS. The
UE may also be configured to search for DCI Format 3 or 3A in the
CSS. Although DCI Format 3 and DCI Format 3A have the same size as
DCI Format 0 and DCI Format 1A, the UE may distinguish the DCI
formats by a CRC scrambled with an ID other than a UE-specific
ID.
An SS.sub.k.sup.(L) is a PDCCH candidate set with a CCE aggregation
level L {1,2,4,8}. The CCEs of PDCCH candidate set m in the SS may
be determined by the following equation. L{(Y.sub.k+m)mod .left
brkt-bot.N.sub.CCE,k/L.right brkt-bot.}+i [Equation 1]
where M.sup.(L) is the number of PDCCH candidates with CCE
aggregation level L to be monitored in the SS, m=0, . . .
M.sup.(L)-1, i is the index of a CCE in each PDCCH candidate, and
i=0, L-1. k=.left brkt-bot.n.sub.s/2.right brkt-bot. where n.sub.s
is the index of a slot in a radio frame.
As described before, the UE monitors both the USS and the CSS to
decode a PDCCH. The CSS supports PDCCHs with CCE aggregation levels
{4, 8} and the USS supports PDCCHs with CCE aggregation levels {1,
2, 4, 8}. [Table 5] illustrates PDCCH candidates monitored by a
UE.
TABLE-US-00005 TABLE 5 Search space S.sub.k.sup.(L) Aggregation
Number of PDCCH Type level L Size [in CCEs] candidates M.sup.(L)
UE- 1 6 6 specific 2 12 6 4 8 2 8 16 2 Common 4 16 4 8 16 2
Referring to [Equation 1], for two aggregation levels, L=4 and L=8,
Y.sub.k is set to 0 in the CSS, whereas Y.sub.k is defined by
[Equation 2] for aggregation level L in the USS.
Y.sub.k=(AY.sub.k-1 mod D [Equation 2]
where Y.sub.-1=n.sub.RNTI.noteq.0, n.sub.RNTI indicating an RNTI
value. A=39827 and D=65537.
1.3 Carrier Aggregation (CA) Environment
1.3.1 CA Overview
A 3GPP LTE system (conforming to Rel-8 or Rel-9) (hereinafter,
referred to as an LTE system) uses Multi-Carrier Modulation (MCM)
in which a single Component Carrier (CC) is divided into a
plurality of bands. In contrast, a 3GPP LTE-A system (hereinafter,
referred to an LTE-A system) may use CA by aggregating one or more
CCs to support a broader system bandwidth than the LTE system. The
term CA is interchangeably used with carrier combining, multi-CC
environment, or multi-carrier environment.
In the present disclosure, multi-carrier means CA (or carrier
combining). Herein, CA covers aggregation of contiguous carriers
and aggregation of non-contiguous carriers. The number of
aggregated CCs may be different for a DL and a UL. If the number of
DL CCs is equal to the number of UL CCs, this is called symmetric
aggregation. If the number of DL CCs is different from the number
of UL CCs, this is called asymmetric aggregation. The term CA is
interchangeable with carrier combining, bandwidth aggregation,
spectrum aggregation, etc.
The LTE-A system aims to support a bandwidth of up to 100 MHz by
aggregating two or more CCs, that is, by CA. To guarantee backward
compatibility with a legacy IMT system, each of one or more
carriers, which has a smaller bandwidth than a target bandwidth,
may be limited to a bandwidth used in the legacy system.
For example, the legacy 3GPP LTE system supports bandwidths {1.4,
3, 5, 10, 15, and 20 MHz} and the 3GPP LTE-A system may support a
broader bandwidth than 20 MHz using these LTE bandwidths. A CA
system of the present disclosure may support CA by defining a new
bandwidth irrespective of the bandwidths used in the legacy
system.
There are two types of CA, intra-band CA and inter-band CA.
Intra-band CA means that a plurality of DL CCs and/or UL CCs are
successive or adjacent in frequency. In other words, the carrier
frequencies of the DL CCs and/or UL CCs are positioned in the same
band. On the other hand, an environment where CCs are far away from
each other in frequency may be called inter-band CA. In other
words, the carrier frequencies of a plurality of DL CCs and/or UL
CCs are positioned in different bands. In this case, a UE may use a
plurality of Radio Frequency (RF) ends to conduct communication in
a CA environment.
The LTE-A system adopts the concept of cell to manage radio
resources. The above-described CA environment may be referred to as
a multi-cell environment. A cell is defined as a pair of DL and UL
CCs, although the UL resources are not mandatory. Accordingly, a
cell may be configured with DL resources alone or DL and UL
resources.
For example, if one serving cell is configured for a specific UE,
the UE may have one DL CC and one UL CC. If two or more serving
cells are configured for the UE, the UE may have as many DL CCs as
the number of the serving cells and as many UL CCs as or fewer UL
CCs than the number of the serving cells, or vice versa. That is,
if a plurality of serving cells are configured for the UE, a CA
environment using more UL CCs than DL CCs may also be
supported.
CA may be regarded as aggregation of two or more cells having
different carrier frequencies (center frequencies). Herein, the
term `cell` should be distinguished from `cell` as a geographical
area covered by an eNB. Hereinafter, intra-band CA is referred to
as intra-band multi-cell and inter-band CA is referred to as
inter-band multi-cell.
In the LTE-A system, a Primacy Cell (PCell) and a Secondary Cell
(SCell) are defined. A PCell and an SCell may be used as serving
cells. For a UE in RRC_CONNECTED state, if CA is not configured for
the UE or the UE does not support CA, a single serving cell
including only a PCell exists for the UE. On the contrary, if the
UE is in RRC_CONNECTED state and CA is configured for the UE, one
or more serving cells may exist for the UE, including a PCell and
one or more SCells.
Serving cells (PCell and SCell) may be configured by an RRC
parameter. A physical-layer ID of a cell, PhysCellId is an integer
value ranging from 0 to 503. A short ID of an SCell, SCellIndex is
an integer value ranging from 1 to 7. A short ID of a serving cell
(PCell or SCell), ServeCellIndex is an integer value ranging from 1
to 7. If ServeCellIndex is 0, this indicates a PCell and the values
of ServeCellIndex for SCells are pre-assigned. That is, the
smallest cell ID (or cell index) of ServeCellIndex indicates a
PCell.
A PCell refers to a cell operating in a primary frequency (or a
primary CC). A UE may use a PCell for initial connection
establishment or connection reestablishment. The PCell may be a
cell indicated during handover. In addition, the PCell is a cell
responsible for control-related communication among serving cells
configured in a CA environment. That is, PUCCH allocation and
transmission for the UE may take place only in the PCell. In
addition, the UE may use only the PCell in acquiring system
information or changing a monitoring procedure. An Evolved
Universal Terrestrial Radio Access Network (E-UTRAN) may change
only a PCell for a handover procedure by a higher-layer
RRCConnectionReconfiguraiton message including mobilityControlInfo
to a UE supporting CA.
An SCell may refer to a cell operating in a secondary frequency (or
a secondary CC). Although only one PCell is allocated to a specific
UE, one or more SCells may be allocated to the UE. An SCell may be
configured after RRC connection establishment and may be used to
provide additional radio resources. There is no PUCCH in cells
other than a PCell, that is, in SCells among serving cells
configured in the CA environment.
When the E-UTRAN adds an SCell to a UE supporting CA, the E-UTRAN
may transmit all system information related to operations of
related cells in RRC_CONNECTED state to the UE by dedicated
signaling. Changing system information may be controlled by
releasing and adding a related SCell. Herein, a higher-layer
RRCConnectionReconfiguration message may be used. The E-UTRAN may
transmit a dedicated signal having a different parameter for each
cell rather than it broadcasts in a related SCell.
After an initial security activation procedure starts, the E-UTRAN
may configure a network including one or more SCells by adding the
SCells to a PCell initially configured during a connection
establishment procedure. In the CA environment, each of a PCell and
an SCell may operate as a CC. Hereinbelow, a Primary CC (PCC) and a
PCell may be used in the same meaning and a Secondary CC (SCC) and
an SCell may be used in the same meaning in embodiments of the
present disclosure.
FIG. 6 illustrates an example of CCs and CA in the LTE-A system,
which are used in embodiments of the present disclosure.
FIG. 6(a) illustrates a single carrier structure in the LTE system.
There are a DL CC and a UL CC and one CC may have a frequency range
of 20 MHz.
FIG. 6(b) illustrates a CA structure in the LTE-A system. In the
illustrated case of FIG. 6(b), three CCs each having 20 MHz are
aggregated. While three DL CCs and three UL CCs are configured, the
numbers of DL CCs and UL CCs are not limited. In CA, a UE may
monitor three CCs simultaneously, receive a DL signal/DL data in
the three CCs, and transmit a UL signal/UL data in the three
CCs.
If a specific cell manages N DL CCs, the network may allocate M
(M.ltoreq.N) DL CCs to a UE. The UE may monitor only the M DL CCs
and receive a DL signal in the M DL CCs. The network may prioritize
L (L.ltoreq.M.ltoreq.N) DL CCs and allocate a main DL CC to the UE.
In this case, the UE should monitor the L DL CCs. The same thing
may apply to UL transmission.
The linkage between the carrier frequencies of DL resources (or DL
CCs) and the carrier frequencies of UL resources (or UL CCs) may be
indicated by a higher-layer message such as an RRC message or by
system information. For example, a set of DL resources and UL
resources may be configured based on linkage indicated by System
Information Block Type 2 (SIB2). Specifically, DL-UL linkage may
refer to a mapping relationship between a DL CC carrying a PDCCH
with a UL grant and a UL CC using the UL grant, or a mapping
relationship between a DL CC (or a UL CC) carrying HARQ data and a
UL CC (or a DL CC) carrying an HARQ ACK/NACK signal.
1.3.2 Cross Carrier Scheduling
Two scheduling schemes, self-scheduling and cross carrier
scheduling are defined for a CA system, from the perspective of
carriers or serving cells. Cross carrier scheduling may be called
cross CC scheduling or cross cell scheduling.
In self-scheduling, a PDCCH (carrying a DL grant) and a PDSCH are
transmitted in the same DL CC or a PUSCH is transmitted in a UL CC
linked to a DL CC in which a PDCCH (carrying a UL grant) is
received.
In cross carrier scheduling, a PDCCH (carrying a DL grant) and a
PDSCH are transmitted in different DL CCs or a PUSCH is transmitted
in a UL CC other than a UL CC linked to a DL CC in which a PDCCH
(carrying a UL grant) is received.
Cross carrier scheduling may be activated or deactivated
UE-specifically and indicated to each UE semi-statically by
higher-layer signaling (e.g. RRC signaling).
If cross carrier scheduling is activated, a Carrier Indicator Field
(CIF) is required in a PDCCH to indicate a DL/UL CC in which a
PDSCH/PUSCH indicated by the PDCCH is to be transmitted. For
example, the PDCCH may allocate PDSCH resources or PUSCH resources
to one of a plurality of CCs by the CIF. That is, when a PDCCH of a
DL CC allocates PDSCH or PUSCH resources to one of aggregated DL/UL
CCs, a CIF is set in the PDCCH. In this case, the DCI formats of
LTE Release-8 may be extended according to the CIF. The CIF may be
fixed to three bits and the position of the CIF may be fixed
irrespective of a DCI format size. In addition, the LTE Release-8
PDCCH structure (the same coding and resource mapping based on the
same CCEs) may be reused.
On the other hand, if a PDCCH transmitted in a DL CC allocates
PDSCH resources of the same DL CC or allocates PUSCH resources in a
single UL CC linked to the DL CC, a CIF is not set in the PDCCH. In
this case, the LTE Release-8 PDCCH structure (the same coding and
resource mapping based on the same CCEs) may be used.
If cross carrier scheduling is available, a UE needs to monitor a
plurality of PDCCHs for DCI in the control region of a monitoring
CC according to the transmission mode and/or bandwidth of each CC.
Accordingly, an appropriate SS configuration and PDCCH monitoring
are needed for the purpose.
In the CA system, a UE DL CC set is a set of DL CCs scheduled for a
UE to receive a PDSCH, and a UE UL CC set is a set of UL CCs
scheduled for a UE to transmit a PUSCH. A PDCCH monitoring set is a
set of one or more DL CCs in which a PDCCH is monitored. The PDCCH
monitoring set may be identical to the UE DL CC set or may be a
subset of the UE DL CC set. The PDCCH monitoring set may include at
least one of the DL CCs of the UE DL CC set. Or the PDCCH
monitoring set may be defined irrespective of the UE DL CC set. DL
CCs included in the PDCCH monitoring set may be configured to
always enable self-scheduling for UL CCs linked to the DL CCs. The
UE DL CC set, the UE UL CC set, and the PDCCH monitoring set may be
configured UE-specifically, UE group-specifically, or
cell-specifically.
If cross carrier scheduling is deactivated, this implies that the
PDCCH monitoring set is always identical to the UE DL CC set. In
this case, there is no need for signaling the PDCCH monitoring set.
However, if cross carrier scheduling is activated, the PDCCH
monitoring set may be defined within the UE DL CC set. That is, the
eNB transmits a PDCCH only in the PDCCH monitoring set to schedule
a PDSCH or PUSCH for the UE.
FIG. 7 illustrates a cross carrier-scheduled subframe structure in
the LTE-A system, which is used in embodiments of the present
disclosure.
Referring to FIG. 7, three DL CCs are aggregated for a DL subframe
for LTE-A UEs. DL CC `A` is configured as a PDCCH monitoring DL CC.
If a CIF is not used, each DL CC may deliver a PDCCH that schedules
a PDSCH in the same DL CC without a CIF. On the other hand, if the
CIF is used by higher-layer signaling, only DL CC `A` may carry a
PDCCH that schedules a PDSCH in the same DL CC `A` or another CC.
Herein, no PDCCH is transmitted in DL CC `B` and DL CC `C` that are
not configured as PDCCH monitoring DL CCs.
2. Control Signal Transmission Through PUCCH (Physical Uplink
Control Channel)
The PUCCH is an uplink control channel used to transmit uplink
control information (UCI). The UCI transmitted on the PUCCH
includes scheduling request (SR) information, HARQ ACK/NACK
information and CQI information.
The amount of control information which a UE can transmit in a
subframe depends on the number of SC-FDMA symbols available for
transmission of control signaling data (at this time, excluding
SC-FDMA symbols used for transmission of reference signals used for
coherent detection of the PUCCH). The LTE/LTE-A system supports 7
different PUCCH formats depending on information which will be
signaled on the PUCCH.
PUCCH may include the following formats to transmit control
information.
(1) Format 1: On-Off keying (OOK) modulation, used for SR
(Scheduling Request)
(2) Format 1a & 1b: Used for ACK/NACK transmission
1) Format 1a: BPSK ACK/NACK for 1 codeword
2) Format 1b: QPSK ACK/NACK for 2 codewords
(3) Format 2: QPSK modulation, used for CQI transmission
(4) Format 2a & Format 2b: Used for simultaneous transmission
of CQI and ACK/NACK
(5) Format 3: Used for multiple ACK/NACK transmission in a carrier
aggregation environment
Table 6 shows a modulation scheme according to PUCCH format and the
number of bits per subframe. Table 7 shows the number of reference
signals (RS) per slot according to PUCCH format. Table 8 shows
SC-FDMA symbol location of RS (reference signal) according to PUCCH
format. In Table 6, PUCCH format 2a and PUCCH format 2b correspond
to a case of normal cyclic prefix (CP).
TABLE-US-00006 TABLE 6 PUCCH No. of bits per format Modulation
scheme subframe, Mbit 1 N/A N/A 1a BPSK 1 1b QPSK 2 2 QPSK 20 2a
QPSK + BPSK 21 2b QPSK + BPSK 22 3 QPSK 48
TABLE-US-00007 TABLE 7 PUCCH format Normal CP Extended CP 1, 1a, 1b
3 2 2, 3 2 1 2a, 2b 2 N/A
TABLE-US-00008 TABLE 8 SC-FDMA symbol location of RS PUCCH format
Normal CP Extended CP 1, 1a, 1b 2, 3, 4 2, 3 2, 3 1, 5 3 2a, 2b 1,
5 N/A
FIG. 8 is a diagram illustrating physical mapping of a PUCCH format
into PUCCH RBs.
Referring to FIG. 8, PUCCH formats 2/2a/2b are mapped and allocated
to edge RBs of a PUCCH band (for example, PUCCH region m=0, 1), and
then PUCCH RB where PUCCH formats 2/2a/2b are combined with PUCCH
formats 1/1a/1b is allocated (for example, PUCCH region m=2). Next,
the PUCCH formats 1/1a/1b are allocated to the PUCCH RBs (for
example, PUCCH region m=3, 4, 5). Information on the number
N.sub.RB.sup.(2) of PUCCH RBs used for the PUCCH formats 2/2a/2b is
transferred from the cell to the UEs by a broadcast signal. FIG. 8
illustrates an example of PUCCH formats which are allocated,
wherein the PUCCH formats actually mapped onto the PUCCH can be
allocated sequentially in accordance with the aforementioned
order.
2.1 CQI Transmission Through PUCCH Format
FIG. 9 is a diagram illustrating PUCCH formats 2/2a/2b in case of a
normal cyclic prefix, and FIG. 10 is a diagram illustrating PUCCH
formats 2/2a/2b in case of an extended cyclic prefix.
The periodicity and frequency resolution used by a UE to report CQI
are both controlled by the eNB. In the time domain, both periodic
and aperiodic CQI reporting are supported. The PUCCH format 2 is
used for periodic CQI reporting only, and the PUSCH is used for
aperiodic reporting of the CQI. At this time, the eNB especially
commands CQI reporting to the UE, and the UE transmits CQI report
to a resource which is scheduled for uplink data transmission.
The PUCCH CQI channel structure for one slot in case of a normal CP
will be understood with reference to FIG. 9. In this case, SC-FDMA
symbols 1 and 5 (i.e., the second and sixth symbols) are used for
DM RS (Demodulation Reference Signal) transmission. The PUCCH CQI
channel structure for one slot in case of an extended CP will be
understood with reference to FIG. 10. In this case, SC-FDMA symbol
3 is used for DM RS transmission. The DM-RS is a reference signal
transmitted by the UE to the uplink and may be referred to as UL
RS.
CQI information of 10 bits channel coded with a 1/2 coding rate is
punctured by (20, k) Reed-Muller (RM) code to give 20 coded bits.
Afterwards, the CQI information is scrambled (for example,
scrambled in a similar way to PUSCH data with a length-31 Gold
sequence) prior to QPSK constellation mapping. One QPSK modulated
symbol is transmitted to each of the 10 SC-FDMA symbols in the
subframe by modulating a cyclic time shift of the base RS sequence
of length-12 prior to OFDM modulation. The 12 equally-spaced cyclic
time shifts allow 12 different UEs to be orthogonally multiplexed
on the same CQI PUCCH RB. The DM RS sequence is similar to the
frequency domain CQI signal sequence but does not include CQI data
modulation.
The UE is configured to periodically report different CQI, PMI, and
RI types on CQI PUCCH by receiving a higher layer signal that
includes a PUCCH resource index n.sub.PUCCH.sup.(2), which
indicates both the cyclic time shift and the PUCCH region which
will be used.
2.2 HARQ ACK/NACK Transmission Through PUCCH Format 1
FIG. 11 is a diagram illustrating PUCCH formats 1a/1b in case of a
normal cyclic prefix, and FIG. 12 is a diagram illustrating PUCCH
formats 1a/1b in case of an extended cyclic prefix.
Referring to FIGS. 11 and 12, three SC-FDMA symbols in the middle
of the slot are used for UL-RS in case of the normal CP, and two
SC-FDMA symbols in the middle of the slot are used for UL-RS in
case of the extended CP. At this time, both 1- and 2-bit ACK/NACKs
are modulated using BPSK and QPSK modulation, respectively.
In case of CQI transmission, the one BPSK/QPSK modulated symbol is
transmitted on each SC-FDMA data symbol by modulating a cyclic time
shift of the base RS sequence of length-12 (i.e. frequency-domain
CDM) prior to OFDM modulation. In addition, time-domain spread
codes with orthogonal (Walsh-Hadamard of DFT) spreading codes are
used to code-division-multiplex UEs. The RSs from the different UEs
are multiplexed in the same way as the data SC-FDMA symbols.
2.3 Multiplexing of CQI and ACK/NACK
In the LTE system, simultaneous transmission of HARQ ACK/NACK and
CQI is enabled by UE-specific higher layer signaling.
In the case that simultaneous transmission is not enabled and that
the UE is configured to report CQI on the PUCCH of the same
subframe that needs HARQ ACK/NACK transmission, CQI report is
dropped and only HARQ ACK/NACK is transmitted using the PUCCH
format 1a/1b.
In the case that simultaneous transmission is enabled, the CQI and
the 1- or 2-bit ACK/NACK information need to be multiplexed on the
same PUCCH RB while maintaining the low CM (Cubic Metric) single
carrier property. The methods used to achieve this are different
for the case of normal CP and extended CP.
In the case of the normal CP, to transmit a 1- or 2-bit HARQ
ACK/NACK together with CQI, the ACK/NACK bits (which are not
scrambled) are BPSK/QPSK modulated as shown in FIG. 13, resulting
in a single HARQ ACK/NACK modulation symbol d.sub.HARQ. FIG. 13 is
a diagram illustrating one of constellation mapping of HARQ
ACK/NACK for a normal CP. At this time, an ACK signal is encoded as
a binary `1` and a NACK signal is encoded as a binary `0`. The
single HARQ ACK/NACK modulation symbol, d.sub.HARQ, is then used to
modulate the second RS symbol (SC-FDMA symbol 5, i.e., RS signaled
by ACK/NACK) in each CQI slot. That is, ACK/NACK is signaled using
the corresponding RS.
In case of the extended CP with one RS symbol per slot, the 1- or
2-bit HARQ ACK/NACK is jointly encoded with the CQI resulting in a
(20, k.sub.CQI+k.sub.A/N) Reed-Muller based block code. A 20-bit
codeword is transmitted on the PUCCH that uses the CQI channel
structure of FIG. 9. The joint coding of the ACK/NACK and CQI is
performed as shown in FIG. 14. The largest number of information
bits supported by the block code is 13. At this time, k.sub.CQI=11
CQI bits and k.sub.A/N=2 bits.
2.4 Multiplexing of SR and ACK/NACK
FIG. 15 is a diagram illustrating one of methods for multiplexing
SR and ACK/NACK signal, and FIG. 16 is a diagram illustrating
constellation mapping of ACK/NACK and SR for PUCCH formats
1/1a/1b.
Referring to FIG. 15, if an SR signal and an ACK/NACK signal are
simultaneously transmitted at the same subframe, the UE transmits
the ACK/NACK signal on the SR PUCCH resource allocated for a
positive SR or transmits ACK/NACK on the ACK/NACK PUCCH resource
allocated in case of a negative SR. The constellation mapping for
simultaneous transmission of ACK/NACK and SR is shown in FIG.
16.
2.5 HARQ ACK/NACK Transmission in TDD System
In case of LTE TDD (Time Division Multiplexing), since the UE can
receive PDSCHs during a plurality of subframes, the UE can feed
HARQ ACK/NACK for multiple PDSCHs back to the eNB. That is, there
are two types of HARQ ACK/NACK transmission schemes as follows.
(1) ACK/NACK Bundling
With ACK/NACK bundling, ACK/NACK responses for multiple data units
are combined by logical-AND operation. For example, if the Rx node
(or receiver) decodes all the data units successfully, the Rx node
transmits ACK using one ACK/NACK unit. Otherwise, if the Rx node
fails in decoding any of the data units, the Rx node may either
transmit NACK using one ACK/NACK unit or transmit nothing for
ACK/NACK.
(2) ACK/NACK Multiplexing
With ACK/NACK multiplexing, contents of the ACK/NACK responses for
multiple data units are identified by the combination of the
ACK/NACK unit used in actual ACK/NACK transmission and the one of
QPSK modulation symbols. For example, if it is assumed that one
ACK/NACK unit carries two bits and two data units are transmitted
in maximum, the ACK/NACK result can be identified at the TX node as
illustrated in the following Table 9.
TABLE-US-00009 TABLE 9 HARQ-ACK(0), HARQ-ACK(1) n.sub.PUCCH.sup.(1)
b(0), b(1) ACK, ACK n.sub.PUCCH, 1.sup.(1) 1, 1 ACK, NACK/DTX
n.sub.PUCCH, 0.sup.(1) 0, 1 NACK/DTX, ACK n.sub.PUCCH, 1.sup.(1) 0,
0 NACK/DTX, NACK n.sub.PUCCH, 1.sup.(1) 1, 0 NACK, DTX n.sub.PUCCH,
0.sup.(1) 1, 0 DTX, DTX N/A N/A
In Table 9, HARQ-ACK(i) indicates the ACK/NACK result for the data
unit i (there are maximum 2 data units, that is, data unit 0 and
data unit 1 in this example). In Table 9, DTX means there is no
data unit transmitted for corresponding HARQ-ACK(i) or the Rx node
does not detect the existence of the data unit corresponding to
HARQ-ACK(i). n.sub.PUCCH,X.sup.(1) indicates the ACK/NACK unit
which is used in actual ACK/NACK transmission, where there are two
ACK/NACK units, n.sub.PUCCH,0.sup.(1) and n.sub.PUCCH,1.sup.(1) in
maximum.
b(0),b(1) indicates two bits carried by the selected ACK/NACK unit.
Modulation symbol which is transmitted through ACK/NACK unit is
decided in accordance with the bits. For example, if the RX node
receives and decodes two data units successfully, the Rx node
transmits two bits, (1, 1), using ACK/NACK unit
n.sub.PUCCH,1.sup.(1). For another example, if the Rx node receives
two data units, fails in decoding of the first data unit
(corresponding to HARQ-ACK(0)), and decodes the second data unit
(corresponding to HARQ-ACK(1)) successfully, the RX node transmits
two bits (0, 0) using n.sub.PUCCH,X.sup.(1).
By linking the actual ACK/NACK contents with the combination of
ACK/NACK unit selection and the actual bit contents used for
transmission of the ACK/NACK unit, ACK/NACK transmission using
single ACK/NACK unit for multiple data units is possible. The
example described in Table 9 can be extended to the ACK/NACK
transmission for more than 2 data units.
In ACK/NACK multiplexing method, NACK and DTX are coupled as
NACK/DTX as shown in Table 9 if at least one ACK exists for all
data units. This is because that combinations of ACK/NACK unit and
QPSK symbol are insufficient to cover all ACK/NACK hypotheses based
on decoupling of NACK and DTX. On the other hand, for the case that
no ACK exists for all data units (in other words, NACK or DTX only
exists for all data units), single definite NACK case is defined as
the case that only one of HARQ-ACK(i) is NACK decoupled with DTX.
In this case, ACK/NACK unit linked to the data unit corresponding
to single definite NACK can also be reserved to transmit the signal
of multiple ACK/NACKs.
When the maximum number of data units which can be transmitted
within a given amount of physical resources becomes larger, the
required ACK/NACK hypotheses for ACK/NACK multiplexing over all the
data units may exponentially increase. Denoting the maximum number
of data units and the number of corresponding ACK/NACK units as N
and N.sub.A, respectively, 2.sup.N ACK/NACK hypotheses are required
for ACK/NACK multiplexing even if DTX case is precluded. On the
other hand, applying the single ACK/NACK unit selection as
described above, ACK/NACK multiplexing can be supported by up to
4N, ACK/NACK hypotheses.
In other words, as the number of data units increases, the single
ACK/NACK unit selection requires relatively larger amount of
ACK/NACK units which yields increased overhead of control channel
resources required to transmit the signal for multiple ACK/NACKs.
For example, if 5 data units (N=5) are used for transmission, 8
ACK/NACK units (N.sub.A=8) should be available for ACK/NACK
transmission because the required number of ACK/NACK hypotheses for
ACK/NACK multiplexing is 2.sup.N=32 (=4N.sub.A).
2.6 Uplink Channel Coding for PUCCH Format 2
In LTE uplink transmission, certain control channels are encoded
using a linear block code as illustrated in Table 10.
TABLE-US-00010 TABLE 10 i M.sub.i, 0 M.sub.i, 1 M.sub.i, 2 M.sub.i,
3 M.sub.i, 4 M.sub.i, 5 M.sub.i, 6 M.sub.i, 7 M.sub.i, 8 M.sub.i, 9
M.sub.i, 10 M.sub.i, 11 M.sub.i, 12 0 1 1 0 0 0 0 0 0 0 0 1 1 0 1 1
1 1 0 0 0 0 0 0 1 1 1 0 2 1 0 0 1 0 0 1 0 1 1 1 1 1 3 1 0 1 1 0 0 0
0 1 0 1 1 1 4 1 1 1 1 0 0 0 1 0 0 1 1 1 5 1 1 0 0 1 0 1 1 1 0 1 1 1
6 1 0 1 0 1 0 1 0 1 1 1 1 1 7 1 0 0 1 1 0 0 1 1 0 1 1 1 8 1 1 0 1 1
0 0 1 0 1 1 1 1 9 1 0 1 1 1 0 1 0 0 1 1 1 1 10 1 0 1 0 0 1 1 1 0 1
1 1 1 11 1 1 1 0 0 1 1 0 1 0 1 1 1 12 1 0 0 1 0 1 0 1 1 1 1 1 1 13
1 1 0 1 0 1 0 1 0 1 1 1 1 14 1 0 0 0 1 1 0 1 0 0 1 0 1 15 1 1 0 0 1
1 1 1 0 1 1 0 1 16 1 1 1 0 1 1 1 0 0 1 0 1 1 17 1 0 0 1 1 1 0 0 1 0
0 1 1 18 1 1 0 1 1 1 1 1 0 0 0 0 0 19 1 0 0 0 0 1 1 0 0 0 0 0 0
If input bits to the linear block code is denoted as a.sub.0,
a.sub.1, a.sub.2, . . . , a.sub.A, after encoding the bits are
denoted by b.sub.0, b.sub.1, b.sub.2, . . . , b.sub.B where B=20.
The following Equation 3 indicates one of methods for generating
encoded bits.
.times..times..times..times..times..times..times..times..times.
##EQU00001##
The encoded bits then are mapped to code-time-frequency resource as
shown in FIG. 17. FIG. 17 is a diagram illustrating methods of
matching control information with a physical resource region. The
first 10 encoded bits are mapped a specific code-time-frequency
resource, and the last 10 encoded bits are mapped to a different
code-time-frequency resource. At this time, the frequency spacing
between first 10 encoded bits and the last 10 encoded bits are
usually large, whereby frequency diversity for the encoded bits can
be obtained.
2.7 Uplink Channel Coding in LTE-A System
As described above, in the LTE system (that is, Rel-8), if UCI is
transmitted to a PUCCH format 2, (20, A) RM coding of Table 10 is
performed for CSI of maximum 13 bits. However, if the UCI is
transmitted to the PUSCH, (32, A) RM coding of Table 11 is
performed for CQI of maximum 11 bits, and truncation or cyclic
repetition is performed to match a code rate which will be
transmitted to the PUSCH.
TABLE-US-00011 TABLE 11 i M.sub.i, 0 M.sub.i, 1 M.sub.i, 2 M.sub.i,
3 M.sub.i, 4 M.sub.i, 5 M.sub.i, 6 M.sub.i, 7 M.sub.i, 8 M.sub.i, 9
M.sub.i, 10 0 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 0 0 0 0 0 0 1 1 2 1 0 0
1 0 0 1 0 1 1 1 3 1 0 1 1 0 0 0 0 1 0 1 4 1 1 1 1 0 0 0 1 0 0 1 5 1
1 0 0 1 0 1 1 1 0 1 6 1 0 1 0 1 0 1 0 1 1 1 7 1 0 0 1 1 0 0 1 1 0 1
8 1 1 0 1 1 0 0 1 0 1 1 9 1 0 1 1 1 0 1 0 0 1 1 10 1 0 1 0 0 1 1 1
0 1 1 11 1 1 1 0 0 1 1 0 1 0 1 12 1 0 0 1 0 1 0 1 1 1 1 13 1 1 0 1
0 1 0 1 0 1 1 14 1 0 0 0 1 1 0 1 0 0 1 15 1 1 0 0 1 1 1 1 0 1 1 16
1 1 1 0 1 1 1 0 0 1 0 17 1 0 0 1 1 1 0 0 1 0 0 18 1 1 0 1 1 1 1 1 0
0 0 19 1 0 0 0 0 1 1 0 0 0 0 20 1 0 1 0 0 0 1 0 0 0 1 21 1 1 0 1 0
0 0 0 0 1 1 22 1 0 0 0 1 0 0 1 1 0 1 23 1 1 1 0 1 0 0 0 1 1 1 24 1
1 1 1 1 0 1 1 1 1 0 25 1 1 0 0 0 1 1 1 0 0 1 26 1 0 1 1 0 1 0 0 1 1
0 27 1 1 1 1 0 1 0 1 1 1 0 28 1 0 1 0 1 1 1 0 1 0 0 29 1 0 1 1 1 1
1 1 1 0 0 30 1 1 1 1 1 1 1 1 1 1 1 31 1 0 0 0 0 0 0 0 0 0 0
In the LTE-A system, a PUCCH format 3 has been introduced to
transmit UCI (A/N and SR) bits of maximum 21 bits, and in the
status of the normal CP, the UE may transmit code bits of 48 bits
by using the PUCCH format 3. Therefore, when the number of UCI bits
is 11 bits or less, (32, A) RM coding is used, and in this case,
cyclic repetition of code bits is used to correspond to code bits
required by the PUCCH format 3. If the number of UCI bits exceeds
11 bits, the number of (32, A) RM code based sequences in Table 11
is not sufficient, whereby two code bits are generated using two
(32, A) RM coding blocks as illustrated in FIG. 18 (this case will
be referred to as Dual RM), and the other bits are transmitted by
truncation and interleaving to reduce the two code bits to
correspond to the number of code bits of the PUCCH format 3.
In the case that the UCI of maximum 21 bits is transmitted to the
PUSCH, truncation or cyclic repetition is performed using (32, A)
RM coding in the same manner as the legacy Rel-8 system to match a
code rate which will be transmitted to the PUSCH when the number of
UCI bits is 11 bits or less, whereas two coded bits are generated
using Dual RM when the number of UCI bits exceeds 11 bits, and
truncation or cyclic repetition is performed for the two coded bits
to match the code rate which will be transmitted to the PUSCH.
Referring to FIG. 18, when the number of input UCI bits corresponds
to 21 bits, the transmitter generates a part 1 and a part 2 by
dividing the corresponding UCI bits. Afterwards, the transmitter
applies (32, A) RM coding to each of the part 1 and the part 2, and
truncates or cyclically repeats the code bits to match with 48 bits
that can be transmitted by the PUCCH format 3. Then, the
transmitter interleaves or concatenates code bits which are output,
whereby the code bits can be transmitted through the PUCCH format
3.
In more detail, a bit configuration order per UCI will be
described. If the PUCCH format 3 is configured to be used for an SR
transport subframe, when SR and A/N are transmitted to the PUCCH
format 3 or the PUSCH, the A/N is first arranged and then the SR is
arranged next to the A/N, whereby UCI bits are configured.
FIG. 19 is a diagram illustrating a method for interleaving output
code bits when dual RM described in FIG. 18 is used. Referring to
FIG. 19, when data blocks (that is, UCI) of lengths A and B are
respectively input to (32, A) and (32, B) RM encoders, the output
code bits are subjected to rate matching of 24 bits to become A0,
A1 , . . . , A23 and B0, B1, . . . , B23.
The code bits A0, A1, . . . , A23 and B0, B1 , . . . , B23 are
input to an interleaver, and the code bits output from the
interleaver are output in pairs in due order to generate bit
streams of A0, A1, B0, B1, A2, A3, B2, B3, . . . , A22, A23, B22
and B23. The bit streams are QPSK modulated and transmitted in
accordance with a PUCCH format 3 transport format, wherein the
first 24 bits (12 QPSK symbols) of the bit streams are mapped into
a first slot and the other 24 bits (12 QPSK symbols) are mapped
into a second slot.
3. Polar Coding
A polar code is known as a channel code that can obtain channel
capacity in a B-DMC (Binary-input Discrete Memory less Channel).
That is, the polar code is a channel code that can obtain channel
capacity having no error when a size N of a code block becomes
large infinitely. An encoder of the polar code can perform a
channel combining procedure and a channel splitting procedure.
The channel combining procedure is to concatenate B-DMCs in
parallel and determine the size of the code block. FIG. 20 is a
diagram illustrating a first level channel combining procedure
performed by polar coding. That is, FIG. 20 illustrates that two W
which are B-DMCs are combined with each other.
In this case, u.sub.1 and u.sub.2 are binary-input source bits, and
y.sub.1 and y.sub.2 are output coded bits. At this time, it is
assumed that an entire equivalent channel is W.sub.2. When N number
of B-DMCs are subjected to channel combining, the respective
channels which are combined can be expressed in a recursive format.
That is, when x.sub.1.sup.N=u.sub.1.sup.NG.sub.N,
x.sub.1.sup.N={x.sub.1, . . . , x.sub.N}, and
u.sub.1.sup.N={u.sub.1, . . . , u.sub.N}, a generator matrix
G.sub.N can be calculated as expressed by the following Equation
4.
.times..times..times. ##EQU00002##
In the Equation 4, R.sub.N indicates a bit-reversal interleaver,
and performs a mapping operation for an input bit s.sub.1.sup.N
into an output bit x.sub.1.sup.N=(s.sub.1, s.sub.3, . . . ,
s.sub.N-1, s.sub.2, . . . , s.sub.N. This relation is shown in FIG.
21. FIG. 21 is a diagram illustrating an Nth level channel
combining procedure performed by polar coding, wherein a size N of
a code block has a limit of 2.sup.n (n is a natural number).
After N number of B-DMCs are subjected to channel combining, a
procedure of defining an equivalent channel for a specific input
can be defined as a channel splitting procedure. The channel
splitting procedure can be expressed as channel transition
probability as expressed by the following Equation 5.
.function..times..times..function..times..times. ##EQU00003##
After the aforementioned channel combining procedure and the
aforementioned channel splitting procedure are performed, a theory
as disclosed in Table 12 below can be derived.
TABLE-US-00012 TABLE 12 Theorem: For any B-DMC W, the channels
{W.sub.N.sup.(i)} polarize in the sense that, for any fixed .delta.
.di-elect cons. (0, 1), as N goes to infinity through powers of
two, the fraction of indices i .di-elect cons. {1, . . . , N} for
which I(W.sub.N.sup.(i)) .di-elect cons. (1 - .delta., 1] goes to
I(W) and the fraction for which I(W.sub.N.sup.(i)) .di-elect cons.
[0, .delta.) goes to 1 - I(W). Hence, as N .fwdarw. .infin.,
channels polarize, either com- pletely noisy or noise free and we
know these channels exactly at the transmitter. So, we fix bad
channels and transmit uncoded bits over good ones.
The theory derived in Table 12 is as follows. If the size N of the
code block becomes great infinitely, the equivalent channel for the
specific input bit is categorized into a noise channel having an
error and a noise free channel having no error. This means that
capacity of the equivalent channel for the specific input bit is
categorized into 0 or I (W).
One of methods for decoding a polar code is a successive
cancellation (SC) decoding method. The SC decoding method is to
obtain a channel transition probability and calculate a likelihood
ratio (LLR) for the input bit based on the channel transition
probability. At this time, the channel transition probability can
be calculated in a recursive format using a recursive property of
the channel combining procedure and the channel splitting
procedure.
Therefore, LLR value can finally be calculated in a recursive
format. First of all, W.sub.N.sup.(i)(y.sub.1.sup.N,
u.sub.1.sup.i-1|u.sub.i) which is a channel transition probability
for the input bit u.sub.i can be obtained through the following
Equations 6 and 7. In this case, u.sub.1.sup.i can be defined as
u.sub.1,o.sup.i, u.sub.1,e.sup.i by being split into an odd index
and an even index.
.times..times..times..times..function..times..times..times..times..times.-
.times..times..times..times..times..times..times..times..times..times..fun-
ction..times..times..times..times..times..times..times..times..times..time-
s..times..function..times..sym..times..times..function..times..times..time-
s..times..times..times..times..times..times..function..times..times..times-
..times..times..times..function..times..sym..times..times..times..times..f-
unction..times..sym..times..times..sym..times..function..times..times..tim-
es..times..times..times.
.times..function..times..times..times..function..times..times..times..tim-
es..function..times..times..times..times..times..times..times..times..time-
s..times..times..times..times..times..times..function..times..times..times-
..times..times..times..times..times..times..times..times..function..times.-
.sym..times..times..function..times..times..times..times..times..times..ti-
mes..function..times..times..times..times..times..times..function..times..-
sym..times..times..function..times..sym..times..times..sym..times..functio-
n..times..times..times..times..times. ##EQU00004##
At this time,
.times..times..function..times..times..function..times..times..sym..times-
..times..times..function..times..times..function..times..times..sym..times-
..times..function..times..times..times..times..times..times..function..tim-
es..times..function..times..times..sym..times..times..times..times..times.-
.times..function..times..times. ##EQU00005## which is LLR for the
input bit can be obtained as expressed by the following Equation
8.
.function..function. ##EQU00006##
Since arithmetic symbols used in all the Equations described in the
embodiments of the present invention are used to refer to the same
meaning as general arithmetic symbols, their detailed description
will be omitted, and may be interpreted as the same definition as
general arithmetic symbols when the corresponding Equations are
interpreted.
The polar encoder and the SC decoder have complexity of O (N log
N), which is varied depending on the length N of the code block.
When an input bit of K bits is assumed in the polar code of the
length N, a coding rate is K/N. At this time, if a generator matrix
of the polar encoder of a data payload size N is defined as
G.sub.N, encoding bits can be expressed as
x.sub.1.sup.N=u.sub.1.sup.NG.sub.N, and K number of bits of
u.sub.1.sup.N correspond to payload bits. It is assumed that a row
index of G.sub.N corresponding to the payload bits is I and a row
index of G.sub.N corresponding to the other N-K bits is F. A
minimum distance of the aforementioned polar code is defined as
expressed by the following Equation 9.
.times..times..times..times..function..di-elect
cons..times..times..function..times..times. ##EQU00007##
In the Equation 9, wt(i) means the number of `1` during binary
extension of i(i=0, 1, . . . , N-1). That is, wt(i) means the
number of `1` when an index (that is, column index I of G.sub.N) of
a channel is expressed as a binary.
3.1 Derivation of Equivalent Channel
The embodiments of the present invention suggest methods for
applying polar coding to a mobile communication system.
As described above, if the channel combining procedure and the
channel splitting procedure are performed, the equivalent channel
is categorized into a noise channel and a noise free channel. At
this time, a data payload should be transmitted to the noise free
channel. That is, the data payload should be transmitted to the
noise free equivalent channel to obtain good performance.
At this time, a method for discovering the noise free equivalent
channel can be determined by obtaining a value z(W)=.SIGMA. {square
root over (W(y|0)W(y|1))} of the equivalent channel for each input
bit. In this case, Z(W) is referred to as a Bhattacharyya
parameter. Z(W) means a value corresponding to an upper-bound of an
error probability when MAP decision is performed after a binary
input 0 or 1 is transmitted. Therefore, values of Z(W) are obtained
and then arranged in an ascending order (small order) to select the
value of Z(W) as much as a desired data payload, whereby the
corresponding data can be transmitted through the noise free
channel
Z(W) can be obtained for a BEC (Binary Erasure Channel) as
expressed by the following Equation 10.
.function..function..times..times..times..function..times..times..functio-
n..times..times..times..function..times..times..times..times..times.
##EQU00008##
When the size of the code block is 8 in case of the BEC having an
erasure probability of 0.5, the value of Z(W) is calculated as
follows using the Equation 10. Z(W)={1.00, 0.68, 0.81, 0.12, 0.88,
0.19, 0.32, 0.00}. Therefore, when the size of the data payload is
2, the data payload is transmitted through an equivalent channel 8
(Z(W)=0.00) and an equivalent channel 4 (Z(W)=0.12).
3.1.1 CRC Addition Method
The transmitter of the wireless access system transmits data by
adding CRC (Cyclic Redundancy Check) bits to a data payload to
detect an error of the data. That is, in the following embodiments,
the data payload may be used to include one or more data bits and
CRC bits unless described separately. Since CRC can detect an
error, if performance of error detection through CRC is relatively
stable, performance of error detection of the data block can be
improved. To this end, when values of Z(W) which are obtained are
listed in an ascending order (small order), CRC bit streams are
arranged in equivalent channels corresponding to a CRC length and
then data bits are arranged, whereby error detection performance
can be improved.
For example, it is assumed that 3-bit data and 2-bit CRC input in a
data payload are subjected to polar coding using a polar encoder of
a length 8 as expressed in the Equation 10. At this time, after CRC
2 bits are arranged in an equivalent channel 8 (that is, channel of
Z(W)=0.00) and an equivalent channel 4 (that is, channel of
Z(W)=0.12) which are noise free channels having the best
performance, the data payload can be transmitted through an
equivalent channel 6 (that is, channel of Z(W)=0.19, an equivalent
channel 7 (that is, channel of Z(W)=0.32) and an equivalent channel
2 (that is, channel of Z(W)=0.68) which are noise free channels
having good performance next to the best performance.
The method for calculating the value of Z(W) as expressed in the
Equation 10 is performed in the BEC. Therefore, an equivalent
channel which will transmit the data payload can be discovered in
an additive white Gaussian nose (AWGN) channel by another method.
However, even in this case, CRC allows the values of Z(W) which are
obtained to be arranged in an ascending order (small order),
whereby the equivalent channels are allocated to CRC bits as much
as the CRC length, and then the data payload can be arranged in the
other equivalent channels.
Also, as still another method, in the case that the receiver does
not assume SC decoding, reliability of the CRC bits and reliability
of the data payload are important equally. Therefore, when the
values of Z(W) are arranged in an ascending order (small order),
the data payload is allocated to the equivalent channel having good
performance and then the CRC bits can be allocated to next
equivalent channels.
It is preferable that the transmitter and the receiver know the
equivalent channel, which will be allocated to the data payload, in
accordance with the size of the data payload including CRC bits,
and a coding rate. It is also preferable that the equivalent
channel which will be allocated to the data payload is previously
calculated by the transmitter.
Therefore, if the transmitter transmits information on the data
payload size and the coding rate to the receiver, the receiver can
perform decoding for a polar coded data signal by acquiring
information on the equivalent channel to which the data payload is
transmitted.
Also, the code block size of the polar encoder of which coding rate
is r=K/N is N and its payload size is K. In this case, a bit stream
corresponding to N-K is a bit stream allocated and transmitted to
the noise channel. It is preferable that the bit stream is
previously determined by the transmitter and the receiver. The bit
stream which will be allocated to the noise channel can be
determined as {0, 0, . . . , 0} or {1, 1, . . . , 1} corresponding
to its size N-K.
3.2 Rate Matching for Encoding Bits
The polar encoder has a size of a code block, which is limited to
2.sup.n (n is a natural number) in view of its property. Therefore,
a rate matching operation of puncturing (or truncation) or
repetition is required depending on transmission numerology of the
system. Hereinafter, the rate matching procedure for coded bits
will be described.
In the embodiments of the present invention, it is assumed that a
size of a data payload (including CRC) generated by a higher layer
satisfies a relation of 2.sup.n<N<2.sup.n+1. At this time, if
a first threshold value THR1 of a codeword size exists and thus a
size of a code bit is greater than the first threshold value, the
transmitter performs encoding through a mother polar encoder of
2.sup.n+1 size and then performs puncturing as much as 2.sup.n+1-N
bits, whereby encoding bit streams of a codeword size N are
generated. At this time, the mother polar encoder means an encoder
which is a reference for performing repetition or puncturing in
accordance with a size of the data payload.
Meanwhile, if the size of the data payload is smaller than a second
threshold value THR2, the transmitter performs encoding through a
mother polar encoder of 2.sup.n size and then generates encoding
bit streams corresponding to the code block size N through
repetition as much as N-2.sup.n bits. At this time, it is
preferable that the size of the data payload satisfies a relation
of K<2.sup.n. In this case, THR1 and THR2 may be the same as
each other or may be different from each other.
Hereinafter, methods for configuring a generator matrix for
performing rate matching by puncturing or repeating codeword bits
considering a weight value, a minimum distance and/or a priority
will be described.
3.2.1 Configuration of Generator Matrix Using Weight Value
If a generator matrix of a polar encoder of a size N for an input
data payload is G.sub.N, encoded bit streams can be expressed as
x.sub.1.sup.N=u.sub.1.sup.NG.sub.N. At this time, a number of "1"
in each column of G.sub.N can be defined as a weight value of each
column. If repetition is performed for G.sub.N by the mother polar
encoder, repetition is performed in the order of column having a
greater weight value, whereby a distance of repeated codewords can
be set to a maximum value.
Likewise, if the transmitter performs puncturing for G.sub.N by the
mother polar encoder, repetition is performed in the order of
column having a smaller weight value, whereby a distance of
punctured codewords can be set to a maximum value. In the
embodiments of the present invention, the mother polar encoder can
be used as the same meaning as a mother generator matrix. Also, the
mother generator matrix can be defined as a first generator matrix,
and a new generator matrix generated from the mother generator
matrix by repetition or puncturing can be defined as a second
generator matrix.
The following Equation 11 is an example of a generator matrix
G.sub.N of a polar encoder of which code block size is N=8.
.times..times..times..times. ##EQU00009##
The weight value of each column in the polar encoder having the
generator matrix as expressed in the Equation 11 is {8, 4, 4, 2, 4,
2, 2, 1}. Therefore, when a codeword of a codeword length C=10 is
transmitted, among respective columns of the generator matrix, the
first column having the greatest weight value and one of second,
third and fifth columns having great weight values next to the
greatest weight value are selected and then transmitted repeatedly.
That is, codeword C=8 is encoded using the generator matrix, and
the other two codewords can be transmitted by selecting the first
column and one of the second, third and fifth columns and
repeatedly encoding the selected columns. For example, when the
transmitter selects the first column and the second column, a new
generator matrix is G'.sub.8=[G.sub.8 G.sub.r],
G.sub.r=[G.sub.8(:,1) G.sub.8(:, 2)]. In this case, G.sub.8(:,x) is
a column vector indicating the xth column.
Likewise, when a codeword of a codeword length C=6 is transmitted,
the eighth column having the smallest weight value and one of
fourth, sixth and seventh columns can be punctured and encoded. At
this time, when the transmitter selects the eighth column and the
fourth column, the generator matrix G.sub.6 is as expressed by the
following Equation 12.
.times..times..times. ##EQU00010##
The Equation 12 indicates a new generator matrix (that is, the
second generator matrix) generated as the eighth column and the
fourth column are punctured in the Equation 11.
If column permutation is performed for the columns of the generator
matrix of the polar encoder in the order of greater weight values
or smaller weight values, puncturing and repetition can easily be
implemented. That is, when the columns of the first generator
matrix are rearranged in the order of column having the greater
weight value, repetition is performed in the order of column index,
and if puncturing is performed in the reverse order of column
index, the minimum distance of the codeword described in the
Equation 9 can be set to a maximum value.
The following Equation 13 represents that permutation is performed
for the generator matrix of the Equation 11 in the order of column
weight value.
.times..times..times..times. ##EQU00011##
3.2.2 Configuration of Generator Matrix Based on Minimum
Distance
Hereinafter, methods for configuring a generator matrix considering
a minimum distance of a polar encoder will be described. In the
generator matrix as expressed in the Equation 11, if the size of
the payload is 2, the payload is transmitted through an equivalent
channel corresponding to the eighth row and the fourth row (row 8,
row 4) of the Equation 11. That is, a matrix such as
##EQU00012## may be a generator sub-matrix corresponding to the
2-bit payload.
If output codewords of 6 bits are generated, the transmitter should
puncture the column corresponding to 2 bits of the polar encoder
from the mother generator matrix which is the first generator
matrix. At this time, if puncturing is performed for the column
having a small weight value, it is preferable that the column
corresponding to 0 of the first row is punctured. If the second and
fourth columns are punctured, the second generator matrix that
generates a payload of 6 bits can be configured as expressed by the
following Equation 14.
.times..times..times..times. ##EQU00013##
That is, column puncturing is performed in the order of smaller
column weight values of a sub-matrix comprised of rows of a
generator matrix corresponding to a payload.
3.2.3 Configuration of Generator Matrix Based on Priority
In another aspect of the present invention, a new generator matrix
can be configured based on a priority of a column index of a mother
generator matrix.
If the same column weight value is applied to columns of the mother
generator matrix, puncturing may be performed in such a manner that
a priority is given in the order of smaller column index or in the
order of greater column index.
If puncturing is performed in the order of greater column index,
the sixth column and the eighth column of the Equation 11 are
punctured, whereby a generator matrix is obtained as expressed by
the following Equation 15.
.times..times..times..times. ##EQU00014##
As described above, if puncturing and repetition are performed,
performance of a newly generated generator matrix may be degraded
due to the same column or row comprised of same elements.
Therefore, puncturing or repetition is performed for the column
corresponding to a priority, whereby the generator matrix can be
configured.
3.3 Method for Transmitting and Receiving Data Using Polar
Coding
Hereinafter, a method for transmitting data using polar coding will
be described based on the aforementioned embodiments of the present
invention.
FIG. 22 is a diagram illustrating an example of a procedure of
transmitting data from a transmitter through polar coding.
The transmitter can derive equivalent channels by repeatedly
performing a channel combining procedure and a channel splitting
procedure. That is, if the channel combining procedure and the
channel splitting procedure are performed, the equivalent channels
are categorized into a noise channel and a noise free channel
(S2210).
The transmitter derives the Bhattacharyya parameter Z(W) described
in the clause 3.1 in accordance with each input bit to discover the
noise free channel from the categorized equivalent channel
(S2220).
The transmitter allocates a data payload comprised of CRC bits and
data bits to the noise free channel. At this time, the transmitter
may improve receiving performance of the receiver by allocating the
CRC bits to the noise free channel better than the data bits
(S2230).
Also, it is assumed that the size of the polar encoder to which the
data payload is input to performed polar coding is N and the size
of the data payload is K. In this case, the transmitter performs
puncturing or truncation if N>k, and performs rate matching by
performing repetition if N<k. Its detailed description will be
understood with reference to the clause 3.2 (S2240).
The transmitter performs interleaving for polar coded code bits (or
code symbols) and transmits the interleaved code bits to the
receiver (S2250).
However, the receiver should know size information k indicating the
size of the data payload input to the polar encoder and a coding
rate k/N of the polar encoder to detect and decode the polar coded
code bits. Therefore, the transmitter can transmit the size
information on the size of the data payload and the coding rate
information of the polar encoder to the receiver (S2260).
The step S2260 may be performed if the transmitter receives data
from the higher layer, or may be performed after the data payload
is input to the polar encoder.
4. Method for Performing HARQ Using Polar Code
If polar coding is used, it is preferable that IR (Incremental
Redundancy)-HARQ scheme to obtain HARQ gain. The IR scheme is a
retransmission scheme for obtaining maximum coding gain by setting
a redundancy version (RV) during retransmission differently from
that during initial transmission.
For example, it is assumed that the data payload size N is not
changed during retransmission. Then, it is preferable to set
maximum polar coding gain by setting puncturing and repetition
patterns of the polar encoder during retransmission. Hereinafter, a
method for performing HARQ when polar coding is used will be
described.
4.1 Configuration of Generator Matrix Based on Puncturing
Hereinafter, a case where transmission of the data payload is
performed through puncturing during retransmission including
initial transmission will be described.
Puncturing is performed as much as the number of bits of the
generator matrix, which will be punctured in the order of the
smaller column weight value. If bit streams corresponding to all
the columns are punctured, the same puncturing pattern is performed
again. A priority may be given to columns having the same weight
value.
For example, the priority may be set in the order of column index
of the generator matrix. The generator matrix of the Equation 11 is
assumed as a mother generator matrix. At this time, puncturing is
performed in the order of the eighth column, the fourth column, the
sixth column, the seventh column, the second column, the third
column, the fifth column, and the first column considering column
index in case of the order of the smaller column weight value and
the same column weight value.
That is, if 2 bits are punctured to transmit codewords of 6 bits
during the first transmission, the transmitter transmits columns
corresponding to the eighth bit and the fourth bit in the Equation
11 by puncturing the columns. Also, if codewords of 5 bits are
transmitted during the second transmission, the transmitter can
transmit the column corresponding to 3 bits in the Equation 11 by
puncturing the corresponding column. For example, the transmitter
can configure the generator matrix by puncturing the columns
corresponding to the sixth bit, the seventh bit and the second bit.
Also, if the third retransmission is required and codewords of 4
bits are retransmitted, the transmitter can configure the generator
matrix by puncturing the column corresponding to 4 bits. For
example, the transmitter can puncture the columns corresponding to
the third bit, the fifth bit, the first bit and the eighth bit.
If the transmitter performs permutation for the columns of the
generator matrix in the order of column weight value, the
transmitter can perform puncturing in the reverse order of column
index.
4.2 Configuration of Generator Matrix Based on Repetition
Hereinafter, a case where transmission of the corresponding data
payload is performed by repetition during retransmission including
initial transmission will be described.
If the generator matrix is configured through repetition,
repetition can be performed as much as the number of repetition
bits in the order of the greater column weight value of the mother
generator matrix. If repetition is performed for bit streams
corresponding to all the columns, the same repetition pattern can
be performed again. Also, a priority may be set to the columns
having the same weight value. In this case, the priority may be set
in the order of column index of the generator matrix.
The generator matrix of the Equation 11 is assumed as a mother
generator matrix. At this time, repetition can be performed in the
order of the first column, the second column, the third column, the
fifth column, the fourth column, the sixth column, the seventh
column and the eighth column considering column index in case of
the order of the greater column weight value and the same column
weight value.
At this time, if the transmitter transmits codewords of 10 bits
during the first transmission, a new generator matrix is configured
by repetition of the column corresponding to 2 bits. If 3 bits are
repeatedly transmitted during the second transmission, the third
bit, the fifth bit and the fourth bit are repeatedly transmitted.
If 4 bits are repeatedly retransmitted during the third
transmission, repetition is performed for the columns corresponding
to the sixth bit, the seventh bit, the eighth bit and the first bit
to configure the generator matrix, whereby the data payload is
transmitted.
If the transmitter performs permutation for the columns of the
generator matrix in the order of column weight value, the
transmitter can perform repetition in the order of column
index.
4.3 Configuration of Generator Matrix Based on Puncturing and
Repetition
Hereinafter, a case where a generator matrix is configured through
puncturing and repetition during initial transmission and
retransmission will be described.
The transmitter can transmit the data payload by independently
using the clauses 4.1 and 4.2 in accordance with puncturing and
repetition patterns of previous transmission. For example, if
current transmission is performed using the repetition pattern, the
scheme described in the clause 4.2 is used considering the
repetition pattern of the previous transmission.
Also, if current transmission is performed using puncturing, the
scheme described in the clause 4.1 can be used considering the
puncturing pattern of transmission corresponding to puncturing of
the previous transmission.
If the transmitter performs permutation for the columns of the
generator matrix in the order of column weight value, the
transmitter can perform puncturing in the reverse order of column
index.
5. Apparatuses
Apparatuses illustrated in FIG. 23 are means that can implement the
methods described before with reference to FIGS. 1 to 22.
A UE may act as a transmitting end on a UL and as a receiving end
on a DL. An eNB may act as a receiving end on a UL and as a
transmitting end on a DL.
That is, each of the UE and the eNB may include a transmitter (Tx)
2340 or 2350 and a Receiver (Rx) 2360 or 2370, for controlling
transmission and reception of information, data, and/or messages,
and an antenna 2300 or 2310 for transmitting and receiving
information, data, and/or messages.
Each of the UE and the eNB may further include a processor 2320 or
2330 for implementing the afore-described embodiments of the
present disclosure and a memory 2380 or 2390 for temporarily or
permanently storing operations of the processor 2320 or 2330.
The embodiments of the present invention can be performed using the
elements and functions of the aforementioned UE and the
aforementioned eNB. For example, the processor of the UE and the
eNB can transmit encoded data by performing polar coding by means
of combination of the methods described in the aforementioned
clauses 1 to 3. Its details will be understood with reference to
the clauses 1 to 3.
The Tx and Rx of the UE and the eNB may perform a packet
modulation/demodulation function for data transmission, a
high-speed packet channel coding function, OFDMA packet scheduling,
TDD packet scheduling, and/or channelization. Each of the UE and
the eNB of FIG. 23 may further include a low-power Radio Frequency
(RF)/Intermediate Frequency (IF) module.
Meanwhile, the UE may be any of a Personal Digital Assistant (PDA),
a cellular phone, a Personal Communication Service (PCS) phone, a
Global System for Mobile (GSM) phone, a Wideband Code Division
Multiple Access (WCDMA) phone, a Mobile Broadband System (MBS)
phone, a hand-held PC, a laptop PC, a smart phone, a Multi
Mode-Multi Band (MM-MB) terminal, etc.
The smart phone is a terminal taking the advantages of both a
mobile phone and a PDA. It incorporates the functions of a PDA,
that is, scheduling and data communications such as fax
transmission and reception and Internet connection into a mobile
phone. The MB-MM terminal refers to a terminal which has a
multi-modem chip built therein and which can operate in any of a
mobile Internet system and other mobile communication systems (e.g.
CDMA 2000, WCDMA, etc.).
Embodiments of the present disclosure may be achieved by various
means, for example, hardware, firmware, software, or a combination
thereof.
In a hardware configuration, the methods according to exemplary
embodiments of the present disclosure may be achieved by one or
more Application Specific Integrated Circuits (ASICs), Digital
Signal Processors (DSPs), Digital Signal Processing Devices
(DSPDs), Programmable Logic Devices (PLDs), Field Programmable Gate
Arrays (FPGAs), processors, controllers, microcontrollers,
microprocessors, etc.
In a firmware or software configuration, the methods according to
the embodiments of the present disclosure may be implemented in the
form of a module, a procedure, a function, etc. performing the
above-described functions or operations. A software code may be
stored in the memory 2380 or 2390 and executed by the processor
2320 or 2330. The memory is located at the interior or exterior of
the processor and may transmit and receive data to and from the
processor via various known means.
Those skilled in the art will appreciate that the present
disclosure may be carried out in other specific ways than those set
forth herein without departing from the spirit and essential
characteristics of the present disclosure. The above embodiments
are therefore to be construed in all aspects as illustrative and
not restrictive. The scope of the invention should be determined by
the appended claims and their legal equivalents, not by the above
description, and all changes coming within the meaning and
equivalency range of the appended claims are intended to be
embraced therein. It is obvious to those skilled in the art that
claims that are not explicitly cited in each other in the appended
claims may be presented in combination as an embodiment of the
present disclosure or included as a new claim by a subsequent
amendment after the application is filed.
INDUSTRIAL APPLICABILITY
The present disclosure is applicable to various wireless access
systems including a 3GPP system, a 3GPP2 system, and/or an IEEE
802.xx system. Besides these wireless access systems, the
embodiments of the present disclosure are applicable to all
technical fields in which the wireless access systems find their
applications.
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