U.S. patent number RE40,036 [Application Number 09/925,942] was granted by the patent office on 2008-01-29 for surface acoustic wave filter.
This patent grant is currently assigned to Fujitsu Limited. Invention is credited to Osamu Ikata, Takashi Matsuda, Tsutomu Miyashita, Yoshio Satoh, Mitsuo Takamatsu.
United States Patent |
RE40,036 |
Satoh , et al. |
January 29, 2008 |
**Please see images for:
( Certificate of Correction ) ** |
Surface acoustic wave filter
Abstract
A SAW filter includes a first SAW resonator having a pair of
terminals and a predetermined resonance frequency (f.sub.rp), the
first SAW resonator being provided in a parallel arm of the SAW
filter. A second SAW resonator has a pair of terminals and a
predetermined resonance frequency (f.sub.rs) approximately equal to
a predetermined antiresonance frequency of the first SAW resonator
(f.sub.ap). The second SAW resonator is provided in a series arm of
the SAW filter. An inductance element is connected in series to the
first SAW resonator.
Inventors: |
Satoh; Yoshio (Kawasaki,
JP), Ikata; Osamu (Kawasaki, JP),
Miyashita; Tsutomu (Kawasaki, JP), Matsuda;
Takashi (Kawasaki, JP), Takamatsu; Mitsuo
(Kawasaki, JP) |
Assignee: |
Fujitsu Limited (Kawasaki,
JP)
|
Family
ID: |
27459581 |
Appl.
No.: |
09/925,942 |
Filed: |
August 10, 2001 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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09314943 |
May 20, 1999 |
Re. 37390 |
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|
07965774 |
Oct 23, 1992 |
5559481 |
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Reissue of: |
08369492 |
Jan 6, 1995 |
05631612 |
May 20, 1997 |
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Foreign Application Priority Data
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Oct 28, 1991 [JP] |
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3-281694 |
Feb 19, 1992 [JP] |
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4-032270 |
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Current U.S.
Class: |
333/193;
310/313R; 333/133 |
Current CPC
Class: |
H03H
9/6483 (20130101); H03H 2250/00 (20130101) |
Current International
Class: |
H03H
9/00 (20060101); H03H 9/64 (20060101); H03H
9/72 (20060101) |
Field of
Search: |
;333/193-196,133
;310/313R,313B,313C,313D |
References Cited
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JP |
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JP |
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5167388 |
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Jul 1993 |
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JP |
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5-206778 |
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Aug 1993 |
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JP |
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|
Primary Examiner: Summons; Barbara
Attorney, Agent or Firm: Staas & Halsey LLP
Parent Case Text
.Iadd.CROSS REFERENCE TO RELATED APPLICATIONS.Iaddend.
.[.This application is a continuation of application No.
07/965,774, filed Oct. 23, 1992, now U.S. Pat. No. 5,559,481,
patented Sep. 24, 1996..]. .Iadd.This application and application
Ser. No. 09/314,943, filed May 20, 1999 (now U.S. Pat. No. RE
37,790), are each reissues of U.S. Pat. No. 5,631,612(application
Ser. No. 08/369,492, filed Jan. 6, 1995). This application is a
continuation of application Ser. No. 09/314,943, filed May 20,
1999, now U.S. Pat. No. RE 37,390, the contents of which are hereby
incorporated by reference, which is a reissue of U.S. Pat. No.
5,631,612(application Ser. No. 08/369,492, filed Jan. 6, 1995),
which is a continuation of application Ser. No. 07/965,774, filed
Oct. 23, 1992, now U.S. Pat. No. 5,559,481. This application is
related to application Ser. No. 09/158,074, filed Sep. 22, 1998,
now U.S. Pat. No. RE 37,375, which is a reissue of U.S. Pat. No.
5,559,481..Iaddend.
Claims
What is claimed is:
1. A band-pass filter having a pair of band-pass filter .[.input.].
.Iadd.common signal .Iaddend.terminals and plural pairs of
band-pass filter .[.output.]. .Iadd.signal .Iaddend.terminals,
comprising: a pair of .[.SAW.]. .Iadd.band-pass .Iaddend.filters
having respective pass bands and comprising a plurality of
.[.one-port SAW.]. .Iadd.acoustic wave .Iaddend.resonators
connected in a .Iadd.multiple .Iaddend.ladder structure, each
having at least a first stage located at a side of the pair of
band-pass filter .[.input.]. .Iadd.common signal .Iaddend.terminals
and .[.a series-arm resonator located at the first stage,.]. a pair
of input terminals and a pair of output terminals; the pair of
band-pass filter .[.input.]. .Iadd.common signal .Iaddend.terminals
being commonly connected to the .[.respective pairs of input
terminals of the pair.]. .Iadd.pair .Iaddend.of .[.SAW.].
.Iadd.band-pass .Iaddend.filters; the plurality of pairs of
band-pass filter .[.output.]. .Iadd.signal .Iaddend.terminals being
.Iadd.respectively .Iaddend.connected to the .[.respective pairs of
output terminals of the pair.]. .Iadd.pair .Iaddend.of .[.SAW.].
.Iadd.band-pass .Iaddend.filters; and an inductance element located
between .[.at least.]. .Iadd.one side of only .Iaddend.one of the
.[.SAW.]. .Iadd.band-pass .Iaddend.filters located at the first
stage and .[.the pair of band-pass filter input terminals and
directly connected between the respective pair of input terminals
of the at least one of the SAW filters and thereby in parallel to
said at least one of the SAW filters.]. .Iadd.one of the common
signal terminals, and no inductance element being located between
the other of the band-pass filters and one of the common signal
terminals.Iaddend..
.[.2. A SAW filter comprising: a plurality of first SAW resonators,
each having a pair of terminals and a predetermined resonance
frequency (f.sub.rp), said first SAW resonators being connected in
respective, parallel arms of the SAW filter; a plurality of second
SAW resonators, each having a pair of terminals and a predetermined
resonance frequency (f.sub.rs) approximately equal to an
antiresonance frequency (f.sub.ap) of each of the first SAW
resonators, said second SAW resonators being provided in series
arms of the SAW filter; and inductance elements respectively
connected in series with the first SAW resonators in the parallel
arms and formed of wires..].
.[.3. The SAW filter as claimed in claim 2, further comprising: a
package accommodating the first and second resonators and the
inductance elements; and lead terminals extending from interiorly
of the package to exteriorly thereof, said wires of the inductance
elements being bonded to said lead terminals..].
.[.4. A band-pass filter having a predetermined pass-band
characteristic and comprising: a plurality of SAW resonators
connected in a ladder formation, said plurality of resonators being
connected in respective serial arms and parallel arms; and bonding
inductance elements, said parallel arms of said ladder formation
being connected to ground via respective said bonding inductance
elements..].
.[.5. The band-pass filter as claimed in claim 4, wherein said
bonding inductance elements comprise wires..].
.[.6. A band-pass filter having a pair of band-pass filter input
terminals and plural pairs of band-pass filter output terminals,
comprising: a pair of SAW filters having respective, different pass
bands and each SAW filter having a pair of SAW filter input
terminals and a pair of SAW filter output terminals and comprising
a plurality of one-port SAW resonators connected in a ladder
structure between the input and output terminals and including at
least a first stage having a series-arm SAW resonator connected to
one of the pair of input terminals; a pair of SAW filters having
respective pass bands and comprising a plurality of one-port SAW
resonators connected in a ladder structure, each having at least a
first stage located at a side of the pair of band-pass filter input
terminals and a series-arm resonator located at the first stage, a
pair of input terminals and a pair of output terminals; the pair of
band-pass filter input terminals being commonly connected to the
respective pairs of input terminals of the pair of SAW filters; the
plurality of pairs of band-pass filter output terminals being
connected to the respective pairs of output terminals of the pair
of SAW filters..].
.[.7. A band-pass filter having a predetermined pass-band
characteristic and comprising: a plurality of SAW resonators
connected in a ladder configuration of respective serial arms and
parallel arms, said plurality of SAW resonators being connected in
respective said serial arms and parallel arms; and bonding
inductance elements respectively connecting said parallel arms of
said ladder configuration to ground..].
.Iadd.8. An acoustic wave filter comprising: a first acoustic wave
resonator having a pair of terminals and a predetermined resonance
frequency (frp), said first acoustic wave resonator being provided
in a parallel arm of the acoustic wave filter on a LiTaO.sub.3
substrate; and a second acoustic wave resonator having a pair of
terminals and a predetermined resonance frequency (frs)
approximately equal to a predetermined antiresonance frequency of
the first acoustic wave resonator (fap), said second acoustic wave
resonator being provided in a series arm of the acoustic wave
filter on the LiTaO.sub.3 substrate; and an inductance element
connected in series with the first acoustic wave resonator in the
parallel arm, the inductance element functioning to increase the
admittance of the parallel arm and decrease the resonance
frequency, wherein the first acoustic wave resonator comprises an
exciting interdigital electrode and first and second reflectors,
each of which comprises either aluminum or an aluminum alloy
containing a few weight percentage of metal, other than aluminum;
and the respective film thicknesses of the exciting interdigital
electrode and the first and second reflectors are in a range of
from 0.06 to 0.09 times the period of the exciting interdigital
electrode..Iaddend.
.Iadd.9. An acoustic wave filter comprising: a first acoustic wave
resonator having a pair of terminals and a predetermined resonance
frequency (frp), said first acoustic wave resonator being provided
in a parallel arm of the acoustic wave filter on a LiTaO.sub.3
substrate; and a second acoustic wave resonator having a pair of
terminals and a predetermined resonance frequency (frs)
approximately equal to a predetermined antiresonance frequency of
the first acoustic wave resonator (fap), said second acoustic wave
resonator being provided in a series arm of the acoustic wave
filter on the LiTaO.sub.3 substrate; and an inductance element
connected in series with the first acoustic wave resonator in the
parallel arm, the inductance element functioning to increase the
admittance of the parallel arm and decrease the resonance
frequency, wherein the first acoustic wave resonator comprises an
exciting interdigital electrode and first and second reflectors,
each of which comprises either gold or a gold alloy containing a
few weight percentage of metal other than gold; and the respective
film thicknesses of the exciting interdigital electrode and the
first and second reflectors are in a range of from 0.0086 to 0.013
times the period of the exciting interdigital
electrode..Iaddend.
.Iadd.10. An acoustic wave filter comprising: a plurality of first
acoustic wave resonators on a single piezoelectric substrate, each
having a pair of terminals and a predetermined resonance frequency
(frp), said first acoustic wave resonators being connected in
respective, parallel arms of the acoustic wave filter; a plurality
of second acoustic wave resonators on the piezoelectric substrate,
each having a pair of terminals and a predetermined resonance
frequency (frs) approximately equal to the predetermined
antiresonance frequency of the first acoustic wave resonator (fap),
said second acoustic wave resonators being provided in a series arm
of the acoustic wave filter; and inductance elements respectively
connected to ground in series with the first acoustic wave
resonators in the parallel arms..Iaddend.
.Iadd.11. A band-pass filter having a pair of band-pass filter
common signal terminals and plural pairs of band-pass filter signal
terminals, comprising: a first band-pass filter having a pass band,
having a band center frequency and comprising a plurality of
acoustic wave resonators connected in a multiple ladder structure,
having at least a first stage located at a side of the pair of
band-pass filter common signal terminals, a pair of input terminals
and a pair of output terminals; a second band-pass filter having a
different pass band from the pass band of the first band-pass
filter, having a band center frequency which is larger than the
band center frequency of the first band-pass filter and comprising
a plurality of acoustic wave resonators connected in a multiple
ladder structure, having at least a first stage located at a side
of the pair of band-pass filter common signal terminals, a pair of
input terminals and a pair of output terminals; the pair of
band-pass filter common signal terminals being commonly connected
to the first and second band-pass filters; the plurality of pairs
of band-pass filter signal terminals being respectively connected
to the first and second band-pass filters; and only one impedance
matching circuit located only between the first stage of the second
band-pass filter and the common signal terminals..Iaddend.
.Iadd.12. The band-pass filter as claimed in claim 11, wherein the
impedance matching circuit includes an inductor..Iaddend.
.Iadd.13. The band-pass filter as claimed in claim 12, wherein the
inductor is formed with a metallic strip line..Iaddend.
.Iadd.14. The band-pass filter as claimed in claim 13, wherein the
metallic strip line is formed on a ceramic package..Iaddend.
.Iadd.15. The band-pass filter as claimed in claim 11, wherein said
impedance matching circuit includes an inductor and a
capacitor..Iaddend.
.Iadd.16. A band-pass filter comprising: a first band-pass filter
having a pass band, having a band center frequency and comprising a
plurality of acoustic wave resonators connected in a multiple
ladder structure, having at least a first stage and a series-arm
resonator located at the first stage, a pair of input terminals and
a pair of output terminals; a second band-pass filter having a
different pass band from the pass band of the first band-pass
filter, having a band center frequency which is larger than the
band center frequency of the first band-pass filter and comprising
a plurality of acoustic wave resonators connected in a multiple
ladder structure, having at least a first stage and a parallel-arm
resonator located at the first stage, a pair of input terminals and
a pair of output terminals; a pair of band-pass filter common
signal terminals commonly connected to the first and second
band-pass filters; a plurality of pairs of band-pass filter signal
terminals respectively connected to the first and second band-pass
filters; a circuit element used for phase rotation and connected
between at least one of the pair of common signal terminals and the
second band-pass filter..Iaddend.
.Iadd.17. The band-pass filter as claimed in claim 16, wherein the
circuit element comprises a line formed on a glass-epoxy substrate
or a ceramic substrate..Iaddend.
.Iadd.18. The band-pass filter as claimed in claim 16, wherein the
circuit element comprises an inductance element..Iaddend.
.Iadd.19. The band-pass filter as claimed in claim 18, wherein the
circuit element further comprises a capacitance element coupled to
the inductance element..Iaddend.
.Iadd.20. A band-pass filter having a predetermined pass-band
characteristic and comprising: a plurality of acoustic wave
resonators connected in a ladder formation, said plurality of
resonators being connected in respective serial arms and parallel
arms; and bonding inductance elements, said parallel arms of said
ladder formation being connected to ground via respective said
bonding inductance elements, wherein: a package in which the
band-pass filter is mounted, contains a piezoelectric substrate and
the ground; and the plurality of acoustic wave resonators are on
the piezoelectric substrate..Iaddend.
.Iadd.21. A band-pass filter having a predetermined pass-band
characteristic and comprising: a plurality of acoustic wave
resonators connected in a ladder formation, said plurality of
resonators being connected in respective serial arms and parallel
arms; and bonding inductance elements, said parallel arms of said
ladder formation being connected to ground via respective said
bonding inductance elements, wherein: a package in which the
band-pass filter is mounted contains a piezoelectric substrate; the
plurality of acoustic wave resonators are on the piezoelectric
substrate; and a first electric resistance (Rs) of an interdigital
electrode of a acoustic wave resonator provided in a series arm, is
smaller than a second electric resistance (Rp) of an interdigital
electrode of a acoustic wave resonator provided in a parallel arm
which is next to the series arm..Iaddend.
.Iadd.22. A band-pass filter having a predetermined pass-band
characteristic and comprising: a plurality of acoustic wave
resonators connected in a ladder formation, said plurality of
resonators being connected in respective serial arms and parallel
arms; and bonding inductance elements, said parallel arms of said
ladder formation being connected to ground via respective said
bonding inductance elements, wherein: the plurality of acoustic
wave resonators are on a piezoelectric substrate; and the bonding
inductance elements are respectively connected to the ground
outside the piezoelectric substrate..Iaddend.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention generally relates to surface acoustic wave
(SAW) filters, and more particularly to a ladder-type SAW filter
suitable for an RF (Radio Frequency) filter provided in pocket and
mobile telephones, such as automobile phone sets and portable
phones.
2. Description of the Prior Art
In Japan, an automobile phone or portable phone system has a
specification in which a transmission frequency band is .+-.8.5 MHz
about a center frequency of 933.5 MHz. The ratio of the above
transmission band to the center frequency is approximately 2%.
Recently, SAW filters have been employed in automobile phone or
portable phone systems. It is required that the SAW filters have
characteristics which satisfy the above specification. More
specifically, it is required that the pass band width is so broad
that 1) the ratio of the pass band to the center frequency is equal
to or greater than 2%, 2) the insertion loss is small and equal to
5 dB-2 dB, and 3) the suppression factor is high and equal to 20
dB-30 dB.
In order to satisfy the above requirements, SAW filters are
substituted for conventional transversal filters. Generally, SAW
elements are so connected that a ladder-type filter serving as a
resonator is formed.
FIG. 1 is an equivalent circuit of a SAW filter disclosed in
Japanese Laid-Open Patent Publication No. 52-19044. A SAW filter 1
shown in FIG. 1 comprises a SAW resonator 3 in a series arm 2, and
a SAW resonator 5 in a parallel arm 4. The equivalent parallel
capacitance C.sub.OB of the resonator 5 in the parallel arm 4 is
larger than the equivalent parallel capacitance C.sub.OA of the
resonator 3 in the series arm 2.
The SAW filter 1 shown in FIG. 1 has a characteristic shown in FIG.
2. A curve 6 shows an attenuation quantity v. frequency
characteristic of the SAW filter 1. As indicated by arrows 7 shown
in FIG. 2, the suppression factor increases as the equivalent
parallel capacitance C.sub.OB increases. However, as the equivalent
parallel capacitance C.sub.OB increases, the band width decreases,
as indicated by arrows 8, and the insertion loss increases, as
indicated by an arrow 9. Hence, the characteristic deteriorates, as
indicated by a broken line 10. When trying to obtain a suppression
factor equal to or larger than 20 dB, the band width is decreased
to that the ratio of the pass band to the center frequency is equal
to or smaller than 1%, and does not satisfy the aforementioned
specification of the 800 MHz-band radio systems.
SUMMARY OF THE INVENTION
It is a general object of the present invention to provide a SAW
filter in which the above disadvantages are eliminated.
A more specific object of the present invention is to provide a SAW
filter having a large band width, a large suppression factor, and a
small insertion loss.
The above objects of the present invention are achieved by a SAW
filter comprising: a first SAW resonator (21, R1A, R1B) having a
pair of terminals and a predetermined resonance frequency
(f.sub.rp), the first SAW resonator being provided in a parallel
arm (24) of the SAW filter; a second SAW resonator (23) having a
pair of terminals and a predetermined resonance frequency
(f.sub.rs) approximately equal to the predetermined antiresonance
frequency of the first SAW resonator (f.sub.ap), the second SAW
resonator being provided in a series arm (24) of the SAW filter;
and an inductance element (25, L1) connected in series to the first
SAW resonator.
BRIEF DESCRIPTION OF THE DRAWINGS
Other objects, features and advantages of the present invention
will become more apparent from the following detailed description
when read in conjunction with the accompanying drawings, in
which:
FIG. 1 is an equivalent circuit diagram of a conventional SAW
filter;
FIG. 2 is a graph of a characteristic of the conventional SAW
filter shown in FIG. 1;
FIG. 3 is a circuit diagram of a SAW filter according to the
present invention;
FIG. 4 is a block diagram of the basic structure of a filter
circuit using a resonator;
FIGS. 5A, 5B and 5C are diagrams showing a one-terminal-pair SAW
resonator;
FIGS. 6A and 6B are diagrams showing frequency characteristics of
impedance and admittance of the one-terminal-pair SAW
resonator;
FIGS. 7A to 7C are diagrams showing an immittance characteristic of
a SAW resonator and a filter characteristic of the filter shown in
FIG. 3 using that SAW resonator;
FIGS. 8A to 8C are diagrams showing the characteristics of the
conventional SAW filter shown in FIG. 1;
FIGS. 9A and 9B are diagrams showing effects obtained when an
inductance is connected in series to a resonator;
FIGS. 10A and 10B are diagrams showing effects obtained when n
one-terminal-pair resonators are connected in series;
FIGS. 11A and 11B are diagrams showing an aperture length
dependence on a parallel-arm resonator;
FIGS. 12A to 12C are diagrams showing an aperture length dependence
on a series-arm resonator;
FIG. 13 is a circuit diagram of a SAW filter according to a first
embodiment of the present invention;
FIG. 14 is a diagram showing a band characteristic of the filter
shown in FIG. 13;
FIGS. 15A and 15B are diagrams showing effects obtained when an
inductance is added to a parallel-arm resonator;
FIG. 16 is a plan view of the structure of the SAW filter shown in
FIG. 13 with a lid removed therefrom;
FIG. 17 is a cross-sectional view taken along a line XVII--XVII
shown in FIG. 16;
FIG. 18 is a diagram of a SAW filter according to a second
embodiment of the present invention;
FIG. 19 is a diagram showing a band characteristic of the filter
shown in FIG. 18;
FIGS. 20A and 20B are diagrams showing effects based on the ratio
of the aperture length of the parallel-arm resonator to the
aperture length of the series-arm resonator;
FIG. 21 is a diagram of a SAW filter according to a third
embodiment of the present invention;
FIG. 22 is a diagram showing a band characteristic of the filter
shown in FIG. 21;
FIG. 23 is a diagram of a SAW filter according to a fourth
embodiment of the present invention;
FIG. 24 is a diagram showing a band characteristic of the filter
shown in FIG. 23;
FIG. 25 is a circuit diagram of a SAW filter according to a fifth
embodiment of the present invention;
FIG. 26 is a diagram showing a band characteristic of the filter
shown in FIG. 25;
FIG. 27 is a circuit diagram of a SAW filter according to a sixth
embodiment of the present invention;
FIG. 28 is a diagram showing a first one-terminal-pair SAW
resonator shown in FIG. 27;
FIG. 29 is a diagram showing a band characteristic of the filter
shown in FIG. 27;
FIG. 30 is a diagram showing the influence of the reflector setting
position on the width of a ripple;
FIG. 31 is a plan view of the structure of the SAW filter shown in
FIG. 27 with a lid removed therefrom;
FIG. 32 is a diagram showing a variation of the first
one-terminal-pair SAW resonator shown in FIG. 27;
FIG. 33 is a diagram showing another variation of the first
one-terminal-pair SAW resonator shown in FIG. 27;
FIG. 34 is a circuit diagram of a SAW filter according to a seventh
embodiment of the present invention;
FIG. 35 is a diagram showing the relation between the film
thickness of the electrode and the ripple occurrence position;
FIG. 36 is a diagram showing a state in which a ripple arising from
reflectors of a parallel-arm resonator has been dropped into a
high-frequency attenuation pole;
FIGS. 37A, 37B and 37C are diagrams showing a film thickness'
dependence on the pass band characteristic of a resonator-type
filter;
FIGS. 38A and 38B are diagrams showing the results of an experiment
concerning the film thickness' dependence on the insertion loss and
the ripple occurrence position;
FIG. 39 is a diagram of a first one-terminal-pair SAW resonator
according to an eighth embodiment of the present invention;
FIG. 40 is a diagram showing a band characteristic of the SAW
filter shown in FIG. 39;
FIG. 41 is a diagram showing a variation of the first
one-terminal-pair SAW resonator used in the eighth embodiment of
the present invention;
FIG. 42 is a plan view of a structure which realizes inductors used
in the filter shown in FIG. 13;
FIG. 43 is a diagram of another structure which realizes inductors
used in the filter shown in FIG. 13;
FIG. 44 is a circuit diagram of a SAW filter according to an
eleventh embodiment of the present invention;
FIG. 45 is a perspective view of the SAW filter shown in FIG.
44;
FIGS. 46A and 46B are diagrams showing an immittance characteristic
of a SAW resonator in which the resonance frequency is higher than
the anti-resonance frequency;
FIGS. 47A, 47B and 47C are diagrams showing variations in the band
characteristic of the ladder-type filter observed when the
difference between the resonance frequency and the antiresonance
frequency increases from zero;
FIGS. 48A and 48B are diagrams showing how to measure the
characteristics of the SAW resonator;
FIG. 49 is a graph showing admittance and immittance
characteristics of SAW resonators in the series arm and the
parallel arm;
FIG. 50 is a diagram showing the frequency dependence on the
product of bx;
FIG. 51 is a diagram showing an equivalent circuit in which a part
of the circuit shown in FIG. 44 is expressed by means of L and
C;
FIG. 52 is a diagram showing the relation between |bx.sub.max| and
.DELTA.f/f.sub.rs;
FIG. 53 is a diagram showing the relation between k.sup.2 and
.tau.;
FIG. 54 is a circuit diagram of a SAW filter according to a twelfth
embodiment of the present invention;
FIG. 55 is a perspective view of the SAW filter shown in FIG.
54;
FIG. 56 is a diagram showing a filter characteristic of the SAW
resonator shown in FIG. 53;
FIG. 57 is a diagram showing a characteristic obtained when an
output-side admittance of the filter shown in FIG. 64 is
reduced;
FIGS. 58A and 58B are circuit diagrams of unit sections;
FIGS. 59A, 59B and 59C are circuit diagrams showing
multi-connections of unit sections;
FIG. 60 is a diagram showing a connection of two four-terminal
circuits and an interface therebetween;
FIGS. 61A, 61B and 61C are circuit diagrams showing unit section
connecting ways;
FIG. 62 is a diagram showing how n unit sections are cascaded;
FIGS. 63A, 63B and 63C are circuit diagrams showing how ladder-type
circuits are configured using the unit sections;
FIG. 64 is a circuit diagram of a .[.conventional.].
.Iadd.comparative example of the .Iaddend.SAW filter;
FIG. 65 is a circuit diagram of a SAW filter according to a
thirteenth embodiment of the present invention;
FIG. 66 is a circuit diagram of a SAW filter according to a
fourteenth embodiment of the present invention;
FIG. 67 is a diagram showing a SAW filter according to a fifteenth
embodiment of the present invention;
FIG. 68 is a perspective view of the SAW filter shown in FIG.
57;
FIG. 69 is a diagram showing a filter characteristic of the filter
shown in FIG. 68;
FIG. 70 is a circuit diagram of a ladder-type filter in which SAW
resonators having different resonance frequencies are respectively
provided in the parallel and series arms;
FIGS. 71A and 71B are diagrams showing a frequency characteristic
of the admittance of the parallel-arm resonator and a frequency
characteristic of the impedance of the series-arm resonator;
FIG. 72 is a circuit diagram of a wave filter according to a
sixteenth embodiment of the present invention;
FIG. 73 is a Smith's chart of the wave filter shown in FIG. 72;
FIG. 74 is a circuit diagram of a wave filter according to a
seventeenth embodiment of the present invention;
FIG. 75 is a Smith's chart of the wave filter shown in FIG. 74;
FIG. 76 is a circuit diagram of a wave filter according to an
eighteenth embodiment of the present invention;
FIG. 77 is a Smith's chart of the wave filter shown in FIG. 76;
FIG. 78 is a circuit diagram of a wave filter according to a
nineteenth embodiment of the present invention; and
FIG. 79 is a Smith's chart of the wave filter shown in FIG. 78.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIG. 3 shows an overview of a SAW filter 20 according to the
present invention. The SAW filter 20 comprises a first SAW
resonator 21 having a pair of terminals, a parallel arm 22, a
second SAW resonator 23 having a pair of terminals, a series arm
24, and an inductor 25. The first resonator 21 connected to the
parallel arm 22 has a predetermined resonance frequency f.sub.rp.
The second resonator 21 connected to the series arm 24 has a
predetermined resonance frequency f.sub.rs approximately equal to
an antiresonance frequency f.sub.ap of the first resonator 21. The
inductor 25 is connected in series to the first resonator 21, and
provided in the parallel arm 22.
The principle of the SAW filter 20 will now be described. Use of
image parameters is convenient to verify whether or not a resonance
circuit has a filter characteristic. The details of image
parameters are described in the following document: Yanagisawa et
al., "Theory and Design of Filters", Sanpo Shuppan, Electronics
Sensho, pp. 192-pp. 203, 1974.
First of all, a basic ladder-type circuit having a filter
characteristic will be described with reference to FIG. 4. Two
black boxes 30 and 31 shown in FIG. 4 are respectively SAW
resonators. For the sake of simplicity, it will now be assumed that
the SAW resonators 30 and 31 are respectively reactance circuits
having no resistance, and that the impedance Z of the resonator 30
provided in the series arm is equal to jx, and the admittance Y of
the resonator 31 provided in the parallel arm is equal to jb.
According to the image parameter method, an image transfer quantity
.gamma. (a complex number) defined in the following equation has
the important meaning: .function..gamma. ##EQU00001## where V.sub.1
and I.sub.1 denote an input voltage and an input current,
respectively, and V.sub.2 and I.sub.2 denote an output voltage and
an output current, respectively. The equation (1) can be rewritten
as follows: .times. .times..gamma..function..alpha..times..beta.
##EQU00002## where A, B, C and D denote parameters of an F matrix
showing the whole circuit shown in FIG. 4. When the value expressed
by the equation (2) is an imaginary number, the two-terminal-pair
circuit shown in FIG. 4 has a pass band characteristic. With the
above value being a real number, the circuit shown in FIG. 4 has an
attenuation characteristic. The ABCD parameters can be rewritten
using the above-mentioned x and b: A=1 B=jx C=jb D=1-bx (3). Hence,
the following equation (4) can be obtained from the equation (2)
using the above ABCD parameters: .times. .times..function..gamma.
##EQU00003## When 0<bx<1, that is, when b and x have the same
sign and are small values, the entire circuit shown in FIG. 4 has a
pass band characteristic. When bx<0 or bx>1, that is, when
the b and x have different signs or the product of bx is a large
value, the circuit shown in FIG. 4 has an attenuation
characteristic.
In order to qualitatively understand the frequency characteristics
of b and x, the impedance and admittance of the SAW resonators will
not be considered.
As shown in FIG. 5A, a SAW resonator having pair of terminals
comprises an interdigital electrode 40 (see "Nikkel Electronics",
November 29, pp.76-pp.98, 1976). A reference number 41 indicates a
pair of electrodes, 42 indicates an aperture length (crossing
width), and 43 indicates an interdigital electrode period. When the
resistance of the interdigital electrode 40 is neglected, the SAW
resonator shown in FIG. 5A has an equivalent circuit 45 shown in
FIG. 8B, in which C.sub.0 denotes the electrostatic capacitance of
the interdigital electrode 40, C.sub.1 and L.sub.1 denote
equivalent constants. Hereinafter, the equivalent circuit 45 is
depicted by symbol 46 shown in FIG. 5C.
FIGS. 6A and 6B qualitatively show an impedance vs. frequency
characteristic (A) of the equivalent circuit shown in FIG. 5B, and
an admittance vs. frequency characteristic (B) thereof. The
characteristics shown in FIGS. 6A and 6B are double resonance
characteristics in which two resonance frequencies f.sub.r and
f.sub.a exist. It will be noted that a resonator having a crystal
has a double resonance characteristic. When the resonators
respectively having a double resonance characteristic are arranged
in the series and parallel arms, respectively, and an antiresonace
frequency f.sub.ap of the parallel arm is made approximately equal
to a resonance frequency f.sub.rs of the series arm, a circuit can
be configured which has a band-pass-type filter characteristic
having the center frequencies f.sub.ap and f.sub.rs. This is
because, as shown in an immittance vs. frequency characteristic
shown in FIG. 7A, the relation 0<bx<1 is satisfied in a
frequency range around the center frequency
f.sub.ap.apprxeq.f.sub.rs and that frequency range is a pass band,
while the relation bx>1 is satisifed in a frequency range
slightly away from the center frequency and the relation bx<0 is
satisfied in a frequency range far away from the center frequency,
the latter two frequency ranges serving as attenuation bands.
Hence, the SAW filter shown in FIG. 4 has a qualitative filter
charateristic 47 shown in FIG. 7B.
A description will now be given of the factors that determine the
band width in the resonator-type SAW filters. As is seen from FIGS.
7A and 7B, the band width is mainly dependent on the difference
between the resonance frequency f.sub.r and the antiresonance
frequency f.sub.a of each of the two resonators. The band width
increases as the above difference increases, while the band width
decreases as the difference decreases. The resonance frequency
f.sub.r and the antiresonance frequency f.sub.a can be determined
using the following equations, using the equivalent circuit
constants shown in FIG. 5B: .pi..times..times..tau..tau.
##EQU00004## where .tau. denotes the capacitance ratio. The ratio
of the pass band to the center frequency (.DELTA.f/f.sub.o) is
mainly dependent on the difference between f.sub.r and f.sub.a, and
is therefore expressed in the following expression, using the
equations (6) and (7):
.DELTA.f/f.sub.o=2(f.sub.a-f.sub.r)/(f.sub.a+f.sub.r)=2/(4.tau.+1)
(8).
It can be seen from the equation (8) that the capacitance ratio
.tau. is the main factor which determines the ratio of the pass
band to the center frequency. However, as set forth in Japanese
Laid-Open Patent Publication No. 52-19044, the capacitance ratio is
much dependent on the type of substrate material used for the
interdigital electrode. For example, an ST-cut crystal having a
small electromechanical coupling coefficient has a capacitance
ratio .tau. equal to or greater than 1300, while a 36.degree. Y-cut
X-propagation LiTaO.sub.3 substrate having a large
electromechanical coupling coefficient has a capacitance ratio
.tau. of approximately 15. The ratio of the pass band to the center
frequency is 0.04% for ST-cut crystal, and 3.3% for the 36.degree.
Y-cut X-propagation LiTaO.sub.3 substrate. Hence, the band width is
much dependent on the substrate material.
The band width decreases as the equivalent parallel capacitance
C.sub.OB increases in order to improve the side lobe suppression
factor according to Japanese Laid-Open Patent Publication No.
52-19044.
The above phenomenon will now be described with reference to FIGS.
8A, 8B and 8C. As is seen from the previous description of the
principle of the present invention, as the admittance value
increases while f.sub.r and f.sub.a of the parallel resonator (see,
FIG. 8C) are kept constant, the product of bx has a negative sign
and increases, as shown in FIG. 8A. However, the bx product
increases around the center frequency, and hence the range of
bx>1 increases. Hence, the pass band in which the relation
0<bx<1 stands is narrowed, and a sufficient pass band cannot
be obtained. This phenomenon is indicated by arrows in FIG. 8B.
The following two conditions must be satisfied in order to
eliminate the above disadvantages. The first condition is to
increase the difference between the resonance frequency f.sub.r and
the antiresonance frequency f.sub.a in at least one of the
resonators provided in the series and parallel arms (see FIG. 8C).
The second condition is to increase either the impedance or
admittance of the above-mentioned one of the resonators. As the
impedance or admittance increases, the side lobe attenuation
quantity increases. When the above two conditions awe satisfied,
the side lobe attenuation quantity can be improved while the pass
band is improved or prevented from being narrowed.
Regarding the first condition, it is effective to provide an
inductor L connected in series to a SAW resonator having a pair of
terminals in order to increase the difference between f.sub.r and
f.sub.a. FIGS. 9A and 9B respectively show an impedance vs.
frequency characteristic of a SAW filter in which an inductor
having an inductance of 8 nH is connected to a resonator, and an
admittance vs. frequency characteristic thereof. The parameters of
the equivalent circuits of the SAW resonators used for obtaining
the characteristics are illustrated in FIGS. 9A and 9B.
FIG. 9A shows an impedance characteristic curve 50 obtained before
the inductor L is connected to the resonator, and an impedance
characteristic curve 51 obtained after the inductor is connected
thereto. FIG. 9B shows an admittance characteristic curve 52
obtained before the inductor L is connected to the resonator, and
an admittance characteristic curve 53 obtained after the inductor L
is connected thereto.
It can be seen from FIG. 9A that the inductance L increases the
distance between the resonance frequency f.sub.r and the
antiresonance frequency f.sub.a. In the graph of FIG. 9A, the
distance is increased by approximately 30 MHz. This is because, as
shown in FIG. 9A, the inductance L functions to shift the impedance
characteristic curve of the original resonator upwards to the plus
side by .omega.L, and hence the resonance frequency f.sub.r changes
to f.sub.r'. In this case, the antiresonance frequency f.sub.a has
little variation. The admittance, which is the reciprocal of the
impedance, changes, as shown in FIG. 9B. In this case, the
resonance frequency f.sub.r changes to f.sub.r'.
Regarding the aforementioned second condition, the admittance value
increases due to the inductance L, as shown in FIG. 9B. However, as
shown in FIG. 9A, the impedance value decreases in frequencies
outside of the pass band. Hence, if the inductance L is added to
the resonator provided in the series arm, it is necessary to
provide an additional means for increasing the impedance value. The
above additional means is, for example, an arrangement in which a
plurality of identical SAW resonators are connected in series to
each other (cascaded).
FIGS. 10A and 10B show an impedance characteristic curve 56 of a
resonance arrangement in which n identical SAW resonators, each
having a pair of terminals, are cascaded. As shown in FIGS. 10A and
10B, the impedance value of the resonance arrangement having the n
cascaded resonators is n times that of the single resonator. The
resonance frequency of the resonator with the inductor L connected
thereto is f.sub.r''. That is, the difference between f.sub.r'' and
f.sub.a of the resonance arrangement with the inductor L connected
thereto is slightly smaller than the difference between f.sub.r'
and f.sub.a of a single resonator with the inductor L connected
thereto. However, the difference between f.sub.r'' and f.sub.a of
the resonance arrangement with the inductor L connected thereto is
larger than that without the inductor L. It is possible to further
increase the difference between the resonance frequency and the
antiresonance frequency by using a larger inductance L.
In order to increase the band width, it is also possible to select
the antiresonance frequency f.sub.ap of the parallel arm resonator
and the resonance frequency f.sub.rs of the series arm resonator so
that f.sub.rs>f.sub.ap. In this case, the condition bx<0
occurs around the center frequency, and hence the aforementioned
pass band condition is not satisfied. Hence, there is a possibility
that an insertion loss and a ripple may increase. However, by
controlling .DELTA.f=f.sub.vs-f.sub.ap, it is possible to
substantially suppress the increase in the insertion loss and the
ripple and to expand the increase in the pass band.
A description will now be given of embodiments of the present
invention. The embodiments which will be described are based on a
simulation. Hence, this simulation will be described first, as well
as the results of comparisons between the experimental results and
the simulation in order to show the validity of the simulation.
The equivalent circuit shown in FIG. 5B easily simulates the
characteristic of the SAW resonator having a pair of terminals,
while that equivalent circuit is not suitable for simulating, with
high accuracy, variations in the number of figure pairs, the
aperture length and the electrode thickness, and the effects of a
reflector. With the above in mind, the inventors have proposed an
improved simulation which uses a Smith's equivalent circuit model
and expands a transfer matrix to analyze the SAW resonators (see O.
Ikata et al., "1990 ULTRASONIC SYMPOSIUM Proceedings, vol. 1,
pp.83-pp.86, 1990; the disclosure of which is hereby incorporated
by reference).
FIG. 11A is a graph showing the results of the simulation
(calculation) for an arrangement in which a SAW resonator having a
pair of terminals is disposed in the parallel arm. FIG. 11B is a
graph showing the results of the experiment on an arrangement in
which a one-terminal-pair SAW resonator including an interdigital
electrode made of Al-2% Cu and having a film thickness of 1600
.ANG. is disposed in a parallel arm, and bonding wires (L=1.5 nH)
having a length of 3 mm are connected to the interdigital
electrode. It can be seen from FIGS. 11A and 11B that the
calculation values match the experiment values with respect to
variations in the resonance points (f.sub.r1, f.sub.r2, f.sub.r3)
as well as the attenuation quantities observed around the resonance
points for different aperture lengths (a=60, 150, 300 .mu.m).
FIG. 12A is a graph showing the results of the simulation for an
arrangement in which a SAW resonator having a pair of terminals is
disposed in the series arm (see, FIG. 12C). The bonding pads used
in the experiment which will be described later were slightly
large, and the simulation was carried out taking into account a
stray capacitance 0.5 pF of the bonding pads. FIG. 12B is a graph
showing the results of the experiment on an arrangement in which a
SAW resonator having a pair of terminals is disposed in the series
arm. It can be seen from FIGS. 12A, 12B and 12C that the
antiresonance frequencies f.sub.a1, f.sub.a2 and f.sub.a3 do not
depend on the aperture length and that the simulation results match
the experimental results regarding variations in the attenuation
quantity around the resonance frequencies.
Hence, it will be apparent from the above that the results of a
simulation of the filter with the combination of the resonators
disposed in the parallel and series arms match the results of the
experiment. The embodiments described below are based on the result
of simulations.
FIG. 13 shows a SAW filter 60 according to a first embodiment of
the present invention. In Japan, an automobile and portable
telephone system has a specification in which the .+-.8.5 MHz range
about a center frequency of 933.5 MHz is a transmission band for
mobile telephones and the .+-.8.5 MHz range about a center
frequency of 878.5 MHz separated from 933.5 MHz by -55 MHz is a
.[.reception.]. .Iadd.rejection .Iaddend.band. The SAW filter 60
according to the first embodiment of the present invention is
designed to be suitable for transmission filters of mobile
telephones.
As shown in FIG. 13, two one-terminal-pair SAW resonators R2 and R4
are arranged in a series arm 61, and three one-terminal-pair SAW
resonators R1, R3 and R5 are respectively arranged in parallel arms
62, 63 and 64. Inductors L1, L2 and L3 are provided in the parallel
arms 62, 63 and 64, and are connected in series to the resonators
R1, R3 and R5, respectively. Each of the resonators R1-R5 has the
interdigital electrode structure shown in FIG. 5A. The number of
finger pairs is 100, and the aperture length is 80 .mu.m. The
electrodes are made of Al-2% Cu, and are 3000 .ANG. thick. The
resonance frequencies of the resonators R1, R3 and R5 respectively
provided in the parallel arms 62, 63 and 64 are 912 MHz, and the
antiresonance frequencies thereof are 934 MHz. The resonance
frequencies of the resonators R2 and R4 respectively provided in
the series arm 61 are 934 MHz, and the antiresonance frequencies
thereof are 962 MHz. The inductors L1, L2 and L3 respectively have
an inductance L of 4 nH.
The SAW filter 60 having the above structure has a band
characteristic indicated by a curve 65 shown in FIG. 14.
Characteristic curves 66 and 67 in FIG. 14 are respectively
obtained when the inductance L is equal to 2 nH and 6 nH.
A curve 70 shown in FIG. 15A illustrates the inductance' dependence
on the band width obtained on the basis of the graphs of FIG. 14.
The band width is defined as the frequency width between the points
on the curve where the insertion loss is 3 dB greater than the
minimum value.
A curve 71 shown in FIG. 15B illustrates a side lobe suppression
factor' dependence on the inductance obtained on the basis of the
graphs of FIG. 14. It can be seen from FIG. 14 that a sufficient
suppression factor is not obtained at a frequency which is 55 MHz
lower than the center frequency when the inductance L is too large.
With the above in mind, an inductance L of 4 nH is selected. The
value of the inductance L is suitable selected in accordance with
the specification of filters.
A curve 68 in FIG. 14 shows a band characteristic of a
configuration in which L1=L2=L3=0 in FIG. 13. It can be seen from
comparison between the band characteristic (curve 65) of the first
embodiment and that (curve 68) of the conventional filter that the
filter 60 according to the first embodiment has a large pass band
width (arrow 75), a large side lobe suppression factor (arrows 76),
and a low insertion loss (arrow 77).
FIGS. 16 and 17 show a SAW filter device 80 which functions as the
SAW filter 60 shown in FIG. 13. The SAW filter device 80 comprises
a ceramic package 81, a filter chip 82, and a lid 83 serving as the
ground. The ceramic package 81 is made of alumina ceramics, and has
dimensions of 5.5 mm (length).times.4 mm (width).times.1.5 mm
(height). Electrode terminals 84.sub.-1-84.sub.-6 made of Au are
formed on the ceramic package 81. The filter chip 82 is made of
LiTaO.sub.3, and has dimensions of 2 mm (length).times.1.55 mm
(width).times.0.5 mm (thickness).
Resonators R1-R5 are arranged on the filter chip 82 so that each of
the resonators R1-R5 does not own propagation paths in common with
other resonators. Each of the resonators R1-R5 has an interdigital
electrode made of Al-2% Cu in which the number of finger pairs is
100, the aperture length is 80 .mu.m, and the film thickness is
3000 .ANG..
Further, two signal line terminals 85.sub.-1 and 85.sub.-2 for
bonding and three ground terminals 85.sub.-3, 85.sub.-4 and
85.sub.-5 for bonding are formed on the surface of the filter chip
82. Reference numbers 86.sub.-1-86.sub.-5 indicate bonding wires
made of Al or Au. The bonding wires 86.sub.-1-86.sub.-5, each
having a diameter of 25 .mu.mo, connects the terminals
84.sub.-1-84.sub.-5 and the terminals 85.sub.-1-85.sub.-5. The
bonding wires 86.sub.-1 and 86.sub.-2 respectively form parts; of
the series arms 61a and 61b. The wire 86.sub.-3 is connected
between the ground electrode terminals 84.sub.-3 and 85.sub.-3, and
the wire 86.sub.-4 is connected between the ground electrode
terminal 84.sub.-4 and 85.sub.-4. The wire 86.sub.-5 is connected
between the ground electrode terminals 84.sub.-5 and 85.sub.-5. The
wires 86.sub.-3-86.sub.-5 are long and, for example, 2.0 mm
long.
According to the theory of high frequencies, a fine, long wire has
an inductance component. According to the theoretical equation of a
ribbon inductor located in a space (see Kuraishi, "Exercise
Microwave Circuit", Tokyo Denki Daigaku Shuppan-Kyoku, pp. 199),
the inductances of the wires 86.sub.-3, 86.sub.-4 and 86.sub.-5 are
approximately equal to 1 nH. .[.It.]. .Iadd.If the high attenuation
and wide pass band are needed, it .Iaddend.is insufficient to
obtain an inductance of 4 nH by means of only the wires. As will be
described later, inductors are formed on the ceramic package 81 and
the filter chip 82. In this manner, the inductors L1, L2 and L3 are
formed.
A description will now be given of a SAW filter according to a
second embodiment of the present invention. FIG. 18 shows a SAW
filter 90 according to the second embodiment of the present
invention. In FIG. 18, parts that are the same as parts shown in
the previously described figures are given the same reference
numbers. The resonator R2 in the series arm 61 has an aperture
length As of 80 .mu.m. A resonator and the inductor L1 connected in
series to each other are provided in the parallel arm 62. The
resonator R1A has an aperture length Ap of 120 .mu.m. The aperture
length Ap is larger than the aperture length As, being 1.5 times
the aperture length As. The numbers Np and Ns of finger pairs of
the resonators R2 and R1A are 100.
The filter 90 shown in FIG. 19 has a band characteristic indicated
by a curve 91 shown in FIG. 19. It can be seen from comparison
between the curve 91 and the characteristic curve 65 of the filter
60 that the filter 90 has an improved side lobe suppression factor
without a change in the pass band width.
FIGS. 20A and 20B show a band characteristic' dependence on the
aperture length in the filter shown in FIG. 18. More particularly,
FIG. 20A shows a curve 92 indicating the dependence on the aperture
length with an inductance L of 4 nH connected to the resonator, and
a curve 93 indicating the dependence on the aperture length without
any inductance. The horizontal axis of FIG. 20A denotes the ratio
Ap/As, and the vertical axis thereof denotes the side lobe
suppression factor (dB). FIG. 20B shows a pass band width s the
ratio Ap/As characteristic. A curve 95 indicates the dependence on
inductance with an inductance L of 4 nH connected to the resonator,
and a curve 96 indicates the dependence on inductance without any
inductance.
The following can be seen from FIGS. 20A and 20B. First, the side
lobe suppression factor increases when making the aperture length
Ap of the resonator R1A in the parallel arm 62 larger than the
aperture length As of the resonator R2 in the series arm 61.
Second, the effect of the aperture length Ap of the resonator R1A
is increased without deterioration of the pass band width by
providing the inductor L1 in the parallel arm 62. It can be seen
from the above that the filter 90 has an improved side lobe
suppression factor while the pass band width is not narrowed, as
compared to the filter 60.
A description will now be given of a third embodiment of the
present invention with reference to FIG. 21, in which parts that
are the same as parts shown in the previously described figures are
given the same reference numbers. A SAW filter 100 shown in FIG. 21
comprises a resonator R1B provided in the parallel arm 62, and the
resonator R2 provided in the series arm 61. The number Ns of finger
pairs of the resonator R2 is 100. The inductor L1 is connected in
series of the resonator R1B. The number Np of finger pairs of the
resonator R1B is 150, and is 1.5 times the number Ns of finger
pairs. The aperture lengths As and Ap of the resonators R2 and R1A
are 80 .mu.m.
The filter 100 shown in FIG. 21 has a band characteristic indicated
by a curve 101 shown in FIG. 22. It can be seen from comparison
between the band characteristic curve 65 of the filter 60 and the
characteristic curve 101 of the filter 100 that the filter 100 has
an improved side lobe suppression factor indicated by arrows 102
without reducing the pass band width. It can also be seen from
comparison between the band characteristic curve 91 of the filter
90 and the characteristic curve 101 that the insertion loss of the
filter 100 is less than that of the filter 90. Hence, the filter
100 has an improved side lobe suppression factor while the pass
band width is reduced, and has an insertion loss smaller than that
of the filter 90.
A description will now be given of a fourth embodiment of the
present invention with reference to FIG. 23, in which parts that
are the same as parts shown in the previously described figures are
given the same reference numbers. A filter 110 according to the
fourth embodiment is intended to increase the difference between
the resonance frequency f.sub.r and the antiresonance frequency
f.sub.a of the resonator in the series arm and to thereby improve
the band characteristic. Two identical resonators R2 are provided
in the series arm 61, and two identical resonators R4 are provided
therein. An inductor Ls having an inductance of 3 nH is connected
in series to the resonators R2, and another inductor Ls having an
inductance of 3 nH is connected in series to the resonators R4. The
resonators R1, R3 and R5 are respectively provided in the parallel
arms 62, 63 and 64. The filter 110 has a band characteristic
indicated by a curve 111 shown in FIG. 24.
A description will now be given of the effects provided by adding
one inductor Ls and two resonators R2 and R4. When one inductor Ls
and two resonators R2 and R4 are omitted from the filter 110, the
remaining circuit configuration consists of five resonators R1, R2,
R3, R4 and R5. The band characteristic of the remaining circuit
configuration is indicated by a curve 68 (see FIG. 14). By adding
one inductor Ls, the pass band width is increased, as indicated by
arrows 112 and the side lobe suppression factor is also increased,
as indicated by arrows 113. Particularly, the pass band width is
large at frequencies higher than the center frequency, and is
increased by approximately 15 MHz. The band characteristic with the
inductor Ls added to the conventional filter 1 is indicated by
curve 114. In this case, a sufficient side lobe suppression factor
is not obtained. Hence, two resonators R2 and R4 are further added
to the conventional filter 1 with the inductor Ls added thereto. As
indicated by arrows 115, the side lobe suppression factor is
improved by approximately 5 dB without reducing the band
characteristic, and a band characteristic curve 111 can be
obtained. It can be seen from comparison between the curves 111 and
68 that the insertion loss is also improved, as indicated by arrows
116. It is possible to use more than two resonators R2 and more
than two resonators R4. Further, as indicated by the two-dot
chained line in FIG. 23, inductors can be provided in the parallel
arms 62-64.
A description will now be given of a fifth embodiment of the
present invention with reference to FIG. 25, in which parts that
are the same as parts shown in the previously described figures are
given the same reference numbers. A SAW filter 120 shown in FIG. 25
comprises five resonators R1-R5, and three inductors L1-L3. The
inductor L1 in the parallel arm 62 has an inductance Lp1 of 4 nH,
and the inductor L2 in the parallel arm 63 has an inductance Lp2 of
5.5 nH. Further, the inductor L3 in the parallel arm 64 has an
inductance Lp3 of 7 nH.
By making the inductors L1, L2 and L3 have different inductance
values, the filter 120 has a band characteristic indicated by a
curve 121 shown in FIG. 26. Let us compare the characteristic curve
121 with the characteristic curve 65 (FIG. 14) of the filter 60
shown in FIG. 13 in which all the inductance values are the same as
each other. It can be seen from the above that the filter 120 has
an improved side lobe characteristic without reducing the pass band
width, as compared to the filter 60. The characteristic curve has
an attenuation pole 123 located around a frequency of 902 MHz,
while the characteristic curve 121 has two attenuation poles 124
and 125 respectively located around 875 MHz and 892 MHz. A
frequency band 126 between the poles 124 and 125 functions as a
blocking range 127.
A description will now be given of a sixth embodiment of the
present invention with reference to FIG. 27, in which parts that
are the same as parts shown in the previously described figures are
given the same reference numbers. A SAW filter 130 shown in FIG. 27
comprises two SAW resonators R2 and R4 provided in the series arm
61, and three SAW resonators R1B, R3B and R5B respectively provided
in the parallel arms 63 and 64.
As shown in FIG. 28, the resonator R1B has an exciting interdigital
electrode 131, and reflectors 132 and 133 respectively disposed on
both sides of the electrode 131. The reflectors 132 and 133 are
positioned so that .beta.=0.4 in which .beta. is obtained from the
following equation: d=(n+.beta.).lamda. where d is the distance
between the center of the electrode 131 and each of the reflectors
132 and 133, n is an arbitrary integer, .beta. is a real number
equal to or less than 1, and .lamda. is the period of the
interdigital electrode 131 corresponding to its resonance
frequency.
The number of finger pairs of each of the reflectors 132 and 133 is
50. The resonators respectively equipped with the reflectors are
indicated by the symbol "*" shown in FIG. 27. The resonators R3B
and R5B respectively provided in the parallel arms 63 and 64
respectively have two reflectors in the same manner as the
resonator R1B.
The filter 130 shown in FIG. 27 has a band characteristic indicated
by a curve 134 shown in FIG. 29. As compared to the characteristic
curve 85 of the filter 80 (FIG. 13), the insertion loss in the
filter 130 is improved, as indicated by an arrow 135. A ripple
r.sub.p arises from the arrangement of the reflectors 132 and
133.
A description will now be given of the reason why the reflectors
132 and 133 are arranged in the above-mentioned manner. The
influence of the ripple r.sub.p observed when .beta. is changed
from 0 to 0.5 is illustrated by a curve 140 shown in FIG. 30. The
smallest ripple width can be obtained at a point 141 at which
.beta. is 0.4.
FIG. 31 shows a SAW filter device 150 functioning as the filter 130
shown in FIG. 27. In FIG. 31, parts that are the same as parts
shown in the previously described figures are given the same
reference numbers as previously. The filter device 150 comprises
reflectors 132, 133, 151, 152, 153 and 154.
Variations of the one-terminal-pair SAW resonators R1B, R3B and R5B
will now be described.
FIG. 32 shows a first variation R1Ba, which comprises interdigital
electrodes 180 and respectively arranged on both sides of the
exciting interdigital electrode 131. Each of the interdigital
electrodes 180 and 181, which functions as a reflector, is an
electrode in which the electric load thereof is of a short-circuit
type.
FIG. 33 shows a second variation R1Bb, which comprises strip array
type electrodes 167 and 166 respectively arranged on both sides of
the electrode 131.
A description will now be given of a seventh embodiment of the
present invention with reference to FIG. 34, in which parts that
are the same as parts shown in the previously described figures are
given the same reference numbers. A SAW filter 170 shown in FIG. 34
comprises two SAW resonators R2 and two resonators R4 respectively
provided in the series arm 61, and three SAW resonators R1B, R3B
and R5B respectively provided in the parallel arms 62, 63 and 64.
Two inductors Ls are provided in the series arm 61, as shown in
FIG. 34.
The filter 170 is obtained by replacing the resonators R1, R3 and
R5 shown in FIG. 23 with the resonators R1B, R3B and R5B shown in
FIG. 28. As has been described previously, the reflectors 132 and
133 shown in FIG. 28 are positioned so that the condition
.beta.=0.4 is satisfied. The filter 170 has a loss of the pass band
smaller than that of the filter 110 shown in FIG. 23, and a
suppressed ripple.
A description will now be given of an eighth embodiment of the
present invention, which is intended to eliminate the ripple
r.sub.p shown in FIG. 29. First of all, a means for effectively
eliminating the ripple r.sub.p arising from the reflectors will be
described.
The inventors simulated the relationship between the frequencies at
which the ripple r.sub.p is observed and the electrode thickness.
In the simulation, the effects resulting from increasing the film
thickness of the electrode are replaced by increasing the ratio
between the acoustic impedance (Z.sub.m) obtained under the
electrode and the acoustic impedance (Z.sub.o) of the free surface.
As described in the aforementioned Ikata document, an increase in
the electrode thickness increases the weight thereof. Hence, it is
possible to consider that an increase in the electrode thickness is
proportional to an increase in a discontinuous quantity of the
acoustic impedance. With the above in mind, the following equation
was prepared:
Q=Z.sub.o/Z.sub.m=V.sub.o/V.sub.m=1+k.sup.2/2+.alpha.(t) (9) where
V.sub.o and V.sub.m respectively denote sound velocities on the
free surface and under the electrode, k.sup.2 is the
electromechanical coupling coefficient, and t is the film thickness
of the electrode. Then .alpha.(t) was changed as a parameter
proportional to the film thickness t.
From the equation (9), the center frequency f.sub.o of the filter
is written as follows: f.sub.o=2f.sub.o'/(1+Q) (10). The equation
(10) is consistent with the well-known experimental result in
which, as the film thickness increases, the center frequency
decreases from the center frequency f.sub.o' obtained when there is
no discontinuity of the acoustic impedance. The results of the
simulation show that, as .alpha.(t) increases, that is, the film
thickness increases, the frequency position at which the ripple
r.sub.p appears shifts toward the high-frequency range of the pass
band, as indicated by an arrow 180 shown in FIG. 35, and finally
drops into the attenuation pole on the high-frequency side of the
pass band. It will be noted that a ripple r.sub.s shown in FIG. 35
is caused by the reflectors of the resonators provided in the
series arm.
FIG. 36 shows an attenuation quantity vs. frequency characteristic
obtained when .alpha.(t)=0.05. A ripple resulting from the
reflectors of the resonators in the parallel arms is located in the
attenuation pole on the high-frequency side of the pass band. That
is, there is no ripple in the pass band. In addition, the graph of
FIG. 36 shows that the insertion loss is very small. In FIG. 36,
the resonance frequencies of the resonators in the parallel and
series arms are calibrated so that they are located at the
frequency position which is 15 MHz higher than the original
frequency position in order to obtain a center frequency of 932
MHz, because the center frequency of the pass band decreases
according to the equation (10).
The inventors fabricated chips and measured the band characteristic
thereof in order to study the relation to the actual film
thickness.
FIGS. 37A, 37B and 37C respectively show band characteristic curves
185, 186 and 187 for film thicknesses of 2000 .ANG., 3000 .ANG. and
4000 .ANG.. In practice, the center frequency is varied by changing
the film thickness. The graphs of FIGS. 37A, 37B and 37C have been
calibrated by changing the period of the interdigital
electrode.
A ripple r.sub.p resulting from the resonators in the parallel arms
is superimposed on the characteristic curve 185 for a film
thickness of 2000 .ANG.. As the film thickness increases, the
ripple r.sub.p shifts to higher frequencies. The experimental
results shown in FIGS. 37A, 37B and 37C are consistent with the
aforementioned results of the simulation.
However, an insertion loss arising from a bulk wave, which cannot
be calculated by simulation, and a resistance loss appear as the
film thickness increases (see Ebata et al., "SURFACE ACOUSTIC WAVE
RESONATOR ON LiTaO.sub.3 SUBSTRATE AND ITS APPLICATION TO
OSCILLATORS FOR USE IN VTR", Journal of the Institute of
Electronics and Communication Engineers of Japan, vol. J66-C, No.1,
pp.23-pp.30, 1988). Further, the correlation between the above
insertion loss and the resistance loss is also a very important
factor.
FIG. 38A shows a curve 190 of the insertion loss resulting from the
bulk wave, and a resistance loss curve 191. A curve 192 shows an
experimental characteristic curve. The insertion loss is
approximately equal to the resistance loss when the film thickness
is 2500 .ANG.. Then, the total loss mainly resulting from the
insertion loss starts to increase when the film thickness is
approximately 3500 .ANG..
A curve 193 shown in FIG. 38B indicates the frequency position
f.sub.rp of the ripple r.sub.p as a function of the identical film
thickness of the exciting electrode 131 and the reflectors 132 and
133 shown in FIG. 28. It is concluded, based on the graphs in FIGS.
38A and 38B of FIG. 38, that the optimum film thickness that
results in no ripple and little insertion loss is between 2600
.ANG. and 4000 .ANG.. The above optimum film thickness can be
normalized by the period .lamda..sub.p (4.4 .mu.m at 932 MHz see
FIG. 28) of the resonators in the parallel arm substantially
determined by the center frequency of the filter. The normalized
optimum film thickness is between 0.06 and 0.09.
The eighth embodiment of the present invention is based on the
results of the above consideration by the inventors.
FIG. 39 shows a first one-terminal-pair SAW resonator 200 used in
the SAW filter according to the eighth embodiment. The resonator
200 comprises an exciting electrode 201, and two reflectors 202 and
203 respectively located on both sides of the electrode 201. The
electrode 201 and the reflectors 202 and 203 are made of aluminum
(Al) or a mixture or alloy of Al and a few percentage of other
metal by weight. The film thickness t.sub.1 of each of the
electrodes and the reflectors 202 and 203 is equal to 0.06-0.09
times the electrode period .lamda..sub.p. A SAW filter, in which
the resonator 200 is applied to each of the resonators R1B, R3B and
R5B shown in FIGS. 27 and FIG. 34, has a band characteristic
indicated by a curve 205 shown in FIG. 40. It can be seen from FIG.
40 that there is no ripple in the pass band. Use of an Al alloy
improves the breakdown power performance, as compared to use of Al,
Cu or Ti can be mixed with Al.
FIG. 41 shows a variation 210 of the SAW resonator 200. The
resonator 210 shown in FIG. 41 comprises an exciting interdigital
electrode 211, and two reflectors 212 and 213 respectively located
on both sides of the electrode 211. The electrode 211 and the
reflectors 212 and 213 are made of Au. The optimum film thickness
of the electrode 211 and the reflectors 212 and 213 is determined,
taking into account the above-mentioned phenomenon caused due to
the influence of an increase in the weights of the electrode 211
and the reflectors 212 and 213. Since the ratio of the density of
Al to that of Au is 2.7/18.9, equal to 0.143, the optimum film
thickness t.sub.2 is determined by multiplying the optimum film
thickness t.sub.1 by 0.143, and is equal to 0.0086-0.013 times the
electrode period .lamda..sub.p. A SAW filter obtained by applying
the resonator 210 to each of the resonators R1B, R3B and R5B has a
band characteristic similar to the characteristic shown in FIG. 40,
and does not have any ripple in the pass band.
A description will now be given of the structure of the inductors
L1, L2 and L3 shown in FIG. 13 according to a ninth embodiment of
the present invention, with reference to FIG. 42, in which parts
that are the same as parts shown in FIG. 16 are given the same
reference numbers. As shown in FIG. 42, zigzag microstrip lines 220
and 221 are formed on the ceramic package 81 and are connected to
the terminals 84.sub.-3 and 84.sub.-5. Ends of the microstrip lines
220 and 221 are connected to the ground. The pattern width of each
of the microstrip lines 220 and 221 is 100 .mu.m, and the distance
between the microstrip lines 220 and 221 and the ground is 0.5 mm.
When the dielectric constant of the ceramic package 81 is equal to
9, the inductance values of the microstrip lines 220 and 221 are
equal to 2 nH.
A description will now be given, with reference to FIG. 43, of a
tenth embodiment of the present invention which is another
structure of the inductors L1, L2 and L3. In FIG. 43, parts that
are the same as parts shown in FIG. 16 are given the same reference
numbers as previously. Two zigzag microstrip lines 230 and 231
respectively connected to the resonators R1 and R2 are formed on
the filter chip 82. Terminals 85.sub.-3 and 85.sub.-5 are connected
to ends of the microstrip lines 230 and 231. Each of the microstrip
lines 230 and 231 is 3000 .ANG. thick, 60 .mu.m wide and 2 mm in
length. When the dielectric constant of the filter chip
(LiTaO.sub.3) 82 is equal to 44, the inductance of the microstrip
lines 230 and 231 are equal to 2.2 nH.
It is possible to form inductors by suitably combining the bonding
wire 86.sub.-3, the microstrip line 220 on the ceramic package 81
and the microstrip line 230 on the filter chip 82.
A description will now be given, with reference to FIG. 44, of a
SAW filter 240 according to an eleventh embodiment of the present
invention. The eleventh embodiment of the present invention is
configured as follows. First, the resonance frequency f.sub.rs of
the resonators in the series arm is made higher than the
antiresonance frequency f.sub.ap of the resonators in the parallel
arms in order to increase the pass band width. Second,
.DELTA.f.ident.f.sub.rs-f.sub.ap is selected so that the pass band
does not have an extremely large loss.
The previously described embodiments of the present invention
require that f.sub.ap=f.sub.rs. However, as long as this condition
is maintained, the pass band cannot be increased. In order to
increase the pass band, the present inventors considered a
condition f.sub.ap<f.sub.rs, as shown in FIGS. 46A and 46B. It
is apparent from FIGS. 46A and 46B that bx<0 within a range
f.sub.ap<f<f.sub.rs and hence this frequency range is changed
to an attenuation band according to the aforementioned theory.
However, in practice, the product bx can be maintained at a very
small value by limiting .DELTA.f=(f.sub.rs-f.sub.ap), and the above
frequency range can practically function as a pass band without any
substantial attenuation.
FIGS. 47A, 47B and 47C show band characteristics of a ladder-type
filter obtained when the .DELTA.f=(f.sub.rs-f.sub.ap) increases
from zero. The filter used in the experiment has a piezo-electric
substrate made of LiTaO.sub.3 having an electromechanical coupling
coefficient of 0.05, and an Al interdigital electrode having a film
thickness of 3000 .ANG.. The structure of the electrode is one of
two basic units connected so as to form a ladder-type structure, as
shown in FIG. 44. Each of the basic units comprises a first
resonator in the parallel arm and a second resonator in the series
arm. In order to form the input and output parts of the filter in
symmetry to each other, a third resonator is provided in another
parallel arm of the final stage. A plurality of basic units are
cascaded so as to form a ladder-type structure in order to increase
the side lobe suppression factor to a practical value.
However, the insertion loss increases as the number of basic units
to be cascaded increases. Hence, it is preferable to determine the
number of basic units to be cascaded, taking into account an actual
filter specification. The filter being considered is intended to
realize a loss equal to or less than 2 dB and a side lobe
suppression factor equal to or higher than 20 dB. The interdigital
electrode of each of the resonators in the parallel and series arms
is designed to have an aperture length of 180 .mu.m and 50 finger
pairs. The ratio P=(Cop/Cos) obtained when Cop and Cos are
electrostatic capacitances of parallel-arm and series-arm,
respectively, is 1 because the electrodes of all the resonators
have identical specifications.
FIG. 47A shows a band characteristic when .DELTA.f=0. FIG. 47B
shows a band characteristic when .DELTA.f=10 MHz. The band
characteristic shown in FIG. 47B is improved so that the pass band
width (in which a loss equal to or less than 2.5 dB is ensured) is
increased to 40 MHz, while the band characteristic shown in FIG.
47A has a pass band width of 22 MHz. It can be seen from FIGS. 47A
and 47B that the pass band width is improved particularly for low
frequencies. Further, the band characteristic shown in FIG. 47B has
an improved side lobe suppression factor. More particularly, the
side lobe suppression factor is improved to 20 dB from 19 dB.
There is a limit regarding improvement due to increase in .DELTA.f.
FIG. 47C shows a band characteristic when .DELTA.f=19 MHz. The pass
band width slightly deteriorates at high frequencies, and this
deterioration is approximately equal to 2.5 dB, which will increase
the ripple in the pass band. In FIG. 47C, a ripple amounting to
approximately 1.0 dB, which is the allowable ripple limit, is
observed. When .DELTA.f is further increased, the insertion loss
and the in-band ripple increase. Hence, an increase of .DELTA.f=19
MHz is the limit.
The product bx obtained when .DELTA.f=19 MHz was examined. In the
experiment, a SAW resonator provided in a parallel arm shown in
FIG. 44 and a SAW resonator provided in a series arm shown therein
were separately fabricated. The admittance of the resonator in the
parallel arm was measured by means of a circuit configuration shown
in FIG. 48A, and the impedance of the resonator in the series arm
was measured by means of a circuit configuration shown in FIG. 48B.
The measurement of admittance and impedance was carried out by
measuring S21 by means of a network analyzer. The measured values
of S21 were inserted into equations shown in FIGS. 48A and 48B, and
the impedance Z.sub.p and the admittance Y.sub.p were
calculated.
A frequency characteristic shown in FIG. 49 was obtained, which
shows the imaginary part of the admittance or impedance, that is,
the value of b or x. The frequency dependence of the product bx is
as shown in FIG. 50. It can be seen from FIG. 50 that the product
bx is negative and is a small value within
f.sub.ap<f<f.sub.rs. The maximum absolute value |bx.sub.max|
of the product bx is given when: ##EQU00005## and was equal to 0.06
for the embodiment being considered. That is, when value
|bx.sub.max| is equal to or smaller than 0.06, and deterioration of
the insertion loss can be reduced and the in-band ripple can be
suppressed to 1 dB or less. If .DELTA.f>19 MHz, the value of
|bx.sub.max| increases, and both the insertion loss and the in-band
ripple will increase to 1 dB or greater. This value is not
practical. As a result, the value of |bx.sub.max| is a an
upper-limit indicator of characteristic deterioration, and
determines the allowable value of .DELTA.f.
The above consideration will be generalized. FIG. 51 is an
equivalent circuit diagram of a ladder-type filter obtained by
approximating the SAW resonators by the double resonance circuits
of LC. The impedance Z.sub.s of the SAW resonator in the series arm
and the admittance Y.sub.p of the SAW resonator in the parallel arm
are expressed as follows:
.function..omega..omega..function..omega..omega..UPSILON.
.times..times..function..omega..omega..omega..omega..times.
.times..times. .times..omega..omega..omega..omega. ##EQU00006##
where .omega..sub.rs, .omega..sub.as, .omega..sub.rp,
.omega..sub.ap are respectively the resonance and antiresonance
frequencies of the series-arm resonator and the resonance and
antiresonance frequencies of the parallel-arm resonator, and .tau.
is the capacitance ratio (inherent in the substrate). The above
resonance and antiresonance frequencies as well as the capacitance
ratio are written as follows:
.omega..times..pi..times..times..times..times..lamda. ##EQU00007##
.omega..times..pi..omega..tau..times..tau. ##EQU00007.2##
.omega..times..pi..times..times..times. ##EQU00007.3##
.omega..times..pi..omega..times..tau. ##EQU00007.4##
.tau..times..times..times..times. ##EQU00007.5##
The product bx is calculated from the equations (11) and (12) as
follows:
bx=-[C.sub.0p(.omega..sub.ap.sup.2-.omega..sup.2)(.omega..sub.rs-.omega..-
sup.2)]/[C.sub.0s(.omega..sub.rp.sup.2-.omega..sup.2)(.omega..sub.as.sup.2-
-.omega..sup.2)] (13) The angular frequency .omega. which makes the
product bx have a pole is obtained from
.delta.(bx)/.delta..omega.=0, and is expressed as follows:
.omega..omega. ##EQU00008## The value obtained by inserting the
above into the equation (13) is the maximum value of the product bx
in the pass band. That is,
bx.sub.max=-[C.sub.0p(1+1/.tau.)]/[C.sub.0s{1+1/(.tau..DELTA..omega./.ome-
ga..sub.rs}.sup.-2] (15) where
.DELTA..omega.=.omega..sub.rs-.omega..sub.ap=2.pi..DELTA.f (16)
FIG. 52 shows a relation between bx.sub.max and .DELTA.f/f.sub.rs
obtained by plotting the equation (15) as a parameter
P=C.sub.0p/C.sub.0s. The hatched area shown in FIG. 52 corresponds
to the condition such that the allowable value of the product bx is
equal to or smaller than 0.06 obtained by the experiment. Hence,
the allowable value .alpha. of .DELTA.f/f.sub.rs dependent on
P=C.sub.0p/C.sub.0s can be determined, and is written as follows,
by inserting |bx.sub.max|=0.06 into the equation (15):
.alpha..function..tau..tau..tau. ##EQU00009##
The capacitance ratio .tau. depends on the substrate material, and
is approximately 15 for 36.degree. Y-cut X-propagation LiTaO.sub.3
according to the experiment. Hence, the equation (17) can be
rewritten as follows: .alpha.=6.67.times.10.sup.-2/(4.22 {square
root over (P)}-1) (18). When P=1, then .alpha.=0.02, and
.DELTA.f=19 MHz for the embodiment shown in FIG. 47 having f.sub.rs
of 948 MHz. That is, the equation (18) stands.
An increase in .DELTA.f is effective for a piezoelectric substrate
material having a small capacitance ratio .tau., that is a
substrate material having a large electromechanical coupling
coefficient. The equation (17) is obtained for such a substrate
material.
The capacitance ratio .tau. is proportional to the reciprocal of
the electromechanical coupling coefficient k.sup.2. The value of
the ratio .tau. for 64.degree. Y-cut X-propagation LiNbO.sub.3
(k.sup.2=0.11) and the value of the ratio .tau. for 41.degree.
Y-cut X-propagation LiNbO.sub.3 are respectively 6.8 and 4.4. The
above values are obtained using the .tau. value of 36.degree. Y-cut
X-propagation LiTaO.sub.3 and k.sup.2=0.05 (see K. Yamanouchi et
al., "Applications for Piezoelectric Leaky Surface Wave", 1990
ULTRASONIC SYMPOSIUM Proceedings, pp.11-pp.18, 1990).
FIG. 53 shows the relation between the capacitance ratio .tau. and
the electromechanical coupling coefficient k.sup.2, which is
obtained using the values of k.sup.2 and .tau. of 36.degree. Y-cut
X-propagation LiTaO.sub.3 and using such a relation that k.sup.2 is
proportional to the reciprocal of .tau..
From the relation shown in FIG. 53, the values of the capacitance
ratios .tau. of 64.degree. and 41.degree. Y-cut X-propagation
LiNbO.sub.3 substrates can be obtained and are equal to 6.8 and
4.4, respectively.
The structure of the embodiment shown in FIGS. 44 and 45 will now
be described. The SAW filter 240 comprises a 36.degree. Y-cut
X-propagation LiTaO.sub.3 substrate 241, and has dimensions of 1.5
mm.times.2 mm.times.0.5 mm. From the input side of the filter 240,
a parallel-arm resonator Rp1, a series-arm resonator Rs1, a
parallel-arm resonator Rp2, a series-arm resonator Rs2, and a
parallel-arm resonator Rp3 are arranged in that order. Each of the
resonators has reflectors (short-circuit type) 242 respectively
provided on both sides of the electrode having an aperture length
of 180 .mu.m and 50 finger pairs. Each of the reflectors 242 has 50
finger pairs.
The parallel-arm resonators are the same as the series-arm
resonators except for the periods of the interdigital electrodes.
The period .lamda..sub.p of the electrode of each parallel-arm
resonator is 4.39 .mu.m (the ratio between the pattern width and
the gap is 1:1 and hence the pattern width is approximately 1.1
.mu.m (=.lamda..sub.p/4), and the period of the electrode of each
series-arm resonator is 4.16 .mu.m (the pattern width is 1.04 .mu.m
(=.lamda..sub.s/4)).
The respective periods are selected using the following equations
so that the resonance frequencies (f.sub.rp, f.sub.rs) of the
respective resonators are equal to the respective predetermined
values (f.sub.rp=893 MHz, f.sub.rs=942 MHz):
.lamda..sub.s=V.sub.m/f.sub.rs .lamda..sub.p=V.sub.m/f.sub.rp where
V.sub.m is the sound velocity of the surface wave propagating in
the 36.degree. Y-cut X-propagation LiTaO.sub.3 crystal for an
electrode thickness of 3000 .ANG., and is experimentally 3920
m/s.
The SAW filter 240 having the above structure has a band-pass
characteristic having a broad pass band and a low loss, as shown in
FIG. 47C, in which .DELTA.f=19 MHz. When only the pattern width
.lamda..sub.p in FIG. 45 is changed to 4.35 .mu.m, then .DELTA.f
becomes 10 MHz, and the characteristic shown in FIG. 47B is
obtained. The electrode is made of an Al-Cu alloy and is 3000 .ANG.
thick, and is arranged so that the surface wave is propagated in
the X direction of the piezoelectric substrate 241.
A description will now be given of piezoelectric substrates other
than 36.degree. Y-cut X-propagation TiTaO.sub.3. The capacitance
ratio .tau. of 64.degree. Y-cut X-propagation LiNbO.sub.3 is 6.8,
and an equation corresponding to the equation (17) is written as
follows: .alpha.=1.47.times.10.sup.-1/(4.37 {square root over
(P)}-1) (19)
The capacitance ratio .tau. of 41.degree. Y-cut X-propagation
LiNbO.sub.3 is 4.4, and an equation corresponding to the equation
(17) is written as follows: .alpha.=2.273.times.10.sup.-1/(4.52
{square root over (P)}-1) (20) As the .tau. value decreases, that
is, the electromechanical coupling coefficient increases, .alpha.
increases, and the characteristic deteriorates little even if
.DELTA.f increases.
A description will now be given of a twelfth embodiment of the
present invention with reference to FIGS. 54 through 57. A SAW
filter 250 according to the twelfth embodiment of the present
invention is a ladder-type SAW filter having a plurality of basic
units (unit sections), each having a SAW resonator in the parallel
arm and a SAW resonator in the series arm, and establishes image
impedance matching between adjacent unit sections in order to
reduce a loss at each connection node. With the above arrangement,
it becomes possible to reduce the insertion loss in the pass
band.
The twelfth embodiment of the present invention was made with the
following consideration. As shown in FIGS. 58A and 58B, a band-pass
characteristic can be obtained by means of at least one
parallel-arm resonator and at least one series-arm resonator. The
ladder-type connection comprising one parallel-arm resonator and
one series-arm resonator is the unit section of the filter.
It is desirable that the resonance frequency of the series-arm
resonator be equal to or higher than the antiresonance frequency of
the parallel-arm resonator. Two unit sections respectively shown in
FIGS. 58A and 58B are available. The series arm of the unit section
shown in FIG. 58A .[.series.]. .Iadd.serves .Iaddend.as the input
terminal, and the series arm of the unit section shown in FIG. 58B
serves as the output terminal. A multi-stage connection comprising
a plurality of unit sections is categorized into one of three types
shown in FIGS. 59A, 59B and 59C. FIG. 59A shows an arrangement in
which either the input or the output is a series arm and the other
is a parallel arm (asymmetrical type). FIG. 59B shows an
arrangement in which both the input and output are parallel arms
(symmetrical type). FIG. 59C shows an arrangement in which both the
input and output are series arms (symmetrical type).
The insertion loss of the multi-stage connection having n unit
sections is n times that of the unit section, and the side lobe
suppression factor thereof is also n times that of the unit
section. Generally, the insertion loss increases, while the side
lobe suppression is improved. Particularly when the insertion loss
is approximately zero, the multi-stage connection is an effective
means. However, the insertion loss will be larger than n times that
of the unit section unless the impedance matching between the
adjacent unit section is good. If the impedance matching is poor,
power is reflected at the interfaces between adjacent unit sections
(each of the interfaces l-l'-n-n'). The reflection of power
increases the insertion loss. When the power reflection occurring
at an interface between adjacent unit sections is denoted by
.GAMMA., the loss is expressed as n10log(.GAMMA.). Hence, it is
important to suppress increase in the insertion loss establishing
an impedance match between adjacent unit sections and suppressing
power reflection at each interface between two adjacent unit
sections.
A descriptions will now be given of a method for matching the
impedance of adjacent unit sections. As shown in FIG. 60, when two
circuits 1 and 2, each having four different terminal constants
(four parameters A, B, C and D of an F matrix), are connected to
each other so that the impedance matching therebetween is
established, the circuits are designed so that image impedances
obtained by viewing the circuits 1 and 2 from an interface b-b' are
equal to each other. An image impedance Z.sub.i1 obtained by
viewing the circuit 1 from the impedance b-b' can be expressed as
follows, using four terminal constants A.sub.1, B.sub.1, C.sub.1
and D.sub.1 of the circuit 1: .times..times. ##EQU00010##
Similarly, an image impedance Z.sub.i2 obtained by viewing the
circuit 2 from the interface b-b' can be expressed as follows:
.times..times. ##EQU00011## The image impedances Z.sub.i1 and
Z.sub.i2 are determined regardless of a load resistance (pure
resistance) R.sub.0.
When the equations (21) and (22) are equal to each other, the
following impedance matching condition can be obtained:
D.sub.1B.sub.1/C.sub.1A.sub.1=A.sub.2B.sub.2/C.sub.2D.sub.2
(23).
FIG. 61A shows a connection having poor impedance matching, and the
condition of the equation (23) is not satisfied. The reflection
factor .GAMMA. obtained by viewing the right circuit from the
interface b-b' is expressed as follows:
.GAMMA.=(Z.sub.sY.sub.p)/(2+Z.sub.sY.sub.p) (24). The values of the
Z.sub.s and Y.sub.p of a practical element are not equal to zero,
and hence the reflection factor .GAMMA. thereof is not zero.
In a connection shown in FIG. 61B, an image impedance Z.sub.i1
obtained by viewing the left circuit from the interface b-b' is
obtained as follows, using the equation (21): .function..times.
##EQU00012## An image impedance Z.sub.i2 obtained by viewing the
right circuit from the interface b-b' can be obtained using the
equation (22). It will be noted that Z.sub.i2=Z.sub.i1. Hence, the
impedance matching is established, and the reflection factor
.GAMMA. at the interface b-b' is zero. The above holds true for a
connection shown in FIG. 61C.
A description will now be given of a method for cascading a
plurality of unit sections in the manner shown in FIG. 61B or 61C.
FIG. 62-(A) shows a circuit comprising n unit sections (n>2), in
which the connection method shown in FIG. 61B and the connection
method shown in FIG. 61C are alternatively employed. It will be
seen from the above description that there is no reflection at each
interface.
The circuit shown in (A) of FIG. 62 can be modified, as shown in
(B) of FIG. 62, in which two resonators respectively in adjacent
parallel nodes are integrated and two adjacent resonators in the
series arm are also integrated. The series-arm resonator closest to
the input of the filter has an impedance value half that of the
resonators located inside the above series-arm resonator.
Similarly, the parallel-arm resonator closest to the output of the
filter has an admittance value half that of the resonators located
inside the above parallel-arm resonator.
FIGS. 63A, 63B and 63C show configurations obtained by applying the
above modifying method shown in FIG. 62 to the configurations shown
in FIGS. 59A, 59B and 59C, respectively. More particularly, FIG.
63A shows an impedance matching method corresponding to the
matching method shown in FIG. 59A, in which either the input or
output of the filter is the series arm and the other is the
parallel arm. In the configuration shown in FIG. 63A, the impedance
of the series-arm resonator located at one end of the filter is
half that of each inner series-arm resonator, and the admittance of
the parallel-arm resonator located at the other end of the filter
is half that of each inner parallel-arm resonator.
FIG. 63B shows an impedance matching method corresponding to the
matching method shown in FIG. 59B. In the configuration shown in
FIG. 63B, each of two parallel-arm resonators located at respective
ends thereof has an admittance value half that of the inner
parallel-arm resonator.
FIG. 63C shows an impedance matching method corresponding to the
matching method shown in FIG. 59C. In the configuration shown in
FIG. 63C, each of the two series-arm resonators located at
respective ends thereof has an impedance value half that of the
inner series-arm resonators.
A further description will now be given of the twelfth embodiment
of the present invention based on the above-mentioned concept. The
SAW filter 250 according to the twelfth embodiment has the
equivalent circuit shown in FIG. 54, and the practical structure
shown in FIG. 55. As shown in FIG. 54, it has three series-arm
resonators Rs1, Rs2 and Rs3, and three parallel-arm resonators Rp1,
Rp2 and Rp3. Each of the six resonators has an identical aperture
length (90 .mu.m), and an identical number of finger pairs (100).
Each of the resonators has two short-circuit-type reflectors
respectively located on two opposite sides of the interdigital
electrode in order to increase Q. Each of the reflectors has
approximately 100 finger pairs. The series-arm resonators Rs1, Rs2
and Rs3 have an identical finger period (.lamda..sub.s) of 4.191
.mu.m. The parallel-arm resonators Rp1, Rp2 and Rp3 have an
identical finger period .lamda..sub.p of 4.38 .mu.m, which is
different from the value of .lamda..sub.s.
FIG. 64 shows a .[.conventional.]. .Iadd.comparative example of the
.Iaddend.SAW filter .[.related to the SAW filter.]. 250 according
to the twelfth embodiment of the present invention .Iadd.as a
related art.Iaddend.. In each of the filters shown in FIGS. 54 and
64, the design specification of each series-arm SAW resonator
indicated by impedance Z.sub.s is such that the aperture length is
90 .mu.m and the number of finger pairs is 100. The design
specification of each parallel-arm SAW resonator indicated by
admittance Y.sub.p is the same as the above design specification.
The piezoelectric substrate crystal is made of 36.degree. Y-cut
X-propagation LiTaO.sub.3. On the crystal substrate, an
interdigital pattern for each SAW resonator formed with an Al alloy
pattern having a thickness of 3000 .ANG. is provided.
Curve 251 of the solid line shown in FIG. 56 indicates the
characteristic of the filter 250. Curve 252 of the broken line
shown in FIG. 56 indicates the characteristic of the filter shown
in FIG. 64. It can be seen from FIG. 56 that the filter 250 has an
insertion loss less than that of the filter shown in FIG. 64, and
particularly the insertion loss at both ends of the pass band in
the filter 250 is greatly improved.
Curve 253 shown in FIG. 57 shows a band characteristic of .[.the
conventional filter shown in FIG. 64, in which.]. .Iadd.such an
embodiment that .Iaddend.the number of finger pairs of only the
parallel-arm resonator indicated by admittance Y.sub.p is reduced
from 100 to 80 to thereby reduce the value of the admittance
Y.sub.p. It can be seen from FIG. 57 that the insertion loss in the
pass band is improved. It may be said that the insertion loss can
be somewhat improved even by reducing the admittance of the
resonator at the end of the filter by a quantity less than 1/2. The
above holds true for impedance.
The embodiment based on the basic structure shown in FIG. 63A has
been described. A variation in which a number of unit sections are
provided at the center of the filter has the same advantages as the
above embodiment.
A description will now be given, with reference to FIG. 65, of a
SAW filter 260 according to a thirteenth embodiment of the present
invention. The SAW filter 260 is based on the basic structure shown
in FIG. 63B, and has the same insertion loss improvement as shown
by the curve 251 in FIG. 56.
FIG. 66 shows a SAW filter 270 according to a fourteenth embodiment
of the present invention. The filter 270 is based on the basic
structure shown in FIG. 63C. The filter 270 has the same insertion
loss improvement as shown by the curve 251 in FIG. 56.
FIGS. 67 and 68 show a SAW filter 280 according to a fifteenth
embodiment of the present invention. The present embodiment is
based on such a consideration that the insertion loss depends on a
resistance component and a conductance component of the
interdigital electrode. With the above in mind, the fifteenth
embodiment is intended to reduce the resistance component of each
series-arm resonator and reduce the conductance component of each
parallel-arm resonator and to thereby reduce the total insertion
loss of a filter in which resonators make a ladder-type
connection.
Referring to FIG. 67, SAW resonators R.sub.s1, R.sub.s2 and
R.sub.s3 are provided in the series arm, and SAW resonators
R.sub.p1, R.sub.p2 and R.sub.p3 are provided in the respective
parallel arms. The resonance frequency f.sub.rs of each of the
resonators in the series arm is different from the resonance
frequency f.sub.rp of each of the resonators in the parallel
arms.
It will now be assumed that the admittance of each parallel-arm
resonator is expressed as follows: Y.sub.p=g+jb (26) where g
denotes a conductance component, and b denotes a susceptance.
Further, it will be assumed that the impedance of each series-arm
resonator is expressed as follows: Z.sub.s=r+jx (27) where r
denotes a resistance component, and x denotes a reactance
component.
Under the above assumptions, the frequency characteristics of g, b,
r and x are as shown in FIG. 71. The susceptance component b
(indicated by the dot chained line) of the admittance Y.sub.p of
the parallel-arm resonator has the largest value at the resonance
frequency f.sub.rp, at which the sign thereof changes from + to -.
Further, the susceptance component b becomes zero at the
antiresonance frequency f.sub.ap, at which the sign thereof changes
from - to +. The conductance component g (one-dot chain line) has
the largest value is at the resonance frequency f.sub.rp, and
rapidly decreases and approaches zero. The value of the conductance
component g assumes only the plus sign.
The reactance component x (indicated by the solid line in FIG. 71)
of the impedance Z.sub.s of the series-arm resonator becomes zero
at the resonance frequency f.sub.rs, and the largest value at the
antiresonance frequency f.sub.as. Further, the sign of the
reactance component x changes from + to -, and approaches zero from
the minus side in a range higher than f.sub.as. The resistance
component r gradually increases from zero to the largest value at
the antiresonance frequency f.sub.as, and then gradually decreases.
The resistance component r assumes only the plus sign.
In order to obtain a filter characteristic, the antiresonance
frequency f.sub.ap of the parallel-arm resonator is equal to or
slightly smaller than the resonance frequency f.sub.rs of the
series-arm resonator.
A graph depicted in the lower portion of FIG. 71 shows the band
characteristic of the filter circuit. The pass band is formed
around f.sub.ap.apprxeq.f.sub.rs, and the other frequency range
serves as an attenuation range. It can also be seen from FIG. 71
that b and x are respectively zero around the center frequency of
the pass band. Hence, the pass band characteristic of the filter is
determined by only r and g, and the following is obtained:
S21=100/(100+r+50rg+2500 g) (28). Since r>0 and g>0, S21
becomes smaller than 1 as both r and g increase, and the insertion
loss written as 20log, S21 also increases. Hence, the insertion
loss decreases as both r and g are closer to zero.
A description will now be given of a consideration concerning which
part of the interdigital electrode is related to the resistance
component r and the conductance component g. The above
consideration takes into account a resistance r.sub.1 inserted in
the equivalent circuit shown in FIG. 5B. The resistance r.sub.1 is
the sum of the electric resistance component of the interdigital
electrode and an acoustic resistance component corresponding to an
energy loss encountered while bulk waves generated from ends of the
fingers are propagated inside the substrate. The resistance
component resulting from emission of bulk waves is little dependent
on the shape of the interdigital electrodes, and is hence
proportional to the electric resistance r.sub.1 of the interdigital
electrode. Particularly, r=r.sub.1 around the center frequency of
x=0.
The conductance component g of the admittance of the parallel-arm
resonator is proportional to the conductance 1/r.sub.1 of the
electric resistance of the interdigital electrode.
The following equation is known: r=l.sub.s.rho..sub.O/(N.sub.sWt)
(29) where .rho..sub.O denotes the resistivity of the fingers of
the interdigital electrodes, W denotes the width of each finger, t
denotes the film thickness of each finger, l.sub.s denotes the
aperture length of the series-arm resonator, and N.sub.s denotes
the number of finger pairs.
The conductance component g is obtained as follows if the same
substrate and the same metallic film as those used in the
series-arm resonator are employed: g=N.sub.pWt/(l.sub.p.rho..sub.O)
(30) where l.sub.p denotes the aperture length of the parallel-arm
resonator, and the N.sub.p denotes the number of finger pairs. It
will be noted that .rho..sub.O, W and t in the parallel-arm
resonator are almost the same as those in the series-arm
resonator.
Hence, an increase in the insertion loss in the equation (28) is
expressed as follows:
r+50rg+2500g=l.sub.s.rho..sub.O/(N.sub.sWt)+50(l.sub.s/l.sub.p)(N.sub.p/N-
.sub.s)+2500N.sub.pWt/(l.sub.p.rho..sub.O). (31)
It can be seen from equation (31) that the insertion loss of the
series-arm resonator becomes smaller as the aperture length l.sub.s
decreases and the number N.sub.s of finger pairs increases, and
that the insertion loss of the parallel-arm resonator becomes
smaller as the aperture length l.sub.p increase and the number
N.sub.p of finger pairs decreases. Particularly, the insertion loss
can be effectively reduced when l.sub.s/l.sub.p<1 and
N.sub.p/N.sub.s<1, that is, when the aperture length of the
series-arm resonator is smaller than that of the parallel-arm
resonator, and the number of finger pairs of the series-arm
resonator is larger than the number of finger pairs of the
parallel-arm resonator.
The reason for the above will now be described. In equation (31),
r=r.sub.s (r.sub.s: electric resistance of the series-arm
resonator), and g=1/r.sub.p (r.sub.p: electric resistance of the
parallel-arm resonator), and therefore the following expression can
be obtained: r+50rg+2500
g=r.sub.s+50(r.sub.s/r.sub.p)+2500(1/r.sub.p). Hence, an increase
in the insertion loss can be suppressed when
(r.sub.s/r.sub.p)<1, that is, r.sub.s<r.sub.p.
If l.sub.s is too short, a loss resulting from diffraction of the
surface wave takes place. If l.sub.p is too long, a decrease in Q
of the parallel-arm resonator due to resistance increase appears,
and the side lobe suppression factor is deteriorated. Hence, there
is a limit on l.sub.s and l.sub.p.
The equation (31) can be modified as follows:
r+50rg+2500g=l.sub.s.rho..sub.0/(N.sub.sWt.sub.s)+50(l.sub.s/l.sub.p)(N.s-
ub.p/N.sub.s)+(t.sub.pt.sub.s)+2500N.sub.pWt.sub.p/(l.sub.p.rho..sub.O)
(32) where t.sub.s denotes the film thickness of the metallic film
forming the interdigital electrode of the series-arm resonator, and
t.sub.p denotes the film thickness of the metallic film forming the
interdigital electrode of the parallel-arm resonator. Hence, the
insertion loss can be reduced when t.sub.p/t.sub.s.
It is possible to use resonators, each having two different
metallic films having different resistivity values (.rho..sub.os,
.rho..sub.op) and to arrange these resonators in the parallel and
series arms, so that .rho..sub.op/.rho..sub.op<1 can be
satisfied. However, this is not practical in terms of mass
productivity.
A further description will be given, with reference to FIGS. 67 and
68, of the fifteenth embodiment based on the above concept. A
piezoelectric substrate 241 is formed of 36.degree. Y-cut
X-propagation LiTaO.sub.3, and an electrode is made of Al and 3000
.ANG. thick.
Conventionally, in each of the parallel and series arms,
l.sub.s=l.sub.p=90 .mu.m, and N.sub.p=N.sub.s=100. In the present
embodiment, l.sub.s=45 .mu.m and N.sub.s=200 in the series arm,
while l.sub.p=180 .mu.m and N.sub.p=50 in the parallel arm. That
is, l.sub.p>l.sub.s, and N.sub.s>N.sub.p. Further,
l.sub.s/l.sub.p=0.25, and N.sub.p/N.sub.s=0.25. The electrostatic
C.sub.O of the interdigital electrode based on the product of the
number of finger pairs and the aperture length is kept
constant.
In FIG. 69, solid line 281 indicates the characteristic of the
present embodiment, and broken line 282 indicates the
characteristic of the conventional filter. The conventional filter
has an insertion loss of 2.5 dB, while the present embodiment has
an insertion loss of 2.0 dB. That is, the insertion loss is
improved by 0.5 dB, in other words, 25%. Further, since an
increased number of finger pairs of the series-arm resonator is
used, the breakdown power performance is improved, and the
applicable maximum power is improved by 20%.
In the present embodiment, a diffraction loss appears when l.sub.s
is equal to or less than 30 .mu.m, and them side lobe starts to
deteriorate when l.sub.p is equal to or larger than 300 .mu.m.
Hence, the l.sub.s and l.sub.p are limited to the above values. It
can be seen from the above that the insertion loss in the pass band
is improved by decreasing the electric resistance of the series-arm
and increasing the electric resistance of the parallel arm
(decreasing the conductance). It is also possible to use a
parallel-arm resonator having a film thickness larger than that of
the series-arm resonator. Even with this structure, it is possible
to reduce the insertion loss in the pass band.
A description will now be given, with reference to FIG. 72, of a
wave filter according to a sixteenth embodiment of the present
invention. The wave filter (branching filter) shown in FIG. 72
comprises two SAW filters F1 and F2 having input terminals
connected to a pair of common signal terminals TO via common nodes
a and b. The SAW filter F1 has a pair of signal terminals T1, and
the SAW filter F2 has a pair of signal terminals T2. A pair of
signal lines l.sub.h and l.sub.c connects the nodes a and b to the
SAW filter F1, and another pair of signal lines l.sub.h and l.sub.c
connects the nodes a and b to the SAW filter
The SAW filter F1 comprises a series-arm SAW resonator Rso, and a
parallel-arm SAW resonator Rp, which resonators are configured as
has been described previously. The resonator Rso is connected to
the common node a, and hence serves as a resonator of the first
stage of the SAW resonator F1. A plurality of pairs, each pair of
series-arm resonator and parallel-arm resonator are cascaded in the
SAW filter F1. The SAW filter F2 is configured in the same manner
as the SAW filter F1.
The SAW filters F1 and F2 respectively have different band center
frequencies. For example, the SAW filter F1 has a band center
frequency f.sub.1 of 887 MHz, and the SAW filter F2 has a band
center frequency f.sub.2 of 932 MHz. In this case, the frequency
f.sub.1 is lower than the frequency f.sub.2.
FIG. 73 is a Smith's chart of the wave filter shown in FIG. 72. In
FIG. 72, P indicates the pass band of the wave filter. A indicates
a low-frequency-side attenuation band, and B indicates a
high-frequency-side attenuation band. It can be seen from FIG. 73
that the characteristic impedance of the circuit shown in FIG. 72
is equal to 50 .OMEGA., while the impedances of the attenuation
bands A and B are greater than 50 .OMEGA.. This means that the wave
filter shown in FIG. 72 has the impedance characteristics of the
respective band-pass filters.
A description will now be given, with reference to FIGS. 74 and 75,
of a wave filter according to a seventeenth embodiment of the
present invention. In FIG. 74, parts that are the same as parts
shown in the previously described figures are given the same
reference symbols.
As has been described previously, the SAW filters F1 and F2 satisfy
the condition f.sub.1<f.sub.2. If the SAW band-pass filters F1
and F2 have characteristics as shown in FIG. 75, the filter F1 is
maintained in a high-impedance state within the pass band frequency
band of the filter F2. In this case, there is no need to provide an
impedance matching circuit M to the filter F1, and the same
characteristic as the characteristic of the filter F2 alone can be
obtained.
However, the filter F2 does not have a high impedance within the
low-frequency attenuation band A thereof, and crosstalk may take
place. Hence, it is necessary to increase the impedance within the
low-frequency attenuation band A of the filter F2.
An impedance matching circuit M for increasing the impedance in the
low-frequency attenuation band A thereof is connected between the
nodes a and b and the filter F2. The impedance matching circuit M
includes an inductor L, which is a high-impedance element for
rotating the phase of signal. The inductor L has an inductance of,
for example, 6 nH. The inductor L can be formed with, for example,
a metallic strip line made of, for example, gold, tungsten, or
copper, and formed on a glass-epoxy or ceramic substrate. The strip
line formed on the glass-epoxy substrate has a width of 0.5 mm and
a length of 11 mm, and the strip line formed on the ceramic
substrate has a width of 0.2 mm and a length of 6 mm.
As shown in FIG. 75, the impedance matching circuit M provided for
the filter F2 rotates the phase in the direction indicated by the
arrow, and the impedance of the filter F2 in the low-frequency
attenuation band A can be increased.
FIG. 76 shows a wave filter according to an eighteenth embodiment
of the present invention. In FIG. 76, parts that are the same as
parts shown in the previously described figures are given the same
reference symbols. The wave filter shown in FIG. 76 can be obtained
by connecting a capacitor C, which corrects the quantity of phase
rotation of the inductor L, in series between the inductor L and
the series-arm resonator Rso. There is a possibility that a
suitable impedance matching may be not obtained by means of only
inductor L. As shown in a Smith's chart shown in FIG. 77, the phase
is rotated in the direction indicated by the arrow shown in FIG. 77
first, and is rotated by means of the inductor L second.
FIG. 78 shows a wave filter according to a nineteenth embodiment of
the present invention. The filter F1 comprises the series-arm SAW
resonator Rso and the parallel-arm SAW resonator Rp, which are
connected so that the series-arm resonator is located at the first
stage of the filter F1. The parallel-arm SAW resonator Rpo of the
filter F is located at the first stage of the filter F. A line S
for use in phase rotation is connected in series to the SAW filter
F2. It is possible to increase the impedance of the filter F1
within the high-frequency attenuation band B thereof even by an
arrangement such that only the filter F1 has the series-arm
resonator of the first stage. In this case, the resonator of the
first stage of the filter F2 is the parallel-arm resonator Rpo
connected in parallel to the pair of common signal terminals T0,
and the low-frequency attenuation band A of the filter F2
(corresponding to the pass band of the filter F1 does not have a
high impedance. Hence, according to the present embodiment, the
phase rotation line S is connected in series to the filter F2.
As shown in FIG. 79, the direction of phase rotation caused by the
line S is opposite to the directions shown in FIGS. 75 and 77.
However, as shown in FIG. 80, suitable matching of the filter F2
can be obtained. In this case, the length of the line S formed on
the glass-epoxy substrate is approximately 25 mm, and the length of
the line S formed on the ceramic substrate is approximately 26
mm.
A variation of the configuration shown in FIG. 78 can be made by
providing the inductor L in the same manner as shown in FIG. 74. It
is also possible to further provide the capacitor C in the same
manner as shown in FIG. 76.
The band center frequencies f.sub.1 and f.sub.2 of the sixteenth
through nineteenth embodiments of the present invention are not
limited to 887 MHz and 932 MHz.
The present invention is not limited to the specifically disclosed
embodiments, and variations and modifications may be made without
departing from the scope of the present invention.
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