U.S. patent number 6,963,532 [Application Number 09/707,590] was granted by the patent office on 2005-11-08 for communication system and method with orthogonal block encoding.
This patent grant is currently assigned to Ericsson Inc.. Invention is credited to Paul W. Dent.
United States Patent |
6,963,532 |
Dent |
November 8, 2005 |
Communication system and method with orthogonal block encoding
Abstract
A communication system and method with orthogonal, block
encoding is provided. Encoded signals are transmitted by repeating
transmissions of symbol blocks with a phase or sign change selected
for each block from a sequence of phase or sign changes. Different
symbols are transmitted using orthogonal sequences. The decoding
uses different orthogonal sequences for separating the received
encoded signals into corresponding separate channels. The
orthogonal encoding is removed from the encoded transmitted signals
and corresponding ones of the repeated symbols are added in
successively received repeated blocks after the orthogonal encoding
is removed. A transmitter uses a digital source encoder to encode
information into symbols, and each symbol is repeated a preselected
number of times to successively produce groups of repeated bits.
Each repeat bit is changed in phase or sing by application of a
sign or phase change determined by a selected assigned orthogonal
code associated with the transmitter. The sign changed bits are
interleaved from a number of such groups to successively generate a
number of blocks, each composed of the different sign or phase
changed bits of the preselected number of repeated groups and
having a collective sign or phase change corresponding to a common
sign change or phase shared by all bits of the block. The
interleaved blocks then modulate a radio signal for
transmission.
Inventors: |
Dent; Paul W. (Pittsboro,
NC) |
Assignee: |
Ericsson Inc. (Research
Triangle Park, NC)
|
Family
ID: |
25409381 |
Appl.
No.: |
09/707,590 |
Filed: |
November 7, 2000 |
Related U.S. Patent Documents
|
|
|
|
|
|
|
Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
|
898392 |
Jul 22, 1997 |
|
|
|
|
Current U.S.
Class: |
370/208; 370/335;
375/E1.002 |
Current CPC
Class: |
H04B
1/707 (20130101); H04L 1/08 (20130101); H04J
13/004 (20130101) |
Current International
Class: |
H04B
1/707 (20060101); H04L 1/08 (20060101); H04J
11/00 (20060101); H04J 011/00 () |
Field of
Search: |
;370/208,342,203,209,320,335 ;375/130,137 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Pham; Chi
Assistant Examiner: Boakye; Alexander O.
Attorney, Agent or Firm: Coats & Bennett, P.L.L.C.
Parent Case Text
This is a continuation of U.S. patent application Ser. No.
08/898,392, filed Jul. 22, 1997.
Claims
What is claimed is:
1. A method for communicating between a plurality of transmitters
and a receiver, said method comprising: orthogonally encoding
information signals at each transmitter, said orthogonally encoding
comprising: coding information signals at said transmitter to
produce corresponding information blocks containing coded
information symbols; repeating each information block a
predetermined number of times according to a predetermined
spectrum-spreading factor; orthogonally encoding said repeated
information blocks by applying a modification factor to each repeat
of said information blocks, the sequence of modification factors
applied to successive repeats of said information blocks forming
one of a set of mutually orthogonal coding sequences; and
transmitting said orthogonally encoded information signals from
said plurality of transmitters.
2. The method according to claim 1 further comprising decoding said
orthogonally encoded information signals by employing different
orthogonal codes respectively for separating the information blocks
corresponding to different information signals into corresponding
separate channels.
3. The method of claim 2 wherein decoding said orthogonally encoded
information signals comprises removing orthogonal encoding from
said encoded information blocks and adding corresponding ones of
said information symbols in repeated information blocks.
4. The method according to claim 3 wherein removing said orthogonal
encoding form said encoded information blocks comprises forming a
summed signal for each symbol within the repeated information
blocks and processing the summed signal with an equalizer.
5. The method of claim 1 wherein applying a modification factor to
each repeat of said information blocks comprises imposing a
sequence of phase changes onto said repeated information
blocks.
6. The method of claim 1 wherein applying a modification factor to
each repeat of said information blocks comprises imposing a sign
change on said repeated information blocks.
7. A receiver to receive a transmitted signal comprising repeated
information blocks that have been orthogonally encoded, said
receiver comprising: a receiver circuit to receive an orthogonally
encoded information signal comprising at least one repeated
information block; and a decoder to decode said orthogonally
encoded repeated information block, said decoder comprising: an
orthogonal code remover for removing the orthogonal encoding from
said repeated information blocks; and an adder for adding
corresponding ones of said symbols in said repeated information
blocks after the orthogonal encoding is removed by said orthogonal
code remover to form a summed signal for each symbol within said
repeated information blocks.
Description
BACKGROUND OF THE INVENTION
The present invention relates generally to a communication system
and method with transmittal signal encoding and particularly to an
orthogonal communication method employing orthogonal encoding.
Deliberate bandwidth expansion, by redundant coding, is presently
employed because of the advantage it confers on the parameters of
performance. However, this advantage can be lost if the
communication channel suffers from delayed echoes, time-dispersion
or multipath effects.
Code Division Multiple Access, or CDMA, is a known technique often
proposed to artificially widen transmission bandwidths. CDMA is an
extension of well known redundant coding techniques such as the
technique of repeat transmissions with majority vote at the
receiver to combine signal repeats. In some applications of CDMA,
also known as Direct Sequence Spread Spectrum, a mixture of simple
repeats, or "dumb spreading," and error correction coding, or
"intelligent spreading," is employed to achieve desired bandwidth
widening ratio.
It is known in the prior art that it is advantageous to use less
intelligent coding and to substitute an element of dumb spreading
in such a way that different signals become orthogonal to one
another and then do not interfere with each other. For example, if
one signal after a suitable amount of intelligent error correction
coding yields a coded bit stream a.sub.1,a.sub.2,a.sub.3,a.sub.4 .
. . and a second signal yields a coded bit stream
b.sub.1,b.sub.2,b.sub.3,b.sub.4.
Then the first signal is transmitted using additional four-times
repeat coding as
a.sub.1,a.sub.1,-a.sub.1,-a.sub.1,a.sub.2,a.sub.2,-a.sub.2,-a.sub.2,a.sub.
3,a.sub.3,-a.sub.3,-a.sub.3,a.sub.4,a.sub.4,-a.sub.4,-a.sub.4 . . .
while the second signal is transmitted with four times repeat
coding as
b.sub.1,-b.sub.1,-b.sub.1,b.sub.1,b.sub.2,-b.sub.2,-b.sub.2,b.sub.2,b.sub.
3,-b.sub.3,-b.sub.3,b.sub.3,b.sub.4,-b.sub.4,-b.sub.4,b.sub.4 . . .
, then a comparison of the sign pattern of the repeat coding
++--++--++--++-- . . . for the first signal and the sign pattern of
the repeat coding +--++--++--++--+ . . . for the second signal,
shows that these differ in sign in exactly half the positions while
agreeing in the other half. Thus, upon combining the repeats with
the proper signs for enhancing one signal, the contribution from
the interfering signal completely cancels, and vice versa. These
signals are known as "mutually orthogonal."
The U.S. digital cellular IS95 system specifies mutual
orthogonality for transmissions from cellular base stations to
mobile phones, using 64-fold repeat coding with one of 64 sign
patterns selected from a set of 64 mutually orthogonal
Walsh-Hadamard codes. The IS95 system uses non-orthogonal
transmission in the direction from mobile phone to cellular base
stations, using instead intelligent error correction coding
comprising convolutional encoding concatenated with orthogonal
Walsh-Hadamard block coding. In the mobile-to-base direction, the
orthogonality between different Walsh-Hadamard codes is used to
discriminate between different 6-bit symbols transmitted from the
same mobile phone, while in the base-to-mobile direction, the
Walsh-Hadamard codes are used to discriminate between symbols
transmitted to different mobile phones.
A disadvantage of the IS95 system of non-orthogonal transmissions
in the mobile-to-base direction is that these signals interfere
with one another if the power of the mobile transmitter is not
strictly controlled as a function of distance from the base station
such that signals from different mobile phones are received at more
or less the same power level. However, the need for strict power
control is alleviated when practicing the invention disclosed in
U.S. Pat. No. 5,151,919 issued to Dent on Sep. 29, 1992, entitled
CDMA Subtractive Demodulation. In U.S. Pat. No. 5,218,619 issued to
Dent on Jun. 8, 1993, entitled CDMA Subtractive Demodulation,
already decoded signals are subtracted more than once to improve
interference subtraction. U.S. Pat. No. 5,353,352 issued to Dent
and Bottomley on Oct. 4, 1994, entitled Multiple Coding for Radio
Communications, describes optimum spread spectrum access codes,
equivalent to the sign patterns discussed above, when orthogonal
signaling is employed within one transmission with
non-orthogonality between different transmissions, such as used in
an IS95 uplink in the mobile-to-base direction. The disclosures of
the above referenced patents are hereby incorporated by reference
here in their entirety.
The reason for the difference between IS95 uplink,
(mobile-to-base), and IS95 downlink, (base-to-mobile), transmission
schemes is that maintaining orthogonality between different
transmissions requires that they be accurately aligned in time,
when the prior art communication schemes are used. If, in the above
example, the first and second signals are aligned with one another
with a one place shift they are shown as follows: ++--++--++--++--
+--++--++--++--+
The two exemplary sign patterns given above are seen now to differ
only at the beginning and the end of symbol blocks, thus severely
compromising orthogonality.
In the downlink, or base-to-mobile, direction all signals originate
at the same base station and thus time alignment can be assured.
When signals in the uplink or mobile-to-base direction originate at
different mobile phones that lie at different distances from the
base station, it is much more difficult to achieve time alignment
of the signals received at the base station.
The European cellular system known as GSM employs dynamic time
alignment of mobile transmissions, wherein individual mobile phones
are commanded by a base station to advance or retard their timing
to bring the signals received into a desired time relationship with
one another. However, the ability to achieve such synchronization
to a high accuracy, for example within fractions of a microsecond,
is limited by multipath signal propagation phenomenon which is a
characteristic of the land-based mobile radio environment.
The multipath signal propagation phenomenon is caused by
reflections of transmitted signals from large objects such as
hillsides and tall buildings, giving rise to delayed echoes. While
it may be possible to synchronize signals transmitted from a mobile
transmitter such that a selected signal ray or echo is time aligned
and thus orthogonal to a ray from another mobile transmitter,
multipath propagation, reflected rays, or echoes, with path delays
different from those of the selected signal rays will not be time
aligned.
The GSM system uses Time Division Multiple Access (TDMA) in which
each mobile signal is allocated a timeslot that does not overlap
with transmissions from other mobiles on the same frequency. A
guard time between slots equal to the longest normally expected
echo delays, plus the use of commanded time advance/retard, reduces
interference between different transmissions caused by multipath
propagation. The interference of an echo with its original signal
has been reduced by using an equalizer that beneficially adds
together energy in different echoes of the same signal. One such
equalizer is described, for example, in U.S. Pat. No. 5,331,666
issued to Dent on Jul. 19, 1994, entitled Adaptive Maximum
Likelihood Demodulator and U.S. Pat. No. 5,335,250 issued to Dent,
et al. on Aug. 2, 1994, entitled Method and Apparatus for
Bidirectional Demodulation of Digitally Modulated Signals, the
disclosures of which are hereby incorporated by reference herein.
The need for a guard time between time slots reduces the bandwidth
capability of the systems while use of an equalizer does not
eliminate all potential multipath propagation problems.
A need therefore still exists for a system and method that
constructs and communicates signals that remain largely orthogonal
to each other even when delayed by different amounts of time due,
for example, to multipath propagation phenomenon.
SUMMARY OF THE INVENTION
The deficiencies of the prior art described above are alleviated
when practicing a communication system and method with orthogonal
encoding in accordance with the present invention. The
communication system and method of the present invention provides
for repetitively transmitting encoded signals with mutually
orthogonally encoded repeated blocks of symbols, the symbols in the
repeated blocks representing coded information. Decoding of the
orthogonally encoded repeated blocks of symbols of the transmitted
encoded signal is provided.
In accordance with one aspect of the invention, a communication
system is described with orthogonal block encoding and comprises a
plurality of transmitters each with means for repetitively
transmitting encoded signals with mutually orthogonally encoded
repeated blocks of symbols respectively representing samples of an
informational source signal produced at the transmitter. A receiver
is provided for receiving the encoded transmitted signals including
means for decoding the orthogonally encoded repeated blocks of
symbols of the transmitted encoded signals received from all the
plurality of transmitters. The decoding is provided by employing
different ones of a plurality of orthogonal codes respectively
associated with different ones of the plurality of transmitters for
separating the received encoded signals into corresponding separate
channels.
In yet another aspect of the invention, the communication system
each of the plurality of transmitters may repeat each bit of
information produced by a digital source encoder a preselected
first number of times to successively produce groups of repeated
bits. A sign change is selectively imposed on the repeated bits of
each of a second number of successive groups of repeated bits in
accordance with an orthogonal code associated with the
transmitter.
Interleaving of the sign changed bits from the second number of
groups is then carried out to successively generate a number of
blocks equal to said first number containing said second number of
symbols each block comprising different coded information bits
sharing a common sign change. A modulated signal is transmitted in
accordance with the generated blocks with sign changes
corresponding to the orthogonal code.
These and other features and advantages of the present invention
will become apparent from the following detailed description, the
accompanying drawings, and the appended claims.
BRIEF DESCRIPTION OF THE DRAWINGS
In the drawings:
FIG. 1 is a simplified functional block diagram of an orthogonal
block encoding communication system of the present invention;
FIG. 2 is an illustration of two of the orthogonally block encoded
signals received at the receiver of the system of FIG. 1 which are
nonsynchronized by an amount to which the orthogonal block encoding
receiver is entirely insensitive;
FIG. 3 is an illustration like that of FIG. 2 but showing a
departure from ideal orthogonality;
FIG. 4 is an illustration like that of FIG. 2 showing the effect of
multipath propagation;
FIG. 5 is a functional block diagram of a transmitter in accordance
with the invention;
FIG. 6 is a functional block diagram of an alternative transmitter
arrangement in accordance with the invention;
FIG. 7 is a functional block diagram of a receiver in accordance
with the invention;
FIG. 8(a) shows a prior art GSM TDMA burst and format of data bits;
and
FIG. 8(b) is an illustration like that of FIG. 8(a) but showing the
delay insensitive orthogonal CDMA transmission of data bits in
accordance with the invention.
DETAILED DESCRIPTION OF THE INVENTION
Referring to FIG. 1, an orthogonal block encoding communication
system 10 of the present invention is seen to include a plurality
of transmitters exemplified by an pair of substantially identical
block encoding transmitters 11 and 12 which broadcast information
carrying signals S11 and S12 in the form of electromagnetic waves.
Preferably these signals S11 and S12 are digital signals, although
the invention contemplates and is usable with analog signals
modulated onto a carrier wave. These signals S11 and S12 are
received by an orthogonal block encoding receiver 14 which decodes
the orthogonally block encoded signals and separates them into
separate output channels. A portion of the orthogonally block
encoded signal S11 from transmitter 11, as shown with a dotted
line, reaches the receiver 14 via an indirect path by reflecting
off of a reflective object 13 on the landscape. Because the length
of the reflective path is greater than the length of the direct
path of the signal S12 the reflected signal S11' arrives at the
receiver 14 at a time later than the arrival of directly received
signal S12. Accordingly, even if signal S12 is synchronized to
arrive at the receiver simultaneously with the arrival of signal
S11, it will not be synchronized with the reflected signal
S11'.
Referring to FIG. 2, an orthogonal block encoding communication
system 10 is shown. The first signal S11, comprises blocks of N
information-bearing samples b.sub.1,b.sub.2,b.sub.3 . . . b.sub.N
which are repeated a number of times with inversion indicated by a
minus sign or without inversion denoted by a plus sign over each
block. Thus, as shown in FIG. 2, S11-1, S11-3, and S11-4, the
first, third and fourth blocks are not inverted, while the second
block S11-2 is inverted. The inversion/non-inversion pattern for
FIG. 2 is therefore represented by the sign pattern +-++.
The second signal S12 comprises a block of signal samples
a.sub.1,a.sub.2,a.sub.3 . . . . a.sub.N which is also repeated with
or without an inversion. In the case of the second signal S12,
there is no inversion for the first, second and third repeats, but
inversion of the fourth repeat, represented by the sign pattern
+++-.
It may be verified that the first and second signals' sign patterns
+-++ and +++- are orthogonal, which means that they agree in as
many places as they disagree.
When the first signal S11, and second signal S12, are both
transmitted at the same time, linear addition of signal samples
occurs in the aether. However, as shown in FIG. 2, the two signals
S11 and S12, or signal blocks S11-1 and S12-1, are not necessarily
time-aligned. In the example of FIG. 2, the samples a.sub.i and
b.sub.i are not aligned and so do not add, while samples a.sub.i
and b.sub.(i+2) are aligned and do add.
The receiver 14 is connected to receive corresponding signal
samples that are repeated in a transmission time T apart. The
receiver 14 preferably converts the signal samples into a suitable
form, such as numerical, which are stored in a receiver sample
memory 15. The receiver 14 processes and combines corresponding
signal samples received a time period T apart by reading them out
of the memory 15 if they are previously received samples. At the
four sample points exemplified in FIG. 2, the sum of the sample
values from signals S11 and S12 are respectively
a.sub.1 +b.sub.3, a.sub.1 -b.sub.3, a.sub.1 +b.sub.3 and -a.sub.1
+b.sub.3.
In combining the samples, the receiver 14 uses addition or
subtraction according to the sign pattern associated with the
signal. In the example of FIG. 2, the sign pattern +-++is used to
receive the first signal S11. Alternatively, the sign pattern +++-
is used to receive the second signal S12.
In receiving the first signal S11 therefore, the receiver 14
forms,
which illustrates that interference from the samples a.sub.1, and
-a.sub.1 of the second signal S12 cancel.
Alternatively, the receiver 14 combines the received samples using
sign pattern +++- to form the second received signal S12,
obtaining,
showing that interference from the samples b.sub.3 and -b.sub.3 of
the first signal S11 cancels.
Thus the two signals S11 and S12 appear orthogonal despite having a
relative time misalignment of two sample intervals. The same
orthogonality will hold for other time misalignments relatively
small compared to the block length of N sample intervals.
Departures from ideal orthogonality when practicing the invention
occur for some repeated bits, the number of bits for which this
occurs being equal to the time misalignment expressed in sample
intervals. Thus, as shown in FIG. 3, when the block duration is
large compared to the time misalignment, departures from
orthogonality affect only a small fraction of the bits. The
receiver 14 combines received samples to decode the samples using
sign pattern +-++. For b.sub.1, the interference from `a` sample
a.sub.3 cancels. However, for decoding b.sub.N, the receiver 14
obtains
The interference from the `a` samples to b.sub.N does not cancel
completely because a.sub.2 ' is a sample from the next set of block
repeats, and is not necessarily equal to a.sub.2. When the number
of repeats is large however, that is greater than four as in the
example, the b.sub.N value will be enhanced by a large multiplier
while the interference from `a` samples will nearly cancel.
Moreover, any underlying error correction coding will tolerate a
few of the `b` values being corrupted by uncancelled interference
from `a` signal values without causing transmission errors in the
underlying information. Thus in practice, with large block sizes, a
large number of repeats, such as the 64 repeats used in IS95, and
the use of further error correction coding, the invention claims
that signal orthogonality is substantially maintained even with
time misalignment between different signals of several sample
intervals.
When a signal, such as the `a` component of the second signal S12,
propagates from a transmitter to a receiver over multiple
propagation paths of different lengths, the signal will be received
multiplied by a complex number C.sub.0 representing phase and
amplitude change over a first path and will be received multiplied
by a complex factor C.sub.1 representing the phase and amplitude of
a delayed path. FIG. 4 illustrates this condition for a relative
path delay of one sample interval. Thus, when decoding sample
a.sub.2, in addition to changes in phase and amplitude to C.sub.0.
a.sub.2 by the first propagation path, it will be further corrupted
by addition of sample a.sub.1 changed in amplitude and phase by the
factor C.sub.1 of the second path. As shown in FIG. 4, the receiver
14 output is then 4(C.sub.0.a.sub.2 +C.sub.1.a.sub.1), which is
just four times the receiver output that would occur without
repeats. The receiver 14 output is thus successively:
. . .
where a.sub.N " means the Nth symbol of the previous block of N
symbols. All outputs except the first depend on two transmitted
samples.
The output sequence may be processed by an equalizer, such as
described in the incorporated references, designed to handle
delayed paths of one or more samples delay. Such an equalizer
processes all samples correctly except for those at the border
between two blocks, such as the first sample given by equation (2)
above. Samples at the edge of blocks are handled appropriately by
such an equalizer. The degree of approximation is better when the
number of combined repeats `M` is larger than four, such that the
first output becomes ##EQU1##
where the error ##EQU2##
tends to zero in relation to C.sub.0.a.sub.1 +C.sub.1.a.sub.N as M
becomes larger. It is possible, however, to effectively model the
dependence of the receiver after combining based on three samples,
a.sub.1,a.sub.N and a.sub.N " in the above example of FIG. 4, and
to construct an equalizer that uses this model in decoding a.sub.1
while using a model dependent on only two transmitted samples
otherwise. Such an equalizer needs to maintain a larger number of
decoding states or "Viterbi states" to resolve the signal
dependence on the additional symbols.
In CDMA systems, the receiver 14 of the invention thus includes
despreading followed by conventional equalization for multipath
propagation. In accordance with the invention, the receiver 14 may
include a Viterbi Maximum Likelihood Sequence estimator form of
equalizer or a Decision Feedback Equalizer (DFE), or,
alternatively, a suitable RAKE receiver which accounts for
multipath propagation in the despreading process. A suitable RAKE
receiver is described in U.S. Pat. No. 5,305,349 issued to Dent on
Apr. 19, 1994 entitledQuantized Coherent RAKE Receiver, which is
hereby incorporated by reference.
A transmitter 16 in accordance with the invention is preferably
constructed as shown in FIG. 5 including a block interleaver 18
which operates on the signal after a final orthogonal
spread-spectrum coding operation is performed by circuit 20. The
circuit 20 includes a bit repeater 22, a direct sequence orthogonal
code generator 24 and a modulo-2 adder 26.
An information source 28 provides information, such as speech or
facsimile signals, to a digital source encoder 30 which converts
the information into digital form. The output of the digital source
encoder 30 is applied to an error correction encoder 32 to render
transmissions more tolerant to noise and interference. The output
bit stream of the encoder 32 (b.sub.1,b.sub.2,b.sub.3 . . . ) is
spread by a bit repeater 22 which samples each bit M times where M
is the desired spreading factor. By then bitwise adding, the
modulo-2 adder 26 bitwise adds to the spread bit stream, a
characteristic orthogonal code allocated for the signal and
generated by the direct sequence orthogonal code generator 24. A
M.times.N block interleaving operation is performed by the block
interleaver 18 on the spread spectrum coded signal on output such
that repeated bits are not transmitted adjacently in time but
rather separated by a block size of N bits. The block interleaver
18 does not add or delete bits, but alters their order of
transmission, for example, by transposing a matrix of N.times.M
bits. Alternatively, the block interleaver 18 is a helical,
diagonal or block-diagonal interleaver rather than a purely block
interleaver. The spread spectrum coded block signal is then applied
to a radio frequency carrier by means of a modulator 33.
The transmitter 16 of FIG. 5 is formed by adding an interleaver 18,
having precise parameters (M,N) adapted to the spread spectrum code
produced by the generator 24, to result in a CDMA transmitter
according to the invention.
FIG. 6 shows an alternative transmitter 35 in accordance with the
invention which includes the information source 28, digital source
encoder 30 and error correction encoder 32.
The embodiments shown in both FIGS. 5 and 6 can include further
interleaving over and beyond the interleaving performed by the
interleaver 18, the purpose of the further interleaving being to
avoid errors in the same sample block appearing consecutively at
the error correction decoder at the receiver 14. Any such
additional within-block interleaving is considered to be part of
the error correction coding process.
The output from the error correction encoder 32 is connected to a
block repeater unit 36 which saves a block of N consecutive bits
and then repeats the block M times. A block sign generator 37
selectively supplies a sign for each repetitive block. Thus, block
sign generator 37 only needs to generate orthogonal codes at the
block rate, not at the rate at which signal samples are generated
or a "chip rate." The sign from block sign generator 37 is combined
with signal samples, such as bit b.sub.3 from the block repetition
unit 36 using an exclusive-OR or modulo-2 adder unit 38.
Alternatively, a module-2 adder is used. A chip-rate scrambling
code is produced from an access code generator 40 to randomize the
output bit stream from the block sign adder 38. The code produced
by the access code generator 40 must be the same for all signals
that are orthogonal, such as signals in the same cell of a cellular
telephone system.
The access code generator 40 can operate in a number of different
embodiments. In a first embodiment, the use of the access code
generator 40 is optional, and may be omitted in some systems.
Signals which are orthogonal to each other are then generally
transmitted in the same cell. If spare orthogonal codes not already
allocated in a cell are available, they can advantageously by
employed in neighboring cells such that a proportion of the
neighboring cell interference is eliminated. CDMA systems of the
prior art are not able to employ such orthogonality between cells
as the transmissions of one cell cannot be synchronized with the
transmissions of neighboring cells. However, when practicing this
invention, lack of precise synchronization is not an impediment to
orthogonality between cells. If, however, the entire set of
mutually orthogonal codes is used in a first cell, then a
neighboring cell uses a second set of codes, orthogonal to each
other but non-orthogonal to the first cells' codes. Such an
additional set of codes preferably has controlled non-orthogonality
with any other set of codes, as may be obtained by using the
technique of the above-referenced U.S. Pat. No. 5,353,352 which may
be embodied in block sign generator 37.
In a second embodiment, the access code generator generates a
chip-rate code of length equal to the block length and repeats it
for the repeated blocks. The code is then changed for the next set
of repeated blocks, and so-on. The property afforded by this second
technique is that multipath signals delayed by a few chips are
despread by the same sign pattern produced jointly by access code
generator 52 and block sign generator 54 in the receiver of FIG. 7.
Thus multipath propagation causes additive intersymbol interference
between the despread symbols output by averager 58, which may be
resolved by the exemplary maximum likelihood equalizer 60. The
access code is preferably the same for all signals in the same cell
while a different access code is employed for signals in different
cells. The access code is preferably chosen according to the
technique disclosed in U.S. Pat. No. 5,353,352 to achieve
controlled non-orthogonality between cells.
In a third embodiment, the access code generator 40 is chosen to
render multipath delayed signals orthogonal to non-delayed signals.
This is achieved by applying like sign changes to any pair of
adjacent chips in half of the block-repeats and unlike sign changes
in the other half of the block-repeats. This has the effect that
delays of +/-one chip relative to a nominal propagation delay
result in multipath signals which are orthogonal to the nominal
propagation path. The multipath signals are then not orthogonal but
rather identically coded to another signal's code. This option is
thus preferably used when only half the available codes are
employed for discriminating between signals in the cell and the
other half of the orthogonal codes are those which appear on +/-one
chip-delayed multipath and thereby discriminate the multipath.
In a fourth embodiment, the access code generator 40 is a random
code generator or none of the above. Then multipath signals are
neither orthogonal to, nor identically coded with undelayed
signals. If it is desired to demodulate multipath signals, then a
RAKE type of equalizer may be employed, in which the receiver
despreads the received signal using different time-shifted outputs
of access code generator 52 and performs different averages for
each using multiple instances of averager 58 to yield multiple
averages each corresponding to signal ray of different propagation
delay. The different rays are then combined in a RAKE equalizer
such as the RAKE receiver using coarsely quantized coefficients as
described in the above-referenced U.S. Pat. No. 5,305,349. This
fourth embodiment is preferably not suggested for application where
degeneration of code orthogonality is affected by relative
propagation delay differences or synchronization errors.
Advantageously, groups of signals that are not orthogonal, such as
signals in different cells of a cellular wireless telephone system,
are provided with different codes.
The receiver 14 of FIGS. 2, 3, and 4 is preferably constructed in
accordance with the invention as shown in FIG. 7. Signals including
desired signals, interfering signals, noise and multipath
distortion signals are received from an antenna 44 and applied to
an input 45 of a downconverter 46. The downconverter 46
downconverts the radio frequency signal to a signal suitable for
processing, preferably a complex baseband signal. Complex baseband
signals may be in Cartesian (X,Y) form having an X, or "I", real
component and a Y, or "Q" imaginary component, or polar form
(R,THETA) or Logpolar form (log(R),THETA) as described in U.S. Pat.
No. 5,048,059, issued to Dent on Sep. 10, 1991, entitled Logpolar
Signal Processing, the disclosure of which is hereby incorporated
by reference. The downconverted samples from output 47 of the
downconverter 46 are then applied to a sign changer 48 which is
connected to an access code generator 52. The downconverted samples
47 are then sign-changed by the adder 48 according to the sign
pattern of an access code provided to the access code generator 52,
to remove the access code applied by a corresponding transmitter
code generator such as access code generator 40 in FIG. 6. When
different codes are applied to I and Q samples at the transmitter
16, FIG. 5, corresponding codes are used for I and Q samples,
respectively, at the receiver 14, in FIG. 7.
The real I and imaginary Q components of the samples from the sign
changer 48 are deinterleaved by a deinterleaver 56 which functions
by blocking together all chips corresponding to repeats of the same
signal sample information bit. The individual repeat signs are made
the same by applying sign changes in the sign changer 50 according
to one of a set of orthogonal sign patterns supplied by a block
sign generator 54. Alternatively, block descrambling is performed
using the access code generator 52. It will be appreciated that two
changes of sign, in the sign changers 48 and 50 respectively, are
equivalent to a single change of sign determined by the product of
the separate signs. Therefore, it does not matter whether the net
sign change is applied before or after deinterleaving as long as
the access code generator 52 or block sign generator 54 or a
combination thereof generates the appropriate sign sequence.
After the repeats are blocked together and the signs of all repeats
are equalized, the repeats are combined together by an averager 58
which preferably averages or adds all repeats in a window of M
bits, where M is the number of repeats. Alternatively, the averager
58 is a low pass filter of bandwidth similar to that of a block
moving averager. The output of averager 58 is then downsampled from
M samples per bit to one sample per bit to yield the bit series
b.sub.1,b.sub.2,b.sub.3 . . . . These samples may contain
Intersymbol Interference (ISI) due to multipath propagation, so
they are next fed to a maximum likelihood equalizer 60. Output
values from the equalizer 60 are preferably in "soft" form in which
1's and 0's are represented by a value indicative of the degree of
"oneness" or "zeroness" rather than hard 1/0 decisions. U.S. Pat.
No. 5,099,499 issued to Hammar describes deriving soft decisions,
the disclosure of which is hereby incorporated by reference. Use of
soft decisions improves the performance of an error correction
decoder 64 which receives equalized signals and produces hard
decisions and "bad frame" indicators to source decoder 66. The
source decoder 66 translates the output bitstream to, for example,
speech signals and uses the bad frame indicators from the error
decoder 64 to mask error events, and to prevent noise bursts from
corrupting perceived speech quality. Further, a deinterleaver 62 is
used between the equalizer 60 and the error correction decoder 64
if a corresponding interleaver is used at the transmitter 16. The
deinterleaving by the deinterleaver 62 is not related to the use of
the deinterleaver 56 to improve orthogonality under conditions of
timing error or multipath.
Commonly assigned U.S. patent application Ser. No. 08/305,727 of
Dent, entitled Simultaneous Demodulation and Decoding Device filed
Sep. 14, 1994, discloses a decodulation technique which performs
all the functions of the equalizer 60, the deinterleaver 62 and the
error correction decoder 64 and may be used in lieu of these
individual units. This disclosure is hereby incorporated by
reference.
Small departures from true orthogonality that remain for some
transmitted symbols when transmitters are not exactly synchronized
are such as described by equation (1). For example, a joint
demodulation method for two signals can proceed as follows:
If the signals b.sub.N and a.sub.2 are described as belonging to a
vector V(i) of current symbols to be demodulated ##EQU3##
where ##EQU4##
and
and V(i-1) is likewise composed of b.sub.N " and a.sub.2 " from
previous blocks of N transmitted symbols, then upon combining
repeats first with the sign pattern for `b` symbols and then for
`a` symbols we obtain sums S.sub.a and S.sub.b as follows:
When all signals are to be demodulated, as in a cellular base
station or satellite ground station, such remaining non
orthogonality can be entirely compensated by joint demodulation,
decision feedback, or alternatively the subtractive demodulation
method of U.S. Pat. No. 5,151,919 which was incorporated by
reference herein above.
Thus, the sum vector S.sub.b, S.sub.a which should be 4Vi is
corrupted by a small amount of the previous vector V(i-1) and the
next vector V(i+1), the amounts being described by "Intervector
Interference" (IVI) coefficients which are matrices M0, M1 and M2
in the following equation:
The center term is descrambled by multiplying equation (4) by the
inverse of the matrix M1 ##EQU6##
to obtain ##EQU7##
which is equal to equation (4) multiplied by M1 .sup.-1.
The effect of the previous vector V(i-1) and the next vector V(i+1)
may be approximately removed by using S'(ī-1) and S'(i+1)
computed using equation (6) and substituting them into equation (6)
to obtain an improved estimate S'(i) of V(i). This process is
iterated to the extent needed to obtain the accuracy desired.
However, more generally, IVI expressed by equation (5) is
unscrambled by use of a matrix transversal equalizer described by
##EQU8##
where L is selectively sized and the equalization matrices H(j) are
chosen to obtain the desired accuracy of equalization.
It is not necessary to over complicate the process of compensating
for residual non-orthogonalities when only a few of the N symbols
per block are affected, particularly when the symbols are further
processed by an error correction decoder. It may suffice to accord
those symbols which are affected by residual non-orthogonality a
soft value indicative of greater symbol uncertainty before applying
them to the error correction decoder.
The present invention is capable of operating with any number of
block repeats and not just the power of two for which
Walsh-Hadamard sign patterns form orthogonal sets. This ability to
generalize the invention relies on the fact that a radio signal is
capable of being changed in phase by any desired amount and not
just by inverting it 180 degrees. A general phase shift of, for
example, 120 degrees can be made and can be represented by
multiplication by the complex factor: ##EQU9##
Presuming that a block of symbols is to be transmitted in three
repeats in accordance with the invention, a first transmitter
transmits its symbol blocks with successive phase shifts of 0, 120,
and 240 degrees applied to the three block repeats. Using the
symbols where S.sub.0, S.sub.1, S.sub.2, represent 0, 120 and 240,
respectively.
S.sub.0 =1, ##EQU10## ##EQU11##
a first transmitter transmits S.sub.0.(b.sub.1,b.sub.2,b.sub.3 . .
. b.sub.N); S.sub.1.(b.sub.1,b.sub.2,b.sub.3 . . . b.sub.N);
S.sub.2.(b.sub.1,b.sub.2,b.sub.3 . . . b.sub.N); where
(b.sub.1,b.sub.2,b.sub.3 . . . b.sub.N) stands for the block of
symbols modulated without a phase shift. A second transmitter
transmits S.sub.0.(a.sub.1,a.sub.2 . . . a.sub.N);
S.sub.2.(a.sub.1,a.sub.2 . . . a.sub.N N); S.sub.1.(a.sub.1,a.sub.2
. . . a.sub.N); where (a.sub.1,a.sub.2 . . . a.sub.N) represents
its modulated symbol block, and a third transmitter transmits
S.sub.0.(C.sub.1,C.sub.2 . . . C.sub.N); S.sub.0.(c.sub.1,c.sub.2 .
. . C.sub.N); S.sub.0.(c.sub.1,c.sub.2 . . . C.sub.N), where
(c.sub.1,c.sub.2 . . . c.sub.N) is the third transmitter's
modulated symbol block.
The three transmissions are orthogonal because the sequences
S.sub.0,S.sub.0,S.sub.0,S.sub.0,S.sub.0,S.sub.0 . . . ;
S.sub.0, S.sub.1,S.sub.2,S.sub.0,S.sub.1,S.sub.2 . . . ; and
S.sub.0, S.sub.2,S.sub.1, S.sub.0,S.sub.2, S.sub.1 . . . ;
are mutually orthogonal even when time-shifted. Such mutually
orthogonal sequences of complex numbers may be called Fourier
sequences and can be of any repeat length L of symbols by forming
them as successive powers of EXP(j2.pi./mL).
The simpler, real-valued Walsh-Hadamard codes are used when the
number of repeats L is a power of two.
In accordance with one aspect of the invention, other orthogonal
sequences may also be constructed, for example by allowing a set of
sequential multipliers for the successive repeats to be neither
complex nor restricted to binary values of +/-1. In particular,
when the multipliers are chosen to be 1 or 0, the orthogonal
sequences
1000000100000001000000 . . .
0100000010000000100000 . . .
0010000000100000001000 . . .
0001000000010000000100 . . .
0000100000001000000010 . . .
0000010000000100000001 . . .
0000001000000010000000 . . .
0000000100000001000000 . . .
arise, which in fact describe an 8-slot TDMA system in which each
signal is transmitted in the slot in which a `1` occurs and not in
which a `0` occurs. Thus, a TDMA system is reproduced as a special
case of the of delay-insensitive orthogonal Code Division Multiple
Access system of the present invention. Likewise when complex
weights are selected from orthogonal Fourier sequences, when symbol
blocks such as (b.sub.1,b.sub.2,b.sub.3 . . . b.sub.N) represent
N-fold repeats of the same symbol `b`, and when each transmitter
output signal is smoothed using a filter, the invention in this
special case provides FDMA signaling in which different
transmissions are mutually orthogonal independent of relative delay
or mistiming by virtue of occupying different, unrelated frequency
channels.
In accordance with another aspect of the invention, TDMA and FDMA
systems can be reproduced as special systems and delay-insensitive
orthogonal CDMA modes may be added to FDMA or TDMA systems by
modification of their coding methods. Referring to FIG. 8(a), a
prior art GSM TDMA signal burst and frame format consists of eight
time slots, each of which contains a signal burst having components
of a syncword surrounded by data bits. In standard GSM, the data
bits in each of the eight timeslots belong to a different
communications link or telephone call. Evolution of GSM to permit
one link to use multiple timeslots provides higher user bit rates,
in which case the data bits in successive slots can be from the
same communications link or call.
Alternatively, FIG. 8(b) shows how, in accordance with the
invention, the same data bits of FIG. 8(a) may be repeated
successively with or without a phase inversion or phase change to
form a delay-insensitive orthogonal CDMA signal. In FIG. 8(b), the
positioning of each repeat preferably straddles two signal bursts,
which advantageously avoids the guard time occurring between time
slots and prevents a block from being split by a syncword. This has
a positive effect on how well orthogonality is preserved under
mistiming conditions, and also avoids the need to apply the
orthogonal phase change sequence to the syncwords S. When a block
straddles two time slots, the block is split by the guard time
where zero energy is transmitted and not by the syncwords. This
results in less reduction of orthogonality under mistiming as the
zero energy symbols of the guard time cause less interference than
the full energy symbols of the syncword, when they overlap data
symbols.
Other arrangements of the repeats within the bursts can of course
be used, and it is not necessary to have eight repeats. For
example, using Fourier sequences, seven repeats could be used with
the eighth timeslot being used for receiving in the mobile
terminal, to avoid a duplexing filter to connect the transmitter
and receiver to the same antenna at the same time.
Those skilled in the art who now have the benefit of the present
disclosure will appreciate that the present invention may take many
forms and embodiments. Some embodiments have been presented so as
to give an understanding of the invention. It is intended that
these embodiments should be illustrative, and not limiting of the
present invention. Rather, it is intended that the invention cover
all modifications, equivalents and alternatives falling within the
spirit and scope of the invention as defined by the appended
claims.
* * * * *